MIC2128
75V, Synchronous Buck Controller Featuring Adaptive
On-Time Control with External Soft Start
Features
General Description
®
• Hyper Speed Control Architecture Enables:
- High Input to Output Voltage Conversion
Ratio Capability
- Any Capacitor™ Stable
- Ultra-Fast Load Transient Response
• Wide 4.5V to 75V Input Voltage Range
• Adjustable Output Voltage from 0.6V to 30V
• 270 kHz to 800 kHz Programmable Switching
Frequency
• Built-in 5V Regulator for Single-Supply Operation
• Auxiliary Bootstrap LDO for Improving System
Efficiency
• Internal Bootstrap Diode
• Adjustable Soft Start Time
• Enable Input and Power Good Output
• Programmable Current Limit
• Hiccup Mode Short-Circuit Protection
• Internal Compensation and Thermal Shutdown
• Supports Safe Start-Up into a Prebiased Output
• AEC-Q100 Qualified (VAO suffix)
The MIC2128 is a constant-frequency synchronous
buck controller featuring a unique adaptive on-time
control architecture with external soft start. The
MIC2128 operates over an input voltage range from
4.5V to 75V.The output voltage is adjustable down to
0.6V with a guaranteed accuracy of ±1%. The device
operates with programmable switching frequency from
270 kHz to 800 kHz.
The MIC2128 features an external soft start pin (SS)
which allows the user to adjust output soft start time to
reduce inrush current from mains during start-up. The
MIC2128 features an auxiliary bootstrap LDO which
improves the system efficiency by supplying the
MIC2128 internal circuit bias power and gate drivers
from output of the converter. A logic level enable (EN)
signal can be used to enable or disable the controller.
The MIC2128 can start-up monotonically into a
prebiased output. The MIC2128 features an open drain
power good signal (PG) which signals when the output
is in regulation and can be used for simple power
supply sequencing purpose.
The MIC2128 offers a full suite of protection features to
ensure protection of the IC during Fault conditions.
These include undervoltage lockout to ensure proper
operation under power-sag conditions, “hiccup” mode
short-circuit protection and thermal shutdown.
The MIC2128 is available in a 16-pin 3 mm x 3 mm
VQFN package, with an operating junction temperature
range from -40°C to +125°C.
Applications
•
•
•
•
•
Networking/Telecom Equipment
Base Station, Servers
Distributed Power Systems
Industrial Power Supplies
Automotive Power Supplies
Typical Application Circuit
VIN
4.5V to 75V
VIN
PVDD
4.7 µF
0.1 µF
Q1
2.2 µFX3
DH
2.2ё
BST
VDD
L1
10 µH
0.1 µF
4.7 µF
MIC2128
ILIM
+ C1
330 µF
1.5 kё
47 µF
0.1 µF
PG
Q2
DL
VIN
VOUT
5V@5A
SW
7.5 kё
EN
4.7 nF
18 kё
SS
10 nF
FB
1 kё
100 kё
EXTVDD
FREQ
VIN
60 kё
2016-2020 Microchip Technology Inc.
AGND
PGND
VOUT
1 µF
Q1,Q2:SiR878ADP
L1: SRP1265A-100M, Bourns
C1: 10SVP330M
DS20005620F-page 1
MIC2128
Package Type
FB
AGND
VDD
VIN
MIC2128
3 x 3 VQFN* (Top View)
16 15 14 13
PG 1
12 SS
ILIM 2
11 FREQ
EP
SW 3
10 EN
BST 4
5
6
7
8
DH
PGND
DL
PVDD
9 EXTVDD
* Includes Exposed Thermal Pad (EP); see Table 3-1.
Functional Block Diagram
EXTVDD
VDD
PVDD
EN
VIN
9
15
8
10
16
LINEAR
REGULATOR
LINEAR
REGULATOR
UVLO
4
BST
5
DH
3
SW
7
DL
2
ILIM
6
PGND
THERMAL
SHUTDOWN
FREQ 11
TON
ESTIMATION
Control
Logic
Negative Current
Limit
COMPENSATION
FB 13
PVDD
gm
1.3 µA
SS 12
VREF
0.6V
CURRENT LIMIT
DETECTION
PG
µA
1
0.9
VREF
FB
14
AGND
DS20005620F-page 2
2016-2020 Microchip Technology Inc.
MIC2128
1.0
ELECTRICAL CHARACTERISTICS
Absolute Maximum Ratings †
VIN, FREQ, ILIM, SW to PGND .................................................................................................................... –0.3V to +76V
VSW to PGND (Transient < 50 ns) ............................................................................................................................... –5V
VDD, PVDD, FB, PG, SS to AGND ................................................................................................................. –0.3V to +6V
EXTVDD to AGND ...................................................................................................................................... –0.3V to +16V
BST to SW .................................................................................................................................................. –0.3V to +6V
BST to AGND ............................................................................................................................................. –0.3V to +82V
EN to AGND ...................................................................................................................................... –0.3V to (VIN +0.3V)
DH, DL to AGND .............................................................................................................................. –0.3V to (VDD +0.3V)
PGND to AGND ........................................................................................................................................... –0.3V to +0.3V
Junction Temperature........................................................................................................................................... +150°C
Storage Temperature (TS)..................................................................................................................... –65°C to +150°C
Lead Temperature (soldering, 10s) ........................................................................................................................ 260°C
ESD Rating(1) ......................................................................................................................................................... 1000V
† Notice: Stresses above those listed under “Maximum Ratings” may cause permanent damage to the device. This is
a stress rating only and functional operation of the device at those or any other conditions above those indicated in the
operational sections of this specification is not intended. Exposure to maximum rating conditions for extended periods
may affect device reliability.
Note 1: Devices are ESD sensitive. Handling precautions recommended. Human body model, 1.5 k in series with
100 pF.
Operating Ratings(1)
Supply Voltage (VIN) ..................................................................................................................................... 4.5V to 75V
SW, FREQ, ILIM, EN........................................................................................................................................... 0V to VIN
EXTVDD ....................................................................................................................................................... 0V to 13.2V
Junction Temperature (TJ)..................................................................................................................... –40°C to +125°C
Package Thermal Resistance (3 mm × 3 mm QFN-16)
Junction to Ambient (JA) .................................................................................................................................. 50.8°C/W
Junction to Case (JC)....................................................................................................................................... 25.3°C/W
Note 1: The device is not ensured to function outside the operating range.
2016-2020 Microchip Technology Inc.
DS20005620F-page 3
MIC2128
ELECTRICAL CHARACTERISTICS
(Note 1)
Electrical Specifications: unless otherwise specified, VIN = 12V, VOUT = 1.2V; VBST - VSW = 5V, TA = +25°C.
Boldface values indicate -40°C TJ +125°C
Parameter
(Note 4)
Symbol
Min.
Typ.
Max.
Units
Test Conditions
VVIN
4.5
—
5.5
V
PVDD and VDD shorted to VIN
(VPVDD = VVIN = VVDD)
5.5
—
75
Power Supply Input
Input Voltage Range (Note 2)
—
Quiescent Supply Current
IQ
—
1.4
1.8
mA
VFB = 1.5V, no switching
Shutdown Supply Current
IVIN(SHDN)
—
0.1
5
µA
EN = Low
—
30
60
µA
EN = Low, VIN = VDD = 5.5V
PVDD,VDD and EXTVDD
PVDD Output Voltage
VPVDD
4.8
5.1
5.4
V
VVIN = 7V to 75V,
IPVDD = 10 mA
VDD UVLO Threshold
VVDD_UVLO_Rise
3.7
4.2
4.5
V
VDD rising
VDD UVLO Hysteresis
VVDD_UVLO_Hys
—
600
—
mV
EXTVDD Bypass Threshold
VEXTVDD_Rise
4.4
4.6
4.85
V
EXTVDD Bypass Hysteresis
VEXTVDD_Hys
—
200
—
mV
—
—
—
250
—
mV
VEXTVDD = 5V, IPVDD = 25mA
VREF
0.597
0.6
0.603
V
EXTVDD Dropout Voltage
VDD falling, Note 5
EXTVDD rising
Reference
Feedback Reference Voltage
FB Bias Current (Note 3)
TJ = 25°C
0.594
0.6
0.606
V
-40°C TJ 125°C
IFB
—
50
500
nA
VFB = 0.6V
VEN_H
1.6
—
—
V
—
VEN_L
—
—
0.6
V
—
VEN_Hys
—
150
—
mV
Note 5
IEN
—
6
30
µA
VEN = 12V
kHz
VFREQ = VVIN, VVIN = 12V
Enable Control
EN Logic Level High
EN Logic Level Low
EN Hysteresis
EN Bias Current
ON Timer
Switching Frequency
Maximum Duty Cycle
Minimum Duty Cycle
fSW
—
800
—
230
270
300
DMAX
—
85
—
VFREQ = 33% of VVIN,
VVIN = 12V
%
VFREQ = VVIN = 12V
DMIN
—
0
—
%
VFB > 0.6V, Note 5
Minimum ON Time
tON(MIN)
—
80
—
ns
—
Minimum OFF Time
tOFF(MIN)
150
230
350
ns
—
ISS
—
1.3
—
µA
—
VOFFSET
–15
0
15
mV
VFB = 0.59V
Soft Start
Soft Start Current Source
Current Limit
Current-Limit Comparator
Offset
Note 1:
2:
3:
4:
5:
Specification for packaged product only.
The application is fully functional at low VDD (supply of the control section) if the external MOSFETs have
low voltage VTH.
Design specification.
Temperature limits apply for automotive AEC-Q100 qualified part.
Not production tested.
DS20005620F-page 4
2016-2020 Microchip Technology Inc.
MIC2128
(Note 1)
ELECTRICAL CHARACTERISTICS
Electrical Specifications: unless otherwise specified, VIN = 12V, VOUT = 1.2V; VBST - VSW = 5V, TA = +25°C.
(Note 4)
Boldface values indicate -40°C T +125°C
J
Parameter
ILIM Source Current
Symbol
Min.
Typ.
Max.
Units
Test Conditions
ICL
85
100
115
µA
TCICL
—
0.3
—
µA/°C
VNCLTH
—
48
—
mV
—
DH On-Resistance, High
State
RDH(PULL-UP)
—
2
3
—
DH On-Resistance, Low
State
RDH(PULL_DOWN)
—
2
4
—
DL On-Resistance, High
State
RDL(PULL-UP)
—
2
4
—
—
0.36
0.8
—
ILIM Source Current Tempco
Negative Current Limit
Comparator Threshold
VFB = 0.59V
Note 5
FET Drivers
DL On-Resistance, Low State RDL(PULL_DOWN)
SW, VIN and BST Leakage
BST Leakage
ILK(BST)
—
—
30
µA
—
VIN Leakage
ILK(VIN)
—
—
50
µA
—
SW Leakage
ILK(SW)
—
—
50
µA
—
Power Good (PG)
PG Threshold Voltage
VPG_Rise
85
—
95
%VOUT
VFB rising
PG Hysteresis
VPG_Hys
—
6
—
%VOUT
VFB falling
PG Delay Time
PG_R_DLY
—
100
—
µs
VFB rising
PG Low Voltage
VOL_PG
—
70
200
mV
VFB < 90% × VNOM,
IPG = 1 mA
Overtemperature Shutdown
TSHDN
—
150
—
°C
Junction temperature rising
Overtemperature Shutdown
Hysteresis
TSHDN_Hys
—
15
—
°C
—
Thermal Protection
Note 1:
2:
3:
4:
5:
Specification for packaged product only.
The application is fully functional at low VDD (supply of the control section) if the external MOSFETs have
low voltage VTH.
Design specification.
Temperature limits apply for automotive AEC-Q100 qualified part.
Not production tested.
2016-2020 Microchip Technology Inc.
DS20005620F-page 5
MIC2128
TEMPERATURE SPECIFICATIONS
Parameters
Sym.
Min.
Typ.
Max.
Units
Conditions
TJ
-40
—
+125
°C
Note 1
TJ(MAX)
—
—
+150
°C
—
TS
-65
—
+150
°C
—
TLEAD
—
—
+260
°C
Soldering, 10s
JA
—
50.8
—
°C/W
—
JC
—
25.3
—
°C/W
—
Temperature Ranges
Operating Junction Temperature
Maximum Junction Temperature
Storage Temperature
Lead Temperature
Package Thermal Resistances
Thermal Resistance, Junction-to-Ambient
16 Lead, 3x3 mm
Junction-to-Case
VQFN
Note 1:
The maximum allowable power dissipation is a function of ambient temperature, the maximum allowable
junction temperature and the thermal resistance from junction-to-air (i.e., TA, TJ, JA). Exceeding the
maximum allowable power dissipation will cause the device operating junction temperature to exceed the
maximum +125°C rating. Sustained junction temperatures above +125°C can impact the device reliability.
DS20005620F-page 6
2016-2020 Microchip Technology Inc.
MIC2128
2.0
TYPICAL CHARACTERISTIC CURVES
Note:
The graphs and tables provided following this note are a statistical summary based on a limited number of
samples and are provided for informational purposes only. The performance characteristics listed herein
are not tested or guaranteed. In some graphs or tables, the data presented may be outside the specified
operating range (e.g., outside specified power supply range) and therefore outside the warranted range.
Note: Unless otherwise indicated, VVIN = 12V, fSW = 300 kHz, RCL = 1.5 k, L = 10 µH, VEXTVDD = VOUT, TA = +25°C
(refer to the Typical Application Circuit circuit).
20
Input Current (ȝA)
Input Supply Current (mA)
25
15
VOUT = 5V
IOUT = 0A
FSW = 300 kHz
VEN = VVIN
10
5
0
6
350
340
330
320
310
300
290
280
270
260
250
VVIN = 48V, with resistor divider
between VIN and AGND at FREQ pin
(100 k and 60 k)
EN = GND
-50
12 18 24 30 36 42 48 54 60 66 72 78
-25
Input Voltage (V)
FIGURE 2-1:
Input Voltage.
FIGURE 2-4:
Temperature.
Input Supply Current vs.
25
50
75
100
Input Shutdown Current vs.
5.4
25
5.3
EXTVDD = GND
PVDD Voltage (V)
Input Supply Current (mA)
30
20
15
VEXTVDD = VOUT
VVIN = 48V
IOUT = 0A
FSW = 300 kHz
VEN = VIN
10
5
0
5.2
5.1
5
IPVDD = 10 mA
VEN = VVIN
EXTVDD = GND
4.9
4.8
-50
-25
0
25
50
75
100
6
12 18 24 30 36 42 48 54 60 66 72 78
Temperature (°C)
FIGURE 2-2:
Temperature.
Input Voltage (V)
FIGURE 2-5:
Input Supply Current vs.
600
5.4
500
5.3
PVDD Voltage (V)
Input Current (ȝA)
0
Temperature (°C)
400
300
200
VVIN = 48V, with resistor divider
between VIN and AGND at FREQ pin
(100 k and 60 k)
EN = GND
100
0
VVIN = 48V
IPVDD = 10 mA
VEN = VVIN
5.2
VEXTVDD
EXTVDD = 12V
5.1
5
EXTVDD = GND
4.9
VEXTVDD = 5V
4.8
6
18
30
42
54
66
78
-50
-25
Input Shutdown Current vs.
2016-2020 Microchip Technology Inc.
0
25
50
75
100
Temperature (°C)
Input Voltage (V)
FIGURE 2-3:
Input Voltage.
PVDD Line Regulation
FIGURE 2-6:
Temperature.
PVDD Voltage vs.
DS20005620F-page 7
MIC2128
Note: Unless otherwise indicated, VVIN = 12V, fSW = 300 kHz, RCL = 1.5 k, L = 10 µH, VEXTVDD = VOUT, TA = +25°C
(refer to the Typical Application Circuit circuit).
5.2
1.6
PVDD Voltage (V)
EXTVDD = GND
Enable Voltage (V)
5
VEXTVDD = 12V
4.8
VEXTVDD = 5V
4.6
4.4
4.2
VVIN = 48V
VEN = VVIN
4
0
10
20
30
40
50
1.4
VEN rising
1.2
1.0
VEN falling
0.8
0.6
60
-50
-25
IPVDD (mA)
FIGURE 2-7:
FIGURE 2-10:
Temperature.
PVDD Load Regulation.
100
125
Enable Threshold vs.
5.4
VVDD rising
4.3
4.1
EN Current (ȝA)
VDD Voltage (V)
75
5.6
4.5
3.9
VVDD falling
3.7
3.5
3.3
IVDD = 0 mA
EXTVDD = GND
5.2
5.0
4.8
4.6
4.4
VVIN = 12V
VEN = 5V
4.2
4.0
3.1
-50
-25
0
25
50
75
100
-50
125
-25
FIGURE 2-8:
Temperature.
VDD UVLO Threshold vs.
Switching frequency (kHz)
VEXTVDD rising
4.6
4.5
VEXTVDD falling
4.4
4.3
4.2
-50
-25
0
25
50
75
100
125
320
310
300
290
280
270
260
250
240
230
220
DS20005620F-page 8
50
75
100
125
EXTVDD Threshold vs.
Enable Bias Current vs.
IOUT = 5A
IOUT = 0A
VOUT = 5V
FSW_SETPOINT = 300 kHz
VEXTVDD = VOUT
VEN = VVIN
6
12 18 24 30 36 42 48 54 60 66 72 78
Input Voltage (V)
Temperature (°C)
FIGURE 2-9:
Temperature.
25
FIGURE 2-11:
Temperature
4.8
4.7
0
Temperature (°C)
Temperature (°C)
EXTVDD Voltage (V)
0
25
50
Temperature (°C)
FIGURE 2-12:
Input Voltage.
Switching Frequency vs.
2016-2020 Microchip Technology Inc.
MIC2128
Note: Unless otherwise indicated, VVIN = 12V, fSW = 300 kHz, RCL = 1.5 k, L = 10 µH, VEXTVDD = VOUT, TA = +25°C
(refer to the Typical Application Circuit circuit).
310
1.38
TA = -40°C
Switching Frequency (kHz)
305
1.36
295
SS Source Current (ȝA)
300
TA = 25°C
290
285
280
VVIN = 48V
VOUT = 5V
FSW_SETPOINT = 300 kHz
VEXTVDD = VOUT
VEN = VVIN
TA = 85°C
275
270
265
0
1
2
3
4
1.34
1.32
1.30
1.28
1.26
1.24
1.22
1.20
5
-50
-25
0
Load Current (A)
Switching Frequency vs.
FIGURE 2-16:
Temperature.
140
606
130
604
Feedback Voltage (mV)
ILIM Source Current (ȝA)
FIGURE 2-13:
Load Current.
120
110
100
90
80
70
50
75
100
125
SS Source Current vs.
602
600
598
596
594
-50
-25
0
25
50
75
100
125
-50
-25
0
Temperature (°C)
FIGURE 2-14:
Temperature.
ILIM Source Current vs.
FIGURE 2-17:
Temperature.
PG Threshold/VREF Ratio
1.2
1.0
0.8
0.6
0.4
0.2
0.0
-50
-25
0
25
50
50
75
100
125
75
100
125
FIGURE 2-15:
Current Limit Comparator
Offset vs. Temperature.
Feedback Voltage vs.
95
94
93
92
91
90
89
88
87
86
85
-50
-25
Temperature (°C)
2016-2020 Microchip Technology Inc.
25
Temperature (°C)
1.4
Current Limit Comparator
Offset Voltgae (mV)
25
Temperature (°C)
0
25
50
75
100
125
Temperature (°C)
FIGURE 2-18:
vs. Temperature.
PG Threshold/VREF Ratio
DS20005620F-page 9
MIC2128
100
90
80
70
60
50
40
30
20
10
0
VOUT=5V
VOUT = 5V
Efficiency (%)
Efficiency (%)
Note: Unless otherwise indicated, VVIN = 12V, fSW = 300 kHz, RCL = 1.5 k, L = 10 µH, VEXTVDD = VOUT, TA = +25°C
(refer to the Typical Application Circuit circuit).
VOUT = 3.3V
VOUT=3.3V
VOUT = 2.5V
VOUT=2.5V
VOUT=1.8V
VOUT = 1.8V
VOUT = 1.5V
VOUT=1.5V
VOUT=1.2V
VOUT = 1.2V
VOUT = 1.0V
VOUT=1.0V
0
0.5
1
1.5
2
2.5
3
3.5
4
4.5
100
90
80
70
60
50
40
30
20
10
0
5
VOUT=5V
VOUT = 5V
VOUT=3.3V
VOUT = 3.3V
VOUT=2.5V
VOUT = 2.5V
VOUT=1.8V
VOUT = 1.8V
VOUT = 1.5V
VOUT=1.5V
VOUT = 1.2V
VOUT=1.2V
VOUT=1V
VOUT = 1.0V
0
0.5
1
Output Current (A)
VOUT = 5V
VOUT=5V
VOUT = 3.3V
VOUT=3.3V
VOUT = 2.5V
VOUT=2.5V
VOUT=1.8V
VOUT = 1.8V
VOUT=1.5V
VOUT = 1.5V
VOUT=1.2V
VOUT = 1.2V
VOUT=1.0V
VOUT = 1.0V
1
1.5
2
2.5
3
3.5
4
4.5
5
VOUT=3.3V
VOUT = 3.3V
VOUT=2.5V
VOUT = 2.5V
VOUT=1.8V
V
= 1.8V
OUT
VOUT=1.5V
VOUT = 1.5V
VOUT=1.2V
VOUT = 1.2V
VOUT=1.0V
VOUT = 1.0V
2
2.5
3
3.5
4
Output Current (A)
FIGURE 2-21:
Efficiency vs. Output
Current (Input Voltage = 36V).
DS20005620F-page 10
4.5
5
0.5
1
1.5
2
2.5
3
3.5
4
4.5
5
FIGURE 2-23:
Efficiency vs. Output
Current (Input Voltage = 60V).
VOUT=5V
VOUT = 5V
1.5
4
VOUT=5V
V
OUT = 5V
VOUT=3.3V
V
OUT = 3.3V
VOUT=2.5V
VOUT = 2.5V
VOUT=1.8V
V
OUT = 1.8V
VOUT=1.5V
V
OUT = 1.5V
VOUT=1.2V
VOUT = 1.2V
VOUT=1.0V
V
OUT = 1.0V
0
Efficiency (%)
Efficiency (%)
100
90
80
70
60
50
40
30
20
10
0
1
3.5
Output Current (A)
FIGURE 2-20:
Efficiency vs. Output
Current (Input Voltage = 24V).
0.5
3
100
90
80
70
60
50
40
30
20
10
0
Output Current (A)
0
2.5
FIGURE 2-22:
Efficiency vs. Output
Current (Input Voltage = 48V).
Efficiency (%)
Efficiency (%)
100
90
80
70
60
50
40
30
20
10
0
0.5
2
Output Current (A)
FIGURE 2-19:
Efficiency vs. Output
Current (Input Voltage=12V).
0
1.5
4.5
5
100
90
80
70
60
50
40
30
20
10
0
VOUT=5V
VOUT = 5V
VOUT=3.3V
VOUT = 3.3V
VOUT = 2.5V
VOUT=2.5V
VOUT=1.8V
VOUT = 1.8V
VOUT=1.5V
VOUT = 1.5V
VOUT=1.2V
VOUT = 1.2V
VOUT=1.0V
VOUT = 1.0V
0
0.5
1
1.5
2
2.5
3
3.5
4
4.5
5
Output Current (A)
FIGURE 2-24:
Efficiency vs. Output
Current (Input Voltage = 75V).
2016-2020 Microchip Technology Inc.
MIC2128
Note: Unless otherwise indicated, VVIN = 12V, fSW = 300 kHz, RCL = 1.5 k, L = 10 µH, VEXTVDD = VOUT, TA = +25°C
(refer to the Typical Application Circuit circuit).
5.1091
Output Voltage (V)
5.0991
VVIN
20V/div
5.0891
VIN = 12V
VIN=12V
5.0791
VIN = 24V
VIN=24V
5.0691
VSW
50V/div
VIN=36V
VIN = 36V
5.0591
VIN=48V
VIN = 48V
5.0491
VIN=60V
VIN = 60V
VOUT
2V/div
VIN = 75V
VIN=75V
5.0391
0
0.5
1
1.5
2
2.5
3
3.5
4
4.5
5
Output Current (A)
FIGURE 2-25:
(VOUT = 5V).
10 ms/div
FIGURE 2-28:
VVIN Turn-On with
Pre-biased Output.
Load Regulation
VEN
2V/div
VVIN
20V/div
VSW
20V/div
VOUT
2V/div
VPG
5V/div
VOUT
2V/div
IL
5A/div
FIGURE 2-26:
IL
5A/div
10 ms/div
VVIN Turn-On.
FIGURE 2-29:
VVIN
20V/div
VSW
20V/div
VOUT
2V/div
10 ms/div
FIGURE 2-27:
VVIN Turn-Off.
2016-2020 Microchip Technology Inc.
IL
5A/div
4 ms/div
EN Turn-On/Turn-Off.
VEN
2V/div
VOUT
2V/div
VPG
5V/div
IL
5A/div
FIGURE 2-30:
2_ms/div
EN Turn-On Delay.
DS20005620F-page 11
MIC2128
Note: Unless otherwise indicated, VVIN = 48V, fSW = 300 kHz, RCL = 1.5 k, L = 10 µH, VEXTVDD = VOUT, TA = +25°C
(refer to the Typical Application Circuit circuit).
VEN
2V/div
VVDD
1V/div
VOUT
2V/div
VPG
5V/div
200 µs/div
FIGURE 2-31:
IL
5A/div
EN Turn-Off Delay.
VOUT
2V/div
VSW
5V/div
FIGURE 2-34:
Rising.
VDD UVLO Threshold-
VVDD
2V/div
VEN
2V/div
VOUT
2V/div
VOUT
2V/div
VSW
5V/div
VSW
50V/div
4 ms/div
FIGURE 2-32:
Output.
20 ms/div
EN Turn-On with Prebiased
VEN
1V/div
100 ms/div
FIGURE 2-35:
Falling.
VDD UVLO Threshold-
VEN
2V/div
VOUT
500 mV/div
VOUT
2V/div
IL
5A/div
VSW
50V/div
FIGURE 2-33:
DS20005620F-page 12
10 ms/div
Enable Thresholds.
4 ms/div
FIGURE 2-36:
Enable into Output Short.
2016-2020 Microchip Technology Inc.
MIC2128
Note: Unless otherwise indicated, VVIN = 48V, fSW = 300 kHz, RCL = 1.5 k, L = 10 µH, VEXTVDD = VOUT, TA = +25°C
(refer to the Typical Application Circuit circuit).
VOUT
2V/div
VVIN
20V/div
VOUT
500 mV/div
IL
5A/div
IL
5A/div
4 ms/div
10 ms/div
FIGURE 2-37:
Power-Up into Output Short.
FIGURE 2-40:
Circuit.
Recovery from Output Short
VOUT
200 mV/div
AC coupled
VOUT
2V/div
IOUT
5A/div
IOUT
5A/div
4ms/div
FIGURE 2-38:
Threshold.
Output Current Limit
FIGURE 2-41:
(IOUT = 0A to 5A).
Load Transient Response
VOUT
200 mV/div
AC coupled
VOUT
2V/div
IOUT
2A/div
IL
5A/div
2 ms/div
FIGURE 2-39:
200 µs/div
Output Short Circuit.
2016-2020 Microchip Technology Inc.
200 µs/div
FIGURE 2-42:
Load Transient Response
(IOUT = 0A to 2.5A).
DS20005620F-page 13
MIC2128
Note: Unless otherwise indicated, VVIN = 48V, fSW = 300 kHz, RCL = 1.5 k, L = 10 µH, VEXTVDD = VOUT, TA = +25°C
(refer to the Typical Application Circuit circuit).
VOUT
50 mV/div
AC coupled
VOUT
200 mV/div
AC coupled
VSW
50V/div
IOUT
2A/div
IL
5A/div
2 µs/div
200 µs/div
FIGURE 2-43:
Load Transient Response
(IOUT = 2.5A to 5A).
FIGURE 2-46:
Load.
Switching Waveform at No
VOUT
50 mV/div
AC coupled
IL
5A/div
VSW
50V/div
VSW
50V/div
VDH
50V/div
IL
5A/div
VDL
5 mV/div
2 µs/div
2 µs/div
FIGURE 2-44:
Load.
Switching Waveform at No
FIGURE 2-47:
Load.
Switching Waveform at Full
IL
5A/div
VSW
50V/div
VDH
50V/div
VDL
5V/div
2 µs/div
FIGURE 2-45:
Load.
DS20005620F-page 14
Switching Waveform at Full
2016-2020 Microchip Technology Inc.
MIC2128
3.0
PIN DESCRIPTION
The descriptions of the pins are listed in Table 3-1.
TABLE 3-1:
PIN FUNCTION TABLE
Pin Number
Pin Name
1
PG
3.1
Pin Function
Open-drain Power Good Output pin
2
ILIM
Current Limit setting resistor connection pin
3
SW
Switch Pin and Current Sense Input for negative current limit
4
BST
Bootstrap Capacitor Connection Pin
5
DH
High-side N-MOSFET gate driver Output
6
PGND
7
DL
8
PVDD
9
EXTVDD
Power Ground
Low-side N-MOSFET gate driver output.
Internal high voltage LDO Output of the MIC2128
Supply to the internal low voltage LDO
10
EN
11
FREQ
Enable Input
12
SS
Soft-Start time setting capacitor connection pin
13
FB
Feedback Input
Analog ground
Switching Frequency Programming Input
14
AGND
15
VDD
Supply for the MIC2128 internal analog circuits
16
VIN
Supply Input for the internal high voltage LDO
17
EP
Exposed Pad
Power Good Output Pin (PG)
Connect PG to VDD through a pull up resistor. PG is low
when the FB voltage is 10% below the 0.6V reference
voltage.
3.2
Current Limit Pin (ILIM)
Connect a resistor from ILIM to SW to set current limit.
Refer to Section 4.3 “Current Limit (ILIM)” for more
details.
3.3
Switch Pin (SW)
SW pin provides return path for the high-side
N-MOSFET gate driver when DH is low and is also
used to sense low-side MOSFET current by monitoring
the SW node voltage for negative current limit function.
Connect SW to the pin where high-side MOSFET
source and the low-side MOSFET drain terminal are
connected together.
3.4
Bootstrap Capacitor Pin (BST)
BST capacitor acts as supply for the high-side
N-MOSFET driver. Connect a minimum of 0.1 µF low
ESR ceramic capacitor between BST and SW. Refer to
Section 4.5 “High-Side MOSFET Gate Drive (DH)”
section for more details.
3.5
High-Side N-MOSFET Gate Driver
Output Pin (DH)
High-side N-MOSFET gate driver Output. Connect DH
to the gate of external high-side N-MOSFET.
3.6
Power Ground Pin (PGND)
PGND provides return path for the internal low-side
N-MOSFET gate driver output and also acts as
reference for current limit comparator. Connect PGND to
the external low-side N-MOSFET source terminal and
to the return terminal of PVDD bypass capacitor.
3.7
Low-Side N-MOSFET Gate Driver
Output Pin (DL)
Low-side N-MOSFET gate driver output. Connect to
the gate terminal of the external low-side N-MOSFET.
2016-2020 Microchip Technology Inc.
DS20005620F-page 15
MIC2128
3.8
Internal High Voltage LDO Output
Pin (PVDD)
Internal high voltage LDO Output of the MIC2128.
PVDD is the supply for the low-side MOSFET driver and
for floating high-side MOSFET driver. Connect a
minimum of 4.7 µF low ESR ceramic capacitor from
PVDD to PGND.
3.9
EXTVDD
Supply to the internal low voltage LDO. Connect
EXTVDD to the output of the Buck converter if it is
between 4.7V to 14V to improve system efficiency.
Bypass EXTVDD with a minimum of 1 µF low ESR
ceramic capacitor. Refer to Section 4.7 “Auxiliary
Bootstrap LDO (EXTVDD)” for more details.
3.10
Enable Input Pin (EN)
EN is a logic input. Connect to logic high to enable the
converter and connect to logic low to disable the
converter.
3.11
Switching Frequency
Programming Input Pin (FREQ)
Switching Frequency Programming Input. Connect to
mid-point of the resistor divider formed between VIN
and AGND to set switching frequency of the converter.
Tie FREQ to VIN to set the switching frequency to
800 kHz. Refer to Section 5.1 “Setting the Switching
Frequency” for more details.
3.12
3.13
Feedback Input Pin (FB)
FB is input to the transconductance amplifier of the
control loop. The control loop regulates the FB voltage
to 0.6V. Connect FB node to mid-point of the resistor
divider between output and AGND.
3.14
Analog Ground Pin (AGND)
AGND is reference to the analog control circuits inside
the MIC2128. Connect AGND to PGND at one point on
PCB.
3.15
Bias Voltage Pin (VDD)
Supply for the MIC2128 internal analog circuits.
Connect VDD to PVDD of the MIC2128 through a low
pass filter. Connect a minimum of 4.7 µF low ESR
ceramic capacitor from VDD to AGND for decoupling.
3.16
Input Voltage Pin (VIN)
Supply Input to the internal high voltage LDO. Connect
to the main power source and bypass to PGND with a
minimum of 0.1 µF low ESR ceramic capacitor.
3.17
Exposed Pad (EP)
Connect to AGND copper plane to improve thermal
performance of the MIC2128.
Soft-Start Time Setting Capacitor
Connection Pin (SS)
Soft-Start time setting capacitor connection pin.
Connect a ceramic capacitor from SS to AGND to set
the output Soft-Start time. Refer to Section 5.3
“Setting the Soft Start Time” section for further
details.
DS20005620F-page 16
2016-2020 Microchip Technology Inc.
MIC2128
4.0
FUNCTIONAL DESCRIPTION
The MIC2128 is an adaptive on-time synchronous buck
controller designed to cover a wide range of input
voltage applications ranging from 4.5V to 75V. An
adaptive on-time control scheme is employed to get
fast transient response and to obtain high voltage
conversion ratios at constant switching frequency.
Overcurrent protection is implemented by sensing
low-side MOSFET's RDS(ON) which eliminates lossy
current sense resistor. The device features external
soft-start, enable input, UVLO, power good output
(PG), secondary bootstrap LDO and thermal shutdown.
4.1
Theory of Operation
The MIC2128 is an adaptive on-time synchronous buck
controller which operates based on ripple at feedback
node. The output voltage is sensed by the MIC2128
feedback pin (FB) and is compared to a 0.6V reference
voltage (VREF) at the low-gain transconductance error
amplifier (gm) as shown in the Functional Block
Diagram. Figure 4-1 shows the MIC2128 control loop
timing during steady-state operation.
The error amplifier behaves as short circuit for the
ripple voltage frequency on the FB pin which causes
the error amplifier output voltage ripple to follow the
feedback voltage ripple. When the transconductance
error amplifier output (VgM) is below the reference
voltage of the comparator, which is same as the error
amplifier reference (VREF), the comparator triggers and
generates an on-time event. The on-time period is
predetermined by the fixed tON estimator circuitry
which is given by the following Equation 4-1:
EQUATION 4-2:
t SW – t OFF MIN
230ns
D MAX = --------------------------------------- = 1 – --------------t SW
tSW
Where:
tSW
= Switching period, equal to 1/fSW
It is not recommended to use the MIC2128 with an OFF
time close to tOFF(MIN) during steady-state operation.
The adaptive on-time control scheme results in a
constant switching frequency over wide range of input
voltage and load current. The actual ON time and
resulting switching frequency varies with the different
rising and falling times of the external MOSFETs. The
minimum controllable ON time (tON(MIN)) results in a
lower switching frequency than the target switching
frequency in high VIN to VOUT ratio applications.
The equation below shows the output-to-input voltage
ratio, below which the MIC2128 lowers the switching
frequency in order to regulate the output-to-set value.
EQUATION 4-3:
V OUT
------------- = t ON MIN f SW
V IN
Where:
VOUT
= Output voltage
VIN
= Input voltage
fSW
= Switching frequency
tON(MIN)
= Minimum controllable ON time (80 ns typ.)
EQUATION 4-1:
VOUT
t ON ESTIMATED = -------------------------V VIN f SW
ȴIL
IL
Where:
ȴVOUT = ESR*ȴIL
VOUT
= Output voltage
VVIN
= Power stage input voltage
fSW
= Switching frequency
At the end of the ON time, the internal high-side driver
turns off the high-side MOSFET and the low-side driver
turns on the low-side MOSFET. The OFF time of the
high-side MOSFET depends on the feedback voltage.
When the feedback voltage decreases, the output of
the gm amplifier (VgM) also decreases. When the output
of the gm amplifier (VgM) is below the reference voltage
of the comparator (which is same as the error amplifier
reference (VREF)) the OFF time ends and ON time is
triggered. If the OFF time determined by the feedback
voltage is less than the minimum OFF time
(tOFF(MIN))of the MIC2128, which is about 230 ns
(typical), the MIC2128 control logic applies the
tOFF(MIN) instead.
The maximum duty cycle can be calculated using the
following Equation 4-2:
2016-2020 Microchip Technology Inc.
VOUT
ȴVFB =ȴVOUT *(VREF/VOUT)
VREF
VFB
ȴVFB
VREF
VgM
MIC212ϴ Triggers ON-Time event if the
error amplifier output (VgM) is below VREF
VDH
Estimated ON-Time
FIGURE 4-1:
Timing.
MIC2128 Control Loop
Figure 4-2 shows operation of the MIC2128 during load
transient. The output voltage drops due to a sudden
increase in load, which results in the error amplifier
output (VgM) falling below VREF. This causes the
comparator to trigger an on-time event. At the end of
the ON time, a minimum OFF time tOFF(MIN) is
DS20005620F-page 17
MIC2128
generated to charge the bootstrap capacitor. The next
ON time is triggered immediately after the tOFF(MIN) if
the error amplifier output voltage (VgM) is still below
VREF due to the low feedback voltage. This operation
results in higher switching frequency during load
transients. The switching frequency returns to the
nominal set frequency once the output stabilizes at new
load current level. The output recovery time is fast and
the output voltage deviation is small in the MIC2128
converter due to the varying duty cycle and switching
frequency.
4.2
Soft Start (SS)
Soft Start reduces the power supply inrush current at
start-up by controlling the output voltage rise time. The
MIC2128 features SS pin which allows the user to set
the soft-start time by connecting a capacitor from the
SS pin to AGND. An internal current source of 1.3 µA
charges this capacitor and generates a linear voltage
which is used as the reference for the internal error
amplifier during Soft Start. Once the voltage on this SS
capacitor is above the internal fixed reference of 0.6V,
the error amplifier uses the fixed 0.6V as reference
instead of the voltage on the external SS capacitor.
Full Load
4.3
Current Limit (ILIM)
The MIC2128 uses the low-side MOSFET RDS(ON) to
sense inductor current. In each switching cycle of the
MIC2128 converter, the inductor current is sensed by
monitoring the voltage across the low-side MOSFET
during the OFF period of the switching cycle during
which low-side MOSFET is ON. An internal current
source of 100 µA generates a voltage across the
external current limit setting resistor RCL as show in the
Figure 4-3.
IL
No Load
VOUT
VREF
VIN
VFB
MIC2128
VREF
DH
VgM
L1
SW
Control
Logic
DL
RCL
PGND
CURRENT LIMIT
DETECTION
VDH
ICL
ILIM
toff(MIN)
FIGURE 4-2:
Response.
MIC2128 Load Transient
Unlike true current-mode control, the MIC2128 uses
the output voltage ripple to trigger an on-time event. In
order to meet the stability requirements, the MIC2128
feedback voltage ripple should be in phase with the
inductor current ripple and large enough to be sensed
by the internal error amplifier. The recommended
feedback voltage ripple is 20 mV~100 mV over the full
input voltage range. If a low-ESR output capacitor is
selected, then the feedback voltage ripple may be too
small to be sensed by the internal error amplifier. Also,
the output voltage ripple and the feedback voltage
ripple are not necessarily in phase with the inductor
current ripple if the ESR of the output capacitor is very
low. For these applications, ripple injection is required
to ensure proper operation. Refer to Section 5.8
“Ripple Injection” for details about the ripple injection
technique.
DS20005620F-page 18
FIGURE 4-3:
Circuit.
MIC2128 Current Limiting
The ILIM pin voltage (VILIM) is the difference of the
voltage across the low-side MOSFET and the voltage
across the resistor (VRCL). The sensed voltage VILIM is
compared with the power ground (PGND) after a
blanking time of 150 ns.
If the absolute value of the voltage drop across the
low-side MOSFET is greater than the absolute value of
the voltage across the current setting resistor (VRCL),
the MIC2128 triggers the current limit event.
Consecutive eight current limit events trigger the
hiccup mode. Once the controller enters into hiccup
mode, it initiates a soft-start sequence after a hiccup
timeout of 4 ms (typical). Both the high-side and
low-side MOSFETs are turned off during hiccup
2016-2020 Microchip Technology Inc.
MIC2128
timeout. The hiccup sequence including the soft start
reduces the stress on the switching FETs and protects
the load and supply from severe short conditions.
The current limit can be programmed by using the
following Equation 4-4.
EQUATION 4-4:
RCL
IL PP
I
+ ---------------- R DS ON + V OFFSET
CLIM
2
= -------------------------------------------------------------------------------------------------I CL
Where:
ILIM
= Load current limit
RDS (ON) = On-resistance of low-side power MOSFET
ILPP
= Inductor peak-to-peak ripple current
VOFFSET = Current-limit comparator offset (15 mV max.)
ICL
= Current-limit source current (100 µA typ)
Since MOSFET RDS(ON) varies from 30% to 40% with
temperature, it is recommended to consider the
RDS(ON) variation while calculating RCL in the above
equation to avoid false current limiting due to increased
MOSFET junction temperature rise. It is also
recommended to connect SW pin directly to the drain
of the low-side MOSFET to accurately sense the
MOSFETs RDS(ON).
To improve the current limit variation, the MIC2128
adjusts the internal current limit source current (ICL) at
a rate of 0.3 µA/°C when the MIC2128 junction
temperature changes to compensate the RDS(ON)
variation of external low-side MOSFET. The
effectiveness of this method depends on the thermal
gradient between the MIC2128 and the external
low-side MOSFET. The lower the thermal gradient, the
better the current limit variation.
A small capacitor (CCL) can be connected from the ILIM
pin to PGND to filter the switch node ringing during the
Off-time allowing a better current sensing. The time
constant of RCL and CCL should be less than the
minimum off time.
4.4
Negative Current Limit
The MIC2128 implements negative current limit by
sensing the SW voltage when the low-side FET is ON.
If the SW node voltage exceeds 48 mV typical, the
device turns off the low-side FET for 500 ns. Negative
current limit value is shown in Equation 4-5.
EQUATION 4-5:
48mV
I NLIM = -------------------R DS ON
4.5
High-Side MOSFET Gate Drive (DH)
The MIC2128's high-side drive circuit is designed to
switch an N-Channel external MOSFET. The MIC2128
Functional Block Diagram shows a bootstrap diode
between PVDD and BST pins. This circuit supplies
energy to the high-side drive circuit. A low ESR ceramic
capacitor should be connected between BST and SW
pins (refer Typical Application Circuit).The capacitor
between BST and SW pins, CBST, is charged while the
low-side MOSFET is on. When the high-side MOSFET
driver is turned on, energy from CBST is used to turn the
MOSFET on. A minimum of 0.1 µF low ESR ceramic
capacitor is recommended between BST and SW pins.
The required value of CBST can be calculated using the
following Equation 4-6.
EQUATION 4-6:
Q G_HS
C BST = ------------------ V CBST
Where:
QG_HS
= High-side MOSFET total gate charge
VCBST
= Voltage drop across the CBST,
generally 50 mV to 100 mV
A small resistor in series with CBST, can be used to slow
down the turn-on time of the high-side N-channel
MOSFET.
4.6
Low-Side MOSFET Gate Drive (DL)
The MIC2128's low-side drive circuit is designed to
switch an N-Channel external MOSFET. The internal
low-side MOSFET driver is powered by PVDD. Connect
a minimum of 4.7 µF low-ESR ceramic capacitor to
supply the transient gate current of the external
MOSFET.
4.7
Auxiliary Bootstrap LDO
(EXTVDD)
The MIC2128 features an auxiliary bootstrap LDO
which improves the system efficiency by supplying the
MIC2128 internal circuit bias power and gate drivers
from the converter output voltage. This LDO is enabled
when the voltage on the EXTVDD pin is above 4.6V
(typical) and at the same time the main LDO which
operates from VIN is disabled to reduce power
consumption. Connect EXTVDD to the output of the
buck converter if it is between 4.7V and 14V. When the
EXTVDD is tied to VOUT, a voltage spike will occur at
the PVDD and VDD during a fast hard short at VOUT.
Larger decoupling ceramic capacitors of 10 µF at PVDD
and VDD are recommended for such a situation.
Where:
INLIM
= Negative current limit
RDS (ON) = On-resistance of low-side power MOSFET
2016-2020 Microchip Technology Inc.
DS20005620F-page 19
MIC2128
5.0
APPLICATIONS INFORMATION
5.2
5.1
Setting the Switching Frequency
The output voltage can be adjusted using a resistor
divider from output to AGND whose mid-point is
connected to FB pin as shown the Figure 5-3.
The MIC2128 is an adjustable-frequency, synchronous
buck controller featuring a unique adaptive on-time
control architecture. The switching frequency can be
adjusted between 270 kHz and 800 kHz by changing
the resistor divider network between VIN and AGND pins
consisting of R1 and R2, as shown in Figure 5-1.
Output Voltage Setting
MIC2128
MIC2128
VOUT
R1
VIN
16
COMPENSATION
VIN
4.5V to 75V
13
gm
FB
R1
11
VREF
0.6V
R2
14
AGND
FIGURE 5-3:
FIGURE 5-1:
Adjustment.
Equation 5-1
frequency:
Switching Frequency
shows
the
estimated
switching
EQUATION 5-1:
The output voltage
Equation 5-2:
can
be
calculated
using
EQUATION 5-2:
R1
V OUT = V REF 1 + ------
R 2
VREF
R2
= fO ------------------R1 + R2
fO is the switching frequency when R1 is 100 k and R2
being open; fO is typically 800 kHz. For more precise
setting, it is recommended to use Figure 5-2:
800
Switching Frequency (kHz)
Output Voltage Adjustment.
Where:
f SW_ADJ
VIN = 24V
700
R2
SOFTSTART
Comparator
FREQ
600
VIN = 48V
500
VIN = 75V
400
= 0.6V
The maximum output voltage that can be programmed
using the MIC2128 is limited to 30V, if not limited by the
maximum duty cycle (see Equation 4-2).
A typical value of R1 is less than 30 k. If R1 is too
large, it may allow noise to be introduced into the
voltage feedback loop and also increases the offset
between the set output voltage and actual output
voltage because of the error amplifier bias current. If R1
is too small in value, it will decrease the efficiency of the
power supply, especially at light loads. Once R1 is
selected, R2 can be calculated using Equation 5-3.
EQUATION 5-3:
VOUT = 5V
R1 = 100 k
IOUT = 5A
300
200
50
500
5000
R2 (k)
FIGURE 5-2:
DS20005620F-page 20
R1
R 2 = ----------------------V OUT
------------- – 1
V REF
Switching Frequency vs. R2.
2016-2020 Microchip Technology Inc.
MIC2128
5.3
Setting the Soft Start Time
5.4.1
The output Soft Start time can be set by connecting a
capacitor from SS to AGND from 2 ms to 100 ms as
shown in Figure 5-4.
MIC2128
HIGH-SIDE MOSFET POWER
LOSSES
The total power loss in the high-side MOSFET
(PHSFET) is the sum of the power losses because of
conduction (PCONDUCTION), switching (PSW), reverse
recovery charge of low-side MOSFET body diode
(PQrr) and MOSFET's output capacitance discharge as
calculated in the Equation 5-5.
EQUATION 5-5:
PHSFET = PCONDUCTION HS + PSW HS + P Qrr + P COSS
2
P CONDUCTION HS = I RMS HS R DS ON_HS
13
COMPENSATION
FB
P SW HS = 0.5 VIN ILOAD tR + t F f SW
ISS
1.3 µA
gm
12
PQrr = V IN Q rr f SW
SS
CSS
Comparator
1
2
P COSS = --- C OSS HS + COSS HS V IN f SW
2
Where:
VREF
0.6V
RDS(ON_HS)
=
VIN
=
Operating input voltage
ILOAD
=
Load current
fSW
=
Operating switching frequency
Qrr
=
The value of the capacitor can be calculated using
Equation 5-4.
Reverse recovery charge of low-side
MOSFET body diode or of external
diode across low-side MOSFET
COSS(HS)
=
Effective high-side MOSFET output
capacitance
EQUATION 5-4:
COSS(LS)
=
Effective low-side
capacitance
IRMS(HS)
=
RMS current of the high-side MOSFET
which can be calculated using
Equation 5-6.
tR, tF
=
The high-side MOSFET turn-on and
turn-off transition times which can be
approximated by Equation 5-8 and
Equation 5-9
FIGURE 5-4:
Setting the Soft Start Time.
ISS tSS
C SS = -------------------V REF
Where:
CSS
= Capacitor from SS pin to AGND
ISS
= Internal Soft Start current (1.3 µA typical)
tSS
= Output Soft Start time
VREF
= 0.6V
On-resistance of the high-side MOSFET
MOSFET
output
EQUATION 5-6:
5.4
MOSFET Selection
Important parameters for MOSFET selection are:
• Voltage rating
• On-resistance
• Total gate charge
The voltage rating for the high-side and low-side
MOSFETs are essentially equal to the power stage
input voltage VIN. A safety factor of 30% should be
added to the VIN(MAX) while selecting the voltage rating
of the MOSFETs to account for voltage spikes due to
circuit parasitic elements.
2016-2020 Microchip Technology Inc.
I RMS HS = I LOAD D
ILOAD is the load current and D is the operating duty
cycle, given by Equation 5-7.
EQUATION 5-7:
VOUT
D = ------------V IN
DS20005620F-page 21
MIC2128
EQUATION 5-12:
EQUATION 5-8:
Q SW HS R DH PULL_UP + RHS GATE
t R = -----------------------------------------------------------------------------------------------------V DD – V TH
ILOAD is the load current and D is the operating duty
cycle.
EQUATION 5-9:
Q SW HS R DH PULL_DOWN + RHS GATE
t F = ------------------------------------------------------------------------------------------------------------V TH
Where:
RDH(PULL-UP)
I RMS LS = I LOAD 1 – D
= High-side gate driver pull-up
resistance
RDH(PULL-DOWN) = High-side gate driver pull-down
resistance
RHS(GATE)
= High-side MOSFET gate resistance
VTH
= Gate threshold voltage of the
high-side MOSFET
QSW(HS)
= Switching gate charge of the
high-side MOSFET which can be
approximated by Equation 5-10.
5.5
Inductor Selection
Inductance value, saturation and RMS currents are
required to select the output inductor. The input and
output voltages and the inductance value determine
the peak-to-peak inductor ripple current.
The lower the inductance value, the higher the
peak-to-peak ripple current through the inductor, which
increases the core losses in the inductor. Higher
inductor ripple current also requires more output
capacitance to smooth out the ripple current. The
greater the inductance value, the lower the
peak-to-peak ripple current, which results in a larger
and more expensive inductor.
A good compromise between size, loss and cost is to
set the inductor ripple current to be equal to 30% of the
maximum output current.
EQUATION 5-10:
QGS HS
Q SW HS = -------------------- + QGD HS
2
The inductance value is calculated by Equation 5-13:
EQUATION 5-13:
Where:
QGS(HS)
= High-side MOSFET gate to source
charge
QGD(HS)
= High-side MOSFET gate to drain charge
VOUT VIN – VOUT
L = -----------------------------------------------------V IN f SW 0.3 IFL
Where:
5.4.2
LOW-SIDE MOSFET POWER
LOSSES
The total power loss in the low-side MOSFET (PLSFET)
is the sum of the power losses because of conduction
(PCONDUCTION(LS)) and body diode conduction during
the dead time (PDT) as calculated in Equation 5-11.
EQUATION 5-11:
PLSFET = PCONDUCTION LS + P DT
VIN
= Input voltage
fSW
= Switching frequency
IFL
= Full load current
VOUT
= Output voltage
For a selected Inductor, the peak-to-peak inductor
ripple current ripple can be calculated using
Equation 5-14.
EQUATION 5-14:
2
P CONDUCTION LS = I RMS LS RDS ON_LS
P DT = 2 V F I LOAD t DT f SW
Where:
RDS(ON_LS) = On-resistance of the low-side MOSFET
VF
= Low-side MOSFET body diode forward
voltage drop
tDT
= Dead time which is approximately 20 ns
fSW
= Switching Frequency
IRMS(LS)
= RMS current of the low-side MOSFET
which can be calculated using
Equation 5-12
DS20005620F-page 22
V
V – V
V IN f SW L
OUT
IN
OUT
I L_PP = -----------------------------------------------------
The peak inductor current is equal to the load current
plus one half of the peak-to-peak inductor current ripple
which is shown in Equation 5-15.
EQUATION 5-15:
I L_PP
IL_PK = I LOAD + ---------------2
2016-2020 Microchip Technology Inc.
MIC2128
The RMS and saturation current ratings of the selected
inductor should be at least equal to the RMS current
and saturation current calculated in the Equation 5-16
and Equation 5-17.
2
2 I L_PP
I LOAD(MAX) + -----------------------12
Where:
ILOAD(MAX)
VOUT_PP
ESR ------------------------- I L_PP
Where:
EQUATION 5-16:
I L_RMS =
EQUATION 5-19:
VOUT_PP
= Peak-to-peak output voltage ripple
IL_PP
= Peak-to-peak inductor current ripple
The required output capacitance to meet steady state
output ripple can be calculated using Equation 5-20.
= Maximum load current
EQUATION 5-20:
I L_PP
C OUT = -------------------------------------------------8 f SW V OUT_PP
EQUATION 5-17:
Where:
RCL I CL + 15mV
IL_SAT = --------------------------------------------R DS(ON)
Where:
RCL
= Current limit resistor
COUT
= Output capacitance value
ICL
= Current-Limit Source Current (96 µA typical)
fSW
= Switching frequency
RDS (ON)
= On-resistance of low-side power MOSFET
Maximizing the efficiency requires the proper selection
of core material and minimizing the winding resistance.
Use of ferrite materials is recommended in the higher
switching frequency applications. Lower cost iron
powder cores may be used but the increase in core
loss reduces the efficiency of the power supply. This is
especially noticeable at low output power. The winding
resistance decreases efficiency at the higher output
current levels. The winding resistance must be
minimized although this usually comes at the expense
of a larger inductor. The power dissipated in the
inductor is equal to the sum of the core and copper
losses. At higher output loads, the core losses are
usually insignificant and can be ignored. At lower
output currents, the core losses can be a significant
contributor. Core loss information is usually available
from the magnetic’s vendor.
The amount of copper loss in the inductor is calculated
by Equation 5-18.
EQUATION 5-18:
2
P INDUCTOR CU = I L_RMS R DCR
5.6
Output Capacitor Selection
The main parameters for selecting the output capacitor
are capacitance value, voltage rating and RMS current
rating. The type of the output capacitor is usually
determined by its equivalent series resistance (ESR).
Recommended capacitor types are ceramic, tantalum,
low-ESR aluminum electrolytic, OS-CON and
POSCAP. The output capacitor ESR also affects the
control loop from a stability point of view. The maximum
value of ESR can be calculated using Equation 5-19.
2016-2020 Microchip Technology Inc.
As described in Section 4.1 “Theory of Operation”,
the MIC2128 requires at least 20 mV peak-to-peak
ripple at the FB pin to ensure that the gm amplifier and
the comparator behave properly. Also, the output
voltage ripple should be in phase with the inductor
current. Therefore, the output voltage ripple caused by
the output capacitor’s value should be much smaller
than the ripple caused by the output capacitor ESR. If
low-ESR capacitors, such as ceramic capacitors, are
selected as the output capacitors, a ripple injection
circuit should be used to provide the enough
feedback-voltage ripple. Refer to the Section 5.8
“Ripple Injection” for details.
The voltage rating of the capacitor should be twice the
output voltage for a tantalum and 20% greater for aluminum electrolytic, ceramic or OS-CON. The output
capacitor RMS current is calculated in Equation 5-21.
EQUATION 5-21:
I L_PP
I C_OUT(RMS) = ---------------12
The power dissipated in the output capacitor is shown
in Equation 5-22.
EQUATION 5-22:
2
P DIS(C_OUT) = IC_OUT(RMS) ESRC_OUT
DS20005620F-page 23
MIC2128
5.7
Input Capacitor Selection
The input capacitor reduces peak current drawn from
the power supply and reduces noise and voltage ripple
on the input. The input voltage ripple depends on the
input capacitance and ESR. The input capacitance and
ESR values can be calculated using Equation 5-23.
EQUATION 5-23:
The applications are divided into three situations
according to the amount of the feedback voltage ripple:
1.
Enough ripple at the feedback due to the large
ESR of the output capacitor (Figure 5-5). The
converter is stable without any additional ripple
injection at the FB node. The feedback voltage
ripple is given by Equation 5-26.
EQUATION 5-26:
I LOAD D 1 – D
C IN = ------------------------------------------------ fSW V IN_C
V IN_ESR
ESRC_IN = ----------------------I L_PK
R
R2 + R 1
2
VFB PP = ----------------- ESR I L_PP
IL_PP is the peak-to-peak value of the inductor current
ripple.
Where:
ILOAD
= Load Current
IL_PK
= Peak Inductor Current
VINC
= Input ripple due to capacitance
VINESR
= Input ripple due to input capacitor ESR
η
= Power conversion efficiency
The input capacitor should be rated for ripple current
rating and voltage rating. The RMS value of input
capacitor current is determined at the maximum output
current. The RMS current rating of the input capacitor
should be greater than or equal to the input capacitor
RMS current calculated using the Equation 5-24.
SW
L
R1
MIC2128
COUT
FB
ESR
R2
FIGURE 5-5:
2.
EQUATION 5-24:
Enough Ripple at FB.
Inadequate ripple at the feedback voltage due to
the small ESR of the output capacitor.
The power dissipated in the input capacitor is
calculated using Equation 5-25.
The output voltage ripple can be fed into the FB pin
through a feed forward capacitor, CFF in this case, as
shown in Figure 5-6. The typical CFF value is between
1 nF and 100 nF. With the feed forward capacitor, the
feedback voltage ripple is very close to the output
voltage ripple which is shown in Equation 5-27.
EQUATION 5-25:
EQUATION 5-27:
I C_IN(RMS) = I LOAD(MAX) D 1 – D
V FB PP = ESR I L_PP
2
PDISS(C_IN) = I C_IN(RMS) ESR C_IN
5.8
Ripple Injection
The minimum recommended ripple at the FB pin for
proper operation of the MIC2128 error amplifier and
comparator is 20 mV. However, the output voltage
ripple is generally designed as 1% to 2% of the output
voltage. For low output voltages, such as a 1V, the
output voltage ripple is only 10 mV to 20 mV, and the
feedback voltage ripple is less than 20 mV. If the
feedback voltage ripple is so small that the gm amplifier
and comparator cannot sense it, then the MIC2128
loses control and the output voltage is not regulated. In
order to have sufficient VFB ripple, ripple injection
method should be applied for low output voltage ripple
applications.
DS20005620F-page 24
SW
L
R1
MIC2128
FB
CFF
COUT
ESR
R2
FIGURE 5-6:
3.
Inadequate Ripple at FB.
Virtually no ripple at the FB pin voltage due to
the very-low ESR of the output capacitors:
2016-2020 Microchip Technology Inc.
MIC2128
In this case, additional ripple can be injected into the
FB pin from the switching node SW via a resistor RINJ
and a capacitor CINJ, as shown in Figure 5-7.
L
SW
RINJ
MIC2128
R1
CFF
CINJ
FB
COUT
ESR
For high duty cycle applications with D > 40%, the
procedures to design the ripple injection circuit
components are as below:
1. For given feedback divider resistor values,
select CFF such that the time constant formed by
CFF and feedback divider is 50% of the switching period as given in Equation 5-31.
EQUATION 5-31:
CFF×RFBEQ = 0.5×TSW
R2
Where RFBEQ is the parallel combination of R1 and
FIGURE 5-7:
Invisible Ripple at FB.
R2, RFBEQ = (R1×R2)/(R1+R2)
The injected ripple at the FB pin in this case is given by
the Equation 5-28:
2.
EQUATION 5-28:
3.
V
1 – D
CFF RINJ f SW
OUT
VFB PP = -----------------------------------------
In the above Equation 5-28, it is assumed that the time
constant associated with the CFF meets the criteria
shown in Equation 5-29.
EQUATION 5-29:
T SW
C FF R 1 R 2 RINJ
The process of sizing the ripple injection resistor and
capacitors is:
Select CINJ in the range of 47 nF to 100 nF,
which can be considered as short for a wide
range of the frequencies.
Select CFF in the range of 0.47 nF to 10 nF, if R1
and R2 are in k range.
Select RINJ according to Equation 5-30.
1.
2.
3.
EQUATION 5-30:
Where:
V OUT 1 – D
R INJ = ------------------------------------------------------CFF fSW V FB PP
VOUT
= Output voltage
D
= Duty cycle
fSW
= Switching frequency
5.9
Calculate RINJ using the Equation 5-30. Make
sure that the injected ripple voltage into FB pin
is in the range of 20mV to 100mV.
Choose CINJ = 100nF or at least 10 times the
CFF value."
Power Dissipation in MIC2128
The MIC2128 features two Low Dropout Regulators
(LDOs) to supply power at the PVDD pin from either VIN
or EXTVDD depending on the voltage at the EXTVDD
pin. PVDD powers MOSFET drivers and VDD pin, which
is recommended to connect to PVDD through a low
pass filter, powers the internal circuitry. In the applications where the output voltage is 5V and above (up to
14V), it is recommended to connect EXTVDD to the
output to reduce the power dissipation in the MIC2128,
to reduce the MIC2128 junction temperature and to
improve the system efficiency.
The power dissipation in the MIC2128 depends on the
internal LDO being in use, gate charge of the external
MOSFETs and switching frequency. The power dissipation and the junction temperature of the MIC2128
can be estimated using the Equation 5-32,
Equation 5-33 and Equation 5-34.
Power dissipation in the MIC2128 when EXTVDD is not
used:
EQUATION 5-32:
P IC = V IN ISW + IQ
Power dissipation in the MIC2128 when EXTVDD is
used:
VFB(pp) = Injected Feedback Ripple (20mV to
100mV)
Once all the ripple injection component values are
calculated, ensure that the criteria shown in the
Equation 5-29 is met.
2016-2020 Microchip Technology Inc.
DS20005620F-page 25
MIC2128
EQUATION 5-33:
PIC = V EXTVDD I SW + I Q
I SW = Q G f SW
Q G = Q G_HS + Q G_LS
Where:
ISW
= Switching current into the VIN pin
IQ
= Quiescent current (1.4 mA typ)
QG
= Total gate charge of the external MOSFETs which is sum of the gate charge of
high-side MOSFET (QG_HS) and the
low-side MOSFET (QG_LS) at 5V gate to
source voltage. Gate charge information
can be obtained from the MOSFETs
datasheet.
VEXTVDD
= Voltage at the EXTVDD pin
(4.6 ≤ VEXTVDD ≤ 14 V typ.)
When the 5V output is used as the input to the EXTVDD pin, the MIC2128 junction temperature reduces
from 113°C to 88°C as calculated in Equation 5-36:
EQUATION 5-36:
P IC = 5V 10mA + 1.5mA
PIC = 0.058W
T J = 0.058W 50.8 C W + 85 C
T J = 88 C
The junction temperature of the MIC2128 can be
estimated using Equation 5-34.
EQUATION 5-34:
T J = P IC JA + T A
Where:
TJ
= Junction temperature
PIC
= Power dissipation
θJA
= Junction Ambient Thermal resistance
(50.8°C/W)
The maximum recommended operating junction
temperature for the MIC2128 is 125°C.
Using the output voltage of the same switching
regulator, when it is in between 4.6V (typ.) to 14V, as
the voltage at the EXTVDD pin, significantly reduces
the power dissipation inside the MIC2128. This
reduces the junction temperature rise as illustrated
below.
For the typical case of VVIN = 48V, VOUT = 5V, the
maximum ambient temperature = 85°C and 10 mA of
ISW.
The condition where EXTVDD is not used is shown in
Equation 5-35:
EQUATION 5-35:
·
P IC = 48V 10mA + 1.5 mA
PIC = 0.552W
T J = 0.552W 50.8 C W + 85 C
T J = 113 C
DS20005620F-page 26
2016-2020 Microchip Technology Inc.
MIC2128
6.0
PCB LAYOUT GUIDELINES
PCB layout is critical to achieve reliable, stable and
efficient performance. The following guidelines should
be followed to ensure proper operation of the MIC2128
converter.
6.1
IC
• The ceramic bypass capacitors, which are
connected to the VDD and PVDD pins, must be
located right at the IC. Use wide traces to connect
to the VDD, PVDD and AGND, PGND pins
respectively.
• The signal ground pin (AGND) must be connected
directly to the ground planes.
• Place the IC close to the point-of-load (POL).
• Signal and power grounds should be kept
separate and connected at only one location.
6.2
Input Capacitor
6.4
Output Capacitor
• Use a copper plane to connect the output
capacitor ground terminal to the input capacitor
ground terminal.
• The feedback trace should be separate from the
power trace and connected as close as possible
to the output capacitor. Sensing a long
high-current load trace can degrade the DC load
regulation.
6.5
MOSFETs
• MOSFET gate drive traces must be short and
wide. The ground plane should be the connection
between the MOSFET source and PGND.
• Chose a low-side MOSFET with a high CGS/CGD
ratio and a low internal gate resistance to
minimize the effect of dv/dt inducted turn-on.
• Use a 4.5V VGS rated MOSFET. Its higher gate
threshold voltage is more immune to glitches than
a 2.5V or 3.3V rated MOSFET.
• Place the input ceramic capacitors as close as
possible to the MOSFETs.
• Place several vias to the ground plane close to
the input capacitor ground terminal.
6.3
Inductor
• Keep the inductor connection to the switch node
(SW) short.
• Do not route any digital lines underneath or close
to the inductor.
• Keep the switch node (SW) away from the
feedback (FB) pin.
• The SW pin should be connected directly to the
drain of the low-side MOSFET to accurately
sense the voltage across the low-side MOSFET.
2016-2020 Microchip Technology Inc.
DS20005620F-page 27
MIC2128
7.0
PACKAGING INFORMATION
7.1
Package Marking Information
16-Pin VQFN (3 x 3 mm)
Example
XXXX
WNNN
Legend: XX...X
WW
NNN
e3
*
2256
Customer-specific information
Week code (week of January 1 is week ‘01’)
Alphanumeric traceability code
Pb-free JEDEC designator for Matte Tin (Sn)
This package is Pb-free. The Pb-free JEDEC designator ( e3 )
can be found on the outer packaging for this package.
●, ▲, ▼ Pin one index is identified by a dot, delta up, or delta down (triangle
mark).
Note:
In the event the full Microchip part number cannot be marked on one line, it will
be carried over to the next line, thus limiting the number of available
characters for customer-specific information. Package may or may not include
the corporate logo.
Underbar (_) symbol may not be to scale.
DS20005620F-page 28
2016-2020 Microchip Technology Inc.
MIC2128
Note:
For the most current package drawings, please see the Microchip Packaging Specification located at
http://www.microchip.com/packaging.
2016-2020 Microchip Technology Inc.
DS20005620F-page 29
MIC2128
NOTES:
DS20005620F-page 30
2016-2020 Microchip Technology Inc.
MIC2128
APPENDIX A:
REVISION HISTORY
Revision F (April 2020)
The following is the list of modifications:
1.
2.
Updated the Electrical Characteristics (Note 1)
table.
Updated Equation 4-6.
Revision E (August 2019)
The following is the list of modifications:
1.
2.
3.
Updated the Features section.
Updated the Electrical Characteristics (Note 1)
table.
Updated the Product Identification System
section.
Revision D (March 2019)
The following is the list of modifications:
1.
2.
Updated the Functional Block Diagram.
Updated the ILIM Source Current values in the
Electrical Characteristics (Note 1) table.
Revision C (June 2018)
The following is the list of modifications:
1.
2.
Updated Section 1.0, Electrical Characteristics.
Added information to Section 3.9 “EXTVDD”
and Section 4.7 “Auxiliary Bootstrap LDO
(EXTVDD)”.
Revision B (January 2017)
The following is the list of modifications:
1.
Updated Product Identification System section.
Revision A (September 2016)
• Original release of this document.
2016-2020 Microchip Technology Inc.
DS20005620F-page 31
MIC2128
NOTES:
2016-2020 Microchip Technology Inc.
DS20005620F-page 32
MIC2128
PRODUCT IDENTIFICATION SYSTEM
To order or obtain information, e.g., on pricing or delivery, refer to the factory or the listed sales office.
PART NO.
X
XX
-XX
XXX
Device Temperature Package Code Media Type Qualification
Device:
MIC2128: 75V, Synchronous Buck Controller Featuring
Adaptive On-Time Control with External Soft Start
Temperature:
Y
= Industrial Temperature Grade (-40°C to +125°C)
Package:
ML
=
16 Lead, 3x3 mm VQFN
Media Type:
TR
=
5000/reel
Qualification:
(Blank) =
VAO
=
DS20005620F-page 33
Examples:
a) MIC2128YML-TR:
75V, Synchronous Buck Controller
Featuring Adaptive On-Time Control
with External Soft Start,
-40°C to +125°C junction temp. range,
16-LD VQFN package, 5000/reel
b) MIC2128YML-TRVAO: 75V, Synchronous Buck Controller
Featuring Adaptive On-Time
Control with External Soft Start,
Automotive AEC-Q100 Qualified,
16-LD VQFN package, 5000/reel
Standard Part
Automotive AEC-Q100 Qualified
2016-2020 Microchip Technology Inc.
MIC2128
NOTES:
DS20005620F-page 34
2016-2020 Microchip Technology Inc.
Note the following details of the code protection feature on Microchip devices:
•
Microchip products meet the specification contained in their particular Microchip Data Sheet.
•
Microchip believes that its family of products is one of the most secure families of its kind on the market today, when used in the
intended manner and under normal conditions.
•
There are dishonest and possibly illegal methods used to breach the code protection feature. All of these methods, to our
knowledge, require using the Microchip products in a manner outside the operating specifications contained in Microchip’s Data
Sheets. Most likely, the person doing so is engaged in theft of intellectual property.
•
Microchip is willing to work with the customer who is concerned about the integrity of their code.
•
Neither Microchip nor any other semiconductor manufacturer can guarantee the security of their code. Code protection does not
mean that we are guaranteeing the product as “unbreakable.”
Code protection is constantly evolving. We at Microchip are committed to continuously improving the code protection features of our
products. Attempts to break Microchip’s code protection feature may be a violation of the Digital Millennium Copyright Act. If such acts
allow unauthorized access to your software or other copyrighted work, you may have a right to sue for relief under that Act.
Information contained in this publication regarding device
applications and the like is provided only for your convenience
and may be superseded by updates. It is your responsibility to
ensure that your application meets with your specifications.
MICROCHIP MAKES NO REPRESENTATIONS OR
WARRANTIES OF ANY KIND WHETHER EXPRESS OR
IMPLIED, WRITTEN OR ORAL, STATUTORY OR
OTHERWISE, RELATED TO THE INFORMATION,
INCLUDING BUT NOT LIMITED TO ITS CONDITION,
QUALITY, PERFORMANCE, MERCHANTABILITY OR
FITNESS FOR PURPOSE. Microchip disclaims all liability
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devices in life support and/or safety applications is entirely at
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hold harmless Microchip from any and all damages, claims,
suits, or expenses resulting from such use. No licenses are
conveyed, implicitly or otherwise, under any Microchip
intellectual property rights unless otherwise stated.
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For information regarding Microchip’s Quality Management Systems,
please visit www.microchip.com/quality.
2016-2020 Microchip Technology Inc.
ISBN: 978-1-5224-5877-7
DS20005620F-page 35
Worldwide Sales and Service
AMERICAS
ASIA/PACIFIC
ASIA/PACIFIC
EUROPE
Corporate Office
2355 West Chandler Blvd.
Chandler, AZ 85224-6199
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Technical Support:
http://www.microchip.com/
support
Web Address:
www.microchip.com
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DS20005620F-page 49
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2020 Microchip Technology Inc.
02/28/20
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