AD677BD

a FEATURES Autocalibrating On-Chip Sample-Hold Function Serial Output 16 Bits No Missing Codes 61 LSB INL –99 dB THD 92 dB S/(N+D) 1 MHz Full Power Bandwidth 16-Bit 100 kSPS Sampling ADC AD677 FUNCTIONAL BLOCK DIAGRAM VIN AGND SENSE VR E F AGND A CHIP 10 9 11 INPUT BUFFERS 16-BIT DAC COMP CAL DAC 8 LOGIC TIMING LEVEL TRANSLATORS 15 BUSY 14 SCLK CAL 16 CLK 2 SAMPLE 1 MICROCODED CONTROLLER SAR 3 SDATA ALU RAM D CHIP AD677 PRODUCT DESCRIPTION PRODUCT HIGHLIGHTS The AD677 is a multipurpose 16-bit serial output analog-todigital converter which utilizes a switched-capacitor/charge redistribution architecture to achieve a 100 kSPS conversion rate (10 µs total conversion time). Overall performance is optimized by digitally correcting internal nonlinearities through on-chip autocalibration. 1. Autocalibration provides excellent dc performance while eliminating the need for user adjustments or additional external circuitry. 2. ± 5 V to ± 10 V input range (± VREF). 3. Available in 16-pin 0.3" skinny DIP or 28-lead SOIC. The AD677 circuitry is segmented onto two monolithic chips— a digital control chip fabricated on Analog Devices DSP CMOS process and an analog ADC chip fabricated on our BiMOS II process. Both chips are contained in a single package. 4. Easy serial interface to standard ADI DSPs. The AD677 is specified for ac (or “dynamic”) parameters such as S/(N+D) Ratio, THD and IMD which are important in signal processing applications. In addition, dc parameters are specified which are important in measurement applications. 7. Industry leading dc performance: 1.0 LSB INL, ± 1 LSB full scale and offset. 5. TTL compatible inputs/outputs. 6. Excellent ac performance: –99 dB THD, 92 dB S/(N+D) peak spurious –101 dB. The AD677 operates from +5 V and ± 12 V supplies and typically consumes 450 mW using a 10 V reference (360 mW with 5 V reference) during conversion. The digital supply (VDD) is separated from the analog supplies (VCC, VEE) for reduced digital crosstalk. An analog ground sense is provided to remotely sense the ground potential of the signal source. This can be useful if the signal has to be carried some distance to the A/D converter. Separate analog and digital grounds are also provided. The AD677 is available in a 16-pin narrow plastic DIP, 16-pin narrow side-brazed ceramic package, or 28-lead SOIC. A parallel output version, the AD676, is available in a 28-pin ceramic or plastic DIP. All models operate over a commercial temperature range of 0°C to +70°C or an industrial range of –40°C to +85°C. REV. A Information furnished by Analog Devices is believed to be accurate and reliable. However, no responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other rights of third parties which may result from its use. No license is granted by implication or otherwise under any patent or patent rights of Analog Devices. One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A. Tel: 617/329-4700 Fax: 617/326-8703 AD677–SPECIFICATIONS AC SPECIFICATIONS (T MIN to TMAX, VCC = +12 V 6 5%, VEE = –12 V 6 5%, VDD = +5 V 6 10%)1 Parameter Min Total Harmonic Distortion (THD)2 @ 83 kSPS, TMIN to TMAX @ 100 kSPS, +25°C @ 100 kSPS, TMIN to TMAX Signal-to-Noise and Distortion Ratio (S/(N+D))2, 3 @ 83 kSPS, TMIN to TMAX @ 100 kSPS, +25°C @ 100 kSPS, TMIN to TMAX Peak Spurious or Peak Harmonic Component Intermodulation Distortion (IMD)4 2nd Order Products 3rd Order Products Full Power Bandwidth Noise –97 –97 –93 89 89 MIN LOGIC INPUTS VIH High Level Input Voltage VIL Low Level Input Voltage IIH High Level Input Current IIL Low Level Input Current CIN Input Capacitance LOGIC OUTPUTS VOH High Level Output Voltage VOL Low Level Output Voltage Max Min AD677K/B Typ Max –92 –92 91 91 89 –101 –99 –99 –95 90 90 –102 –98 1 160 DIGITAL SPECIFICATIONS (for all grades T Parameter AD677J/A Typ dB dB dB 92 92 90 –101 dB dB dB dB –102 –98 1 160 dB dB MHz µV rms to TMAX, VCC = +12 V 6 5%, VEE = –12 V 6 5%, VDD = +5 V 6 10%) Test Conditions Min VIH = VDD VIL = 0 V 2.0 –0.3 –10 –10 Typ Max Units VDD + 0.3 0.8 +10 +10 V V µA µA pF 10 IOH = 0.1 mA IOH = 0.5 mA IOL = 1.6 mA –95 –95 Units VDD – 1 V 2.4 0.4 V V V NOTES 1 VREF = 10.0 V, Conversion Rate = 100 kSPS, f lN = 1.0 kHz, V IN = –0.05 dB, Bandwidth = 50 kHz unless otherwise indicated. All measurements referred to a 0 dB (20 V p-p) input signal. Values are post-calibration. 2 For other input amplitudes, refer to Figure 12. 3 For dynamic performance with different reference values see Figure 11. 4 fa = 1008 Hz, fb = 1055 Hz. See Definition of Specifications section and Figure 16. Specifications subject to change without notice. –2– REV. A AD677 DC SPECIFICATIONS (T 1 MIN to TMAX, VCC = +12 V 6 5%, VEE = –12 V 6 5%, VDD = +5 V 6 1O%) Parameter AD677J/A Typ Max Min TEMPERATURE RANGE J, K Grades A, B Grades ACCURACY Resolution Integral Nonlinearity (INL) @ 83 kSPS, TMIN to TMAX @ 100 kSPS, +25°C @ 100 kSPS, TMIN to TMAX Differential Nonlinearity (DNL)–No Missing Codes Bipolar Zero Error2 Positive, Negative FS Errors2 @ 83 kSPS @ 100 kSPS, +25°C @ 100 kSPS 0 –40 +70 +85 16 AD677K/B Typ Max 0 –40 +70 +85 16 ±1 ±1 ±2 16 ±2 ±2 ±2 ±4 TEMPERATURE DRIFT3 Bipolar Zero Postive Full Scale Negative Full Scale VOLTAGE REFERENCE INPUT RANGE4 (VREF) Min ±4 16 ±4 ±4 ± 0.5 ± 0.5 ± 0.5 5 °C °C Bits ±1 +1 ±2 ± 1.5 ± 1.5 ±1 ±3 ±1 ±1 ±4 ±3 ±3 ± 0.5 ± 0.5 ± 0.5 10 Units 5 LSB LSB LSB Bits LSB LSB LSB LSB LSB LSB LSB 10 V ± VREF V 50* 5 ANALOG INPUT Input Range (VIN) Input Impedance Input Settling Time Input Capacitance During Sample Aperture Delay Aperture Jitter ± VREF * 2 * 2 6 100 6 100 µs pF ns ps ± 0.5 ± 0.5 ± 0.5 ± 0.5 ± 0.5 ± 0.5 LSB LSB LSB 50* POWER SUPPLIES Power Supply Rejection6 VCC = +12 V ± 5% VEE = –12 V ± 5% VDD = +5 V ± 10% Operating Current VREF = +5 V ICC IEE IDD Power Consumption VREF = +10 V ICC IEE IDD Power Consumption 14.5 14.5 3 360 18 18 5 480 14.5 14.5 3 360 18 18 5 480 mA –mA mA mW 18 18 3 450 24 24 5 630 18 18 3 450 24 24 5 630 mA –mA mA mW NOTES 1 VREF = 10.0 V, Conversion Rate = 100 kSPS unless otherwise noted. Values are post-calibration. 2 Values shown apply to any temperature from T MIN to TMAX after calibration at that temperature at nominal supplies. 3 Values shown are based upon calibration at +25°C with no additional calibration at temperature. Values shown are the typical variation from the value at +25 °C. 4 See “APPLICATIONS” section for recommended voltage reference circuit, and Figure 11 for dynamic performance with other reference voltage values. 5 See “APPLICATIONS” section for recommended input buffer circuit. 6 Typical deviation of bipolar zero, –full scale or +full scale from min to max rating. *For explanation of input characteristics, see “ANALOG INPUT” section. Specifications subject to change without notice. REV. A –3– AD677 TIMING SPECIFICATIONS (T MIN to TMAX, VCC = +12 V 6 5%, VEE = –12 V 6 5%, VDD = +5 V 6 10%)1 Parameter 2, 3 Conversion Period CLK Period4 Calibration Time Sampling Time Last CLK to SAMPLE Delay5 SAMPLE Low SAMPLE to Busy Delay 1st CLK Delay CLK Low6 CLK High6 CLK to BUSY Delay CLK to SDATA Valid CLK to SCLK High SCLK Low SDATA to SCLK High CAL High Time CAL to BUSY Delay Symbol Min tC tCLK tCT tS tLCS tSL tSS tFCD tCL tCH tCB tCD tCSH tSCL tDSH tCALH tCALB 10 480 Typ Max Units 1000 µs ns tCLK µs µs ns ns ns ns ns ns ns ns ns ns ns ns 85532 2 2.1 100 30 75 180 100 180 80 80 300 175 300 15 50 50 50 50 50 100 50 50 50 NOTES 1 See the “CONVERSION CONTROL” and “AUTOCALIBRATION” sections for detailed explanations of the above timing. 2 Depends upon external clock frequency; includes acquisition time and conversion time. The maximum conversion period is specified to account for the droop of the internal sample/hold function. Operation at slower rates may degrade performance. 3 tC = tFCD + 16 × tCLK + tLCS. 4 580 ns is recommended for optimal accuracy over temperature (not necessary during calibration cycle). 5 If SAMPLE goes high before the 17th CLK pulse, the device will start sampling approximately 100 ns after the rising edge of the 17th CLK pulse. 6 tCH + tCL = tCLK and must be greater than 480 ns. CAL (INPUT) tCALH tCT tCALB BUSY (OUTPUT) tFCD tCB CLK* (INPUT) 2 1 3 85531 85530 tCH 85532 tCL tCLK *SHADED PORTIONS OF INPUT SIGNALS ARE OPTIONAL. FOR BEST PERFORMANCE, WE RECOMMEND THAT THESE SIGNALS BE HELD LOW EXCEPT WHEN EXPLICITY SHOWN HIGH. Figure 1. Calibration Timing tS tC tSL SAMPLE* (INPUT) tS tSB BUSY (OUTPUT) tCB tFCD tLCS tCH CLK* (INPUT) 1 tCLK tCL 2 15 3 16 17 tCSH SCLK (OUTPUT) tSCL tDSH tCD SDATA (OUTPUT) OLD BIT 16 MSB BIT 2 BIT 13 BIT 14 BIT 15 BIT 16 *SHADED PORTIONS OF INPUT SIGNALS ARE OPTIONAL. FOR BEST PERFORMANCE, WE RECOMMEND THAT THESE SIGNALS BE HELD LOW EXCEPT WHEN EXPLICITY SHOWN HIGH. Figure 2. General Conversion Timing –4– REV. A AD677 ORDERING GUIDE Model Temperature Range S/(N+D) Max INL Package Description Package Option* AD677JN AD677KN AD677JD AD677KD AD677JR AD677KR AD677AD AD677BD 0°C to +70°C 0°C to +70°C 0°C to +70°C 0°C to +70°C 0°C to +70°C 0°C to +70°C –40°C to +85°C –40°C to +85°C 89 dB 90 dB 89 dB 90 dB 89 dB 90 dB 89 dB 90 dB Typ Only ± 1.5 LSB Typ Only ± 1.5 LSB Typ Only ± 1.5 LSB Typ Only ± 1.5 LSB Plastic 16-Pin DIP Plastic 16-Pin DIP Ceramic 16-Pin DIP Ceramic 16-Pin DIP Plastic 28-Lead SOIC Plastic 28-Lead SOIC Ceramic 16-Pin DIP Ceramic 16-Pin DIP N-16 N-16 D-16 D-16 R-28 R-28 D-16 D-16 *D = Ceramic DIP; N = Plastic DIP; R = Small Outline IC (SOIC). ABSOLUTE MAXIMUM RATINGS* VCC to VEE . . . . . . . . . . . . . . . . . . . . . . . . . . . . –0.3 V to +26.4 V VDD to DGND . . . . . . . . . . . . . . . . . . . . . . . . . . . . –0.3 V to +7 V Vcc to AGND . . . . . . . . . . . . . . . . . . . . . . . . . . . –0.3 V to +18 V VEE to AGND . . . . . . . . . . . . . . . . . . . . . . . . . . –18 V to +0.3 V AGND to DGND . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . +0.3 V Digiul Inputs to DGND . . . . . . . . . . . . . . . . . . . . . . 0 to +5.5 V Analog Inputs, VREF to AGND . . . . . . . . . . . . . . . . . . . . . . . . . . . . (VCC +0.3 V) to (VEE –0.3 V) Soldering . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . +300°C, 10 sec Storage Temperature . . . . . . . . . . . . . . . . . . . . . –65°C to +150°C *Stresses greater than those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. This is a stress rating only and functional operation of the device at these or any other conditions above those indicated in the operational section of this specification is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability. CAUTION ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily accumulate on the human body and test equipment and can discharge without detection. Although the AD677 features proprietary ESD protection circuitry, permanent damage may occur on devices subjected to high energy electrostatic discharges. Therefore, proper ESD precautions are recommended to avoid performance degradation or loss of functionality. REV. A –5– WARNING! ESD SENSITIVE DEVICE AD677 PIN DESCRIPTION DIP Pin SOIC Pin Type Name Description 1 1 SAMPLE DI VIN Acquisition Control Pin. Active HIGH. During conversion, SAMPLE controls the suite of the internal sample-hold amplifier and the falling edge initiates conversion. During calibration, SAMPLE should be held LOW. If HIGH during calibration, diagnostic information will appear on SDATA. 2 2 CLK DI Master Clock Input. The AD677 requires 17 clock pulses to execute a conversion. CLK is also used to derive SCLK. 3 3 SDATA DO Serial Output Data Controlled by SCLK. 4 6, 7 DGND P Digital Ground. 5 8 VCC P +12 V Analog Supply Voltage. 8 12 AGND P Analog Ground. .9 15 AGND SENSE AI Analog Ground Sense. 10 16 VIN AI Analog Input Voltage. 11 17 VREF AI External Voltage Reference Input. 12 21 VEE P –12 V Analog Supply Voltage. 13 22, 23 VDD P +5 V Logic Supply Voltage. 14 26 SCLK DO Clock Output for Data Read, derived from CLK. 15 27 BUSY DO Status Line for Converter. Active HIGH, indicating a conversion or calibration in progress. 16 28 CAL DI Calibration Control Pin. 6, 7 4, 5, 9, 10, 11, 13, 14, 18, 19, 20, 24, 25 NC _ No Connection. No connections should be made to these pins. Type: AI = Analog Input DI = Digital Input DO = Digital Output P = Power SAMPLE CLK 1 16 CAL 2 15 BUSY SDATA 3 DGND 4 VCC 5 14 SCLK AD677 13 VDD TOP VIEW (Not to Scale) 12 VEE 1 28 CAL CLK 2 27 BUSY SDATA 3 26 SCLK NC 4 25 NC NC 5 24 NC DGND1 6 23 VDD1 DGND2 7 AD677 22 VDD2 VCC 8 TOP VIEW (Not to Scale) 21 VEE 20 NC VREF NC 6 11 NC 7 10 VIN AGND 8 9 SAMPLE AGND SENSE NC = NO CONNECT DIP Pinout NC 9 NC 10 19 NC NC 11 18 NC AGND 12 17 VREF NC 13 16 VIN NC 14 15 AGND SENSE NC = NO CONNECT SOIC Pinout –6– REV. A Definition of Specifications–AD677 NYQUIST FREQUENCY INTERMODULATION DISTORTION (IMD) An implication of the Nyquist sampling theorem, the “Nyquist frequency’’ of a converter is that input frequency which is one half the sampling frequency of the converter. With inputs consisting of sine waves at two frequencies, fa and fb, any device with nonlinearities will create distortion products, of order (m+n), at sum and difference frequencies of mfa ± nfb, where m, n = 0, 1, 2, 3 . . . . Intermodulation terms are those for which m or n is not equal to zero. For example, the second order terms are (fa + fb) and (fa – fb), and the third order terms are (2 fa + fb), (2 fa – fb), (fa + 2 fb) and (fa – 2 fb). The IMD products are expressed as the decibel ratio of the rms sum of the measured input signals to the rms sum of the distortion terms. The two signals applied to the converter are of equal amplitude, and the peak value of their sum is –0.5 dB from full scale. The IMD products are normalized to a 0 dB input signal. TOTAL HARMONIC DISTORTION Total harmonic distortion (THD) is the ratio of the rms sum of the harmonic components to the rms value of a full-scale input signal and is expressed in percent (%) or decibels (dB). For input signals or harmonics that are above the Nyquist frequency, the aliased components are used. SIGNAL-TO-NOISE PLUS DISTORTION RATIO Signal-to-noise plus distortion is defined to be the ratio of the rms value of the measured input signal to the rms sum of all other spectral components below the Nyquist frequency, including harmonics but excluding dc. APERTURE DELAY Aperture delay is the time required after SAMPLE pin is taken LOW for the internal sample-hold of the AD677 to open, thus holding the value of VIN. +/– FULL-SCALE ERROR The last + transition (from 011 . . . 10 to 011 . . . 11) should occur for an analog voltage 1.5 LSB below the nominal full scale (4.99977 volts for a ± 5 V range). The full-scale error is the deviation of the actual level of the last transition from the ideal level. BIPOLAR ZERO ERROR Bipolar zero error is the difference between the ideal midscale input voltage (0 V) and the actual voltage producing the midscale output code. DIFFERENTIAL NONLINEARITY (DNL) In an ideal ADC, code transitions are one LSB apart. Differential nonlinearity is the maximum deviation from this ideal value. It is often specified in terms of resolution for which no missing codes are guaranteed. INTEGRAL NONLINEARITY (INL) The ideal transfer function for an ADC is a straight line bisecting the center of each code drawn between “zero” and “full scale.” The point used as “zero” occurs 1/2 LSB before the most negative code transition. “Full scale” is defined as a level 1.5 LSB beyond the most positive code transition. Integral nonlinearity is the worst-case deviation of a code center average from the straight line. BANDWIDTH The full-power bandwidth is that input frequency at which the amplitude of the reconstructed fundamental is reduced by 3 dB for a full-scale input. REV. A APERTURE JITTER Aperture jitter is the variation in the aperture delay from sample to sample. POWER SUPPLY REJECTION DC variations in the power supply voltage will affect the overall transfer function of the ADC, resulting in zero error and fullscale error changes. Power supply rejection is the maximum change in either the bipolar zero error or full-scale error value. Additionally, there is another power supply variation to consider. AC ripple on the power supplies can couple noise into the ADC, resulting in degradation of dynamic performance. This is displayed in Figure 15. INPUT SETTLING TIME Settling time is a function of the SHA’s ability to track fast slewing signals. This is specified as the maximum time required in track mode after a full-scale step input to guarantee rated conversion accuracy. NOISE/DC CODE UNCERTAINTY Ideally, a fixed dc input should result in the same output code for repetitive conversions. However, as a consequence of unavoidable circuit noise within the wideband circuits in the ADC, there is a range of output codes which may occur for a given input voltage. If you apply a dc signal to the ADC and record a large number of conversions, the result will be a distribution of codes. If you fit a Gaussian probability distribution to the histogram, the standard deviation is approximately equivalent to the rms input noise of the ADC. –7– AD677 FUNCTIONAL DESCRIPTION The AD677 is a multipurpose 16-bit analog-to-digital converter and includes circuitry which performs an input sample/hold function, ground sense, and autocalibration. These functions are segmented onto two monolithic chips—an analog signal processor and a digital controller. Both chips are contained within the AD677 package. The AD677 employs a successive-approximation technique to determine the value of the analog input voltage. However, instead of the traditional laser-trimmed resistor-ladder approach, this device uses a capacitor-array, charge redistribution technique. Binary-weighted capacitors subdivide the input sample to perform the actual analog-to-digital conversion. The capacitor array eliminates variation in the linearity of the device due to temperature-induced mismatches of resistor values. Since a capacitor array is used to perform the data conversions, the sample/hold function is included without the need for additional external circuitry. Initial errors in capacitor matching are eliminated by an autocalibration circuit within the AD677. This circuit employs an on-chip microcontroller and a calibration DAC to measure and compensate capacitor mismatch errors. As each error is determined, its value is stored in on-chip memory (RAM). Subsequent conversions use these RAM values to improve conversion accuracy. The autocalibration routine may be invoked at any time. Autocalibration insures high performance while eliminating the need for any user adjustments and is described in detail below. The microcontroller controls all of the various functions within the AD677. These include the actual successive approximation algorithm, the autocalibration routine, the sample/hold operation, and the internal output data latch. AUTO CALIBRATION The AD677 achieves rated performance without the need for user trims or adjustments. This is accomplished through the use of on-chip autocalibration. In the autocalibration sequence, sample/hold offset is nulled by internally connecting the input circuit to the ground sense circuit. The resulting offset voltage is measured and stored in RAM for later use. Next, the capacitor representing the most significant bit (MSB) is charged to the reference voltage. This charge is then transferred to a capacitor of equal size (composed of the sum of the remaining lower weight bits). The voltage that results represents the amount of capacitor mismatch. A calibration digital-to-analog converter (DAC) adds an appropriate value of error correction voltage to cancel this mismatch. This correction factor is also stored in RAM. This process is repeated for each of the eight remaining capacitors representing the top nine bits. The accumulated values in RAM are then used during subsequent conversions to adjust conversion results accordingly. As shown in Figure 1, when CAL is taken HIGH the AD677 internal circuitry is reset, the BUSY pin is driven HIGH, and the ADC prepares for calibration. This is an asynchronous hardware reset and will interrupt any conversion or calibration currently in progress. Actual calibration begins when CAL is taken LOW and completes in 85,532 clock cycles, indicated by BUSY going LOW. During calibration, it is preferable for SAMPLE to be held LOW. If SAMPLE is HIGH, diagnostic data will appear on SDATA. This data is of no value to the user. In most applications, it is sufficient to calibrate the AD677 only upon power-up, in which case care should be taken that the power supplies and voltage reference have stabilized first. If calibration is not performed, the AD677 may come up in an unknown state, or performance could degrade to as low as 10 bits. CONVERSION CONTROL The AD677 is controlled by two signals: SAMPLE and CLK, as shown in Figure 2. It is assumed that the part has been calibrated and the digital I/O pins have the levels shown at the start of the timing diagram. A conversion consists of an input acquisition followed by 17 clock pulses which execute the 16-bit internal successive approximation routine. The analog input is acquired by taking the SAMPLE line HIGH for a minimum sampling time of tS. The actual sample taken is the voltage present on VIN one aperture delay after the SAMPLE line is brought LOW, assuming the previous conversion has completed (signified by BUSY going LOW). Care should be taken to ensure that this negative edge is well defined and jitter free in ac applications to reduce the uncertainty (noise) in signal acquisition. With SAMPLE going LOW, the AD677 commits itself to the conversion—the input at VIN is disconnected from the internal capacitor array, BUSY goes HIGH, and the SAMPLE input will be ignored until the conversion is completed (when BUSY goes LOW). SAMPLE must be held LOW for a minimum period of time tSL. A period of time tFCD after bringing SAMPLE LOW, the 17 CLK cycles are applied; CLK pulses that start before this period of time are ignored. BUSY goes HIGH tSB after SAMPLE goes LOW, signifying that a conversion is in process, and remains HIGH until the conversion is completed. As indicated in Figure 2, the twos complement output data is presented MSB first. This data may be captured with the rising edge of SCLK or the falling edge of CLK, beginning with pulse #2. The AD677 will ignore CLK after BUSY has gone LOW and SDATA or SCLK will not change until a new sample is acquired. CONTINUOUS CONVERSION For maximum throughput rate, the AD677 can be operated in a continuous convert mode. This is accomplished by utilizing the fact that SAMPLE will no longer be ignored after BUSY goes LOW, so an acquisition may be initiated even during the HIGH time of the 17th CLK pulse for maximum throughput rate while enabling full settling of the sample/hold circuitry. If SAMPLE is already HIGH during the rising edge of the 17th CLK, then an acquisition is immediately initiated approximately 100 ns after the rising edge of the 17th clock pulse. Care must be taken to adhere to the minimum/maximum timing requirements in order to preserve conversion accuracy. GENERAL CONVERSION GUIDELINES During signal acquisition and conversion, care should be taken with the logic inputs to avoid digital feedthrough noise. It is possible to run CLK continuously, even during the sample period. However, CLK edges during the sampling period, and especially when SAMPLE goes LOW, may inject noise into the sampling process. The AD677 is tested with no CLK cycles during the sampling period. The BUSY signal can be used to prevent the clock from running during acquisition, as illustrated –8– REV. A AD677 in Figure 3. In this circuit BUSY is used to reset the circuitry which divides the system clock down to provide the AD677 CLK. This serves to interrupt the clock until after the input signal has been acquired, which has occurred when BUSY goes HIGH. When the conversion is completed and BUSY goes LOW, the circuit in Figure 3 truncates the 17th CLK pulse width which is tolerable because only its rising edge is critical. 12.288MHz SYSTEM CLOCK 11 3Q 2Q 7 4 1D 3D 12 9 CLK CLR 1 1Q 2 BUSY 6 1QD 2QC 9 Output Code 0 Figure 5a. Input to the A/D is Corrupted by IR Drop in Ground Leads: VIN = VS + ∆V. BOARD LAYOUT Designing with high resolution data converters requires careful attention to board layout. Trace impedance is a significant issue. A 1.22 mA current through a 0.5 Ω trace will develop a voltage drop of 0.6 mV, which is 4 LSBs at the 16-bit level for a 10 V full-scale span. In addition to ground drops, inductive and capacitive coupling need to be considered, especially when high accuracy analog signals share the same board with digital signals. Analog and digital signals should not share a common return path. Each signal should have an appropriate analog or digital return routed close to it. Using this approach, signal loops enclose a small area, minimizing the inductive coupling of noise. Wide PC tracks, large gauge wire, and ground planes are highly recommended to provide low impedance signal paths. Separate analog and digital ground planes are also desirable, with a single interconnection point at the AD677 to minimize interference between analog and digital circuitry. Analog signals should be routed as far as possible from digital signals and should cross them, if at all, only at right angles. A solid analog ground plane around the AD677 will isolate it from large switching ground currents. For these reasons, the use of wire wrap circuit construction will not provide adequate performance; careful printed circuit board construction is preferred. AGND SHIELDED CABLE AD677 VIN SOURCE VS AGND SENSE AGND GROUND LEAD I GROUND > 0 TO POWER SUPPLY GND Figure 5b. AGND SENSE Eliminates the Problem in Figure 5a. shielded in a noisy environment to avoid capacitive coupling. If inductive (magnetic) coupling is expected to be dominant such as where motors are present, twisted-pair wires should be used instead. The digital ground pin is the reference point for all of the digital signals that operate the AD677. This pin should be connected to the digital common point in the system. As Figure 4 illustrated, the analog and digital grounds should be connected together at one point in the system, preferably at the AD677. GROUNDING The AD677 has three grounding pins, designated ANALOG GROUND (AGND), DIGITAL GROUND (DGND) and ANALOG GROUND SENSE (AGND SENSE). The analog ground pin is the “high quality” ground reference point for the device, and should be connected to the analog common point in the system. AGND SENSE is intended to be connected to the input signal ground reference point. This allows for slight differences in level between the analog ground point in the system and the input signal ground point. However no more than 100 mV is recommended between the AGND and the AGND SENSE pins for specified performance. Using AGND SENSE to remotely sense the ground potential of the signal source can be useful if the signal has to be carried some distance to the A/D converter. Since all IC ground currents have to return to the power supply and no ground leads are free from resistance and inductance, there are always some voltage differences from one ground point in a system to another. Over distance this voltage difference can easily amount to several LSBs (in a 10 V input span, 16-bit system each LSB is about 0.15 mV). This would directly corrupt the A/D input signal if the A/D measures its input with respect to power ground (AGND) as shown in Figure 5a. To solve this problem the AD677 offers an AGND SENSE pin. Figure 5b shows how the AGND SENSE can be used to eliminate the problem in Figure 5a. Figure 5b also shows how the signal wires should be VOLTAGE REFERENCE The AD677 requires the use of an external voltage reference. The input voltage range is determined by the value of the reference voltage; in general, a reference voltage of n volts allows an input range of ± n volts. The AD677 is specified for a voltage reference between +5 V and +10 V. A 10 V reference will typically require support circuitry operated from ± 15 V supplies; a 5.0 V reference may be used with ± 12 V supplies. Signal-tonoise performance is increased proportionately with input signal range (see Figure 12). In the presence of a fixed amount of system noise, increasing the LSB size (which results from increasing the reference voltage) will increase the effective S/(N+D) performance. Figure 11 illustrates S/(N+D) as a function of reference voltage. In contrast, dc accuracy will be optimal at lower reference voltage values (such as 5 V) due to capacitor nonlinearity at higher voltage values. During a conversion, the switched capacitor array of the AD677 presents a dynamically changing current load at the voltage reference as the successive-approximation algorithm cycles through various choices of capacitor weighting. (See the following section “Analog Input” for a detailed discussion of the VREF input characteristics.) The output impedance of the reference circuitry must be low so that the output voltage will remain sufficiently constant as the current drive changes. In some applications, this may require that the output of the voltage reference be buffered by an amplifier with low impedance at relatively high frequencies. In choosing a voltage reference, consideration should be –10– REV. A AD677 made for selecting one with low noise. A capacitor connected between REF IN and AGND will reduce the demands on the reference by decreasing the magnitude of high frequency components required to be sourced by the reference. regulator prevents very large voltage spikes from entering the regulators. Any power line noise which the regulators cannot eliminate will be further filtered by an RC filter (10 Ω/10 µF) having a –3 dB point at 1.6 kHz. For best results the regulators should be within a few centimeters of the AD677. Figures 6 and 7 represent typical design approaches. ANALOG INPUT +12V As previously discussed, the analog input voltage range for the AD677 is ± VREF. For purposes of ground drop and common mode rejection, the VIN and VREF inputs each have their own ground. VREF is referred to the local analog system ground (AGND), and VIN is referred to the analog ground sense pin (AGND SENSE) which allows a remote ground sense for the input signal. 2 VIN 8 CN 6 AD586 1.0µF VREF 10µF AD677 0.1µF 4 AGND Figure 6. Figure 6 shows a voltage reference circuit featuring the 5 V output AD586. The AD586 is a low cost reference which utilizes a buried Zener architecture to provide low noise and drift. Over the 0°C to +70°C range, the AD586M grade exhibits less than 1.0 mV output change from its initial value at +25°C. A noise reduction capacitor, CN, reduces the broadband noise of the AD586 output, thereby optimizing the overall performance of the AD677. It is recommended that a 10 µF to 47 µF high quality tantalum capacitor and a 0.1 µF capacitor be tied between the VREF input of the AD677 and ground to minimize the impedance on the reference. Using the AD677 with ± 10 V input range (VREF = 10 V) typically requires ± 15 V supplies to drive op amps and the voltage reference. If ± 12 V is not available in the system, regulators such as 78L12 and 79L12 can be used to provide power for the AD677. This is also the recommended approach (for any input range) when the ADC system is subjected to harsh environments such as where the power supplies are noisy and where voltage spikes are present. Figure 7 shows an example of such a system based upon the 10 V AD587 reference, which provides a 300 µV LSB. Circuitry for additional protection against power supply disturbances has been shown. A 100 µF capacitor at each 2 VIN 10µF VO 6 NR 8 GND 4 0.1µF In most cases, these characteristics require the use of an external op amp to drive the input of the AD677. Care should be taken with op amp selection; even with modest loading conditions, most available op amps do not meet the low distortion requirements necessary to match the performance capabilities of the AD677. Figure 8 represents a circuit, based upon the AD845, which will provide excellent overall performance. For applications optimized more for low distortion and low noise, the AD845 of Figure 8 may be replaced by the AD743. AD587 10Ω The AD677 analog inputs (VIN, VREF and AGND SENSE) exhibit dynamic characteristics. When a conversion cycle begins, each analog input is connected to an internal, discharged 50 pF capacitor which then charges to the voltage present at the corresponding pin. The capacitor is disconnected when SAMPLE is taken LOW, and the stored charge is used in the subsequent conversion. In order to limit the demands placed on the external source by this high initial charging current, an internal buffer amplifier is employed between the input and this capacitance for a few hundred nanoseconds. During this time the input pin exhibits typically 20 kΩ input resistance, 10 pF input capacitance and ± 40 µA bias current. Next, the input is switched directly to the now precharged capacitor and allowed to fully settle. During this time the input sees only a 50 pF capacitor. Once the sample is taken, the input is internally floated so that the external input source sees a very high input resistance and a parasitic input capacitance of typically only 2 pF. As a result, the only dominant input characteristic which must be considered is the high current steps which occur when the internal buffers are switched in and out. 1k Ω 1µF ±5V INPUT 10Ω +15V 78L12 100µF 0.01µF VCC VDD +5V VEE 10Ω –15V 100µF 3 10µF 0.1µF AD677 4 6 –12V AGND SENSE 79L12 0.01µF VIN 0.1µF AGND VIN 10µF 0.1µF Figure 8. VIN Figure 7. REV. A 0.1µF 7 AD845 499 Ω VREF AD677 0.1µF 100µF 2 0.1µF 10µF 10Ω +12V 1k Ω –11– AD677 AC parameters, which include S/(N+D), THD, etc., reflect the AD677’s effect on the spectral content of the analog input signal. Figures 11 through 18 provide information on the AD677’s ac performance under a variety of conditions. A perfect n-bit ADC with no errors will yield a theoretical quantization noise of q/√12, where q is the weight of the LSB. This relationship leads to the well-known equation for theoretical full-scale rms sine wave signal-to-noise plus distortion level of S/(N + D) = 6.02 n + 1.76 dB, here n is the bit resolution. An actual ADC, however, will yield a measured S/(N + D) less than the theoretical value. Solving this equation for n using the measured S/(N + D) value yields the equation for effective number of bits (ENOB): ENOB = [S / ( N + D )] ACTUAL – 1.76 dB 6.02 As a general rule, averaging the results from several conversions reduces the effects of noise, and therefore improves such parameters as S/(N+D). AD677 performance may be optimized by operating the device at its maximum sample rate of 100 kSPS and digitally filtering the resulting bit stream to the desired signal bandwidth. This succeeds in distributing noise over a wider frequency range, thus reducing the noise density in the frequency band of interest. This subject is discussed in the following section. OVERSAMPLING AND NOISE FILTERING The Nyquist rate for a converter is defined as one-half its sampling rate. This is established by the Nyquist theorem, which requires that a signal be sampled at a rate corresponding to at least twice its highest frequency component of interest in order to preserve the informational content. Oversampling is a conversion technique in which the sampling frequency is more than twice the frequency bandwidth of interest. In audio applications, the AD677 can operate at a 2 × FS oversampling rate, where FS = 48 kHz. In quantized systems, the informational content of the analog input is represented in the frequency spectrum from dc to the Nyquist rate of the converter. Within this same spectrum are higher frequency noise and signal components. Antialias, or low pass, filters are used at the input to the ADC to reduce these noise and signal components so that their aliased components do not corrupt the baseband spectrum. However, wideband noise contributed by the AD677 will not be reduced by the antialias filter. The AD677 quantization noise is evenly distributed from dc to the Nyquist rate, and this fact can be used to minimize its overall affect. FS is the sampling frequency, and Fa is the signal bandwidth of interest. For audio bandwidth applications, the AD677 is capable of operating at a 2 × oversample rate (96 kSPS), which typically produces an improvement in S/(N+D) of 3 dB compared with operating at the Nyquist conversion rate of 48 kSPS. Oversampling has another advantage as well; the demands on the antialias filter are lessened. In summary, system performance is optimized by running the AD677 at or near its maximum sampling rate of 100 kHz and digitally filtering the resulting spectrum to eliminate undesired frequencies. DC PERFORMANCE The self-calibration scheme used in the AD677 compensates for bit weight errors that may exist in the capacitor array. This mismatch in capacitor values is adjusted (using the calibration coefficients) during conversion and provides for excellent dc linearity performance. Figure 19 illustrates the DNL plot of a typical AD677 at +25°C. A histogram test is a statistical method for deriving an A/D converter’s differential nonlinearity. A ramp input is sampled by the ADC and a large number of conversions are taken and stored. Theoretically the codes would all be the same size and, therefore, have an equal number of occurrences. A code with an average number of occurrences would have a DNL of “0”. A code with more or less than average will have a DNL of greater than or less than zero LSB. A DNL of –1 LSB indicates missing code (zero occurrences). Figure 20 illustrates the code width distribution of the DNL plots of Figure 19. DC CODE UNCERTAINTY Ideally, a fixed dc input should result in the same output code for repetitive conversions. However, as a consequence of unavoidable circuit noise within the wideband circuits in the ADC, there is range of output codes which may occur for a given input voltage. If you apply a dc signal to the AD677 and record 10,000 conversions, the result will be a distribution of codes as shown in Figure 9 (using a 10 V reference). If you fit a Gaussian probability distribution to the histogram, the standard deviation is approximately equivalent to the rms input noise of ADC. 7649 7000 The AD677 quantization noise effects can be reduced by oversampling—sampling at a rate higher than that defined by the Nyquist theorem. This spreads the noise energy over a bandwidth wider than the frequency band of interest. By judicious selection of a digital decimation filter, noise frequencies outside the bandwidth of interest may be eliminated. The process of analog to digital conversion inherently produces noise, known as quantization noise. The magnitude of this noise is a function of the resolution of the converter, and manifests itself as a limit to the theoretical signal-to-noise ratio achievable. This limit is described by S/(N + D) = (6.02n + 1.76 + 10 log FS/2FA) dB, where n is the resolution of the converter in bits, AAA AAA AAA AAA AAA AAA AAA AAAAAAAAAAAA 8000 NUMBER OF CODE HITS AC PERFORMANCE 6000 5000 4000 3000 2000 1000 1267 1081 3 0 –2 –1 0 1 DEVIATION FROM CORRECT CODE – LSBs Figure 9. Distribution of Codes from 10,000 Conversions Relative to the Correct Code –12– REV. A AD677 The standard deviation of this distribution is approximately 0.5 LSBs. If less uncertainty is desired, averaging multiple conversions will narrow this distribution by the inverse of the square root of the number of samples; i.e., the average of 4 conversions would have a standard deviation of 0.25 LSBs. 105 100 90 80 THD 70 dB DSP INTERFACE Figure 10 illustrates the use of the Analog Devices ADSP-2101 digital signal processor with the AD677. The ADSP-2101 FO (flag out) pin of Serial Port 1 (SPORT 1) is connected to the SAMPLE line and is used to control acquisition of data. The ADSP-2101 timer is used to provide precise timing of the FO pin. 60 S/(N+D) 50 40 30 20 10 –80 ADSP-2101 FO SAMPLE RFS0 BUSY AMPLITUDE – dB The AD677 BUSY signal is connected to RF0 to notify SPORT0 when a new data word is coming. SPORT0 should be configured in normal, external, noninverting framing mode and can be programmed to generate an interrupt after the last data bit is received. To maximize the conversion rate, SAMPLE should be brought HIGH immediately after the last data bit is received. dB 82 3.5 4.5 5.5 6.5 7.5 8.5 9.5 10.0 VREF – Volts Figure 11. S/(N+D) and THD vs. VREF, fS = 100 kHz (Calibration is not guaranteed below +5 VREF) REV. A –40 –60 –80 –100 –140 0 5 10 15 20 25 30 FREQUENCY – kHz 35 40 45 50 Figure 13. 4096 Point FFT at 100 kSPS, fIN = 1 kHz, VREF = 5 V 0 –20 –40 –60 –80 –100 0 5 10 15 20 25 30 FREQUENCY – kHz 35 40 45 Figure 14. 4096 Point FFT at 100 kSPS, fIN = 1 kHz, VREF = 10 V S/(N+D) 2.5 0 –20 –140 THD 86 –10 –120 AA AA AA 90 –20 –120 AMPLITUDE – dB The SCLK pin of the ADSP-2101 SPORT0 provides the CLK input for the AD677. The clock should be programmed to be approximately 2 MHz to comply with AD677 specifications. To minimize digital feedthrough, the clock should be disabled (by setting Bit 14 in SPORT0 control register to 0) during data acquisition. Since the clock floats when disabled, a pulldown resistor of 12 kΩ–15 kΩ should be connected to SCLK to ensure it will be LOW at the falling edge of SAMPLE. To maximize the conversion rate, the serial clock should be enabled immediately after SAMPLE is brought LOW (hold mode). 94 –30 0 DT0 Figure 10. ADSP-2101 Interface 98 –40 Figure 12. S/(N+D) and THD vs. Input Amplitude, fS = 100 kHz TFS0 102 –50 SDATA DR0 106 –60 INPUT LEVEL – dB CLK SCLK0 SERIAL PORT 0 –70 AD677 –13– 48 AD677 106 +5V THD, 5V 104 90 THD, 10V 102 80 +12V 100 98 –12V 60 96 dB S/(N+D) – dB 70 S/(N+D), 10V 94 50 92 40 S/(N+D), 5V 90 30 88 86 20 0 100 1k 10k RIPPLE FREQUENCY – Hz 100k 1M –40 Figure 15. AC Power Supply Rejection (fIN = 1.06 kHz) fSAMPLE = 96 kSPS, VRIPPLE = 0.13 V p-p AMPLITUDE – dB AMPLITUDE – dB –30 –50 –70 –90 –110 –130 0 5 10 15 20 25 30 FREQUENCY – kHz 35 40 45 –0.2 –0.4 –0.6 –0.8 –1.0 0 32000 NUMBER OF CODES WITH EACH DNL 102 100 98 dB 5 10 THD, 10V S/(N+D), 10V 92 90 S/(N+D), 5V 26000 22000 18000 12000 8000 510 530 CLK PERIOD – ns 550 570 590 Figure 17. AC Performance vs. Clock Period, TA = +85°C (5 V and 10 V Reference) 45 50 55 60 65 30671 14113 2993 0 490 25 30 35 40 FREQUENCY – kHz AA AA AA AAA AA AA AAA AA A AA AA AA AA A AA AA AA AA AA AAAAAA AAAAAAAAA 2500 2 86 470 20 14645 14000 4000 88 450 15 Figure 19. DNL Plot at VREF = 10 V, TA = +25°C, fS = 100 kSPS THD, 5V 94 80 0.2 0.0 106 96 60 1.0 0.8 0.6 0.4 48 Figure 16. IMD Plot for fIN = 1008 Hz (fa), 1055 Hz (fb) at 96 kSPS 104 0 20 40 TEMPERATURE – Degree °C Figure 18. AC Performance Using Minimum Clock Period vs. Temperature (tCLK = 480 ns), 5 V and 10 V Reference 0 –150 –20 –.35 392 152 –.25 –.15 –.05 0 .05 .15 DNL – LSBs .25 .35 60 6 .40 Figure 20. DNL Error Distribution (Taken from Figure 19) –14– REV. A AD677 OUTLINE DIMENSIONS Dimensions shown in inchcs and (mm) D-16 16-Lead Side Brazed Ceramic DIP Package 0.005 (0.13) MIN 0.080 (2.03) MAX 9 16 0.310 (7.87) 0.220 (5.59) PIN 1 1 8 0.840 (21.34) MAX 0.060 (1.52) 0.015 (0.38) 0.200 (5.08) MAX 0.150 (3.81) MIN 0.200 (5.08) 0.125 (3.18) SEATING PLANE 0.023 (0.58) 0.110 (2.79) 0.070 (1.78) 0.014 (0.36) 0.090 (2.29) 0.030 (0.76) 0.015 (0.38) 0.008 (0.20) 0.320 (8.13) 0.290 (7.37) N-16 16-Lead Plastic DIP 9 16 0.280 (7.11) 0.240 (6.10) PIN 1 1 8 0.840 (21.33) 0.745 (18.93) 0.060 (1.52) 0.015 (0.38) 0.210 (5.33) MAX 0.150 (3.81) MIN 0.200 (5.05) 0.125 (3.18) 0.022 (0.558) 0.014 (0.356) 0.100 (2.54) BSC 0.070 (1.77) 0.045 (1.15) 0.325 (8.25) 0.300 (7.62) 0.195 (4.95) 0.115 (2.93) 0.015 (0.381) 0.008 (0.204) SEATING PLANE R-28 28-Lead Wide Body SOIC (SOIC-28) 0.7125 (18.10) 0.6969 (17.70) 28 15 0.2992 (7.60) 0.2914 (7.40) PIN 1 0.4193 (10.65) 0.3937 (10.00) 14 1 0.1043 (2.65) 0.0926 (2.35) 0.0500 (1.27) BSC 0.0291 (0.74) 0.0098 (0.25) X 45° 0°- 8° 0.0118 (0.30) 0.0040 (0.10) REV. A 0.0192 (0.49) 0.0138 (0.35) 0.0125 (0.32) 0.0091 (0.23) –15– 0.0500 (1.27) 0.0157 (0.40) –16– PRINTED IN U.S.A. C1786–18–4/93