AD8330ARQZ

Low Cost, DC to 150 MHz, Variable Gain Amplifier AD8330 Data Sheet FUNCTIONAL BLOCK DIAGRAM Fully differential signal path, also used with single-sided signals Inputs from 0.3 mV to 1 V rms, rail-to-rail outputs Differential RIN = 1 kΩ; ROUT (each output) 75 Ω Automatic offset compensation (optional) Linear-in-dB and linear-in-magnitude gain modes 0 dB to 50 dB, for 0 V < VDBS < 1.5 V (30 mV/dB) Inverted gain mode: 50 dB to 0 dB at −30 mV/dB ×0.03 to ×10 nominal gain for 15 mV < VMAG < 5 V Constant bandwidth: 150 MHz at all gains Low noise: 5 nV/√Hz typical at maximum gain Low distortion: ≤−62 dBc typical Low power: 20 mA typical at VS of 2.7 V to 6 V Available in a space-saving, 3 mm × 3 mm LFCSP package ENBL OFST CNTR BIAS AND VREF CM AND OFFSET CONTROL VGA CORE OUTPUT STAGES INHI OPLO INLO MODE OPHI GAIN INTERFACE OUTPUT CONTROL VDBS CMGN COMM VMAG CMOP 03217-101 FEATURES Figure 1. APPLICATIONS Pre-ADC signal conditioning 75 Ω cable driving adjust AGC amplifiers GENERAL DESCRIPTION The AD8330 is a wideband variable gain amplifier for applications requiring a fully differential signal path, low noise, well-defined gain, and moderately low distortion, from dc to 150 MHz. The input pins can also be driven from a single-ended source. The peak differential input is ±2 V, allowing sine wave operation at 1 V rms with generous headroom. The output pins can drive single-sided loads essentially rail-to-rail. The differential output resistance is 150 Ω. The output swing is a linear function of the voltage applied to the VMAG pin that internally defaults to 0.5 V, providing a peak output of ±2 V. This can be raised to 10 V p-p, limited by the supply voltage. The basic gain function is linear-in-dB, controlled by the voltage applied to Pin VDBS. The gain ranges from 0 dB to 50 dB for control voltages between 0 V and 1.5 V—a slope of 30 mV/dB. The gain linearity is typically within ±0.1 dB. By changing the logic level on Pin MODE, the gain decreases over the same range, with an opposite slope. A second gain control port is provided at the VMAG pin and allows the user to vary the numeric gain from a factor of 0.03 to 10. All the parameters of the AD8330 have low sensitivities to temperature and supply voltages. Using VMAG, the basic 0 dB to 50 dB range can be repositioned to any value from 20 dB higher (that is, 20 dB to 70 dB) to at least Rev. H 30 dB lower (that is, –30 dB to +20 dB) to suit the application, thereby providing an unprecedented gain range of over 100 dB. A unique aspect of the AD8330 is that its bandwidth and pulse response are essentially constant for all gains, over both the basic 50 dB linear-in-dB range, but also when using the linearin-magnitude function. The exceptional stability of the HF response over the gain range is of particular value in those VGA applications where it is essential to maintain accurate gain lawconformance at high frequencies. An external capacitor at Pin OFST sets the high-pass corner of an offset reduction loop, whose frequency can be as low as 5 Hz. When this pin is grounded, the signal path becomes dc-coupled. When used to drive an ADC, an external common-mode control voltage at Pin CNTR can be driven to within 0.5 V of either ground or VS to accommodate a wide variety of requirements. By default, the two outputs are positioned at the midpoint of the supply, VS/2. Other features, such as two levels of power-down (fully off and a hibernate mode), further extend the practical value of this exceptionally versatile VGA. The AD8330 is available in 16-lead LFCSP and 16-lead QSOP packages and is specified for operation from −40°C to +85°C. Document Feedback Information furnished by Analog Devices is believed to be accurate and reliable. However, no responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other rights of third parties that may result from its use. Specifications subject to change without notice. No license is granted by implication or otherwise under any patent or patent rights of Analog Devices. Trademarks and registered trademarks are the property of their respective owners. One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A. Tel: 781.329.4700 ©2002–2016 Analog Devices, Inc. All rights reserved. Technical Support www.analog.com AD8330 Data Sheet TABLE OF CONTENTS Features .............................................................................................. 1  Applications Information .............................................................. 26  Applications ....................................................................................... 1  ADC Driving............................................................................... 26  Functional Block Diagram .............................................................. 1  Simple AGC Amplifier .............................................................. 26  General Description ......................................................................... 1  Wide Range True RMS Voltmeter............................................ 27  Revision History ............................................................................... 2  Evaluation Board ............................................................................ 29  Specifications..................................................................................... 3  General Description ................................................................... 29  Absolute Maximum Ratings............................................................ 5  Basic Operation .......................................................................... 29  ESD Caution .................................................................................. 5  Options ........................................................................................ 30  Pin Configurations and Function Descriptions ........................... 6  Measurement Setup.................................................................... 30  Typical Performance Characteristics ............................................. 8  AD8330-EVALZ Board Design................................................. 30  Theory of Operation ...................................................................... 15  Outline Dimensions ....................................................................... 32  Circuit Description..................................................................... 15  Ordering Guide .......................................................................... 32  Using the AD8330 ...................................................................... 21  REVISION HISTORY 5/2016—Rev. G to Rev. H Changes to Figure 2 and Table 3 ..................................................... 6 Moved Figure 3 ................................................................................. 7 Changes to Table 4 ............................................................................ 7 Change to Figure 45 ....................................................................... 15 Changes to Simple AGC Amplifier Section ................................ 27 Updated Outline Dimensions ....................................................... 32 Changes to Ordering Guide .......................................................... 32 5/2014—Rev. F to Rev. G Changes to Table 1 ............................................................................ 3 11/2012—Rev. E to Rev. F Changes to Figure 1 .......................................................................... 1 Changes to Output (Input) Common-Mode Control ............... 20 Updated Outline Dimensions ....................................................... 31 Changes to Ordering Guide .......................................................... 31 3/2010—Rev. D to Rev. E Changes to Figure 2 and Table 3 ..................................................... 6 Changes to Figure 69 ...................................................................... 28 Changes to Figure 71 ...................................................................... 29 Changes to Figure 72 ...................................................................... 30 Deleted Table 7; Renumbered Sequentially ................................ 31 Changes to Ordering Guide .......................................................... 32 1/2008—Rev. C to Rev. D Changes to Figure 28 and Figure 29............................................. 12 Added Evaluation Board Section ................................................. 28 Changes to Ordering Guide .......................................................... 33 6/2006—Rev. B to Rev. C Updated Format .................................................................. Universal Changes to Figure 1 ...........................................................................1 Deleted Figure 2; Renumbered Sequentially .................................1 Changes to Specifications Section ...................................................3 Change to Absolute Maximum Ratings .........................................5 Changes to Typical Performance Characteristics Summary Statement ............................................................................................7 Changes to Figure 14 and Figure 15................................................8 Changes to Figure 31 and Figure 32............................................. 11 Updated Outline Dimensions ....................................................... 28 10/2004—Rev. A to Rev. B Changes to Absolute Maximum Ratings Section and Ordering Guide Section .....................................................................................4 Change to TPC 14 .............................................................................8 Note Added to CP-16 Package ...................................................... 26 4/2003—Rev. 0 to Rev. A Updated Outline Dimensions ....................................................... 26 10/2002—Revision 0: Initial Version Rev. H | Page 2 of 32 Data Sheet AD8330 SPECIFICATIONS VS = 5 V, TA = 25°C, CL = 12 pF on OPHI and OPLO, RL = ∞, VDBS = 0.75 V, VMODE = high, VMAG = Pin VMAG open circuit (0.5 V), VOFST = 0 V, differential operation, unless otherwise noted. Table 1. Parameter INPUT INTERFACE Full-Scale Input Input Resistance Input Capacitance Voltage Noise Spectral Density Common-Mode Voltage Level Input Offset Drift Permissible CM Range 1 Common-Mode AC Rejection OUTPUT INTERFACE Small Signal –3 dB Bandwidth Peak Slew Rate Peak-to-Peak Output Swing Common-Mode Voltage Offset Voltage Offset Correction Enabled Offset Correction Disabled Voltage Noise Spectral Density Differential Output Impedance HD2 2 HD32 OUTPUT OFFSET CONTROL AC-Coupled Offset High-Pass Corner Frequency COMMON-MODE CONTROL Usable Voltage Range Input Resistance DECIBEL GAIN CONTROL Normal Voltage Range Elevated Range Gain Scaling Gain Linearity Error Absolute Gain Error Bias Current Incremental Resistance Gain Settling Time to 0.5 dB Error Mode Up/Down Mode Up Logic Level Mode Down Logic Level Test Conditions/Comments Pin INHI, Pin INLO VDBS = 0 V, differential drive VDBS = 1.5 V Pin-to-pin Either pin to COMM f = 1 MHz, VDBS = 1.5 V; inputs ac-shorted Min Typ ±1.4 ±4.5 800 ±2 ±6.3 1k 4 5 3.0 1 2 2.75 Pin OFST connected to Pin COMM VMAG ≥ 2 V (peaks are supply limited) Pin CNTR O/C Pin OFST connected to ground f = 1 MHz, VDBS = 0 V Pin-to-pin VOUT = 1 V p-p, f = 10 MHz, RL = 1 kΩ VOUT = 1 V p-p, f = 10 MHz, RL = 1 kΩ Pin OFST CHPF on Pin OFST (0 V < VDBS < 1.5 V) CHPF = 3.3 nF, from OFST to CNTR (scales as 1/CHPF) Pin CNTR ±1.8 ±4 2.4 120 −60 −55 150 1500 ±2 ±4.5 2.5 MHz V/µs V V V VDBS stepped from 0.05 V to 1.45 V or 1.45 V to 0.05 V Pin MODE Gain increases with VDBS, MODE = O/C Gain decreases with VDBS Rev. H | Page 3 of 32 1.2 k 3.25 VS 1 8 62 150 −62 −53 ±2.2 2.6 ±5 ±40 180 10 100 0.5 From Pin CNTR to VS/2 VDBS, CMGN, and MODE pins CMGN connected to COMM CMGN O/C (VCMGN rises to 0.2 V) Mode high or low 0.3 V ≤ VDBS ≤ 1.2 V VDBS = 0 V Flows out of Pin VDBS 27 −0.35 −2 Unit V mV Ω pF nV/√Hz V mV rms µV/°C V dB dB 0 f = 1 MHz, 0.1 V rms f = 50 MHz Pin OPHI, Pin OPLO 0 V < VDBS < 1.5 V VDBS = 0 V Max mV mV nV/√Hz Ω dBc dBc mV rms kHz 4 4.5 V kΩ 0 to 1.5 0.2 to 1.7 30 ±0.1 ±0.5 100 100 250 V V mV/dB dB dB nA MΩ ns 33 +0.35 +2 1.5 0.5 V V AD8330 Parameter LINEAR GAIN INTERFACE Peak Output Scaling, Gain vs. VMAG Gain Multiplication Factor vs. VMAG Usable Input Range Default Voltage Incremental Resistance Bandwidth CHIP ENABLE Logic Voltage for Full Shutdown Logic Voltage for Hibernate Mode Logic Voltage for Full Operation Current in Full Shutdown Current in Hibernate Mode Minimum Time Delay 3 POWER SUPPLY Supply Voltage Quiescent Current Data Sheet Test Conditions/Comments Pin VMAG, Pin CMGN See the Circuit Description section Gain is nominal when VMAG = 0.5 V Min Typ Max Unit 3.8 4.0 ×2 4.2 V/V 5 0.52 V V kΩ MHz 0.5 1.7 V V V µA mA µs 0 0.48 VMAG O/C For VMAG ≥ 0.1 V Pin ENBL Output pins remain at CNTR 1.3 2.3 0.5 4 150 1.5 20 1.5 1.7 100 VPSI, VPOS, VPSO, COMM, and CMOP pins 2.7 VDBS = 0.75 V 20 6 27 V mA The use of an input common-mode voltage significantly different from the internally set value is not recommended due to its effect on noise performance. See Figure 56. See the Typical Performance Characteristics section for more detailed information on distortion in a variety of operating conditions. 3 For minimum sized coupling capacitors. 1 2 Rev. H | Page 4 of 32 Data Sheet AD8330 ABSOLUTE MAXIMUM RATINGS Table 2. Parameter Supply Voltage Power Dissipation 16-Lead QSOP Package1 16-Lead LFCSP Package Input Voltage at Any Pin Storage Temperature Range θJA 16-Lead QSOP Package 16-Lead LFCSP Package θJC 16-Lead QSOP Package Operating Temperature Range Lead Temperature (Soldering 60 sec) 1 Rating 6V 0.62 W 1.67 W VS + 200 mV −65°C to +150°C Stresses at or above those listed under Absolute Maximum Ratings may cause permanent damage to the product. This is a stress rating only; functional operation of the product at these or any other conditions above those indicated in the operational section of this specification is not implied. Operation beyond the maximum operating conditions for extended periods may affect product reliability. ESD CAUTION 105.4°C/W 60°C/W 39°C/W −40°C to +85°C 300°C 4-layer JEDEC Board (252P). Rev. H | Page 5 of 32 AD8330 Data Sheet 13 CNTR 14 VPOS 16 ENBL 15 OFST PIN CONFIGURATIONS AND FUNCTION DESCRIPTIONS VPSI 1 INLO 3 12 VPSO AD8330 TOP VIEW (Not to Scale) 11 OPHI 10 OPLO 9 CMOP VMAG 8 COMM 7 VDBS 5 CMGN 6 MODE 4 NOTES 1. THE EXPOSED PAD IS NOT CONNECTED INTERNALLY. FOR INCREASED RELIABILITY OF THE SOLDER JOINTS AND MAXIMUM THERMAL CAPABILITY, IT IS RECOMMENDED THAT THE PAD BE SOLDERED TO THE GROUND PLANE. 03217-003 INHI 2 Figure 2. 16-Lead LFCSP Pin Configuration Table 3. 16-Lead LFCSP Pin Function Descriptions Pin No. 1 2 3 4 5 6 7 8 9 10 11 12 13 14 15 Mnemonic VPSI INHI INLO MODE VDBS CMGN COMM VMAG CMOP OPLO OPHI VPSO CNTR VPOS OFST 16 ENBL EPAD Description Positive Supply for Input Stages. Differential Signal Input, Positive Polarity. Differential Signal Input, Negative Polarity. Logic Input: Selects Gain Slope. High = gain up vs. VDBS. Input for Linear-in-dB Gain Control Voltage, VDBS. Common Baseline for Gain Control Interfaces. Ground for Input and Gain Control Bias Circuitry. Input for Gain/Amplitude Control, VMAG. Ground for Output Stages. Differential Signal Output, Negative Polarity. Differential Signal Output, Positive Polarity. Positive Supply for Output Stages. Common-Mode Output Voltage Control. Positive Supply for Inner Stages. Internal Offset Compensation Feature. When the OFST pin is unconnected, this feature is enabled. When the OFST pin is grounded, this feature is disabled. See the Offset Compensation section. Power Enable, Active High. Exposed Pad. The exposed pad is not connected internally. For increased reliability of the solder joints and maximum thermal capability, it is recommended that the pad be soldered to the ground plane. Rev. H | Page 6 of 32 Data Sheet AD8330 O FS T 1 16 VPOS E N BL 2 15 CN T R 14 VPSO VPSI 3 AD8330 TOP VIEW 13 OPHI (Not to Scale) 12 OPLO I N LO 5 MODE 6 11 CMOP VDBS 7 10 VMAG CMGN 8 9 COMM 03217-004 INHI 4 Figure 3. 16-Lead QSOP Pin Configuration Table 4. 16-Lead QSOP Pin Function Descriptions Pin No. 1 Mnemonic OFST 2 3 4 5 6 7 8 9 10 11 12 13 14 15 16 ENBL VPSI INHI INLO MODE VDBS CMGN COMM VMAG CMOP OPLO OPHI VPSO CNTR VPOS Description Internal Offset Compensation Feature. When the OFST pin is unconnected, this feature is enabled. When the OFST pin is grounded, this feature is disabled. See the Offset Compensation section. Power Enable, Active High. Positive Supply for Input Stages. Differential Signal Input, Positive Polarity. Differential Signal Input, Negative Polarity. Logic Input: Selects Gain Slope. High = gain up vs. VDBS. Input for linear-in-dB Gain Control Voltage, VDBS. Common Baseline for Gain Control Interfaces. Ground for Input and Gain Control Bias Circuitry. Input for Gain/Amplitude Control, VMAG. Ground for Output Stages. Differential Signal Output, Negative Polarity. Differential Signal Output, Positive Polarity. Positive Supply for Output Stages. Common-Mode Output Voltage Control. Positive Supply for Inner Stages. Rev. H | Page 7 of 32 AD8330 Data Sheet TYPICAL PERFORMANCE CHARACTERISTICS VS = 5 V, TA = 25°C, CL = 12 pF, VDBS = 0.75 V, VMODE = high (or O/C) VMAG = O/C (0.5 V), RL = ∞, VOFST = 0, differential operation, unless otherwise noted. 2.0 50 45 LO MODE 40 HI MODE 1.0 GAIN ERROR (dB) 35 30 25 20 0.5 0 –0.5 0 0.25 0.75 VDBS (V) 0.50 1.00 1.25 1.50 –2.0 03217-005 0 0 0.2 Figure 4. Gain vs. VDBS 20 9 0.6 0.8 1.0 VDBS (V) 1.2 1.4 1.6 2340 UNITS MODE = LO 15 8 10 % OF UNITS 6 5 4 3 5 0 –30.6 –30.5 –30.4 –30.3 –30.2 –30.1 –30.0 –29.9 –29.8 –29.7 –29.6 –29.5 –29.4 –29.3 –29.2 –29.1 –29.0 20 2 10 1 5 0 2 1 3 4 5 VMAG (V) MODE = HI 15 0 29.1 29.2 29.3 29.4 29.5 29.6 29.7 29.8 29.9 30.0 30.1 30.2 30.3 30.4 30.5 30.6 GAIN SCALING (mV/dB) Figure 5. Linear Gain Multiplication Factor vs. VMAG Figure 8. Gain Slope Histogram 1.0 60 0.8 50 1.2V 0.9V 30 0.4 0.6V GAIN (dB) 20 T = –40°C 0 –0.2 0V 0 –20 T = +25°C –0.6 0.3V 10 –10 T = +85°C –0.4 –30 –0.8 –40 0 0.2 0.4 0.6 0.8 1.0 1.2 1.4 1.6 VDBS (V) Figure 6. Gain Linearity Error Normalized at 25°C vs. VDBS, at Three Temperatures, f = 1 MHz –50 100k 03217-007 –1.0 VDBS = 1.5V 40 0.6 0.2 03217-009 7 03217-006 GAIN MULTIPLICATION FACTOR 0.4 Figure 7. Gain Error vs. VDBS at Various Frequencies 10 GAIN ERROR (dB) 10MHz 100MHz –1.5 5 1MHz 1MHz –1.0 10 03217-008 15 0 50MHz 100MHz 10MHz, 50MHz 1M 10M FREQUENCY (Hz) 100M 500M 03217-010 GAIN (dB) NORMALIZED @ VDBS = 0.75V 1.5 Figure 9. Frequency Response in 10 dB Steps for Various Values of VDBS Rev. H | Page 8 of 32 Data Sheet 50 AD8330 40 1048 UNITS ENABLE MODE 1.52V 30 20 0.48V 20 0.15V 10 0.048V 0 0.015V % OF UNITS GAIN (dB) 25 VMAG = 4.8V 15 10 –10 –20 5 1.0 03217-014 0.9 0.8 0.7 0.6 0.5 0.4 0.3 0.2 0 0.1 –0.1 –0.2 –0.3 Figure 13. Differential Input Offset Histogram 10 VDBS = 0.1V 0 OUTPUT BALANCE ERROR (dB) 8 GROUP DELAY (ns) –0.4 DIFFERENTIAL OFFSET (mV) Figure 10. Frequency Response for Various Values of VMAG, VDBS = 0.75 V 10 –0.5 0 –0.6 500M –0.7 100M 10M FREQUENCY (Hz) –0.8 1M 03217-011 –40 100k –0.9 –30 6 4 2 –10 –20 –30 –40 –50 –60 –70 10M FREQUENCY (Hz) 100M 300M –90 100k 03217-012 1M Figure 11. Group Delay vs. Frequency 1M 10M FREQUENCY (Hz) 100M 03217-015 –80 0 100k Figure 14. Output Balance Error vs. Frequency for a Representative Part 0 200 190 –1 OUTPUT IMPEDANCE (Ω) T = –40°C –3 –4 T = +25°C –5 160 150 140 130 120 –6 T = +85°C 0 0.2 0.4 0.6 0.8 1.0 VDBS (V) 1.2 1.4 110 1.6 03217-013 –7 170 100 100k Figure 12. Differential Output Offset vs. VDBS for Three Temperatures, for a Representative Part 1M 10M FREQUENCY (Hz) 100M Figure 15. Output Impedance vs. Frequency Rev. H | Page 9 of 32 300M 03217-016 OFFSET VOLTAGE (mV) 180 –2 AD8330 90 Data Sheet 6000 VDBS = 1.5V 80 VDBS = 0.75V 70 5000 NOISE (nV/√Hz) 60 CMRR (dB) 50 VDBS = 0V 40 VDBS = 1.5V f = 1MHz OFST: ENABLED DISABLED 30 20 10 4000 3000 2000 1000 1M 10M FREQUENCY (Hz) 0 03217-017 –10 50k 100k 100M 80 2.5 2.0 VMAG = 0.5V f = 1MHz 70 T = +25°C 1200 1.5 Figure 19. Output Referred Noise vs. VMAG T = +85°C f = 1MHz VMAG = 0.5V 1 .0 VMAG (V) Figure 16. CMRR vs. Frequency 1500 0.5 0 03217-020 0 T = +85°C NOISE (nV/√Hz) NOISE (nV/√Hz) 60 900 T = –40°C 600 50 40 T = +25°C 30 T = –40°C 20 300 0 0.2 0.4 0.6 0.8 1.0 VDBS (V) 1.2 1.4 0 03217-018 0 1.6 0.2 0.4 0.6 0.8 1.0 1.2 1.4 1.6 VDBS (V) Figure 17. Output Referred Noise vs. VDBS for Three Temperatures Figure 20. Input Referred Noise vs. VDBS for Three Temperatures 700 180 f = 1MHz f = 1MHz 160 600 140 NOISE (nV/√Hz) 500 400 300 200 VMAG = 0.125V 120 100 VMAG = 0.5V 80 60 40 0 20 0 0.5 1.0 1.5 2.0 VMAG (V) 2.5 0 VMAG = 2V 0 0.2 0.4 0.6 0.8 1.0 1.2 1.4 1.6 VDBS (V) Figure 18. Output Referred Noise vs. VMAG, VDBS = 0.75 V Figure 21. Input Referred Noise vs. VDBS for Three Values of VMAG Rev. H | Page 10 of 32 03217-022 100 03217-019 NOISE (nV/√Hz) 0 03217-021 10 Data Sheet 7 AD8330 0 VDBS = 1.5V –20 DISTORTION (dBc) 5 4 3 2 –30 –40 HD3, RL = 1kΩ –50 –60 HD2, RL = 1kΩ –70 1M 100M 10M FREQUENCY (Hz) –80 03217-023 0 100k 0 Figure 22. Input Referred Noise vs. Frequency f = 10MHz –10 –20 –20 DISTORTION (dBc) –30 –40 HD3 –50 HD2 –60 HD2 AND HD3, RL = 150Ω1 –30 –40 HD3, RL = 1kΩ –50 –60 –70 HD2, RL = 1kΩ –70 1M 10M 100M FREQUENCY (Hz) –80 03217-024 –80 100k 1OUTPUT Figure 23. Harmonic Distortion vs. Frequency 0 4 2 3 VOUT (V p-p) 5 Figure 26. Harmonic Distortion vs. VOUT , VMAG = 2.0 V 0 VDBS = 0.75V VOUT = 1V p-p RL = 1kΩ –10 AMPLITUDE HARD LIMITED 1 0 03217-027 DISTORTION (dBc) 1.5 1.2 0 VDBS = 0.75V VOUT = 1V p-p –10 RL = 1kΩ f = 10MHz VOUT = 1V p-p –10 RL = 1kΩ –20 DISTORTION (dBc) –20 –30 –40 HD3 –50 –60 HD2 –70 0 10 20 30 CLOAD (pF) –30 HD3 –40 –50 HD2 –60 40 50 03217-025 DISTORTION (dBc) 0.6 0.9 VOUT (V p-p) Figure 25. Harmonic Distortion vs. VOUT , VMAG = 0.5 V 0 –80 0.3 03217-026 1 Figure 24. Harmonic Distortion vs. CLOAD –70 0 0.2 0.4 0.6 0.8 1.0 VDBS (V) 1.2 Figure 27. Harmonic Distortion vs. VDBS Rev. H | Page 11 of 32 1.4 1.6 03217-028 NOISE (nV/√Hz) f = 10MHz –10 6 AD8330 Data Sheet 33 30 23 25 28 –10 3 20 23 –20 –7 –30 –17 –40 0.4 0.6 0.8 1.0 1.2 1.4 OIP3 (dBV rms) 10 13 –27 5 8 –37 1.6 0 VDBS (V) 0 0.4 0.2 Figure 28. Input V1dB Compression vs. VDBS 33 10 23 0 13 35 3 –20 –7 –30 –17 3 VMAG (V) 4 5 6 –27 3 33 28 20 23 f = 50MHz 15 18 10 13 5 8 0 0.2 Figure 29. Output V1dB Compression vs. VMAG 0.4 0.6 0.8 VMAG (V) 1.0 1.2 1.4 3 1.6 Figure 32. OIP3 vs. VMAG 0 1.5 VDBS = 0.75V –10 VOUT = 1V p-p 1.0 –20 –30 0.5 VOUT (V) –40 –50 –60 VDBS = 0V 0 –0.5 –70 –90 1M 10M FREQUENCY (Hz) 100M Figure 30. IMD3 Distortion vs. Frequency –1.5 –50 –25 0 25 TIME (ns) 50 75 Figure 33. Full-Scale Transient Response, VDBS = 0 V Rev. H | Page 12 of 32 100 03217-034 –1.0 –80 03217-031 IMD3 (dBc) 1.6 38 f = 10MHz 25 0 03217-030 2 OIP3 (dBV rms) P1dB (REF 50Ω) –10 1 1.4 43 30 0 1.2 40 f = 10MHz –40 0.8 1.0 VDBS (V) Figure 31. OIP3 vs. VDBS 20 INPUT V1dB COMPRESSION (dBV rms) 0.6 03217-032 0.2 18 OIP3 (dBm) 0 f = 50MHz 15 03217-033 –50 P1dB (REF 50Ω) 13 0 OIP3 (dBm) f = 10MHz f = 10MHz 03217-029 INPUT V1dB COMPRESSION (dBV rms) 10 Data Sheet AD8330 1.5 1V 1.0 VOUT (V) 0.5 VDBS = 0.75V 0 –0.5 –25 0 25 TIME (ns) 50 75 100 1V 400ns 03217-038 –1.5 –50 03217-035 –1.0 Figure 37. VDBS Interface Response, Top: VDBS, Bottom: VOUT Figure 34. Full-Scale Transient Response, VDBS = 0.75 V, f = 1 MHz, VOUT = 2 V p-p 1.5 2V 1.0 VOUT (V) 0.5 VDBS = 1.5V 0 –1.0 –25 0 25 TIME (ns) 50 75 100 400ns 03217-036 1mV –1.5 –50 03217-039 –0.5 Figure 38. VMAG Interface Response, Top: VMAG, Bottom: VOUT Figure 35. Full-Scale Transient Response, VDBS = 1.5 V, f = 1 MHz, VOUT = 2 V p-p 1V 500mV CL = 12pF VMAG = 5V VMAG = 0.5V CL = 54pF CL = 24pF 100mV Figure 36. Transient Response vs. Various Load Capacitances, G = 25 dB Rev. H | Page 13 of 32 12.5ns Figure 39. Transient Response vs. VMAG 03217-040 12.5ns 03217-037 VMAG = 0.05V AD8330 Data Sheet 26 4.00V OUTPUT INPUT 25ns +85°C 20 +25°C 18 –40°C 16 03217-041 50mV 22 14 Figure 40. Overdrive Response, VDBS = 1.5 V, VMAG = 0.5 V, 18.5 dB Overdrive 0 0.2 0.4 0.6 0.8 1.0 VDBS (V) 1.2 1.4 1.6 Figure 43. Supply Current vs. VDBS at Three Temperatures 2V 3.125V 2.5V 1.875V 3.125V 400ns 1.875V 03217-042 1V 100ns Figure 41. ENBL Interface Response. Top: VENBL; Bottom: VOUT, f = 10 MHz –10 –20 VDBS = 0.75V –30 PSRR (dB) –40 VPSI –50 –60 VPOS –70 VPSO –80 –90 –110 1M 10M FREQUENCY (Hz) 100M 200M 03217-043 –100 Figure 42. PSRR vs. Frequency Rev. H | Page 14 of 32 Figure 44. CNTR Transient Response, Top: Input to CNTR, Bottom: VOUT Single-Ended 03217-045 2.5V 03217-044 SUPPLY CURRENT (mA) 24 Data Sheet AD8330 INPUT IS xlD CIRCUIT DESCRIPTION Many monolithic variable gain amplifiers use techniques that share common principles that are broadly classified as translinear. This term refers to circuit cells whose functions depend directly on the very predictable properties of bipolar junction transistors, notably the linear dependence of their transconductance on collector current. Since the discovery of these cells in 1967, and their commercial exploitation in products developed during the early 1970s, accurate wide bandwidth analog multipliers, dividers, and variable gain amplifiers have invariably employed translinear principles. By varying IN, the overall function is that of a two-quadrant analog multiplier, exhibiting a linear relationship to both the signal modulation factor (x) and this numerator current. On the other hand, by varying ID, a two-quadrant analog divider is realized, having a hyperbolic gain function with respect to the input factor, x, controlled by this denominator current. The AD8330 exploits both modes of operation. However, because a hyperbolic gain function is generally of less value than one in which the decibel gain is a linear function of a control input, a special interface is included to provide either increasing or decreasing exponential control of ID. (1–x) IN 2 + Q4 Q2 ID (1+x) IN 2 – Q1 Q3 NUMERATOR DENOMINATOR BIAS CURRENT BIAS CURRENT IN Figure 45. Basic Core ENBL OFST VPSI BIAS AND VREF INHI AD8330 VPOS CNTR VPSO CM MODE AND OFFSET CONTROL OPHI OUTPUT STAGES MODE VGA CORE GAIN INTERFACE VDBS CMGN OPLO OUTPUT CONTROL CMOP COMM VMAG 03217-047 INLO Figure 45 shows a basic representative cell comprising just four transistors. This, or a very closely related form, is at the heart of most translinear multipliers, dividers, and VGAs. The key concepts are as follows: In practice, the realization of the full potential of this circuit involves many other factors, but these three elementary ideas remain essential. LOOP AMPLIFIER (1–x) ID 2 (1+x) ID 2 Although these techniques are well understood, the realization of a high performance variable gain amplifier (VGA) requires special technologies and attention to many subtle details in its design. The AD8330 is fabricated on a proprietary silicon-oninsulator, complementary bipolar IC process and draws on decades of experience in developing many leading edge products using translinear principles to provide an unprecedented level of versatility. First, the ratio of the currents in the left-hand and right-hand pairs of transistors is identical, represented by the modulation factor, x, with values between −1 and +1. Second, the input signal is arranged to modulate the fixed tail current, ID, to cause the variable value of x, introduced in the left-hand pair, to be replicated in the right-hand pair, and, thus, generate the output by modulating its nominally fixed tail current, IN. Third, the current gain of this cell is exactly G = IN/ID over many decades of variable bias current. OUTPUT IS xlN G = IN/ID 03217-046 THEORY OF OPERATION Figure 46. Block Schematic Overall Structure Figure 46 shows a block schematic of the AD8330 locating the key sections. More detailed descriptions of its structure and features are provided throughout the Theory of Operation section; however, Figure 46 provides a general overview of its capabilities. The VGA core contains a more elaborate version of the cell shown in Figure 45. The current, ID, is controlled exponentially (linear-in-decibels) through the decibel gain interface at Pin VDBS and its local common, Pin CMGN. The gain span (that is, the decibel difference between maximum and minimum values) provided by this control function is slightly more than 50 dB. The absolute gain from input to output is a function of source and load impedance, and depends on the voltage on a second gain control pin (VMAG), explained in the Normal Operating Conditions section. Rev. H | Page 15 of 32 AD8330 Data Sheet To minimize confusion, normal operating conditions are defined as follows: • • • • • The input pins are voltage driven (the source impedance is assumed to be zero). The output pins are open circuited (the load impedance is assumed to be infinite). Pin VMAG is unconnected setting up the output bias current (IN in the four-transistor gain cell) to its nominal value. Pin CMGN is grounded. MODE is either tied to a logic high or left unconnected, to set the up gain mode. The effects of other operating conditions are considered separately. Throughout this data sheet, the end-to-end voltage gain for the normal operating conditions is referred to as the basic gain. Under these conditions, it runs from 0 dB when VDBS = 0 V (where this voltage is more exactly measured with reference to Pin CMGN, which is not necessarily tied to ground) up to 50 dB for VDBS = 1.5 V. The gain does not fold over when the VDBS pin is driven below ground or above its nominal full-scale value. The input is accepted at the INHI/INLO differential port. These pins are internally biased to roughly the midpoint of the supply, VS (it is actually ~2.75 V for VS = 5 V, VDBS = 0 V, and 1.5 V for VS = 3 V), but the AD8330 is able to accept a forced commonmode value, from zero to VS, with certain limitations. This interface provides good common-mode rejection up to high frequencies (see Figure 16) and, thus, can be driven in either a single-sided or differential manner. However, operation using a differential drive is preferable, and this is assumed in the specifications, unless otherwise stated. The pin-to-pin input resistance is specified as 950 Ω ± 20%. The driving-point impedance of the signal source can range from zero up to values considerably in excess of this resistance, with a corresponding variation in noise figure (see Figure 53). In most cases, the input is coupled via two capacitors, chosen to provide adequate low frequency transmission. This results in the minimum input noise that increases when some other common-mode voltage is forced onto these pins. The short-circuit, input-referred noise at maximum gain is approximately 5 nV/√Hz. Output Pin OPHI and Output Pin OPLO operate at a commonmode voltage at the midpoint of the supply, VS/2, within a few millivolts. This ensures that an analog-to-digital converter (ADC) attached to these outputs operates within the often narrow range permitted by their design. When a common-mode voltage other than VS/2 is required at this interface, it can easily be forced by applying an externally provided voltage to the output centering pin, CNTR. This voltage can run from zero to the full supply, though the use of such extreme values leaves only a small range for the differential output signal swing. The differential impedance measured between OPHI and OPLO is 150 Ω ± 20%. It follows that both the gain and the full-scale voltage swing depend on the load impedance; both are nominally halved when this is also 150 Ω. A fixed impedance output interface, rather than an op amp style voltage-mode output, is preferable in high speed applications because the effects of complex reactive loads on the gain and phase can be better controlled. The top end of the AD8330 ac response is optimally flat for a 12 pF load on each pin, but this is not critical, and the system remains stable for any value of load capacitance including zero. Another useful feature of this VGA in connection with the driving of an ADC is that the peak output magnitude can be precisely controlled by the voltage on Pin VMAG. Usually, this voltage is internally preset to 500 mV, and the peak differential unloaded output swing is ±2 V ± 3%. However, any voltage from zero to at least 5 V can be applied to this pin to alter the peak output in an exactly proportional way. Because either output pin can swing rail-to-rail, which in practice means down to at least 0.35 V and to within the same voltage below the supply, the peak-to-peak output between these pins can be as high as 10 V using VS = 6 V. CM MODE FEEDBACK VPSI VPSO TRANSIMPEDANCE OUTPUT STAGE INHI 500Ω OPHI ΔV = 0 ROUT = 150Ω ΔV = 0 OPLO INLO 500Ω O/P CM-MODE NORMALLY AT VP/2 CNTR LINEAR-IN-dB INTERFACE MAGNITUDE INTERFACE MODE VMAG VDBS VDBS COMM 100µA 12.65µA–4mA OR 4mA–12.65µA VMAG 5kΩ COMM Figure 47. Schematic of Key Components Linear-in-dB Gain Control (VDBS) All Analog Devices, Inc., VGAs featuring a linear-in-dB gain law, such as the X-AMP® family, provide exact, constant gain scaling over the fully specified gain range, and the deviation from the ideal response is within a small fraction of a dB. For the AD8330, the scaling of both of its gain interfaces is substantially independent of process, supply voltage, or temperature. The basic gain, GB, is simply GB (dB ) = VDBS 30 mV where VDBS is in volts. Rev. H | Page 16 of 32 (1) 03217-048 Normal Operating Conditions Data Sheet AD8330 Alternatively, this can be expressed as a numerical gain magnitude provided by VMAG. The latter modifies the basic numerical gain GBN to generate a total gain, expressed here in magnitude terms. VDBS GBN = 10 0.6 V GT = GBN (2) The gain can be increased or decreased by changing the voltage, VMAG, applied to the VMAG pin. The internally set default value of 500 mV is derived from the same band gap reference that determines the decibel scaling. The tolerance on this voltage, and mismatches in certain on-chip resistors, cause small gain errors (see the Specifications section). Though not all applications of VGAs demand accurate gain calibration, it is a valuable asset in many situations, for example, in reducing design tolerances. Figure 47 shows the core circuit in more detail. The range and scaling of VDBS is independent of the supply voltage, and the gain control pin, VDBS, presents a high incremental input resistance (~100 MΩ) with a low bias current (~100 nA), making the AD8330 easy to drive from a variety of gain control sources. VOUT = 2 × GIN × VIN × VMAG GAIN (dB) (4) from which it is apparent that the AD8330 implements a linear, two-quadrant multiplier with a bipolar VIN and a unipolar VMAG. Because the AD8330 is a dc-coupled system, it can be used in many applications where a wideband two-quadrant multiplier function is required, from dc up to about 100 MHz from either input (VIN or VMAG). As VMAG is varied, so also is the peak output magnitude, up to a point where this is limited by the absolute output limit imposed by the supply voltage. In the absence of the latter effect, the peak output into an open-circuited load is just VOUT_PK = ±4 VMAG The AD8330 supports many features that further extend the versatility of this VGA in wide bandwidth gain control systems. For example, the logic pin, MODE, allows the slope of the gain function to be inverted, so that the basic gain starts at +50 dB for a gain voltage, VDBS, of zero and runs down to 0 dB when this voltage is at its maximum specified value of 1.5 V. The basic forms of these two gain control modes are shown in Figure 48. MODE PIN HIGH, GAIN INCREASES WITH VDBS MODE PIN LOW, GAIN DECREASES WITH VDBS 40 (3) Using this to calculate the output voltage, Inversion of the Gain Slope 50 VMAG 0. 5 V 30 20 (5) whereas for a load resistance of RL directly across OPHI and OPLO, it is VOUT _ PK = ±2 VMAG RL (6) (RL + 150 ) These capabilities are illustrated in Figure 49, where VS = 6 V, RL = O/C, VDBS = 0 V, VIN is swept from −2.5 V dc to +2.5 V dc, and VMAG is set to 0.25 V, 0.5 V, 1 V, and 2 V. Except for the last value of VMAG, the peak output follows Equation 5. This exceeds the supply-limited value when VMAG = 2 V and the peak output is ±5.65 V, that is, ±6 V − 0.35 V. Figure 50 demonstrates the high speed multiplication capability. The signal input is a 100 MHz, 0.1 V sine wave, VDBS is set to 0.6 V, and VMAG is a square wave at 5 MHz alternating from 0.25 V to 1 V. The output is ideally a sine wave switching in amplitude between 0.5 V and 2 V. 8 VMAG = 2V 6 1V 0.25 0.50 0.75 1.0 VDBS (V) 1.25 1.50 0.5V 2 VOUT (V) 0 03217-049 4 0 Figure 48. Two Gain Directions of the AD8330 Gain Magnitude Control (VMAG) –2 In addition to the basic linear-in-dB control, two more gain control features are provided. The voltage applied to Pin VMAG provides accurate linear-in-magnitude gain control with a very rapid response. The bandwidth of this interface is >100 MHz. When this pin is unconnected, VMAG assumes its default value of 500 mV (see Figure 47) to set up the basic 0 dB to 50 dB range. However, any voltage from ~15 mV to 5 V can be applied either to lower the gain by up to 30 dB or to raise it by 20 dB. The combined gain span is thus 100 dB, that is, the 50 dB basic gain span provided by VDBS plus a 60 dB linear-in-magnitude span –4 Rev. H | Page 17 of 32 0.25V 0 –6 –8 –3 –2 –1 0 VIN ( V) 1 2 Figure 49. Effect of VMAG on Gain and Peak Output 3 03217-050 10 AD8330 Data Sheet VIN 0.10 Amplitude/Phase Response 0.05 The ac response of the AD8330 is remarkably consistent not only over the full 50 dB of its basic gain range, but also with changes of gain due to alteration of VMAG, as demonstrated in Figure 51. This is an overlay of two sets of results: first, with a very low VMAG of 16 mV that reduces the overall gain by 30 dB [20 × log10(500 mV/16 mV)]; second, with VMAG = 5 V that increases the gain by 20 dB = 20 × log10(5 V/0.5 V). 0 –0.05 –0.10 VMAG 1.2 1.0 0.8 0.6 0.4 0.2 0 90 VOUT –200 –100 0 100 200 TIME (ns) 300 50 30 10 –10 –30 –50 100k 0 Another gain related feature allows both gain control ranges to be accurately raised by 200 mV. To enable this offset, open circuit CMGN (Pin 6, LFCSP; Pin 8, QSOP) and add a 0.1 µF capacitor to ground. In use, the nominal range for VDBS extends from 0.2 V to 1.7 V and VMAG from 0.2 V to 5.2 V. These specifications apply for any supply voltage. This allows the use of DACs whose output range does not include ground as sources for the gain control function(s). Note that the 200 mV that appears on this pin affects the response to an externally applied VMAG, but when Pin VMAG is unconnected, the internally set default value of 0.5 V still applies. Furthermore, Pin CMGN can, if desired, be driven by a usersupplied voltage to reposition the baseline for VDBS (or for an externally applied VMAG) to any other voltage up to 500 mV. In all cases, the gain scaling, its law conformance, and temperature stability are unaffected. Two Classes of Variable Gain Amplifiers Note that there are two broad classes of VGAs. The first type is designed to cope with a very wide range of input amplitudes and, by virtue of its gain control function, compress this range down to an essentially constant output. This is the function needed in an AGC system. Such a VGA is called an IVGA, referring to a structure optimized to address a wide range of input amplitudes. By contrast, an OVGA is optimized to deliver a wide range of output values while operating with an essentially constant input amplitude. This function might be needed, for example, in providing a variable drive to a power amplifier. It is apparent from the foregoing sections that the AD8330 is both an IVGA and an OVGA in one package. This is an unusual and possibly confusing degree of versatility for a VGA; therefore, these two distinct control functions are described at separate points throughout this data sheet to explain the operation and applications of this product. It is, nevertheless, useful to briefly describe the capabilities of these features when used together. PHASE (Degrees) Figure 50. Using VMAG in Modulation Mode 1M 10M 100M 300M G = +70dB –50 –100 –150 –200 –250 –300 –350 10k G = –20dB 100k 1M 10M FREQUENCY (Hz) 100M 300M 03217-052 –300 GAIN (dB) 70 03217-051 2.5 2.0 1.5 1.0 0.5 0 –0.5 –1.0 –1.5 –2.0 –400 Figure 51. AC Performance over a 100 dB Gain Range Obtained by Using Two Values of VMAG This 50 dB step change in gain produces two sets of gain curves, having a total gain span of 100 dB. It is apparent that the amplitude and phase response are essentially independent of the gain over this wide range, an aspect of the AD8330 performance potential unprecedented in any prior VGA. It is unusual for an application to require such a wide range of gains; and, as a practical matter, the peak output voltage for VMAG = 16 mV is reduced by the factor 16/500, compared to its nominal value of ±2 V, to only ±64 mV. As previously noted, most applications of VGAs require that they operate in a mode that is predominantly of either an IVGA or OVGA style, rather than mixed modes. With this limitation in mind, and simply to illustrate the unusual possibilities afforded by the AD8330, note that, with appropriate drive to VDBS and VMAG in tandem, the gain span is a remarkable 120 dB, extending from −50 dB to +70 dB, as shown in Figure 52 for operation at 1 MHz and 100 MHz. In this case, VDBS and VMAG are driven from a common control voltage, VGAIN, that varies from 1.2 mV to 5 V, with 30% (1.5/5) of VGAIN applied to VDBS, and 100% applied to VMAG. The gain varies in a linear-in-dB manner with VDBS, although the response from VMAG is linear-in-magnitude. Consequently, the overall numerical gain as the product of these two functions is GAIN = VGAIN / 0.5 V × 0.3 ×10 VGAIN 0. 6 V In rare cases where such a wide gain range is of value, the calibration is still accurate and the temperature is stable. Rev. H | Page 18 of 32 (7) Data Sheet AD8330 80 perfect, the noise figure cannot be better than 3 dB. The 1 kΩ internal termination resistance results in a minimum noise figure of 3 dB for an RS of 1 kΩ if the amplifier were noise-free. However, this is not the case, and the minimum noise figure occurs at a slightly different value of RS (for an example, see Figure 53 and the Using the AD8330 section). 40 20 0 –20 –40 15 10k 14 1k 13 100 12 10 0.1 VGAIN (V) 1 10 Figure 52. Gain Control Function and Input Referred Noise Spectral Density over a 120 dB Range 10 9 8 7 Noise, Input Capacity, and Dynamic Range Below midgain (25 dB, VDBS = 0.75 V), noise in the output section dominates, and the total input noise is 11 nV/√Hz, or 4.9 µV rms in a 200 kHz bandwidth, and the peak input is 78 mV rms. Thus, the dynamic range increases to 84 dB. At minimum gain, the input noise is up to 120 nV/√Hz, or 53.7 mV rms in the assumed 200 kHz bandwidth, while the input capacity is ±2 V, or +1.414 V rms (sine), a dynamic range of 88.4 dB. In calculating the dynamic range for other channel bandwidths, ∆f, subtract 10 log10(∆f/200 kHz) from these illustrative values. A system operating with a 2 MHz bandwidth, for example, exhibits dynamic range values that are uniformly 10 dB lower; used in an audio application with a 20 kHz bandwidth, they are 10 dB higher. Noise figure is a misleading metric for amplifiers that are not impedance matched at their input, which is the special condition resulting only when both the voltage and current components of a signal, that is, the signal power, are used at the input port. When a source of impedance (RS) is terminated using a resistor of RS (a condition that is not to be confused with matching), only one of these components is used, either the current (as in the AD8330) or the voltage. Then, even if the amplifier is 6 5 10 100 RS (Ω) 1k 10k Figure 53. Noise Figure for Source Resistance of 50 Ω to 5 kΩ, at f = 10 MHz (Lower) and 100 MHz (Simulation) 144 140 DYNAMIC RANGE (dB/√Hz) The design of variable gain amplifiers invariably incurs some compromises in noise performance. However, the structure of the AD8330 is such that this penalty is minimal. Examination of the simplified schematic (Figure 47) shows that the input voltage is converted to current-mode form by the two 500 Ω resistors at Pin INHI and Pin INLO, whose combined Johnson noise contributes 4.08 nV/√Hz. The total input noise at full gain, when driven from a low impedance source, is typically 5 nV/√Hz after accounting for the voltage and current noise contributions of the loop amplifier. For a 200 kHz channel bandwidth, this amounts to 2.24 μV rms. The peak input at full gain is ±6.4 mV, or +4.5 mV rms for a sine wave signal. The signal-to-noise ratio at full input, that is, the dynamic range, for these conditions is, thus, 20 log10(4.5 mV/2.24 μV), or 66 dB. The value of VMAG has essentially no effect on the input referred noise, but it is assumed to be 0.5 V. 03217-054 0.01 11 CONSTANT 1V rms OUTPUT, BOTH CASES 136 132 128 X-AMP WITH 40dB OF GAIN AND AN INPUT NSD OF nV/√Hz 124 120 116 0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1.0 1.1 1.2 1.3 1.4 1.5 VDBS (V) 03217-055 1 0.001 03217-053 NOISE (nV/√Hz) –60 100k NOISE FIGURE GAIN (dB) 60 Figure 54. Dynamic Range in dB/√Hz vs. VDBS (VMAG = 0.5 V, 1 V rms Output) Compared with a Representative X-AMP (Simulation) Dynamic Range The ratio of peak output swing, expressed in rms terms, to the output-referred noise spectral density provides a measure of dynamic range, in dB/√Hz. For a certain class of variable gain amplifiers, exemplified by the Analog Devices X-AMP® family, the dynamic range is essentially independent of the gain setting because the peak output swing and noise are both constant. The AD8330 provides a different dynamic range profile because there is no longer a constant relationship between these two parameters. Figure 54 compares the dynamic range of the AD8330 to a representative X-AMP. Input Common-Mode Range and Rejection Ratio AC-couple the input pins, INHI and INLO, in most applications to achieve the stated noise performance. In general, when direct coupling is used, care must be taken in setting the dc voltage level at these inputs, and particularly when minimizing noise is Rev. H | Page 19 of 32 AD8330 Data Sheet critical. This objective is complicated by the fact that the common-mode level varies with the basic gain voltage, VDBS. Figure 55 shows this relationship for a supply voltage of 5 V, for temperatures of −40°C, +25°C, and +85°C. Figure 56 shows the input noise spectral density (RS = 0) vs. the input commonmode voltage, for VDBS = 0.5 V, 0.6 V, 0.75 V, and 1.5 V. It is apparent that there is a broad range over which the noise is unaffected by this dc level. The input CMRR is excellent (see Figure 16). T = +85°C VNOISE_OUT = (0.1 + 0.32 VMAG) µV/√Hz for RS = 0 and VDBS = 1.5 V, assuming an input noise of 5 nV/√Hz. The output noise for very small values of VMAG (at or below 15 mV) is not precise, partly because the small input offset associated with this interface has a large effect on the gain. The AD8330 includes an offset compensation feature that is operational in the default condition (no connection to Pin OFST). This loop introduces a high-pass filter function into the signal path, whose −3 dB corner frequency is at 3.0 T = –40°C 2.9 f HPF = 2.8 2.7 2.6 0 0.2 0.4 0.6 0.8 VDBS (V) 1.0 1.2 1.4 1.6 VDBS = 0.6V 16 14 VDBS = 0.75V 12 SIMULATION 6 4 0 0.4 VDBS = 1.5V 0.8 1.2 1.6 2.0 2.4 2.8 3.2 3.6 4.0 COMMON-MODE VOLTAGE AT INHI, INLO (V) 4.4 C HP (C in μF) HP (12) The offset compensation feature can be disabled simply by grounding the OFST pin. This provides a dc-coupled signal path, with no other effects on the overall ac response. Input offsets must be externally nulled in this mode of operation, as shown in Figure 58. 10 8 330 µ A small amount of peaking at this corner when using small capacitor values can be avoided by adding a series resistor. Useful combinations are CHP = 3 nF, RHP = 180 Ω, f = 100 kHz; CHP = 33 nF, RHP = 10 Ω, f = 10 kHz; CHP = 0.33 μF, RHP = 0 Ω, f = 1 kHz; CHP = 3.3 μF, RHP = 0 Ω, f = 100 Hz. 20 4.8 03217-057 INPUT REFERRED NOISE (nV/√Hz) f HPF = VDBS = 0.5V 18 (11) This evaluates to 26 22 1 (2π RINT CHP ) where: CHP is the external capacitance added from OFST to CNTR. RINT is an internal resistance of approximately 480 Ω, having a maximum uncertainty of about ±20%. Figure 55. Common-Mode Voltage at Input Pins vs. VDBS, for VS = 5 V, T = −40°C, + 25°C, and + 85°C 24 (10) Offset Compensation T = +25°C 3.1 03217-056 DC VOLTAGE AT INHI, INLO (V) 3.2 For example, using a reduced value of VMAG = 0.25 V that lowers all gain values by 6 dB, the peak output swing is ±1 V (differentially) and the output noise spectral density evaluates to 102.5 nV/√Hz. The peak output swing is no different at full gain, but the noise becomes Effects of Loading on Gain and AC Response Figure 56. Input Noise vs. Common-Mode Input Voltage for VDBS = 0.5 V, 0.6 V, 0.75 V, and 1.5 V Output Noise and Peak Swing The output noise of the AD8330 is the input noise multiplied by the overall gain, including any optional change to the voltage, VMAG, applied to Pin VMAG. The peak output swing is also proportional to this voltage, which, at low gains and high values of VMAG, affects the output noise. The scaling for VDBS = 0 V is as follows: VOUT_PK = ±4 VMAG (8) VNOISE_OUT = (85 + 70 VMAG) nV/√Hz (9) The differential output impedance (RO) is 150 Ω, and the frequency response of the output stage is optimized for operation with a certain load capacitance on each output pin (OPHI and OPLO) to ground, in combination with a load resistance (RL) directly across these pins. In the absence of these capacitances, there is a small amount of peaking at the top extremity of the ac response. Suitable combinations are: RL = ∞, CL = 12 pF; RL = 150 Ω, CL = 25 pF; RL = 75 Ω, CL = 40 pF; or RL = 50 Ω, CL = 50 pF. The gain calibration is specified for an open-circuited load, such as the high input resistance of an ADC. When resistively loaded, all gain values are nominally lowered as follows: G LOADED = Rev. H | Page 20 of 32 GUNLOADED R L (150 Ω + R L ) (13) Data Sheet AD8330 Gain Errors Due to On-Chip Resistor Tolerances In all cases where external resistors are used, keep in mind that all on-chip resistances, including the RO and the input resistance (RI), are subject to variances of up to ±20%. the output stage where decoupling can be useful in maintaining a glitch-free output. Figure 57 shows the general case, where VPSI and VPSO are each provided with their own decoupling network, but this is not needed in all cases. VS 2.7V TO 6V RD1 These variances need to be accounted for when calculating the gain with input and output loading. This sensitivity can be avoided by adjusting the source and load resistances to bear an inverse relationship as follows: CHPF ENBL CD1 VPSI If RS = αRI, then make RL = RO/α; or, if RL = αRO, then make RS = RI/α The simplest case is when RS = 1 kΩ and RL = 150 Ω, therefore, the gain is 12 dB lower than the basic value. The reduction of peak swing at the load can be corrected by using VMAG = 1 V, thereby restoring 6 dB of gain; using VMAG = 2 V restores the full basic gain and doubles the peak available output swing. The input common-mode voltage, VCMI, at Pin INHI and Pin INLO is slaved to the output. It bears a y = mx + b linear and offset relationship to VCNTR as shown in Equation 14 where y = VCMI, m = 0.757, x = VCNTR, and b = 1.12 V for VDBS = 0.75 V and T = 25°C. VCMI = 0.757 VCNTR + 1.12 V (14) The effects of VDBS and ambient temperature on VCMI are shown in Figure 55. Thus, the default value for VCMI for VDBS = 0.75V, T = 25°C and VS = 5 V is 3.01 V. USING THE AD8330 This section describes a few general aspects of using the AD8330. Applying the AD8330 to a wide variety of circumstances requires very few precautions. As in all high frequency circuits, careful observation of the ground nodes associated with each function is important. Three positive supply pins are provided: VPSI supports the input circuitry that often operates at a relatively high sensitivity; VPOS supports general bias sources and needs no decoupling; and VPSO biases BIAS AND V-REF VPOS INPUT, 0V TO ±2V MAX VGA CORE MODE VDBS BASIC GAIN BIAS VDBS: 0V TO 1.5V CD3 OPHI OUTPUT, ±2V MAX OUTPUT STAGES OPLO INLO NC CNTR VPSO CM MODE AND OFFSET CONTROL INHI Output (Input) Common-Mode Control The output voltages are nominally positioned at the midpoint of the supply, VS/2, over the range 2.7 V < VS < 6 V, and this voltage appears at Pin CNTR, which is not normally expected to be loaded (the source resistance is ~4 kΩ). However, some circumstances require a small change in this voltage, and a resistor from CNTR to ground can lower this voltage, whereas a resistor to the supply raises it. On the other hand, this pin can be driven by an external voltage source to set the common-mode level to satisfy, for example, the needs of a following ADC. Any value from 0.5 V above ground to 0.5 V below the supply is permissible. Of course, when using an extreme common-mode level, the available output swing is limited, and it is recommended that a value equal or close to the default of VCNTR = VS/2 be used. There may be a few millivolts of offset between the applied voltage and the actual common-mode level at the output pins. OFST RD2 CD2 GAIN INTERFACE CMGN OUTPUT CONTROL CMOP COMM VMAG NC GROUND 03217-058 Thus, when RL = 150 Ω, the gain is reduced by 6 dB; for RL = 75 Ω, the reduction is 9.5 dB; and for RL = 50 Ω, it is 12 dB. Figure 57. Power Supply Decoupling and Basic Connections Because of the differential nature of the signal path, power supply decoupling is, in general, much less critical than in a single-sided amplifier; and where the minimization of boardlevel components is especially crucial, it is possible that these pins need no decoupling at all. On the other hand, when the signal source is single-sided, giving extra attention to the decoupling on Pin VPSI is sometimes required. Likewise, care is required in decoupling the VPSO pin if the output is loaded on only one of its two output pins. The general common (COMM) and the output stage common (CMOP) are usually grounded as shown in the Figure 57; however, the Applications Information section shows how a negative supply can optionally be used. The AD8330 is enabled by taking the ENBL pin to a logical high (or, in all cases, the supply). The UP gain mode is enabled either by leaving the MODE pin unconnected or taking it to a logical high. When the opposite gain direction is needed, ground the MODE pin or drive it to a logical low. The CHPF capacitor determines the low-pass corner of the offset loop; this is preferably tied to the CNTR pin that in turn, must be decoupled to ground. The gain interface common pin (CMGN) is grounded, and the output magnitude control pin (VMAG) is left unconnected, or can optionally be connected to a 500 mV source for basic gain calibration. Connections to the input and output pins are not shown in Figure 57 because of the many options that are available. When the AD8330 is used to drive an ADC, connect the OPHI and OPLO pins directly to the differential inputs of a suitable converter, such as an AD9214. Rev. H | Page 21 of 32 AD8330 Data Sheet When the loads to be driven introduce a dc resistive path to ground, coupling capacitors must be used. These must be of sufficient value to pass the lowest frequency components of the signal without excessive attenuation. Keep in mind that the voltage swing on such loads alternates both above and below ground, requiring that the subsequent component must be able to cope with negative signal excursions. Gain and Swing Adjustments When Loaded The output can also be coupled to a load via a transformer to achieve a higher load power by impedance transformation. For example, using a 2:1 turns ratio, a 50 Ω final load presents a 200 Ω load on the output. The gain loss (relative to the basic value with no termination) is 20 log10{(200+150)/200} or 4.86 dB, which can be restored by raising the voltage on the VMAG pin by a factor of 104.86/20 or × 1.75, from its basic value of 0.5 V to 0.875 V. This also restores the peak swing at the 200 Ω level to ±2 V, or ±1 V into the 50 Ω final load. Whenever a stable supply voltage is available, additional voltage swing can be provided by adding a resistor from the VMAG pin to the supply. The calculation is based on knowing that the internal bias is delivered via a 5 kΩ source; because an additional 0.375 V is needed, the current in this external resistor must be 0.375 V/ 5 kΩ = 75 μA. Thus, using a 5 V supply, a resistor of 5 V − 0.875 V/ 75 μA = 55 kΩ is used. Based on this example, the corrections for other load conditions are easy to calculate. If the effects on gain and peak output swing due to supply variations cannot be tolerated, VMAG must be driven by an accurate voltage. DC-Coupled Signal Path In many cases, where the VGA is not required to provide its lowest noise, the full common-mode input range of zero to VS can be used without problems, avoiding the need for any ac coupling means. However, such direct coupling at both the input and output does not automatically result in a fully dc-coupled signal path. The internal offset compensation loop must also be disengaged by connecting the OFST pin to ground. Note that at the maximum basic gain of 50 dB (×316), every millivolt of offset at the input, arising from whatever source, causes an output offset of 316 mV, which is an appreciable fraction of the peak output swing. Because the offset correction loop is placed after the front-end variable gain sections of the AD8330, the most effective way of dealing with such offsets is at the input pins, as shown in Figure 58. For example, assume, for illustrative purposes, that the resistances associated with each side of the source in a certain application are 50 Ω. If this source has a very low (op amp) output impedance, insert the extra resistors, with a negligible noise penalty and an attenuation of only 0.83 dB. The resistor values shown provide a trim range of about ±2 mV. Using Single-Sided Sources and Loads Where the source provides a single-sided output, either INHI or INLO can be used for the input, with a polarity change when using INLO. The unused pin must be connected either through a capacitor to ground, or through a dc bias point that corresponds closely to the dc level on the active signal pin. The input CMRR over the full frequency range is illustrated in Figure 59. In some cases, an additional element such as a SAW filter (having a single-sided balanced configuration) or a flux-coupled transformer can be interposed. Where this element must be terminated in the correct impedance, other than 1 kΩ, it is necessary to add either shunt or series resistors at this interface. The dc common-mode voltage at the input pins varies with the supply, the basic gain bias, and temperature (see Figure 55); for this reason, many applications need to use coupling capacitors from the source that are large enough to support the lowest frequencies to be transmitted. Using one capacitor at each input pin, their minimum values can be readily found from the expression CIN_CPL= (320 µF/fHPF) VS 2.7V TO 6V RD1 Input Coupling (15) where fHPF is the –3dB frequency expressed in hertz. Thus, for an fHPF of 10 kHz, 33 nF capacitors are used. CD2 ENBL CD1 VPSI RS ASSUMED 50kΩ TO BE 50Ω ON EACH SIDE Occasionally, it is possible to avoid the use of coupling capacitors when the dc level of the driving source is within a certain range, as shown in Figure 56. This range extends from 3.5 V to 4.5 V when using a 5 V supply, and at high basic gains, where the effect of an incorrect dc level degrades the noise level due to internal aspects of the input stage. For example, suppose the driver, IC, is an LNA having an output topology in which its load resistors are taken to the supply, and the output is buffered by emitter followers. This presents a source for the AD8330 that can be directly coupled. Rev. H | Page 22 of 32 75kΩ OFST BIAS AND V-REF VPOS CM MODE AND OFFSET CONTROL INHI BASIC GAIN BIAS VDBS: 0V TO 1.5V OUTPUT, ±2V MAX OUTPUT STAGES INLO VDBS CD3 VPSO OPHI VGA CORE MODE CNTR RD2 OPLO GAIN INTERFACE CMGN OUTPUT CONTROL CMOP COMM VMAG NC GROUND Figure 58. Input Offset Nulling in a DC-Coupled System 03217-059 If an adjustment is needed to this common-mode level, it can be introduced by applying that voltage to the CNTR pin, or, more simply, by using a resistor from this pin to either ground or the supply (see the Applications Information section). The CNTR pin can also supply the common-mode voltage to an ADC that supports such a feature. Data Sheet 80 VDBS = 1.5V VDBS = 0.75V In Case 3 and Case 4, a further factor of ×1.33 is needed to make up the 2.5 dB loss, that is, raise VMAG to 2 V. With the restoration of gain, the peak output swing at the load is, likewise restored to ±2 V. OFST: ENABLED DISABLED 70 CMRR (dB) 60 50 40 Pulse Operation VDBS = 0V 30 20 10 1M 10M FREQUENCY (Hz) 1 00 M 03217-060 0 –10 50k 100k Because of the considerable variation between applications, only general recommendations can be made with regard to minimizing pulse overshoot and droop. The former can be optimized by adding small load capacitances, if necessary; the latter requires the use of sufficiently large capacitors (C1). Figure 59. Input CMRR vs. Frequency for Various Values of VDBS 30 LINE 1 LINE 3 10 0 LINE 4 –10 LINE 2 –20 PHASE (Degrees) –30 0 LINE 2 –100 LINE 3 –200 LINE 4 –300 –400 LINE 1 –500 –600 1M 10M FREQUENCY (Hz) 100M 500M 03217-061 GAIN (dB) 20 When using the AD8330 in applications where its transient response is of greater interest and the outputs are conveyed to their loads via coaxial cables, the added capacitances can slightly differ in value, and can be placed either at the sending or load end of the cables, or divided between these nodes. Figure 61 shows an illustrative example where dual, 1 meter, 75 Ω cables are driven through dc-blocking capacitors and are independently terminated at ground level. Figure 60. AC Gain and Phase for Various Loading Conditions When driving a single-sided load, either OPHI or OPLO can be used. These outputs are very symmetric, so the only effect of this choice is to select the desired polarity. However, when the frequency range of interest extends to the upper limits of the AD8330, attach a dummy resistor of the same value to the unused output. Figure 60 illustrates the ac gain and phase response for various loads and VDBS = 0.75 V. Line 1 shows the unloaded (CL = 12 pF) case for reference; the gain is 6 dB lower (20 dB) using only the single-sided output. Adding a 75 Ω load from OPHI to an ac ground results in Line 2. The gain becomes a factor of ×1.5 V or 3.54 dB lower, but artifacts of the output common-mode control loop appear in both the magnitude and phase response. Figure 62 shows typical results for VDBS = 0.24 V, a square wave input amplitude of 450 mV (the actual combination is not important), a rise time of 2 ns, and VMAG raised to 2.0 V. In the upper waveforms, the load capacitors are both zero, and a small amount of overshoot is visible; with 40 pF, the response is cleaner. A shunt capacitance of 20 pF from OPHI to OPLO has a similar effect. Coupling capacitors for this demonstration are sufficiently large to prevent any visible droop over this time scale. The outputs at the load side eventually assume a mean value of zero, with negative and positive excursions depending on the duty cycle. CD2 ENBL VPSI OFST BIAS AND V-REF VPOS CNTR CM MODE AND OFFSET CONTROL RD2 VS 2.7V–6V CD3 VPSO RL1 C1 INHI OPHI VGA CORE INLO MODE Adding a dummy 75 Ω to OPLO results in Line 3: the gain is a further 2.5 dB lower, at about 14 dB. The CM artifacts are no longer present but a small amount of peaking occurs. If objectionable, this can be eliminated by raising both of the capacitors on the output pins to 25 pF, as shown in Line 4 of Figure 60. The gain reduction incurred both by using only one output and by the additional effect of loading can be overcome by taking advantage of the VMAG feature, provided primarily for just such circumstances. Thus, to restore the basic gain in the first case (Line 1), apply a 1 V source to this pin; to restore the gain in the second case, this voltage must be raised by a factor of ×1.5 to 1.5 V. Rev. H | Page 23 of 32 VDBS CL1 OUTPUT STAGES C1 OPLO GAIN INTERFACE CMGN CL2 RL2 OUTPUT CONTROL CMOP COMM VMAG NC 03217-062 90 AD8330 Figure 61. Driving Dual Cables with Grounded Loads AD8330 Table 5. Preserving Absolute Gain 0 5ns 10ns 15ns 20ns 25ns 03217-063 1.2 1.0 0.8 0.6 0.4 0.2 0 –0.2 0.2 0 –0.2 –0.4 –0.6 –0.8 –1.0 –1.2 1.2 1.0 0.8 0.6 0.4 0.2 0 –0.2 0.2 0 –0.2 –0.4 –0.6 –0.8 –1.0 –1.2 Data Sheet Figure 62. Typical Pulse Response for Figure 61 The bandwidth from Pin VMAG to these outputs is somewhat higher than from the normal input pins. Thus, when this pin is used to rapidly modulate the primary signal, some further experimentation with response optimization may be required. In general, the AD8330 is very tolerant of a wide range of loading conditions. Preserving Absolute Gain Although the AD8330 is not laser trimmed, its absolute gain calibration, based mainly on ratios, is very good. Full details are found in the Specifications section and in the typical performance curves (see the Typical Performance Characteristics section). Nevertheless, having finite input and output impedances, the gain is necessarily dependent on the source and load conditions. The loss that is incurred when either of these is finite causes an error in the absolute gain. The absolute gain can also be uncertain due to the approximately ±20% tolerance in the absolute value of the input and output impedances. Often, such losses and uncertainties can be tolerated and accommodated by a correction to the gain control bias. On the other hand, the error in the loss can be essentially nulled by using appropriate modifications to either the source impedance (RS) or the load impedance (RL), or both (in some cases by padding them with series or shunt components). The formulation for this correction technique was previously described. However, to simplify its use, Table 5 shows spot values for combinations of RS and RL resulting in an overall loss that is not dependent on sample-to-sample variations in on chip resistances. Furthermore, this fixed and predictable loss can be corrected by an adjustment to VMAG, as indicated in Table 5. RS (Ω) 10 15 20 30 50 75 100 150 200 300 500 750 1k 1.5 k 2k Uncorrected Loss Factor dB 0.980 0.17 0.971 0.26 0.961 0.34 0.943 0.51 0.907 0.85 0.865 1.26 0.826 1.66 0.756 2.43 0.694 3.17 0.592 4.56 0.444 7.04 0.327 9.72 0.250 12.0 0.160 15.9 0.111 19.1 RL (Ω) 15 k 10 k 7.5 k 5.0 k 3.0 k 2.0 k 1.5 k 1.0 k 750 500 300 200 150 100 75 VMAG Required to Correct Loss 0.510 0.515 0.520 0.530 0.551 0.578 0.605 0.661 0.720 0.845 1.125 1.531 2.000 3.125 4.500 Calculation of Noise Figure The AD8330 noise is a consequence of its intrinsic voltage noise spectral density (ENSD) and the current noise spectral density (INSD). Their combined effect generates a net input noise, VNOISE_IN, that is a function of the input resistance of the device (RI), nominally 1 kΩ, and the differential source resistance (RS) as follows: VNOISE _ IN = {ENSD2 + I NSD2 (RI + RS )2 } (16) Note that purely resistive source and input impedances as a concession to simplicity is assumed. A more thorough treatment of noise mechanisms, for the case where the source is reactive, is beyond the scope of these brief notes. Also note that VNOISE_IN is the voltage noise spectral density appearing across INHI and INLO, the differential input pins. In preparing for the calculation of the noise figure, VSIG is defined as the open-circuit signal voltage across the source, and VIN is defined as the differential input to the AD8330. The relationship is simply VIN = VSIG RI (17) (RI + RS ) At maximum gain, ENSD is 4.1 nV/√Hz, and INSD is 3 pA/√Hz. Thus, the short-circuit voltage noise is VNOISE _ IN = {(4.1 n V/ Hz ) + (3pA / Hz ) (1 kΩ + 0) } = 2 2 2 5.08 nV/√Hz (18) Next, examine the net noise when RS = RI = 1 kΩ, often incorrectly called the matching condition, rather than source impedance termination, which is the actual situation in this case. Repeating the procedure, V NOISE _ IN = = 7.3 nV/√Hz Rev. H | Page 24 of 32 (4.1 nV / Hz ) + (3 pA / 2 Hz ) (1 kΩ + 1 kΩ) 2 2 (19) Data Sheet AD8330 The noise figure is the decibel representation of the noise factor, NFAC, commonly defined as follows: N FAC = SNR at Input SNR at Output (20) However, this is equivalent to N FAC = SNR at the Source SNR at the Input Pins (21) Let VNSD be the voltage noise spectral density √kTRS due to the source resistance. Using Equation 17 gives N FAC = = VSIG {RI /(RI + RS )}/ VNSD VIN /{VNOISE _ IN RS /(RI + RS )} P1dB and V1dB RI VNOISE _ IN RS VNSD (22) Then, using the result from Equation 19 for a source resistance of 1 kΩ, having a noise-spectral density of 4.08 nV/√Hz produces N FAC = (1 kΩ)(7.3 nV / (1 kΩ)(4.08 nV/ Hz ) = 1.79 ) Hz (23) Finally, converting this to decibels using NFIG = 10 log10(NFAC) (24) Thus, the resultant noise figure in this example is 5.06 dB, which is somewhat lower than the value shown in Figure 53 for this operating condition. Noise as a Function of VDBS The chief consequence of lowering the basic gain using VDBS is that the current noise spectral density INSD increases with the square root of the basic gain magnitude, GBN such that INSD = (3 pA/√Hz)(√GBN) In practice, however, the effect of device mismatches and junction resistances in the core cell, and other mechanisms in its supporting circuitry inevitably cause distortion, further aggravated by other effects in the later output stages. Some of these effects are very consistent from one sample to the next, while those due to mismatches (causing predominantly even-order distortion components) are quite variable. Where the highest linearity (and lowest noise) is demanded, consider using one of the XAMP products such as the AD603 (single-channel), AD604 (dual-channel), or AD8332 (wideband dual-channel with ultralow noise LNAs). (25) Therefore, at the minimum basic gain of ×0, INSD rises to 53.3 pA/√Hz. However, the noise figure rises to 17.2 db if it is recalculated using the procedures in Equation 16 through Equation 24. Distortion Considerations Continuously variable gain amplifiers invariably employ nonlinear circuit elements; consequently, it is common for their distortion to be higher than well-designed fixed gain amplifiers. The translinear multiplier principles used in the AD8330, in theory, yield extremely low distortion, a result of the fundamental linearization technique that is an inherent aspect of these circuits. In addition to the nonlinearities that arise within the core of the AD8330, at moderate output levels, another metric that is more commonly stated for RF components that deliver appreciable power to a load is the 1 dB compression point. This is defined in a very specific manner: it is that point at which, with increasing output level, the power delivered to the load eventually falls to a value that is 1 dB lower than it would be for a perfectly linear system. (Although this metric is sometimes called the 1 dB gain compression point, it is important to note that this is not the output level at which the incremental gain has fallen by 1 dB). As shown in Figure 49, the output of the AD8330 limits quite abruptly, and the gain drops sharply above the clipping level. The output power, on the other hand, using an external resistive load, RL, continues to increase. In the most extreme case, the waveform changes from the sinusoidal form of the test signal, with an amplitude just below the clipping level, VCLIP, to a square wave of precisely the same amplitude. The change in power over this range is from (VCLIP/√2)2/RL to (VCLIP)2/RL, that is, a factor of 2, or 3 dB in power terms. It can be shown that for an ideal limiting amplifier, the 1 dB compression point occurs for an overdrive factor of 2 dB. For example, if the AD8330 is driving a 150 Ω load and VMAG is set to 2 V, the peak output is nominally ±4 V (as noted previously, the actual value, when loaded. can differ because of a mismatch between on-chip and external resistors), or 2.83 V rms for a sine wave output that corresponds to a power of 53.3 mW, that is, 17.3 dBm in 150 Ω. Thus, the P1dB level, at 2 dB above clipping, is 19.3 dBm. Though not involving power transfer, it is sometimes useful to state the V1dB, which is the output voltage (unloaded or loaded) that is 2 dB above clipping for a sine waveform. In the above example, this voltage is still 2.83 V rms, which can be expressed as 9.04 dBV (0 dBV corresponds to a 1 V sine wave). Thus, the V1dB is at 11.04 dBV. Rev. H | Page 25 of 32 AD8330 Data Sheet APPLICATIONS INFORMATION from VMAG to ground. An overrange condition is signaled by a high state on Pin OR of the AD9214. DFS/GAIN is unconnected in this example producing an offset-binary output. To provide a twos complement output, connect it to the REF pin. The versatility of the AD8330, its very constant ac response over a wide range of gains, the large signal dynamic range, output swing, single supply operation, and low power consumption commend this VGA to a diverse variety of applications. Only a few can be described here, including the most basic uses and some unusual ones. For ADCs running at sampling rates substantially below the bandwidth of the AD8330, an intervening noise filter is recommended to limit the noise bandwidth. A one-pole filter can easily be created with a single differential capacitor between the OPHI and OPLO outputs. For a corner frequency of fC, the capacitor must have a value of ADC DRIVING The AD8330 is well-suited to drive a high speed converter. There are many high speed converters available, but to illustrate the general features, the example in this data sheet uses one of the least expensive, the AD9214. This is available in three grades for operation at 65 MHz, 80 MHz, and 105 MHz; the AD9214BRSZ-80 is a good complement to the general capabilities of this VGA. CFILT = 1/942 fC (26) For example, a 10 MHz corner requires about 100 pF. SIMPLE AGC AMPLIFIER Figure 64 illustrates the use of the inverted gain mode and the offset gain range (0.2 V < VDBS < 1.7 V) in supporting a low cost AGC loop. Q1 is used as a detector. When OPHI is sufficiently higher than CNTR, due to the signal swing, it conducts and charges C1. This raises VDBS and rapidly lowers the gain. Note that MODE is grounded (see Figure 48). The minimum voltage needed across R1 to set up the full gain is 0.2 V because CMGN is dc open-circuited (this does not alter VMAG) and the maximum voltage is 1.7 V. Figure 63 shows the connections to drive an ADC. A 3.3 V supply is used for both parts. The ADC requires that its input pins be positioned at one third of the supply, or 1.1 V. Given that the default output level of the VGA is one-half the supply or 1.65 V, a small correction is introduced by the 8 kΩ resistor from CNTR to ground. The ADC specifications require that the common-mode input be within ±0.2 V of the nominal 1.1 V; variations of up to ±20% in the AD8330 on-chip resistors change this voltage by only ±70 mV. With the connections shown in Figure 63, the AD9214 is able to receive an input of 2 V p-p; the peak output of the AD8330 can be reduced if desired by adding a resistor VS, 3.3V 0.1µF 3.3Ω 0.1µF OVERRANGE 3.3Ω CHPF 0.1µF OFST VPOS CNTR VPSO CM MODE AND OFFSET CONTROL BIAS AND V-REF AVDD OR DrVDD PWRDN D8 DFS/GAIN INHI INPUT, ±2V MAX OPHI VGA CORE D7 AIN D6 OUTPUT STAGES D5 AD9214BRS-80 OPLO INLO D4 D3 AIN D2 REFSENSE NC MODE VDBS GAIN BIAS, VDBS , 0V–1.5V GAIN INTERFACE CMGN OUTPUT CONTROL CMOP COMM NC D1 REF D0 AGND VMAG D9 DATA OUTPUTS ENBL VPSI 0.1µF CLK DGND 0.1µF ANALOG GROUND CLOCK Figure 63. Driving an Analog-to-Digital Converter (Preliminary) Rev. H | Page 26 of 32 DIGITAL GROUND 03217-064 10Ω 8kΩ Data Sheet AD8330 VS, 2.7V–6V This simple detector exhibits a temperature variation in the differential output amplitude of about 4 mV/°C. It provides a fast attack time (an increase in the input is quickly leveled to the nominal output, due to the high peak currents in Q1) and a slow release time (a decrease in the input is not restored as quickly). The voltage at the VDBS pin can be used as an RSSI output, scaled 30 mV/dB. Note that the attack time can be halved by adding a second transistor, labeled Q2 in Figure 64. For operation at lower frequencies, the AGC hold capacitor must be increased. 33nF 10Ω ENBL VPSI 0.1µF OFST VPOS CM MODE AND OFFSET CONTROL BIAS AND V-REF OUTPUT STAGES VGA CORE INLO MODE VPSO 0.1µF OPHI INHI INPUT, 5mV TO 1V rms 4.7Ω CNTR SEE TEXT OUTPUT, ~1V rms OPLO GAIN INTERFACE VDBS CMGN Q2 OUTPUT CONTROL CMOP COMM WIDE RANGE TRUE RMS VOLTMETER Q1 VMAG 0.1µF NC C1 0.1µF 0.1µF 03217-065 R1 10kΩ Figure 64. Simple AGC Amplifier (Preliminary) When the loop is settled, the average current in Q1 is VDBS/R1, which varies from 20 µA at maximum gain (VDBS = 0.2 V) to 170 µA at minimum gain (VDBS = 1.7 V). This change in the Q1 current causes an increase of ~0.25 dB over the full gain range in the differential output of nominally 0.75 dBV at midrange (3.08 V p-p), corresponding to a 200:1 compression ratio. This is plotted in Figure 65 for a representative 100 kHz input. 1.0 The AD8362 is an rms responding detector providing a dynamic range of 60 dB from low frequencies to 2.7 GHz. This can increase to 110 dB using an AD8330 as a preconditioner, provided the noise bandwidth is limited by an interstage low-pass or bandpass filter. The VGA also provides an input port that is easier to drive than the 200 Ω input of the AD8362. Figure 67 shows the general scheme. Both the AD8330 and AD8362 provide linear-in-decibel control interfaces. Thus, when the output of the AD8362 is used to control the gain of the AD8330, the functional form is unaffected. The overall scaling is 33 mV/dB. Figure 68 shows the time domain response using a loop filter capacitor of 10 nF, for inputs ranging from 10 μV to 1 V rms, that is, a 100 dB measurement range. 1.75 1.50 VDBS 1.25 1.00 0.75 0.8 GAIN ERROR (dB) LEVELED OUTPUT (dBV) 0.9 0.7 0.6 0.50 0.25 0 3 2 1 0 –1 –30 –20 INPUT TO AD8330 (dBV) –10 0 –2 OUTPUT –3 –4 Figure 65. AGC Output vs. Input Amplitude (Simulation) The upper panel in Figure 66 shows the time-domain output for fourteen 3 dB steps in input amplitude from 5.4 mV to 1.7 V. The waveforms in Figure 65 show the AGC voltage (VDBS). Rev. H | Page 27 of 32 0 10 20 30 40 50 60 70 80 TIME (µs) 90 100 110 120 130 140 150 Figure 66. Time Domain Waveforms (Simulation) 03217-067 –40 03217-066 0.5 –50 AD8330 Data Sheet 5V 3.3Ω 3.3Ω 0.1µF AD8362 0.1µF VPOS VPS1 CNTR VPSO INHI OPHI CFLT 18nF AD8330 INPUT INLO OPLO MODE CMOP CMGN COMM VMAG 2 CHPF VREF 15 3 DECL VTGT 14 4 INHI VPOS 13 5 INLO VOUT 12 6 DECL VSET 11 7 PWDN ACOM 10 8 COMM CLPF 9 0.1µF 3.6V VOUT 0.1µF 10µF 6.04kΩ 4.02kΩ Figure 67. Wide Range True RMS Voltmeter (Preliminary) 4 3 OUTPUT (V) VDBS ACOM 16 2 1 0 0 0.4 0.8 1.2 1.6 2.0 2.4 2.8 TIME (ms) 3.2 3.6 4.0 4.4 4.8 Figure 68. Time Domain Response of RMS Voltmeter (Simulation) Rev. H | Page 28 of 32 03217-068 OFST COMM 10µF 3.6V ENBL 1 03217-069 0.1µF 3.3Ω Data Sheet AD8330 EVALUATION BOARD GENERAL DESCRIPTION BASIC OPERATION The AD8330-EVALZ is an easy-to-use accessory that enables a hands-on evaluation of the AD8330 variable gain amplifier (VGA). It includes test pins for connections to all of the functional device inputs. Figure 69 is a full size photograph of the board. The input SMA connector IN is terminated with a 49.9 Ω resistor (see Figure 70). For convenience, the board includes an AD8131 high speed differential amplifier to convert a singleended signal source to the differential input of the AD8330. If desired, the AD8131 can be removed and the AD8330 can be driven at one of its inputs from a single-ended source. The AD8330 output is observed at the SMA connectors OUT_HI and OUT_LO or by using the 2-pin header OUT_HI/OUT_LO adjacent to the device. 03217-070 The AD8330 requires only a +5 V power supply; however, because of the AD8131 buffer bipolar power supply requirements, ±5 V supplies are required to power the board. The current required for the board is approximately 40 mA from the +5 V supply and 10 mA from the −5 V supply. Figure 69. Photograph of the AD8330 Evaluation Board C13 1nF FILTER_OFFSET FLTR OFST 1 +5V R4 0Ω C1 + 10µF 10V C18 0.1µF 2 1 R1 24.9Ω A1 6 CNTR VPSI VPSO OPHI INHI +5V GAIN_SLOPE 16 15 14 13 AD8330 INLO OPLO +5V C12 0.1µF C19 0.1µF +5V C11 0.1µF C9 12pF DUT1 5 C5 0.1µF 12 C8 12pF UP 6 MODE CMOP VDBS VMAG CMGN COMM CNTR C4 0.1µF OUT_HI OUT_TEST OUT_ LO C10 0.1µF 11 DOWN VDBS –5V C17 10µF 10V ENBL IN_TEST 5 C16 10nF + VPOS 4 C3 0.1µF AD8131 8 3 4 OFST C20 0.1µF 7 C6 0.1µF 8 CMGN Figure 70. Schematic Diagram Rev. H | Page 29 of 32 10 9 C7 0.1µF VMAG 03217-071 IN 2 C2 0.1µF 3 C14 10nF C15 0.1µF R2 49.9Ω ENBL GND1 GND2 GND3 GND4 R3 1kΩ AD8330 Data Sheet OPTIONS MEASUREMENT SETUP Table 6 lists the jumpers on the board and their functions. The basic board connections for a typical measurement are shown in Figure 71. To minimize circuit-loading effects, a low capacitance FET probe is recommended for observing input or output waveforms. Two-pin headers, IN_TEST and OUT_TEST, are provided for this purpose. The SMA connectors OUT_HI and OUT_LO can also be used, but the user may need to account for load capacitance effects. Table 6. Functions of Jumpers OFST UP DOWN Function Connects a high-pass filter to the offset control loop pin. This jumper is normally not installed. Disables the offset correction loop. This jumper is installed for dc or low frequency operation. Mode up. Install for ascending gain with increasing VDBS gain control voltage. Mode down. Install for descending gain with increasing VDBS gain control voltage. AD8330-EVALZ BOARD DESIGN The AD8330-EVALZ is a 4-layer design for maximum groundplane area. The evaluation board side silkscreen and wiring patterns are shown in Figure 72 through Figure 77. NETWORK ANALYZER PROBE POWER SUPPLY SIGNAL INPUT POWER SUPPLY DIFFERENTIAL PROBE +5V GND PRECISION VOLTAGE REFERENCES (FOR VDBS, VMAG) Figure 71. Typical Connections Rev. H | Page 30 of 32 03217-072 Name FLTR AD8330 Figure 72. Component Side Silkscreen 03217-076 03217-073 Data Sheet Figure 73. Component Side Wiring 03217-077 03217-074 Figure 75. Wiring Side Silkscreen 03217-075 03217-078 Figure 76. Wiring Side Pattern Figure 77. Inner Layer 2 Figure 74. Ground Plane Rev. H | Page 31 of 32 AD8330 3.10 3.00 SQ 2.90 PIN 1 INDICATOR 0.30 0.25 0.20 0.50 BSC 12 13 16 PIN 1 INDICATOR 1 1.65 1.50 SQ 1.45 EXPOSED PAD 9 0.50 0.40 0.30 TOP VIEW 0.80 0.75 0.70 5 8 BOTTOM VIEW 0.05 MAX 0.02 NOM COPLANARITY 0.08 0.20 REF SEATING PLANE 4 0.20 MIN FOR PROPER CONNECTION OF THE EXPOSED PAD, REFER TO THE PIN CONFIGURATION AND FUNCTION DESCRIPTIONS SECTION OF THIS DATA SHEET. 01-26-2012-A OUTLINE DIMENSIONS COMPLIANT TO JEDEC STANDARDS MO-220-WEED-6. Figure 78. 16-Lead Lead Frame Chip Scale Package [LFCSP] 3 mm × 3 mm Body and 0.75 mm Package Height (CP-16-27) Dimensions shown in millimeters 0.197 (5.00) 0.193 (4.90) 0.189 (4.80) 9 1 8 0.244 (6.20) 0.236 (5.99) 0.228 (5.79) 0.010 (0.25) 0.006 (0.15) 0.069 (1.75) 0.053 (1.35) 0.065 (1.65) 0.049 (1.25) 0.010 (0.25) 0.004 (0.10) COPLANARITY 0.004 (0.10) 0.158 (4.01) 0.154 (3.91) 0.150 (3.81) 0.025 (0.64) BSC 0.012 (0.30) 0.008 (0.20) SEATING PLANE 8° 0° 0.050 (1.27) 0.016 (0.41) COMPLIANT TO JEDEC STANDARDS MO-137-AB CONTROLLING DIMENSIONS ARE IN INCHES; MILLIMETER DIMENSIONS (IN PARENTHESES) ARE ROUNDED-OFF INCH EQUIVALENTS FOR REFERENCE ONLY AND ARE NOT APPROPRIATE FOR USE IN DESIGN. 0.020 (0.51) 0.010 (0.25) 0.041 (1.04) REF 09-12-2014-A 16 Figure 79. 16-Lead Shrink Small Outline Package [QSOP] (RQ-16) Dimensions shown in inches and (millimeters) ORDERING GUIDE Model 1 AD8330ACPZ-R2 AD8330ACPZ-RL AD8330ACPZ-R7 AD8330ARQZ AD8330ARQZ-RL AD8330ARQZ-R7 AD8330-EVALZ 1 Temperature Range −40°C to +85°C −40°C to +85°C −40°C to +85°C −40°C to +85°C −40°C to +85°C −40°C to +85°C Package Description 16-Lead Lead Frame Chip Scale Package [LFCSP] 16-Lead Lead Frame Chip Scale Package [LFCSP] 16-Lead Lead Frame Chip Scale Package [LFCSP] 16-Lead Shrink Small Outline Package [QSOP] 16-Lead Shrink Small Outline Package [QSOP] 16-Lead Shrink Small Outline Package [QSOP] Evaluation Board Z = RoHS Compliant Part. ©2002–2016 Analog Devices, Inc. All rights reserved. Trademarks and registered trademarks are the property of their respective owners. D03217-0-5/16(H) Rev. H | Page 32 of 32 Package Option CP-16-27 CP-16-27 CP-16-27 RQ-16 RQ-16 RQ-16 Branding JFZ JFZ JFZ