AD9022SQ

a FEATURES Monolithic 12-Bit 20 MSPS A/D Converter Low Power Dissipation: 1.4 Watts On-Chip T/H and Reference High Spurious-Free Dynamic Range TTL Logic APPLICATIONS Radar Receivers Digital Communications Digital Instrumentation Electro-Optics ANALOG INPUT T/H 12-Bit 20 MSPS Monolithic A/D Converter AD9022 FUNCTIONAL BLOCK DIAGRAM 5-BIT ADC DIGITAL 12 ERROR CORRECTION TTL AD9022 DAC 5-BIT ADC ENCODE +5V T/H 16 DAC 4-BIT ADC –5.2V GND 8 +2V REF PRODUCT DESCRIPTION The AD9022 is a high speed, high performance, monolithic 12-bit analog-to-digital converter. All necessary functions, including track-and-hold (T/H) and reference, are included on-chip to provide a complete conversion solution. It is a companion unit to the AD9023; the primary difference between the two is that all logic for the AD9022 is TTL-compatible, while the AD9023 utilizes ECL logic for digital inputs and outputs. Pinouts for the two parts are nearly identical. Operating from +5 V and –5.2 V supplies, the AD9022 provides excellent dynamic performance. Sampling at 20 MSPS with AIN = 1 MHz, the spurious-free dynamic range (SFDR) is typically 76 dB; with AIN = 9.6 MHz, SFDR is 74 dB. SNR is typically 65 dB. The onboard T/H has a 110 MHz bandwidth and, more importantly, is designed to provide excellent dynamic performance for analog input frequencies above Nyquist. This feature is necessary in many undersampling signal processing applications, such as in direct IF-to-digital conversion. To maintain dynamic performance at higher IFs, monolithic RF track-and-holds (such as the AD9100 and AD9101 Samplifier™) can be used with the AD9022 to process signals up to and beyond 70 MHz. With DNL typically less than 0.5 LSB and 20 ns transient response settling time, the AD9022 provides excellent results when low-frequency analog inputs must be oversampled (such as CCD digitization). The full scale analog input is ± 1 V with a 300 Ω input impedance. The analog input can be driven directly from the signal source, or can be buffered by the AD96xx series of low noise, low distortion buffer amplifiers. All timing is internal to the AD9022; the clock signal initiates the conversion cycle. For best results, the encode command should contain as little jitter as possible. High speed layout practices must be followed to ensure optimum A/D performance. The AD9022 is built on a trench isolated bipolar process and utilizes an innovative multipass architecture (see the block diagram). The unit is packaged in 28-lead ceramic DIPs and gullwing surface mount packages. The AD9022 is specified to operate over the industrial (–25°C to +85°C) and military (–55°C to +125°C) temperature ranges. Samplifier is a trademark of Analog Devices, Inc. REV. B Information furnished by Analog Devices is believed to be accurate and reliable. However, no responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other rights of third parties which may result from its use. No license is granted by implication or otherwise under any patent or patent rights of Analog Devices. One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A. Tel: 781/329-4700 World Wide Web Site: http://www.analog.com Fax: 781/326-8703 © Analog Devices, Inc., 1998 AD9022–SPECIFICATIONS ELECTRICAL CHARACTERISTICS Parameter (Conditions) RESOLUTION DC ACCURACY Differential Nonlinearity Integral Nonlinearity No Missing Codes Offset Error Gain Error Thermal Noise ANALOG INPUT Input Voltage Range Input Resistance Input Capacitance Analog Bandwidth SWITCHING PERFORMANCE 1 Minimum Conversion Rate Maximum Conversion Rate Aperture Delay (tA) Aperture Uncertainty (Jitter) Output Delay (tOD) ENCODE INPUT Logic Compatibility Logic “1” Voltage Logic “0” Voltage Logic “1” Current Logic “0” Current Input Capacitance Pulsewidth (High) Pulsewidth (Low) DYNAMIC PERFORMANCE Transient Response Overvoltage Recovery Time Harmonic Distortion Analog Input @ 1.2 MHz @ 1.2 MHz @ 4.3 MHz @ 9.6 MHz @ 9.6 MHz Signal-to-Noise Ratio 2 Analog Input @ 1.2 MHz @ 1.2 MHz @ 4.3 MHz @ 9.6 MHz @ 9.6 MHz Signal-to-Noise Ratio 2 (Without Harmonics) Analog Input @ 1.2 MHz @ 1.2 MHz @ 4.3 MHz @ 9.6 MHz @ 9.6 MHz +25°C Full +25°C Full Full +25°C Full +25°C Full +25°C I VI I VI VI I VI I VI V Temp (+VS = +5 V; –VS = –5.2 V; Encode = 20 MSPS, unless otherwise noted) Test Level AD9022AQ/AZ Min Typ Max 12 0.75 1.0 1.3 2.5 1.6 3.0 Guaranteed 5 25 15 35 0.5 2.5 0.6 3.5 0.57 ± 1.024 300 360 5 110 0.6 AD9022BQ/BZ Min Typ Max 12 0.5 1.0 1.3 2.0 1.6 3.0 Guaranteed 5 25 15 35 0.5 2.5 0.6 3.5 0.57 ± 1.024 300 360 5 110 0.4 AD9022SQ/SZ Min Typ Max 12 0.75 1.0 1.3 2.5 1.6 3.0 Guaranteed 5 25 15 35 0.5 2.5 0.6 3.5 0.57 ± 1.024 300 360 5 110 0.6 Units Bits LSB LSB LSB LSB mV mV % FS % FS LSB, rms V Ω pF MHz MSPS MSPS ns ps, rms ns Full +25°C +25°C +25°C Full +25°C +25°C Full IV V V IV VI IV V VI 240 240 240 4 20 0.55 0.71 0.85 6 15 27.5 TTL 4 20 0.55 0.71 0.85 6 15 27.5 TTL 2.0 4 20 0.55 0.71 0.85 6 15 27.5 TTL 2.0 Full Full Full Full +25°C +25°C +25°C +25°C +25°C +25°C Full +25°C +25°C Full +25°C Full +25°C +25°C Full +25°C Full +25°C +25°C Full VI VI VI VI V IV IV V V I V V I V I V V I V 2.0 8 8 6 22.5 20 20 20 65 73 70 73 72 68 64 63 64 63 62 0.8 20 20 125 125 8 8 6 22.5 20 20 20 70 75 72 75 74 72 66 65 66 65 63 0.8 20 20 125 125 22.5 20 8 8 6 0.8 20 20 125 125 V V µA µA pF ns ns ns ns dBc dBc dBc dBc dBc dB dB dB dB dB 20 20 65 73 70 73 72 68 64 63 64 63 62 63 69 63 62 64 62 61 63 61 I V V I V 63 62 66 64 66 65 63 65 64 67 66 66 66 65 63 62 66 64 66 65 63 dB dB dB dB dB –2– REV. B AD9022 Parameter (Conditions) Two-Tone Intermodulation Distortion Rejection3 DIGITAL OUTPUTS1 Logic Compatibility Logic “1” Voltage Logic “0” Voltage Output Coding POWER SUPPLY +VS Supply Voltage +VS Supply Current –VS Supply Voltage –VS Supply Current Power Dissipation Power Supply Rejection Ratio (PSRR) 4 Temp +25°C Test Level V AD9022AQ/AZ Min Typ Max 74 TTL Full Full VI VI 2.4 0.5 Offset Binary 4.75 5.0 100 –5.45 –5.2 180 1.4 32 2.4 0.5 Offset Binary 5.25 120 –4.95 220 1.9 AD9022BQ/BZ Min Typ Max 74 TTL 2.4 0.5 Offset Binary 4.75 5.0 100 –5.45 –5.2 180 1.4 32 5.25 120 –4.95 220 1.9 AD9022SQ/SZ Min Typ Max 74 TTL V V Units dBc Full Full Full Full Full Full VI VI VI VI VI V 5.25 4.75 5.0 120 100 –4.95 –5.45 –5.2 220 180 1.9 1.4 32 mA mA mA mA W mV/V NOTES 1 AD9022 load is a single LS latch. 2 RMS signal-to-rms noise with analog input signal 1 dB below full scale at specified frequency. Tested at 55% duty cycle. 3 Intermodulation measured with analog input frequencies of 8.9 MHz and 9.8 MHz at 7 dB below full scale. 4 PSRR is sensitivity of offset error to power supply variations within the 5% limits shown. Specifications subject to change without notice. N ANALOG IN ta t a = 0.7 TYPICAL N+1 N+2 ENCODE tOD tOD = 15–27.5 TYPICAL DATA OUTPUT N–3 N–2 N–1 N AD9022 Timing Diagram ABSOLUTE MAXIMUM RATINGS 1 ORDERING GUIDE Model Temperature Range Package Description Package Option Q-28 Z-28 Q-28 Z-28 +VS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . +6 V –VS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . –6 V Analog Input . . . . . . . . . . . . . . . . . . . . . . . . . –1.5 V to +1.5 V Digital Inputs . . . . . . . . . . . . . . . . . . . . . . . . . . . . . +VS to 0 V Digital Output Current . . . . . . . . . . . . . . . . . . . . . . . . . 20 mA Operating Temperature Range AD9022AQ/AZ/BQ/BZ . . . . . . . . . . . . . . . –25°C to +85°C AD9022SQ/SZ . . . . . . . . . . . . . . . . . . . . . –55°C to +125°C Maximum Junction Temperature2 . . . . . . . . . . . . . . . . +175°C Lead Temperature (Soldering, 10 sec) . . . . . . . . . . . . +300°C Storage Temperature Range . . . . . . . . . . . . –65°C to +150°C NOTES 1 Absolute maximum ratings are limiting values to be applied individually, and beyond which the serviceability of the circuit may be impaired. Functional operability is not necessarily implied. Exposure to absolute maximum rating conditions for an extended period of time may affect device reliability. 2 Typical thermal impedances: “Q” Package (Ceramic DIP): θJC = 10°C/W; θJA = 35°C/W. “Z” Package (Gullwing Surface Mount): θJC = 13°C/W; θJA = 45°C/W. AD9022AQ/BQ –25°C to +85°C AD9022AZ/BZ –25°C to +85°C AD9022SQ AD9022SZ 28-Lead Ceramic DIP 28-Pin Ceramic Leaded Chip Carrier –55°C to +125°C 28-Lead Ceramic DIP –55°C to +125°C 28-Pin Ceramic Leaded Chip Carrier REV. B –3– AD9022 EXPLANATION OF TEST LEVELS Test Level PIN FUNCTION DESCRIPTIONS Pin No. 1–3 4 5 6 7 8 Name D3–D1 D0 (LSB) NC +VS GND ENCODE Function Digital output bits of ADC; TTL compatible. Least significant bit of ADC output; TTL compatible. No Connection Internally +5 V Power Supply Ground Encode clock input to ADC. Internal T/H is placed in hold mode (ADC is encoding) on rising edge of encode signal. Ground +5 V Power Supply Ground Noninverting input to T/H amplifier. –5.2 V Power Supply +5 V Power Supply –5.2 V Power Supply Ground Should be connected to –V S through 0.1 µF capacitor. I – 100% production tested. II – 100% production tested at +25°C, and sample tested at specified temperatures. AC testing done on sample basis. III – Sample tested only. IV – Parameter is guaranteed by design and characterization testing. V – Parameter is a typical value only. VI – All devices are 100% production tested at +25°C. 100% production tested at temperature extremes for extended temperature devices; guaranteed by design and characterization testing for industrial devices. 9 DIE LAYOUT AND MECHANICAL INFORMATION 10 11 12 13 14 15 16 17 18 19–25 26 27 28 GND +VS GND AIN –VS +VS –VS GND COMP Die Dimensions . . . . . . . . . . . . . . . . 205 × 228 × 21 (± 1) mils Pad Dimensions . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4 × 4 mils Metalization . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Aluminum Backing . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . None Substrate Potential . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . –VS Transistor Count . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4,080 Passivation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Oxynitride Bond Wire . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Aluminum D11 (MSB) Most significant bit of ADC output; TTL compatible. D10–D4 +VS –VS GND Digital output bits of ADC; TTL compatible. +5 V Power Supply –5.2 V Power Supply Ground PIN DESIGNATIONS D3 1 D2 2 D1 3 D0 (LSB) 4 NC 5 +VS 6 28 GND 27 –VS 26 +VS 25 D4 24 D5 AD9022 23 D6 TOP VIEW 22 D7 ENCODE 8 (Not to Scale) 21 D8 GND 7 GND 9 +VS 10 GND 11 AIN 12 –VS 13 +VS 14 20 D9 19 D10 18 D11(MSB) 17 COMP 16 GND 15 –VS NC = NO CONNECT COMPENSATION (PIN 17) SHOULD BE CONNECTED TO –VS THROUGH 0.01 F –4– REV. B AD9022 DEFINITIONS OF SPECIFICATIONS +VS Analog Bandwidth The analog input frequency at which the spectral power of the fundamental frequency (as determined by FFT analysis) is reduced by 3 dB. Aperture Delay +VS The delay between the rising edge of the ENCODE command and the instant at which the analog input is sampled. Aperture Uncertainty (Jitter) ANALOG INPUT 180 120 10pF The sample-to-sample variation in aperture delay. Differential Nonlinearity –VS The deviation of any code from an ideal 1 LSB step. Harmonic Distortion Analog Input +VS 100 ENCODE 900 The rms value of the fundamental divided by the rms value of the worst harmonic component. Integral Nonlinearity The deviation of the transfer function from a reference line measured in fractions of 1 LSB using a “best straight line” determined by a least square curve fit. Minimum Conversion Rate The encode rate at which the SNR of the lowest analog signal frequency tested drops by no more than 3 dB below the guaranteed limit. Maximum Conversion Rate –VS Encode Input COMPENSATION The encode rate at which parametric testing is performed. Output Propagation Delay The delay between the 50% point of the rising edge of the ENCODE command and the time when all output data bits are within valid logic levels. Overvoltage Recovery Time 50 –VS The amount of time required for the converter to recover to 12-bit accuracy after an analog input signal 150% of full scale is reduced to the full-scale range of the converter. Power Supply Rejection Ratio (PSRR) 20pF –VS The ratio of a change in input offset voltage to a change in power supply voltage. Signal-to-Noise Ratio (SNR) Compensation +VS The ratio of the rms signal amplitude to the rms value of “noise,” which is defined as the sum of all other spectral components, including harmonics but excluding dc, with an analog input signal 1 dB below full scale. Signal-to-Noise Ratio (Without Harmonics) 11k 12k DIGITAL OUTPUT The ratio of the rms signal amplitude to the rms value of “noise,” which is defined as the sum of all other spectral components, excluding the first five harmonics and dc, with an analog input signal 1 dB below full scale. Transient Response –VS The time required for the converter to achieve 12-bit accuracy when a step function is applied to the analog input. Two-Tone Intermodulation Distortion (IMD) Rejection Output Stage Figure 1. Equivalent Circuits The ratio of the power of either of two input signals to the power of the strongest third-order IMD signal. REV. B –5– AD9022–Typical Performance Characteristics –76 WORST CASE HARMONIC DISTORTION – dBc –75 –74 –73 +25 C ROOM 70 +25 C 65 60 –55 C SNR – dB –55 C –72 –71 55 50 45 +125 C –70 +125 C –69 –68 0 1 2 3 4 5 6 7 8 ANALOG INPUT FREQUENCY – MHz 9 10 40 35 1.24 2.3 5.3 7.3 9.6 11.3 13.3 15.3 17.3 19.3 ANALOG INPUT FREQUENCY – MHz Figure 2. Harmonic Distortion vs. Analog Input Frequency Figure 5. Signal-to-Noise Ratio vs. Analog Input Frequency 85 AIN = 1.2MHz 80 90 80 70 AIN = 1.2MHz ENCODE = 20MHz SFDR 60 50 SNR 40 30 20 HARMONICS AND SNR – dB 75 70 SNR 65 60 10 55 5.0 0 –50 7.5 10.0 12.5 15.0 17.5 20.0 ENCODE RATE – MSPS 22.5 25.0 SFDR AND SNR – dB WORST HARMONICS –45 –40 –35 –30 –25 –20 –15 –10 –5 –1 INPUT LEVEL – dB Figure 3. SNR and Harmonics vs. Encode Rate Figure 6. SFDR and SNR vs. Analog Input Level 2.0 DIFFERENTIAL NONLINEARITY – LSBs 80 AIN = 1.2MHz FS = 20MSPS 1.5 1.0 0.5 0 –0.5 –1.0 –1.5 –2.0 1024 2048 OUTPUT CODE 70 60 SFDR AND SNR – dB AIN = 9.6MHz ENCODE = 20MHz SFDR 50 SNR 40 30 20 10 0 –50 3072 4096 –45 –40 –35 –30 –25 –20 –15 –10 –5 –1 INPUT LEVEL – dB Figure 4. Differential Nonlinearity vs. Output Code Figure 7. SFDR and SNR vs. Analog Input Level –6– REV. B AD9022 0 –10 –20 FULL SCALE – dB –30 –40 –50 –60 –70 –80 –90 –100 0 FREQUENCY – MHz 10 AIN = 1.2MHz AIN = –1.0dBFS SNR = 66.7dB THD = 77.51dB SFDR = 79.49dBFS is present when the unit is strobed with an ENCODE command. The conversion process begins on the rising edge of this pulse, which should conform to the minimum and maximum pulsewidth requirements shown in the specifications. Operation below the recommended encode rate (4 MSPS) may result in excessive droop in the internal T/H devices–leading to large dc and ac errors. The held analog value of the first track-and-hold is applied to a 5-bit flash converter and a second T/H. The 5-bit flash converter resolves the most significant bits (MSBs) of the held analog voltage. These five bits are reconstructed via a 5-bit DAC and subtracted from the original T/H output signal to form a residue signal. A second T/H holds the amplified residue signal while it is encoded with a second 5-bit flash ADC. Again the five bits are reconstructed and subtracted from the second T/H output to form a residue signal. This residue is amplified and encoded with a 4-bit flash ADC to provide the three least significant bits (LSBs) of the digital output and one bit of error correction. Digital Error Correction logic aligns the data from the three flash converters and presents the result as a 12-bit parallel digital word. The output stage of the AD9022 is TTL. Output data may be strobed on the rising edge of the ENCODE command. AD9022 IN RECEIVER APPLICATIONS Figure 8. FFT Plot 0 –10 –20 FULL SCALE – dB –30 –40 –50 –60 –70 –80 –90 –100 0 FREQUENCY – MHz 10 AIN = 9.6MHz AIN = –1.0dBFS SNR = 66.05dB THD = 74.28dB SFDR = 75.32dBFS Figure 9. FFT Plot 0 AIN1 = 8.9MHz AIN2 = 9.8MHz AIN1 = 7.0dBFS AIN2 = 7.0dBFS SFDR = 80.62dBFS 20 FULL SCALE – dB 40 Advances in semiconductor processes have resulted in low cost digital signal processing (DSP) and analog signal processing which can help create cost effective alternative receiver designs. Today, an all-digital receiver allows tuning, demodulation, and detection of receiver signals in the digital domain. By digitizing IF signals directly, and utilizing digital techniques, it becomes possible to make significant improvements in receiver design. For high frequency IFs, the ADC is the key to the receiver’s performance. Unfortunately, the specifications frequently used by receiver designers and analog-to-digital (ADC) manufacturers are often very different. Noise Figure and Intercept Point are common measures of noise and linearity in analog RF system design. ADCs are more frequently specified in terms of SNR and harmonic distortion. Noise 60 80 Noise figure (NF) is a measure of receiver sensitivity and is defined as the degradation of signal-to-noise ratio (SNR) as a signal passes through a device. In equation form: NF = SNR (in) – SNR (out) Noise figure is a bandwidth invariant parameter for reasonably narrow bandwidths in most devices. The system noise figure for a combination of amplifiers and mixers, for instance, can be analyzed without regard to the information bandwidth. Thermal noise contribution from the ADC behaves in a similar fashion; however, the spectral density of quantization noise is a function of the sample rate. In addition, the spectral density of the quantization noise is flat only in an ADC with perfect linearity, i.e., perfect 1 LSB step sizes. To analyze the system noise performance, ADC noise figure is calculated by normalizing the SNR of the ADC output to a 1 Hz bandwidth. This result is given by: SNR (/Hz) = SNR + 10 log10 (FS/2) where FS is the sample rate. –7– 100 120 0.0 2.0 4.0 6.0 FREQUENCY – MHz 8.0 10.0 Figure 10. Two-Tone FFT THEORY OF OPERATION Refer to the block diagram. The AD9022 employs a three-pass subranging architecture and digital error correction. This combination of design techniques ensures 12-bit accuracy at relatively low power. Analog input signals are immediately attenuated through a resistor divider and applied directly to the sampling bridge of the track-and-hold (T/H). The T/H holds whatever analog value REV. B AD9022 This will be true only for converters in which perfect quantization noise dominates. There may be an upper sample rate, above which the thermal noise of the converter is the dominant source of noise. In this case, normalization would be based on the noise bandwidth of the ADC. For an AD9022 with a typical SNR of 64 dB and a sample rate of 20 MSPS, the normalized SNR is equal to 134 dB (64 + 70). Both thermal and quantization noise contribute to this number. The SNR of the input is assumed to be limited by the thermal noise of the input resistance, or –174 dBm/Hz. The input signal level is +10 dBm (2 V p-p into 50 Ω). Noise figure of the ADC can be calculated by: NF = SNR (in) – SNR (out) = [+10 – (174)] – 134 = 50 dB Most ADCs detect input voltage levels, not power. Consequently, the input SNR can be determined more accurately by determining the ratio of the signal voltage to the noise voltage of the terminating resistor. However, both the input signal and noise voltage delivered to the ADC are also a function of the source impedance. The dependence of NF on sample rate, linearity, source and terminating impedances, and the number of assumptions required, highlight the weakness of using NF as a figure of merit for an ADC. The rather large number that results bolsters this belief by indicating the ADC is often the weakest link in the signal processing path. Linearity AD9022 NOISE PERFORMANCE High speed, wide bandwidth ADCs such as the AD9022 are optimized for dynamic performance over a wide range of analog input frequencies. However, there are many applications (Imaging, Instrumentation, etc.) where dc precision is also important. Due to the wide input bandwidth of the AD9022 for a given input voltage, there will be a range of output codes which may occur. This is caused by unavoidable circuit noise within the wideband circuits in the ADC. If a dc signal is applied to the ADC and several thousand outputs are recorded, a distribution of codes such as that shown in the histogram below may result. 2.0 RELATIVE FREQUENCY OF OCCURRENCE 1.5 1.0 0.5 0 –0.5 –1.0 –1.5 –2.0 ONE STANDARD DEVIATION = RMS NOISE LEVEL x–3 x–2 x–1 x x+1 x+2 x+3 The Third Order intercept point for a linear device (with some nonlinearity) is a good way to predict 3rd order spurious signals as a function of input signal level. For an ADC, however, this in an invalid concept except with signals near full scale. As the input signal is reduced, the performance burden shifts from the input track-and-hold (T/H) to the encoder. This creates a nonlinear function, as contrasted with the third order intercept behavior, which predicts an improvement in dynamic range as the signal level is decreased. For signals near full scale, the intercept point is calculated the same as any device: Intercept Point = [Harmonic Suppression/(N –1)] + Input Power where N = the order of the IMD (3 in this case) AD9022 Intercept Point = 80/2 + 3 dBm (7 dBm below full scale) = 43 dBm For signals below this level, the spurious free dynamic range (SFDR) curves shown in the data sheet are a more accurate predictor of dynamic range. The SFDR curve is generated by measuring the ratio of the signal (either tone in the two-tone measurement) to the worst spurious signal, which is observed as the analog input signal amplitude is swept. The worst spurious signal is usually the second harmonic or 3rd order IMD. Actual results are shown on several plots. The straightline with a slope of one is constructed at the point where the worst SFDR touches the line. This line, extrapolated to full scale, gives the SFDR of the ADC. This value can then be used to predict the dynamic range by simply subtracting the input level from the SFDR. It should be noted that all SFDR lines are constructed to be valid only below a certain level below full scale. Above these points, the linearity of the device is dominated by the nonlinearities of the front end and best predicted by the intercept point. OUTPUT CODE Figure 11. ADC Equivalent Input Noise The correct code appears most of the time, but adjacent codes also appear with reduced probability. If a normal probability density curve is fitted to this Gaussian distribution of codes, the standard deviation will be equal to the equivalent input rms noise of the ADC. The rms noise may also be approximated by converting the SNR, as measured by a low frequency FFT, to an equivalent input noise. This method is accurate only if the SNR performance is dominated by random thermal noise (the low frequency SNR without harmonics is the best measure). Sixty-three dB equates to 1 LSB rms for a 2 V p-p (0.707 V rms) input signal. The AD9022 has approximately 0.5 LSB of rms noise or a noise limited SNR of 69 dB, indicating that noise alone does not limit the SNR performance of the device (quantization noise and linearity are also major contributors). This thermal noise may come from several sources. The drive source impedance should be kept low to minimize resistor thermal noise. Some of the internal ADC noise is generated in the wideband T/H. Sampling ADCs generally have input bandwidths which exceed the Nyquist frequency of one-half the sampling rate. (The AD9022 has an input bandwidth of over 100 MHz, even though the sampling rate is limited to 20 MSPS.) Wide bandwidth is required to minimize gain and phase distortion and to permit adequate settling times in the internal amplifiers and T/Hs. But a certain amount of unavoidable noise is generated in the T/H and other wideband circuits within the ADC; this causes variation in output codes for dc inputs. Good layout, grounding and decoupling techniques are essential to prevent external noise from coupling into the ADC and further corrupting performance. –8– REV. B AD9022 USING THE AD9022 Layout Information Preserving the accuracy and dynamic performance of the AD9022 requires that designers pay special attention to the layout of the printed circuit board. Analog paths should be kept as short as possible and be properly terminated to avoid reflections. The analog input connection should be kept away from digital signal paths; this reduces the amount of digital switching noise, which is capacitively coupled into the analog section. Digital signal paths should also be kept short, and run lengths should be matched to avoid propagation delay mismatch. The AD9022 digital outputs should be buffered or latched close to the device (