AD9200LQFP-EVAL

a FEATURES CMOS 10-Bit, 20 MSPS Sampling A/D Converter Pin-Compatible with AD876 Power Dissipation: 80 mW (3 V Supply) Operation Between 2.7 V and 5.5 V Supply Differential Nonlinearity: 0.5 LSB Power-Down (Sleep) Mode Three-State Outputs Out-of-Range Indicator Built-In Clamp Function (DC Restore) Adjustable On-Chip Voltage Reference IF Undersampling to 135 MHz PRODUCT DESCRIPTION Complete 10-Bit, 20 MSPS, 80 mW CMOS A/D Converter AD9200 A single clock input is used to control all internal conversion cycles. The digital output data is presented in straight binary output format. An out-of-range signal (OTR) indicates an overflow condition which can be used with the most significant bit to determine low or high overflow. The AD9200 can operate with supply range from 2.7 V to 5.5 V, ideally suiting it for low power operation in high speed portable applications. The AD9200 is specified over the industrial (–40°C to +85°C) and commercial (0°C to +70°C) temperature ranges. PRODUCT HIGHLIGHTS Low Power The AD9200 is a monolithic, single supply, 10-bit, 20 MSPS analog-to-digital converter with an on-chip sample-and-hold amplifier and voltage reference. The AD9200 uses a multistage differential pipeline architecture at 20 MSPS data rates and guarantees no missing codes over the full operating temperature range. The input of the AD9200 has been designed to ease the development of both imaging and communications systems. The user can select a variety of input ranges and offsets and can drive the input either single-ended or differentially. The sample-and-hold (SHA) amplifier is equally suited for both multiplexed systems that switch full-scale voltage levels in successive channels and sampling single-channel inputs at frequencies up to and beyond the Nyquist rate. AC coupled input signals can be shifted to a predetermined level, with an onboard clamp circuit (AD9200ARS, AD9200KST). The dynamic performance is excellent. The AD9200 has an onboard programmable reference. An external reference can also be chosen to suit the dc accuracy and temperature drift requirements of the application. The AD9200 consumes 80 mW on a 3 V supply (excluding the reference power). In sleep mode, power is reduced to below 5 mW. Very Small Package The AD9200 is available in both a 28-lead SSOP and 48-lead LQFP packages. Pin Compatible with AD876 The AD9200 is pin compatible with the AD876, allowing older designs to migrate to lower supply voltages. 300 MHz On-Board Sample-and-Hold The versatile SHA input can be configured for either singleended or differential inputs. Out-of-Range Indicator The OTR output bit indicates when the input signal is beyond the AD9200’s input range. Built-In Clamp Function Allows dc restoration of video signals with AD9200ARS and AD9200KST. FUNCTIONAL BLOCK DIAGRAM CLAMP CLAMP IN CLK AVDD DRVDD STBY SHA AIN REFTS REFBS REFTF REFBF VREF REFSENSE 1V OUTPUT BUFFERS OTR D9 (MSB) D0 (LSB) AVSS DRVSS A/D D/A A/D D/A A/D D/A A/D D/A SHA GAIN SHA GAIN SHA GAIN SHA GAIN A/D MODE THREESTATE CORRECTION LOGIC AD9200 REV. E Information furnished by Analog Devices is believed to be accurate and reliable. However, no responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other rights of third parties which may result from its use. No license is granted by implication or otherwise under any patent or patent rights of Analog Devices. One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A. Tel: 781/329-4700 World Wide Web Site: http://www.analog.com Fax: 781/326-8703 © Analog Devices, Inc., 1999 AD9200–SPECIFICATIONS Parameter RESOLUTION CONVERSION RATE DC ACCURACY Differential Nonlinearity Integral Nonlinearity Offset Error Gain Error REFERENCE VOLTAGES Top Reference Voltage Bottom Reference Voltage Differential Reference Voltage Reference Input Resistance1 ANALOG INPUT Input Voltage Range Input Capacitance Aperture Delay Aperture Uncertainty (Jitter) Input Bandwidth (–3 dB) Full Power (0 dB) DC Leakage Current INTERNAL REFERENCE Output Voltage (1 V Mode) Output Voltage Tolerance (1 V Mode) Output Voltage (2 V Mode) Load Regulation (1 V Mode) POWER SUPPLY Operating Voltage Supply Current Power Consumption Power-Down Gain Error Power Supply Rejection FS DNL INL EZS EFS REFTS REFBS (AVDD = +3 V, DRVDD = +3 V, FS = 20 MHz (50% Duty Cycle), MODE = AVDD, 2 V Input Span from 0.5 V to 2.5 V, External Reference, TMIN to TMAX unless otherwise noted) Typ 10 20 ± 0.5 ± 0.75 0.4 1.4 1 GND 2 10 4.2 ±1 ±2 1.2 3.5 Max Units Bits MHz LSB LSB % FSR % FSR REFTS = 2.5 V, REFBS = 0.5 V Condition Symbol Min AVDD V AVDD – 1 V V p-p kΩ kΩ REFTS V pF ns ps MHz µA V mV V mV V V mA mW mW % FS REFTS, REFBS: MODE = AVDD Between REFTF and REFBF: MODE = AVSS REFBS Min = GND: REFTS Max = AVDD Switched AIN CIN tAP tAJ BW REFBS 1 4 2 300 23 Input = ± FS REFSENSE = VREF REFSENSE = GND 1 mA Load Current VREF VREF 1 ± 10 2 0.5 3 3 26.6 80 4 1 ± 25 2 5.5 5.5 33.3 100 AVDD 2.7 DRVDD 2.7 IAVDD PD AVDD = 3 V, MODE = AVSS AVDD = DRVDD = 3 V, MODE = AVSS STBY = AVDD, MODE and CLOCK = AVSS PSRR DYNAMIC PERFORMANCE (AIN = 0.5 dBFS) Signal-to-Noise and Distortion SINAD f = 3.58 MHz f = 10 MHz Effective Bits f = 3.58 MHz f = 10 MHz Signal-to-Noise SNR f = 3.58 MHz f = 10 MHz Total Harmonic Distortion THD f = 3.58 MHz f = 10 MHz Spurious Free Dynamic Range SFDR f = 3.58 MHz f = 10 MHz Two-Tone Intermodulation Distortion IMD Differential Phase DP Differential Gain DG 54.5 57 54 9.1 8.6 dB dB Bits Bits dB dB dB dB dB dB dB Degree % f = 44.49 MHz and 45.52 MHz NTSC 40 IRE Mod Ramp 55 57 56 –66 –58 –69 –61 68 0.1 0.05 –59 –61 – 2– REV. E AD9200 Parameter DIGITAL INPUTS High Input Voltage Low Input Voltage DIGITAL OUTPUTS High-Z Leakage Data Valid Delay Data Enable Delay Data High-Z Delay LOGIC OUTPUT (with DRVDD = 3 V) High Level Output Voltage (IOH = 50 µA) High Level Output Voltage (IOH = 0.5 mA) Low Level Output Voltage (IOL = 1.6 mA) Low Level Output Voltage (IOL = 50 µA) LOGIC OUTPUT (with DRVDD = 5 V) High Level Output Voltage (IOH = 50 µA) High Level Output Voltage (IOH = 0.5 mA) Low Level Output Voltage (IOL = 1.6 mA) Low Level Output Voltage (IOL = 50 µA) CLOCKING Clock Pulsewidth High Clock Pulsewidth Low Pipeline Latency CLAMP2 Clamp Error Voltage Clamp Pulsewidth NOTES 1 See Figures 1a and 1b. 2 Available only in AD9200ARS and AD9200KST. Specifications subject to change without notice. Symbol VIH VIL IOZ tOD tDEN tDHZ VOH VOH VOL VOL VOH VOH VOL VOL tCH tCL Min 2.4 Typ Max Units V V µA ns ns ns V V V V V V V V ns ns Cycles Condition 0.3 –10 25 25 13 +2.95 +2.80 +0.4 +0.05 +4.5 +2.4 +0.4 +0.1 22.5 22.5 3 +10 Output = GND to VDD CL = 20 pF EOC tCPW ± 20 2 ± 40 mV µs CLAMPIN = 0.5 V–2.7 V, R IN = 10 Ω CIN = 1 µF (Period = 63.5 µs) 10k REFTS 10k REFTS AD9200 REFTF REFBF 0.4 VDD REFBS MODE AD9200 4.2k REFBS AVDD MODE Figure 1a. Figure 1b. REV. E –3– AD9200 ABSOLUTE MAXIMUM RATINGS* Parameter With Respect to Min –0.3 –0.3 –0.3 –6.5 –0.3 –0.3 –0.3 –0.3 –0.3 –0.3 –0.3 –0.3 –65 Max +6.5 +6.5 +0.3 +6.5 AVDD + 0.3 AVDD + 0.3 DRVDD + 0.3 AVDD + 0.3 AVDD + 0.3 AVDD + 0.3 AVDD + 0.3 AVDD + 0.3 +150 +150 +300 Units V V V V V V V V V V V V °C °C °C *Stresses above those listed under Absolute Maximum Ratings may cause permanent damage to the device. This is a stress rating only; functional operation of the device at these or any other conditions above those indicated in the operational sections of this specification is not implied. Exposure to absolute maximum ratings for extended periods may effect device reliability. AVDD AVSS DRVDD DRVSS AVSS DRVSS AVDD DRVDD MODE AVSS CLK AVSS Digital Outputs DRVSS AIN AVSS VREF AVSS REFSENSE AVSS REFTF, REFTB AVSS REFTS, REFBS AVSS Junction Temperature Storage Temperature Lead Temperature 10 sec ORDERING GUIDE Temperature Range Package Description 28-Lead SSOP 28-Lead SSOP 48-Lead LQFP 48-Lead LQFP 28-Lead SSOP (Reel) 28-Lead SSOP (Reel) 48-Lead LQFP (Reel) 48-Lead LQFP (Reel) Evaluation Board Evaluation Board Package Options* RS-28 RS-28 ST-48 ST-48 RS-28 RS-28 ST-48 ST-48 Model AD9200JRS 0°C to +70°C AD9200ARS –40°C to +85°C AD9200JST 0°C to +70°C AD9200KST 0°C to +70°C AD9200JRSRL 0°C to +70°C AD9200ARSRL –40°C to +85°C AD9200JSTRL 0°C to +70°C AD9200KSTRL 0°C to +70°C AD9200 SSOP-EVAL AD9200 LQFP-EVAL *RS = Shrink Small Outline; ST = Thin Quad Flatpack. AVDD DRVDD AVDD AVDD AVDD AVDD DRVSS DRVSS AVSS AVSS AVSS AVSS AVSS a. D0–D9, OTR b. Three-State, Standby, Clamp AVDD REFBS REFTF AVSS AVDD REFBF REFTS AVSS c. CLK AVDD AVDD AVSS AVDD AVSS AVSS d. AIN e. Reference AVDD AVDD AVDD AVDD AVSS AVSS AVSS AVSS f. CLAMPIN g. MODE h. REFSENSE i. VREF Figure 2. Equivalent Circuits CAUTION ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily accumulate on the human body and test equipment and can discharge without detection. Although the AD9200 features proprietary ESD protection circuitry, permanent damage may occur on devices subjected to high energy electrostatic discharges. Therefore, proper ESD precautions are recommended to avoid performance degradation or loss of functionality. WARNING! ESD SENSITIVE DEVICE –4– REV. E AD9200 PIN CONFIGURATIONS 28-Lead Shrink Small Outline (SSOP) AVSS 1 DRVDD 2 D0 3 D1 4 D2 5 D3 6 28 AVDD 27 AIN 26 VREF 25 REFBS D0 1 D1 2 D2 3 D3 4 D4 5 NC 6 NC 7 D5 8 D6 9 D7 10 D8 11 D9 12 NC = NO CONNECT 13 14 15 16 17 18 19 20 21 22 23 24 PIN 1 IDENTIFIER 36 35 34 33 32 48-Lead Plastic Thin Quad Flatpack (LQFP) NC DRVDD AVSS AVDD VREF AIN NC NC NC NC 48 47 46 45 44 43 42 41 40 39 38 37 NC NC NC REFBS REFBF NC MODE NC REFTF REFTS CLAMPIN CLAMP REFSENSE NC AD9200 24 REFBF 23 MODE TOP VIEW D4 7 (Not to Scale) 22 REFTF D5 8 D6 9 D7 10 D8 11 D9 12 OTR 13 DRVSS 14 21 REFTS 20 CLAMPIN 19 CLAMP 18 REFSENSE 17 STBY 16 THREE-STATE 15 CLK AD9200 TOP VIEW (Not to Scale) 31 30 29 28 27 26 25 NC NC THREE-STATE OTR NC NC NC DRVSS CLK NC NC PIN FUNCTION DESCRIPTIONS SSOP Pin No. 1 2 3 4 5 6 7 8 9 10 11 12 13 14 15 16 17 18 19 20 21 22 23 24 25 26 27 28 LQFP Pin No. 44 45 1 2 3 4 5 8 9 10 11 12 16 17 22 23 24 26 27 28 29 30 32 34 35 38 39 42 Name AVSS DRVDD D0 D1 D2 D3 D4 D5 D6 D7 D8 D9 OTR DRVSS CLK THREE-STATE STBY REFSENSE CLAMP CLAMPIN REFTS REFTF MODE REFBF REFBS VREF AIN AVDD Description Analog Ground Digital Driver Supply Bit 0, Least Significant Bit Bit 1 Bit 2 Bit 3 Bit 4 Bit 5 Bit 6 Bit 7 Bit 8 Bit 9, Most Significant Bit Out-of-Range Indicator Digital Ground Clock Input HI: High Impedance State. LO: Normal Operation HI: Power-Down Mode. LO: Normal Operation Reference Select HI: Enable Clamp Mode. LO: No Clamp Clamp Reference Input Top Reference Top Reference Decoupling Mode Select Bottom Reference Decoupling Bottom Reference Internal Reference Output Analog Input Analog Supply REV. E –5– STBY AD9200 DEFINITIONS OF SPECIFICATIONS Integral Nonlinearity (INL) Offset Error Integral nonlinearity refers to the deviation of each individual code from a line drawn from “zero” through “full scale”. The point used as “zero” occurs 1/2 LSB before the first code transition. “Full scale” is defined as a level 1 1/2 LSB beyond the last code transition. The deviation is measured from the center of each particular code to the true straight line. Differential Nonlinearity (DNL, No Missing Codes) The first transition should occur at a level 1 LSB above “zero.” Offset is defined as the deviation of the actual first code transition from that point. Gain Error An ideal ADC exhibits code transitions that are exactly 1 LSB apart. DNL is the deviation from this ideal value. It is often specified in terms of the resolution for which no missing codes (NMC) are guaranteed. The first code transition should occur for an analog value 1 LSB above nominal negative full scale. The last transition should occur for an analog value 1 LSB below the nominal positive full scale. Gain error is the deviation of the actual difference between first and last code transitions and the ideal difference between the first and last code transitions. Pipeline Delay (Latency) The number of clock cycles between conversion initiation and the associated output data being made available. New output data is provided every rising edge. Typical Characterization Curves Span from 0.5 V to 2.5 V, External Reference, unless otherwise noted) 1.0 (AVDD = +3 V, DRVDD = +3 V, FS = 20 MHz (50% Duty Cycle), MODE = AVDD, 2 V Input 60 55 –0.5 AMPLITUDE –6.0 AMPLITUDE 0.5 50 45 SNR– dB DNL 0 40 35 30 25 –20.0 AMPLITUDE –0.5 –1.0 0 128 256 384 512 640 CODE OFFSET 768 896 1024 20 1.00E+05 1.00E+06 1.00E+07 INPUT FREQUENCY – Hz 1.00E+08 Figure 3. Typical DNL Figure 5. SNR vs. Input Frequency 1.0 60 –0.5 AMPLITUDE 55 0.5 50 45 –6.0 AMPLITUDE SINAD – dB INL 0 40 35 30 25 –20.0 AMPLITUDE –0.5 –1.0 0 128 256 384 512 640 CODE OFFSET 768 896 1024 20 1.00E+05 1.00E+06 1.00E+07 INPUT FREQUENCY – Hz 1.00E+08 Figure 4. Typical INL Figure 6. SINAD vs. Input Frequency –6– REV. E AD9200 –30 CLOCK = 20MHz –35 80.0 –20.0 AMPLITUDE POWER CONSUMPTION – mW 1.00E+08 –40 –45 THD – dB 80.5 79.5 79.0 –50 –55 –60 –65 –70 –75 –80 1.00E+05 –0.5 AMPLITUDE –6.0 AMPLITUDE 78.5 78.0 77.5 77.0 1.00E+06 1.00E+07 INPUT FREQUENCY – Hz 0 2 4 6 8 10 12 14 CLOCK FREQUENCY – MHz 16 18 20 Figure 7. THD vs. Input Frequency Figure 10. Power Consumption vs. Clock Frequency (MODE = AVSS) 1M –70 –60 –50 FIN = 1MHz 900k 800k 700k 600k HITS THD – dB –40 500k 400k 300k 200k 499856 –30 –20 –10 100k 0 100E+03 10E+06 1E+06 CLOCK FREQUENCY – Hz 100E+06 54383 N–1 N CODE 54160 N+1 0 Figure 8. THD vs. Clock Frequency Figure 11. Grounded Input Histogram 1.005 1.004 1.003 1.002 20 0 CLOCK = 20MHz –20 –40 VREF – V –60 1.001 –80 1.000 0.999 0.998 –40 –100 –120 –140 0E+0 1E+6 2E+6 3E+6 4E+6 5E+6 6E+6 7E+6 8E+6 9E+6 10E+6 SINGLE TONE FREQUENCY DOMAIN –20 0 20 40 TEMPERATURE – °C 60 80 100 Figure 9. Voltage Reference Error vs. Temperature Figure 12. Single-Tone Frequency Domain REV. E –7– AD9200 0 –3 –6 –9 –12 –15 –18 –21 –24 –27 1.0E+6 APPLYING THE AD9200 THEORY OF OPERATION The AD9200 implements a pipelined multistage architecture to achieve high sample rate with low power. The AD9200 distributes the conversion over several smaller A/D subblocks, refining the conversion with progressively higher accuracy as it passes the results from stage to stage. As a consequence of the distributed conversion, the AD9200 requires a small fraction of the 1023 comparators used in a traditional flash type A/D. A sample-and-hold function within each of the stages permits the first stage to operate on a new input sample while the second, third and fourth stages operate on the three preceding samples. 100.0E+6 10.0E+6 FREQUENCY – Hz 1.0E+9 SIGNAL AMPLITUDE – dB OPERATIONAL MODES Figure 13. Full Power Bandwidth 25 20 15 10 5 IB – A REFBS = 0.5V REFTS = 2.5V CLOCK = 20MHz The AD9200 is designed to allow optimal performance in a wide variety of imaging, communications and instrumentation applications, including pin compatibility with the AD876 A/D. To realize this flexibility, internal switches on the AD9200 are used to reconfigure the circuit into different modes. These modes are selected by appropriate pin strapping. There are three parts of the circuit affected by this modality: the voltage reference, the reference buffer, and the analog input. The nature of the application will determine which mode is appropriate: the descriptions in the following sections, as well as the Table I should assist in picking the desired mode. 0 –5 –10 –15 –20 –25 0 0.5 1.0 2.0 1.5 INPUT VOLTAGE – V 2.5 3.0 Figure 14. Input Bias Current vs. Input Voltage Table I. Mode Selection Modes TOP/BOTTOM Input Connect AIN AIN CENTER SPAN AIN AIN Differential AIN Is Input 1 Input Span 1V 2V 1V 2V 1V MODE Pin AVDD AVDD REFSENSE Pin REF REFTS REFBS AGND AGND AVDD/2 AVDD/2 AVDD/2 29 Figure 18 19 20 Short REFSENSE, REFTS and VREF Together AGND Short REFTS and VREF Together AVDD/2 AVDD/2 AVDD/2 AVDD/2 Short VREF and REFSENSE Together AVDD/2 AGND No Connect AVDD/2 Short VREF and REFSENSE Together REFTS and REFBS Are Shorted Together for Input 2 2V External Ref AIN AVDD/2 AGND AVDD No Connect No Connect AVDD/2 AVDD/2 21, 22 23 30 2 V max AVDD AGND Span = REFTS – REFBS (2 V max) Short to VREFTF Short to VREFBF Short to VREFBF AD876 AIN 2V Float or AVSS AVDD No Connect Short to VREFTF –8– REV. E AD9200 SUMMARY OF MODES VOLTAGE REFERENCE AIN REFTS 1 V Mode the internal reference may be set to 1 V by connecting REFSENSE and VREF together. 2 V Mode the internal reference my be set to 2 V by connecting REFSENSE to analog ground External Divider Mode the internal reference may be set to a point between 1 V and 2 V by adding external resistors. See Figure 16f. External Reference Mode enables the user to apply an external reference to REFTS, REFBS and VREF pins. This mode is attained by tying REFSENSE to VDD. REFERENCE BUFFER SHA A/D CORE AD9200 REFBS Figure 15. AD9200 Equivalent Functional Input Circuit In single-ended operation, the input spans the range, REFBS ≤ AIN ≤ REFTS where REFBS can be connected to GND and REFTS connected to VREF. If the user requires a different reference range, REFBS and REFTS can be driven to any voltage within the power supply rails, so long as the difference between the two is between 1 V and 2 V. In differential operation, REFTS and REFBS are shorted together, and the input span is set by VREF, (REFTS – VREF/2) ≤ AIN ≤ (REFTS + VREF/2) where VREF is determined by the internal reference or brought in externally by the user. The best noise performance may be obtained by operating the AD9200 with a 2 V input range. The best distortion performance may be obtained by operating the AD9200 with a 1 V input range. REFERENCE OPERATION Center Span Mode midscale is set by shorting REFTS and REFBS together and applying the midscale voltage to that point The MODE pin is set to AVDD/2. The analog input will swing about that midscale point. Top/Bottom Mode sets the input range between two points. The two points are between 1 V and 2 V apart. The Top/Bottom Mode is enabled by tying the MODE pin to AVDD. ANALOG INPUT Differential Mode is attained by driving the AIN pin as one differential input and shorting REFTS and REFBS together and driving them as the second differential input. The MODE pin is tied to AVDD/2. Preferred mode for optimal distortion performance. Single-Ended is attained by driving the AIN pin while the REFTS and REFBS pins are held at dc points. The MODE pin is tied to AVDD. Single-Ended/Clamped (AC Coupled) the input may be clamped to some dc level by ac coupling the input. This is done by tying the CLAMPIN to some dc point and applying a pulse to the CLAMP pin. MODE pin is tied to AVDD. SPECIAL AD876 Mode enables users of the AD876 to drop the AD9200 into their socket. This mode is attained by floating or grounding the MODE pin. INPUT AND REFERENCE OVERVIEW The AD9200 can be configured in a variety of reference topologies. The simplest configuration is to use the AD9200’s onboard bandgap reference, which provides a pin-strappable option to generate either a 1 V or 2 V output. If the user desires a reference voltage other than those two, an external resistor divider can be connected between VREF, REFSENSE and analog ground to generate a potential anywhere between 1 V and 2 V. Another alternative is to use an external reference for designs requiring enhanced accuracy and/or drift performance. A third alternative is to bring in top and bottom references, bypassing VREF altogether. Figures 16d, 16e and 16f illustrate the reference and input architecture of the AD9200. In tailoring a desired arrangement, the user can select an input configuration to match drive circuit. Then, moving to the reference modes at the bottom of the figure, select a reference circuit to accommodate the offset and amplitude of a full-scale signal. Table I outlines pin configurations to match user requirements. Figure 16, a simplified model of the AD9200, highlights the relationship between the analog input, AIN, and the reference voltages, REFTS, REFBS and VREF. Like the voltages applied to the resistor ladder in a flash A/D converter, REFTS and REFBS define the maximum and minimum input voltages to the A/D. The input stage is normally configured for single-ended operation, but allows for differential operation by shorting REFTS and REFBS together to be used as the second input. REV. E –9– AD9200 V* +FS –FS AIN SHA 10k 10k A2 10k –F/S RANGE OBTAINED FROM VREF PIN OR EXTERNAL REF 10k REFBF A/D CORE 4.2k TOTAL 0.1 F 10 F MIDSCALE AD9200 AIN AD9200 SHA 10k 10k MODE MODE (AVDD) REFTF 0.1 F AVDD/2 +F/S RANGE OBTAINED FROM VREF PIN OR EXTERNAL REF REFTS REFBS REFTF 0.1 F REFTS REFBS 10k INTERNAL REF MIDSCALE OFFSET VOLTAGE IS DERIVED FROM INTERNAL OR EXTERNAL REF A2 A/D CORE 4.2k TOTAL 0.1 F 10 F 10k REFBF 0.1 F 0.1 F * MAXIMUM MAGNITUDE OF V IS DETERMINED BY INTERNAL REFERENCE a. Top/Bottom Mode b. Center Span Mode MAXIMUM MAGNITUDE OF V IS DETERMINED BY INTERNAL REFERENCE AND TURNS RATIO V AIN SHA AVDD/2 10k 10k REFTS REFBS 10k INTERNAL 10k REF REFBF A2 A/D CORE 4.2k TOTAL 0.1 F 10 F REFTF 0.1 F AD9200 MODE AVDD/2 0.1 F c. Differential Mode VREF (2V) A1 1V VREF (1V) 1V 0.1 F REFSENSE 1.0 F A1 10k 0.01 F REFSENSE 1.0 F AD9200 AVSS AD9200 10k AVSS d. 1 V Reference e. 2 V Reference A1 1V VREF (= 1 + RA/RB) RA REFSENSE RB 0.1 F 1.0 F A1 1V VREF REFSENSE AVDD AD9200 INTERNAL 10K REF RESISTORS ARE SWITCHED OPEN BY THE PRESENSE OF RA AND RB. AVSS AD9200 f. Variable Reference (Between 1 V and 2 V) Figure 16. g. Internal Reference Disable (Power Reduction) –10– REV. E AD9200 The actual reference voltages used by the internal circuitry of the AD9200 appear on REFTF and REFBF. For proper operation, it is necessary to add a capacitor network to decouple these pins. The REFTF and REFBF should be decoupled for all internal and external configurations as shown in Figure 17. Figure 19 shows the single-ended configuration for 2 V p-p operation. REFSENSE is connected to GND, resulting in a 2 V reference output. 2V 0V REFTF 10 F 0.1 F AIN SHA 10k REFTS REFBS 10k 10k 10k A2 A/D CORE AD9200 MODE AVDD 0.1 F REFTF AD9200 REFBF 0.1 F 0.1 F 4.2k TOTAL 0.1 F 10 F 0.1 F REFBF Figure 17. Reference Decoupling Network Note: REFTF = reference top, force REFBF = reference bottom, force REFTS = reference top, sense REFBS = reference bottom, sense INTERNAL REFERENCE OPERATION VREF 1.0 F 0.1 F REF SENSE A1 1V Figures 18, 19 and 20 show example hookups of the AD9200 internal reference in its most common configurations. (Figures 18 and 19 illustrate top/bottom mode while Figure 20 illustrates center span mode). Figure 29 shows how to connect the AD9200 for 1 V p-p differential operation. Shorting the VREF pin directly to the REFSENSE pin places the internal reference amplifier, A1, in unity-gain mode and the resultant reference output is 1 V. In Figure 18 REFBS is grounded to give an input range from 0 V to 1 V. These modes can be chosen when the supply is either +3 V or +5 V. The VREF pin must be bypassed to AVSS (analog ground) with a 1.0 µF tantalum capacitor in parallel with a low inductance, low ESR, 0.1 µF ceramic capacitor. 1V 0V 10k REFTS REFBS 10k 10k REFBF VREF 1.0 F 0.1 F REF SENSE A1 1V 10k A2 AIN SHA Figure 19. Internal Reference, 2 V p-p Input Span (Top/Bottom Mode) Figure 20 shows the single-ended configuration that gives the good high frequency dynamic performance (SINAD, SFDR). To optimize dynamic performance, center the common-mode voltage of the analog input at approximately 1.5 V. Connect the shorted REFTS and REFBS inputs to a low impedance 1.5 V source. In this configuration, the MODE pin is driven to a voltage at midsupply (AVDD/2). Maximum reference drive is 1 mA. An external buffer is required for heavier loads. AD9200 MODE AVDD 0.1 F 2V 1V AIN SHA 10k REFTS 10k A2 AD9200 MODE AVDD/2 0.1 F REFTF REFTF A/D CORE 4.2k TOTAL 0.1 F 10 F +1.5V REFBS A/D CORE 4.2k TOTAL 0.1 F 10 F 0.1 F 10k 10k REFBF VREF 1.0 F 0.1 F REF SENSE A1 1V 0.1 F Figure 18. Internal Reference 1 V p-p Input Span (Top/Bottom Mode) Figure 20. Internal Reference 1 V p-p Input Span, (Center Span Mode) REV. E –11– AD9200 EXTERNAL REFERENCE OPERATION 4V VIN 2V REFTS 4V 10 F 2V 0.1 F 0.1 F 0.1 F REFTF Using an external reference may provide more flexibility and improve drift and accuracy. Figures 21 through 23 show examples of how to use an external reference with the AD9200. To use an external reference, the user must disable the internal reference amplifier by connecting the REFSENSE pin to VDD. The user then has the option of driving the VREF pin, or driving the REFTS and REFBS pins. The AD9200 contains an internal reference buffer (A2), that simplifies the drive requirements of an external reference. The external reference must simply be able to drive a 10 kΩ load. Figure 21 shows an example of the user driving the top and bottom references. REFTS is connected to a low impedance 2 V source and REFBS is connected to a low impedance 1 V source. REFTS and REFBS may be driven to any voltage within the supply as long as the difference between them is between 1 V and 2 V. 2V 1V 10k REFTS REFBS 10k REF SENSE MODE 10k REFBF 10k A2 A/D CORE 4.2k TOTAL 0.1 F AIN SHA REFTF 0.1 F AD9200 REFBF REFBS VREF AVDD REFSENSE MODE Figure 23a. External Reference—2 V p-p Input Span REFTS +5V 6 C4 0.1 F 7 C3 0.1 F C2 10 F C6 0.1 F REFTF AD9200 REFT 5 8 AD9200 REFBS 2V 1V 2 C5 0.1 F 6 REFBF C1 0.1 F 4 10 F REFB 3 AVDD 0.1 F Figure 23b. Kelvin Connected Reference Using the AD9200 Figure 21. External Reference Mode—1 V p-p Input Span STANDBY OPERATION Figure 22 shows an example of an external reference generating 2.5 V at the shorted REFTS and REFBS inputs. In this instance, a REF43 2.5 V reference drives REFTS and REFBS. A resistive divider generates a 1 V VREF signal that is buffered by A3. A3 must be able to drive a 10 kΩ, capacitive load. Choose this op amp based on noise and accuracy requirements. AD9200 AIN REFTS REFBS 10 F 1.5k A3 1.0 F 0.1 F 1k +5V AVDD 0.1 F 0.1 F MODE REFBF 0.1 F VREF REFTF 0.1 F 10 F 0.1 F AVDD AVDD The ADC may be placed into a powered down (sleep) mode by driving the STBY (standby) pin to logic high potential and holding the clock at logic low. In this mode the typical power drain is approximately 4 mW. If there is no connection to the STBY pin, an internal pull-down circuit will keep the ADC in a “wake-up” mode of operation. The ADC will “wake up” in 400 ns (typ) after the standby pulse goes low. 3.0V 2.5V 2.0V CLAMP OPERATION REFSENSE REF43 0.1 F AVDD/2 Figure 22. External Reference Mode—1 V p-p Input Span 2.5 VCM The AD9200ARS and AD9200KST parts feature an optional clamp circuit for dc restoration of video or ac coupled signals. Figure 24 shows the internal clamp circuitry and the external control signals needed for clamp operation. To enable the clamp, apply a logic high to the CLAMP pin. This will close the switch SW1. The clamp amplifier will then servo the voltage at the AIN pin to be equal to the clamp voltage applied at the CLAMPIN pin. After the desired clamp level is attained, SW1 is opened by taking CLAMP back to a logic low. Ignoring the droop caused by the input bias current, the input capacitor CIN will hold the dc voltage at AIN constant until the next clamp interval. The input resistor RIN has a minimum recommended value of 10 Ω, to maintain the closed-loop stability of the clamp amplifier. The allowable voltage range that can be applied to CLAMPIN depends on the operational limits of the internal clamp amplifier. When operating off of 3 volt supplies, the recommended clamp range is between 0.5 volts and 2.0 volts. Figure 23a shows an example of the external references driving the REFTF and REFBF pins that is compatible with the AD876. REFTS is shorted to REFTF and driven by an external 4 V low impedance source. REFBS is shorted to REFBF and driven by a 2 V source. The MODE pin is connected to GND in this configuration. –12– REV. E AD9200 The input capacitor should be sized to allow sufficient acquisition time of the clamp voltage at AIN within the CLAMP interval, but also be sized to minimize droop between clamping intervals. Specifically, the acquisition time when the switch is closed will equal: V  T ACQ = RIN CIN ln  C   VE  back porch to truncate the SYNC below the AD9200’s minimum input voltage. With a CIN = 1 µF, and RIN = 20 Ω, the acquisition time needed to set the input dc level to one volt with 1 mV accuracy is about 140 µs, assuming a full 1 volt VC. With a 1 µF input coupling capacitor, the droop across one horizontal can be calculated: IBIAS = 10 µA, and t = 63.5 µs, so dV = 0.635 mV, which is less than one LSB. After the input capacitor is initially charged, the clamp pulsewidth only needs to be wide enough to correct small voltage errors such as the droop. The fine scale settling characteristics of the clamp circuitry are shown in Table II. Depending on the required accuracy, a CLAMP pulsewidth of 1 µs–3 µs should work in most applications. The OFFSET values ignore the contribution of offset from the clamp amplifier; they simply compare the output code with a “final value” measured with a much longer CLAMP pulse duration. Table II. where VC is the voltage change required across CIN, and VE is the error voltage. VC is calculated by taking the difference between the initial input dc level at the start of the clamp interval and the clamp voltage supplied at CLAMPIN. VE is a systemdependent parameter, and equals the maximum tolerable deviation from VC. For example, if a 2-volt input level needs to be clamped to 1 volt at the AD9200’s input within 10 millivolts, then VC equals 2 – 1 or 1 volt, and VE equals 10 mV. Note that once the proper clamp level is attained at the input, only a very small voltage change will be required to correct for droop. The voltage droop is calculated with the following equation: I dV = BIAS t CIN () CLAMP 10 µs 5 µs 4 µs 3 µs 2 µs 1 µs OFFSET