8-Bit, 210 MSPS TxDAC® D/A Converter
AD9748
FEATURES
APPLICATIONS
High performance member of pin-compatible
TxDAC product family
Linearity
0.1 LSB DNL
0.1 LSB INL
Twos complement or straight binary data format
Differential current outputs: 2 mA to 20 mA
Power dissipation: 135 mW @ 3.3 V
Power-down mode: 15 mW @ 3.3 V
On-chip 1.20 V reference
CMOS-compatible digital interface
32-lead LFCSP
Edge-triggered latches
Fast settling: 11 ns to 0.1% full-scale
Communications
Direct digital synthesis (DSS)
Instrumentation
FUNCTIONAL BLOCK DIAGRAM
3.3V
RSET
3.3V
AVDD
150pF
1.2V REF
REFIO
FS ADJ
CURRENT
SOURCE
ARRAY
AD9748
DVDD
DCOM
SEGMENTED
SWITCHES
CLK+
CLK–
ACOM
LSB
SWITCHES
IOUTA
IOUTB
LATCHES
3.3V
MODE
CMODE
CLKVDD
CLKCOM
SLEEP
DIGITAL DATA INPUTS (DB7–DB0)
03211-001
0.1μF
Figure 1.
GENERAL DESCRIPTION
The AD9748 1 is an 8-bit resolution, wideband, third generation
member of the TxDAC series of high performance, low power
CMOS digital-to-analog converters (DACs). The TxDAC
family, consisting of pin-compatible 8-, 10-, 12-, and 14-bit
DACs, is specifically optimized for the transmit signal path of
communication systems. All of the devices share the same
interface options, small outline package, and pinout, providing
an upward or downward component selection path based on
performance, resolution, and cost. The AD9748 offers
exceptional ac and dc performance while supporting update
rates up to 210 MSPS.
The AD9748’s low power dissipation makes it well suited for
portable and low power applications. Its power dissipation can
be further reduced to 60 mW with a slight degradation in
performance by lowering the full-scale current output. In
addition, a power-down mode reduces the standby power
dissipation to approximately 15 mW. A segmented current
source architecture is combined with a proprietary switching
technique to reduce spurious components and enhance
dynamic performance.
Edge-triggered input latches and a 1.2 V temperaturecompensated band gap reference have been integrated to
provide a complete monolithic DAC solution. The digital inputs
support 3 V CMOS logic families.
PRODUCT HIGHLIGHTS
1.
2.
3.
4.
5.
6.
1
32-lead LFCSP.
The AD9748 is the 8-bit member of the pin-compatible
TxDAC family, which offers excellent INL and DNL
performance.
Differential or single-ended clock input (LVPECL or
CMOS), supports 210 MSPS conversion rate.
Data input supports twos complement or straight binary
data coding.
Low power: Complete CMOS DAC function operates on
135 mW from a 2.7 V to 3.6 V single supply. The DAC fullscale current can be reduced for lower power operation,
and a sleep mode is provided for low power idle periods.
On-chip voltage reference: The AD9748 includes a 1.2 V
temperature-compensated band gap voltage reference.
Protected by U.S. Patent Numbers 5568145, 5689257, and 5703519.
Rev. A
Information furnished by Analog Devices is believed to be accurate and reliable. However, no
responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other
rights of third parties that may result from its use. Specifications subject to change without notice. No
license is granted by implication or otherwise under any patent or patent rights of Analog Devices.
Trademarks and registered trademarks are the property of their respective owners.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 781.329.4700
www.analog.com
Fax: 781.461.3113
© 2005 Analog Devices, Inc. All rights reserved.
AD9748
TABLE OF CONTENTS
Features .............................................................................................. 1
Applying the AD9748 ................................................................ 15
Applications....................................................................................... 1
Differential Coupling Using a Transformer............................... 15
Functional Block Diagram .............................................................. 1
Differential Coupling Using an Op Amp................................ 16
General Description ......................................................................... 1
Single-Ended, Unbuffered Voltage Output............................. 16
Product Highlights ........................................................................... 1
Single-Ended, Buffered Voltage Output Configuration........ 16
Revision History ............................................................................... 2
Power and Grounding Considerations, Power Supply
Rejection...................................................................................... 17
Specifications..................................................................................... 3
DC Specifications ......................................................................... 3
Dynamic Specifications ............................................................... 4
Digital Specifications ................................................................... 5
Absolute Maximum Ratings............................................................ 6
Thermal Characteristics .............................................................. 6
ESD Caution.................................................................................. 6
Pin Configuration and Function Descriptions............................. 7
Terminology ...................................................................................... 8
Typical Performance Characteristics ............................................. 9
Functional Description .................................................................. 11
Reference Operation .................................................................. 11
Reference Control Amplifier .................................................... 11
DAC Transfer Function ............................................................. 12
Analog Outputs........................................................................... 12
Digital Inputs .............................................................................. 13
Clock Input.................................................................................. 13
DAC Timing................................................................................ 14
Evaluation Board ............................................................................ 18
General Description................................................................... 18
Outline Dimensions ....................................................................... 24
Ordering Guide .......................................................................... 24
REVISION HISTORY
12/05—Rev. 0 to Rev. A
Updated Format..................................................................Universal
Changes to General Description and Product Highlights ...........1
Changes to Table 1.............................................................................3
Changes to Table 2.............................................................................4
Changes to Table 4.............................................................................6
Inserted Figure 7; Renumbered Sequentially.................................9
Changes to Figure 8 and Figure 9....................................................9
Changes to Functional Description, Reference Operation, and
Reference Control Amplifier Sections......................................... 11
Inserted Figure 16; Renumbered Sequentially ........................... 11
Changes to DAC Transfer Function Section............................... 12
Changes to Digital Inputs Section................................................ 13
Changes to Figure 22, Figure 23, and Figure 24 ......................... 14
Changes to Figure 25...................................................................... 15
Changes to Figure 26, Figure 27, Figure 28, and Figure 29....... 16
Updated Outline Dimensions....................................................... 24
Changes to Ordering Guide .......................................................... 24
2/03—Revision 0: Initial Version
Power Dissipation....................................................................... 14
Rev. A | Page 2 of 24
AD9748
SPECIFICATIONS
DC SPECIFICATIONS
TMIN to TMAX, AVDD = 3.3 V, DVDD = 3.3 V, CLKVDD = 3.3 V, IOUTFS = 20 mA, unless otherwise noted.
Table 1.
Parameter
RESOLUTION
DC ACCURACY 1
Integral Linearity Error (INL)
Differential Nonlinearity (DNL)
ANALOG OUTPUT
Offset Error
Gain Error (Without Internal Reference)
Gain Error (With Internal Reference)
Full-Scale Output Current 2
Output Compliance Range
Output Resistance
Output Capacitance
REFERENCE OUTPUT
Reference Voltage
Reference Output Current 3
REFERENCE INPUT
Input Compliance Range
Reference Input Resistance (External Reference)
Small Signal Bandwidth
TEMPERATURE COEFFICIENTS
Offset Drift
Gain Drift (Without Internal Reference)
Gain Drift (With Internal Reference)
Reference Voltage Drift
POWER SUPPLY
Supply Voltages
AVDD
DVDD
CLKVDD
Analog Supply Current (IAVDD)
Digital Supply Current (IDVDD) 4
Clock Supply Current (ICLKVDD)
Supply Current Sleep Mode (IAVDD)
Power Dissipation4
Power Dissipation 5
Power Supply Rejection Ratio—AVDD 6
Power Supply Rejection Ratio—DVDD6
OPERATING RANGE
Min
8
Typ
Max
Unit
Bits
±0.25
±0.25
±0.1
±0.1
+0.25
+0.25
LSB
LSB
+0.02
+0.5
+0.5
20.0
+1.25
% of FSR
% of FSR
% of FSR
mA
V
kΩ
pF
1.26
V
nA
1.25
7
0.5
V
kΩ
MHz
0
±50
±100
±50
ppm of FSR/°C
ppm of FSR/°C
ppm of FSR/°C
ppm/°C
−0.02
−0.5
−0.5
2.0
−1.0
±0.1
±0.1
100
5
1.14
1.20
100
0.1
2.7
2.7
2.7
−1
−0.04
−40
1
3.3
3.3
3.3
33
8
5
5
135
145
3.6
3.6
3.6
36
9
7
6
145
+1
+0.04
+85
Measured at IOUTA, driving a virtual ground.
Nominal full-scale current, IOUTFS, is 32 times the IREF current.
3
An external buffer amplifier with input bias current 100 kΩ).
Figure 18. External Reference Configuration
REFERENCE CONTROL AMPLIFIER
The AD9748 contains a control amplifier that is used to regulate
the full-scale output current, IOUTFS. The control amplifier is
configured as a V-I converter, as shown in Figure 17, so that its
current output, IREF, is determined by the ratio of the VREFIO and
an external resistor, RSET, as stated in Equation 4. IREF is copied
to the segmented current sources with the proper scale factor to
set IOUTFS, as stated in Equation 3.
Rev. A | Page 11 of 24
AD9748
The control amplifier allows a wide (10:1) adjustment span of
IOUTFS over a 2 mA to 20 mA range by setting IREF between 62.5
μA and 625 μA. The wide adjustment span of IOUTFS provides
several benefits. The first relates directly to the power
dissipation of the AD9748, which is proportional to IOUTFS (see
the Power Dissipation section). The second relates to a 20 dB
adjustment, which is useful for system gain control purposes.
The small signal bandwidth of the reference control amplifier is
approximately 500 kHz and can be used for low frequency small
signal multiplying applications.
DAC TRANSFER FUNCTION
The AD9748 provides complementary current outputs, IOUTA
and IOUTB. IOUTA provides a near full-scale current output,
IOUTFS, when all bits are high (that is, DAC CODE = 255), while
IOUTB, the complementary output, provides no current. The
current output appearing at IOUTA and IOUTB is a function of
both the input code and IOUTFS and can be expressed as:
Substituting the values of IOUTA, IOUTB, IREF, and VDIFF can be
expressed as:
VDIFF = {(2 × DAC CODE − 255)/256}
(32 × RLOAD/RSET) × VREFIO
(8)
Equation 7 and Equation 8 highlight some of the advantages of
operating the AD9748 differentially. First, the differential
operation helps cancel common-mode error sources associated
with IOUTA and IOUTB, such as noise, distortion, and dc
offsets. Second, the differential code dependent current and
subsequent voltage, VDIFF, is twice the value of the single-ended
voltage output (that is, VOUTA or VOUTB), thus providing twice the
signal power to the load.
Note that the gain drift temperature performance for a singleended (VOUTA and VOUTB) or differential output (VDIFF) of the
AD9748 can be enhanced by selecting temperature tracking
resistors for RLOAD and RSET due to their ratiometric relationship,
as shown in Equation 8.
B
IOUTA = (DAC CODE/256) × IOUTFS
(1)
ANALOG OUTPUTS
IOUTB = (255 − DAC CODE)/256 × IOUTFS
(2)
The complementary current outputs in each DAC, IOUTA,
and IOUTB can be configured for single-ended or differential
operation. IOUTA and IOUTB can be converted into
complementary single-ended voltage outputs, VOUTA and VOUTB,
via a load resistor, RLOAD, as described in the DAC Transfer
Function section by Equation 5 through Equation 8. The
differential voltage, VDIFF, existing between VOUTA and VOUTB, can
also be converted to a single-ended voltage via a transformer or
differential amplifier configuration. The ac performance
of the AD9748 is optimum and specified using a differential
transformer-coupled output in which the voltage swing at
IOUTA and IOUTB is limited to ±0.5 V.
where DAC CODE = 0 to 255 (that is, decimal representation).
As mentioned previously, IOUTFS is a function of the reference
current IREF, which is nominally set by a reference voltage,
VREFIO, and external resistor, RSET. It can be expressed as:
IOUTFS = 32 × IREF
(3)
where
IREF = VREFIO/RSET
(4)
The two current outputs typically drive a resistive load directly
or via a transformer. If dc coupling is required, then IOUTA
and IOUTB should be directly connected to matching resistive
loads, RLOAD, that are tied to analog common, ACOM. Note
that RLOAD can represent the equivalent load resistance seen by
IOUTA or IOUTB as would be the case in a doubly terminated
50 Ω or 75 Ω cable. The single-ended voltage output appearing
at the IOUTA and IOUTB nodes is simply
VOUTA = IOUTA × RLOAD
(5)
VOUTB = IOUTB × RLOAD
(6)
Note that the full-scale value of VOUTA and VOUTB should not
exceed the specified output compliance range to maintain
specified distortion and linearity performance.
VDIFF = (IOUTA − IOUTB) × RLOAD
(7)
The distortion and noise performance of the AD9748 can be
enhanced when it is configured for differential operation. The
common-mode error sources of both IOUTA and IOUTB can
be significantly reduced by the common-mode rejection of a
transformer or differential amplifier. These common-mode
error sources include even-order distortion products and noise.
The enhancement in distortion performance becomes more
significant as the frequency content of the reconstructed
waveform increases and/or its amplitude decreases. This is due
to the first-order cancellation of various dynamic commonmode distortion mechanisms, digital feedthrough, and noise.
Performing a differential-to-single-ended conversion via a
transformer also provides the ability to deliver twice the
reconstructed signal power to the load (assuming no source
termination). Because the output currents of IOUTA and
IOUTB are complementary, they become additive when
processed differentially. A properly selected transformer allows
the AD9748 to provide the required power and voltage levels to
different loads.
Rev. A | Page 12 of 24
AD9748
The output impedance of IOUTA and IOUTB is determined
by the equivalent parallel combination of the PMOS switches
associated with the current sources and is typically 100 kΩ in
parallel with 5 pF. It is also slightly dependent on the output
voltage (that is, VOUTA and VOUTB) due to the nature of a PMOS
device. As a result, maintaining IOUTA and/or IOUTB at a
virtual ground via an I-V op amp configuration results in the
optimum dc linearity. Note that the INL/DNL specifications for
the AD9748 are measured with IOUTA maintained at a virtual
ground via an op amp.
IOUTA and IOUTB also have a negative and positive voltage
compliance range that must be adhered to in order to achieve
optimum performance. The negative output compliance range
of −1 V is set by the breakdown limits of the CMOS process.
Operation beyond this maximum limit can result in a breakdown
of the output stage and affect the reliability of the AD9748.
The positive output compliance range is slightly dependent on
the full-scale output current, IOUTFS. It degrades slightly from its
nominal 1.2 V for an IOUTFS = 20 mA to 1 V for an IOUTFS = 2 mA.
The optimum distortion performance for a single-ended or
differential output is achieved when the maximum full-scale
signal at IOUTA and IOUTB does not exceed 0.5 V.
DIGITAL INPUTS
The AD9748 digital section consists of eight input bit channels
and a clock input. The 8-bit parallel data inputs follow standard
positive binary coding, where DB7 is the most significant bit
(MSB) and DB0 is the least significant bit (LSB). IOUTA
produces a full-scale output current when all data bits are at
Logic 1. IOUTB produces a complementary output with the
full-scale current split between the two outputs as a function of
the input code.
DVDD
CLOCK INPUT
A configurable clock input allows for one single-ended and two
differential modes. The mode selection is controlled by the
CMODE input, as summarized in Table 6. Connecting CMODE
to CLKCOM selects the single-ended clock input. In this mode,
the CLK+ input is driven with rail-to-rail swings and the CLK−
input is left floating. If CMODE is connected to CLKVDD, then
the differential receiver mode is selected. In this mode, both
inputs are high impedance. The final mode is selected by
floating CMODE. This mode is also differential, but internal
terminations for positive emitter-coupled logic (PECL) are
activated. There is no significant performance difference
between any of the three clock input modes.
Table 6. Clock Mode Selection
CMODE Pin
CLKCOM
CLKVDD
Float
Clock Input Mode
Single-ended
Differential
PECL
In the single-ended input mode, the CLK+ pin must be driven
with rail-to-rail CMOS levels. The quality of the DAC output is
directly related to the clock quality, and jitter is a key concern.
Any noise or jitter in the clock translates directly into the DAC
output. Optimal performance is achieved if the clock input has
a sharp rising edge, because the DAC latches are positive edge
triggered.
In the differential input mode, the clock input functions as a
high impedance differential pair. The common-mode level of
the CLK+ and CLK− inputs can vary from 0.75 V to 2.25 V, and
the differential voltage can be as low as 0.5 V p-p. This mode
can be used to drive the clock with a differential sine wave,
because the high gain bandwidth of the differential inputs
convert the sine wave into a single-ended square wave
internally.
The final clock mode allows for a reduced external component
count when the DAC clock is distributed on the board using
PECL logic. The internal termination configuration is shown in
Figure 20. These termination resistors are untrimmed and can
vary up to ±20%. However, matching between the resistors
should generally be better than ±1%.
Figure 19. Equivalent Digital Input
The digital interface is implemented using an edge-triggered
master/slave latch. The DAC output updates on the rising edge
of the clock and is designed to support a clock rate as high as
210 MSPS. The clock can be operated at any duty cycle that
meets the specified latch pulse width. The setup and hold times
can also be varied within the clock cycle as long as the specified
minimum times are met, although the location of these
transition edges can affect digital feedthrough and distortion
performance. Best performance is typically achieved when the
input data transitions on the falling edge of a 50% duty cycle clock.
Rev. A | Page 13 of 24
AD9748
CLK+
CLOCK
RECEIVER
CLK–
50Ω
TO DAC CORE
50Ω
VTT = 1.3V NOM
Figure 20. Clock Termination in PECL Mode
03211-017
03211-016
DIGITAL
INPUT
AD9748
Input Clock and Data Timing Relationship
Dynamic performance in a DAC is dependent on the
relationship between the position of the clock edges and the
time at which the input data changes. The AD9748 is rising
edge triggered, and so exhibits dynamic performance sensitivity
when the data transition is close to this edge. In general, the
goal when applying the AD9748 is to make the data transition
close to the falling clock edge. This becomes more important as
the sample rate increases. Figure 21 shows the relationship of
SFDR to clock placement with different sample rates. Note that
at the lower sample rates, more tolerance is allowed in clock
placement, while at higher rates, more care must be taken.
Conversely, IDVDD is dependent on both the digital input
waveform, fCLOCK, and digital supply DVDD. Figure 23 shows IDVDD
as a function of full-scale sine wave output ratios (fOUT/fCLOCK)
for various update rates with DVDD = 3.3 V.
35
30
25
IAVDD (mA)
DAC TIMING
20
15
80
10
75
0
70
2
4
6
SFDR (dB)
65
10
12
IOUTFS (mA)
14
16
18
20
Figure 22. IAVDD vs. IOUTFS
60
55
20
50MHz SFDR
50
18
210MSPS
45
16
40
14
2
4
6
8
CLOCK PLACEMENT (ns)
10
12
03211-018
0
IDVDD (mA)
35
30
8
03211-019
20MHz SFDR
Figure 21. SFDR vs. Clock Placement @ fOUT = 20 MHz and 50 MHz
(fCLOCK = 165 MSPS)
165MSPS
12
10
125MSPS
8
6
65MSPS
4
Sleep Mode Operation
The power dissipation, PD, of the AD9748 is dependent on
several factors that include the:
0.1
RATIO (fOUT/fCLOCK)
1
Figure 23. IDVDD vs. Ratio @ DVDD = 3.3 V
11
10
9
DIFF
8
ICLKVDD (mA)
POWER DISSIPATION
7
6
PECL
5
SE
4
3
Power supply voltages (AVDD, CLKVDD, and DVDD)
Full-scale current output (IOUTFS)
Update rate (fCLOCK)
Reconstructed digital input waveform
2
1
0
The power dissipation is directly proportional to the analog
supply current, IAVDD, and the digital supply current, IDVDD. IAVDD
is directly proportional to IOUTFS, as shown in Figure 22, and is
insensitive to fCLOCK.
Rev. A | Page 14 of 24
0
50
100
150
200
fCLOCK (MSPS)
Figure 24. ICLKVDD vs. fCLOCK and Clock Mode
250
03211-042
•
•
•
•
0
0.01
03211-041
2
The AD9748 has a power-down function that turns off the output
current and reduces the supply current to less than 6 mA over the
specified supply range of 2.7 V to 3.6 V and the temperature range.
This mode can be activated by applying a Logic Level 1 to the
SLEEP pin. The SLEEP pin logic threshold is equal to 0.5 Ω
AVDD. This digital input also contains an active pull-down
circuit that ensures that the AD9748 remains enabled if this
input is left disconnected. The AD9748 takes less than 50 ns to
power down and approximately 5 μs to power back up.
AD9748
APPLYING THE AD9748
DIFFERENTIAL COUPLING USING A TRANSFORMER
Output Configurations
An RF transformer can be used to perform a differential-tosingle-ended signal conversion, as shown in Figure 25. A
differentially coupled transformer output provides the optimum
distortion performance for output signals whose spectral
content lies within the transformer’s pass band. An RF transformer,
such as the Mini-Circuits® T1–1T, provides excellent rejection of
common-mode distortion (that is, even-order harmonics) and
noise over a wide frequency range. It also provides electrical
isolation and the ability to deliver twice the power to the load.
Transformers with different impedance ratios can also be used
for impedance matching purposes. Note that the transformer
provides ac coupling only.
A single-ended output is suitable for applications requiring a
unipolar voltage output. A positive unipolar output voltage
results if IOUTA and/or IOUTB is connected to an appropriately
sized load resistor, RLOAD, referred to ACOM. This configuration
can be more suitable for a single-supply system requiring a dccoupled, ground referred output voltage. Alternatively, an
amplifier could be configured as an I-V converter, thus
converting IOUTA or IOUTB into a negative unipolar voltage.
This configuration provides the best dc linearity because
IOUTA or IOUTB is maintained at a virtual ground.
IOUTA
MINI-CIRCUITS
T1-1T
AD9748
RLOAD
IOUTB
OPTIONAL RDIFF
03211-022
The following sections illustrate some typical output
configurations for the AD9748. Unless otherwise noted, it is
assumed that IOUTFS is set to a nominal 20 mA. For applications
requiring the optimum dynamic performance, a differential
output configuration is suggested. A differential output
configuration can consist of either an RF transformer or a
differential op amp configuration. The transformer configuration
provides the optimum high frequency performance and is
recommended for any application that allows ac coupling. The
differential op amp configuration is suitable for applications
requiring dc coupling, bipolar output, signal gain, and/or
level shifting within the bandwidth of the chosen op amp.
Figure 25. Differential Output Using a Transformer
The center tap on the primary side of the transformer must be
connected to ACOM to provide the necessary dc current path
for both IOUTA and IOUTB. The complementary voltages
appearing at IOUTA and IOUTB (that is, VOUTA and VOUTB)
swing symmetrically around ACOM and should be maintained
with the specified output compliance range of the AD9748. A
differential resistor, RDIFF, can be inserted in applications where
the output of the transformer is connected to the load, RLOAD,
via a passive reconstruction filter or cable. RDIFF is determined
by the transformer’s impedance ratio and provides the proper
source termination that results in a low VSWR. Note that
approximately half the signal power is dissipated across RDIFF.
B
Rev. A | Page 15 of 24
AD9748
DIFFERENTIAL COUPLING USING AN OP AMP
SINGLE-ENDED, UNBUFFERED VOLTAGE OUTPUT
An op amp can also be used to perform a differential-to-singleended conversion, as shown in Figure 26. The AD9748 is
configured with two equal load resistors, RLOAD, of 25 Ω. The
differential voltage developed across IOUTA and IOUTB is
converted to a single-ended signal via the differential op amp
configuration. An optional capacitor can be installed across
IOUTA and IOUTB, forming a real pole in a low-pass filter. The
addition of this capacitor also enhances the op amp’s distortion
performance by preventing the DAC’s high slewing output from
overloading the op amp’s input.
Figure 28 shows the AD9748 configured to provide a unipolar
output range of approximately 0 V to 0.5 V for a doubly
terminated 50 Ω cable because the nominal full-scale current,
IOUTFS, of 20 mA flows through the equivalent RLOAD of 25 Ω.
In this case, RLOAD represents the equivalent load resistance seen
by IOUTA or IOUTB. The unused output (IOUTA or IOUTB)
can be connected to ACOM directly or via a matching RLOAD.
Different values of IOUTFS and RLOAD can be selected as long as
the positive compliance range is adhered to. One additional
consideration in this mode is the integral nonlinearity (INL),
discussed in the Analog Outputs section. For optimum INL
performance, the single-ended, buffered voltage output
configuration is suggested.
500Ω
AD9748
225Ω
IOUTA
AD8047
225Ω
AD9748
COPT
IOUTFS = 20mA
500Ω
50Ω
25Ω
03211-023
25Ω
VOUTA = 0V TO 0.5V
IOUTA
50Ω
IOUTB
03211-025
IOUTB
25Ω
Figure 26. DC Differential Coupling Using an Op Amp
Figure 28. 0 V to 0.5 V Unbuffered Voltage Output
The common-mode rejection of this configuration is typically
determined by the resistor matching. In this circuit, the differential
op amp circuit using the AD8047 is configured to provide some
additional signal gain. The op amp must operate off a dual
supply because its output is approximately ±1 V. A high speed
amplifier capable of preserving the differential performance
of the AD9748 while meeting other system level objectives (that
is, cost or power) should be selected. The op amp’s differential
gain, gain setting resistor values, and full-scale output swing
capabilities should all be considered when optimizing this circuit.
The differential circuit shown in Figure 27 provides the
necessary level shifting required in a single-supply system. In
this case, AVDD, which is the positive analog supply for both
the AD9748 and the op amp, is also used to level shift the
differential output of the AD9748 to midsupply (that is,
AVDD/2). The AD8041 is a suitable op amp for this application.
500Ω
Figure 29 shows a buffered single-ended output configuration
in which the op amp U1 performs an I-V conversion on the
AD9748 output current. U1 maintains IOUTA (or IOUTB) at a
virtual ground, minimizing the nonlinear output impedance
effect on the DAC’s INL performance as described in the
Analog Outputs section. Although this single-ended
configuration typically provides the best dc linearity
performance, its ac distortion performance at higher DAC
update rates can be limited by U1’s slew rate capabilities. U1
provides a negative unipolar output voltage, and its full-scale
output voltage is simply the product of RFB and IOUTFS. The fullscale output should be set within U1’s voltage output swing
capabilities by scaling IOUTFS and/or RFB. An improvement in ac
distortion performance can result with a reduced IOUTFS because
U1 is required to sink less signal current.
225Ω
COPT
AD8041
225Ω
IOUTB
COPT
25Ω
1kΩ
25Ω
1kΩ
RFB
200Ω
AD9748
AVDD
03211-024
IOUTA
IOUTFS = 10mA
IOUTA
U1
VOUT = IOUTFS × RFB
IOUTB
Figure 27. Single-Supply DC Differential Coupled Circuit
200Ω
Figure 29. Unipolar Buffered Voltage Output
Rev. A | Page 16 of 24
03211-026
AD9748
SINGLE-ENDED, BUFFERED VOLTAGE OUTPUT
CONFIGURATION
AD9748
POWER AND GROUNDING CONSIDERATIONS,
POWER SUPPLY REJECTION
Many applications seek high speed and high performance under
less than ideal operating conditions. In these application circuits,
the implementation and construction of the printed circuit
board is as important as the circuit design. Proper RF techniques
must be used for device selection, placement, and routing as
well as power supply bypassing and grounding to ensure
optimum performance. Figure 35 to Figure 38 illustrate the
recommended printed circuit board ground, power, and signal
plane layouts implemented on the AD9748 evaluation board.
One factor that can measurably affect system performance is
the ability of the DAC output to reject dc variations or ac noise
superimposed on the analog or digital dc power distribution.
This is referred to as the power supply rejection ratio (PSRR).
For dc variations of the power supply, the resulting performance
of the DAC directly corresponds to a gain error associated with
the DAC’s full-scale current, IOUTFS. AC noise on the dc supplies
is common in applications where the power distribution is
generated by a switching power supply. Typically, switching
power supply noise occurs over the spectrum from tens of
kilohertz to several megahertz. The PSRR vs. frequency of the
AD9748 AVDD supply over this frequency range is shown in
Figure 30.
85
Note that the ratio in Figure 30 is calculated as amps out/volts
in. Noise on the analog power supply has the effect of modulating
the internal switches, and therefore the output current. The
voltage noise on AVDD, therefore, is added in a nonlinear
manner to the desired IOUT. Due to the relative different size of
these switches, the PSRR is very code dependent. This can produce
a mixing effect that can modulate low frequency power supply
noise to higher frequencies. Worst-case PSRR for either one of
the differential DAC outputs occurs when the full-scale current
is directed toward that output. As a result, the PSRR measurement
in Figure 30 represents a worst-case condition in which the
digital inputs remain static and the full-scale output current of
20 mA is directed to the DAC output being measured.
The following illustrates the effect of supply noise on the analog
supply. Suppose a switching regulator with a switching
frequency of 250 kHz produces 10 mV of noise and, for simplicity’s
sake (ignoring harmonics), all of this noise is concentrated at
250 kHz. To calculate how much of this undesired noise appears as
current noise superimposed on the DAC’s full-scale current,
IOUTFS, users must determine the PSRR in dB using Figure 30 at
250 kHz. To calculate the PSRR for a given RLOAD, such that the
units of PSRR are converted from A/V to V/V, adjust the curve
in Figure 30 by the scaling factor 20 Ω log (RLOAD). For instance,
if RLOAD is 50 Ω, then the PSRR is reduced by 34 dB (that is,
PSRR of the DAC at 250 kHz, which is 85 dB in Figure 30,
becomes 51 dB VOUT/VIN).
80
Proper grounding and decoupling should be a primary
objective in any high speed, high resolution system. The
AD9748 features separate analog and digital supplies and
ground pins to optimize the management of analog and digital
ground currents in a system. In general, AVDD, the analog
supply, should be decoupled to ACOM, the analog common,
as close to the chip as physically possible. Similarly, DVDD, the
digital supply, should be decoupled to DCOM as close to the
chip as physically possible.
75
65
60
55
50
45
0
2
4
6
8
FREQUENCY (MHz)
10
Figure 30. Power Supply Rejection Ratio (PSRR)
12
For those applications that require a single 3.3 V supply for both
the analog and digital supplies, a clean analog supply can be
generated using the circuit shown in Figure 31. The circuit
consists of a differential LC filter with separate power supply
and return lines. Lower noise can be attained by using low ESR
type electrolytic and tantalum capacitors.
FERRITE
BEADS
TTL/CMOS
LOGIC
CIRCUITS
AVDD
100μF
ELECT.
10μF–22μF
TANT.
0.1μF
CER.
ACOM
3.3V
POWER SUPPLY
Figure 31. Differential LC Filter for Single 3.3 V Applications
Rev. A | Page 17 of 24
03211-028
40
03211-027
PSRR (dB)
70
AD9748
EVALUATION BOARD
GENERAL DESCRIPTION
The AD9748 evaluation boards allow for easy setup and testing
of the product in the LFCSP package. Careful attention to layout
and circuit design, combined with a prototyping area, allows the
user to evaluate the AD9748 easily and effectively in any
application where high resolution, high speed conversion is
required.
This board allows the user the flexibility to operate the AD9748
in various configurations. Possible output configurations
include transformer coupled, resistor terminated, and single
and differential outputs. The digital inputs are designed to be
driven from various word generators, with the on-board option
to add a resistor network for proper load termination. Provisions
are also made to exercise the power-down feature of the
AD9748 and select the clock and data modes.
Rev. A | Page 18 of 24
AD9748
RED
TP12
TB1
C3
0.1μF
TB1
CVDD
1
BLK
C2
10μF
6.3V
TP2
2
C10
0.1μF
2
4
1
3
6
5
8
7
DB10X
10
9
DB9X
11
DB8X
13
DB7X
15
DB6X
17
DB5X
19
DB4X
21
DB3X
23
DB2X
25
DB1X
27
DB0X
12
L2 BEAD
TB3
16
DVDD
1
C7
0.1μF
TB3
14
RED
TP13
18
20
BLK
C6
0.1μF
C4
10μF
6.3V
TP4
2
22
24
26
28
RED
TP5
L3 BEAD
C9
0.1μF
TB4
32
AVDD
1
BLK
36
C8
0.1μF
C5
10μF
6.3V
TP6
34
38
40
2
DB13X
DB12X
DB11X
29
31
33
35
JP3
CKEXTX
37
39
J1
R3
100Ω
R4
100Ω
R15
100Ω
R16
100Ω
R17
100Ω
R18
100Ω
R19
100Ω
DB13X
DB12X
DB11X
DB10X
DB9X
DB8X
DB7X
DB6X
DB5X
DB4X
DB3X
DB2X
DB1X
DB0X
CKEXTX
R21
100Ω
R24
100Ω
R25
100Ω
R26
100Ω
R27
100Ω
R20
100Ω
1 RP3
22Ω 16
2 RP3
22Ω 15
3 RP3
22Ω 14
4 RP3
22Ω 13
5 RP3
22Ω 12
6 RP3
7 RP3
22Ω 11
22Ω 10
8 RP3
22Ω 9
1 RP4
22Ω 16
2 RP4
22Ω 15
3 RP4
22Ω 14
4 RP4
22Ω 13
5 RP4
22Ω 12
6 RP4
7 RP4
22Ω 11
22Ω 10
8 RP4
22Ω 9
DB13
DB12
DB11
DB10
DB9
DB8
DB7
DB6
DB5
DB4
DB3
DB2
DB1
DB0
CKEXT
R28
100Ω
03211-029
TB4
30
HEADER STRAIGHT UP MALE NO SHROUD
L1 BEAD
Figure 32. Evaluation Board Schematic—Power Supply and Digital Inputs
Rev. A | Page 19 of 24
AD9748
AVDD
DVDD
CVDD
C19
0.1μF
C17
0.1μF
C32
0.1μF
SLEEP
TP11
WHT
R29
10kΩ
DB7
DB6
DVDD
DB5
DB4
DB3
DB2
DB1
DB0
CVDD
CLK
CLKB
CMODE
1
2
3
4
5
6
7
8
9
10
11
12
13
14
15
16
DB7
DB6
DVDD
DB5
DB4
DB3
DB2
DB1
DB0
DCOM
U1
CVDD
CLK
CLKB
CCOM
CMODE
MODE
DB8
DB9
DB10
DB11
DB12
DB13
DCOM1
SLEEP
FS ADJ
REFIO
ACOM
IA
IB
ACOM1
AVDD
AVDD1
32
31
30
29
28
27
26
25
DB8
DB9
DB10
DB11
DB12
DB13
R11
50Ω
DNP
C13
24
23
22
TP3
TP1
WHT
WHT
JP8
IOUT
3
21
20
19
18
17
TP7
R30
10Ω
S3
AGND: 3, 4, 5
5
2
6
1
AVDD
T1 – 1T
C11
0.1μF
JP9
AD9748LFCSP
WHT
4
T1
DNP
C12
R10
50Ω
CVDD
R1
2kΩ
0.1%
JP1
03211-030
MODE
Figure 33. Evaluation Board Schematic—Output Signal Conditioning
CVDD
1
7
U4
C20
10μF
16V
2
AGND: 5
CVDD: 8
C35
0.1μF
CVDD
R5
120Ω
3
JP2
CKEXT
CLK
4
U4
6
AGND: 5
CVDD: 8
R2
120Ω
C34
0.1μF
S5
AGND: 3, 4, 5
R6
50Ω
03211-031
CLKB
Figure 34. Evaluation Board Schematic—Clock Input
Rev. A | Page 20 of 24
03211-032
AD9748
03211-033
Figure 35. Evaluation Board Layout—Primary Side
Figure 36. Evaluation Board Layout—Secondary Side
Rev. A | Page 21 of 24
03211-034
AD9748
03211-035
Figure 37. Evaluation Board Layout—Ground Plane
Figure 38. Evaluation Board Layout—Power Plane
Rev. A | Page 22 of 24
03211-036
AD9748
03211-037
Figure 39. Evaluation Board Layout Assembly—Primary Side
Figure 40. Evaluation Board Layout Assembly—Secondary Side
Rev. A | Page 23 of 24
AD9748
OUTLINE DIMENSIONS
0.60 MAX
5.00
BSC SQ
0.60 MAX
PIN 1
INDICATOR
TOP
VIEW
0.50
BSC
4.75
BSC SQ
0.50
0.40
0.30
32
1
3.25
3.10 SQ
2.95
EXPOSED
PAD
(BOTTOM VIEW)
17
16
9
8
0.25 MIN
3.50 REF
0.80 MAX
0.65 TYP
12° MAX
1.00
0.85
0.80
PIN 1
INDICATOR
25
24
0.05 MAX
0.02 NOM
SEATING
PLANE
0.30
0.23
0.18
0.20 REF
COPLANARITY
0.08
COMPLIANT TO JEDEC STANDARDS MO-220-VHHD-2
Figure 41. 32-Lead Lead Frame Chip Scale Package [LFCSP_VQ]
5 mm × 5 mm Body, Very Thin Quad
(CP-32-2)
Dimensions shown in millimeters
ORDERING GUIDE
Model
AD9748ACP
AD9748ACPRL7
AD9748ACPZ 1
AD9748ACPZRL71
AD9748ACP-PCB
1
Temperature Range
−40°C to +85°C
−40°C to +85°C
−40°C to +85°C
−40°C to +85°C
Package Description
32-Lead LFCSP_VQ
32-Lead LFCSP_VQ
32-Lead LFCSP_VQ
32-Lead LFCSP_VQ
Evaluation Board
Z = Pb-free part.
© 2005 Analog Devices, Inc. All rights reserved. Trademarks and
registered trademarks are the property of their respective owners.
C03211–0–12/05(A)
Rev. A | Page 24 of 24
Package Option
CP-32-2
CP-32-2
CP-32-2
CP-32-2
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