Fractional-N/Integer-N PLL Synthesizer
ADF4151
Data Sheet
FEATURES
GENERAL DESCRIPTION
Fractional-N synthesizer and integer-N synthesizer
RF bandwidth to 3.5 GHz
3.0 V to 3.6 V power supply
1.8 V logic compatibility
Separate charge pump supply (VP) allows extended tuning
voltage (up to 5.5 V) in 3 V systems
Programmable dual-modulus prescaler of 4/5 or 8/9
Programmable RF output phase
3-wire serial interface
Analog and digital lock detect
Switched bandwidth fast lock mode
Cycle slip reduction
The ADF4151 allows implementation of fractional-N or
integer-N phase-locked loop (PLL) frequency synthesizers
if used with an external voltage controlled oscillator (VCO),
loop filter, and external reference frequency.
The ADF4151 is used with external VCO parts and is footprint
and software compatible with the ADF4350. The part consists
of a low noise digital phase frequency detector (PFD), a precision
charge pump, and a programmable reference divider. There is
a Σ-Δ based fractional interpolator to allow programmable
fractional-N division. The INT, FRAC, and MOD registers
define an overall N divider [N = (INT + (FRAC/MOD))]. The
RF output phase is programmable for applications that require
a particular phase relationship between the output and the
reference. The ADF4151 also features cycle slip reduction
circuitry, leading to faster lock times without the need for
modifications to the loop filter.
APPLICATIONS
Wireless infrastructure (W-CDMA, TD-SCDMA, WiMax, GSM,
PCS, DCS, DECT)
Test equipment
Wireless LANs, CATV equipment
Clock generation
Control of all the on-chip registers is through a simple 3-wire
interface. The device operates with a power supply ranging
from 3.0 V to 3.6 V that can be powered down when not in use.
The ADF4151 is available in a 5 mm × 5 mm package.
FUNCTIONAL BLOCK DIAGRAM
SDVDD
REFIN
CLK
DATA
LE
×2
DOUBLER
AVDDx
10-BIT R
COUNTER
DVDD
VP
RSET
MULTIPLEXER
÷2
DIVIDER
MUXOUT
LOCK
DETECT
FLO SWITCH
SW
LD
DATA REGISTER
FUNCTION
LATCH
CHARGE
PUMP
CPOUT
PHASE
COMPARATOR
INTEGER
REG
FRACTION
REG
MODULUS
REG
THIRD-ORDER
FRACTIONAL
INTERPOLATOR
RFIN+
N COUNTER
ADF4151
CE
AGND
CPGND
SDGND
DGND
10265-001
RFIN–
Figure 1.
Rev. B
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responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other
rights of third parties that may result from its use. Specifications subject to change without notice. No
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Fax: 781.461.3113
©2011 Analog Devices, Inc. All rights reserved.
ADF4151
Data Sheet
TABLE OF CONTENTS
Features .............................................................................................. 1
Register 1 ..................................................................................... 17
Applications ....................................................................................... 1
Register 2 ..................................................................................... 17
General Description ......................................................................... 1
Register 3 ..................................................................................... 19
Functional Block Diagram .............................................................. 1
Register 4 ..................................................................................... 19
Revision History ............................................................................... 2
Register 5 ..................................................................................... 19
Specifications..................................................................................... 3
Initialization Sequence .............................................................. 19
Timing Characteristics ................................................................ 5
RF Synthesizer—A Worked Example ...................................... 20
Absolute Maximum Ratings ............................................................ 6
Modulus ....................................................................................... 20
Transistor Count ........................................................................... 6
Reference Doubler and Reference Divider ............................. 20
ESD Caution .................................................................................. 6
12-Bit Programmable Modulus ................................................ 20
Pin Configuration and Function Descriptions ............................. 7
Cycle Slip Reduction for Faster Lock Times ........................... 21
Typical Performance Characteristics ............................................. 9
Spurious Optimization and Fast lock ...................................... 21
Circuit Description ......................................................................... 11
Fast Lock Timer and Register Sequences ................................ 21
Reference Input Section ............................................................. 11
Fast Lock—An Example ............................................................ 22
RF N Divider ............................................................................... 11
Fast Lock—Loop Filter Topology............................................. 22
INT, FRAC, MOD, and R Counter Relationship.................... 11
Spur Mechanisms ....................................................................... 22
INT N Mode ................................................................................ 11
Spur Consistency and Fractional Spur Optimization ........... 23
R Counter .................................................................................... 11
Phase Resync ............................................................................... 23
Phase Frequency Detector (PFD) and Charge Pump ............ 11
Applications Information .............................................................. 24
MUXOUT and Lock Detect ...................................................... 12
Direct Conversion Modulator .................................................. 24
Input Shift Registers ................................................................... 12
Interfacing ................................................................................... 25
Program Modes .......................................................................... 12
PCB Design Guidelines for Chip Scale Package .................... 25
Register Maps .............................................................................. 13
Outline Dimensions ....................................................................... 26
Register 0 ..................................................................................... 17
Ordering Guide .......................................................................... 26
REVISION HISTORY
12/11—Rev. A to Rev. B
Changes to Normalized 1/f Noise Parameter, Table 1 ................. 4
11/11—Rev. 0 to Rev. A
Changes to Figure 28 ...................................................................... 23
10/11—Revision 0: Initial Version
Rev. B | Page 2 of 28
Data Sheet
ADF4151
SPECIFICATIONS
AVDD = DVDD = SDVDD = 3.3 V ± 10%; VP = AVDD to 5.5 V; AGND = DGND = 0 V; TA = TMIN to TMAX, unless otherwise noted. Operating
temperature range is −40°C to +85°C.
Table 1.
Parameter
REFIN CHARACTERISTICS
Input Frequency
Input Sensitivity
Input Capacitance
Input Current
RF INPUT CHARACTERISTICS
RF Input Frequency (RFIN)
Prescaler Output Frequency
MAXIMUM PFD FREQUENCY
Fractional-N Mode
Low Spur Mode
Low Noise Mode
Integer-N Mode
CHARGE PUMP
ICP Sink/Source
High Value
Low Value
RSET Range
ICP Leakage
Sink and Source Matching
ICP vs. VCP
ICP vs. Temperature
LOGIC INPUTS
Input High Voltage, VINH
Input Low Voltage, VINL
Input Current, IINH/IINL
Input Capacitance, CIN
LOGIC OUTPUTS
Output High Voltage, VOH
Output High Current, IOH
Output Low Voltage, VO
POWER SUPPLIES
AVDD
DVDD, SDVDD
VP
DIDD + AIDD 2
VPIDD2
Low Power Sleep Mode
Min
B Version
Typ
10
0.7
Max
Unit
Conditions/Comments
250
AVDD
For f < 10 MHz, ensure slew rate > 21 V/µs
Biased at AVDD/2 1
±60
MHz
V p-p
pF
µA
3.5
750
GHz
MHz
26
32
32
MHz
MHz
MHz
10
0.5
For lower frequencies, ensure slew rate > 400 V/µs
−10 dBm ≤ RF input power ≤ +5 dBm
RSET = 5.1 kΩ
4.5
0.281
2.7
10
1
2
1.5
2
1.5
0.6
±1
5.0
DVDD − 0.4
3.0
mA
mA
kΩ
nA
%
%
%
VCP = VP/2
0.5 V ≤ VCP ≤ VP − 0.5 V
0.5 V ≤ VCP ≤ VP − 0.5 V
VCP = VP/2
V
V
µA
pF
500
0.4
V
µA
V
3.6
V
5.5
50
V
mA
mA
µA
CMOS output chosen
IOL = 500 µA
AVDD
AVDD
40
2
1
Rev. B | Page 3 of 28
VP = 5 V
ADF4151
Parameter
NOISE CHARACTERISTICS
Normalized In-Band Phase Noise
Floor (PNSYNTH) 3
Normalized 1/f Noise (PN1_f) 4
Normalized In-Band Phase Noise
Floor (PNSYNTH)3
Normalized 1/f Noise (PN1_f)4
Spurious Signals Due to PFD
Frequency 5
Data Sheet
Min
B Version
Typ
Max
Unit
Conditions/Comments
−221
dBc/Hz
PLL loop BW = 500 kHz (ABP = 3 ns)
−118
−220
dBc/Hz
dBc/Hz
−115
dBc/Hz
−107
dBc
10 kHz offset. Normalized to 1 GHz (ABP = 3 ns)
PLL loop BW = 500 kHz (ABP = 6 ns);
low noise mode
10 kHz offset; normalized to 1 GHz (ABP = 6 ns);
low noise mode
PFD = 25 MHz
AC coupling ensures AVDD/2 bias.
TA = 25°C; AVDD = DVDD = 3.6 V; prescaler = 4/5; fREFIN = 130 MHz; fPFD = 26 MHz; fRF = 1.742 GHz.
3
The synthesizer phase noise floor is estimated by measuring the in-band phase noise at the output of the VCO and subtracting 20 log N (where N is the N divider
value) and 10 log FPFD. PNSYNTH = PNTOT – 10 log fPFD – 20 log N
4
The PLL phase noise is composed of 1/f (flicker) noise plus the normalized PLL noise floor. The formula for calculating the 1/f noise contribution at an RF frequency (fRF)
and at a frequency offset (f) is given by PN = P1_f + 10 log(10 kHz/f) + 20 log(fRF/1 GHz). Both the normalized phase noise floor and flicker noise are modeled in ADIsimPLL
5
Spurious measured on EVAL-ADF4151EB1Z with RF buffer between VCO output and RF input by-passed, using a Rohde & Schwarz FSUP signal source analyzer.
1
2
Rev. B | Page 4 of 28
Data Sheet
ADF4151
TIMING CHARACTERISTICS
AVDD1, AVDD2 = DVDD = SDVDD = 3.3 V ± 10%; VP = AVDD to 5.5 V; AGND = DGND = 0 V; TA = TMIN to TMAX, unless otherwise noted.
Operating temperature range is −40°C to +85°C.
Table 2.
Parameter
t1
t2
t3
t4
t5
t6
t7
Limit (B Version)
20
10
10
25
25
10
20
Unit
ns min
ns min
ns min
ns min
ns min
ns min
ns min
t4
Test Conditions/Comments
LE setup time
DATA to CLK setup time
DATA to CLK hold time
CLK high duration
CLK low duration
CLK to LE setup time
LE pulse width
t5
CLK
t3
t2
DATA
DB31 (MSB)
DB30
DB2 (LSB)
(CONTROL BIT C3)
DB1 (LSB)
(CONTROL BIT C2)
DB0 (LSB)
(CONTROL BIT C1)
t7
LE
t1
10265-002
t6
LE
Figure 2. Timing Diagram
Rev. B | Page 5 of 28
ADF4151
Data Sheet
ABSOLUTE MAXIMUM RATINGS
TA = 25°C, unless otherwise noted.
Table 3.
Parameter
AVDD1, AVDD2 to GND1
AVDD1, AVDD2 to DVDD
VP to AVDD1, AVDD2
Digital I/O Voltage to GND1
Analog I/O Voltage to GND1
REFIN to GND1
Operating Temperature Range
Storage Temperature Range
Maximum Junction Temperature
LFCSP θJA Thermal Impedance
(Paddle-Soldered)
Reflow Soldering
Peak Temperature
Time at Peak Temperature
1
Rating
−0.3 V to +3.9 V
−0.3 V to +0.3 V
−0.3 V to +5.8 V
−0.3 V to VDD + 0.3 V
−0.3 V to VDD + 0.3 V
−0.3 V to VDD + 0.3 V
−40°C to +85°C
−65°C to +125°C
150°C
Stresses above those listed under Absolute Maximum Ratings
may cause permanent damage to the device. This is a stress
rating only; functional operation of the device at these or any
other conditions above those indicated in the operational
section of this specification is not implied. Exposure to absolute
maximum rating conditions for extended periods may affect
device reliability.
TRANSISTOR COUNT
36685 (CMOS) and 967 (bipolar)
ESD CAUTION
27.3°C/W
260°C
40 sec
GND = AGND = DGND = 0 V.
Rev. B | Page 6 of 28
Data Sheet
ADF4151
32
31
30
29
28
27
26
25
SDVDD
SDGND
MUXOUT
REFIN
DVDD
DGND
DGND
LD
PIN CONFIGURATION AND FUNCTION DESCRIPTIONS
24
1
2
PIN 1
INDICATOR
3
23
22
4
ADF4151
21
5
TOP VIEW
(Not to Scale)
20
6
19
7
18
8
17
NC
NC
RSET
AGND
NC
NC
AGND
AVDD2
NOTES
1. NC = NO CONNECT. DO NOT CONNECT TO THIS PIN.
2. THE LFCSP HAS AN EXPOSED PADDLE THAT MUST
BE CONNECTED TO GND.
10265-003
AGND
AVDD1
AGND
NC
NC
RFIN+
RFIN−
AVDD2
9
10
11
12
13
14
15
16
CLK
DATA
LE
CE
SW
VP
CPOUT
CPGND
Figure 3. Pin Configuration
Table 4. Pin Function Descriptions
Pin No.
1
Mnemonic
CLK
2
DATA
3
LE
4
CE
5
6
SW
VP
7
CPOUT
8
9, 11, 18,
21
10
CPGND
AGND
12, 13, 19,
20, 23, 24
14
15
NC
16, 17
AVDD2
AVDD1
RFIN+
RFIN−
Description
Serial Clock Input. Data is clocked into the 32-bit shift register on the CLK rising edge. This input is a high
impedance CMOS input.
Serial Data Input. The serial data is loaded, MSB first, with the three LSBs as the control bits. This input is a high
impedance CMOS input.
Load Enable, CMOS Input. When LE goes high, the data stored in the shift register is loaded into the register
that is selected by the three LSBs.
Chip Enable. A logic low on this pin powers down the device and puts the charge pump into three-state
mode. Taking the pin high powers up the device depending on the status of the power-down bits.
Fast Lock Switch. Make a connection to this pin from the loop filter when using the fast lock mode.
Charge Pump Power Supply. This pin should be greater than or equal to AVDD. In systems where AVDDx is 3 V, it
can be set to 5.5 V and used to drive a VCO with a tuning range of up to 5.5 V.
Charge Pump Output. When enabled, this provides ±ICP to the external loop filter. The output of the loop filter
is connected to VTUNE to drive the external VCO.
Charge Pump Ground. This is the ground return pin for CPOUT.
Analog Ground. This is a ground return pin for AVDD1 and AVDD2.
Analog Power Supply. This pin ranges from 3.0 V to 3.6 V. Decoupling capacitors to the analog ground plane
are to be placed as close as possible to this pin. AVDD must have the same value as DVDD.
No connect. Do not connect to this pin.
Input to the RF Input. This small signal input is ac-coupled to the external VCO.
Complementary Input to the RF Input. This pin must be decoupled to the ground plane with a small bypass
capacitor, typically 100 pF.
Analog Power Supply. This pin ranges from 3.0 V to 3.6 V. Decoupling capacitors to the analog ground plane
are to be placed as close as possible to this pin. AVDDx must have the same value as DVDD.
Rev. B | Page 7 of 28
ADF4151
Pin No.
22
Mnemonic
RSET
Data Sheet
Description
Connecting a resistor between this pin and GND sets the charge pump output current. The nominal voltage
bias at the RSET pin is 0.49 V. The relationship between ICP and RSET is
I CP =
25
LD
26, 27
28
DGND
DVDD
29
REFIN
30
MUXOUT
31
32
SDGND
SDVDD
EP
22.95
RSET
where:
RSET = 5.1 kΩ.
ICP = 4.5 mA.
Lock Detect Output Pin. This pin outputs a logic high to indicate PLL lock; a logic low output indicates loss of PLL
lock.
Digital Ground. Ground return path for DVDD.
Digital Power Supply. This pin should be the same voltage as AVDD. Decoupling capacitors to the ground plane
should be placed as close as possible to this pin.
Reference Input. This is a CMOS input with a nominal threshold of VDD/2 and a dc equivalent input resistance
of 100 kΩ. This input can be driven from a TTL or CMOS crystal oscillator, or it can be ac-coupled.
Multiplexer Output. This multiplexer output allows either the lock detect, the scaled RF, or the scaled reference
frequency to be accessed externally.
Digital Sigma-Delta (Σ-Δ) Modulator Ground. Ground return path for the Σ-Δ modulator.
Power Supply Pin for the Digital Σ-Δ Modulator. Should be the same voltage as AVDDx. Decoupling capacitors
to the ground plane are to be placed as close as possible to this pin.
The exposed pad must be connected to GND.
Rev. B | Page 8 of 28
Data Sheet
ADF4151
TYPICAL PERFORMANCE CHARACTERISTICS
0
–90
+85°C
+25°C
–40°C
–5
LOW SPUR MODE
–91
–92
PHASE NOISE (dBc/Hz)
POWER (dBm)
–10
–15
–20
–25
–30
–93
–94
–95
–96
LOW NOISE MODE
–97
–98
–35
1.0
1.5
2.0
2.5
3.0
3.5
4.0
FREQUENCY (GHz)
ICP MISMATCH (%)
1.13mA
0.56mA
0.56mA
1.13mA
2.25mA
4.5mA
SOURCE
0.5
1.0
1.5
2.0
2.5
VCP (V)
SINK
3.0
3.5
4.0
4.5
5.0
10265-005
ICP (mA)
2.25mA
0
2.63
2.64
2.65
2.66
2.67
2.68
2.69
2.70
Figure 6. In-Band Phase Noise Measured at 10 kHz Offset
for Low Noise Mode and Low Spur Mode,
PFD = 25 MHz, PLL Loop Bandwidth = 50 kHz
4.5mA
0.28mA
0.28mA
2.62
FREQUENCY (GHz)
Figure 4. RF Input Sensitivity
6.0
5.5
5.0
4.5
4.0
3.5
3.0
2.5
2.0
1.5
1.0
0.5
0
–0.5
–1.0
–1.5
–2.0
–2.5
–3.0
–3.5
–4.0
–4.5
–5.0
–5.5
–6.0
2.61
Figure 5. Charge Pump Output Characteristics, VP = 5 V, Selected ICP Values
Between 0.28 mA (Min) and 4.5 mA (Max), RSET = 5.1 kΩ
6.0
5.5
5.0
4.5
4.0
3.5
3.0
2.5
2.0
1.5
1.0
0.5
0
–0.5
–1.0
–1.5
–2.0
–2.5
–3.0
–3.5
–4.0
–4.5
–5.0
–5.5
–6.0
ICP
ICP
ICP
ICP
ICP
0
0.5
1.0
1.5
2.0
= 0.28mA
= 0.56mA
= 1.13mA
= 2.25mA
= 4.5mA
2.5
VCP (V)
3.0
3.5
4.0
4.5
5.0
10265-007
0.5
10265-004
0
–100
2.60
10265-006
–99
–40
Figure 7. Charge Pump Output Mismatch vs. VCP , Selected ICP Values Between
0.28 mA (Min) and 4.5 mA (Max), RSET = 5.1 kΩ
Rev. B | Page 9 of 28
Data Sheet
–60
–60
–80
–80
PHASE NOISE (dBc/Hz)
–100
–120
–140
–160
–100
–120
–140
10k
100k
1M
10M
FREQUENCY OFFSET (Hz)
–180
1k
10265-008
–80
–80
PHASE NOISE (dBc/Hz)
PHASE NOISE (dBc/Hz)
–60
–100
–120
–140
10M
–100
–120
–140
–160
–160
100k
1M
10M
–180
1k
10265-009
10k
FREQUENCY OFFSET (Hz)
Figure 9. Fractional-N Phase Noise and Spur Performance; Low Noise Mode;
VCOOUT = 1755.2 MHz, REFIN = 100 MHz, PFD = 25 MHz, Loop Filter
Bandwidth = 50 kHz, Channel Spacing = 200 kHz, FRAC = 26, MOD = 125
–60
–80
–80
–120
–140
100k
1M
10M
Figure 12. Fractional-N Phase Noise and Spur Performance; Low Noise Mode;
VCOOUT = 905.2 MHz, REFIN = 100 MHz, PFD = 25 MHz, Loop Filter
Bandwidth= 20 kHz, Channel Spacing = 200 kHz, FRAC = 26, MOD = 125
–60
–100
10k
FREQUENCY OFFSET (Hz)
PHASE NOISE (dBc/Hz)
PHASE NOISE (dBc/Hz)
1M
Figure 11. Integer-N Phase Noise and Spur Performance;
Low Noise Mode; VCOOUT = 900 MHz, REFIN = 100 MHz,
PFD = 25 MHz, Loop Filter Bandwidth = 20 kHz
–60
–160
–100
–120
–140
–160
10k
100k
1M
FREQUENCY OFFSET (Hz)
10M
–180
1k
10265-010
–180
1k
100k
FREQUENCY OFFSET (Hz)
Figure 8. Integer-N Phase Noise and Spur Performance;
Low Noise Mode; VCOOUT = 1750 MHz, REFIN = 100 MHz,
PFD = 25 MHz, Loop Filter Bandwidth = 50 kHz
–180
1k
10k
10265-012
–180
1k
10265-011
–160
Figure 10. Fractional-N Phase Noise and Spur Performance; Low Spur Mode;
VCOOUT = 1755.2 MHz, REFIN = 100 MHz, PFD = 25 MHz, Loop Filter
Bandwidth = 50 kHz, Channel Spacing = 200 kHz, FRAC = 26, MOD = 125
10k
100k
1M
FREQUENCY OFFSET (Hz)
10M
10265-013
PHASE NOISE (dBc/Hz)
ADF4151
Figure 13. Fractional-N Phase Noise and Spur Performance; Low Spur Mode;
VCOOUT = 905.2 MHz, REFIN = 100 MHz, PFD = 25 MHz, Loop Filter Bandwidth
= 20 kHz, Channel Spacing = 200 kHz, FRAC = 26, MOD = 125
Rev. B | Page 10 of 28
Data Sheet
ADF4151
CIRCUIT DESCRIPTION
REFERENCE INPUT SECTION
N = INT + FRAC/MOD
RF N DIVIDER
The reference input stage is shown in Figure 14. SW1 and SW2
are normally closed switches. SW3 is normally open. When
power-down is initiated, SW3 is closed and SW1 and SW2 are
opened. This ensures that there is no loading of the REFIN pin
on power-down.
FROM
VCO OUTPUT/
OUTPUT DIVIDERS
TO PFD
N COUNTER
THIRD-ORDER
FRACTIONAL
INTERPOLATOR
POWER-DOWN
CONTROL
MOD
REG
FRAC
VALUE
10265-015
INT
REG
100kΩ
NC
SW2
REFIN NC
TO R COUNTER
Figure 15. RF INT Divider
BUFFER
SW1
INT N MODE
10265-014
SW3
NO
The RF N divider allows a division ratio in the PLL feedback
path. Division ratio is determined by the INT, FRAC, and MOD
values, which build up this divider.
If the FRAC = 0 and DB8 in Register 2 (LDF) is set to 1, the
synthesizer operates in integer-N mode. The DB8 in Register 2
(LDF) should be set to 1 to get integer-N digital lock detect.
Additionally, lower phase noise is possible if the antibacklash
pulse width is reduced to 3 ns. This mode is not valid for
fractional-N applications.
INT, FRAC, MOD, AND R COUNTER RELATIONSHIP
R COUNTER
The INT, FRAC, and MOD values, in conjunction with the R
counter, make it possible to generate output frequencies that
are spaced by fractions of the PFD frequency. See the RF
Synthesizer—A Worked Example section for more
information. The RF VCO frequency (RFOUT) equation is
The 10-bit R counter allows the input reference frequency
(REFIN) to be divided down to produce the reference clock
to the PFD. Division ratios from 1 to 1023 are allowed.
Figure 14. Reference Input Stage
RF N DIVIDER
(1)
where:
RFOUT is the output frequency of the external voltage controlled
oscillator (VCO).
INT is the preset divide ratio of the binary 16-bit counter
(23 to 32,767 for 4/5 prescaler, 75 to 65,535 for 8/9 prescaler).
FRAC is the numerator of the fractional division (0 to MOD − 1).
MOD is the preset fractional modulus (2 to 4095 for low noise
mode, 50 to 4095 for low spur mode).
fPFD = REFIN × [(1 + D)/(R × (1 + T))]
The phase frequency detector (PFD) takes inputs from the R
counter and N counter and produces an output proportional to
the phase and frequency difference between them. Figure 16 is
a simplified schematic of the phase frequency detector. The PFD
includes a programmable delay element that sets the width of
the antibacklash pulse, which can be either 6 ns (default, for
fractional-N applications) or 3 ns (for integer-N mode). This
pulse ensures that there is no dead zone in the PFD transfer
function and gives a consistent reference spur level.
(2)
where:
REFIN is the reference input frequency.
D is the REFIN doubler bit.
R is the preset divide ratio of the binary 10–bit programmable
reference counter (1 to 1023).
T is the REFIN divide-by-2 bit (0 or 1).
HIGH
D1
Q1
UP
U1
+IN
CLR1
DELAY
HIGH
U3
CHARGE
PUMP
CLR2
DOWN
D2
Q2
U2
–IN
Figure 16. PFD Simplified Schematic
Rev. B | Page 11 of 28
CP
10265-016
RFOUT = fPFD × (INT + (FRAC/MOD))
PHASE FREQUENCY DETECTOR (PFD) AND
CHARGE PUMP
ADF4151
Data Sheet
MUXOUT AND LOCK DETECT
shift register. There are three LSBs: DB2, DB1, and DB0, as
shown in Figure 2. The truth table for these bits is shown in
Table 5. Figure 18 shows a summary of how the latches are
programmed.
The output multiplexer on the ADF4151 allows the user
to access various internal points on the chip. The state of
MUXOUT is controlled by M3, M2, and M1 (for details, see
Figure 21). Figure 17 shows the MUXOUT section in block
diagram form.
Table 5. C3, C2, and C1 Truth Table
R COUNTER INPUT
C3
0
0
0
0
1
1
DVDD
THREE-STATE-OUTPUT
DVDD
DGND
R COUNTER OUTPUT
MUX
CONTROL
MUXOUT
N COUNTER OUTPUT
Control Bits
C2
0
0
1
1
0
0
C1
0
1
0
1
0
1
Register
Register 0 (R0)
Register 1 (R1)
Register 2 (R2)
Register 3 (R3)
Register 4 (R4)
Register 5 (R5)
PROGRAM MODES
ANALOG LOCK DETECT
DIGITAL LOCK DETECT
Figure 19 through Figure 24 show how the program modes are
to be set up in the ADF4151.
10265-017
RESERVED
DGND
Figure 17. MUXOUT Schematic
INPUT SHIFT REGISTERS
The ADF4151 digital section includes a 10-bit RF R counter,
a 16-bit RF N counter, a 12-bit FRAC counter, and a 12-bit
modulus counter. Data is clocked into the 32-bit shift register
on each rising edge of CLK. The data is clocked in MSB first.
Data is transferred from the shift register to one of six latches
on the rising edge of LE. The destination latch is determined
by the state of the three control bits (C3, C2, and C1) in the
A number of settings in the ADF4151 are double buffered.
These include the modulus value, phase value, R counter
value, reference doubler, reference divide-by-2, and current
setting. This means that two events must occur before the
part uses a new value of any of the double-buffered settings.
First, the new value is latched into the device by writing to the
appropriate register. Second, a new write must be performed
on Register R0. For example, any time the modulus value is
updated, Register R0 must be written to, thus ensuring that the
modulus value is loaded correctly.
Rev. B | Page 12 of 28
Data Sheet
ADF4151
REGISTER MAPS
RESERVED
REGISTER 0
16-BIT INTEGER VALUE (INT)
CONTROL
BITS
12-BIT FRACTIONAL VALUE (FRAC)
DB31 DB30 DB29 DB28 DB27 DB26 DB25 DB24 DB23 DB22 DB21 DB20 DB19 DB18 DB17 DB16 DB15 DB14 DB13 DB12 DB11 DB10 DB9 DB8 DB7 DB6 DB5 DB4 DB3
0
N16
N15
N14
N13
N12
N11
N10
N9
N8
N7
N6
N5
N4
N3
N2
N1
F12
F11
F10
F9
F8
F7
F6
F5
F4
F3
F2
F1
DB2
DB1
DB0
C3(0) C2(0) C1(0)
RESERVED
PRESCALER
PHASE ADJUST
REGISTER 1
DBR1
12-BIT PHASE VALUE (PHASE)
CONTROL
BITS
DBR 1
12-BIT MODULUS VALUE (MOD)
P12
P11
P10
P9
P8
P7
P6
P5
P4
P3
P2
P1
M12
M11
M10
M9
M8
M7
DB2
DB1
DB0
M6
M5
M4
M3
M2
M1
C3(0) C2(0) C1(1)
COUNTER
RESET
PR1
CP THREESTATE
PH1
POWER-DOWN
0
PD
POLARITY
0
LDF
0
LDP
DB31 DB30 DB29 DB28 DB27 DB26 DB25 DB24 DB23 DB22 DB21 DB20 DB19 DB18 DB17 DB16 DB15 DB14 DB13 DB12 DB11 DB10 DB9 DB8 DB7 DB6 DB5 DB4 DB3
CONTROL
BITS
DBR 1
10-BIT R COUNTER
RESERVED
MUXOUT
RDIV2
LOW
NOISE AND
LOW SPUR
MODES
REFERENCE
DOUBLER DBR 1
RESERVED
REGISTER 2
DBR 1
CHARGE
PUMP
CURRENT
SETTING
DBR 1
DB31 DB30 DB29 DB28 DB27 DB26 DB25 DB24 DB23 DB22 DB21 DB20 DB19 DB18 DB17 DB16 DB15 DB14 DB13 DB12 DB11 DB10 DB9 DB8 DB7 DB6 DB5 DB4 DB3
0
L2
L1
M3
M2
M1
RD2
RD1
R10
R9
R8
R7
R6
R5
R4
R3
R2
R1
0
CP4
CP3
CP2
CP1
U6
U5
U4
U3
U2
U1
DB2
DB1
DB0
C3(0) C2(1) C1(0)
RESERVED
RESERVED
CSR
ABP
RESERVED
CHARGE
CANCEL
REGISTER 3
CLK
DIV
MODE
CONTROL
BITS
12-BIT CLOCK DIVIDER VALUE
DB31 DB30 DB29 DB28 DB27 DB26 DB25 DB24 DB23 DB22 DB21 DB20 DB19 DB18 DB17 DB16 DB15 DB14 DB13 DB12 DB11 DB10 DB9 DB8 DB7 DB6 DB5 DB4 DB3
0
0
0
0
0
0
0
0
0
F3
F2
0
0
F1
0
C2
C1
D12
D11
D10
D9
D8
D7
D6
D5
D4
D3
D2
D1
DB2
DB1
DB0
C3(0) C2(1) C1(1)
REGISTER 4
CONTROL
BITS
RESERVED
DB31 DB30 DB29 DB28 DB27 DB26 DB25 DB24 DB23 DB22 DB21 DB20 DB19 DB18 DB17 DB16 DB15 DB14 DB13 DB12 DB11 DB10 DB9 DB8 DB7 DB6 DB5 DB4 DB3
0
0
0
0
0
0
0
0
1
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
DB2
DB1
DB0
C3(1) C2(0) C1(0)
REGISTER 5
CONTROL
BITS
RESERVED
DB31 DB30 DB29 DB28 DB27 DB26 DB25 DB24 DB23 DB22 DB21 DB20 DB19 DB18 DB17 DB16 DB15 DB14 DB13 DB12 DB11 DB10 DB9 DB8 DB7 DB6 DB5 DB4 DB3
0
1 DBR
0
0
0
0
0
0
0
D15
D14
0
0
0
0
0
0
0
0
0
= DOUBLE BUFFERED REGISTER—BUFFERED BY THE WRITE TO REGISTER 0.
Figure 18. Register Summary
Rev. B | Page 13 of 28
0
0
0
0
0
0
0
0
0
0
DB2
DB1
DB0
C3(1) C2(0) C1(1)
10265-018
LD PIN
MODE
RESERVED
Data Sheet
RESERVED
ADF4151
16-BIT INTEGER VALUE (INT)
DB31 DB30 DB29 DB28 DB27 DB26 DB25 DB24 DB23 DB22 DB21 DB20 DB19 DB18 DB17 DB16 DB15 DB14 DB13 DB12 DB11 DB10 DB9 DB8
N16
N15
N14
N13
N12
N11
N10
N9
N8
N7
N6
N5
N4
N3
N2
N1
F12
F11
F10
F9
F8
F7
F6
DB7 DB6
F5
F4
DB5 DB4
F3
F2
DB3
F1
DB2
N16
N15
...
N5
N4
N3
N2
N1
INTEGER VALUE (INT)
F12
F11
..........
F2
F1
FRACTIONAL VALUE (FRAC)
0
0
...
0
0
0
0
0
NOT ALLOWED
0
0
..........
0
0
0
0
0
...
0
0
0
0
1
NOT ALLOWED
0
0
..........
0
1
1
0
0
...
0
0
0
1
0
NOT ALLOWED
0
0
..........
1
0
2
.
.
...
.
.
.
.
.
...
0
0
..........
1
1
3
0
0
...
1
0
1
1
0
NOT ALLOWED
.
.
..........
.
.
.
0
0
...
1
0
1
1
1
23
.
.
..........
.
.
.
0
0
...
1
1
0
0
0
24
.
.
..........
.
.
.
.
.
...
.
.
.
.
.
...
1
1
..........
0
0
4092
1
1
...
1
1
1
0
1
65533
1
1
..........
0
1
4093
1
1
...
1
1
1
1
0
65534
1
1
..........
1
0
4094
1
1
...
1
1
1
1
1
65535
1
1
.........
1
1
4095
DB1
DB0
C3(0) C2(0) C1(0)
10265-019
0
CONTROL
BITS
12-BIT FRACTIONAL VALUE (FRAC)
INTmin = 75 WITH PRESCALER = 8/9
RESERVED
PRESCALER
PHASE ADJUST
Figure 19. Register 0 (R0)
DBR
12-BIT PHASE VALUE (PHASE)
DB31 DB30 DB29 DB28 DB27 DB26 DB25 DB24 DB23 DB22 DB21 DB20 DB19 DB18 DB17 DB16 DB15 DB14 DB13 DB12 DB11 DB10 DB9 DB8
0
0
PH1
1
OFF
ON
P12
P11
P10
P9
P8
P7
P6
P5
P4
P3
P2
P1
M12
M11
M10
M9
M8
M7
M6
DB7 DB6
M5
M4
P1
PRESCALER
P12
P11
..........
P2
P1
PHASE VALUE (PHASE)
M12
M11
..........
M2
M1
0
4/5
0
0
..........
0
0
0
0
0
..........
1
0
2
1
8/9
0
0
..........
0
1
1 (RECOMMENDED)
0
0
..........
1
1
3
0
0
..........
1
0
2
.
.
..........
.
.
.
0
0
..........
1
1
3
.
.
..........
.
.
.
.
.
..........
.
.
.
.
.
..........
.
.
.
1
1
..........
0
0
4092
.
.
..........
.
.
.
1
1
..........
0
1
4093
.
.
..........
.
.
.
1
1
..........
1
0
4094
1
1
..........
0
0
4092
1
1
..........
1
1
4095
1
1
..........
0
1
4093
1
1
..........
1
0
4094
1
1
..........
1
1
4095
PH1 PHASE ADJUST
0
PR1
DB5 DB4
M3
M2
DB3
M1
DB2
DB1
DB0
C3(0) C2(0) C1(1)
INTERPOLATOR MODULUS (MOD)
10265-020
0
CONTROL
BITS
DBR
12-BIT MODULUS VALUE (MOD)
Figure 20. Register 1 (R1)
Rev. B | Page 14 of 28
COUNTER
RESET
CP THREESTATE
POWER-DOWN
LDP
DBR
CHARGE
PUMP
CURRENT
SETTING
LDF
DBR
10-BIT R COUNTER
RESERVED
MUXOUT
RDIV2
LOW
NOISE AND
LOW SPUR
MODES
PD
POLARITY
ADF4151
REFERENCE
DOUBLER DBR
RESERVED
Data Sheet
DB31 DB30 DB29 DB28 DB27 DB26 DB25 DB24 DB23 DB22 DB21 DB20 DB19 DB18 DB17 DB16 DB15 DB14 DB13 DB12 DB11 DB10 DB9 DB8 DB7 DB6 DB5 DB4 DB3
0
L2
L1
M3
M2
M1
RD2
RD1
R10
R9
R8
R7
R6
R5
R4
R3
R2
R1
0
CP4
CP3
CP2
CP1
U6
U5
U4
U3
U2
U1
CONTROL
BITS
DB2
DB1
RD2
REFERENCE
DOUBLER
U6
LDF
U1
COUNTER
RESET
LOW NOISE MODE
0
DISABLED
0
FRAC-N
0
DISABLED
1
RESERVED
1
ENABLED
1
INT-N
1
ENABLED
1
0
RESERVED
1
1
LOW SPUR MODE
L2
NOISE MODE
0
0
0
RD1
R10
REFERENCE DIVIDE BY 2
0
DISABLED
CP4
CP3
CP2
CP1
ICP (mA)
5.1kΩ
1
ENABLED
0
0
0
0
0
0
0
0
1
1
1
1
1
1
1
1
0
0
0
0
1
1
1
1
0
0
0
0
1
1
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0.28
0.56
0.84
1.13
1.41
1.69
1.97
2.25
2.53
2.81
3.09
3.38
3.66
3.94
4.22
4.5
R9
..........
R2
R1
0
0
..........
0
1
1
0
0
..........
1
0
2
.
.
..........
.
.
.
.
.
..........
.
.
.
.
.
..........
.
.
.
1
1
..........
0
0
1020
1
1
..........
0
1
1021
1
1
..........
1
0
1022
1
1
..........
1
1
1023
M3
M2
M1
OUTPUT
0
0
0
THREE-STATE OUTPUT
0
0
1
DVDD
0
1
0
DGND
0
1
1
R DIVIDER OUTPUT
1
0
0
N DIVIDER OUTPUT
1
0
1
ANALOG LOCK DETECT
1
1
0
DIGITAL LOCK DETECT
1
1
1
RESERVED
R DIVIDER (R)
CP
THREE-STATE
U5
LDP
U2
0
10ns
0
DISABLED
1
6ns
1
ENABLED
U4
PD POLARITY
U3
POWER-DOWN
0
NEGATIVE
0
DISABLED
1
POSITIVE
1
ENABLED
10265-021
L1
DB0
C3(0) C2(1) C1(0)
Figure 21. Register 2 (R2)
Rev. B | Page 15 of 28
RESERVED
CSR
CHARGE
CANCEL
RESERVED
RESERVED
Data Sheet
ABP
ADF4151
CLK
DIV
MODE
DB31 DB30 DB29 DB28 DB27 DB26 DB25 DB24 DB23 DB22 DB21 DB20 DB19 DB18 DB17 DB16 DB15 DB14 DB13 DB12 DB11 DB10 DB9 DB8
0
0
0
0
0
0
0
0
0
F3
F2
0
0
F1
C2
C1
D12
D11
D10
D9
D8
D7
D6
DB7 DB6
D5
CYCLE SLIP
REDUCTION
D12
D11
..........
D2
D1
0
0
..........
0
0
0
0
DISABLED
0
0
..........
0
1
1
1
ENABLED
0
0
..........
1
0
2
0
0
..........
1
1
3
.
.
..........
.
.
.
.
.
..........
.
.
.
.
.
..........
.
.
.
1
1
..........
0
0
4092
F1
F2
0
CONTROL
BITS
12-BIT CLOCK DIVIDER VALUE
D4
DB5 DB4
D3
D2
DB3
D1
DB2
DB1
DB0
C3(0) C2(1) C1(1)
CLOCK DIVIDER VALUE
C2
C1
CLOCK DIVIDER MODE
0
0
CLOCK DIVIDER OFF
0
1
FAST LOCK ENABLE
1
1
..........
0
1
4093
1
0
RESYNC ENABLE
1
1
..........
1
0
4094
1
1
RESERVED
1
1
..........
1
1
4095
CHARGE
CANCELLATION
0
DISABLED
1
ENABLED
0
6ns (FRAC-N)
1
3ns (INT_N)
10265-022
ANTIBACKLASH
PULSE WIDTH
F3
Figure 22. Register 3 (R3)
DB31 DB30 DB29 DB28 DB27 DB26 DB25 DB24 DB23 DB22 DB21 DB20 DB19 DB18 DB17 DB16 DB15 DB14 DB13 DB12 DB11 DB10 DB9 DB8 DB7 DB6 DB5 DB4 DB3
0
0
0
0
0
0
0
0
1
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
DB2
DB1
DB0
C3(1) C2(0) C1(0)
10265-023
CONTROL
BITS
RESERVED
Figure 23. Register 4 (R4)
LD PIN
MODE
RESERVED
CONTROL
BITS
RESERVED
DB31 DB30 DB29 DB28 DB27 DB26 DB25 DB24 DB23 DB22 DB21 DB20 DB19 DB18 DB17 DB16 DB15 DB14 DB13 DB12 DB11 DB10 DB9 DB8
0
0
0
0
0
0
0
D15
D14
0
0
D1 5
D1 4
LOCK DETECT PIN OPERATION
0
0
LOW
0
1
DIGITAL LOCK DETECT
1
0
LOW
1
1
HIGH
0
0
0
0
0
0
0
0
0
0
0
0
0
0
DB5 DB4
0
0
DB3
0
DB2
DB1
DB0
C3(1) C2(0) C1(1)
10265-024
0
DB7 DB6
Figure 24. Register 5 (R5)
Rev. B | Page 16 of 28
Data Sheet
ADF4151
REGISTER 0
12-Bit Phase Value (PHASE)
Control Bits
These bits control what is loaded as the phase word. The word
must be less than the MOD value programmed in Register 1.
The word is used to program the RF output phase from 0° to
360° with a resolution of 360°/MOD. See the Phase Resync
section for more information. In most applications, the phase
relationship between the RF signal and the reference is not
important. In such applications, the phase value can be used to
optimize the fractional and subfractional spur levels. See the
Spur Consistency and Fractional Spur Optimization section for
more information.
With Bits[C3:C1] set to 0, 0, 0, Register 0 is programmed.
Figure 19 shows the input data format for programming this
register.
16-Bit Integer Value (INT)
These 16 bits set the INT value, which determines the integer
part of the feedback division factor. They are used in Equation 1
(see the INT, FRAC, MOD, and R Counter Relationship
section). All integer values from 23 to 32,767 are allowed for 4/5
prescaler. For 8/9 prescaler, the minimum integer value is 75, and
the maximum value is 65,535.
12-Bit Fractional Value (FRAC)
The 12 FRAC bits set the numerator of the fraction that is input
to the Σ-Δ modulator. This, along with INT, specifies the new
frequency channel that the synthesizer locks to, as shown in the
RF Synthesizer—A Worked Example section. FRAC values from
0 to MOD − 1 cover channels over a frequency range equal to
the PFD reference frequency.
REGISTER 1
This programmable register sets the fractional modulus. This
is the ratio of the PFD frequency to the channel step resolution
on the RF output. See the RF Synthesizer—A Worked Example
section for more information.
REGISTER 2
With Bits[C3:C1] set to 0, 1, 0, Register 2 is programmed.
Figure 21 shows the input data format for programming
this register.
With Bits[C3:C1] set to 0, 0, 1, Register 1 is programmed.
Figure 20 shows the input data format for programming
this register.
Low Noise and Spur Modes
Phase Adjust
The noise modes on the ADF4151 are controlled by DB30 and
DB29 in Register 2 (see Figure 21). The noise modes allow the
user to optimize a design either for improved spurious performance or for improved phase noise performance.
The phase adjust bit, enabled by programming a 1 to DB28,
permits adjustments to the output phase of a given output
frequency. If enabled, it does not perform a phase resync
function on updating R0. If set to 0, the phase resync (if
enabled in R3, Bits[DB16:DB15]) occurs on every update
of R0.
Prescaler Value
The dual modulus prescaler (P/P + 1), along with the INT,
FRAC, and MOD counters, determines the overall division
ratio from the VCO output to the PFD input.
Operating at CML levels, it takes the clock from the VCO
output and divides it down for the counters. It is based on a
synchronous 4/5 core. When set to 4/5, the maximum RF
frequency allowed is 3 GHz. Therefore, when operating the
ADF4151 above 3 GHz, this must be set to 8/9. The prescaler
limits the INT value, where:
In the ADF4151, PR1 in Register 1 sets the prescaler values.
12-Bit Modulus Value (MOD)
Control Bits
Control Bits
P = 4/5, NMIN = 23
P = 8/9, NMIN = 75
If neither the phase resync nor the spurious optimization
functions are being used, it is recommended that the phase
word be set to 1.
When the lowest spur setting is chosen, dither is enabled. This
randomizes the fractional quantization noise so it resembles
white noise rather than spurious noise. As a result, the part is
optimized for improved spurious performance. This operation
would normally be used when the PLL closed-loop bandwidth
is wide, for fast locking applications. (Wide-loop bandwidth is
seen as a loop bandwidth greater than 1/10 of the RFOUT channel
step resolution (fRES)). A wide loop filter does not attenuate the
spurs to the same level as a narrow-loop bandwidth.
For best noise performance, use the lowest noise setting option.
As well as disabling the dither, it also ensures that the charge
pump is operating in an optimum region for noise performance.
This setting is extremely useful where a narrow-loop filter
bandwidth is available. The synthesizer ensures extremely low
noise, and the filter attenuates the spurs. The typical performance
characteristics give the user an idea of the trade-off in a typical
W-CDMA setup for the different noise and spur settings.
Rev. B | Page 17 of 28
ADF4151
Data Sheet
MUXOUT
Lock Detect Precision (LDP)
The on-chip multiplexer is controlled by Bits[DB28:DB26] (see
Figure 21).
When DB7 is set to 0, the fractional-N digital lock detect is
activated. In this case after setting DB7 to 0, 40 consecutive PFD
cycles of 10 ns must occur before digital lock detect is set. When
DB7 is programmed to 1, 40 consecutive reference cycles of 6 ns
must occur before digital lock detect goes high. Setting DB8
(LDF) to 1 causes the activation of the integer-N digital lock
detect. In this case, after setting DB7 (LDP) to 0, five
consecutive cycles of 10 ns must occur before digital lock detect
is set. When DB7 is set to 1, five consecutive cycles of 6 ns must
occur. Recommended settings of both the LDP and LDF bits are
shown in Table 6.
Reference Doubler
Setting DB25 to 0 feeds the REFIN signal directly to the 10-bit
R counter, disabling the doubler. Setting this bit to 1 multiplies
the REFIN frequency by a factor of 2 before feeding into the
10-bit R counter. When the doubler is disabled, the REFIN
falling edge is the active edge at the PFD input to the fractional
synthesizer. When the doubler is enabled, both the rising and
falling edges of REFIN become active edges at the PFD input.
When the doubler is enabled and the lowest spur mode is
chosen, the in-band phase noise performance is sensitive to the
REFIN duty cycle. The phase noise degradation can be as much
as 5 dB for the REFIN duty cycles outside a 45% to 55% range.
The phase noise is insensitive to the REFIN duty cycle in the
lowest noise mode. The phase noise is insensitive to the REFIN
duty cycle when the doubler is disabled.
When the doubler is enabled, the maximum allowable REFIN
frequency is 30 MHz.
RDIV2
Setting the DB24 bit to 1 inserts a divide-by-2 toggle flip-flop
between the R counter and PFD, which extends the maximum
REFIN input rate. This function allows a 50% duty cycle signal
to appear at the PFD input, which is necessary for cycle slip
reduction.
10-Bit R Counter
The 10-bit R counter allows the input reference frequency
(REFIN) to be divided down to produce the reference clock to
the PFD. Division ratios from 1 to 1023 are allowed.
Current Setting
Bits[DB12:DB9] set the charge pump current setting. This
should be set to the charge pump current that the loop filter
is designed with (see Figure 21).
LDF
Setting DB8 to 1 enables integer-N digital lock detect, when
the FRAC part of the divider is zero; setting DB8 to 0 enables
fractional-N digital lock detect.
Table 6. Recommended LDF/LDP Bit Settings
Mode
Integer-N
Fractional-N Low Noise Mode
Fractional-N Low Spur Mode
DB8
(LDF)
1
0
0
DB7
(LDP)
1
1
0
Phase Detector Polarity
DB6 sets the phase detector polarity. When a passive loop filter
or noninverting active loop filter is used, set this bit to 1. If an
active filter with an inverting characteristic is used, this bit
should be set to 0.
Power-Down (PD)
DB5 provides the programmable power-down mode. Setting this
bit to 1 performs a power-down. Setting this bit to 0 returns the
synthesizer to normal operation. When in software power-down
mode, the part retains all information in its registers. Only if the
supply voltages are removed are the register contents lost.
When a power-down is activated, the following events occur:
•
•
•
•
•
The synthesizer counters are forced to their load state
conditions.
The charge pump is forced into three-state mode.
The digital lock detect circuitry is reset.
The RFOUT buffers are disabled.
The input register remains active and capable of loading
and latching data.
Charge Pump (CP) Three-State
DB4 puts the charge pump into three-state mode when
programmed to 1. It should be set to 0 for normal operation.
Counter Reset
DB3 is the R counter and N counter reset bit for the ADF4151.
When this bit is 1, the RF synthesizer N counter and R counter
are held in reset. For normal operation, this bit should be set to 0.
Rev. B | Page 18 of 28
Data Sheet
ADF4151
REGISTER 3
REGISTER 4
Control Bits
Control Bits
With Bits[C3:C1] set to 0, 1, 1, Register 3 is programmed.
Figure 22 shows the input data format for programming
this register.
With Bits[C3: C1] set to 1, 0, 0, Register 4 is programmed.
Figure 23 shows the input data format for programming this
register.
Antibacklash Pulse Width
This register is reserved and has to be programmed with the
values as shown in Figure 23. Bits[DB31:DB24] and [DB22:DB3]
must be programmed to 0, while Bit DB23 must be set to 1.
Setting DB22 to 0 sets the PFD antibacklash pulse width to 6 ns.
This is the recommended mode for fractional-N use. By setting
this bit to 1, the 3 ns pulse width is used and results in a phase
noise and spur improvement in integer-N operation. For
fractional-N mode it is not recommended to use this smaller
setting.
Charge Cancellation Mode Pulse Width
Setting DB21 to 1 enables charge pump charge cancellation.
This has the effect of reducing PFD spurs in integer-N mode.
In fractional-N mode, this bit should not be used. This results
in a phase noise and fractional spur improvement.
REGISTER 5
Control Bits
With Bits[C3:C1] set to 1, 0, 1, Register 5 is programmed.
Figure 24 shows the input data form for programming this
register.
Lock Detect PIN Operation
Bits[DB23:DB22] set the operation of the lock detect pin (see
Figure 24).
Cycle Slip Reduction (CSR) Enable
INITIALIZATION SEQUENCE
Setting DB18 to 1 enables cycle slip reduction. This is a method
for improving lock times. Note that the signal at the phase frequency detector (PFD) must have a 50% duty cycle for cycle slip
reduction to work. The charge pump current setting must also
be set to a minimum. See the Cycle Slip Reduction for Faster
Lock Times section for more information.
The following sequence of registers is the correct sequence for
initial power up of the ADF4151 after the correct application
of voltages to the supply pins:
Clock Divider Mode
Bits[DB16:DB15] must be set to 1, 0 to activate phase resync or
0, 1 to activate fast lock. Setting Bits[DB16:DB15] to 0, 0
disables the clock divider. See Figure 22.
1.
2.
3.
4.
5.
6.
12-Bit Clock Divider Value
The 12-bit clock divider value sets the timeout counter for
activation of phase resync. See the Phase Resync section for
more information. It also sets the timeout counter for fast lock.
See the Fast Lock Timer and Register Sequences section for
more information.
Rev. B | Page 19 of 28
Register 5
Register 4
Register 3
Register 2
Register 1
Register 0
ADF4151
Data Sheet
RF SYNTHESIZER—A WORKED EXAMPLE
MODULUS
The following is an example of how to program the ADF4151
synthesizer:
The choice of modulus (MOD) depends on the reference signal
(REFIN) available and the channel resolution (fRES) required at
the RF output. For example, a GSM system with 13 MHz REFIN
sets the modulus to 65. This means that the RF output resolution
(fRES) is the 200 kHz (13 MHz/65) necessary for GSM. With dither
off, the fractional spur interval depends on the modulus values
chosen (see Table 7).
RFOUT = [INT + (FRAC/MOD)] × [fPFD]/RF Divider
(3)
where:
RFOUT is the RF frequency output.
INT is the integer division factor.
FRAC is the fractionality.
MOD is the modulus.
RF Divider is the output divider that divides down the VCO
frequency.
fPFD = REFIN × [(1 + D)/(R × (1 + T))]
(4)
where:
REFIN is the reference frequency input.
D is the RF REFIN doubler bit.
R is the RF reference division factor.
T is the reference divide-by-2 bit (0 or 1).
12-BIT PROGRAMMABLE MODULUS
Unlike most other fractional-N PLLs, the ADF4151 allows the
user to program the modulus over a 12-bit range. This means
that the user can set up the part in many different configurations
for the application, when combined with the reference doubler
and the 10-bit R counter.
fPFD
RFOUT
N
DIVIDER
10265-025
VCO
Figure 25. Loop Closed Before Output Divider
A channel resolution (fRES) of 200 kHz is required at the output
of the VCO.
MOD = REFIN/fRES
MOD = 10 MHz/200 kHz = 50
From Equation 4
fPFD = [10 MHz × (1 + 0)/1] = 10 MHz
(5)
2112.6 MHz = 10 MHz × (INT + FRAC/50)
(6)
where:
INT = 211
FRAC = 13
The reference doubler on chip allows the input reference signal
to be doubled. This is useful for increasing the PFD comparison
frequency. Making the PFD frequency higher improves the
noise performance of the system. Doubling the PFD frequency
usually improves noise performance by 3 dB. It is important
to note that the PFD cannot operate above maximum value (see
Table 1) due to a limitation in the speed of the Σ-Δ circuit of the
N-divider.
The reference divide-by-2 divides the reference signal by 2,
resulting in a 50% duty cycle PFD frequency. This is necessary
for the correct operation of the cycle slip reduction (CSR)
function. See the Cycle Slip Reduction for Faster Lock Times
section for more information.
For example, in a UMTS system, where 2112.6 MHz RF
frequency output (RFOUT) is required, a 10 MHz reference
frequency input (REFIN) is available, and a 200 kHz channel
resolution (fRESOUT) is required on the RF output. A 2.1 GHz
VCO is suitable to cover the required fractional frequency of
2112.6 MHz.
PFD
REFERENCE DOUBLER AND REFERENCE DIVIDER
For example, consider an application that requires 1.75 GHz RF
and 200 kHz channel step resolution. The system has a 13 MHz
reference signal.
One possible setup is feeding the 13 MHz directly to the PFD
and programming the modulus to divide by 65. This results in
the required 200 kHz resolution.
Another possible setup is using the reference doubler to create
26 MHz from the 13 MHz input signal. The 26 MHz is then fed
into the PFD, programming the modulus to divide by 130. This
also results in 200 kHz resolution and offers superior phase
noise performance over the previous setup.
The programmable modulus is also very useful for multistandard applications. If a dual-mode phone requires PDC
and GSM 1800 standards, the programmable modulus is a
great benefit. PDC requires 25 kHz channel step resolution,
whereas GSM 1800 requires 200 kHz channel step resolution.
Rev. B | Page 20 of 28
Data Sheet
ADF4151
A 13 MHz reference signal can be fed directly to the PFD, and
the modulus can be programmed to 520 when in PDC mode
(13 MHz/520 = 25 kHz).
The modulus needs to be reprogrammed to 65 for GSM 1800
operation (13 MHz/65 = 200 kHz).
It is important that the PFD frequency remain constant (13 MHz).
This allows the user to design one loop filter for both setups
without running into stability issues. It is important to remember that the ratio of the RF frequency to the PFD frequency
principally affects the loop filter design, not the actual channel
spacing.
CYCLE SLIP REDUCTION FOR FASTER LOCK TIMES
As outlined in the Low Noise and Spur Mode section, the
ADF4151 contains a number of features that allow optimization
for noise performance. However, in fast locking applications,
the loop bandwidth generally needs to be wide, and, therefore,
the filter does not provide much attenuation of the spurs. If
the cycle slip reduction feature is enabled, the narrow-loop
bandwidth is maintained for spur attenuation but faster lock
times are still possible.
Cycle Slips
If the phase error increases again to a point where another cycle
slip is likely, the ADF4151 turns on another charge pump cell.
This continues until the ADF4151 detects that the VCO
frequency has gone past the desired frequency. The extra charge
pump cells are turned off one by one until all the extra charge
pump cells have been disabled and the frequency is settled with
the original loop filter bandwidth.
Up to seven extra charge pump cells can be turned on. In most
applications, it is enough to eliminate cycle slips altogether,
giving much faster lock times.
Setting Bit DB18 in the Register 3 to 1 enables cycle slip
reduction. Note that the PFD requires a 45% to 55% duty
cycle for CSR to operate correctly.
SPURIOUS OPTIMIZATION AND FAST LOCK
Narrow-loop bandwidths can filter unwanted spurious signals,
but these usually have a long lock time. A wider loop bandwidth
achieves faster lock times, but a wider loop bandwidth may lead
to increased spurious signals inside the loop bandwidth.
The fast lock feature can achieve the same fast lock time as the
wider bandwidth, but with the advantage of a narrow final loop
bandwidth to keep spurs low.
Cycle slips occur in integer-N/fractional-N synthesizers when
the loop bandwidth is narrow compared to the PFD frequency.
The phase error at the PFD inputs accumulates too fast for the
PLL to correct, and the charge pump temporarily pumps in the
wrong direction. This slows down the lock time dramatically.
The ADF4151 contains a cycle slip reduction feature that
extends the linear range of the PFD, allowing faster lock
times without modifications to the loop filter circuitry.
FAST LOCK TIMER AND REGISTER SEQUENCES
When the circuitry detects that a cycle slip is about to occur,
it turns on an extra charge pump current cell. This outputs a
constant current to the loop filter or removes a constant
current from the loop filter (depending on whether the VCO
tuning voltage needs to increase or decrease to acquire the
new frequency). The effect is that the linear range of the PFD
is increased. Loop stability is maintained because the current
is constant and is not a pulsed current.
1.
If the fast lock mode is used, a timer value must be loaded into
the PLL to determine the duration of the wide bandwidth mode.
When Bits[DB16:DB15] in Register 3 are set to 0, 1 (fast
lock enable), the timer value is loaded by the 12-bit clock
divider value. The following sequence must be programmed
to use fast lock:
2.
Rev. B | Page 21 of 28
Initialization sequence (see the Initialization Sequence
section); occurs only once after powering up the part.
Load Register 3 by setting Bits[DB16:DB15] to 0, 1 and
the chosen fast lock timer value, Bits[DB14:DB3]. Note that
the length of time the PLL remains in wide bandwidth is
equal to the fast lock timer/fPFD.
ADF4151
Data Sheet
FAST LOCK—AN EXAMPLE
SPUR MECHANISMS
If a PLL has a reference frequency of 13 MHz, a fPFD of 13 MHz
and a required lock time of 50 µs, the PLL is set to wide bandwidth
for 40 µs. This example assumes a modulus of 65 for channel
spacing of 200 kHz.
This section describes the three different spur mechanisms that
arise with a fractional-N synthesizer and how to minimize them
in the ADF4151.
If the time period set for the wide bandwidth is 40 µs, then
The fractional interpolator in the ADF4151 is a third-order Σ-Δ
modulator (SDM) with a modulus (MOD) that is programmable
to any integer value from 2 to 4095. In low spur mode (dither
enabled), the minimum allowable value of MOD is 50. The
SDM is clocked at the PFD reference rate (fPFD) that allows PLL
output frequencies to be synthesized at a channel step resolution
of fPFD/MOD.
Fast Lock Timer Value = Time In Wide Bandwidth × fPFD/MOD
Fast Lock Timer Value = 40 µs × 13 MHz/65 = 8
Therefore, 8 must be loaded into the clock divider value in
Register 3 in Step 1 of the sequence described in the Fast Lock
Timer and Register Sequences section.
FAST LOCK—LOOP FILTER TOPOLOGY
To use fast lock mode, the damping resistor in the loop filter
is reduced to ¼ of its value while in wide bandwidth mode. To
achieve the wider loop filter bandwidth, the charge pump
current increases by a factor of 16. To maintain loop stability,
the damping resistor must be reduced a factor of ¼. To enable
fast lock, the SW pin is shorted to the GND pin by setting
Bits[DB16:DB15] in Register 3 to values 0, 1. The following two
topologies are available:
ADF4151
R2
CPOUT
VCO
C1
C3
C2
R1
SW
Table 7. Fractional Spurs with Dither Off
Condition (Dither Off)
If MOD is divisible by 2, but not 3
If MOD is divisible by 3, but not 2
If MOD is divisible by 6
Otherwise
Repeat
Length
2 × MOD
3 × MOD
6 × MOD
MOD
Spur Interval
Channel step/2
Channel step/3
Channel step/6
Channel step
In low spur mode (dither on), the repeat length is extended to
221 cycles, regardless of the value of MOD, which makes the
quantization error spectrum look like broadband noise. This
may degrade the in-band phase noise at the PLL output by as
much as 10 dB. For lowest noise, dither off is a better choice,
particularly when the final loop bandwidth is low enough to
attenuate even the lowest frequency fractional spur.
Integer Boundary Spurs
R1A
10265-026
•
The damping resistor (R1) is divided into two values (R1
and R1A) that have a ratio of 1:3 (see Figure 26).
An extra resistor (R1A) is connected directly from SW,
as shown in Figure 27. The extra resistor is calculated
such that the parallel combination of an extra resistor
and the damping resistor (R1) is reduced to ¼ of the
original value of R1 (see Figure 27).
In low noise mode (dither off), the quantization noise from the
Σ-Δ modulator appears as fractional spurs. The interval between
spurs is fPFD/L, where L is the repeat length of the code sequence
in the digital Σ-Δ modulator. For the third-order modulator
used in the ADF4151, the repeat length depends on the value
of MOD, as listed in Table 7.
Figure 26. Fast Lock Loop Filter Topology—Topology 1
ADF4151
R2
CPOUT
VCO
C1
C2
R1A
R1
C3
SW
Another mechanism for fractional spur creation is the interactions
between the RF VCO frequency and the reference frequency.
When these frequencies are not integer related (the point of a
fractional-N synthesizer) spur sidebands appear on the VCO
output spectrum at an offset frequency that corresponds to the
beat note or difference frequency between an integer multiple of
the reference and the VCO frequency. These spurs are attenuated
by the loop filter and are more noticeable on channels close to
integer multiples of the reference where the difference frequency
can be inside the loop bandwidth; therefore, the name integer
boundary spurs.
10265-027
•
Fractional Spurs
Figure 27. Fast Lock Loop Filter Topology—Topology 2
Rev. B | Page 22 of 28
Data Sheet
ADF4151
Reference Spurs
Reference spurs are generally not a problem in fractional-N
synthesizers because the reference offset is far outside the loop
bandwidth. However, any reference feedthrough mechanism
that bypasses the loop can cause a problem. Feedthrough of low
levels of on-chip reference switching noise, through the RFIN
pin back to the VCO, can result in reference spur levels as high
as −90 dBc. PCB layout must ensure adequate isolation between
VCO traces and the input reference to avoid a possible
feedthrough path on the board.
SPUR CONSISTENCY AND FRACTIONAL SPUR
OPTIMIZATION
With dither off, the fractional spur pattern due to the quantization noise of the SDM also depends on the particular phase
word with which the modulator is seeded.
The phase word can be varied to optimize the fractional and
subfractional spur levels on any particular frequency. Thus, a
look-up table of phase values corresponding to each frequency
can be constructed for use when programming the ADF4151.
If a look-up table is not used, keep the phase word at a constant
value to ensure consistent spur levels on any particular frequency.
PHASE RESYNC
The output of a fractional-N PLL can settle to any one of the
MOD phase offsets with respect to the input reference, where
MOD is the fractional modulus. The phase resync feature in the
ADF4151 produces a consistent output phase offset with respect
to the input reference. This is necessary in applications where the
output phase and frequency are important, such as digital beam
forming. See the Phase Programmability section for how to
program a specific RF output phase when using phase resync.
Phase resync is enabled by setting Bit DB16, Bit DB15 in
Register 3 to 1, 0. When phase resync is enabled, an internal
timer generates sync signals at intervals of tSYNC given by the
following formula:
tSYNC = CLK_DIV_VALUE × MOD × tPFD
where:
CLK_DIV_VALUE is the decimal value programmed in
Bits[DB14:DB3] of Register 3 and can be any integer in the
range of 1 to 4095.
MOD is the modulus value programmed in Bits[DB14:DB3] of
Register 1 (R1).
tPFD is the PFD reference period.
When a new frequency is programmed, the second sync pulse
after the LE rising edge is used to resynchronize the output
phase to the reference. The tSYNC time must be programmed to
a value that is at least as long as the worst-case lock time. This
guarantees that the phase resync occurs after the last cycle slip
in the PLL settling transient.
In the example shown in Figure 28, the PFD reference is
25 MHz and MOD is 125 for a 200 kHz channel spacing. tSYNC
is set to 400 µs by programming the clock divider value,
CLK_DIV_VALUE, to 80.
LE
SYNC
(INTERNAL)
tSYNC
LAST CYCLE SLIP
FREQUENCY
PLL SETTLES TO
INCORRECT PHASE
PLL SETTLES TO
CORRECT PHASE
AFTER RESYNC
–100
0
100
200 300
400 500 600
TIME (µs)
700
800
900 1000
10265-028
PHASE
Figure 28. Phase Resync Example
Phase Programmability
The phase word in Register 1 controls the RF output phase. As
this word is swept from 0 to MOD, the RF output phase sweeps
over a 360° range in steps of 360°/MOD.
Rev. B | Page 23 of 28
ADF4151
Data Sheet
APPLICATIONS INFORMATION
DIRECT CONVERSION MODULATOR
The LO ports of the ADL5375 can be driven from the VCO
output. To ensure that all three RF ports (VCO output, RFIN and
LOIP) are connected to 50 Ω impedance, the matching network
of three 18 Ω resistors must be placed as in Figure 29. AC
coupling of the RF signal is implemented by the capacitors
connected in serial with the 18 Ω resistors . It is possible, as
well, to use a balun to convert from a single-ended LO input to
the differential LO inputs for the ADL5375.
Direct conversion architectures are increasingly being used to
implement base station transmitters. Figure 29 shows how Analog
Devices, Inc., parts can be used to implement such a system.
The circuit block diagram shows the AD9788 TxDAC® being
used with the ADL5375. The use of dual integrated DACs, such
as the AD9788 with its specified ±0.02 dB and ±0.004 dB gain
and offset matching characteristics, ensures minimum error
contribution (over temperature) from this portion of the
signal chain. The signal for the I channel of the quadrature
modulator is taken from the OUT1 differential outputs of the
AD9788, and the OUT2 differential outputs provide the signal
for the Q channel of the quadrature modulator ADL5375.
If the I and Q inputs are driven in quadrature by 2 V p-p
signals, the resulting output power from the modulator is
approximately 2 dBm.
The local oscillator (LO) is implemented using the ADF4151.
The low-pass filter was designed using ADIsimPLL™ for a channel
spacing of 200 kHz and a closed-loop bandwidth of 35 kHz.
51Ω
REFIO
51Ω
OUT1_P
MODULATED
DIGITAL
DATA
AD9788
LOW-PASS
FILTER
OUT1_N
TxDAC
OUT2_P
LOW-PASS
FILTER
OUT2_N
FSADJ
51Ω
51Ω
2kΩ
VDD
VP
ADL5375
IBBP
LOCK
DETECT
IBBN
10
16
AVDD1 AVDD2
17
AVDD2
4
28
32
6
CE DVDD SDV DD VP MUXOUT
1nF 1nF
51Ω
29 REF IN
1
CLK
2
DATA
3
LE
LD
100pF
RFIN+ 14
18Ω
18Ω
1nF
LOIN
18Ω
RFIN– 15
LOIP
QUADRATURE
PHASE
SPLITTER
RFOUT
1nF
100pF
100pF
DSOP
VVCO
ADF4151
QBBP
VCO OUT
VCC
QBBN
VTUNE
22 RSET
VCO
4.7kΩ
680Ω
CPOUT
7
SW
5
39nF
2700pF
CPGND
AGND
AGND
AGND
AGND
DGND
8
9
11
18
21
26
1200pF
360Ω
DGND SDGND
27
31
10265-029
SPI-COMPATIBLE SERIAL BUS
FREF IN
25
30
Figure 29. Direct Conversion Modulator
Rev. B | Page 24 of 28
Data Sheet
ADF4151
INTERFACING
Blackfin BF527 Interface
The ADF4151 has a simple SPI-compatible serial interface
for writing to the device. CLK, DATA, and LE control the data
transfer. When LE goes high, the 32 bits that have been clocked
into the appropriate register on each rising edge of CLK are
transferred to the appropriate latch. See Figure 2 for the timing
diagram and Table 5 for the register address table.
Figure 31 shows the interface between the ADF4151 and the
Blackfin ADSP-BF527 digital signal processor (DSP). The
ADF4151 needs a 32-bit serial word for each latch write. The
easiest way to accomplish this using the Blackfin family is to use
the autobuffered transmit mode of operation with alternate
framing. This provides a means for transmitting an entire block
of serial data before an interrupt is generated. Set up the word
length for eight bits and use four memory locations for each
32-bit word. To program each 32-bit latch, store the four 8-bit
bytes, enable the autobuffered mode, and write to the transmit
register of the DSP. This last operation initiates the autobuffer
transfer. As in the microcontroller case, just make sure that the
clock speeds are within the maximum limits outlined in Table 2.
ADuC812 Interface
Figure 30 shows the interface between the ADF4151 and the
ADuC812 MicroConverter®. Because the ADuC812 is based
on an 8051 core, this interface can be used with any 8051-based
microcontroller. The MicroConverter is set up for SPI master
mode with CPHA = 0. To initiate the operation, the I/O port
driving LE is brought low. Each latch of the ADF4151 needs a
32-bit word, which is accomplished by writing four 8-bit bytes
from the MicroConverter to the device. When the fourth byte
has been written, the LE input should be brought high to
complete the transfer.
MOSI
MOSI
DATA
LE
CE
I/O FLAGS
MUXOUT
(LOCK DETECT)
CLK
DATA
LE
I/O PORTS
CLK
10265-031
SCLOCK
SCLK
GPIO
ADF4151
ADuC812
ADF4151
ADSP-BF527
Figure 31. ADSP-BF527 to ADF4151 Interface
CE
PCB DESIGN GUIDELINES FOR CHIP SCALE
PACKAGE
10265-030
MUXOUT
(LOCK DETECT)
Figure 30. ADuC812 to ADF4151 Interface
I/O port lines on the ADuC812 are also used to control powerdown (CE input) and detect lock (MUXOUT configured as
lock detect and polled by the port input). When operating in
the described mode, the maximum SCLOCK rate of the
ADuC812 is 4 MHz. This means that the maximum rate at
which the output frequency can be changed is 125 kHz.
The lands on the chip scale package (CP-32-7) are rectangular.
The PCB pad for these must be 0.1 mm longer than the package
land length and 0.05 mm wider than the package land width.
The land is to be centered on the pad. This ensures that the
solder joint size is maximized. The bottom of the chip scale
package has a central thermal pad.
The thermal pad on the PCB must be at least as large as the
exposed pad. On the PCB, there is to be a minimum clearance
of 0.25 mm between the thermal pad and the inner edges of the
pad pattern. This ensures that shorting is avoided.
Thermal vias can be used on the PCB thermal pad to improve
the thermal performance of the package. If vias are used, they
are to be incorporated in the thermal pad at 1.2 mm pitch grid.
The via diameter must be between 0.3 mm and 0.33 mm, and
the via barrel must be plated with one ounce copper to plug
the via.
Rev. B | Page 25 of 28
ADF4151
Data Sheet
OUTLINE DIMENSIONS
0.30
0.25
0.18
32
25
0.50
BSC
1
24
TOP VIEW
0.80
0.75
0.70
8
16
0.05 MAX
0.02 NOM
COPLANARITY
0.08
0.20 REF
SEATING
PLANE
3.25
3.10 SQ
2.95
EXPOSED
PAD
17
0.50
0.40
0.30
PIN 1
INDICATOR
9
BOTTOM VIEW
0.25 MIN
FOR PROPER CONNECTION OF
THE EXPOSED PAD, REFER TO
THE PIN CONFIGURATION AND
FUNCTION DESCRIPTIONS
SECTION OF THIS DATA SHEET.
COMPLIANT TO JEDEC STANDARDS MO-220-WHHD.
112408-A
PIN 1
INDICATOR
5.10
5.00 SQ
4.90
Figure 32. 32-Lead Lead Frame Chip Scale Package [LFCSP_WQ]
5 mm × 5 mm Body, Very Thin Quad
(CP-32-7)
Dimensions shown in millimeters
ORDERING GUIDE
Model 1
ADF4151BCPZ
ADF4151BCPZ-RL7
EVAL-ADF4151EB1Z
1
Temperature Range
−40°C to +85°C
−40°C to +85°C
Package Description
32-Lead Lead Frame Chip Scale Package [LFCSP_WQ]
32-Lead Lead Frame Chip Scale Package [LFCSP_WQ]
Evaluation Board
Z = RoHS Compliant Part.
Rev. B | Page 26 of 28
Package Option
CP-32-7
CP-32-7
Data Sheet
ADF4151
NOTES
Rev. B | Page 27 of 28
ADF4151
Data Sheet
NOTES
©2011 Analog Devices, Inc. All rights reserved. Trademarks and
registered trademarks are the property of their respective owners.
D10265-0-12/11(B)
Rev. B | Page 28 of 28
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