950 MHz to 1575 MHz Quadrature Modulator
with Integrated Fractional-N PLL and VCO
ADRF6750
FEATURES
GENERAL DESCRIPTION
I/Q modulator with integrated fractional-N PLL and VCO
Gain control span: 47 dB in 1 dB steps
Output frequency range: 950 MHz to 1575 MHz
Output 1 dB compression: 8.5 dBm
Output IP3: 23 dBm
Noise floor: −162 dBm/Hz
Baseband modulation bandwidth: 250 MHz (1 dB)
Output frequency resolution: 1 Hz
Functions with external VCO for extended frequency range
SPI and I2C-compatible serial interfaces
Power supply: 5 V/310 mA
The ADRF6750 is a highly integrated quadrature modulator,
frequency synthesizer, and programmable attenuator. The
device covers an operating frequency range from 950 MHz
to 1575 MHz for use in satellite, cellular and broadband
communications.
The ADRF6750 modulator includes a high modulus fractional-N
frequency synthesizer with integrated VCO, providing better
than 1 Hz frequency resolution, and a 47 dB digitally controlled
output attenuator with 1 dB steps.
Control of all the on-chip registers is through a user-selected
SPI interface or I2C interface. The device operates from a single
power supply ranging from 4.75 V to 5.25 V.
FUNCTIONAL BLOCK DIAGRAM
VCC1
REGOUT
VREG1
VREG2
VREG3
VREG4
VREG5
VREG6
LOMONP
LOMONN
VCC2
VCC3
VCC4
3.3V
REGULATOR
IBBP
IBBN
CCOMP1
CCOMP2
CCOMP3
47dB
GAIN CONTROL
RANGE
0°/90°
OUTPUT
STAGE
VCO
CORE
RFOUT
VTUNE
TESTLO
TESTLO
TXDIS
QBBP
QBBN
RSET
REFERENCE
REFIN
×2
DOUBLER
5-BIT
DIVIDER
÷2
+
PHASE
FREQUENCY
– DETECTOR
REFIN
CHARGE
PUMP
CURRENT SETTING
N-COUNTER
SDI/SDA
CLK/SCL
SDO
CS
SPI/
I2C
INTERFACE
THIRD-ORDER
FRACTIONAL
INTERPOLATOR
FRACTIONAL
REGISTER
MODULUS
225
RFCP4 RFCP3 RFCP2 RFCP1
CP
LF3
LF2
LDET
INTEGER
REGISTER
AGND
08201-001
ADRF6750
DGND
Figure 1.
Rev. A
Information furnished by Analog Devices is believed to be accurate and reliable. However, no
responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other
rights of third parties that may result from its use. Specifications subject to change without notice. No
license is granted by implication or otherwise under any patent or patent rights of Analog Devices.
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One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 781.329.4700
www.analog.com
Fax: 781.461.3113
©2010 Analog Devices, Inc. All rights reserved.
ADRF6750
TABLE OF CONTENTS
Features .............................................................................................. 1
I2C Interface ................................................................................ 21
General Description ......................................................................... 1
SPI Interface ................................................................................ 23
Functional Block Diagram .............................................................. 1
Program Modes .......................................................................... 25
Revision History ............................................................................... 2
Register Map ................................................................................... 27
Specifications..................................................................................... 3
Register Map Summary ............................................................. 27
Timing Characteristics ................................................................ 5
Register Bit Descriptions ........................................................... 28
Absolute Maximum Ratings............................................................ 7
Suggested Power-Up Sequence ..................................................... 31
ESD Caution .................................................................................. 7
Initial Register Write Sequence ................................................ 31
Pin Configuration and Function Descriptions ............................. 8
Evaluation Board ............................................................................ 32
Typical Performance Characteristics ........................................... 10
General Description ................................................................... 32
Theory of Operation ...................................................................... 18
Hardware Description ............................................................... 32
Overview...................................................................................... 18
PCB Artwork............................................................................... 35
PLL Synthesizer and VCO ......................................................... 18
Bill of Materials ........................................................................... 38
Quadrature Modulator .............................................................. 20
Outline Dimensions ....................................................................... 39
Attenuator .................................................................................... 21
Ordering Guide .......................................................................... 39
Voltage Regulator ....................................................................... 21
EXTERNAL vco OPERATION ................................................ 21
REVISION HISTORY
4/10—Rev. 0 to Rev. A
Changes to Table 5 ............................................................................ 9
Changes to LOMON Outputs Section ......................................... 33
Changes to Ordering Guide .......................................................... 39
1/10—Revision 0: Initial Version
Rev. A | Page 2 of 40
ADRF6750
SPECIFICATIONS
VCC = 5 V, TA = 25°C, I/Q inputs = 0.9 V p-p differential sine waves in quadrature on a 500 mV dc bias, baseband frequency = 1 MHz,
REFIN = 10 MHz, PFD = 20 MHz, loop bandwidth = 50 kHz, and LOMONx is off, unless otherwise noted.
Table 1.
Parameter
RF OUTPUT
Operating Frequency Range
Nominal Output Power
Gain Flatness
Output P1dB
Output IP3
Output Return Loss
LO Carrier Feedthrough
2× LO Carrier Feedthrough
Sideband Suppression
Noise Floor
Harmonics
REFERENCE CHARACTERISTICS
Input Frequency
Input Sensitivity
Input Capacitance
Input Current
CHARGE PUMP
ICP Sink/Source
High Value
Low Value
Absolute Accuracy
RSET Value
VCO Gain
SYNTHESIZER SPECIFICATIONS
Frequency Resolution
Spurs
Phase Noise 1
Integrated Phase Noise1
Frequency Settling1
Maximum Frequency Step for
No Autocalibration
Phase Detector Frequency
Test Conditions/Comments
RFOUT pin
Min
Typ
950
VIQ = 0.9 V p-p differential
Any 40 MHz
Unit
1575
MHz
dBm
dB
dBm
dBm
dB
dBc
dBm
dBc
dBm/Hz
dBc/Hz
dBm/Hz
dBc
300
165
VREG
10
±100
MHz
MHz
V p-p
pF
μA
−1.6
±0.5
8.5
23
−12
−45
−45
−45
−162
−147
−170
−60
f1BB = 3.5 MHz, f2BB = 4.5 MHz, POUT = −6 dBm per tone
Attenuator setting = 0 dB
Attenuator setting = 0 dB to 47 dB
Attenuator setting = 0 dB to 47 dB
I/Q inputs = 0 V p-p differential, Attenuator setting = 0 dB
Attenuator setting = 0 dB to 21 dB, carrier offset = 15 MHz
Attenuator setting = 21 dB to 47 dB, carrier offset = 15 MHz
REFIN pin
With R/2 divider enabled
With R/2 divider disabled
AC-coupled
Max
10
10
0.4
Programmable
With RSET = 4.7 kΩ
5
312.5
4.0
4.7
25
With RSET = 4.7 kΩ
KVCO
mA
μA
%
kΩ
MHz/V
1
Integer boundary < loop bandwidth
>10 MHz offset from carrier
Frequency = 950 MHz to 1575 MHz
100 Hz offset
1 kHz offset
10 kHz offset
100 kHz offset
1 MHz offset
>15 MHz offset
1 kHz to 8 MHz integration bandwidth
Maximum frequency error = 100 Hz
Frequency step with no autocalibration routine;
Register CR24, Bit 0 = 1
−55
−85
−80
−88
−93
−107
−133
−152
0.4
170
10
Rev. A | Page 3 of 40
Hz
dBc
dBc
100
dBc/Hz
dBc/Hz
dBc/Hz
dBc/Hz
dBc/Hz
dBc/Hz
°rms
μs
kHz
30
MHz
ADRF6750
Parameter
GAIN CONTROL
Gain Range
Step Size
Relative Step Accuracy
Absolute Step Accuracy 2
Output Settling Time
OUTPUT DISABLE
Off Isolation
Turn-On Settling Time
Turn-Off Settling Time
MONITOR OUTPUT
Nominal Output Power
BASEBAND INPUTS
I and Q Input Bias Level
1 dB Bandwidth
LOGIC INPUTS
Input High Voltage, VINH
Input Low Voltage, VINL
Input High Voltage, VINH
Input Low Voltage, VINL
Input Current, IINH/IINL
Input Capacitance, CIN
LOGIC OUTPUTS
Output High Voltage, VOH
Output Low Voltage, VOL
POWER SUPPLIES
Voltage Range
Supply Current
Operating Temperature
1
2
Test Conditions/Comments
Min
Fixed frequency, adjacent steps
All attenuation steps
Over full frequency range, adjacent steps
47 dB attenuation step
Any step; output power settled to ±0.2 dB
TXDIS pin
RF OUT, attenuator setting = 0 dB to 47 dB, TXDIS high
LO, Attenuator setting = 0 dB to 47 dB, TXDIS high
2 x LO, Attenuator setting = 0 dB to 47 dB, TXDIS high
TXDIS high to low (90% of envelope)
TXDIS low to high (to −55 dBm)
LOMONP, LOMONN pins
Typ
Max
Unit
47
1
dB
dB
±0.3
±1.5
−2.0
10
dB
dB
dB
μs
−110
−90
−50
180
270
dBm
dBm
dBm
ns
ns
−24
dBm
500
250
mV
MHz
IBBP, IBBN, QBBP, QBBN pins
CS, TXDIS pins
CS, TXDIS pins
SDI/SDA, CLK/SCL pins
SDI/SDA, CLK/SCL pins
CS, TXDIS, SDI/SDA, CLK/SCL pins
CS, TXDIS, SDI/SDA, CLK/SCL pins
1.4
Rev. A | Page 4 of 40
0.4
0.4
V
V
V
2.1
SDO, LDET pins; IOH = 500 μA
2.8
SDO, LDET pins; IOL = 500 μA
SDA (SDI/SDA); IOL = 3 mA
VCC1, VCC2, VCC3, VCC4, VREG1, VREG2, VREG3, VREG4,
VREG5, VREG6, and REGOUT pins
REGOUT normally connected to VREG1, VREG2, VREG3,
VREG4, VREG5, and VREG6
VCC1, VCC2, VCC3, and VCC4
4.75
REGOUT, VREG1, VREG2, VREG3, VREG4, VREG5, and VREG6
VCC1, VCC2, VCC3, and VCC4 combined; REGOUT connected to VREG1, VREG2, VREG3, VREG4, VREG5, and VREG6
−40
LBW = 50 kHz at LO = 1200 MHz; ICP = 2.5 mA.
All other attenuation steps have an absolute error of 10MHz OFFSET FREQUENCY (dBc)
–60
–70
–90
–100
–110
–120
–130
–140
–150
–70
–80
–90
–100
–110
1600
1625
LO FREQUENCY (MHz)
Figure 32. Spurs > 10 MHz from Carrier vs. LO Frequency,
Supply, and Temperature
Figure 29. Phase Noise Performance Distribution at Worst-Case Conditions
0.50
–40
+25°C; 5.00V
0.45
+85°C; 4.75V
–45
0.40
+85°C; 5.25V
–40°C; 4.75V
–50
RMS JITTER (Degrees)
INTEGER BOUNDARY SPUR (dBc)
08201-127
1500
1400
1300
1200
100M
1100
10M
10k
100k
1M
OFFSET FREQUENCY (Hz)
1000
1k
08201-130
–120
–160
100
900
PHASE NOISE (dBc/Hz)
–80
PFD SPURS AT 20MHz OFFSET
REFERENCE SPURS AT 10MHz OFFSET
–40°C; 5.25V
–55
–60
–65
0.35
0.30
0.25
0.20
0.15
0.10
–70
0.05
LO FREQUENCY (MHz)
Figure 30. Integer Boundary Spur Performance vs. LO Frequency,
Supply, and Temperature
08201-131
1550
1575
1500
1450
1400
1350
1300
1250
1200
1150
1100
1050
1000
950
1550
1575
LO FREQUENCY (MHz)
08201-125
1450
1350
1250
1150
1050
950
0
Figure 33. Integrated Phase Noise vs. LO Frequency at
Nominal Conditions
80
60
NOMINAL
WORST CASE
70
50
NOMINAL
WORST CASE
OCCURENCE (%)
50
40
30
40
30
20
20
0
–85
–80
–75
–70
–65
–60
–55
–50
–45
–40
INTEGER BOUNDARY SPURS (dBc)
Figure 31. Integer Boundary Spur Distribution at Nominal
and Worst-Case Conditions
0
0.275 0.300 0.325 0.350 0.375 0.400 0.425 0.450 0.475 0.500
RMS JITTER (Degrees)
Figure 34. Integrated Phase Noise at Nominal and
Worst-Case Conditions
Rev. A | Page 14 of 40
08201-137
10
10
08201-126
OCCURENCE (%)
60
ADRF6750
100M
45
10M
40
1M
35
OCCURENCE (%)
50
100k
ACQUISITION
TO 100Hz
10k
START OF ACQUISITION
ON CR0 WRITE
1k
30
25
20
10
50
–5
45
–10
40
–15
35
ATTENUATOR RELATIVE STEP ACCURACY (dB)
0.8
0.6
0.4
0.2
0
–0.2
–0.4
–0.6
–0.8
2.25
08201-140
2.00
1.75
0.5
0.3
0.1
–0.1
–0.3
–0.5
–0.7
–0.9
–1.1
–1.3
Figure 37. Attenuator Relative Step Accuracy over all Attenuation Steps
vs. LO Frequency, Nominal Conditions
2000
1900
1800
1700
08201-136
LO FREQUENCY (MHz)
1600
1500
1400
1300
1200
1100
500
–1.5
08201-134
1550
1575
1500
1450
1400
1350
1300
1250
1150
1200
1100
–1.0
1050
1.50
Figure 39. Attenuator Relative Step Accuracy Across Full Output
Frequency Range Distribution at Nominal and Worst-Case Conditions
1.0
1000
1.25
ATTENUATOR RELATIVE STEP ACCURACY ACROSS
FULL OUTPUT FREQUENCY RANGE (dB)
Figure 36. Attenuator Gain vs. LO Frequency by Gain Code,
All Attenuator Code Steps
950
1.00
–2.25
08201-133
1550
1575
1500
1450
1400
1350
1250
1300
1200
1100
0
1150
5
–50
1050
–45
1000
10
950
15
–40
0.75
20
–35
LO FREQUENCY (MHz)
08201-135
25
0.50
–30
30
0
–25
0.25
–20
NOMINAL
WORST CASE
–0.25
OCCURENCE (%)
0
LO FREQUENCY (MHz)
1.0
Figure 38. Attenuator Relative Step Accuracy Distribution at Nominal
and Worst-Case Conditions
–0.50
Figure 35. PLL Frequency Settling Time at Worst-Case Low Frequency
with Lock Detect Shown
–0.75
TIME (µs)
0
–1.0 –0.8 –0.6 –0.4 –0.2
0
0.2
0.4
0.6
0.8
ATTENUATOR RELATIVE STEP ACCURACY (dB)
–1.00
100 125 150 175 200 225 250
1000
75
–1.25
50
900
25
–1.50
0
800
0.1
–50 –25
5
–1.75
CR23[3] = 0
700
LDET
–2.00
CR23[3] = 1
08201-132
LDET
1
OUPTUT POWER (dBm)
WORST CASE
15
100
10
ATTENUATOR RELATIVE STEP ACCURACY (dB)
NOMINAL
600
FREQUENCY ERROR (Hz)
1G
Figure 40. Attenuator Relative Step Accuracy over all Attenuation Steps
vs. LO Frequency for External VCO Mode, Nominal Conditions
Rev. A | Page 15 of 40
0.5
1.0
0.8
GAIN FLATNESS IN ANY 40MHz (dB)
0
–0.5
–1.0
–1.5
–2.0
–2.5
0.4
0.2
0
–0.2
–0.4
–0.6
LO FREQUENCY (MHz)
1550
1575
08201-149
1500
1450
1400
1350
1300
1250
1150
1200
1100
1050
950
08201-139
1550
1575
1500
1450
1400
1350
1300
1250
1150
1200
1100
1050
1000
1000
1.0
LO FREQUENCY (MHz)
Figure 44. Gain Flatness in any 40 MHz for all Attenuation Steps vs.
LO Frequency at Nominal Conditions
Figure 41. Attenuator Absolute Step Accuracy over all Attenuation Steps
vs. LO Frequency, Nominal Conditions
5.0
70
NOMINAL
WORST CASE
60
4.5
4.0
SETTLING TIME (µs)
50
OCCURENCE (%)
0.6
–0.8
–3.0
950
ATTENUATOR ABSOLUTE STEP ACCURACY (dB)
ADRF6750
40
30
20
SETTLING TIME TO 0.2dB
SETTLING TIME TO 0.5dB
3.5
3.0
INCREASING STEP SIZE
2.5
2.0
1.5
1.0
10
–0.4
–0.6
–0.8
–1.0
–1.2
–1.4
–1.6
–1.8
–2.0
–2.2
–2.4
–2.6
–2.8
–3.0
–3.2
–3.4
ATTENUATOR ABSOLUTE STEP ACCURACY (dB)
08201-138
0
0
Figure 45. Attenuator Settling Time to 0.2 dB and 0.5 dB for Small Steps
(1 dB to 6 dB) at Nominal Conditions
20
1.5
18
1.0
SETTLING TIME TO 0.2dB
SETTLING TIME TO 0.5dB
16
SETTLING TIME (µs)
0.5
0
–0.5
–1.0
–1.5
14
12
INCREASING STEP SIZE
10
8
6
4
–2.0
0
2000
1900
1800
1700
7dB TO 47dB ATTENUATOR STEP SIZES
08201-142
LO FREQUENCY (MHz)
1600
1500
1400
1300
1200
1100
1000
900
800
700
600
–2.5
Figure 43. Attenuator Absolute Step Accuracy over all Attenuation Steps
vs. LO Frequency for External VCO Mode, Nominal Conditions
08201-144
2
500
ATTENUATOR ABSOLUTE STEP ACCURACY (dB)
Figure 42. Attenuator Absolute Step Accuracy Distribution at Nominal
and Worst-Case Conditions
1dB TO 6dB ATTENUATOR STEP SIZES
08201-143
0.5
Figure 46. Attenuator Settling Time to 0.2 dB and 0.5 dB for Large Steps
(7 dB to 47 dB) at Nominal Conditions
Rev. A | Page 16 of 40
ADRF6750
100
80
NOMINAL SETTLING TIME TO 0.2dB
NOMINAL SETTLING TIME TO 0.5dB
WORST-CASE SETTLING TIME TO 0.2dB
WORST-CASE SETTLING TIME TO 0.5dB
90
80
70
NOMINAL SETTLING TIME TO 0.2dB
NOMINAL SETTLING TIME TO 0.5dB
WORST-CASE SETTLING TIME TO 0.2dB
WORST-CASE SETTLING TIME TO 0.5dB
60
OCCURENCE (%)
OCCURENCE (%)
70
60
50
40
50
40
30
30
20
20
10
10
0.5
1.0
1.5
2.0
2.5
3.0
3.5
4.0
4.5
5.0
ATTENUATOR SETTLING TIME (µs)
Figure 47. Attenuator Settling Time to 0.2 dB and 0.5 dB Distribution
at Nominal and Worst-Case Conditions for Typical Small Step
0
3
6
9
12
15
18
21
24
27
30
ATTENUATOR SETTLING TIME (µs)
08201-148
0
08201-146
0
0
Figure 50. Attenuator Settling Time to 0.2 dB and 0.5 dB Distribution at
Nominal and Worst-Case Conditions for Worst-Case Large Step
(47 dB to 0 dB)
100
0
90
–10
OUTPUT POWER (dBm)
80
OCCURENCE (%)
70
60
NOMINAL SETTLING TIME TO 0.2dB
NOMINAL SETTLING TIME TO 0.5dB
WORST-CASE SETTLING TIME TO 0.2dB
WORST-CASE SETTLING TIME TO 0.5dB
50
40
30
–20
TURN-ON = 180ns
–30
TURN-OFF = 270ns
–40
–50
20
–60
10
2
4
6
8
10
12
14
16
ATTENUATOR SETTLING TIME (µs)
18
20
08201-145
0
–70
0
100
90
80
NOMINAL SETTLING TIME TO 0.2dB
NOMINAL SETTLING TIME TO 0.5dB
WORST-CASE SETTLING TIME TO 0.2dB
WORST-CASE SETTLING TIME TO 0.5dB
OCCURENCE (%)
60
50
40
30
20
0
2
4
6
8
10
12
14
16
ATTENUATOR SETTLING TIME (µs)
18
20
08201-147
10
0
1.0
1.5
2.0
2.5
3.0
3.5
4.0
4.5
5.0
TXDIS SETTLING TIME (µs)
Figure 48. Attenuator Settling Time to 0.2 dB and 0.5 dB Distribution
at Nominal and Worst-Case Conditions for Worst-Case Small Step
(36 dB to 42 dB)
70
0.5
Figure 49. Attenuator Settling Time to 0.2 dB and 0.5 dB Distribution at
Nominal and Worst-Case Conditions for Typical Large Step (0 dB to 47 dB)
Rev. A | Page 17 of 40
Figure 51. TXDIA Turn-On Settling Time at Worst-Case Supply
and Temperature
08201-151
TXDIS
0
ADRF6750
OVERVIEW
The ADRF6750 device can be divided into the following basic
building blocks:
TO
PFD
÷2
Figure 53. Reference Input Path
fPFD = fREFIN × [(1 + D)/(R × (1 + T))]
(2)
where:
fREFIN is the reference input frequency.
D is the doubler bit.
R is the programmed divide ratio of the binary 5-bit
programmable reference divider (1 to 32).
T is the divide-by-2 bit (0 or 1).
Each of these building blocks is described in detail in the
sections that follow.
PLL SYNTHESIZER AND VCO
RF Fractional-N Divider
Overview
The phase-locked loop (PLL) consists of a fractional-N frequency
synthesizer with a 25-bit fixed modulus, allowing a frequency
resolution of less than 1 Hz over the entire frequency range. It
also has an integrated voltage-controlled oscillator (VCO) with
a fundamental output frequency ranging from 1900 MHz to
3150 MHz. This allows the PLL to generate a stable frequency at
2× LO, which is then divided down to provide a local oscillator
(LO) frequency ranging from 950 MHz to 1575 MHz to the
quadrature modulator.
Reference Input Section
The reference input stage is shown in Figure 52. SW1 and SW2
are normally closed switches. SW3 is normally open. When
power-down is initiated, SW3 is closed, and SW1 and SW2 are
open. This ensures that there is no loading of the REFIN pin at
power-down.
POWER-DOWN
CONTROL
100kΩ
TO
R-DIVIDER
SW2
REFIN NC
5-BIT
R-DIVIDER
The PFD frequency equation is
PLL synthesizer and VCO
Quadrature modulator
Attenuator
Voltage regulator
I2C/SPI interface
NC
×2
DOUBLER
The RF fractional-N divider allows a division ratio in the PLL
feedback path that can range from 23 to 4095. The relationship
between the fractional-N divider and the LO frequency is
described in the following section.
INT and FRAC Relationship
The integer (INT) and fractional (FRAC) values make it
possible to generate output frequencies that are spaced by
fractions of the phase frequency detector (PFD) frequency.
See the Example—Changing the LO Frequency section for
more information.
The LO frequency equation is
LO = fPFD × (INT + (FRAC/225))
where:
LO is the local oscillator frequency.
fPFD is the PFD frequency.
INT is the integer component of the required division factor
and is controlled by the CR6 and CR7 registers.
FRAC is the fractional component of the required division
factor and is controlled by the CR0 to CR3 registers.
BUFFER
RF N-DIVIDER
NC
08201-006
SW1
SW3
(1)
FROM VCO
OUTPUT
DIVIDERS
N = INT + FRAC/225
TO
PFD
N-COUNTER
THIRD-ORDER
FRACTIONAL
INTERPOLATOR
Figure 52. Reference Input Stage
Reference Input Path
The on-chip reference frequency doubler allows the input
reference signal to be doubled. This is useful for increasing the
PFD comparison frequency. Making the PFD frequency higher
improves the noise performance of the system. Doubling the
PFD frequency usually improves the in-band phase noise
performance by 3 dBc/Hz.
The 5-bit R-divider allows the input reference frequency
(REFIN) to be divided down to produce the reference clock
to the PFD. Division ratios from 1 to 32 are allowed.
An additional divide-by-2 function in the reference input path
allows for a greater division range.
INT
REG
FRAC
VALUE
08201-007
•
•
•
•
•
FROM
REFIN
PIN
08201-008
THEORY OF OPERATION
Figure 54. RF Fractional-N Divider
Phase Frequency Detector (PFD) and Charge Pump
The PFD takes inputs from the R-divider and the N-counter and
produces an output proportional to the phase and frequency difference between them (see Figure 55 for a simplified schematic).
The PFD includes a fixed delay element that sets the width of
the antibacklash pulse, ensuring that there is no dead zone in
the PFD transfer function.
Rev. A | Page 18 of 40
ADRF6750
HI
D1
Q1
UP
The autocalibration time is set to 50 μs. During this time, the
VCO VTUNE is disconnected from the output of the loop filter
and is connected to an internal reference voltage. A typical
frequency acquisition is shown in Figure 57.
U1
+IN
CLR1
DELAY
CHARGE
PUMP
U3
1G
CP
CLR2
DOWN
D2
Q2
08201-009
U2
–IN
Figure 55. PFD Simplified Schematic
Lock Detect (LDET)
LDET (Pin 44) signals when the PLL has achieved lock to an
error frequency of less than 100 Hz. On a write to Register CR0,
a new PLL acquisition cycle starts, and the LDET signal goes
low. When lock has been achieved, this signal returns high.
10M
1M
AUTOCAL
TIME (µs)
100k
10k
ACQUISITION TO 100Hz
1k
100
10
0
20 40 60 80 100 120 140 160 180 200 220 240 260 280 300
TIME (µs)
08201-158
HI
FREQUENCY ERROR (Hz)
100M
Figure 57. PLL Acquisition
Voltage-Controlled Oscillator (VCO)
The VCO core in the ADRF6750 consists of two separate VCOs,
each with 16 overlapping bands. Figure 56 shows an acquisition
plot demonstrating both the VCO overlap at roughly 1260 MHz
and the multiple overlapping bands within each VCO. The
choice of two 16-band VCOs allows a wide frequency range to
be covered without a large VCO sensitivity (KVCO) and resultant
poor phase noise and spurious performance. Note that the VCO
range is larger than the 2× LO frequency range of the part to
ensure that the device has enough margin to cover the full
frequency range over all conditions.
After autocalibration, normal PLL action resumes and the
correct frequency is acquired to within a frequency error of
100 Hz in 170 μs typically.
For a maximum cumulative step of 100 kHz, autocalibration
can be turned off by Register CR24, Bit 0. This enables cumulative PLL acquisitions of 100 kHz or less to occur without the
autocalibration procedure, which improves acquisition times
significantly (see Figure 58).
100k
2.5
FREQUENCY ERROR (Hz)
2.3
2.1
VTUNE (V)
1.9
1.7
1.5
1.3
10k
1k
ACQUISITION TO 100Hz
100
1.1
0.5
800
10
08201-057
0.7
900
1000
1100
1200
1300
1400
1500
1600
0
50
100
TIME (µs)
1700
150
200
08201-159
0.9
Figure 58. PLL Acquisition Without Autocalibration for 100 kHz Step
LO FREQUENCY (MHz)
Figure 56. VTUNE vs. LO Frequency
The correct VCO and band are chosen automatically by the
VCO and band select circuitry when Register CR0 is updated.
This is referred to as autocalibration.
The VCO displays a variation of KVCO as VTUNE varies within
the band and from band to band. Figure 59 shows how the
KVCO varies across the full LO frequency range. Also shown
is the average value for each of the frequency bands. Figure 59
is useful when calculating the loop filter bandwidth and
individual loop filter components.
Rev. A | Page 19 of 40
ADRF6750
40
CURRENT OUTPUT DAC
(EXAMPLE: AD9779)
ADRF6750
35
IBBP
50Ω
25
50Ω
OUT1_N
IBBN
20
OUT2_N
15
QBBN
50Ω
10
5
08201-013
50Ω
OUT2_P
QBBP
Figure 61. Establishing DC Bias Level on Baseband Inputs
LO FREQUENCY (MHz)
08201-160
1550
1575
1450
1350
1250
1150
1050
950
0
Figure 59. KVCO vs. LO Frequency
QUADRATURE MODULATOR
Overview
A basic block diagram of the ADRF6750 quadrature modulator
circuit is shown in Figure 60. The VCO generates a signal at the
2× LO frequency, which is then divided down to give a signal at the
LO frequency. This signal is then split into in-phase and quadrature
components to provide the LO signals that drive the mixers.
The differential baseband inputs (QBBP, QBBN, IBBN, and
IBBP) consist of the bases of PNP transistors, which present
a high impedance of about 30 kΩ in parallel with roughly 2 pF
of capacitance. The impedance looks like 30 kΩ below 1 MHz
and starts to roll off at higher frequency. A 100 Ω differential
termination is recommended at the baseband inputs, and this
dominates the input impedance as seen by the input baseband
signal. This ensures that the input impedance, as seen by the
input circuit, remains flat across the baseband bandwidth. See
Figure 62 for a typical configuration.
CURRENT OUTPUT DAC
(EXAMPLE: AD9779)
V-TO-I
IBBP
IBBN
ADRF6750
OUT1_P
IBBP
50Ω
VCO
50Ω
LOWPASS
FILTER
100Ω
OUT1_N
RFOUT TO
ATTENUATOR
BALUN
QUAD
PHASE
SPLITTER
IBBN
÷2
OUT2_N
QBBN
50Ω
V-TO-I
08201-012
QBBP
QBBN
50Ω
OUT2_P
LOWPASS
FILTER
100Ω
QBBP
08201-014
VCO SENSITIVITY (MHz/V)
OUT1_P
30
Figure 60. Block Diagram of the Quadrature Modulator
Figure 62. Typical Baseband Input Configuration
The I and Q baseband input signals are converted to currents by
the V-to-I stages, which then drive the two mixers. The outputs
of these mixers combine to feed the output balun, which provides a
single-ended output. This single-ended output is then fed to the
attenuator and, finally, to the external RFOUT signal pin.
The swing of the AD9779 output currents ranges from 0 mA to
20 mA. The ac voltage swing is 1 V p-p single-ended or 2 V p-p
differential with the 50 Ω resistors in place. The 100 Ω differential termination resistors at the baseband inputs have the effect
of limiting this swing without changing the dc bias condition of
500 mV. The low-pass filter is used to filter the DAC outputs
and remove images when driving a modulator.
Baseband Inputs
The baseband inputs, QBBP, QBBN, IBBP, and IBBN, must be
driven from a differential source. The nominal drive level of
0.9 V p-p differential (450 mV p-p on each pin) should be
biased to a common-mode level of 500 mV dc.
To set the dc bias level at the baseband inputs, refer to Figure 61.
The average output current on each of the AD9779 outputs is
10 mA. A current of 10 mA flowing through each of the 50 Ω
resistors to ground produces the desired dc bias of 500 mV at
each of the baseband inputs.
Another consideration is that the baseband inputs actually
source a current of 240 μA out of each of the four inputs. This
current must be taken into account when setting up the dc bias
of 500 mV. In the initial example based on Figure 61, an error
of 12 mV occurs due to the 240 μA current flowing through
the 50 Ω resistor. Analog Devices, Inc., recommends that the
accuracy of the dc bias should be 500 mV ±25 mV. It is also
important that this 240 μA current have a dc path to ground.
Rev. A | Page 20 of 40
ADRF6750
Optimization
The carrier feedthrough and the sideband suppression performance of the ADRF6750 can be improved over the numbers
specified in Table 1 by using the following optimization
techniques.
Carrier Feedthrough Nulling
Carrier feedthrough results from dc offsets that occur between
the P and N inputs of each of the differential baseband inputs.
Normally these inputs are set to a dc bias of approximately 500 mV.
However, if a dc offset is introduced between the P and N inputs of
either or both I and Q inputs, the carrier feedthrough is affected
in either a positive or a negative fashion. Note that the dc bias
level remains at 500 mV (average P and N level). The I channel
offset is often held constant while the Q channel offset is varied
until a minimum carrier feedthrough level is obtained. Then,
while retaining the new Q channel offset, the I channel offset is
adjusted until a new minimum is reached. This is usually performed at a single frequency and, thus, is not optimized over
the complete frequency range. Multiple optimizations at different
frequencies must be performed to ensure optimum carrier feedthrough across the full frequency range.
Sideband Suppression Nulling
Sideband suppression results from relative gain and relative
phase offsets between the I channel and Q channel and can
be optimized through adjustments to those two parameters.
Adjusting only one parameter improves the sideband suppression
only to a point. For optimum sideband suppression, an iterative
adjustment between phase and amplitude is required.
ATTENUATOR
The digital attenuator consists of six attenuation blocks: 1 dB,
2 dB, 4 dB, 8 dB, and two 16 dB blocks; each is separately
controlled. Each attenuation block consists of field effect
transistor (FET) switches and resistors that form either a pishaped or a T-shaped attenuator. By controlling the states of the
FET switches through the control lines, each attenuation block
can be set to the pass state (0 dB) or the attenuation state (n dB).
The various combinations of the six blocks provide the
attenuation states from 0 dB to 47 dB in 1 dB increments.
VOLTAGE REGULATOR
The voltage regulator is powered from a 5 V supply that is
provided by VCC1 (Pin 11) and produces a 3.3 V nominal
regulated output voltage, REGOUT, on Pin 12. This pin must
be connected (external to the IC) to the VREG1 through VREG6
package pins.
The regulator output (REGOUT) should be decoupled by
a parallel combination of 10 pF and 220 μF capacitors. The
220 μF capacitor, which is recommended for best performance,
decouples broadband noise, leading to better phase noise. Each
VREGx pin should have the following decoupling capacitors:
100 nF multilayer ceramic with an additional 10 pF in parallel,
both placed as close as possible to the DUT power supply pins.
X7R or X5R capacitors are recommended. See the Evaluation
Board section for more information.
EXTERNAL VCO OPERATION
The ADRF6750 can be operated with an external VCO. This
can be useful if the user wants to improve the phase noise
performance or extend the frequency range. Note that the
external VCO needs to operate at a frequency of 2× LO.
To operate the ADRF6750 with an external VCO, follow
these steps:
1.
Connect the charge pump output (Pin 9) to the loop filter
and onward to the external VCO input.
The KVCO of the external VCO needs to be taken into
account when calculating the loop bandwidth and loop
filter components. Note that a 50 kHz loop bandwidth is
recommended when using the internal VCO. This takes
into account the phase noise performance of the internal
VCO. It is possible for an external VCO to provide better
phase noise performance and a 50 kHz loop bandwidth
may not be optimal in that case. When selecting a loop
bandwidth, consider rms jitter, phase noise performance,
and acquisition time. ADISimPLL™ can be used to optimize the loop bandwidth with a variety of external VCOs.
2. Connect the output of the external VCO to the TESTLO
and TESTLO input pins.
It is likely that a low-pass filter will be needed to filter the
output of the external VCO. This is very important if the
external VCO has poor second harmonic performance.
Second harmonic performance directly impacts sideband
suppression performance. For example, −30 dBc second
harmonic performance leads to −30 dBc sideband suppression. Both TESTLO and TESTLO need to be dc biased. A
dc bias of 1.7 V to 3.3 V is recommended. The REGOUT
output provides a 3.3 V output voltage.
3. Select external VCO operation by setting the following bits:
• Set Register CR27[3] = 1. This bit multiplexes the
TESTLO and TESTLO through to the quadrature
modulator.
• Set Register CR28[5] = 1. This bit powers down the
internal VCO and connects the external VCO to
the PLL.
4. Set the correct polarity for the PFD based on the slope of
the KVCO. The default is for positive polarity. This bit is
accessed by Register CR12[3].
When selecting an external VCO, at times it is difficult to select
one with an appropriate frequency range and KVCO. One solution may be the ADF4350, which can function as VCO only
with a range of 137.5 MHz to 4.4 GHz. Note that the ADF4350
requires an autocalibration time of 100 μs which directly
impacts acquisition time.
I2C INTERFACE
The ADRF6750 supports a 2-wire, I2C-compatible serial bus
that drives multiple peripherals. The serial data (SDA) and serial
Rev. A | Page 21 of 40
ADRF6750
first byte indicates that the master writes information to the
peripheral. Logic 1 on the LSB of the first byte indicates that the
master reads information from the peripheral.
clock (SCL) inputs carry information between any devices that
are connected to the bus. Each slave device is recognized by
a unique address. The ADRF6750 has two possible 7-bit slave
addresses for both read and write operations. The MSB of the
7-bit slave address is set to 1. Bit 5 of the slave address is set by
the CS pin (Pin 27). Bits[4:0] of the slave address are set to all
0s. The slave address consists of the seven MSBs of an 8-bit
word. The LSB of the word sets either a read or a write operation (see Figure 63). Logic 1 corresponds to a read operation,
whereas Logic 0 corresponds to a write operation.
The ADRF6750 acts as a standard slave device on the bus. The
data on the SDA pin (Pin 29) is eight bits long, supporting the
7-bit addresses plus the R/W bit. The ADRF6750 has 34 subaddresses to enable the user-accessible internal registers. Therefore,
it interprets the first byte as the device address and the second
byte as the starting subaddress. Autoincrement mode is supported,
which allows data to be read from or written to the starting subaddress and each subsequent address without manually addressing
the subsequent subaddress. A data transfer is always terminated
by a stop condition. The user can also access any unique subaddress
register on a one-by-one basis without updating all registers.
To control the device on the bus, the following protocol must
be followed. The master initiates a data transfer by establishing
a start condition, defined by a high-to-low transition on SDA
while SCL remains high. This indicates that an address/data
stream follows. All peripherals respond to the start condition
and shift the next eight bits (the 7-bit address and the R/W bit).
The bits are transferred from MSB to LSB. The peripheral that
recognizes the transmitted address responds by pulling the data
line low during the ninth clock pulse. This is known as an
acknowledge bit. All other devices then withdraw from the bus
and maintain an idle condition. During the idle condition, the
device monitors the SDA and SCL lines waiting for the start
condition and the correct transmitted address. The R/W bit
determines the direction of the data. Logic 0 on the LSB of the
Stop and start conditions can be detected at any stage of the data
transfer. If these conditions are asserted out of sequence with
normal read and write operations, they cause an immediate jump
to the idle condition. If an invalid subaddress is issued by the
user, the ADRF6750 does not issue an acknowledge and returns
to the idle condition. In a no acknowledge condition, the SDA
line is not pulled low on the ninth pulse. See Figure 64 and
Figure 65 for sample write and read data transfers, Figure 66 for
the timing protocol, and Figure 2 for a more detailed timing
diagram.
R/W
CTRL
SLAVE ADDRESS[6:0]
A5
SET BY
PIN 27
(CS)
0
0
0
0
0
X
0 = WR
1 = RD
08201-016
1
MSB = 1
Figure 63. Slave Address Configuration
SLAVE ADDR, LSB = 0 (WR) A(S) SUBADDR
A(S) DATA A(S)
DATA A(S)
P
08201-017
S
S = START BIT
A(S) = ACKNOWLEDGE BY SLAVE
P = STOP BIT
Figure 64. I2C Write Data Transfer
SLAVE ADDR, LSB = 0 (WR) A(S) SUBADDR
A(S) S SLAVE ADDR, LSB = 1 (RD) A(S) DATA
A(M)
DATA A(M) P
P = STOP BIT
A(M) = NO ACKNOWLEDGE BY MASTER
A(M) = ACKNOWLEDGE BY MASTER
08201-018
S
S = START BIT
A(S) = ACKNOWLEDGE BY SLAVE
Figure 65. I2C Read Data Transfer
START BIT
SDA
SLAVE ADDRESS
A6
SUBADDRESS
A5
A7
STOP BIT
DATA
A0
D7
D0
S
WR
SLAVE
ADDR[4:0]
ACK
ACK
SUBADDR[6:1]
Figure 66. I2C Data Transfer Timing
Rev. A | Page 22 of 40
ACK
DATA[6:1]
P
08201-002
SCL
ADRF6750
SPI INTERFACE
SPI Serial Interface Functionality
The ADRF6750 also supports the SPI protocol. The part powers
up in I2C mode but is not locked in this mode. To stay in I2C
mode, it is recommended that the user tie the CS line to either
3.3 V or GND, thus disabling SPI mode. It is not possible to lock
the I2C mode, but it is possible to select and lock the SPI mode.
The SPI serial interface of the ADRF6750 consists of the CS,
SDI (SDI/SDA), CLK (CLK/SCL), and SDO pins. CS is used to
select the device when more than one device is connected to the
serial clock and data lines. CLK is used to clock data in and out
of the part. The SDI pin is used to write to the registers. The
SDO pin is a dedicated output for the read mode. The part
operates in slave mode and requires an externally applied serial
clock to the CLK pin. The serial interface is designed to allow
the part to be interfaced to systems that provide a serial clock
that is synchronized to the serial data.
To select and lock the SPI mode, three pulses must be sent to the
CS pin, as shown in Figure 67. When the SPI protocol is locked
in, it cannot be unlocked while the device is still powered up. To
reset the serial interface, the part must be powered down and
powered up again.
Figure 68 shows an example of a write operation to the ADRF6750.
Data is clocked into the registers on the rising edge of CLK using
a 24-bit write command. The first eight bits represent the write
command 0xD4, the next eight bits are the register address, and
the final eight bits are the data to be written to the specific register.
Figure 69 shows an example of a read operation. In this example,
a shortened 16-bit write command is first used to select the
appropriate register for a read operation, the first eight bits
representing the write command 0xD4 and the final eight bits
representing the specific register. Then the CS line is pulsed low
for a second time to retrieve data from the selected register
using a 16-bit read command, the first eight bits representing
the read command 0xD5 and the final eight bits representing
the contents of the register being read. Figure 3 shows the
timing for both SPI read and SPI write operations.
Serial Interface Selection
The CS pin controls selection of the I2C or SPI interface.
Figure 67 shows the selection process that is required to lock
the SPI mode. To communicate with the part using the SPI
protocol, three pulses must be sent to the CS pin. On the third
rising edge, the part selects and locks the SPI protocol. Consistent
with most SPI standards, the CS pin must be held low during all
SPI communication to the part and held high at all other times.
A
B
C
CS
(STARTING
HIGH)
SPI LOCKED ON
THIRD RISING EDGE
A
B
C
SPI LOCKED ON
THIRD RISING EDGE
Figure 67. Selecting the SPI Protocol
Rev. A | Page 23 of 40
SPI FRAMING
EDGE
08201-019
CS
(STARTING
LOW)
SPI FRAMING
EDGE
ADRF6750
•••
CS
•••
CLK
D7
D6
D5
START
D4
D3
D2
D1
D0
D7
D6
D5
D4
WRITE
COMMAND [0xD4]
D3
D2
D1
D0
D2
D1
D0
•••
REGISTER
ADDRESS
CS
(CONTINUED)
•
•
•
CLK
(CONTINUED)
•
•
•
SDI
(CONTINUED)
•
•
•
D7
D6
D5
D4
D3
08201-020
SDI
STOP
DATA
BYTE
Figure 68. SPI Byte Write Example
•••
CS
•••
CLK
SDI
D7
D6
D5
START
D4
D3
D2
D1
D0
D7
D6
D5
D4
WRITE
COMMAND [0xD4]
D3
D2
D1
D0
•••
REGISTER
ADDRESS
CS
SDI
D7
D6
D5
D4
D3
D2
D1
D0
X
X
X
X
X
X
X
X
SDO
X
X
X
X
X
X
X
X
D7
D6
D5
D4
D3
D2
D1
D0
START
READ
COMMAND [0xD5]
DATA
BYTE
Figure 69. SPI Byte Read Example
Rev. A | Page 24 of 40
STOP
08201-021
CLK
ADRF6750
PROGRAM MODES
Charge Pump Current
The ADRF6750 has 34 8-bit registers to allow program control
of a number of functions. Either an SPI or an I2C interface
can be used to program the register set. For details about the
interfaces and timing, see Figure 63 to Figure 69. The registers
are documented in Table 6 to Table 24.
Register CR9, Bits[7:4], specify the charge pump current
setting. With an RSET value of 4.7 kΩ, the maximum charge
pump current is 5 mA. The following equation applies:
Several settings in the ADRF6750 are double-buffered. These
settings include the FRAC value, the INT value, the 5-bit
R-divider value, the reference frequency doubler, the R/2
divider, and the charge pump current setting. This means that
two events must occur before the part uses a new value for
any of the double-buffered settings. First, the new value is
latched into the device by writing to the appropriate register.
Next, a new write must be performed on Register CR0. When
Register CR0 is written, a new PLL acquisition takes place.
For example, updating the fractional value involves a write to
Register CR3, Register CR2, Register CR1, and Register CR0.
Register CR3 should be written to first, followed by Register CR2
and Register CR1 and, finally, Register CR0. The new acquisition
begins after the write to Register CR0. Double buffering ensures
that the bits written to do not take effect until after the write to
Register CR0.
12-Bit Integer Value
Register CR7 and Register CR6 program the integer value (INT)
of the feedback division factor. The INT value is a 12-bit number
whose MSBs are programmed through Register CR7, Bits[3:0].
The LSBs are programmed through Register CR6, Bits[7:0]. The
INT value is used in Equation 1 to set the LO frequency. Note
that these registers are double-buffered.
ICPmax = 23.5/RSET
The charge pump current has 16 settings from 312.5 μA to 5 mA.
For the loop filter that is specified in the application solution, a
charge pump current of 2.5 mA (Register CR9[7:4] = 7) gives a
loop bandwidth of 50 kHz, which is the recommended loop
bandwidth setting.
Transmit Disable Control (TXDIS)
The transmit disable control (TXDIS) is used to disable the RF output. TXDIS is normally held low. When asserted (brought high), it
disables the RF output. Register CR14 is used to control which
circuit blocks are powered down when TXDIS is asserted. To meet
both the off isolation power specifications and the turn-on/
turn-off settling time specifications, a value of 0x1B should be
loaded into Register CR14. This effectively ensures that the
attenuator is always enabled when TXDIS is asserted, even if other
circuitry is disabled.
Power-Down/Power-Up Control Bits
The three programmable power-up and power-down control
bits are as follows:
•
•
25-Bit Fractional Value
Register CR3 to Register CR0 program the fractional value
(FRAC) of the feedback division factor. The FRAC value is a
25-bit number whose MSB is programmed through Register CR3,
Bit 0. The LSB is programmed through Register CR0, Bit 0. The
FRAC value is used in Equation 1 to set the LO frequency. Note
that these registers are double-buffered.
Reference Input Path
The reference input path consists of a reference frequency doubler,
a 5-bit reference divider, and a divide-by-2 function (see Figure 53).
The doubler is programmed through Register CR10, Bit 5. The
5-bit divider is enabled by programming Register CR5, Bit 4,
and the division ratio is programmed through Register CR10,
Bits[4:0]. The R/2 divider is programmed through Register CR10,
Bit 6. Note that these registers are double-buffered.
When using a 10 MHz reference input frequency, enable the
doubler and disable the 5-bit divider and divide-by-2 to ensure
a PFD frequency of 20 MHz. As mentioned in the Reference
Input Path section, making the PFD frequency higher improves
the system noise performance.
•
Register CR12, Bit 2. Master power control bit for the PLL,
including the VCO. This bit is normally set to a default
value of 0 to power up the PLL.
Register CR27, Bit 2. Controls the LO monitor outputs,
LOMONP and LOMONN. The default is 0 when the monitor
outputs are powered down. Setting this bit to 1 powers up
the monitor outputs to one of −6 dBm, −12 dBm, −18 dBm,
or −24 dBm, as controlled by Register CR27, Bits[1:0].
Register CR29, Bit 0. Controls the quadrature modulator
power. The default is 0, which powers down the modulator.
Write a 1 to this bit to power up the modulator.
Lock Detect (LDET)
Lock detect is enabled by setting Register CR23, Bit 4, to 1.
Register CR23, Bit 3 sets the number of up/down pulses
generated by the PFD before lock detect is declared. The default
is 3072 pulses, which is selected when Bit 3 is set to 0. A more
aggressive setting of 2048 is selected when Bit 3 is set to 1. This
improves the lock detect time by 50 μs. Note, however, that it
does not affect the acquisition time to 100 Hz. Register CR23,
Bit 2 should be set to 0 for best operation. This bit sets up the
PFD up/down pulses to a coarse or low precision setting.
Rev. A | Page 25 of 40
ADRF6750
VCO Autocalibration
The VCO uses an autocalibration technique to select the correct
VCO and band, as explained in the Voltage-Controlled Oscillator
(VCO) section. Register CR24, Bit 0, controls whether the autocalibration is enabled. For normal operation, autocalibration needs
to be enabled. However, if using cumulative frequency steps of
100 kHz or less, autocalibration can be disabled by setting this
bit to 1 and then a new acquisition is initiated by writing to
Register CR0.
Attenuator
The attenuator can be programmed from 0 dB to 47 dB in steps
of 1 dB. Control is through Register CR30, Bits[5:0].
Revision Readback
The revision of the silicon die can be read back via Register CR33.
Rev. A | Page 26 of 40
ADRF6750
REGISTER MAP
REGISTER MAP SUMMARY
Table 6. Register Map Summary
Register Address (Hex)
0x00
0x01
0x02
0x03
0x04
0x05
0x06
0x07
0x08
0x09
0x0A
0x0B
0x0C
0x0D
0x0E
0x0F
0x10
0x11
0x12
0x13
0x14
0x15
0x16
0x17
0x18
0x19
0x1A
0x1B
0x1C
0x1D
0x1E
0x1F
0x20
0x21
Register Name
CR0
CR1
CR2
CR3
CR4
CR5
CR6
CR7
CR8
CR9
CR10
CR11
CR12
CR13
CR14
CR15
CR16
CR17
CR18
CR19
CR20
CR21
CR22
CR23
CR24
CR25
CR26
CR27
CR28
CR29
CR30
CR31
CR32
CR33
Type
Read/write
Read/write
Read/write
Read/write
Read/write
Read/write
Read/write
Read/write
Read/write
Read/write
Read/write
Read/write
Read/write
Read/write
Read/write
Read/write
Read/write
Read/write
Read/write
Read/write
Read/write
Read/write
Read/write
Read/write
Read/write
Read/write
Read/write
Read/write
Read/write
Read/write
Read/write
Read only
Read only
Read only
Rev. A | Page 27 of 40
Description
Fractional Word 4
Fractional Word 3
Fractional Word 2
Fractional Word 1
Reserved
5-bit reference divider enable
Integer Word 2
Integer Word 1 and muxout control
Reserved
Charge pump current setting
Reference frequency control
Reserved
PLL power-up
Reserved
TXDIS control
Reserved
Reserved
Reserved
Reserved
Reserved
Reserved
Reserved
Reserved
Lock detector control
Autocalibration
Reserved
Reserved
LO monitor output and External VCO control
Internal VCO power-down
Modulator
Attenuator
Reserved
Reserved
Revision code
ADRF6750
REGISTER BIT DESCRIPTIONS
Table 11. Register CR5 (Address 0x05), 5-Bit Reference
Divider Enable
Table 7. Register CR0 (Address 0x00), Fractional Word 4
Bit
7
6
5
4
3
2
1
0
1
1
Description
Fractional Word F7
Fractional Word F6
Fractional Word F5
Fractional Word F4
Fractional Word F3
Fractional Word F2
Fractional Word F1
Fractional Word F0 (LSB)
Bit
7
6
5
4
3
2
1
0
Double-buffered. Loaded on the write to Register CR0.
Table 8. Register CR1 (Address 0x01), Fractional Word 3
Bit
7
6
5
4
3
2
1
0
1
Description1
Fractional Word F15
Fractional Word F14
Fractional Word F13
Fractional Word F12
Fractional Word F11
Fractional Word F10
Fractional Word F9
Fractional Word F8
1
Bit
7
6
5
4
3
2
1
0
Table 9. Register CR2 (Address 0x02), Fractional Word 2
1
Description1
Fractional Word F23
Fractional Word F22
Fractional Word F21
Fractional Word F20
Fractional Word F19
Fractional Word F18
Fractional Word F17
Fractional Word F16
1
Double-buffered. Loaded on the write to Register CR0.
Bit
[7:4]
Double-buffered. Loaded on the write to Register CR0.
Bit
7
6
5
4
3
2
1
0
Description
Reserved
Reserved
Reserved
Reserved
Reserved
Reserved
Reserved
Fractional Word F24 (MSB)1
Description1
Integer Word N7
Integer Word N6
Integer Word N5
Integer Word N4
Integer Word N3
Integer Word N2
Integer Word N1
Integer Word N0
Table 13. Register CR7 (Address 0x07), Integer Word 1 and
Muxout Control
Table 10. Register CR3 (Address 0x03), Fractional Word 1
1
Double-buffered. Loaded on the write to Register CR0.
Table 12. Register CR6 (Address 0x06), Integer Word 2
Double-buffered. Loaded on the write to Register CR0.
Bit
7
6
5
4
3
2
1
0
Description
Reserved
Reserved
Reserved
5-bit R-divider enable1
0 = disable 5-bit R-divider (default)
1 = enable 5-bit R-divider
Reserved
Reserved
Reserved
Reserved
3
2
1
0
1
Description
Muxout control
0000 = tristate
0001 = logic high
0010 = logic low
1101 = RCLK/2
1110 = NCLK/2
Integer Word N111
Integer Word N101
Integer Word N91
Integer Word N81
Double-buffered. Loaded on the write to Register CR0.
Double-buffered. Loaded on the write to Register CR0.
Rev. A | Page 28 of 40
ADRF6750
Table 14. Register CR9 (Address 0x09), Charge Pump
Current Setting
Bit
[7:4]
3
2
1
0
1
Description
Charge pump current1
0000 = 0.31 mA (default)
0001 = 0.63 mA
0010 = 0.94 mA
0011 = 1.25 mA
0100 = 1.57 mA
0101 = 1.88 mA
0110 = 2.19 mA
0111 = 2.50 mA
1000 = 2.81 mA
1001 = 3.13 mA
1010 = 3.44 mA
1011 = 3.75 mA
1100 = 4.06 mA
1101 = 4.38 mA
1110 = 4.69 mA
1111 = 5.00 mA
Reserved
Reserved
Reserved
Reserved
Table 16. Register CR12 (Address 0x0C), PLL Power-Up
Bit
7
6
5
4
3
2
1
0
Table 17. Register CR14 (Address 0x0E), TXDIS Control
Bit
7
6
5
4
3
Double-buffered. Loaded on the write to Register CR0.
Table 15. Register CR10 (Address 0x0A), Reference
Frequency Control
2
1
Bit
7
6
0
5
[4:0]
1
Description
Reserved1
R/2 divider enable1
0 = bypass R/2 divider (default)
1 = enable R/2 divider
R-doubler enable1
0 = disable doubler (default)
1 = enable doubler
5-bit R-divider setting1
00000 = divide by 32 (default)
00001 = divide by 1
00010 = divide by 2
…
11110 = divide by 30
11111 = divide by 31
Description
Reserved
Reserved
Reserved
Reserved
Reserved
Power down PLL
0 = power up PLL (default)
1 = power down PLL
Reserved
Reserved
Description
Reserved
Reserved
TxDis_attenuator
0 = attenuator always enabled (default)
1 = disable attenuator when TXDIS = 1
TxDis_LOBuf
0 = LOBuf always enabled (default)
1 = disable LOBuf when TXDIS = 1
TxDis_QuadDiv
0 = QuadDiv always enabled (default)
1 = disable QuadDiv when TXDIS = 1
Reserved
TxDis_LOX2
0 = LOX2 always enabled (default)
1 = Disable LOX2 when TXDIS = 1
TxDis_RFMON
0 = RFMON always enabled (default)
1 = Disable RFMON when TXDIS = 1
Table 18. Register CR23 (Address 0x17), Lock Detector Control
Bit
7
6
5
4
3
Double-buffered. Loaded on the write to Register CR0.
2
1
0
Rev. A | Page 29 of 40
Description
Reserved
Reserved
Reserved
Lock detector enable
0 = lock detector disabled (default)
1 = lock detector enabled
Lock detector up/down count
0 = 3072 up/down pulses
1 = 2048 up/down pulses
Lock detector precision
0 = low, coarse (16 ns)
1 = high, fine (6 ns)
Reserved
Reserved
ADRF6750
Table 19. Register CR24 (Address 0x18), Autocalibration
Table 22. Register CR29 (Address 0x1D), Modulator
Bit
7
6
5
4
3
2
1
0
Bit
7
6
5
4
3
2
1
0
Description
Reserved
Reserved
Reserved
Reserved
Reserved
Reserved
Reserved
Disable autocalibration
0 = enable autocalibration (default)
1 = disable autocalibration
Table 20. Register CR27 (Address 0x1B), LO Monitor Output
and External VCO Control
Bit
7
6
5
4
3
2
[1:0]
Description
Reserved
Reserved
Reserved
Reserved
External VCO control
0 = internal VCO selected
1 = external VCO selected
Power up LO monitor output
0 = power down (default)
1 = power up
Monitor output power into 50 Ω
00 = −24 dBm (default)
01 = −18 dBm
10 = −12 dBm
11 = −6 dBm
Description
Reserved
Reserved
Reserved
Reserved
Reserved
Reserved
Reserved
Power up modulator
0 = power down (default)
1 = power up
Table 23. Register CR30 (Address 0x1E), Attenuator
Bit
7
6
[5:0]
Description
Reserved
Reserved
Attenuator A5 to Attenuator A0
000000 = 0 dB
000001 = 1 dB
000010 = 2 dB
…
011111 = 31 dB
110000 = 32 dB
110001 = 33 dB
…
111101 = 45 dB
111110 = 46 dB
111111 = 47 dB
Table 24. Register CR33 (Address 0x21), Revision Code1
Table 21. Register CR28 (Address 0x1C), Internal VCO
Power-Down
Bit
7
6
5
4
3
2
1
0
Description
Reserved
Reserved
Internal VCO power-down
0 = power up (default)
1 = power down
Reserved
Reserved
Reserved
Reserved
Reserved
Bit
7
6
5
4
3
2
1
0
1
Description
Revision code
Revision code
Revision code
Revision code
Revision code
Revision code
Revision code
Revision code
Read-only register.
Rev. A | Page 30 of 40
ADRF6750
SUGGESTED POWER-UP SEQUENCE
INITIAL REGISTER WRITE SEQUENCE
After applying power to the part, perform the initial register write
sequence that follows. Note that Register CR33, Register CR32,
and Register CR31 are read-only registers. Also note that all writable registers should be written to on power-up. Refer to the
Register Map section for more details on all registers.
1.
Write Register CR30: 0x00. Set attenuator to 0 dB gain.
2.
Write Register CR29: 0x00. Modulator is powered down.
The modulator is powered down by default to ensure that
no spurious signals can occur on the RF output when the
PLL is carrying out its first acquisition. The modulator
should be powered up only when the PLL is locked.
25. Write Register CR6: 0xXX. Set according to Equation 1 in
the Theory of Operation section.
26. Write Register CR5: 0x00. Disable the 5-bit reference divider.
27. Write Register CR4: 0x01. Reserved register.
28. Write Register CR3: 0x0X. Set according to Equation 1 in
the Theory of Operation section.
29. Write Register CR2: 0xXX. Set according to Equation 1 in
the Theory of Operation section.
30. Write Register CR1: 0xXX. Set according to Equation 1 in
the Theory of Operation section.
31. Write Register CR0: 0xXX. Set according to Equation 1 in
the Theory of Operation section. Register CR0 must be the
last register written for all the double-buffered bit writes to
take effect.
3.
Write Register CR28: 0x01. Power up the internal VCO.
Write 0x21 if using an external VCO.
4.
Write Register CR27: 0x00. Power down the LO monitor
and select the internal VCO. Write 0x08 to select an
external VCO.
5.
Write Register CR26: 0x00. Reserved register.
6.
Write Register CR25: 0x32. Reserved register.
33. Write Register CR29: 0x01. Power up modulator. The write
to Register CR29 does not need to be followed by a write to
Register CR0 because this register is not double-buffered.
7.
Write Register CR24: 0x18. Enable autocalibration.
Example—Changing the LO Frequency
8.
Write Register CR23: 0x70. Enable lock detector and
choose the recommended lock detect timing.
9.
Write Register CR22: 0x00. Reserved register.
Following is an example of how to change the LO frequency
after the initialization sequence. Using an example in which
the PLL is locked to 1200 MHz, the following conditions apply:
32. Monitor the LDET output or wait 170 μs to ensure that the
PLL is locked.
•
•
10. Write Register CR21: 0x00. Reserved register.
11. Write Register CR20: 0x00. Reserved register.
fPFD = 20 MHz (assumed)
Divide ratio N = 60, so INT = 60 decimal and FRAC = 0
The INT registers contain the following values:
Register CR7 = 0x00 and Register CR6 = 0x3C
12. Write Register CR19: 0x00. Reserved register.
13. Write Register CR18: 0x00. Reserved register.
The FRAC registers contain the following values:
Register CR3 = 0x00, Register CR2 = 0x00,
Register CR1 = 0x00, and Register CR0 = 0x00
14. Write Register CR17: 0x00. Reserved register.
15. Write Register CR16: 0x00. Reserved register.
16. Write Register CR15: 0x00. Reserved register.
To change the LO frequency to 1230 MHz, the divide ratio N
must be set to 61.5. Therefore, INT must be set to 61 decimal
and FRAC must be set to 16777216 by writing to the following
registers:
17. Write Register CR14: 0x1B. The attenuator is always
enabled, even when TXDIS is asserted.
18. Write Register CR13: 0x18. Reserved register.
19. Write Register CR12: 0x08. PLL powered up.
20. Write Register CR11: 0x00. Reserved register.
1.
Set the INT registers as follows:
Register CR7 = 0x00, Register CR6 = 0x3D
2.
Set the FRAC registers as follows:
Register CR3 = 0x01, Register CR2 = 0x00,
Register CR1 = 0x00, Register CR0 = 0x00
21. Write Register CR10: 0x21. The reference frequency doubler
is enabled, and the 5-bit divider and R/2 divider are bypassed.
22. Write Register CR9: 0x70. With the recommended loop
filter component values and RSET = 4.7 kΩ, as shown in
Figure 71, the charge pump current is set to 2.5 mA for
a loop bandwidth of 50 kHz.
Note that Register CR0 should be the last write in this sequence.
Writing to Register CR0 causes all double-buffered registers to
be updated, including the INT and FRAC registers, and starts a
new PLL acquisition.
23. Write Register CR8: 0x00. Reserved register.
If the cumulative frequency step is 100 kHz or less, the user can
turn off autocalibration. This process involves an additional
write of 0x19 to Register CR24, resulting in a smoother
frequency step and shorter acquisition time.
24. Write Register CR7: 0x0X. Set according to Equation 1 in
the Theory of Operation section. Also sets the MUXOUT
pin to tristate.
Rev. A | Page 31 of 40
ADRF6750
EVALUATION BOARD
GENERAL DESCRIPTION
Recommended Decoupling for Supplies
This board is designed to allow the user to evaluate the
performance of the ADRF6750. It contains the following:
The external 5 V supply is decoupled initially by a 10 μF capacitor
and then further by a parallel combination of 100 nF and 10 pF
capacitors that are placed as close to the DUT as possible for good
local decoupling. The regulator output should be decoupled by a
parallel combination of 10 pF and 220 μF capacitors. The 220 μF
capacitor decouples broadband noise, which leads to better phase
noise and is recommended for best performance. Case Size C
220 μF capacitors are used to minimize area. A parallel combination of 100 nF and 10 pF capacitors should be placed on each
VREGx pin. Again, these capacitors are placed as close to the pins
as possible. The impedance of all these capacitors should be low
and constant across a broad frequency range. Surface-mount
multilayered ceramic chip (MLCC) Class II capacitors provide
very low ESL and ESR, which assist in decoupling supply noise
effectively. They also provide good temperature stability and good
aging characteristics. Capacitance also changes vs. applied bias
voltage. Larger case sizes have less capacitance change vs. applied
bias voltage and also lower ESR but higher ESL. The 0603 size
capacitors provide a good compromise. X5R and X7R capacitors
are examples of these types of capacitors and are recommended
for decoupling.
•
•
I/Q modulator with integrated fractional-N PLL and VCO
SPI and I2C interface connectors
DC biasing and filter circuitry for the baseband inputs
Low-pass loop filter circuitry
10 MHz reference clock
Circuitry to support differential signaling to the TESTLO
inputs, including dc biasing circuitry
Circuitry to monitor the LOMON outputs
SMA connectors for power supplies and the RF output
The evaluation board comes with associated software to allow
easy programming of the ADRF6750.
HARDWARE DESCRIPTION
For more information, refer to the circuit diagram in Figure 71.
Power Supplies
An external 5 V supply (DUT +5 V) drives both an on-chip
3.3 V regulator and the quadrature modulator.
The regulator feeds the VREG1 through VREG6 pins on the
chip with 3.3 V. These pins power the PLL circuitry.
SPI and I2C Interface
The SPI interface connector is a 9-way, D-type connector that can
be connected to the printer port of a PC. Figure 70 shows the
PC cable diagram that must be used with the provided software.
The external reference clock generator can be driven by a 3 V
supply or by a 5 V supply. These supplies can be connected via
an SMA connector, VCO +V.
1
6
7
8
2
3
There is also an option to use the I2C interface by using the I2C
receptacle connector. This is a standard I2C connector. Pull-up
resistors are required on the signal lines. The CS pin can be used
to set the slave address of the ADRF6750. CS high sets the slave
address to 0x60, and CS low sets the slave address to 0x40.
CLK
1
2 14
DATA
3 15
LE
4 16
4
9
5 17
18
6
5
9-WAY
FEMALE
D-TYPE
GND
7
PC
19
20
8
21
9
10 22
23
11
24
12
13
25
25-WAY
MALE
D-TYPE
TO PC
PRINTER PORT
Figure 70. SPI PC Cable Diagram
Rev. A | Page 32 of 40
08201-022
•
•
•
•
•
•
ADRF6750
Baseband Inputs
The pair of I and Q baseband inputs are served by SMA inputs
so that they can be driven directly from an external generator,
which can also provide the dc bias required. An option is
provided to supply this dc bias through Connector J1, as well.
There is also an option to filter the baseband inputs, although
filtering may not be required, depending on the quality of the
baseband source.
These inputs also require a dc bias; the following two options
are provided:
•
•
Loop Filter
A fourth-order loop filter is provided at the output of the charge
pump and is required to adequately filter noise from the Σ-Δ
modulator used in the N-divider. With the charge pump current
set to a midscale value of 2.5 mA and using the on-chip VCO,
the loop bandwidth is approximately 60 kHz, and the phase
margin is 55°. C0G capacitors are recommended for use in the
loop filter because they have low dielectric absorption, which is
required for fast and accurate settling time. The use of non-C0G
capacitors may result in a long tail being introduced into the
settling time transient.
Reference Input
The reference input can be supplied by a 10 MHz Taitien clock
generator or by an external clock through the use of Connector J7.
The frequency range of the reference input is from 10 MHz to
20 MHz; if the lower frequency clock is used, the on-chip reference
frequency doubler should be used to set the PFD frequency to
20 MHz to optimize phase noise performance.
TESTLO Inputs
These pins are differential test inputs that allow a variety of
debug options. On this board, the capability is provided to drive
these pins with an external 2× LO signal that is then applied to
an Anaren balun to provide a differential input signal.
When driving the TESTLO pins, the PLL can be bypassed, and the
modulator can be driven directly by this external 2× LO signal.
A dc bias point of 3.3 V through a series inductor path.
A resistor in parallel is provided to de-Q any resonance.
A dc bias point, which can be varied from 0 V to 3.3 V
through a resistor divider network. Note that these resistors
should be large in value to ensure that the current drawn is
small and that the resistors have little effect on the input
resistance.
If these pins are not used, ground them by inserting 0 Ω resistors
in R47 and R54.
LOMON Outputs
These pins are differential LO monitor outputs that provide a
replica of the internal LO frequency at 1× LO. The single-ended
power in a 50 Ω load can be programmed to −24 dBm, −18 dBm,
−12 dBm, or −6 dBm. These open-collector outputs must be
terminated to 3.3 V. Because both outputs must be terminated
to 50 Ω, options are provided to terminate to 3.3 V using onboard 50 Ω resistors or by series inductors (or a ferrite bead),
in which case the 50 Ω termination is provided by the measuring
instrument. If not used, these outputs should be tied to REGOUT.
CCOMPx Pins
The CCOMPx pins are internal compensation nodes that must
be decoupled to ground with a 100 nF capacitor.
MUXOUT
MUXOUT is a test output that allows different internal nodes
to be monitored. It is a CMOS output stage that requires no
termination.
Lock Detect (LDET)
Lock detect is a CMOS output that indicates the state of the
PLL. A high level indicates a locked condition, and a low level
indicates a loss of lock condition.
TXDIS
This input disables the RF output. It can be driven from an external stimulus or simply connected high or low by Jumper J18.
RF Output (RFOUT)
RFOUT is the RF output of the ADRF6750. RFOUT MOD
should be grounded in the user application.
Rev. A | Page 33 of 40
ADRF6750
USER-DEFINED VALUE
08201-072
Figure 71. Applications Circuit Schematic
Rev. A | Page 34 of 40
ADRF6750
PCB ARTWORK
08201-073
Component Placement
08201-074
Figure 72. Evaluation Board, Top Side Component Placement
Figure 73. Evaluation Board, Bottom Side Component Placement
Rev. A | Page 35 of 40
ADRF6750
08201-075
PCB Layer Information
08201-076
Figure 74. Evaluation Board, Top Side—Layer 1
Figure 75. Evaluation Board, Bottom Side—Layer 4
Rev. A | Page 36 of 40
08201-077
ADRF6750
08201-078
Figure 76. Evaluation Board, Ground—Layer 2
Figure 77. Evaluation Board Power—Layer 3
Rev. A | Page 37 of 40
ADRF6750
BILL OF MATERIALS
Table 25. Bill of Materials
Qty
1
1
1
1
2
13
Description
ADRF6750 LFCSP, 56-lead 8 mm × 8 mm
VCO, 10 MHz
Connector, 9-pin, D-sub plug, SDEX9PNTD
Connector, I2C, SEMCONN receptacle
Capacitor, 10 μF, 25 V, tantalum, TAJ-C
Capacitor, 10 pF, 50 V, ceramic, C0G, 0402
Manufacturer
Analog Devices
Jauch
ITW McMurdo
Molex
AVX
Murata
Part Number
ADRF6750ACPZ
O 10.0-VX3Y-T1
FEC 150750
15830064
FEC 197518
FEC 8819564
Capacitor, 100 nF, 25 V, X7R, ceramic, 0603
AVX
FEC 317287
Capacitor, 220 μF, 6.3 V, tantalum, Case Size C
Capacitor spacing, 0402 (do not install)
Capacitor, 1 nF, 50 V, XR7, ceramic, 0603
Capacitor, 47 nF, 50 V, Xr7, ceramic, 1206
Capacitor, 680 pF, 50 V, NPO, ceramic, 0603
Capacitor, 1 nF, 50 V, C0G, ceramic, 0402
Capacitor, 100 pF, 50 V, C0G, ceramic, 0402
SMA end launch connector
AVX
FEC 197087
Murata
Murata
Murata
Murata
Murata
Johnson/Emerson
FEC 722170
FEC 1740542
FEC 430997
FEC 8819556
FEC 8819572
142-0701-851
3
Reference Designator
DUT
Y2
SPI
CONN
C1, C21
C2, C4, C6, C8, C10, C12, C14, C16,
C18, C19, C48, C53, C55
C3, C5, C7, C9, C11, C13, C15, C17,
C22, C47, C49 to C52, C54
C20
C30 to C33
C26
C24
C23, C25
C38, C39
C40, C44, C46, C57
J1 to J5, J7, J10 to J12, J14, J15,
TXDIS
J18, J20, J21
Jumper, 3-pin + shunt
Harwin
4
4
4
5
2
1
2
2
1
2
3
4
3
L1, L2
L3, L4
R2 to R5
R6 to R9, R36
R10, R11
R13
R14, R39
R12, R16
R15
R17, R18
R35, R44, R45
R48 to R51
R59 to R61
Inductor, 20 nH, 0402, LQW series
Inductor, 10 μH, 0805, LQM series
Resistor spacing, 0603 (user-defined values)
Resistor, 0 Ω, 1/16 W, 1%, 0402
Resistor, 0402, spacing (do not install)
Resistor, 4.7 kΩ, 1/10 W, 1%, 0603
Resistor, 1.2 kΩ, 1/10 W, 5%, 0603
Resistor, 270 Ω, 1/16 W, 1%, 0603
Resistor, 300 Ω, 1/16 W, 1%, 0603
Resistor, 0603, spacing (do not install)
Resistor, 51 Ω, 1/16 W, 5%, 0402
Resistor, 330 Ω, 1/10 W, 5%, 0805
Resistor, 100 Ω, 1/10 W, 5%, 0805
Murata
Murata
FEC 148533 and
FEC 150411
LQW15AN20N
LQM21FN1N100M
Vishay Draloric
FEC 1158241
Bourns
Yageo
Multicomp
Multicomp
CR0603-FX-472
FEC 9233393
FEC 9330917
FEC 93330968
Bourns
Bourns
Bourns
CR0402-JW-510
CR0805-JW-331
CR0805-JW-101
15
1
4
1
1
2
4
4
12
Rev. A | Page 38 of 40
ADRF6750
OUTLINE DIMENSIONS
8.00
BSC SQ
0.30
0.23
0.18
0.60 MAX
0.60 MAX
56
43
42
1
PIN 1
INDICATOR
PIN 1
INDICATOR
TOP
VIEW
4.95
4.80 SQ
4.65
EXPOSED
PAD
(BOTTOM VIEW)
7.75
BSC SQ
0.50
0.40
0.30
14
29
28
15
0.30 MIN
0.80 MAX
0.65 TYP
12° MAX
SEATING
PLANE
0.50 BSC
0.05 MAX
0.02 NOM
COPLANARITY
0.08
0.20 REF
FOR PROPER CONNECTION OF
THE EXPOSED PAD, REFER TO
THE PIN CONFIGURATION AND
FUNCTION DESCRIPTIONS
SECTION OF THIS DATA SHEET.
COMPLIANT TO JEDEC STANDARDS MO-220-VLLD-2
041807-B
1.00
0.85
0.80
6.50
REF
Figure 78. 56-Lead Lead Frame Chip Scale Package [LFCSP_VQ]
8 mm × 8 mm Body, Very Thin Quad
(CP-56-3)
Dimensions shown in millimeters
ORDERING GUIDE
Model 1
ADRF6750ACPZ-R7
ADRF6750-EVALZ
1
Temperature Range
−40°C to +85°C
Package Description
56-Lead Lead Frame Chip Scale Package [LFCSP_VQ], 7" Tape and Reel
Evaluation Board
Z = RoHS Compliant Part.
Rev. A | Page 39 of 40
Package Option
CP-56-3
ADRF6750
NOTES
I2C refers to a communications protocol originally developed by Philips Semiconductors (now NXP Semiconductors).
©2010 Analog Devices, Inc. All rights reserved. Trademarks and
registered trademarks are the property of their respective owners.
D08201-0-4/10(0)
Rev. A | Page 40 of 40
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