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ADS8881
SBAS547D – MAY 2013 – REVISED AUGUST 2015
ADS8881x 18-Bit, 1-MSPS, Serial Interface, microPower, Miniature,
True-Differential Input, SAR Analog-to-Digital Converter
1 Features
2 Applications
•
•
•
•
•
•
•
1
•
•
•
•
•
•
•
•
Sample Rate: 1 MHz
No Latency Output
Unipolar, True-Differential Input Range:
–VREF to +VREF
Wide Common-Mode Voltage Range:
0 V to VREF with 90-dB CMRR (min)
SPI™-Compatible Serial Interface with
Daisy-Chain Option
Excellent AC and DC Performance:
– ADS8881C:
INL: ±1 LSB (typ), ±2.0 LSB (max)
DNL: ±1.0 LSB (max), 18-Bit NMC
SNR: 100 dB, THD: –115 dB
– ADS8881I:
INL: ±1.5 LSB (typ), ±3.0 LSB (max)
DNL: +1.5 and –1 LSB (max), 18-Bit NMC
SNR: 100 dB, THD: –115 dB
Wide Operating Range:
– AVDD: 2.7 V to 3.6 V
– DVDD: 2.7 V to 3.6 V
(Independent of AVDD)
– REF: 2.5 V to 5 V (Independent of AVDD)
– Operating Temperature:
ADS8881C : 0°C to +70°C
ADS8881I : –40°C to +85°C
Low-Power Dissipation:
– 5.5 mW at 1 MSPS
– 0.55 mW at 100 kSPS
– 55 µW at 10 kSPS
Power-Down Current (AVDD): 50 nA
Full-Scale Step Settling to 18 Bits: 290 ns
Packages: MSOP-10 and VSON-10
Automatic Test Equipment (ATE)
Instrumentation and Process Controls
Precision Medical Equipment
Low-Power, Battery-Operated Instruments
3 Description
The ADS8881 is an 18-bit, 1-MSPS, true-differential
input, analog-to-digital converter (ADC). The device
operates with a 2.5-V to 5-V external reference,
offering a wide selection of signal ranges without
additional input signal scaling. The reference voltage
setting is independent of, and can exceed, the analog
supply voltage (AVDD).
The device
that also
cascading
indicator bit
easy.
offers an SPI-compatible serial interface
supports daisy-chain operation for
multiple devices. An optional busymakes synchronizing with the digital host
The device supports unipolar, true-differential analog
input signals with a differential input swing of –VREF to
VREF. This true-differential analog input structure
allows for a common-mode voltage of any value in
the range of 0 V to VREF (when both inputs are within
the operating input range of –0.1 V to VREF
+ 0.1 V).
Device operation is optimized for very low-power
operation. Power consumption directly scales with
speed. This feature makes the ADS8881 excellent for
lower-speed applications.
Device Information(1)
PART NUMBER
PACKAGE
ADS8881x
BODY SIZE (NOM)
VSSOP (10)
3.00 mm × 3.00 mm
VSON (10)
3.00 mm × 3.00 mm
(1) For all available packages, see the orderable addendum at
the end of the datasheet.
True-Differential Input Range
Traditional Input Range
ADS8881 Input Range
VDIFF
+VREF
2.5 V to 5 V
2.7 V to 3.6 V
REF
AVDD
2.7 V to 3.6 V
VREF
VCM
0V
0 V - VREF
DVDD
AINP
VCM
DIN
ADS8881
VREF
SCLK
DOUT
0V
AINM
CONVST
0 V - VREF
VREF/2
GND
-VREF
1
An IMPORTANT NOTICE at the end of this data sheet addresses availability, warranty, changes, use in safety-critical applications,
intellectual property matters and other important disclaimers. PRODUCTION DATA.
ADS8881
SBAS547D – MAY 2013 – REVISED AUGUST 2015
www.ti.com
Table of Contents
1
2
3
4
5
6
7
8
Features ..................................................................
Applications ...........................................................
Description .............................................................
Revision History.....................................................
Companion Products.............................................
Device Comparison ...............................................
Pin Configurations and Functions .......................
Specifications.........................................................
8.1
8.2
8.3
8.4
8.5
8.6
8.7
8.8
8.9
9
1
1
1
2
4
5
6
7
Absolute Maximum Ratings ...................................... 7
ESD Ratings.............................................................. 7
Recommended Operating Conditions....................... 7
Thermal Information .................................................. 7
Electrical Characteristics........................................... 8
Timing Requirements: 3-Wire Operation ................ 10
Timing Requirements: 4-Wire Operation ................ 11
Timing Requirements: Daisy-Chain ........................ 12
Typical Characteristics ............................................ 13
Parametric Measurement Information ............... 20
9.1 Equivalent Circuits .................................................. 20
10 Detailed Description ........................................... 21
10.1 Overview ............................................................... 21
10.2 Functional Block Diagram ..................................... 21
10.3 Feature Description............................................... 21
10.4 Device Functional Modes...................................... 24
11 Application and Implementation........................ 33
11.1 Application Information.......................................... 33
11.2 Typical Applications .............................................. 36
12 Power-Supply Recommendations ..................... 45
12.1 Power-Supply Decoupling..................................... 45
12.2 Power Saving ........................................................ 45
13 Layout................................................................... 47
13.1 Layout Guidelines ................................................. 47
13.2 Layout Example .................................................... 47
14 Device and Documentation Support ................. 48
14.1
14.2
14.3
14.4
14.5
14.6
14.7
Device Support......................................................
Documentation Support ........................................
Receiving Notification of Documentation Updates
Community Resources..........................................
Trademarks ...........................................................
Electrostatic Discharge Caution ............................
Glossary ................................................................
48
48
48
48
49
49
49
15 Mechanical, Packaging, and Orderable
Information ........................................................... 49
4 Revision History
NOTE: Page numbers for previous revisions may differ from page numbers in the current version.
Changes from Revision C (July 2014) to Revision D
Page
•
Added Companion Products and Device Comparison sections............................................................................................. 4
•
Changed ESD Ratings table to current standards, added HBM and CDM data ................................................................... 7
•
Added timing specifications for different operating temperature ranges for the tconv, td-CK-DO, and tquiet parameters in
the Timing Requirements: 3-Wire Operation table .............................................................................................................. 10
•
Added timing specifications for different operating temperature ranges for the tconv parameter in Timing
Requirements: 4-Wire Operation table ................................................................................................................................. 11
•
Added timing specifications for different operating temperature ranges for the tconv parameter in Timing
Requirements: Daisy-Chain table ........................................................................................................................................ 12
2
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SBAS547D – MAY 2013 – REVISED AUGUST 2015
Changes from Revision B (December 2014) to Revision C
Page
•
Changed format to meet latest data sheet standards; added new sections, moved existing sections ................................. 1
•
Changed ADS8881 to ADS8881C, added ADS8881I ........................................................................................................... 1
•
Separated ADS8881C and ADS8881I specifications in Excellent AC and DC Performance Features bullet ....................... 1
•
Changed Device Information table to current standards ........................................................................................................ 1
•
Added Recommended Operating Conditions table ................................................................................................................ 7
•
Changed LSB footnote to include how to convert LSB to ppm ............................................................................................. 8
•
Changed fSCLK parameter maximum specification from 66.6 MHz to 70 MHz in Timing Requirements: 3-Wire
Operation table. .................................................................................................................................................................... 10
•
Changed tSCLK parameter minimum specification from 15 ns to 14.3 ns in Timing Requirements: 3-Wire Operation
table. ..................................................................................................................................................................................... 10
•
Added more information about validity of data on SCLK edges in all interface modes ....................................................... 25
•
Changed diagrams and text for better explanation of the daisy-chain feature in the Daisy-Chain Mode section ............... 30
•
Changed Equation 2 and Equation 3 .................................................................................................................................. 34
•
Added Layout Guidelines section ......................................................................................................................................... 47
Changes from Revision A (July 2013) to Revision B
Page
•
Changed Wide Common-Mode Voltage Range Features bullet ............................................................................................ 1
•
Added note 2 to Family Information table............................................................................................................................... 5
•
Changed External Reference Input, Reference input current parameter typical specification from 350 to 300 .................... 8
•
Added External Reference Input, Reference leakage current parameter to Electrical Characteristics.................................. 8
•
Changed Power-Supply Requirements, Power-supply voltage parameter digital interface supply range as a function
of SCLK in Electrical Characteristics ...................................................................................................................................... 9
•
Added Digital Inputs, Digital input leakage current parameter to Electrical Characteristics .................................................. 9
•
Added true-differential input feature details to Analog Input section.................................................................................... 22
•
Deleted shading from Figure 64 ........................................................................................................................................... 35
•
Deleted shading from Figure 65 ........................................................................................................................................... 36
•
Deleted shading from Figure 67 ........................................................................................................................................... 38
•
Deleted shading from Figure 69 ........................................................................................................................................... 40
•
Deleted shading from Figure 70 ........................................................................................................................................... 40
•
Deleted shading from Figure 72 ........................................................................................................................................... 43
•
Added power scaling with throughput feature details to Power Saving section................................................................... 45
Changes from Original (May 2013) to Revision A
•
Page
Changed document status to Production Data; pre-RTM changes made throughout document .......................................... 1
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ADS8881
SBAS547D – MAY 2013 – REVISED AUGUST 2015
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5 Companion Products
PART NUMBER
4
NAME
REF5045
Low Noise, Very Low Drift, Precision Voltage Reference
THS4281
Very Low-Power High Speed Rail-To-Rail Input/Output Voltage Feedback Operational Amplifier
REF2925
2.5V 100ppm/Degrees C, 50uA in SOT23-3 Series (Bandgap) Voltage Reference
AFE5808A
Fully Integrated, 8 Channel Ultrasound Analog Front End with Passive CW Mixer
LMK04803
Low Noise Clock Jitter Cleaner With Dual Cascaded PLLs and Integrated 1.9 GHz VCO
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SBAS547D – MAY 2013 – REVISED AUGUST 2015
6 Device Comparison
THROUGHPUT
18-BIT, TRUE-DIFFERENTIAL
16-BIT, SINGLE-ENDED
16-BIT, TRUE-DIFFERENTIAL
100 kSPS
ADS8887
ADS8866
ADS8867
250 kSPS
—
ADS8339
—
400 kSPS
ADS8885
ADS8864
ADS8865
500 kSPS
—
ADS8319
ADS8318
680 kSPS
ADS8883
ADS8862
ADS8863
1 MSPS
ADS8881
ADS8860
ADS8861
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ADS8881
SBAS547D – MAY 2013 – REVISED AUGUST 2015
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7 Pin Configurations and Functions
DGS Package
VSSOP-10
Top View, Not to Scale
DRC Package
VSON-10
Top View, Not to Scale
REF
1
10
DVDD
AVDD
2
9
DIN
SCLK
AINP
3
8
SCLK
7
DOUT
AINN
4
7
DOUT
6
CONVST
GND
5
6
CONVST
REF
1
10
DVDD
AVDD
2
9
DIN
AINP
3
8
AINN
4
GND
5
Thermal
PAD
Pin Functions
PIN
NAME
NO.
I/O
AINN
4
Analog input
Inverting analog signal input
AINP
3
Analog input
Noninverting analog signal input
AVDD
2
Analog
CONVST
6
Digital input
Convert input. This pin also functions as the CS input in 3-wire interface mode; see the
Description and Timing Requirements sections for more details.
DIN
9
Digital input
Serial data input. The DIN level at the start of a conversion selects the mode of operation
(such as CS or daisy-chain mode). This pin also serves as the CS input in 4-wire interface
mode; see the Description and Timing Requirements sections for more details.
DOUT
7
Digital output
Serial data output
DVDD
10
Power supply
Digital interface power supply. This pin must be decoupled to GND with a 1-μF capacitor.
GND
5
Analog, digital
Device ground. Note that this pin is a common ground pin for both the analog power supply
(AVDD) and digital I/O supply (DVDD). The reference return line is also internally connected to
this pin.
REF
1
Analog
SCLK
8
Digital input
Clock input for serial interface. Data output (on DOUT) are synchronized with this clock.
Thermal
pad
—
Thermal pad
Exposed thermal pad (only for the DRC package option). Texas Instruments recommends
connecting the thermal pad to the printed circuit board (PCB) ground.
6
DESCRIPTION
Analog power supply. This pin must be decoupled to GND with a 1-μF capacitor.
Positive reference input. This pin must be decoupled with a 10-μF or larger capacitor.
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SBAS547D – MAY 2013 – REVISED AUGUST 2015
8 Specifications
8.1 Absolute Maximum Ratings
over operating free-air temperature range (unless otherwise noted) (1)
MIN
MAX
UNIT
AINP to GND or AINN to GND
–0.3
REF + 0.3
V
AVDD to GND or DVDD to GND
–0.3
4
V
REF to GND
–0.3
5.7
V
Digital input voltage to GND
–0.3
DVDD + 0.3
V
V
Digital output to GND
Operating temperature, TA
–0.3
DVDD + 0.3
ADS8881C
0
70
ADS8881I
–40
85
–65
150
Storage temperature, Tstg
(1)
°C
°C
Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. These are stress ratings
only, which do not imply functional operation of the device at these or any other conditions beyond those indicated under Recommended
Operating Conditions. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
8.2 ESD Ratings
VALUE
Human-body model (HBM), per ANSI/ESDA/JEDEC JS-001
V(ESD)
(1)
(2)
Electrostatic discharge
(1)
UNIT
±2000
Charged-device model (CDM), per JEDEC specification JESD22C101 (2)
V
±500
JEDEC document JEP155 states that 500-V HBM allows safe manufacturing with a standard ESD control process.
JEDEC document JEP157 states that 250-V CDM allows safe manufacturing with a standard ESD control process.
8.3 Recommended Operating Conditions
over operating free-air temperature range (unless otherwise noted)
MIN
NOM
MAX
UNIT
AVDD
Analog power supply
3
V
DVDD
Digital power supply
3
V
VREF
Reference voltage
5
V
8.4 Thermal Information
ADS8881
THERMAL METRIC
DGS (VSSOP)
DRC (VSON)
UNIT
10 PINS
10 PINS
RθJA
Junction-to-ambient thermal resistance
151.9
111.1
°C/W
RθJC(top)
Junction-to-case (top) thermal resistance
45.4
46.4
°C/W
RθJB
Junction-to-board thermal resistance
72.2
45.9
°C/W
ψJT
Junction-to-top characterization parameter
3.3
3.5
°C/W
ψJB
Junction-to-board characterization parameter
70.9
45.5
°C/W
RθJC(bot)
Junction-to-case (bottom) thermal resistance
N/A
N/A
°C/W
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SBAS547D – MAY 2013 – REVISED AUGUST 2015
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8.5 Electrical Characteristics
All minimum and maximum specifications are at AVDD = 3 V, DVDD = 3 V, VREF = 5 V, VCM = VREF / 2 V,
and fSAMPLE = 1 MSPS, over the operating free-air temperature range, unless otherwise noted.
Typical specifications are at TA = 25°C, AVDD = 3 V, and DVDD = 3 V.
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
–VREF
VREF
V
AINP
–0.1
VREF + 0.1
AINN
–0.1
VREF + 0.1
ANALOG INPUT
Full-scale input span (1) (2)
Operating input range (1) (2)
VCM
Input common-mode range
CI
Input capacitance
AINP – AINN
0
VREF / 2
AINP and AINN terminal to GND
VREF
59
V
V
pF
EXTERNAL REFERENCE INPUT
VREF
Input range
Reference input current
ADS8881C
ADS8881I
3
5
2.5
5
During conversion, 1-MHz sample rate, midcode
Reference leakage current
CREF
Decoupling capacitor at the
REF input
Input leakage current
V
300
μA
250
nA
22
µF
5
nA
18
Bits
10
During acquisition for dc input
SYSTEM PERFORMANCE
Resolution
NMC
No missing codes
DNL
Differential linearity
INL
Integral linearity (4)
EO
Offset error (5)
18
–0.99
±0.6
1
ADS8881I
–0.99
±0.7
1.5
ADS8881C
–2
±1.2
2
ADS8881I
–3
±1.5
3
–4
±1
4
Offset error drift with
temperature
EG
±1.5
Gain error
–0.01
Gain error drift with
temperature
CMRR
Common-mode rejection
ratio
PSRR
Power-supply rejection ratio
Bits
ADS8881C
±0.005
0.01
mV
%FSR
ppm/°C
100
At mid-code
Transition noise
LSB (3)
µV/°C
±0.15
90
LSB (3)
dB
80
dB
0.7
LSB
SAMPLING DYNAMICS
tconv
Conversion time
500
tACQ
Acquisition time
290
710
ns
Maximum throughput rate
with or without latency
(1)
(2)
(3)
(4)
(5)
8
ns
1000
kHz
Aperture delay
4
ns
Aperture jitter, RMS
5
ps
Step response
Settling to 18-bit accuracy
290
ns
Overvoltage recovery
Settling to 18-bit accuracy
290
ns
Ideal input span, does not include gain or offset error.
Specified for VCM = VREF / 2; see the Analog Input section for the effect of VCM on the full-scale input range.
LSB = least significant bit. 1 LSB at 18-bits is approximately 3.8 ppm.
This parameter is the endpoint INL, not best-fit.
Measured relative to actual measured reference.
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SBAS547D – MAY 2013 – REVISED AUGUST 2015
Electrical Characteristics (continued)
All minimum and maximum specifications are at AVDD = 3 V, DVDD = 3 V, VREF = 5 V, VCM = VREF / 2 V,
and fSAMPLE = 1 MSPS, over the operating free-air temperature range, unless otherwise noted.
Typical specifications are at TA = 25°C, AVDD = 3 V, and DVDD = 3 V.
PARAMETER
TEST CONDITIONS
MIN
TYP
98
99.9
MAX
UNIT
DYNAMIC CHARACTERISTICS
At 1 kHz, VREF = 5 V
SINAD
Signal-to-noise + distortion
(6)
At 10 kHz, VREF = 5 V
98.7
At 100 kHz, VREF = 5 V
93.3
At 1 kHz, VREF = 5 V
SNR
THD
Signal-to-noise ratio
(6)
Total harmonic distortion
(6) (7)
SFDR
Spurious-free dynamic
range (6)
BW–3dB
–3-dB small-signal bandwidth
98.5
dB
100
At 10 kHz, VREF = 5 V
99.5
At 100 kHz, VREF = 5 V
93.5
At 1 kHz, VREF = 5 V
–115
At 10 kHz, VREF = 5 V
–112
At 100 kHz, VREF = 5 V
–102
At 1 kHz, VREF = 5 V
115
At 10 kHz, VREF = 5 V
112
At 100 kHz, VREF = 5 V
102
dB
dB
dB
30
MHz
POWER-SUPPLY REQUIREMENTS
AVDD
Power-supply
voltage
Supply current
PVA
DVDD
AVDD
Power dissipation
Analog supply
2.7
3
3.6
Digital supply range for SCLK > 40 MHz
2.7
3
3.6
Digital supply range for SCLK < 40 MHz
1.65
1.8
3.6
1-MHz sample rate, AVDD = 3 V
1.8
2.4
1-MHz sample rate, AVDD = 3 V
5.5
7.2
100-kHz sample rate, AVDD = 3 V
0.55
10-kHz sample rate, AVDD = 3 V
IAPD
Device power-down
current (8)
V
mA
mW
55
μW
50
nA
DIGITAL INPUTS: LOGIC FAMILY (CMOS)
VIH
High-level input voltage
VIL
Low-level input voltage
ILK
Digital input leakage current
1.65 V < DVDD < 2.3 V
0.8 × DVDD
DVDD + 0.3
2.3 V < DVDD < 3.6 V
0.7 × DVDD
DVDD + 0.3
1.65 V < DVDD < 2.3 V
–0.3
0.2 × DVDD
2.3 V < DVDD < 3.6 V
–0.3
0.3 × DVDD
±10
V
V
±100
nA
0.8 × DVDD
DVDD
V
V
DIGITAL OUTPUTS: LOGIC FAMILY (CMOS)
VOH
High-level output voltage
IO = 500-μA source, CLOAD = 20 pF
VOL
Low-level output voltage
IO = 500-μA sink, CLOAD = 20 pF
0
0.2 × DVDD
ADS8881C
0
70
–40
85
TEMPERATURE RANGE
TA
(6)
(7)
(8)
Operating free-air
temperature
ADS8881I
°C
All specifications expressed in decibels (dB) refer to the full-scale input (FSR) and are tested with an input signal 0.5 dB below full-scale,
unless otherwise specified.
Calculated on the first nine harmonics of the input frequency.
The device automatically enters a power-down state at the end of every conversion, and remains in power-down during the acquisition
phase.
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8.6 Timing Requirements: 3-Wire Operation
All specifications are at AVDD = 3 V, DVDD = 3 V, and over the operating free-air temperature range, unless otherwise noted.
MIN
tACQ
Acquisition time
TYP MAX UNIT
290
ns
TA in the range –40°C to 85°C
500
710
TA in the range 0°C to 70°C
500
700
tconv
Conversion time
tconv
Conversion time
1/fsample
Time between conversions
twh-CNV
Pulse duration: CONVST high
fSCLK
SCLK frequency
tSCLK
SCLK period
14.3
tclkl
SCLK low time
0.45
0.55 tSCLK
tclkh
SCLK high time
0.45
0.55 tSCLK
th-CK-DO
SCLK falling edge to current data invalid
td-CK-DO
ns
500
ns
1000
ns
10
ns
70 MHz
ns
3
SCLK falling edge to next data valid delay
ns
TA in the range –40°C to 85°C
13.4
TA in the range 0°C to 70°C
11.7
TA in the range 25°C to 50°C
10.7
ns
td-CNV-DO
Enable time: CONVST low to MSB valid
12.3
ns
td-CNV-DOhz
Disable time: CONVST high or last SCLK falling edge to DOUT 3-state (CS mode)
13.2
ns
tquiet
Quiet time
TA in the range –40°C to 85°C
20
TA in the range 0°C to 70°C
13
ns
1/fsample
DIN = HIGH
tconv-max
tACQ
tclkh
th-CK-DO
CONVST
tclkl
tquiet
œœ
SCLK
1
2
td-CNV-DO
DOUT
3
16
17
18
D1
D0
tSCLK
œœ
D17
D16
D15
D2
œœ
twh-CNV-min
td-CK-DO
td-CK-DOhz
Figure 1. 3-Wire Operation: CONVST Functions as Chip Select
NOTE: Figure 1 shows the timing diagram for the 3-Wire CS Mode Without a Busy Indicator interface option.
However, the timing parameters specified in Timing Requirements: 3-Wire Operation are also applicable for the
3-Wire CS Mode With a Busy Indicator interface option, unless otherwise specified; see the Device Functional
Modes section for specific details for each interface option.
10
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SBAS547D – MAY 2013 – REVISED AUGUST 2015
8.7 Timing Requirements: 4-Wire Operation
All specifications are at AVDD = 3 V, DVDD = 3 V, and over the operating free-air temperature range, unless otherwise noted.
MIN
tACQ
Acquisition time
TYP MAX UNIT
290
ns
TA in the range –40°C to 85°C
500
710
TA in the range 0°C to 70°C
500
700
tconv
Conversion time
ns
tconv
Conversion time
1/fsample
Time between conversions
twh-DI
Pulse duration: DIN high
twl-CNV
Pulse width: CONVST low
td-DI-DO
Delay time: DIN low to MSB valid
12.3
ns
td-DI-DOhz
Delay time: DIN high or last SCLK falling edge to DOUT 3-state
13.2
ns
tsu-DI-CNV
Setup time: DIN high to CONVST rising edge
th-DI-CNV
Hold time: DIN high from CONVST rising edge (see Figure 63)
500
ns
1000
ns
10
ns
20
ns
7.5
ns
0
ns
1/fsample
tACQ
tconv-max
CONVST
tsu-DI-CNV
twl-CNV
DIN
œœ
SCLK
1
DOUT
D17
2
3
D16
D15
16
17
18
D1
D0
œœ
twh-DI-min
td-DI-DO
D2
œœ
td-DI-DOhz
Figure 2. 4-Wire Operation: DIN Functions as Chip Select
NOTE: Figure 2 shows the timing diagram for the 4-Wire CS Mode Without a Busy Indicator interface option.
However, the timing parameters specified in Timing Requirements: 4-Wire Operation are also applicable for the
4-Wire CS Mode With a Busy Indicator interface option, unless otherwise specified; see the Device Functional
Modes section for specific details for each interface option.
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8.8 Timing Requirements: Daisy-Chain
All specifications are at AVDD = 3 V, DVDD = 3 V, and over the operating free-air temperature range, unless otherwise noted.
MIN
tACQ
Acquisition time
TYP MAX UNIT
290
ns
TA in the range -40°C to 85°C
500
710
TA in the range 0°C to 70°C
500
700
tconv
Conversion time
tconv
Conversion time
1/fsample
Time between conversions
tsu-CK-CNV
Setup time: SCLK valid to CONVST rising edge
th-CK-CNV
Hold time: SCLK valid from CONVST rising edge
tsu-DI-CNV
Setup time: DIN low to CONVST rising edge (see Figure 2)
th-DI-CNV
Hold time: DIN low from CONVST rising edge (see Figure 63)
tsu-DI-CK
Setup time: DIN valid to SCLK falling edge
ns
500
ns
1000
ns
5
ns
5
ns
7.5
ns
0
ns
1.5
ns
1/fsample
tconv-
tACQ
max
CONVST
th-CK-CNV
SCLK
DIN 1 = LOW
1
2
17
18
19
20
35
36
D17
D16
D1
D0
tsu-DI-CK
tsu-CK-CNV
DOUT 1,
DIN 2
D17
D16
D1
D0
DOUT 2
D17
D16
D1
D0
Device 2 Data
Device 1 Data
Figure 3. Daisy-Chain Operation: Two Devices
NOTE: Figure 3 shows the timing diagram for the Daisy-Chain Mode Without a Busy Indicator interface option.
However, the timing parameters specified in Timing Requirements: Daisy-Chain are also applicable for the DaisyChain Mode With a Busy Indicator interface option, unless otherwise specified; see the Device Functional Modes
section for specific details for each interface option.
12
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8.9 Typical Characteristics
At TA = 25°C, AVDD = 3 V, DVDD = 3 V, VREF = 5 V, and fSAMPLE = 1 MSPS, unless otherwise noted.
1
Typical Differential Nonlinearity (LSB)
Typical Integral Nonlinearity (LSB)
3
AVDD = 3 V
REF = 2.5 V
TA = 25ƒC
2
1
0
±1
±2
±3
±131072
0.5
0.25
0
-0.25
-0.5
-0.75
-1
±131072
131071
ADC Output Code
C002
Figure 5. Typical DNL (VREF = 2.5 V)
1
3
AVDD = 3 V
REF = 5 V
TA = 25ƒC
2
Typical Differential Nonlinearity (LSB)
Typical Integral Nonlinearity (LSB)
131071
ADC Output Code
C001
Figure 4. Typical INL (VREF = 2.5 V)
1
0
±1
±2
±3
±131072
131071
ADC Output Code
AVDD = 3 V
REF = 5 V
TA = 25ƒC
0.75
0.5
0.25
0
-0.25
-0.5
-0.75
-1
±131072
131071
ADC Output Code
C003
Figure 6. Typical INL (VREF = 5 V)
C004
Figure 7. Typical DNL (VREF = 5 V)
3
2
AVDD = 3 V
REF = 5 V
2
Differential Nonlinearity (LSB)
Integral Nonlinearity (LSB)
AVDD = 3 V
REF = 2.5 V
TA = 25ƒC
0.75
1
0
-1
-2
-3
AVDD = 3 V
REF = 5 V
1.5
1
0.5
0
-0.5
-1
-40
-15
10
35
Free-Air Temperature (oC)
60
85
-40
C00
Figure 8. INL vs Temperature
-15
10
35
60
Free-Air Temperature (oC)
85
C00
Figure 9. DNL vs Temperature
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Typical Characteristics (continued)
At TA = 25°C, AVDD = 3 V, DVDD = 3 V, VREF = 5 V, and fSAMPLE = 1 MSPS, unless otherwise noted.
2
AVDD = 3 V
TA = 25oC
2
Differential Nonlinearity (LSB)
Integral Nonlinearity (LSB)
3
1
0
-1
-2
AVDD = 3 V
TA = 25oC
1.5
1
0.5
0
-0.5
-1
-3
2.5
3
3.5
4
Reference Voltage (V)
4.5
2.5
5
3
C00
Figure 10. INL vs Reference Voltage
4.5
5
C00
Figure 11. DNL vs Reference Voltage
40
60
AVDD = 3 V
REF = 2.5 V
TA = 25oC
AVDD = 3 V
REF = 5 V
TA = 25oC
50
Hits per Code (%)
30
Hits per Code (%)
3.5
4
Reference Voltage (V)
20
40
30
20
10
10
0
0
10
12
14
16
ADC Output Code
18
20
4
Figure 12. DC Input Histogram (VREF = 2.5 V)
10
C01
0
AVDD = 3 V
REF = 2.5 V
TA = 25ƒC
fIN = 1 kHz
SNR = 95.4 dB
THD = ±119 dB
±40
±60
±80
±40
±60
±100
±120
±80
±100
±120
±140
±140
±160
±160
±180
±180
±200
0
100
200
300
Input Frequency (kHz)
400
AVDD = 3 V
REF = 5 V
TA = 25ƒC
fIN = 1 kHz
SNR = 100 dB
THD = ±115 dB
±20
Power (dB)
±20
Power (dB)
8
ADC Output Code
Figure 13. DC Input Histogram (VREF = 5 V)
0
500
±200
0
C011
Figure 14. Typical FFT (VREF = 2.5 V)
14
6
C00
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100
200
300
Input Frequency (kHz)
400
500
C012
Figure 15. Typical FFT (VREF = 5 V)
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Typical Characteristics (continued)
At TA = 25°C, AVDD = 3 V, DVDD = 3 V, VREF = 5 V, and fSAMPLE = 1 MSPS, unless otherwise noted.
102
Signal-to-Noise and Distortion (dBFS)
Signal-to-Noise Ratio (dBFS)
102
101
100
99
98
97
96
95
fIN = 1 kHz
94
101
100
99
98
97
96
95
fIN = 1 kHz
94
2.5
3
3.5
4
Reference Voltage (V)
4.5
5
2.5
Figure 16. SNR vs Reference Voltage
4.5
5
C01
-109
Total Harmonic Distortion (dBFS)
fIN = 1 kHz
17.5
Effective Number of Bits
3.5
4
Reference Voltage (V)
Figure 17. SINAD vs Reference Voltage
18
17
16.5
16
15.5
15
14.5
fIN = 1 kHz
-111
-113
-115
-117
-119
-121
-123
-125
14
2.5
3
3.5
4
Reference Voltage (V)
4.5
2.5
5
3
C01
Figure 18. ENOB vs Reference Voltage
3.5
4
Reference Voltage (V)
4.5
5
C01
Figure 19. THD vs Reference Voltage
102
126
fIN = 1 kHz
124
Signal-to-Noise Ratio (dBFS)
Spurious-Free Dynamic Range (dBFS)
3
C01
122
120
118
116
114
112
fIN = 1 kHz
101
100
99
98
97
96
95
94
110
2.5
3
3.5
4
Reference Voltage (V)
4.5
5
-40
C01
Figure 20. SFDR vs Reference Voltage
-15
10
35
Free-Air Temperature (oC)
60
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C01
Figure 21. SNR vs Temperature
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Typical Characteristics (continued)
At TA = 25°C, AVDD = 3 V, DVDD = 3 V, VREF = 5 V, and fSAMPLE = 1 MSPS, unless otherwise noted.
18
fIN = 1 kHz
101
fIN = 1 kHz
17.5
Effective Number of Bits
Signal-to-Noise and Distortion (dBFS)
102
100
99
98
97
96
95
17
16.5
16
15.5
15
14.5
94
14
-40
-15
10
35
Free-Air Temperature (oC)
60
85
-40
Figure 22. SINAD vs Temperature
Spurious-Free Dynamic Range (dBFS)
Total Harmonic Distortion (dBFS)
fIN = 1 kHz
-113
-115
-117
-119
-121
-123
-125
-15
10
35
Free-Air Temperature (oC)
60
85
C02
126
fIN = 1 kHz
124
122
120
118
116
114
112
85
-40
-15
C02
Figure 24. THD vs Temperature
10
35
Free-Air Temperature (oC)
60
85
C02
Figure 25. SFDR vs Temperature
101
Signal-to-Noise and Distortion (dBFS)
106
Signal-toNoise Ratio (dBFS)
60
110
-40
104
102
100
98
96
94
92
100
99
98
97
96
95
94
93
90
0
20
40
60
Input Frequency (kHz)
80
100
0
C02
Figure 26. SNR vs Input Frequency
16
10
35
Free-Air Temperature (oC)
Figure 23. ENOB vs Temperature
-109
-111
-15
C01
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20
40
60
Input Frequency (kHz)
80
100
C02
Figure 27. SINAD vs Input Frequency
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Typical Characteristics (continued)
At TA = 25°C, AVDD = 3 V, DVDD = 3 V, VREF = 5 V, and fSAMPLE = 1 MSPS, unless otherwise noted.
18
-100
Total Harmonic Distortion (dBFS)
Effective Number of Bits
17.5
17
16.5
16
15.5
15
14.5
14
0
20
40
60
Input Frequency (kHz)
80
-103
-106
-109
-112
-115
-118
-121
-124
100
0
20
C02
124
121
2.3
118
115
112
109
106
100
C02
2.2
2.1
2
1.9
1.8
1.7
103
1.6
100
0
20
40
60
Input Frequency (kHz)
80
-40
100
-15
C02
Figure 30. SFDR vs Input Frequency
10
35
Free-Air Temperature (oC)
60
85
C02
Figure 31. Supply Current vs Temperature
6
2
5.9
1.8
Analog Supply Current (mA)
Power Consumption (mW)
80
Figure 29. THD vs Input Frequency
2.4
Analog Supply Current (mA)
Spurious-Free Dynamic Range (dBFS)
Figure 28. ENOB vs Input Frequency
40
60
Input Frequency (kHz)
5.8
5.7
5.6
5.5
5.4
5.3
5.2
5.1
1.6
1.4
1.2
1
0.8
0.6
0.4
0.2
5
-40
-15
10
35
Free-Air Temperature (oC)
60
85
0
0
C02
Figure 32. Power Consumption vs Temperature
200
400
600
Throughput (kSPS)
800
1000
Figure 33. Supply Current vs Throughput
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Typical Characteristics (continued)
At TA = 25°C, AVDD = 3 V, DVDD = 3 V, VREF = 5 V, and fSAMPLE = 1 MSPS, unless otherwise noted.
200
175
5
Power-Down Current (mA)
Power Consumption (mW)
6
4
3
2
1
150
125
100
75
50
25
0
0
0
200
400
600
Throughput (kSPS)
800
-40
1000
-15
C03
Figure 34. Power Consumption vs Throughput
10
35
Free-Air Temperature (oC)
60
85
C03
Figure 35. Power-Down Current vs Temperature
4
0.01
3
0.006
Gain Error (%FS)
Offset (mV)
2
1
0
-1
-2
0.002
-0.002
-0.006
-3
-4
-0.01
-40
-15
10
35
Free-Air Temperature (oC)
60
85
-40
103
16000
101
14000
99
12000
97
95
93
60
85
C03
AVDD = 3 V
REF = 5 V
TA = 25oC
15000 Devices
10000
8000
6000
91
4000
89
2000
87
0
0.01
0.1
1
Input Frequency (kHz)
10
100
-0.01
-0.005
0
0.005
0.01
Gain Error (% FS)
C03
Figure 38. CMRR vs Input Frequency
18
10
35
Free-Air Temperature (oC)
Figure 37. Gain Error vs Temperature
18000
Frequency
Common-Mode Rejection Ratio (dB)
Figure 36. Offset vs Temperature
105
85
0.001
-15
C03
C03
Figure 39. Typical Distribution of Gain Error
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Typical Characteristics (continued)
At TA = 25°C, AVDD = 3 V, DVDD = 3 V, VREF = 5 V, and fSAMPLE = 1 MSPS, unless otherwise noted.
16000
8000
AVDD = 3 V
REF = 5 V
TA = 25oC
15000 Devices
7000
12000
5000
Frequency
Frequency
6000
AVDD = 3 V
REF = 5 V
TA = 25ƒC
15000 Devices
14000
4000
3000
10000
8000
6000
2000
4000
1000
2000
0
0
-4
-3
-2
-1
0
1
Offset (mV)
2
3
±1.0
4
C03
Figure 40. Typical Distribution of Offset Error
0.0
0.5
1.0
±0.5
Differential Nonlinearity Min and Max (LSB)
1.5
C038
Figure 41. Typical Distribution of Differential Nonlinearity
(Minimum and Maximum)
14000
AVDD = 3 V
REF = 5 V
TA = 25ƒC
15000 Devices
12000
Frequency
10000
8000
6000
4000
2000
0
±3.0 ±2.5 ±2.0 ±1.5 ±1.0 ±0.5 0.0 0.5 1.0 1.5 2.0 2.5 3.0
Integral Nonlinearity Min and Max (LSB)
C039
Figure 42. Typical Distribution of Integral
Nonlinearity (Minimum and Maximum)
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9 Parametric Measurement Information
9.1 Equivalent Circuits
500 µA
IOL
1.4 V
DOUT
20 pF
500 µA
IOH
Figure 43. Load Circuit for Digital Interface Timing
DIN
CONVST
SCLK
VIH
VIL
VOH
VOH
VOL
VOL
SDO
Figure 44. Voltage Levels for Timing
20
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10 Detailed Description
10.1 Overview
The ADS8881 is a high-speed, successive approximation register (SAR), analog-to-digital converter (ADC) from
a 16- and 18-bit device family. This compact device features high performance. Power consumption is inherently
low and scales linearly with sampling speed. The architecture is based on charge redistribution that inherently
includes a sample-and-hold (S/H) function.
The ADS8881 supports a true-differential analog input across two pins (INP and INN). When a conversion is
initiated, the differential input on these pins is sampled on the internal capacitor array. While a conversion is in
progress, both the INP and INN inputs are disconnected from the internal circuit.
The ADS8881 uses an internal clock to perform conversions. The device reconnects the sampling capacitors to
the INP and INN pins after conversion and then enters an acquisition phase. During the acquisition phase, the
device is powered down and the conversion result can be read.
The device digital output is available in SPI-compatible format, thus making interfacing with microprocessors,
digital signal processors (DSPs), or field-programmable gate arrays (FPGAs) easy.
10.2 Functional Block Diagram
Figure 45 shows the detailed functional block diagram for the device.
AVDD
REF
DVDD
REF
CONVST
AINP
Sample
and
Hold
AINN
SCLK
SAR
ADC
ADC
SPI
DOUT
DIN
AGND
REFM
DGND
GND
GND
Figure 45. Detailed Block Diagram
10.3 Feature Description
10.3.1 Analog Input
As shown in Figure 45, the device features a differential analog input. Both positive and negative inputs are
individually sampled on 55-pF sampling capacitors and the device converts for the voltage difference between
the two sampled values: VINP – VINN.
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Feature Description (continued)
Most differential input SAR ADCs prohibit the input common-mode voltage, VCM (that is, the average voltage
between the inputs), at AINP or AINM from varying more than approximately 10% beyond the mid-scale input
value. As shown in Figure 46, the device has a unique common-mode voltage detection and rejection block that
does not have this restriction and thus allows VCM to be set to any value between 0 V and VREF without
degrading device performance.
REF
AINP
+
±
2
I
N
T
E
R
F
A
C
E
Binary
Search
Algorithm
(SAR)
+
±
AINM
Common Mode Voltage
Detection and Rejection
Block
Figure 46. Conceptual Diagram: True Differential Input Structure
Table 1 shows the full-scale input range of the device as a function of input common-mode voltage. The device
offers a maximum dynamic range for VCM = VREF / 2. The differential input with wide common-mode range allows
connecting differential signals from sensors without any signal conditioning.
Table 1. Full-Scale Input Range
ABSOLUTE INPUT RANGE
VCM
FULL SCALE INPUT RANGE (VFS)
VAINP
VAINN
VCM < VREF / 2
0 to 2 × VCM
0 to 2 × VCM
(–2 × VCM) to (2 × VCM)
VCM = VREF / 2
0 to VREF
0 to VREF
(–VREF) to (VREF)
VCM > VREF / 2
(2 × VCM – VREF) to VREF
(2 × VCM – VREF) to VREF
[–2 × (VCM – VREF)] to [2 × (VCM – VREF)]
Figure 47 shows an equivalent circuit of the input sampling stage. The sampling switch is represented by a 96-Ω
resistance in series with the ideal switch; see the ADC Input Driver section for more details on the recommended
driving circuits.
Device in Hold Mode
96
AINP
4 pF
55 pF
REF
4 pF
55 pF
96
GND
GND
AINN
Figure 47. Input Sampling Stage Equivalent Circuit
Figure 45 and Figure 47 illustrate electrostatic discharge (ESD) protection diodes to REF and GND from both
analog inputs. Make sure that these diodes do not turn on by keeping the analog inputs within the specified
range.
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10.3.2 Reference
The device operates with an external reference voltage and switches binary-weighted capacitors onto the
reference terminal (REF pin) during the conversion process. The switching frequency is proportional to the
internal conversion clock frequency but the dynamic charge requirements are a function of the absolute value of
the input voltage and reference voltage. This dynamic load must be supported by a reference driver circuit
without degrading the noise and linearity performance of the device. During the acquisition process, the device
automatically powers down and does not take any dynamic current from the external reference source. The basic
circuit diagram for such a reference driver circuit for precision ADCs is shown in Figure 48; see the ADC
Reference Driver section for more details on the application circuits.
RREF_FLT
Buffer
CREF_FLT
RBUF_FLT
Voltage
Reference
REF
CBUF_FLT
ADC
Figure 48. Reference Driver Schematic
10.3.3 Clock
The device uses an internal clock for conversion. Conversion duration may vary but is bounded by the minimum
and maximum value of tconv, as specified in the Timing Requirements section. An external SCLK is only used for
a serial data read operation. Data are read after a conversion completes and when the device is in acquisition
phase for the next sample.
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10.3.4 ADC Transfer Function
The ADS8881 is a unipolar, differential input device. The device output is in twos compliment format.
Figure 49 shows ideal characteristics for the device. The full-scale range for the ADC input (AINP – AINN) is
equal to twice the reference input voltage to the ADC (2 × VREF). The LSB for the ADC is given by Equation 1.
1 LSB = [2 × (VREF / 218)]
(1)
ADC Code (Hex)
1FFFF
00000
3FFFF
20001
20000
±VREF
+ 1 LSB
±1 LSB
VREF
± 1 LSB
0
VIN
Differential Analog Input
(AINP AINN)
Figure 49. Differential Transfer Characteristics
10.4 Device Functional Modes
The ADS8881 is a low pin-count device. However, the device offers six different options for interfacing with the
digital host.
These options can be broadly classified as being either CS mode (in either a 3- or 4-wire interface) or daisychain mode. The device operates in CS mode if DIN is high at the CONVST rising edge. If DIN is low at the
CONVST rising edge, or if DIN and CONVST are connected together, the device operates in daisy-chain mode.
In both modes, the device can either operate with or without a busy indicator, where the busy indicator is a bit
preceding the output data bits that can be used to interrupt the digital host and trigger the data transfer.
The 3-wire interface in CS mode is useful for applications that need galvanic isolation on-board. The 4-wire
interface in CS mode allows the user to sample the analog input independent of the serial interface timing and,
therefore, allows easier control of an individual device while having multiple, similar devices on-board. The daisychain mode is provided to hook multiple devices in a chain similar to a shift register and is useful in reducing
component count and the number of signal traces on the board.
10.4.1 CS Mode
CS mode is selected if DIN is high at the CONVST rising edge. There are four different interface options
available in this mode: 3-wire CS mode without a busy indicator, 3-wire CS mode with a busy indicator, 4-wire
CS mode without a busy indicator, and 4-wire CS mode with a busy indicator. The following sections discuss
these interface options in detail.
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Device Functional Modes (continued)
10.4.1.1 3-Wire CS Mode Without a Busy Indicator
This interface option is most useful when a single ADC is connected to an SPI-compatible digital host. In this
interface option, DIN can be connected to DVDD and CONVST functions as CS (as shown in Figure 50). As
shown in Figure 51, a CONVST rising edge forces DOUT to 3-state, samples the input signal, and causes the
device to enter a conversion phase. Conversion is done with the internal clock and continues regardless of the
state of CONVST. As a result, CONVST (functioning as CS) can be pulled low after the start of the conversion to
select other devices on the board. However, CONVST must return high before the minimum conversion time
(tconv-min) elapses and is held high until the maximum possible conversion time (tconv-max) elapses. A high level on
CONVST at the end of the conversion ensures the device does not generate a busy indicator.
DVDD
DIN
CONVST
CNV
SCLK
CLK
DOUT
SDI
ADC
Digital Host
Figure 50. Connection Diagram: 3-Wire CS Mode Without a Busy Indicator (DIN = 1)
1/fsample
DIN = HIGH
CONVST = 1
CONVST
œœ
SCLK
1
DOUT
D17
2
3
D16
D15
16
17
18
D1
D0
œœ
D2
œœ
tACQ
tconv-max
tconv-min
ADC
STATE
Acquiring
Sample N
Conversion Result of Sample N Clocked-out
while Acquiring Sample N+1
Converting
Sample N
End-of-Conversion
Figure 51. Interface Timing Diagram: 3-Wire CS Mode (DIN = 1)
When conversion is complete, the device enters an acquisition phase and powers down. CONVST (functioning
as CS) can be brought low after the maximum conversion time (tconv-max) elapses. On the CONVST falling edge,
DOUT comes out of 3-state and the device outputs the MSB of the data. The lower data bits are output on
subsequent SCLK falling edges. Fast sampling rates require high frequency SCLK and data must be read at
SCLK falling edges. For slow sampling rates and SCLK frequency ≤ 36 MHz, data can be read at either SCLK
falling or rising edges. Note that with any SCLK frequency, reading data at SCLK falling edge requires the digital
host to clock in the data during the th_CK_DO-min time frame. DOUT goes to 3-state after the 18th SCLK falling
edge or when CONVST goes high, whichever occurs first.
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Device Functional Modes (continued)
10.4.1.2 3-Wire CS Mode With a Busy Indicator
This interface option is most useful when a single ADC is connected to an SPI-compatible digital host and an
interrupt-driven data transfer is desired. In this interface option, DIN can be connected to DVDD and CONVST
functions as CS (as shown in Figure 52). The pull-up resistor on the DOUT pin ensures that the IRQ pin of the
digital host is held high when DOUT goes to 3-state. As shown in Figure 53, a CONVST rising edge forces
DOUT to 3-state, samples the input signal, and causes the device to enter a conversion phase. Conversion is
done with the internal clock and continues regardless of the state of CONVST. As a result, CONVST (functioning
as CS) can be pulled low after the start of the conversion to select other devices on the board. However,
CONVST must be pulled low before the minimum conversion time (tconv-min) elapses and must remain low until
the maximum possible conversion time (tconv-max) elapses. A low level on the CONVST input at the end of a
conversion ensures the device generates a busy indicator.
DVDD
CNV
CONVST
CLK
SCLK
DIN
DVDD
DOUT
SDI
ADC
IRQ
Digital Host
Figure 52. Connection Diagram: 3-Wire CS Mode With a Busy Indicator
1/fsample
DIN = DVDD
CONVST
CONVST = 0
œœ
SCLK
1
2
3
D17
D16
17
18
19
D1
D0
œœ
SDO Pulled-up
DOUT
BUSY
D2
œœ
tACQ
tconv-max
tconv-min
ADC
STATE
Acquiring
Sample N
Conversion Result of Sample N Clocked-out
while Acquiring Sample N+1
Converting
Sample N
End-of-Conversion
Figure 53. Interface Timing Diagram: 3-Wire CS Mode With a Busy Indicator (DIN = 1)
When conversion is complete, the device enters an acquisition phase and powers down, DOUT comes out of 3state, and the device outputs a busy indicator bit (low level) on the DOUT pin. This configuration provides a highto-low transition on the IRQ pin of the digital host. The data bits are clocked out, MSB first, on the subsequent
SCLK falling edges. Fast sampling rates require high frequency SCLK and data must be read at SCLK falling
edges. For slow sampling rates and SCLK frequency ≤ 36 MHz, data can be read at either SCLK falling or rising
edges. Note that with any SCLK frequency, reading data at SCLK falling edge requires the digital host to clock in
the data during the th_CK_DO-min time frame. DOUT goes to 3-state after the 19th SCLK falling edge or when
CONVST goes high, whichever occurs first.
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Device Functional Modes (continued)
10.4.1.3 4-Wire CS Mode Without a Busy Indicator
This interface option is useful when one or more ADCs are connected to an SPI-compatible digital host.
Figure 54 shows the connection diagram for single ADC; see Figure 56 for the connection diagram for two ADCs.
CS
CNV
DIN
CONVST
DOUT
SDI
SCLK
CLK
ADC
Digital Host
Figure 54. Connection Diagram: Single ADC With 4-Wire CS Mode Without a Busy Indicator
In this interface option, DIN is controlled by the digital host and functions as CS. As shown in Figure 55, with DIN
high, a CONVST rising edge selects CS mode, forces DOUT to 3-state, samples the input signal, and causes the
device to enter a conversion phase. In this interface option, CONVST must be held at a high level from the start
of the conversion until all data bits are read. Conversion is done with the internal clock and continues regardless
of the state of DIN. As a result, DIN (functioning as CS) can be pulled low to select other devices on the board.
However, DIN must be pulled high before the minimum conversion time (tconv-min) elapses and remains high until
the maximum possible conversion time (tconv-max) elapses. A high level on DIN at the end of the conversion
ensures the device does not generate a busy indicator.
1/fsample
tconv-max
tACQ
tconv-min
CONVST
DIN = 1
DIN
œœ
SCLK
1
2
DOUT
D17
D16
17
18
œœ
ADC
STATE
End-ofConversion
Acquiring
Sample N
D1
D0
œœ
Read Sample N
Converting
Sample N
Acquiring Sample N+1
Figure 55. Interface Timing Diagram: Single ADC With 4-Wire CS Mode
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Device Functional Modes (continued)
When conversion is complete, the device enters acquisition phase and powers down. DIN (functioning as CS)
can be brought low after the maximum conversion time (tconv-max) elapses. On the DIN falling edge, DOUT comes
out of 3-state and the device outputs the MSB of the data. The lower data bits are output on subsequent SCLK
falling edges. Fast sampling rates require high frequency SCLK and data must be read at SCLK falling edges.
For slow sampling rates and SCLK frequency ≤ 36 MHz, data can be read at either SCLK falling or rising edges.
Note that with any SCLK frequency, reading data at SCLK falling edge requires the digital host to clock in the
data during the th_CK_DO-min time frame. DOUT goes to 3-state after the 18th SCLK falling edge or when DIN goes
high, whichever occurs first.
As shown in Figure 56, multiple devices can be hooked together on the same data bus. In this case, as shown in
Figure 57, the DIN of the second device (functioning as CS for the second device) can go low after the first
device data are read and the DOUT of the first device is in 3-state.
Care must be taken so that CONVST and DIN are not both low together at any time during the cycle.
CS1
CS2
CNV
CONVST
DIN
CONVST
DIN
DOUT
DOUT
SCLK
SDI
SCLK
CLK
ADC #1
ADC #2
Digital Host
Figure 56. Connection Diagram: Two ADCs With 4-Wire CS Mode Without a Busy Indicator
1/fsample
tconv-max
tACQ
tconv-min
CONVST
DIN = 1
DIN
(ADC 1)
DIN = 1
DIN
(ADC 2)
œœ
SCLK
1
2
DOUT
D17
D16
œœ
17
18
19
20
D0
D17
D16
ADC
STATE
Acquiring
Sample N
Converting
Sample N
36
œœ
œœ
End-ofConversion
35
D1
D1
D0
œœ
œœ
Read Sample N
ADC 1
Read Sample N
ADC 2
Acquiring Sample N+1
Figure 57. Interface Timing Diagram: Two ADCs With 4-Wire CS Mode
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Device Functional Modes (continued)
10.4.1.4 4-Wire CS Mode With a Busy Indicator
This interface option is most useful when a single ADC is connected to an SPI-compatible digital host and an
interrupt-driven data transfer is desired. In this interface option, the analog sample is least affected by clock jitter
because the CONVST signal (used to sample the input) is independent of the data read operation. In this
interface option, DIN is controlled by the digital host and functions as CS (as shown in Figure 58). The pull-up
resistor on the DOUT pin ensures that the IRQ pin of the digital host is held high when DOUT goes to 3-state. As
shown in Figure 59, when DIN is high, a CONVST rising edge selects CS mode, forces DOUT to 3-state,
samples the input signal, and causes the device to enter a conversion phase. In this interface option, CONVST
must be held high from the start of the conversion until all data bits are read. Conversion is done with the internal
clock and continues regardless of the state of DIN. As a result, DIN (acting as CS) can be pulled low to select
other devices on the board. However, DIN must be pulled low before the minimum conversion time (tconv-min)
elapses and remains low until the maximum possible conversion time (tconv-max) elapses. A low level on the DIN
input at the end of a conversion ensures the device generates a busy indicator.
CS
DIN
CNV
CONVST
CLK
SCLK
DVDD
DOUT
SDI
IRQ
ADC
Digital Host
Figure 58. Connection Diagram: 4-Wire CS Mode With a Busy Indicator
1/fsample
tACQ
tconv-max
tconv-min
CONVST
DIN =0
DIN
œœ
SCLK
1
2
3
D17
D16
17
18
19
D1
D0
œœ
DOUT
ADC
STATE
SDO Pulled-up
Acquiring
Sample N
Converting
Sample N
BUSY
D2
œœ
Conversion Result of Sample N Clocked-out
while Acquiring Sample N+1
Figure 59. Interface Timing Diagram: 4-Wire CS Mode With a Busy Indicator
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Device Functional Modes (continued)
When conversion is complete, the device enters an acquisition phase and powers down, DOUT comes out of 3state, and the device outputs a busy indicator bit (low level) on the DOUT pin. This configuration provides a highto-low transition on the IRQ pin of the digital host. The data bits are clocked out, MSB first, on the subsequent
SCLK falling edges. Fast sampling rates require high frequency SCLK and data must be read at SCLK falling
edges. For slow sampling rates and SCLK frequency ≤ 36 MHz, data can be read at either SCLK falling or rising
edges. Note that with any SCLK frequency, reading data at SCLK falling edge requires the digital host to clock in
the data during the th_CK_DO-min time frame. DOUT goes to 3-state after the 19th SCLK falling edge or when DIN
goes high, whichever occurs first. Care must be taken so that CONVST and DIN are not both low together at any
time during the cycle.
10.4.2 Daisy-Chain Mode
Daisy-chain mode is selected if DIN is low at the time of a CONVST rising edge or if DIN and CONVST are
connected together. Similar to CS mode, this mode features operation with or without a busy indicator. The
following sections discuss these interface modes in detail.
10.4.2.1 Daisy-Chain Mode Without a Busy Indicator
This interface option is most useful in applications where multiple ADC devices are used but the digital host has
limited interfacing capability. Figure 60 shows a connection diagram with N ADCs connected in the daisy-chain.
The CONVST pins of all ADCs in the chain are connected together and are controlled by a single pin of the
digital host. Similarly, the SCLK pins of all ADCs in the chain are connected together and are controlled by a
single pin of the digital host. The DIN pin for ADC 1 (DIN-1) is connected to GND. The DOUT pin of ADC 1
(DOUT-1) is connected to the DIN pin of ADC 2 (DIN-2), and so on. The DOUT pin of the last ADC in the chain
(DOUT-N) is connected to the SDI pin of the digital host.
CNV
CONVST
DIN1
DOUT1
SCLK
CONVST
DIN2
DOUT2
CONVST
}
SCLK
DINN-2
DOUTN-1
SCLK
CONVST
DINN-1
DOUTN
SDI
SCLK
CLK
ADC 1
ADC 2
}
ADC N 1
ADC N
Digital Host
Figure 60. Connection Diagram: Daisy-Chain Mode Without a Busy Indicator (DIN = 0)
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Device Functional Modes (continued)
As shown in Figure 61, the device DOUT pin is driven low when DIN and CONVST are low together. With DIN
low, a CONVST rising edge selects daisy-chain mode, samples the analog input, and causes the device to enter
a conversion phase. In this interface option, CONVST must remain high from the start of the conversion until all
data bits are read. When started, the conversion continues regardless of the state of SCLK, however SCLK must
be low at the CONVST rising edge so that the device does not generate a busy indicator at the end of the
conversion.
tconv-min
1/fsample
tconv-max
tACQ
CONVST
œœ
SCLK
1
2
œœ
17
18
19
20
35
36
DIN-1 = LOW
DOUT-1
& DIN-2
D17
D17
DOUT-2
End-ofConversion
ADC
STATE
Acquiring
6DPSOH µQ¶
Converting
6DPSOH µQ¶
ADC 1 data
œœ
D16
D1
œœ
D16
œœ
D0
D1
D0
ADC 1 data
œœ
D16
D1
œœ
D17
œœ
ADC 2 data
ADC 2 GDWD IRU VDPSOH µQ¶
D0
ADC 1 GDWD IRU VDPSOH µQ¶
$FTXLULQJ 6DPSOH µQ+1¶
Figure 61. Interface Timing Diagram: For Two Devices in Daisy-Chain Mode
At the end of conversion, every ADC in the chain loads its own conversion result into the internal, 18-bit, shift
register and also outputs the MSB bit of this conversion result on its own DOUT pin. All ADCs enter an
acquisition phase and power-down. On every subsequent SCLK falling edge, the internal shift register of each
ADC latches the data available on its DIN pin and shifts out the next bit of data on its DOUT pin. Therefore, the
digital host receives the data of ADC N, followed by the data of ADC N–1, and so on (in MSB-first fashion). A
total of 18 x N SCLK falling edges are required to capture the outputs of all N devices in the chain. Fast sampling
rates require high frequency SCLK and data must be read at SCLK falling edges. For slow sampling rates and
SCLK frequency ≤ 36 MHz, data can be read at either SCLK falling or rising edges. Note that with any SCLK
frequency, reading data at SCLK falling edge requires the digital host to clock in the data during the th_CK_DO-min
time frame.
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Device Functional Modes (continued)
10.4.2.2 Daisy-Chain Mode With a Busy Indicator
This interface option is most useful in applications where multiple ADC devices are used but the digital host has
limited interfacing capability and an interrupt-driven data transfer is desired. Figure 62 shows a connection
diagram with N ADCs connected in the daisy-chain. The CONVST pins of all ADCs in the chain are connected
together and are controlled by a single pin of the digital host. Similarly, the SCLK pins of all ADCs in the chain
are connected together and are controlled by a single pin of the digital host. The DIN pin for ADC 1 (DIN-1) is
connected to its CONVST. The DOUT pin of ADC 1 (DOUT-1) is connected to the DIN pin of ADC 2 (DIN-2), and
so on. The DOUT pin of the last ADC in the chain (DOUT-N) is connected to the SDI and IRQ pins of the digital
host.
CNV
CONVST
DIN1
DOUT1
CONVST
DIN2
SCLK
DOUT2
CONVST
}
SCLK
DINN-1
CONVST
DOUTN-1
DINN
SCLK
IRQ
DOUTN
SDI
SCLK
CLK
ADC 1
ADC 2
}
ADC N
ADC N 1
Digital Host
Figure 62. Connection Diagram: Daisy-Chain Mode With a Busy Indicator (DIN = 0)
As shown in Figure 63, the device DOUT pin is driven low when DIN and CONVST are low together. A CONVST
rising edge selects daisy-chain mode, samples the analog input, and causes the device to enter a conversion
phase. In this interface option, CONVST must remain high from the start of the conversion until all data bits are
read. When started, the conversion continues regardless of the state of SCLK, however SCLK must be high at
the CONVST rising edge so that the device generates a busy indicator at the end of the conversion.
tconv-min
1/fsample
tconv-max
tACQ
CONVST
œœ
SCLK
1
th-DI-CNV
DIN-1 =
CONVST
DOUT-1
& DIN-2
BUSY
BUSY
DOUT-2
End-ofConversion
ADC
STATE
Acquiring
6DPSOH µQ¶
Converting
6DPSOH µQ¶
2
œœ
18
19
20
ADC 1 data
œœ
D17
D1
D0
œœ
D17
œœ
21
36
37
ADC 1 data
œœ
D1
D0
D17
œœ
ADC 2 data
ADC 2 GDWD IRU VDPSOH µQ¶
D16
D1
D0
œœ
ADC 1 GDWD IRU VDPSOH µQ¶
$FTXLULQJ 6DPSOH µQ+1¶
Figure 63. Interface Timing Diagram: For Two Devices in Daisy-Chain Mode With a Busy Indicator
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Device Functional Modes (continued)
At the end of conversion, every ADC in the chain loads its own conversion result into the internal, 18-bit, shift
register and also forces its DOUT pin high, thereby providing a low-to-high transition on the IRQ pin of the digital
host. All ADCs enter an acquisition phase and power-down. On every subsequent SCLK falling edge, the internal
shift register of each ADC latches the data available on its DIN pin and shifts out the next bit of data on its DOUT
pin. Therefore, the digital host receives the interrupt signal followed by the data of ADC N followed by the data of
ADC N–1, and so on (in MSB-first fashion). A total of (18 × N) + 1 SCLK falling edges are required to capture the
outputs of all N devices in the chain. Fast sampling rates require a high-frequency SCLK and data must be read
at the SCLK falling edges. For slow sampling rates and SCLK frequency ≤ 36 MHz, data can be read at either
SCLK falling or rising edges. Note that with any SCLK frequency, reading data at the SCLK falling edge requires
the digital host to clock in the data during the th_CK_DO-min time frame. Note that the busy indicator bits of ADC 1 to
ADC N–1 do not propagate to the next device in the chain.
NOTE: For SCLK ≤ 36 MHz, SPI mode-3 (CPOL = 1, CPHA = 1) allows reading the conversion results of N
ADCs in 18 × N SCLK cycles because the busy indicator bit is not clocked in by the host.
11 Application and Implementation
NOTE
Information in the following applications sections is not part of the TI component
specification, and TI does not warrant its accuracy or completeness. TI’s customers are
responsible for determining suitability of components for their purposes. Customers should
validate and test their design implementation to confirm system functionality.
11.1 Application Information
The two primary circuits required to maximize the performance of a high-precision, successive approximation
register (SAR), analog-to-digital converter (ADC) are the input driver and the reference driver circuits. This
section details some general principles for designing these circuits, followed by some application circuits
designed using the ADS8881.
11.1.1 ADC Reference Driver
The external reference source to the ADS8881 must provide low-drift and very accurate voltage for the ADC
reference input and support the dynamic charge requirements without affecting the noise and linearity
performance of the device. The output broadband noise of most references can be in the order of a few hundred
μVRMS. Therefore, to prevent any degradation in the noise performance of the ADC, the output of the voltage
reference must be appropriately filtered by using a low-pass filter with a cutoff frequency of a few hundred hertz.
After band-limiting the noise of the reference circuit, the next important step is to design a reference buffer that
can drive the dynamic load posed by the reference input of the ADC. The reference buffer must regulate the
voltage at the reference pin such that the value of VREF stays within the 1-LSB error at the start of each
conversion. This condition necessitates the use of a large capacitor, CBUF_FLT (see Figure 48) for regulating the
voltage at the reference input of the ADC. The amplifier selected to drive the reference pin must have an
extremely low offset and temperature drift with a low output impedance to drive the capacitor at the ADC
reference pin without any stability issues.
11.1.2 ADC Input Driver
The input driver circuit for a high-precision ADC mainly consists of two parts: a driving amplifier and a fly-wheel
RC filter. The amplifier is used for signal conditioning of the input voltage and its low output impedance provides
a buffer between the signal source and the switched capacitor inputs of the ADC. The RC filter helps attenuate
the sampling charge injection from the switched-capacitor input stage of the ADC and functions as an antialiasing
filter to band-limit the wideband noise contributed by the front-end circuit. Careful design of the front-end circuit is
critical to meet the linearity and noise performance of a high-precision, 18-bit ADC such as the ADS8881.
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Application Information (continued)
11.1.2.1 Input Amplifier Selection
Selection criteria for the input amplifiers is highly dependent on the input signal type as well as the performance
goals of the data acquisition system. Some key amplifier specifications to consider while selecting an appropriate
amplifier to drive the inputs of the ADC are:
• Small-signal bandwidth. Select the small-signal bandwidth of the input amplifiers to be as high as possible
after meeting the power budget of the system. Higher bandwidth reduces the closed-loop output impedance
of the amplifier, thus allowing the amplifier to more easily drive the low cutoff frequency RC filter (see the
Antialiasing Filter section) at the inputs of the ADC. Higher bandwidth also minimizes the harmonic distortion
at higher input frequencies. In order to maintain the overall stability of the input driver circuit, select the
amplifier bandwidth as described in Equation 2:
§
1
Unity Gain Bandwidth t 4 u ¨¨
© 2S u ( RFLT RFLT ) u C FLT
•
·
¸¸
¹
(2)
Noise. Noise contribution of the front-end amplifiers must be as low as possible to prevent any degradation in
SNR performance of the system. As a rule of thumb, to ensure that the noise performance of the data
acquisition system is not limited by the front-end circuit, the total noise contribution from the front-end circuit
must be kept below 20% of the input-referred noise of the ADC. Noise from the input driver circuit is bandlimited by designing a low cutoff frequency RC filter, as explained in Equation 3.
§ V 1 _ AM P_ PP ·
¨
¸
NG u 2 u ¨ f
¸¸
6
.
6
¨
©
¹
2
en2 _ RM S u
S
uf
2
3 dB
d
1 VREF
u
u 10
5
2
§ SNR dB ·
¨
¸
20
©
¹
where:
•
•
•
•
•
V1 / f_AMP_PP is the peak-to-peak flicker noise in µV,
en_RMS is the amplifier broadband noise density in nV/√Hz,
f–3dB is the 3-dB bandwidth of the RC filter, and
NG is the noise gain of the front-end circuit, which is equal to 1 in a buffer configuration.
THD AMP d THD ADC
•
(3)
Distortion. Both the ADC and the input driver introduce nonlinearity in a data acquisition block. As a rule of
thumb, to ensure that the distortion performance of the data acquisition system is not limited by the front-end
circuit, the distortion of the input driver must be at least 10 dB lower than the distortion of the ADC, as shown
in Equation 4.
10 dB
(4)
Settling Time. For dc signals with fast transients that are common in a multiplexed application, the input signal
must settle within an 18-bit accuracy at the device inputs during the acquisition time window. This condition is
critical to maintain the overall linearity performance of the ADC. Typically, the amplifier data sheets specify
the output settling performance only up to 0.1% to 0.001%, which may not be sufficient for the desired 18-bit
accuracy. Therefore, always verify the settling behavior of the input driver by TINA™-SPICE simulations
before selecting the amplifier.
11.1.2.2 Antialiasing Filter
Converting analog-to-digital signals requires sampling an input signal at a constant rate. Any higher frequency
content in the input signal beyond half the sampling frequency is digitized and folded back into the low-frequency
spectrum. This process is called aliasing. Therefore, an analog, antialiasing filter must be used to remove the
harmonic content from the input signal before being sampled by the ADC. An antialiasing filter is designed as a
low-pass, RC filter, for which the 3-dB bandwidth is optimized based on specific application requirements. For dc
signals with fast transients (including multiplexed input signals), a high-bandwidth filter is designed to allow
accurately settling the signal at the inputs of the ADC during the small acquisition time window. For ac signals,
keep the filter bandwidth low to band-limit the noise fed into the input of the ADC, thereby increasing the signalto-noise ratio (SNR) of the system.
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Application Information (continued)
Besides filtering the noise from the front-end drive circuitry, the RC filter also helps attenuate the sampling
charge injection from the switched-capacitor input stage of the ADC. A differential filter capacitor, CFLT, is
connected across the inputs of the ADC (as shown in Figure 64). This capacitor helps reduce the sampling
charge injection and provides a charge bucket to quickly charge the internal sample-and-hold capacitors during
the acquisition process. As a rule of thumb, the value of this capacitor must be at least 10 times the specified
value of the ADC sampling capacitance. For the ADS8881, the input sampling capacitance is equal to 59 pF,
thus the value of CFLT must be greater than 590 pF. The capacitor must be a COG- or NPO-type because these
capacitor types have a high-Q, low-temperature coefficient, and stable electrical characteristics under varying
voltages, frequency, and time.
RFLT ” 22
f
3 dB
2S u R FLT
1
R FLT u CFLT
CFLT • 590 pF
V
AINP
+
Device
AINM
GND
RFLT ” 22
Figure 64. Antialiasing Filter
Note that driving capacitive loads can degrade the phase margin of the input amplifiers, thus making the amplifier
marginally unstable. To avoid amplifier stability issues, series isolation resistors (RFLT) are used at the output of
the amplifiers. A higher value of RFLT is helpful from the amplifier stability perspective, but adds distortion as a
result of interactions with the nonlinear input impedance of the ADC. Distortion increases with source impedance,
input signal frequency, and input signal amplitude. Therefore, the selection of RFLT requires balancing the stability
and distortion of the design. For the ADS8881, TI recommends limiting the value of RFLT to a maximum of 22 Ω
in order to avoid any significant degradation in linearity performance. The tolerance of the selected resistors can
be chosen as 1% because the use of a differential capacitor at the input balances the effects resulting from any
resistor mismatch.
The input amplifier bandwidth must be much higher than the cutoff frequency of the antialiasing filter. TI strongly
recommends performing a SPICE simulation to confirm that the amplifier has more than 40° phase margin with
the selected filter. Simulation is critical because even with high-bandwidth amplifiers, some amplifiers might
require more bandwidth than others to drive similar filters. If an amplifier has less than a 40° phase margin with
22-Ω resistors, using a different amplifier with higher bandwidth or reducing the filter cutoff frequency with a
larger differential capacitor is advisable.
This section describes some common application circuits using the ADS8881. These data acquisition (DAQ)
blocks are optimized for specific input types and performance requirements of the system. For simplicity, powersupply decoupling capacitors are not shown in these circuit diagrams; see the Power Supply section for
suggested guidelines.
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11.2 Typical Applications
11.2.1 DAQ Circuit for a 1-µs, Full-Scale Step Response
Reference Drive Circuit
20 k
1 µF
THS4281
+
0.2
OPA+
-
1k
+
OPA333
1 µF
+
1k
+
OPA+
10 µF
REF5045
(See Reference Datasheet
for Detailed Pin Configuration)
OPA+
1 µF
+
OPA+
VIN+
+
AVDD
+
OPA350
10
REF AVDD
VCM
V
+
AINP
1 nF
CONVST
Device
AINM
GND
+
VIN-
-
CONVST
10
OPA350
+
+
SAR ADC
Input Driver
OPA+
Figure 65. DAQ Circuit for a 1-µs, Full-Scale Step Response
11.2.1.1 Design Requirements
Step input signals are common in multiplexed applications when switching between different channels. In a
worst-case scenario, one channel is at the negative full-scale (NFS) and the other channel is at the positive fullscale (PFS) voltage, in which case the step size is the full-scale range (FSR) of the ADC when the MUX channel
is switched.
Design an application circuit optimized for using the ADS8881 to achieve
• full-scale step input settling to 18-bit accuracy and
• INL of < ±2 LSB and
• maximum specified throughput of 1 MSPS
11.2.1.2 Detailed Design Procedure
The application circuit is shown in Figure 65.
In such applications, the primary design requirement is to ensure that the full-scale step input signal settles to 18bit accuracy at the ADC inputs. This condition is critical to achieve the excellent linearity specifications of the
ADC. Therefore, the bandwidth of the antialiasing RC filter must be large enough to allow optimal settling of the
input signal during the ADC acquisition time. The filter capacitor helps reduce the sampling charge injection at
the ADC inputs, but degrades the phase margin of the driving amplifier, thereby leading to stability issues.
Amplifier stability is maintained by the series isolation resistor.
The application circuit in Figure 65 shows the schematic of a complete reference driver circuit that generates a
voltage of 4.5 V dc using a single 5-V supply. This circuit is suitable to drive the reference of the ADS8881 at
higher sampling rates up to 1 MSPS. The reference voltage of 4.5 V in this design is generated by the highprecision, low-noise REF5045 circuit. The output broadband noise of the reference is heavily filtered by a lowpass filter with a 3-dB cutoff frequency of 160 Hz.
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Typical Applications (continued)
The reference buffer is designed with the THS4281 and OPA333 in a composite architecture to achieve superior
dc and ac performance at a reduced power consumption, compared to using a single high-performance amplifier.
The THS4281 is a high-bandwidth amplifier with a very low output impedance of 1 Ω at a frequency of 1 MHz.
The low output impedance makes the THS4281 a good choice for driving a high capacitive load to regulate the
voltage at the reference input of the ADC. The high offset and drift specifications of the THS4281 are corrected
by using a dc-correcting amplifier (OPA333) inside the feedback loop. The composite scheme inherits the
extremely low offset and temperature drift specifications of the OPA333.
For the input driving amplifiers, key specifications include rail-to-rail input and output swing, high bandwidth, high
slew rate, and fast settling time. The OPA350 CMOS amplifier meets all these specification requirements for this
circuit with a single-supply and low quiescent current. The component values of the antialiasing filter are selected
to meet the settling requirements of the system as well as to maintain the stability of the input driving amplifiers.
11.2.1.3 Application Curve
1.5
Integral Nonlinearity (LSB)
1.25
1
0.75
0.5
0.25
0
-0.25
-0.5
-0.75
-1
-1.25
-1.5
±131072
131071
Code
C001
18-bit INL
Figure 66. Limited Point Linearity
For step-by-step design procedure, circuit schematics, bill of materials, PCB files, simulation results, and test
results, see 18-Bit Data Acquisition (DAQ) Block Optimized for 1-μs Full-Scale Step Response (TIDU012).
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11.2.2 Low-Power DAQ Circuit for Excellent Dynamic Performance at 1 MSPS
Reference Drive Circuit
20 k
1 µF
THS4281
+
0.2
OPA+
-
1k
+
OPA333
1 µF
+
1k
+
OPA+
10 µF
REF5045
(See Reference Datasheet
for Detailed Pin Configuration)
OPA+
1 µF
+
OPA+
VIN+
+
AVDD
+
OPA320
22
REF AVDD
VCM
V
+
AINP
2.2 nF
CONVST
Device
AINM
GND
+
VIN-
-
CONVST
22
OPA320
+
+
SAR ADC
OPA+
Input Driver
Figure 67. DAQ Circuit for Lowest Power and Excellent Dynamic Performance at 1 MSPS
11.2.2.1 Design Requirements
Design an application circuit optimized for using the ADS8881 to achieve
• ENOB > 17 bits and
• < 35 mW of total power consumption and
• maximum specified throughput of 1 MSPS
11.2.2.2 Detailed Design Procedure
The application circuit in Figure 67 shows the schematic of a complete reference driver circuit that generates a
voltage of 4.5 V dc using a single 5-V supply. This circuit is suitable to drive the reference of the ADS8881 at
higher sampling rates up to 1 MSPS. The reference voltage of 4.5 V in this design is generated by the highprecision, low-noise REF5045 circuit. The output broadband noise of the reference is heavily filtered by a lowpass filter with a 3-dB cutoff frequency of 160 Hz.
The reference buffer is designed with the THS4281 and OPA333 in a composite architecture to achieve superior
dc and ac performance at a reduced power consumption, compared to using a single high-performance amplifier.
The THS4281 is a high-bandwidth amplifier with a very low output impedance of 1 Ω at a frequency of 1 MHz.
The low output impedance makes the THS4281 a good choice for driving a high capacitive load to regulate the
voltage at the reference input of the ADC. The high offset and drift specifications of the THS4281 are corrected
by using a dc-correcting amplifier (OPA333) inside the feedback loop. The composite scheme inherits the
extremely low offset and temperature drift specifications of the OPA333.
In such applications, the input driver must be low in power and noise as well as able to support rail-to-rail input
and output swing with a single supply. A high amplifier bandwidth is also preferred to help attenuate highfrequency distortion. However, oftentimes bandwidth and noise are traded off with the power consumption of the
amplifier. This circuit uses the OPA320 as the front-end driving amplifier because this device has a relatively low
noise density of 7 nV/√Hz for a maximum-specified quiescent current of 1.45 mA per channel.
The noise contribution from the front-end amplifier is band-limited by the 3-dB bandwidth of the RC filter and is
designed to be 1.65 MHz in this application. Again, the component values of the antialiasing filter are carefully
selected to maintain the stability of the input driving amplifiers.
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11.2.2.3 Application Curve
16000
14000
Number of Hits
12000
10000
8000
6000
4000
2000
0
-11
-10
-9
-8
-7
-6
Code
-5
-4
C002
Vdiff close to 0 V, 32768 data points, standard deviation = 0.82 bits,
ENOB (dc) = 17.18 bits
Figure 68. DC Input Histogram
For step-by-step design procedure, circuit schematics, bill of materials, PCB files, simulation results, and test
results, see 18-Bit, 1-MSPS Data Acquisition (DAQ) Block Optimized for Lowest Power (SLAU513).
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11.2.3 DAQ Circuit for Lowest Distortion and Noise Performance at 1 MSPS
Reference Drive Circuit
20 k
1 µF
THS4281
+
0.2
OPA+
-
1k
+
OPA333
+
1 µF
1k
REF5045
+
(See Reference Datasheet
for Detailed Pin Configuration)
OPA+
OPA+
10 µF
1 µF
1k
1k
AVDD
OPA+
VIN+
VCM
+
10
+
REF AVDD
V
+
AINP
THS4521
+
-
10 nF
AINM
CONVST
GND
10
+
VIN-
CONVST
Device
1k
1k
SAR ADC
Input Driver
Figure 69. Differential Input DAQ Circuit for Lowest Distortion and Noise at 1 MSPS
Reference Drive Circuit
20 k
1 µF
THS4281
+
0.2
OPA+
-
1k
+
OPA333
1 µF
+
1k
REF5045
+
(See Reference Datasheet
for Detailed Pin Configuration)
OPA+
OPA+
10 µF
1 µF
1k
1k
AVDD
OPA+
VIN+
+
THS4521
+
-
10
REF AVDD
+
10 nF
CONVST
Device
AINM
10
1k
V
+
AINP
GND
1k
CONVST
SAR ADC
Input Driver
Figure 70. Single-Ended to Differential DAQ Circuit for Lowest Distortion and Noise at 1 MSPS
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11.2.3.1 Design Requirements
Design an application circuit optimized for using the ADS8881 to achieve
• > 98.5-dB SNR, < –110-dB THD and
• ± 1.5-LSB linearity and
• maximum specified throughput of 1 MSPS
11.2.3.2 Detailed Design Procedure
The application circuits are shown in Figure 69 and Figure 70. In both applications, the input signal is processed
through a high-bandwidth, low-distortion, fully-differential amplifier (FDA) designed in an inverting gain
configuration and a low-pass RC filter before being fed into the ADC.
The reference driver circuit, shown in Figure 69 and Figure 70, generates a voltage of 4.5 V dc using a single 5V supply. This circuit is suitable to drive the reference of the ADS8881 at higher sampling rates up to 1 MSPS.
The reference voltage of 4.5 V in this design is generated by the high-precision, low-noise REF5045 circuit. The
output broadband noise of the reference is heavily filtered by a low-pass filter with a 3-dB cutoff frequency of 160
Hz.
The reference buffer is designed with the THS4281 and OPA333 in a composite architecture to achieve superior
dc and ac performance at a reduced power consumption, compared to using a single high-performance amplifier.
The THS4281 is a high-bandwidth amplifier with a very low output impedance of 1 Ω at a frequency of 1 MHz.
The low output impedance makes the THS4281 a good choice for driving a high capacitive load to regulate the
voltage at the reference input of the ADC. The high offset and drift specifications of the THS4281 are corrected
by using a dc-correcting amplifier (OPA333) inside the feedback loop. The composite scheme inherits the
extremely low offset and temperature drift specifications of the OPA333.
As a rule of thumb, the distortion from the input driver must be at least 10 dB less than the ADC distortion. The
distortion resulting from variation in the common-mode signal is eliminated by using the FDA in an inverting gain
configuration that establishes a fixed common-mode level for the circuit. This configuration also eliminates the
requirement of a rail-to-rail swing at the amplifier input. Therefore, these circuits use the low-power THS4521 as
an input driver that provides exceptional ac performance because of its extremely low-distortion and highbandwidth specifications.
In addition, the components of the antialiasing filter are such that the noise from the front-end circuit is kept low
without adding distortion to the input signal.
The circuit in Figure 69 shows a fully-differential DAQ block optimized for low distortion and noise using the
THS4521 and ADS8881. This front-end circuit configuration requires a differential signal at the input of the FDA
and provides a differential output to drive the ADC inputs. The common-mode voltage of the input signal provided
to the ADC is set by the VOCM pin of the THS4521 (not shown in Figure 69). To use the complete dynamic range
of the ADC, VOCM can be set to VREF / 2 by using a simple resistive divider. However, note that the ADS8881
allows the common-mode input voltage (VCM) to be set to any value in the range of 0 V to VREF.
The circuit in Figure 70 shows a single-ended to differential DAQ block optimized for low distortion and noise
using the THS4521 and the ADS8881. This front-end circuit configuration requires a single-ended ac signal at
the input of the FDA and provides a fully-differential output to drive the ADC inputs. The common-mode voltage
of the input signal provided to the ADC is set by the VOCM pin of the THS4521 (not shown in Figure 70). To use
the complete dynamic range of the ADC, VOCM can be set to VREF / 2 by using a simple resistive divider.
However, note that the ADS8881 allows the common-mode input voltage (VCM) to be set to any value in the
range of 0 V to VREF.
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11.2.3.3 Application Curve
0
Signal Power (dB)
±20
±40
±60
±80
±100
±120
±140
±160
0
100
200
300
400
Input Frequency (kHz)
500
C003
fIN = 10 kHz, SNR = 99 dB, THD = –112 dB
Figure 71. FFT Plot
For step-by-step design procedure, circuit schematics, bill of materials, PCB files, simulation results, and test
results, see 18-Bit, 1-MSPS Data Acquisition (DAQ) Block Optimized for Lowest Distortion and Noise
(SLAU515).
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11.2.4 Ultralow-Power DAQ Circuit at 10 kSPS
1k
Reference Drive Circuit
1 µF
AVDD
-
4.7
OPA313
10 k
REF3325
+ +
(See Reference Datasheet
for Detailed Pin Configuration)
22 µF
1 µF
AVDD
10 k
AVDD
10 nF
REF AVDD
AINP
CONVST
Device
AINM
-
1k
CONVST
GND
+
OPA333
+ +
VIN+
VCM
AVDD
4.7 nF
VIN+
+ +
1k
OPA333
Device running at 10 kSPS.
-
10 nF
10 k
Input Driver
Figure 72. Ultralow-Power DAQ Circuit at 10 kSPS
11.2.4.1 Design Requirements
Portable and battery-powered applications require ultralow-power consumption and do not need very high
throughput from the ADC.
Design a single-supply, data acquisition circuit optimized for using the ADS8881 to achieve
• ENOB > 16 bits and
• Ultralow-power consumption of < 1 mW at throughput of 10 kSPS.
11.2.4.2 Detailed Design Procedure
The application circuit in Figure 72 shows the schematic of a complete reference driver circuit that generates a
voltage of 2.5 V dc using a single 3.3-V supply. This ultralow power reference block is suitable to drive the
ADS8881 for power-sensitive applications at a relatively lower throughput. This design uses the high-precision
REF3325 circuit that provides an accurate 2.5-V reference voltage at an extremely low quiescent current of 5 µA.
The output broadband noise of the reference is heavily filtered by a low-pass filter with a 3-dB cutoff frequency of
16 Hz.
The reference buffer is designed with the low-power OPA313 that can operate from a 3.3-V supply at an
extremely low quiescent current of 50 µA. The wideband noise contribution from the amplifier is limited by a
lowpass filter of a cutoff frequency equal to 1.5 kHz, formed by a 4.7-Ω resistor in combination with a 22-μF
capacitor. The 4.7-Ω series resistor creates an additional drop in the reference voltage that is corrected by a
dual-feedback configuration.
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The input driver circuit uses extremely low-power, dual amplifiers (such as the OPA2333) with a maximum
quiescent current of 28 µA per channel to drive the ADC inputs. The input amplifiers are configured in a modified
unity-gain buffer configuration. The filter capacitor at the ADC inputs attenuates the sampling charge-injection
noise from the ADC but effects the stability of the input amplifiers by degrading the phase margin. This
attenuation requires a series isolation resistor to maintain amplifier stability. The value of the series resistor is
directly proportional to the open-loop output impedance of the driving amplifier to maintain stability, which is high
(in the order of kΩ) in the case of low-power amplifiers such as the OPA333. Therefore, a high value of 1 kΩ is
selected for the series resistor at the ADC inputs. However, this series resistor creates an additional voltage drop
in the signal path, thereby leading to linearity and distortion issues. The dual-feedback configuration used in
Figure 72 corrects for this additional voltage drop and maintains system performance at ultralow-power
consumption.
11.2.4.3 Application Curve
10000
9000
Number of Hits
8000
7000
6000
5000
4000
3000
2000
1000
0
-23 -22 -21 -20 -19 -18 -17 -16 -15 -14 -13 -12 -11
Code
C004
Vdiff close to 0 V, 32768 data points, standard deviation = 1.7 bits,
ENOB (dc) = 16.3 bits
Figure 73. DC Input Histogram
For step-by-step design procedure, circuit schematics, bill of materials, PCB files, simulation results, and test
results, see 18-Bit, 10kSPS Data Acquisition (DAQ) Block Optimized for Ultra Low Power < 1mW (SLAU514).
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12 Power-Supply Recommendations
The device has two separate power supplies: AVDD and DVDD. The internal circuits of the device operate on
AVDD; DVDD is used for the digital interface. AVDD and DVDD can be independently set to any value within the
permissible range.
12.1 Power-Supply Decoupling
Decouple the AVDD and DVDD pins with GND, using individual 1-µF decoupling capacitors placed in close
proximity to the pin, as shown in Figure 74.
Digital
Supply
REF
Analog
Supply
DVDD
AVDD
1 µF
1 µF
DIN
AINP
SCLK
AINN
DOUT
GND
CONVST
Figure 74. Supply Decoupling
12.2 Power Saving
The device has an auto power-down feature that powers down the internal circuitry at the end of every
conversion. Referring to Figure 75, the input signal is acquired on the sampling capacitors when the device is in
a power-down state (tacq); at the same time, the result for the previous conversion is available for reading. The
device powers up on the start of the next conversion. During conversion phase (tconv), the device also consumes
current from the reference source (connected to the REF pin).
tTHROUGHPUT
Device Phase
tCONV
tACQ
œœ
IREF
tACQ
~50000X
~50000X
œœ
IAVDD
tCONV
2 x tTHROUGHPUT
~1200X
~1200X
~2X
IAVG(AVDD+REF)
Figure 75. Power Scaling With Throughput
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Power Saving (continued)
The conversion time, tconv, is independent of the SCLK frequency. When operating the device at speeds lower
than the maximum rated throughput, the conversion time, tconv, does not change; the device spends more time in
power-down state. Therefore, as shown in Figure 76, the device power consumption from the AVDD supply and
the external reference source is directly proportional to the speed of operation. Extremely low AVDD power-down
current (50 nA, typical) and extremely low external reference leakage current (250 nA, typical), make this device
ideal for very low throughput applications (such as pulsed measurements).
Power Consumption (mW)
6
5
4
3
2
1
0
0
200
400
600
Throughput (kSPS)
800
1000
C03
Figure 76. Power Scaling With Throughput
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13 Layout
13.1 Layout Guidelines
Figure 77 shows a board layout example for the device. Appropriate layout that interconnects accompanying
capacitors and converters with low inductance is critical for achieving optimum performance. Thus, a PCB board
with at least four layers is recommended to keep all critical components on the top layer and interconnected to a
solid (low inductance) analog ground plane at the subsequent inner layer using 15-mil vias. Avoid crossing digital
lines with the analog signal path and keep the analog input signals and the reference input signals away from
noise sources. As shown in Figure 77, the analog input and reference signals are routed on the left side of the
board and the digital connections are routed on the right side of the device.
As a result of dynamic currents during conversion and data transfer, each supply pin (AVDD and DVDD) must
have a decoupling capacitor to keep the supply voltage stable. To maximize decoupling capabilities, inductance
between each supply capacitor and the supply pin of the converter is kept less than 5 nH by placing the
capacitor within 0.2-inches from the pin and connecting it with 20-mil traces and a 15-mil grounding via, as
shown in Figure 77. TI recommends using one 1-μF ceramic capacitor at each supply pin. Avoid placing vias
between the supply pin and its decoupling capacitor.
Dynamic currents are also present at the REF pin during the conversion phase and very good decoupling is
critical to achieve optimum performance. The inductance between the reference capacitor and the REF pin is
kept less than 2 nH by placing the capacitor within 0.1-inches from the pin and connecting it with 20-mil traces
and multiple 15-mil grounding vias, as shown in Figure 77. A single, 10-μF, X7R-grade, 0805-size, ceramic
capacitor with at least a 10-V rating is recommended for good performance over the rated temperature range.
Avoid using additional lower value capacitors because the interactions between multiple capacitors may affect
the ADC performance at higher sampling rates. A small, 0.1-Ω to 0.47-Ω, 0603-size resistor placed in series with
the reference capacitor (as shown in Figure 77) keeps the overall impedance low and constant, especially at very
high frequencies.
The fly-wheel RC filters are placed immediately next to the input pins. Among ceramic surface-mount capacitors,
COG (NPO) ceramic capacitors provide the best capacitance precision. The type of dielectric used in COG
(NPO) ceramic capacitors provides the most stable electrical properties over voltage, frequency, and temperature
changes.
R
AVDD
13.2 Layout Example
EF
GND
DVDD
REF
1: REF
Positive Input
Differential
Inputs
Negative Input
DVDD
GND
1PF
1PF
0.1O t 0.47O
AVDD
10PF
GND
47O
10: DVDD
2: AVDD
9: SDI
47O
3: AINP
8: SCLK
4: AINN
7: SDO
5: GND
6: CONVST
GND
SDO
47O
Figure 77. Recommended Layout
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14 Device and Documentation Support
14.1 Device Support
14.1.1 Development Support
• Using SAR ADC TINA Models: Static behavior, power scaling
• Using SAR ADC TINA Models: Much ado about settling
• SAR ADC Response Times: Respond quickly, control quickly
• Input Considerations for SAR ADCs
14.2 Documentation Support
14.2.1 Related Documentation
For related documentation see the following:
• OPAx333 1.8-V, microPower, CMOS Operational Amplifiers, Zero-Drift Series Data Sheet (SBOS351)
• THS452x Very Low Power, Negative Rail Input, Rail-To-Rail Output, Fully Differential Amplifier Data Sheet
(SBOS458)
• THS4281 Very Low-Power, High-Speed, Rail-to-Rail Input and Output Voltage-Feedback Operational
Amplifier Data Sheet (SLOS432)
• 1-MHz, Micro-Power, Low-Noise, RRIO,1.8-V CMOS Operational Amplifier Precision Value Line Series Data
Sheet (SBOS649)
• OPAx350 High-Speed, Single-Supply, Rail-to-Rail Operational Amplifiers MicroAmplifier Series Data Sheet
(SBOS099)
• 1.8V, 7MHz, 90dB CMRR, Single-Supply, Rail-to-Rail I/O Operational Amplifier Data Sheet (SBOS259)
• 18-Bit Data Acquisition (DAQ) Block Optimized for 1-μs Full-Scale Step Response TI Precision Design
(TIDU012)
• 18-Bit, 1-MSPS Data Acquisition (DAQ) Block Optimized for Lowest Power TI Precision Design (SLAU513)
• 18 bit, 10kSPS Data Acquisition (DAQ) Block Optimized for Ultra Low Power < 1 mW TI Precision Design
(SLAU514)
• 18-Bit, 1MSPS Data Acquisition Block (DAQ) Optimized for Lowest Distortion and Noise TI Precision Design
(SLAU515)
• Ultra Low Power, 18 bit Precision ECG Data Acquisition System TI Precision Design (SLAU516)
• ADS8881 IBIS Model (SBAM172)
• ADS8881 TINA-TI Reference Design (SBAM173)
• ADS8881EVM-PDK User's Guide (SBAU211)
14.3 Receiving Notification of Documentation Updates
To receive notification of documentation updates, navigate to the device product folder on ti.com. In the upper
right corner, click on Alert me to register and receive a weekly digest of any product information that has
changed. For change details, review the revision history included in any revised document.
14.4 Community Resources
The following links connect to TI community resources. Linked contents are provided "AS IS" by the respective
contributors. They do not constitute TI specifications and do not necessarily reflect TI's views; see TI's Terms of
Use.
TI E2E™ Online Community TI's Engineer-to-Engineer (E2E) Community. Created to foster collaboration
among engineers. At e2e.ti.com, you can ask questions, share knowledge, explore ideas and help
solve problems with fellow engineers.
Design Support TI's Design Support Quickly find helpful E2E forums along with design support tools and
contact information for technical support.
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14.5 Trademarks
E2E is a trademark of Texas Instruments.
TINA is a trademark of Texas Instruments Inc..
SPI is a trademark of Motorola Inc.
All other trademarks are the property of their respective owners.
14.6 Electrostatic Discharge Caution
This integrated circuit can be damaged by ESD. Texas Instruments recommends that all integrated circuits be handled with
appropriate precautions. Failure to observe proper handling and installation procedures can cause damage.
ESD damage can range from subtle performance degradation to complete device failure. Precision integrated circuits may be more
susceptible to damage because very small parametric changes could cause the device not to meet its published specifications.
14.7 Glossary
SLYZ022 — TI Glossary.
This glossary lists and explains terms, acronyms, and definitions.
15 Mechanical, Packaging, and Orderable Information
The following pages include mechanical, packaging, and orderable information. This information is the most
current data available for the designated devices. This data is subject to change without notice and revision of
this document. For browser-based versions of this data sheet, refer to the left-hand navigation.
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Copyright © 2013–2015, Texas Instruments Incorporated
Product Folder Links: ADS8881
49
PACKAGE OPTION ADDENDUM
www.ti.com
10-Dec-2020
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
(4/5)
(6)
ADS8881CDGS
ACTIVE
VSSOP
DGS
10
80
RoHS & Green
NIPDAU
Level-2-260C-1 YEAR
0 to 70
8881C
ADS8881CDGSR
ACTIVE
VSSOP
DGS
10
2500
RoHS & Green
NIPDAU
Level-2-260C-1 YEAR
0 to 70
8881C
ADS8881CDRCR
ACTIVE
VSON
DRC
10
3000
RoHS & Green
NIPDAU
Level-2-260C-1 YEAR
0 to 70
8881C
ADS8881CDRCT
ACTIVE
VSON
DRC
10
250
RoHS & Green
NIPDAU
Level-2-260C-1 YEAR
0 to 70
8881C
ADS8881IDGS
ACTIVE
VSSOP
DGS
10
80
RoHS & Green
NIPDAU
Level-2-260C-1 YEAR
-40 to 85
8881
ADS8881IDGSR
ACTIVE
VSSOP
DGS
10
2500
RoHS & Green
NIPDAU
Level-2-260C-1 YEAR
-40 to 85
8881
ADS8881IDRCR
ACTIVE
VSON
DRC
10
3000
RoHS & Green
NIPDAU
Level-2-260C-1 YEAR
-40 to 85
8881
ADS8881IDRCT
ACTIVE
VSON
DRC
10
250
RoHS & Green
NIPDAU
Level-2-260C-1 YEAR
-40 to 85
8881
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of
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