KIT
ATION
EVALU
E
L
B
A
IL
AVA
19-2373; Rev 0; 4/02
Low-Cost, High-Frequency, Current-Mode PWM
Buck Controller
The MAX1953/MAX1954/MAX1957 is a family of versatile, economical, synchronous current-mode, pulse-width
modulation (PWM) buck controllers. These step-down
controllers are targeted for applications where cost and
size are critical.
The MAX1953 operates at a fixed 1MHz switching frequency, thus significantly reducing external component
size and cost. Additionally, excellent transient response
is obtained using less output capacitance. The MAX1953
operates from low 3V to 5.5V input voltage and can supply up to 10A of output current. Selectable current limit is
provided to tailor to the external MOSFETs’ on-resistance
for optimum cost and performance. The output voltage is
adjustable from 0.8V to 0.86VIN.
With the MAX1954, the drain-voltage range on the highside FET is 3V to 13.2V and is independent of the supply
voltage. It operates at a fixed 300kHz switching frequency and can be used to provide up to 25A of output current with high efficiency. The output voltage is adjustable
from 0.8V to 0.86VHSD.
The MAX1957 features a tracking output voltage range of
0.4V to 0.86VIN and is capable of sourcing or sinking
current for applications such as DDR bus termination
and PowerPC™/ASIC/DSP core supplies. The MAX1957
operates from a 3V to 5.5V input voltage and at a fixed
300kHz switching frequency.
The MAX1953/MAX1954/MAX1957 provide a COMP pin
that can be pulled low to shut down the converter in
addition to providing compensation to the error amplifier.
An input undervoltage lockout (ULVO) is provided to
ensure proper operation under power-sag conditions to
prevent the external power MOSFETs from overheating.
Internal digital soft-start is included to reduce inrush current. The MAX1953/MAX1954/MAX1957 are available in
tiny 10-pin µMAX packages.
Applications
Features
♦ Low-Cost Current-Mode Controllers
♦ Fixed-Frequency PWM
♦ MAX1953
1MHz Switching Frequency
Small Component Size, Low Cost
Adjustable Current Limit
♦ MAX1954
3V to 13.2V Input Voltage
25A Output Current Capability
93% Efficiency
300kHz Switching Frequency
♦ MAX1957
Tracking 0.4V to 0.86VIN Output Voltage Range
Sinking and Sourcing Capability of 3A
♦ Shutdown Feature
♦ All N-Channel MOSFET Design for Low Cost
♦ No Current-Sense Resistor Needed
♦ Internal Digital Soft-Start
♦ Thermal Overload Protection
♦ Small 10-Pin µMAX Package
Ordering Information
PART
TEMP RANGE
PIN-PACKAGE
MAX1953EUB
-40°C to +85°C
10 µMAX
MAX1954EUB
-40°C to +85°C
10 µMAX
MAX1957EUB
-40°C to +85°C
10 µMAX
Pin Configurations
TOP VIEW
Printers and Scanners
Graphic Cards and Video Cards
PCs and Servers
Microprocessor Core Supply
Low-Voltage Distributed Power
ILIM 1
10 BST
COMP
2
FB
3
GND
4
7
PGND
IN
5
6
DL
MAX1953EUB
9
LX
8
DH
Telecommunications and Networking
µMAX
†Pg
PowerPC is a trademark of Motorola, Inc.
Pin Configurations continued at end of data sheet.
________________________________________________________________ Maxim Integrated Products
For pricing, delivery, and ordering information, please contact Maxim/Dallas Direct! at
1-888-629-4642, or visit Maxim’s website at www.maxim-ic.com.
1
MAX1953/MAX1954/MAX1957
General Description
MAX1953/MAX1954/MAX1957
Low-Cost, High-Frequency, Current-Mode PWM
Buck Controller
ABSOLUTE MAXIMUM RATINGS
IN, FB to GND...........................................................-0.3V to +6V
LX to BST..................................................................-6V to +0.3V
BST to GND ............................................................-0.3V to +20V
DH to LX ....................................................-0.3V to (VBST + 0.3V)
DL, COMP to GND.......................................-0.3V to (VIN + 0.3V)
HSD, ILIM, REFIN to GND ........................................-0.3V to 14V
PGND to GND .......................................................-0.3V to +0.3V
IDH, IDL ................................................................±100mA (RMS)
Continuous Power Dissipation (TA = +70°C)
(derate 5.6mW/°C above +70°C) ..................................444mW
Operating Temperature Range ...........................-40°C to +85°C
Junction Temperature ......................................................+150°C
Storage Temperature Range .............................-65°C to +150°C
Lead Temperature (soldering, 10s) .................................+300°C
Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional
operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to
absolute maximum rating conditions for extended periods may affect device reliability.
ELECTRICAL CHARACTERISTICS
(VIN = 5V, VBST - VLX = 5V, TA = -40°C to +85°C, unless otherwise noted. Typical values are at TA = +25°C.) (Note 1)
PARAMETER
CONDITIONS
Operating Input Voltage Range
HSD Voltage Range
MAX1954 only (Note 2)
Quiescent Supply Current
VFB = 1.5V, no switching
MIN
TYP
MAX
UNITS
3.0
5.5
V
3.0
13.2
V
1
2
mA
Standby Supply Current (MAX1953/ MAX1957) VIN = VBST = 5.5V, COMP = GND
220
350
µA
Standby Supply Current (MAX1954)
VIN = VBST = 5.5V, VHSD = 13.2V,
COMP = GND
220
350
µA
Undervoltage Lockout Trip Level
Rising and falling VIN, 3% hysteresis
2.78
2.95
V
0.86 x
VIN
V
Output Voltage Adjust Range (VOUT)
2.50
0.8
ERROR AMPLIFIER
FB Regulation Voltage
TA = 0°C to +85°C (MAX1953/MAX1954)
0.788
0.8
0.812
TA = -40°C to +85°C (MAX1953/MAX1954)
0.776
0.8
0.812
VREFIN
- 8mV
VREFIN
VREFIN
+ 8mV
70
110
160
µS
MAX1957 only
Transconductance
FB Input Leakage Current
VFB = 0.9V
5
500
nA
REFIN Input Bias Current
VREFIN = 0.8V, MAX1957 only
5
500
nA
1.5
V
FB Input Common-Mode Range
-0.1
REFIN Input Common-Mode Range
MAX1957 only
-0.1
1.5
V
Current-Sense Amplifier Voltage Gain Low
ILIM = GND (MAX1953 only)
5.67
6.3
6.93
V/V
3.15
3.5
3.85
V/V
Current-Sense Amplifier Voltage Gain
2
V
VILIM = VIN or ILIM = open (MAX1953 only)
MAX1954/MAX1957
_______________________________________________________________________________________
Low-Cost, High-Frequency, Current-Mode PWM
Buck Controller
(VIN = 5V, VBST - VLX = 5V, TA = -40°C to +85°C, unless otherwise noted. Typical values are at TA = +25°C.) (Note 1)
PARAMETER
ILIM Input Impedance
Current-Limit Threshold
CONDITIONS
MIN
TYP
MAX
UNITS
MAX1953 only
50
125
200
kΩ
VPGND - VLX, ILIM = GND (MAX1953 only)
85
105
125
VPGND - VLX, ILIM = open (MAX1953 only)
190
210
235
VPGND - VLX, ILIM = IN (MAX1953 only)
290
320
350
VPGND – VLX (MAX1954/MAX1957 only)
190
210
235
mV
OSCILLATOR
Switching Frequency
Maximum Duty Cycle
Minimum Duty Cycle
MAX1953
0.8
1
1.2
MHz
MAX1954/MAX1957
240
300
360
kHz
Measured at DH
86
89
96
%
MAX1953, measured at DH
15
18
MAX1954/MAX1957, measured at DH
4.5
5.5
%
SOFT-START
MAX1953
Soft-Start Period
4
MAX1954/MAX1957
ms
3.4
FET DRIVERS
DH On-Resistance, High State
2
3
Ω
DH On-Resistance, Low State
1.5
3
Ω
DL On-Resistance, High State
2
3
Ω
0.8
2
Ω
LX, BST Leakage Current
VBST = 10.5V, VLX = VIN = 5.5V,
MAX1953/MAX1957
20
µA
LX, BST, HSD Leakage Current
VBST = 18.7V, VLX = 13.2V, VIN = 5.5V
VHSD = 13.2V (MAX1954 only)
30
µA
DL On-Resistance, Low State
THERMAL PROTECTION
Thermal Shutdown
Rising temperature
Thermal Shutdown Hysteresis
160
°C
15
°C
SHUTDOWN CONTROL
COMP Logic Level Low
3V < VIN < 5.5V
COMP Logic Level High
3V < VIN < 5.5V
COMP Pullup Current
0.25
V
100
µA
0.8
V
Note 1: Specifications to -40°C are guaranteed by design and not production tested.
Note 2: HSD and IN are externally connected for applications where VHSD < 5.5V.
_______________________________________________________________________________________
3
MAX1953/MAX1954/MAX1957
ELECTRICAL CHARACTERISTICS (continued)
Typical Operating Characteristics
(TA = +25°C, unless otherwise noted.)
MAX1954
EFFICIENCY vs. LOAD CURRENT
VOUT = 2.5V
90
80
EFFICIENCY (%)
85
VIN = 5V
75
70
65
60
80
VOUT = 1.7V
70
60
50
VOUT = 2.5V
CIRCUIT OF FIGURE 1
55
50
0.1
0.1
1
60
VIN = 5V
CIRCUIT OF FIGURE 3
40
10
0.1
1
LOAD CURRENT (A)
MAX1953
OUTPUT VOLTAGE vs. LOAD CURRENT
MAX1954
EFFICIENCY vs. LOAD CURRENT
VOUT = 1.8V
OUTPUT VOLTAGE (V)
90
85
80
75
70
65
MAX1953 toc05
2.60
MAX1953 toc04
100
EFFICIENCY (%)
VOUT = 1.25V
70
LOAD CURRENT (A)
LOAD CURRENT (A)
95
80
50
VIN = 5V
CIRCUIT OF FIGURE 2
40
10
1
90
EFFICIENCY (%)
90
100
MAX1953 toc02
VIN = 3.3V
95
100
MAX1953 toc01
100
MAX1957
EFFICIENCY vs. LOAD CURRENT
MAX1953 toc03
MAX1953
EFFICIENCY vs. LOAD CURRENT
EFFICIENCY (%)
2.55
VIN = 5V
2.50
VIN = 3.3V
2.45
60
VIN = 12V
CIRCUIT OF FIGURE 4
55
50
0
5
10
15
CIRCUIT OF FIGURE 1
2.40
0
25
20
0.5
1.0
1.5
2.5
3.0
LOAD CURRENT (A)
MAX1954
OUTPUT VOLTAGE vs. LOAD CURRENT
MAX1954
OUTPUT VOLTAGE vs. LOAD CURRENT
1.75
OUTPUT VOLTAGE (V)
2.50
VHSD = VIN = 5V
2.45
MAX1953 toc07
1.80
MAX1953 toc06
2.55
1.70
VHSD = VIN = 5V
1.65
1.60
2.40
CIRCUIT OF FIGURE 2
CIRCUIT OF FIGURE 2
1.55
2.35
0
1
2
3
4
LOAD CURRENT (A)
4
2.0
LOAD CURRENT (A)
2.60
OUTPUT VOLTAGE (V)
MAX1953/MAX1954/MAX1957
Low-Cost, High-Frequency, Current-Mode PWM
Buck Controller
5
6
0
1
2
3
4
5
LOAD CURRENT (A)
_______________________________________________________________________________________
6
10
Low-Cost, High-Frequency, Current-Mode PWM
Buck Controller
1.25
VIN = 5V
MAX1953 toc10
1.74
2.55
OUTPUT VOLTAGE (V)
OUTPUT VOLTAGE (V)
1.30
1.76
MAX1953 toc09
2.60
MAX1953 toc08
1.35
ILOAD = 3A
2.50
ILOAD = 0
1.72
ILOAD = 0
1.70
ILOAD = 5A
1.68
2.45
1.20
1.66
-1
0
1
1.64
2.40
3.0
3
2
3.5
4.0
4.5
3.5
MAX1954
OUTPUT VOLTAGE vs. INPUT VOLTAGE
2.51
4.5
5.0
5.5
MAX1957
OUTPUT VOLTAGE vs. INPUT VOLTAGE
1.29
MAX1953 toc11
2.52
4.0
INPUT VOLTAGE (V)
INPUT VOLTAGE (V)
LOAD CURRENT (A)
1.27
OUTPUT VOLTAGE (V)
ILOAD = 0
2.50
2.49
3.0
5.5
5.0
ILOAD = 5A
2.48
MAX1953 toc12
-2
OUTPUT VOLTAGE (V)
ILOAD = 0
1.25
1.23
ILOAD = 3A
1.21
2.47
CIRCUIT OF FIGURE 3
CIRCUIT OF FIGURE 2
2.46
1.19
3.5
4.0
4.5
5.0
3.0
3.5
4.0
4.5
5.0
INPUT VOLTAGE (V)
MAX1953
FREQUENCY vs. INPUT VOLTAGE
MAX1954/MAX1957
FREQUENCY vs. INPUT VOLTAGE
VOUT = 2.5V
1.06
5.5
INPUT VOLTAGE (V)
320
310
FREQUENCY (kHz)
1.04
TA = -40°C
1.02
1.00
VOUT = 1.25V
315
5.5
MAX1953 toc14
3.0
MAX1953 toc13
-3
CIRCUIT OF FIGURE 2
CIRCUIT OF FIGURE 1
CIRCUIT OF FIGURE 3
1.15
FREQUENCY (MHz)
OUTPUT VOLTAGE (V)
MAX1954
OUTPUT VOLTAGE vs. INPUT VOLTAGE
MAX1953
OUTPUT VOLTAGE vs. INPUT VOLTAGE
MAX1957
OUTPUT VOLTAGE vs. LOAD CURRENT
TA = -40°C
305
300
TA = +25°C
295
290
285
TA = +85°C
0.98
TA = +85°C
280
TA = +25°C
275
0.96
270
3.0
3.5
4.0
4.5
INPUT VOLTAGE (V)
5.0
5.5
3.0
3.5
4.0
4.5
5.0
5.5
INPUT VOLTAGE (V)
_______________________________________________________________________________________
5
MAX1953/MAX1954/MAX1957
Typical Operating Characteristics (continued)
(TA = +25°C, unless otherwise noted.)
MAX1953/MAX1954/MAX1957
Low-Cost, High-Frequency, Current-Mode PWM
Buck Controller
Typical Operating Characteristics (continued)
(TA = +25°C, unless otherwise noted.)
MAX1954
LOAD TRANSIENT
MAX1953
LOAD TRANSIENT
MAX1953 toc16
MAX1953 toc15
VOUT
AC-COUPLED
100mV/div
VOUT
AC-COUPLED
100mV/div
3A
ILOAD
1.5A
5A
2.5A
ILOAD
CIRCUIT OF FIGURE 1
400µs/div
400µs/div
MAX1953
NO-LOAD SWITCHING WAVEFORMS
MAX1957
LOAD TRANSIENT
MAX1953 toc18
MAX1953 toc17
VOUT
AC-COUPLED
50mV/div
3A
ILOAD
ILX
2A/div
LX
5V/div
DL
5V/div
DH
5V/div
-3A
400µs/div
6
2µs/div
_______________________________________________________________________________________
Low-Cost, High-Frequency, Current-Mode PWM
Buck Controller
MAX1953
SHORT-CIRCUIT SWITCHING WAVEFORMS
MAX1953
FULL-LOAD SWITCHING WAVEFORMS
MAX1954/MAX1957
NO-LOAD SWITCHING WAVEFORMS
MAX1953 toc20
MAX1953 toc19
ILX
2A/div
LX
5V/div
DL
5V/div
DH
5V/div
MAX1953 toc21
ILX
5A/div
LX
5V/div
DL
5V/div
ILX
2A/div
LX
10V/div
DL
5V/div
DH
10V/div
5V/div
DH
2µs/div
2µs/div
4µs/div
MAX1954/MAX1957
SHORT-CIRCUIT SWITCHING WAVEFORMS
MAX1954/MAX1957
FULL-LOAD SWITCHING WAVEFORMS
MAX1953 toc22
ILX
MAX1953 toc23
2A/div
ILX
5A/div
LX
10V/div
LX
10V/div
DL
5V/div
DL
5V/div
DH
10V/div
DH
10V/div
4µs/div
4µs/div
_______________________________________________________________________________________
7
MAX1953/MAX1954/MAX1957
Typical Operating Characteristics (continued)
(TA = +25°C, unless otherwise noted.)
Low-Cost, High-Frequency, Current-Mode PWM
Buck Controller
MAX1953/MAX1954/MAX1957
Pin Description
PIN
MAX1953
8
MAX1954
MAX1957
NAME
FUNCTION
1
—
—
ILIM
ILIM Sets the Current-Limit Threshold for the Low-Side N-Channel
MOSFET, as well as the Current-Sense Amplifier Gain. Connect to IN
for 320mV, leave floating for 210mV, or connect to GND for 105mV
current-limit threshold.
—
1
—
HSD
HSD Senses the Voltage at the Drain of the High-Side N-Channel
MOSFET. Connect to the high-side MOSFET drain using a Kelvin
connection.
—
—
1
REFIN
REFIN Sets the FB Regulation Voltage. Drive REFIN with the desired
FB regulation voltage using an external resistor-divider. Bypass to
GND with a 0.1µF capacitor.
2
2
2
COMP
Compensation and Shutdown Control Pin. Connect an RC network to
compensate control loop. Drive to GND to shut down the IC.
3
3
3
FB
4
4
4
GND
Feedback Input. Regulates at VFB = 0.8V (MAX1953/MAX1954) or
REFIN (MAX1957). Connect FB to a resistor-divider to set the output
voltage (MAX1953/MAX1954). Connect to output through a decoupling
resistor (MAX1957).
Ground
5
5
5
IN
Input Voltage (3V to 5.5V). Provides power for the IC. For the
MAX1953/MAX1957, IN serves as the current-sense input for the highside MOSFET. Connect to the drain of the high-side MOSFET
(MAX1953/MAX1957). Bypass IN to GND close to the IC with a
0.22µF (MAX1954) capacitor. Bypass IN to GND close to the IC with
10µF and 4.7µF in parallel (MAX1953/MAX1957) capacitors. Use
ceramic capacitors.
6
6
6
DL
Low-Side Gate-Drive Output. Drives the synchronous-rectifier MOSFET.
Swings from PGND to VIN.
7
7
7
PGND
8
8
8
DH
High-Side Gate-Drive Output. Drives the high-side MOSFET. DH is a
floating driver output that swings from VLX to VBST.
9
9
9
LX
Master Controller Current-Sense Input. Connect LX to the junction of
the MOSFETs and inductor. LX is the reference point for the current
limit.
10
10
10
BST
Boost Capacitor Connection for High-Side Gate Driver. Connect a
0.1µF ceramic capacitor from BST to LX and a Schottky diode to IN.
Power Ground. Connect to source of the synchronous rectifier close to
the IC.
_______________________________________________________________________________________
Low-Cost, High-Frequency, Current-Mode PWM
Buck Controller
IN
THERMAL
LIMIT
MAX1953
MAX1954
MAX1957
UVLO
HSD
(MAX1954
ONLY)
SLOPE
COMPENSATION
SHUTDOWN
COMPARATOR
0.5V
BST
COMP
ERROR
AMPLIFIER
FB
DH
PWM
CONTROL
CIRCUITRY
CURRENTSENSE
CIRCUITRY
GND
LX
IN
DL
REFIN
(MAX1957
ONLY)
REFERENCE
AND
SOFT-START
DAC
PGND
CLOCK
SHORT-CIRCUIT
CURRENT-LIMIT
CIRCUITRY
CURRENT-LIMIT
COMPARATOR
ILIM
(MAX1953
ONLY)
Typical Operating Circuit
INPUT
3V TO 5.5V
IN
ILIM
BST
DH
MAX1953
LX
OUTPUT
0.8V TO 0.86VIN
COMP
DL
PGND
GND
FB
_______________________________________________________________________________________
9
MAX1953/MAX1954/MAX1957
Functional Diagram
MAX1953/MAX1954/MAX1957
Low-Cost, High-Frequency, Current-Mode PWM
Buck Controller
Detailed Description
The MAX1953/MAX1954/MAX1957 are single-output,
fixed-frequency, current-mode, step-down, PWM, DCDC converter controllers. The MAX1953 switches at
1MHz, allowing the use of small external components for
small applications. Table 1 lists suggested components.
The MAX1954 switches at 300kHz for higher efficiency
and operates from a wider range of input voltages.
Figure 1 is the MAX1953 typical application circuit. The
MAX1953/MAX1954/MAX1957 are designed to drive a
pair of external N-channel power MOSFETs in a synchronous buck topology to improve efficiency and cost
compared with a P-channel power MOSFET topology.
The on-resistance of the low-side MOSFET is used for
short-circuit current-limit sensing, while the high-side
MOSFET on-resistance is used for current-mode feedback and current-limit sensing, thus eliminating the
need for current-sense resistors. The MAX1953 has
three selectable short-circuit current-limit thresholds:
105mV, 210mV, and 320mV. The MAX1954 and
MAX1957 have 210mV fixed short-circuit current-limit
thresholds. The MAX1953/MAX1954/MAX1957 accept
input voltages from 3V to 5.5V. The MAX1954 is configured with a high-side drain input (HSD) allowing an
extended input voltage range of 3V to 13.2V that is
independent of the input supply (Figure 2). The
MAX1957 is tailored for tracking output voltage applications such as DDR bus termination supplies, referred to
as VTT. It utilizes a resistor-divider network connected
to REFIN to keep the 1/2 ratio tracking between VTT
and VDDQ (Figure 3). The MAX1957 can source and
sink up to 3A. Figure 4 shows the MAX1954 20A circuit.
DC-DC Converter Control Architecture
The MAX1953/MAX1954/MAX1957 step-down converters use a PWM, current-mode control scheme. An internal transconductance amplifier establishes an integrated
error voltage. The heart of the PWM controller is an openloop comparator that compares the integrated voltagefeedback signal against the amplified current-sense
signal plus the slope compensation ramp, which are
summed into the main PWM comparator to preserve
inner-loop stability and eliminate inductor staircasing. At
each rising edge of the internal clock, the high-side
MOSFET turns on until the PWM comparator trips or the
maximum duty cycle is reached. During this on-time, current ramps up through the inductor, storing energy in a
magnetic field and sourcing current to the output. The
current-mode feedback system regulates the peak
inductor current as a function of the output voltage error
signal. The circuit acts as a switch-mode transconductance amplifier and pushes the output LC filter pole normally found in a voltage-mode PWM to a higher
frequency.
During the second half of the cycle, the high-side MOSFET turns off and the low-side MOSFET turns on. The
inductor releases the stored energy as the current ramps
down, providing current to the output. The output capacitor stores charge when the inductor current exceeds the
required load current and discharges when the inductor
current is lower, smoothing the voltage across the load.
Under overload conditions, when the inductor current
exceeds the selected current-limit (see the Current Limit
Circuit section), the high-side MOSFET is not turned on
at the rising clock edge and the low-side MOSFET
remains on to let the inductor current ramp down.
The MAX1953/MAX1954/MAX1957 operate in a forcedPWM mode. As a result, the controller maintains a constant switching frequency, regardless of load, to allow for
easier postfiltering of the switching noise.
Table 1. Suggested Components
DESIGNATION
10
MAX1953
MAX1954
MAX1957
20A CIRCUIT
C1
10µF, 6.3V X5R CER
Taiyo Yuden
JMK212BJ106MG
0.22µF, 10V X7R CER
Kemet
C0603C224M8RAC
3 x 22µF, 6.3V X5R CER
Taiyo Yuden
JMK316BJ226ML
0.22µF, 10V X7R CER
Kemet
C0603C224M8RAC
C2
0.1µF, 50V X7R CER
Taiyo Yuden
UMK107BJ104KA
10µF, 6.3V X5R CER
Taiyo Yuden
JMK212BJ106MG
0.1µF, 50V X7R CER
Taiyo Yuden
UMK107BJ104KA
10µF, 6.3V X5R CER
Taiyo Yuden
JMK212BJ106MG
C3
10µF, 6.3V X5R CER
Taiyo Yuden
JMK212BJ106MG
0.1µF, 50V X7R CER
Taiyo Yuden
UMK107BJ104KA
270µF, 2V SP Polymer
Panasonic
EEFUEOD271R
10µF, 6.3V X5R CER
Taiyo Yuden
JMK212BJ106MG
______________________________________________________________________________________
Low-Cost, High-Frequency, Current-Mode PWM
Buck Controller
MAX1953/MAX1954/MAX1957
Table 1. Suggested Components (continued)
DESIGNATION
MAX1953
C4
10µF, 6.3V X5R CER
Taiyo Yuden
JMK212BJ106MG
C5
4.7µF, 6.3V X5R CER
Taiyo Yuden
JMK212BJ475MG
C6
10µF, 6.3V X5R CER
Taiyo Yuden
JMK212BJ106MG
MAX1954
180µF, 4V SP Polymer
Panasonic
EEFUEOG181R
MAX1957
20A CIRCUIT
270µF, 2V SP Polymer
Panasonic
EEFUEOD271R
10µF, 6.3V X5R CER
Taiyo Yuden
JMK212BJ106MG
—
270µF, 2V SP Polymer
Panasonic
EEFUEOD271R
10µF, 6.3V X5R CER
Taiyo Yuden
JMK212BJ106MG
—
10µF, 6.3V X5R CER
Taiyo Yuden
JMK212BJ106MG
10µF, 6.3V X5R CER
Taiyo Yuden
JMK212BJ106MG
C7
—
—
4.7µF, 6.3V X5R CER
Taiyo Yuden
JMK212BJ475MG
0.1µF, 50V X7R CER
Taiyo Yuden
UMK107BJ104KA
C8
—
—
0.1µF, 50V X7R CER
Taiyo Yuden
UMK107BJ104KA
270µF, 2V SP polymer
Panasonic
EEFUEOD271R
C9-C13
—
—
C14
—
—
CC
270pF, 10V X7R CER
Kemet
C0402C271M8RAC
Cf
—
—
270µF, 2V SP polymer
Panasonic
EEFUEOD271R
1500pF, 50V X7R CER
Murata
GRM39X7R152K50
—
1000pF, 10V X7R CER
Kemet
C0402C102M8RAC
470pF, 50V X7R CER
Murata
GRM39X7R471K50
560pF, 10V X7R CER
Kemet
C0402C561M8RAC
47pF, 10V C0G CER
Kemet
C0402C470K8GAC
68pF, 50V COG CER
Murata
GRM39COG680J50
15pF, 10V C0G CER
Kemet
C0402C150K8GAC
D1
Schottky diode
Central Semiconductor
CMPSH1-4
Schottky diode
Central Semiconductor
CMPSH1-4
Schottky diode
Central Semiconductor
CMPSH1-4
Schottky diode
Central Semiconductor
CMPSH1-4
L1
1µH 3.6A
Toko 817FY-1R0M
2.7µH 6.6A
Coilcraft
DO3316-272HC
2.7µH 6.6A
Coilcraft
DO3316-272HC
0.8µH 27.5A
Sumida
CEP125U-0R8
Dual MOSFET 20V 5A
Fairchild
FDS6898A
Dual MOSFET 20V
Fairchild
FDS6890A
Dual MOSFET 20V
Fairchild
FDS6898A
N-channel 30V
International Rectifier
IRF7811W
N1-N2
N3-N4
—
—
R1
16.9kΩ 1%
9.09kΩ 1%
R2
8.06kΩ 1%
8.06kΩ 1%
R3
RC
—
N-channel 30V
Siliconix Si4842DY
2kΩ 1%
10kΩ 1%
2kΩ 1%
8.06kΩ 1%
10kΩ 5%
33kΩ 5%
62kΩ 5%
51.1kΩ 5%
270kΩ 5%
______________________________________________________________________________________
11
MAX1953/MAX1954/MAX1957
Low-Cost, High-Frequency, Current-Mode PWM
Buck Controller
VIN
3V TO 5.5V
C6
10µF
C1
10µF
C5
4.7µF
IN
ILIM
D1
N1
BST
DH
RC
33kΩ
MAX1953
L1
1µH
C2
0.1µF
LX
COMP
CC
270pF
VOUT
2.5V AT 3A
R1
16.9kΩ
DL
C3
10µF
C4
10µF
PGND
GND
R2
8.06Ω
FB
Figure 1. Typical Application Circuit for the MAX1953
VIN
3V TO 5.5V
C2
10µF
C1
0.22µF
VHSD
5.5V TO 13.2V
D1
IN
HSD
RC
62kΩ
BST
DH
MAX1954
DL
Cf
47pF
C3
0.1µF
L1
2.7µH
VOUT
1.7V AT 3A
LX
COMP
CC
1000pF
N1
R1
9.09kΩ
C4
180µF
PGND
GND
FB
R2
8.06kΩ
Figure 2. Typical Application Circuit for the MAX1954
12
______________________________________________________________________________________
Low-Cost, High-Frequency, Current-Mode PWM
Buck Controller
MAX1953/MAX1954/MAX1957
VIN
3V TO 5.5V
C6
10µF
C1
3 ✕ 22µF
C7
4.7µF
RC
51.1kΩ
COMP
VDDQ
IN
N1
BST
Cf
68pF
CC
470pF
D1
DH
R1
2kΩ
MAX1957
L1
2.7µH
C2
0.1µF
LX
C14
1500pF R3
10kΩ
REFIN
DL
C8
0.1µF
R2
2kΩ
VTT = 1/2 VDDQ
C3
270µF
C4
270µF
C5
270µF
PGND
GND
FB
Figure 3. Typical Application Circuit for the MAX1957
VHSD
10.8V TO 13.2V
VIN
3V TO 5.5V
C2
10µF
C3
10µF
C4
10µF
C5
10µF
C6
10µF
D1
C1
0.22µF
HSD
IN
RC
270kΩ
BST
DH
MAX1954
LX
COMP
CC
560pF
DL
Cf
15pF
N1
N2
L1
0.8µH
C7
0.1µF
N3
VOUT
1.8V AT 20A
N4
R1
10kΩ
C8
270µF
C9
270µF
C10
270µF
C11
270µF
C12
270µF
C13
270µF
PGND
GND
FB
R2
8.06kΩ
Figure 4. 20A Circuit
______________________________________________________________________________________
13
MAX1953/MAX1954/MAX1957
Low-Cost, High-Frequency, Current-Mode PWM
Buck Controller
Current-Sense Amplifier
Synchronous Rectifier Driver (DL)
The MAX1953/MAX1954/MAX1957s’ current-sense circuit amplifies (AV = 3.5 typ) the current-sense voltage
(the high-side MOSFET’s on-resistance (RDS(ON)) multiplied by the inductor current). This amplified currentsense signal and the internal-slope compensation
signal are summed (VSUM) together and fed into the
PWM comparator’s inverting input. The PWM comparator shuts off the high-side MOSFET when V SUM
exceeds the integrated feedback voltage (VCOMP).
Synchronous rectification reduces conduction losses in
the rectifier by replacing the normal Schottky catch
diode with a low-resistance MOSFET switch. The
MAX1953/MAX1954/MAX1957 use the synchronous
rectifier to ensure proper startup of the boost gatedriver circuit and to provide the current-limit signal. The
DL low-side waveform is always the complement of the
DH high-side drive waveform. A dead-time circuit monitors the DL output and prevents the high-side MOSFET
from turning on until DL is fully off, thus preventing
cross-conduction or shoot-through. In order for the
dead-time circuit to work properly, there must be a lowresistance, low-inductance path from the DL driver to
the MOSFET gate. Otherwise, the sense circuitry in the
MAX1953/MAX1954/MAX1957 can interpret the MOSFET gate as OFF when gate charge actually remains.
The dead time at the other edge (DH turning off) is
determined through gate sensing as well.
Current-Limit Circuit
The current-limit circuit employs a lossless current-limiting algorithm that uses the low-side and high-side
MOSFETs’ on-resistances as the sensing elements. The
voltage across the high-side MOSFET is monitored for
current-mode feedback, as well as current limit. This
signal is amplified by the current-sense amplifier and is
compared with a current-sense voltage. If the currentsense signal is larger than the set current-limit voltage,
the high-side MOSFET turns off. Once the high-side
MOSFET turns off, the low-side MOSFET is monitored
for current limit. If the voltage across the low-side MOSFET (RDS(ON) ✕ IINDUCTOR) does not exceed the shortcircuit current limit, the high-side MOSFET turns on
normally. In this condition, the output drops smoothly
out of regulation. If the voltage across the low-side
MOSFET exceeds the short-circuit current-limit threshold at the beginning of each new oscillator cycle, the
MAX1953/MAX1954/MAX1957 do not turn on the highside MOSFET.
In the case where the output is shorted, the low-side
MOSFET is monitored for current limit. The low-side
MOSFET is held on to let the current in the inductor
ramp down. Once the voltage across the low-side
MOSFET drops below the short-circuit current-limit
threshold, the high-side MOSFET is pulsed. Under this
condition, the frequency of the MAX1953/MAX1954/
MAX1957 appears to decrease because the on-time of
the low-side MOSFET extends beyond a clock cycle.
The actual peak output current is greater than the
short-circuit current-limit threshold by an amount equal
to the inductor ripple current. Therefore, the exact current-limit characteristic and maximum load capability
are a function of the low-side MOSFET on-resistance,
inductor value, input voltage, and output voltage.
The short-circuit current-limit threshold is preset for the
MAX1954/MAX1957 at 210mV. The MAX1953, however,
has three options for the current-limit threshold: connect ILIM to IN for a 320mV threshold, connect ILIM to
GND for 105mV, or leave floating for 210mV.
14
High-Side Gate-Drive Supply (BST)
Gate-drive voltage for the high-side switch is generated
by a flying capacitor boost circuit (Figure 5). The
capacitor between BST and LX is charged from the VIN
supply up to VIN, minus the diode drop while the lowside MOSFET is on. When the low-side MOSFET is
switched off, the stored voltage of the capacitor is
stacked above LX to provide the necessary turn-on
voltage (VGS) for the high-side MOSFET. The controller
then closes an internal switch between BST and DH to
turn the high-side MOSFET on.
Undervoltage Lockout
If the supply voltage at IN drops below 2.75V, the
MAX1953/MAX1954/MAX1957 assume that the supply
voltage is too low to make valid decisions, so the UVLO
circuitry inhibits switching and forces the DL and DH
gate drivers low. After the voltage at IN rises above
2.8V, the controller goes into the startup sequence and
resumes normal operation.
Startup
The MAX1953/MAX1954/MAX1957 start switching when
the voltage at IN rises above the UVLO threshold.
However, the controller is not enabled unless all four of
the following conditions are met:
• VIN exceeds the 2.8V UVLO threshold.
• The internal reference voltage exceeds 92% of its
nominal value (VREF > 1 V).
• The internal bias circuitry powers up.
• The thermal overload limit is not exceeded.
______________________________________________________________________________________
Low-Cost, High-Frequency, Current-Mode PWM
Buck Controller
Setting the Output Voltage
IN
To set the output voltage for the MAX1953/MAX1954,
connect FB to the center of an external resistor-divider
connected between the output to GND (Figures 1 and
2). Select R2 between 8kΩ and 24kΩ, and then calculate R1 by:
BST
DH
MAX1953
MAX1954
MAX1957
LX
⎛V
⎞
R1 = R2 × ⎜ OUT − 1⎟
⎝ VFB
⎠
DL
Figure 5. DH Boost Circuit
Once these conditions are met, the step-down controller
enables soft-start and starts switching. The soft-start circuitry gradually ramps up to the feedback-regulation
voltage in order to control the rate-of-rise of the output
voltage and reduce input surge currents during startup.
The soft-start period is 1024 clock cycles (1024/f S,
MAX1954/MAX1957) or 4096 clock cycles (4096/f S,
MAX1953) and the internal soft-start DAC ramps the
voltage up in 64 steps. The output reaches regulation
when soft-start is completed, regardless of output
capacitance and load.
Shutdown
The MAX1953/MAX1954/MAX1957 feature a low-power
shutdown mode. Use an open-collector transistor to
pull COMP low to shut down the IC. During shutdown,
the output is high impedance. Shutdown reduces the
quiescent current (IQ) to approximately 220µA.
Thermal Overload Protection
Thermal overload protection limits total power dissipation
in the MAX1953/MAX1954/MAX1957. When the junction
temperature exceeds TJ = +160°C, an internal thermal
sensor shuts down the device, allowing the IC to cool.
The thermal sensor turns the IC on again after the junction temperature cools by 15°C, resulting in a pulsed output during continuous thermal overload conditions.
where VFB = 0.8V. R1 and R2 should be placed as
close to the IC as possible.
For the MAX1957, connect FB directly to the output
through a decoupling resistor of 10kΩ to 21kΩ (Figure
3). The output voltage is then equal to the voltage at
REFIN. Again, this resistor should be placed as close to
the IC as possible.
Determining the Inductor Value
There are several parameters that must be examined
when determining which inductor is to be used. Input
voltage, output voltage, load current, switching frequency, and LIR. LIR is the ratio of inductor current ripple to
DC load current. A higher LIR value allows for a smaller
inductor, but results in higher losses and higher output
ripple. A good compromise between size, efficiency,
and cost is an LIR of 30%. Once all of the parameters
are chosen, the inductor value is determined as follows:
L=
(
VOUT × VIN − VOUT
)
( )
VIN × fS × ILOAD MAX × LIR
where fS is the switching frequency. Choose a standard
value close to the calculated value. The exact inductor
value is not critical and can be adjusted in order to
make trade-offs among size, cost, and efficiency. Lower
inductor values minimize size and cost, but they also
increase the output ripple and reduce the efficiency due
to higher peak currents. By contrast, higher inductor values increase efficiency, but eventually resistive losses
due to extra turns of wire exceed the benefit gained
from lower AC current levels.
For any area-restricted applications, find a low-core
loss inductor having the lowest possible DC resistance.
Ferrite cores are often the best choice, although powdered iron is inexpensive and can work well at 300kHz.
______________________________________________________________________________________
15
MAX1953/MAX1954/MAX1957
Design Procedures
MAX1953/MAX1954/MAX1957
Low-Cost, High-Frequency, Current-Mode PWM
Buck Controller
The chosen inductor’s saturation current rating must
exceed the expected peak inductor current (IPEAK).
Determine IPEAK as:
⎛ LIR ⎞
IPEAK = ILOAD(MAX ) + ⎜
⎟ × ILOAD(MAX )
⎝ 2 ⎠
Setting the Current Limit
The MAX1953/MAX1954/MAX1957 use a lossless current-sense method for current limiting. The voltage
drops across the MOSFETs created by their on-resistances are used to sense the inductor current.
Calculate the current-limit threshold as follows:
VCS =
0.8V
A CS
where ACS is the gain of the current-sense amplifier.
ACS is 6.3 for the MAX1953 when ILIM is connected to
GND and 3.5 for the MAX1954/MAX1957, and for the
MAX1953 when ILIM is connected to IN or floating. The
0.8V is the usable dynamic range of COMP (VCOMP).
Initially, the high-side MOSFET is monitored. Once the
voltage drop across the high-side MOSFET exceeds VCS,
the high-side MOSFET is turned off and the low-side
MOSFET is turned on. The voltage across the low-side
MOSFET is then monitored. If the voltage across the lowside MOSFET exceeds the short-circuit current limit, a
short-circuit condition is determined and the low-side
MOSFET is held on. Once the monitored voltage falls
below the short-circuit current-limit threshold, the
MAX1953/MAX1954/MAX1957 switch normally. The shortcircuit current-limit threshold is fixed at 210mV for the
MAX1954/ MAX1957 and is selectable for the MAX1953.
When selecting the high-side MOSFET, use the following method to verify that the MOSFET’s RDS(ON) is sufficiently low at the operating junction temperature (TJ):
RDS(ON)N1 ≤
0.8V
A CS × IPEAK
The voltage drop across the low-side MOSFET at the
valley point and at ILOAD(MAX) is:
⎛ LIR ⎞
VVALLEY = RDS(ON) × (ILOAD(MAX) − ⎜
⎟ × ILOAD(MAX ) )
⎝ 2 ⎠
where RDS(ON) is the maximum value at the desired
maximum operating junction temperature of the MOS-
16
FET. A good general rule is to allow 0.5% additional
resistance for each °C of MOSFET junction temperature
rise. The calculated VVALLEY must be less than VCS.
For the MAX1953, connect ILIM to GND for a shortcircuit current-limit voltage of 105mV, to VIN for 320mV
or leave ILIM floating for 210mV.
MOSFET Selection
The MAX1953/MAX1954/MAX1957 drive two external,
logic-level, N-channel MOSFETs as the circuit switch
elements. The key selection parameters are:
• On-Resistance (RDS(ON)): The lower, the better.
• Maximum Drain-to-Source Voltage (VDSS): Should
be at least 20% higher than the input supply rail at
the high side MOSFET’s drain.
• Gate Charges (Qg, Qgd, Qgs): The lower, the better.
For a 3.3V input application, choose a MOSFET with a
rated RDS(ON) at VGS = 2.5V. For a 5V input application,
choose the MOSFETs with rated RDS(ON) at VGS ≤ 4.5V.
For a good compromise between efficiency and cost,
choose the high-side MOSFET (N1) that has conduction
losses equal to switching loss at the nominal input voltage and output current. The selected low-side and highside MOSFETs (N2 and N1, respectively) must have
RDS(ON) that satisfies the current-limit setting condition
above. For N2, make sure that it does not spuriously turn
on due to dV/dt caused by N1 turning on, as this would
result in shoot-through current degrading the efficiency.
MOSFETs with a lower Qgd/Qgs ratio have higher immunity to dV/dt.
For proper thermal management design, the power dissipation must be calculated at the desired maximum
operating junction temperature, TJ(MAX). N1 and N2
have different loss components due to the circuit operation. N2 operates as a zero-voltage switch; therefore,
major losses are the channel conduction loss (PN2CC)
and the body diode conduction loss (PN2DC):
USE RDS(ON)AT TJ(MAX)
V
PN2CC = (1 − OUT ) × I2LOAD × RDS(ON)
VIN
PN2DC = 2 × ILOAD × VF × tDT × fS
where VF is the body diode forward-voltage drop, tdt is
the dead time between N1 and N2 switching transitions, and fS is the switching frequency.
______________________________________________________________________________________
Low-Cost, High-Frequency, Current-Mode PWM
Buck Controller
⎞
⎛V
PN1CC = ⎜ OUT ⎟ × I2 LOAD × RDS(ON) USE RDS(ON) AT TJ(MAX)
⎝ VIN ⎠
⎛Q
+ QGD ⎞
PN2SW = VIN × ILOAD × ⎜ GS
⎟ × fS
⎝ IGATE
⎠
(
Output Capacitor
)
where IGATE is the average DH driver output current
capability determined by:
IGATE ≅
1
VIN
×
2
RDH + RGATE
where RDH is the high-side MOSFET driver’s on-resistance (3Ω max) and RGATE is the internal gate resistance of the MOSFET (~ 2Ω):
PN1DR = QG × VGS × fS ×
RGATE
RGATE + RDH
where VGS ~ VIN. In addition to the losses above, allow
about 20% more for additional losses due to MOSFET
output capacitances and N2 body diode reverse recovery charge dissipated in N1 that exists, but is not well
defined in the MOSFET data sheet. Refer to the MOSFET data sheet for the thermal-resistance specification
to calculate the PC board area needed to maintain the
desired maximum operating junction temperature with
the above calculated power dissipations.
The minimum load current must exceed the high-side
MOSFET’s maximum leakage current over temperature
if fault conditions are expected.
Input Capacitor
The input filter capacitor reduces peak currents drawn
from the power source and reduces noise and voltage
ripple on the input caused by the circuit’s switching.
The input capacitor must meet the ripple current
requirement (IRMS) imposed by the switching currents
defined by the following equation:
IRMS =
ILOAD ×
mended due to their low ESR and ESL at high frequency,
with relatively low cost. Choose a capacitor that exhibits
less than 10°C temperature rise at the maximum operating RMS current for optimum long-term reliability.
VOUT × (VIN − VOUT )
VIN
I RMS has a maximum value when the input voltage
equals twice the output voltage (VIN = 2 x VOUT), where
IRMS(MAX) = ILOAD/2. Ceramic capacitors are recom-
The key selection parameters for the output capacitor
are the actual capacitance value, the equivalent series
resistance (ESR), the equivalent series inductance
(ESL), and the voltage-rating requirements. These parameters affect the overall stability, output voltage ripple,
and transient response. The output ripple has three
components: variations in the charge stored in the output capacitor, the voltage drop across the capacitor’s
ESR, and the voltage drop across the ESL caused by
the current into and out of the capacitor:
VRIPPLE = VRIPPLE(ESR) + VRIPPLE(C) + VRIPPLE(ESL)
The output voltage ripple as a consequence of the ESR,
ESL, and output capacitance is:
VRIPPLE(ESR) = IP−P × ESR
VRIPPLE(C)
IP−P
8 × COUT × fS
⎛V ⎞
VRIPPLE(ESL) = ⎜ IN ⎟ ESL
⎝ L ⎠
⎛ V −V
⎞ ⎛
⎞
OUT × VOUT
IP−P = ⎜ IN
⎟ ⎜
⎟
⎝ fS × L ⎠ ⎝ VIN ⎠
where IP-P is the peak-to-peak inductor current (see the
Determining the Inductor Value section). These equations are suitable for initial capacitor selection, but final
values should be chosen based on a prototype or evaluation circuit.
As a general rule, a smaller current ripple results in less
output voltage ripple. Since the inductor ripple current
is a factor of the inductor value and input voltage, the
output voltage ripple decreases with larger inductance,
and increases with higher input voltages. Ceramic
capacitors are recommended for the MAX1953 due to
its 1MHz switching frequency. For the MAX1954/
MAX1957, using polymer, tantalum, or aluminum electrolytic capacitors is recommended. The aluminum
electrolytic capacitor is the least expensive; however, it
has higher ESR. To compensate for this, use a ceramic
capacitor in parallel to reduce the switching ripple and
noise. For reliable and safe operation, ensure that the
capacitor’s voltage and ripple-current ratings exceed
the calculated values.
______________________________________________________________________________________
17
MAX1953/MAX1954/MAX1957
N1 operates as a duty-cycle control switch and has the
following major losses: the channel conduction loss
(PN1CC), the voltage and current overlapping switching
loss (PN1SW), and the drive loss (PN1DR).
MAX1953/MAX1954/MAX1957
Low-Cost, High-Frequency, Current-Mode PWM
Buck Controller
The MAX1953/MAX1954/MAX1957s’ response to a load
transient depends on the selected output capacitors. In
general, more low-ESR output capacitance results in
better transient response. After a load transient, the
output voltage instantly changes by ESR ✕ ∆I LOAD.
Before the controller can respond, the output voltage
deviates further, depending on the inductor and output
capacitor values. After a short period of time (see the
Typical Operating Characteristics), the controller
responds by regulating the output voltage back to its
nominal state. The controller response time depends on
its closed-loop bandwidth. With a higher bandwidth,
the response time is faster, preventing the output voltage from further deviation from its regulating value.
Compensation Design
The MAX1953/MAX1954/MAX1957 use an internal
transconductance error amplifier whose output compensates the control loop. The external inductor, highside MOSFET, output capacitor, compensation resistor,
and compensation capacitors determine the loop stability. The inductor and output capacitors are chosen
based on performance, size, and cost. Additionally, the
compensation resistor and capacitors are selected to
optimize control-loop stability. The component values
shown in the Typical Application Circuits (Figures 1
through 4) yield stable operation over the given range
of input-to-output voltages and load currents.
The controller uses a current-mode control scheme that
regulates the output voltage by forcing the required
current through the external inductor. The MAX1953/
MAX1954/MAX1957 use the voltage across the highside MOSFET’s on-resistance (RDS(ON)) to sense the
inductor current. Current-mode control eliminates the
double pole in the feedback loop caused by the inductor and output capacitor, resulting in a smaller phase
shift and requiring less elaborate error-amplifier compensation. A simple single-series RC and CC is all that
is needed to have a stable high bandwidth loop in
applications where ceramic capacitors are used for
output filtering. For other types of capacitors, due to the
higher capacitance and ESR, the frequency of the zero
created by the capacitance and ESR is lower than the
desired close loop crossover frequency. Another compensation capacitor should be added to cancel this
ESR zero.
The basic regulator loop may be thought of as a power
modulator, output feedback divider, and an error amplifier. The power modulator has DC gain set by gmc x
RLOAD, with a pole and zero pair set by RLOAD, the output capacitor (COUT), and its equivalent series resistance (RESR).
18
Below are equations that define the power modulator:
GMOD = gmc ×
RLOAD × (fS × L)
(
)
RLOAD + fS × L
where RLOAD = VOUT/IOUT(MAX), and gmc = 1/(ACS ✕
RDS(ON)), where ACS is the gain of the current-sense
amplifier and RDS(ON) is the on-resistance of the highside power MOSFET. ACS is 6.3 for the MAX1953 when
ILIM is connected to GND, and 3.5 for the MAX1954/
MAX1957 and for the MAX1953 when ILIM is connected to VIN or floating. The frequencies at which the pole
and zero due to the power modulator occur are determined as follows:
fpMOD =
1
⎛R
⎞
LOAD × fS × L + RESR
⎟
2π × COUT × ⎜
⎜
⎟
R
f
L
+
×
LOAD
S
⎝
⎠
(
)
(
fzMOD =
)
1
2π × COUT × RESR
The feedback voltage-divider used has a gain of GFB =
VFB/VOUT, where VFB is equal to 0.8V. The transconductance error amplifier has DC gain, GEA(DC) = gm ✕
RO. RO is typically 10MΩ. A dominant pole is set by the
compensation capacitor (C C ), the amplifier output
resistance (RO), and the compensation resistor (RC). A
zero is set by the compensation resistor (RC) and the
compensation capacitor (CC).
There is an optional pole set by Cf and RC to cancel the
output capacitor ESR zero if it occurs before crossover
frequency (fC):
1
2 π × CC × (RO + RC )
1
fzEA =
2π × C C × R C
1
fpEA =
2π × C f × R C
fpdEA =
The crossover frequency (fC) should be much higher
than the power modulator pole f pMOD . Also, the
crossover frequency should be less than 1/5 the
switching frequency:
f
fpMOD
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