bq24721
bq24721C
www.ti.com.................................................................................................................................................. SLUS683F – NOVEMBER 2005 – REVISED APRIL 2009
ADVANCED MULTI-CHEMISTRY AND MULTI-CELL SYNCHRONOUS SWITCH-MODE
CHARGER AND SYSTEM POWER SELECTOR
FEATURES
APPLICATIONS
• High Efficiency NMOS-NMOS Synchronous
Buck Converter With User-Selectable 300 kHz
or 500 kHz frequency
• bq24721C Offers Softer Turn-On, Stronger
Turn-Off
• SBS-Like(1) SMBus Interface for Control and
Status Communications With Host
• Programmable Battery Voltage, Charge
Current, and AC Adapter Current via SBS-Like
SMBus Interface
• 0.4% Charge Voltage Regulation Accuracy
• 3% Charge Current Regulation Accuracy
• 3% Adapter Current Regulation Accuracy
Dynamic Power Management (DPM)
• 2% Accuracy Integrated Charge and AC
Adapter 20× Current Amplifier Output
• 3-Cell and 4-Cell Li-Ion Voltage Regulation
9 V, 12 V–14.4 V, 16 V–19.2 V
• Battery Pack Voltage Operating Range
0 V–19.2 V
• AC Adapter Operating Range 8 V–28 V
• 99.5% Max Duty Cycle
• Internal Soft Start
• Integrated 5% 5-V LDO When AC Adapter
Applied
• 6-V Drive Supply Voltage for Increased
Efficiency
• Reverse Battery to Adapter Discharge
Protection
• Battery/Adapter to System Power Selector
Function
• Charge and Adapter Overcurrent Protection
• Battery Thermistor Sense, TS, Comparators
• Available in 32-Pin 5x5-mm QFN Package
•
•
•
DESCRIPTION
The bq24721 is a high efficiency synchronous battery
pack charger with high level of integration for portable
applications. This device implements a high
performance analog front-end that interfaces to the
system power management micro-controller through
a simplified SBS-like SMBus interface.
The dynamic power management (DPM) function
modifies the charge current depending on system
load conditions, avoiding ac adapter overload.
High accuracy current sense amplifiers enable
accurate measurement of either the charge current or
the ac adapter current, allowing termination of
nonsmart packs and monitoring of overall system
power.
The adapter isolation diode can be bypassed or
entirely replaced with an external MOSFET using a
control signal provided by the bq24721, thus reducing
overall power dissipation.
Integrated features such as charger soft start, charge
overcurrent protection, and IC temperature monitoring
provide a second level of protection, in addition to
pack and system protection functions.
REGN
LODRV
PGND
ALARM
28
27
26
25
HIDRV
PH
30
29
PVCC
BTST
32
31
5x5 QFN PACKAGE (TOP VIEW)
CHGEN
1
24
BATDRV
ACDRV
2
23
SYS
ACN
3
22
SYNP
ACP
4
21
SYNN
ACDET
5
20
SRP
BYPASS
6
19
SRN
EAO
7
18
BAT
EAI
8
17
IOUT
13
14
15
16
SCL
TS
ISYNSET
VCC
SDA
11
12
VREF5
9
bq24721,
bq24721C
10
(1) SBS-Like interface is not 100% SBS compliant. SBS-Like
interface is SMBus1.1 complaint but does not support Packet Error
Correction (PEC). The control and status registers were changed
to simplify and enhance notebook charger control. An 8-bit address
(0x12) is used. See Table 1 for a comparison between SBS-like vs
SBS Specification.
Portable Notebook Computers
Portable DVD Players
Webpads, PC Tablets
FBO
2
AGND
1
1
2
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas
Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
PowerPAD is a trademark of Texas Instruments.
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of the Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
Copyright © 2005–2009, Texas Instruments Incorporated
bq24721
bq24721C
SLUS683F – NOVEMBER 2005 – REVISED APRIL 2009.................................................................................................................................................. www.ti.com
These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
ORDERING INFORMATION
PART NO.
PACKAGE
THERMISTOR
SENSE
BATTERY SHORTED
(VERY LOW BATTERY
VOLTAGE) OPERATION
bq24721
32 PIN
5x5mm QFN
TS
bq24721C
32 PIN
5x5mm QFN
TS
ORDERING NUMBER
(TAPE AND REEL)
QUANTITY
Charge Current Changes to
C/8
bq24721RHBR
3000
bq24721RHBT
250
Charge Current Changes to
C/8
bq24721CRHBR
3000
bq24721CRHBT
250
PACKAGE THERMAL DATA
(1)
(2)
PACKAGE (1)
θJA
TA ≤ 40°C
POWER RATING
DERATING FACTOR
ABOVE TA = 25°C
RHB (2)
36°C/W
2.36 W
0.028 W/°C
For the most current package and ordering information, see the Package Option Addendum at the end of this document, or see the TI
Web site at www.ti.com.
This data is based on using the JEDEC High-K board and the exposed die pad is connected to a copper pad on the board. This is
connected to the ground plane by a 2×3 via matrix.
DEVICE INFORMATION
TERMINAL FUNCTIONS
TERMINAL
NO.
CHGEN
Charge enable logic level low input. Logic HI on the CHGEN pin disables the charger. Logic LO on the
CHGEN pin enables the charger. When the SMBus control register = bit0, CHGEN is also LO.
2
ACDRV
AC adapter to system switch driver output. Connect directly to the gate of the ACFET PMOS power
FET. Connect the FET source to the PVCC node and negative side of the input current-sense resistor.
Connect the FET drain to the system load side. Recommend placing a 100-kΩ resistor from the gate to
the source of the Bypass FET. If needed, an optional capacitor from gate to source of the ACFET is
used to help slow down the ON and OFF times. The internal gate drive is asymmetrical allowing a
quick turn-off and slower turn-off in addition to the internal break-before-make logic with respect to the
BATDRV.
3
ACN
Adapter current sense resistor, negative input. An optional 0.1-µF ceramic capacitor is placed from this
pin to AGND for common-mode filtering. An optional 0.1-µF ceramic capacitor is placed from ACN to
ACP to provide differential-mode filtering.
4
ACP
Adapter current sense resistor, positive input. Place this on the adapter side of the input current sense
resistor. Recommend placing a 0.1-µF ceramic capacitor from ACP to AGND to provide common-mode
filtering.
ACDET
AC adapter detected sense voltage input. Connect a voltage divider resistor from adapter input (before
Bypass FET) to ACDET, and another resistor from ACDET to AGND, in order to program adapter
detect threshold of 1.2 V. ACDET threshold should be greater than maximum battery regulation
voltage, and lower than the minimum adapter voltage.
6
BYPASS
Gate drive for the adapter input BYPASS switch to prevent reverse discharge from the battery to the
input. Connect this pin directly to the gate of the input bypass PMOS power FET. The source of the
FET is connected to the adapter input voltage node. Recommend placing a 10-kΩ resistor from the
gate to the source of the BYPASS FET. The drain of the FET is connected to the positive node of the
input current-sense resistor. An optional capacitor can be placed from the gate to the source to
slow-down the switching times. Adjusting the turn-on and turn-off times is typically not needed for this
FET.
7
EAO
Error amplifier output for compensation. Connect the feedback compensation components from EAO to
EAI. Typically a capacitor in parallel with a series resistor and capacitor. See the compensation
calculation procedures. This node is internally compared to the PWM saw-tooth oscillator.
1
5
2
DESCRIPTION
NAME
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Copyright © 2005–2009, Texas Instruments Incorporated
Product Folder Link(s): bq24721 bq24721C
bq24721
bq24721C
www.ti.com.................................................................................................................................................. SLUS683F – NOVEMBER 2005 – REVISED APRIL 2009
TERMINAL FUNCTIONS (continued)
TERMINAL
NO.
DESCRIPTION
NAME
8
EAI
Error amplifier input for compensation, also connect the feedback compensation components from EAI
to EAO. Connect the input compensation components from FBO to EAI. See the compensation
calculation procedures.
9
FBO
Feedback output for compensation. Connect the input compensation components from FBO to EAI.
Typically, a resistor in parallel with a series resistor and capacitor. See the compensation calculation
procedures.
10
AGND
Analog ground. Ground connection for low current sensitive analog and digital signals. Only connect to
the PGND node by connecting to the PowerPAD™ underneath the IC.
11
VREF5
5-V regulated voltage output, used for internal bias and the reference for programming the TS
thermistor sense network. Used to indicate adapter present status. It is enabled by ac detected.
Connect a 1-µF ceramic capacitor from VREF5 pin to AGND as close to IC as possible.
12
VCC
IC analog positive supply. Connect to adapter input, or diode, or by putting a diode from adapter input
and a diode from battery pack to VCC. Put a 1-µF ceramic capacitor from VCC to AGND, as close to
the IC as possible.
13
SDA
SMBus Data input. Connect to SMBus data line from the host controller. A 10-kΩ pullup resistor to the
host controller supply rail is needed.
14
SCL
SMBus Clock input. Connect to SMBus clock line from the host controller. A 10-kΩ pullup resistor to
the host controller supply rail is needed.
15
TS
Thermistor sense input. Use a voltage divider from VREF5 to TS and AGND. Place a resistor from
VREF to TS, and a resistor from TS to AGND to program the hot and cold battery pack thermistor
temperatures. Charge is disabled when outside the hot/cold window. The TS pin is also used to detect
if a battery is connected.
16
ISYNSET
Program current threshold for synchronous to nonsynchronous regulation transition. Place a resistor
from ISYNSET to AGND to program the charge undercurrent threshold to force nonsynchronous
converter operation at low output current and prevent negative inductor current. Threshold should be
set from inductor current ripple to full value of inductor current ripple.
17
IOUT
Battery charger or adapter current amplifier output. Current sense amplifier that outputs a voltage 20x
the current sense resistor differential voltage. The output can be selected by SMBus charge control
register (0x12) bit3 to be the input adapter current (ACP-ACN), or the battery charge current
(SRP-SRN). Place a 0.1-µF capacitor from IOUT to AGND for filtering the output ripple. Optionally, add
an RC filter after the output filter for further filtering.
18
BAT
Battery voltage remote sense. Directly connect a kelvin sense trace from the battery pack positive
terminal to the BAT pin to accurately sense the battery pack voltage. Place a 0.1-µF capacitor from
BAT to AGND close to the IC to filter high frequency noise.
19
SRN
Charge current sense resistor, negative input. Connect to the charge current sense resistor negative
terminal. Optionally, add a 0.1-µF ceramic capacitor from SRN to AGND near the IC for common-mode
filter.
20
SRP
Charge current sense resistor, positive input. Connect to the charge current sense resistor positive
terminal. Recommend placing a 0.1-µF ceramic capacitor from SRP to AGND near the IC for
common-mode filter. Optionally, place a 0.1-µF ceramic capacitor from SRP to SRN near the IC for
differential-mode filter.
21
SYNN
Charge overcurrent and charge undercurrent negative sense input. Connect to the charge current
sense resistor negative terminal. If sensing the same sense resistor as SRN, the user can connect
directly to the SRN pin and no further filter capacitors are needed. To sense a different sense resistor,
add a 0.1-µF ceramic capacitor from SYNN to AGND near the IC for common-mode filter.
22
SYNP
Charge overcurrent and charge undercurrent negative sense input. Connect to the charge current
sense resistor positive terminal. If sensing the same sense resistor as SRP, the user can connect
directly to the SRP pin, and no further filter capacitors are needed. To sense a different sense resistor,
add a 0.1-µF ceramic capacitor from SYNP to AGND near the IC for common-mode filter, and place a
0.1-µF ceramic capacitor from SYNP to SYNN near the IC for differential-mode filter.
23
SYS
System load, voltage sense. Connect directly to the system load node and the source of the BAT
PMOS power FET.
Copyright © 2005–2009, Texas Instruments Incorporated
Product Folder Link(s): bq24721 bq24721C
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bq24721C
SLUS683F – NOVEMBER 2005 – REVISED APRIL 2009.................................................................................................................................................. www.ti.com
TERMINAL FUNCTIONS (continued)
TERMINAL
NO.
BATDRV
Battery to system switch driver output. Gate drive for the battery to system load BAT PMOS power FET
to isolate the system from the battery to prevent current flow from the system to the battery, while
allowing a low impedance path from battery to system while discharging the battery pack to the system
load. Connect this pin directly to the gate of the input BAT PMOS power FET. Connect the source of
the FET to the system load voltage node. Connect the drain of the FET to the battery pack positive
node. Recommend placing a 100-kΩ resistor from the gate to the source of the BAT FET. An optional
capacitor is placed from the gate to the source to slow-down the switching times. The internal gate
drive is asymmetrical allowing a quick turn-off and slower turn-off in addition to the internal
break-before-make logic with respect to the ACDRV.
25
ALARM
Alarm indicating charger status change, open-drain output. The ALARM is pulled low (LO) whenever
the SMBus status register (0x13) has a change. The ALARM output is cleared (HI) when the SMBus
status (0x13) register is read, or there is a reset. This is used to alert the host and initiate an interrupt
with the host instead of having to continuously poll the charger. A 10-kΩ pull-up resistor to the host
controller supply rail is needed.
26
PGND
Power ground. Ground connection for the high-current power converter nodes. Only connect to the
AGND node by connecting to the PowerPAD™ underneath the IC.
27
LODRV
PWM low side driver output. Connect directly to the gate of the low-side NMOS power FET with a short
trace.
28
REGN
Low-side driver gate voltage regulator and source for high-side driver bootstrap voltage. Add a 1-µF
ceramic capacitor from REGN pin to PGND pin, close to the IC. Place a small signal Schottky diode
from REGN to BTST for bootstrap voltage.
29
PH
High-side driver negative supply. Connect directly to the source of the high-side NMOS FET with a
short trace. This node is the common connection between the high-side FET, low-side FET, and output
inductor. Connect a 0.1-µF boot-strap ceramic capacitor from BTST to PH.
30
HIDRV
PWM high side driver output. Connect directly to the gate of the high-side NMOS power FET with a
short trace.
BTST
High-side driver positive supply, connect pos-side of boot-strap capacitor. Connect a 0.1-µF bootstrap
capacitor from the BTST pin to the PH node. Also, connect a bootstrap diode with the anode connected
to the REGN pin and the cathode connected to the BTST pin. An optional 4.7-Ω - 15-Ω series resistor
is placed between the BTST pin and the bootstrap-diode/capacitor junction to slow-down the turn-on
time of the high-side FET for reducing ringing due to high dv/dt of the phase node.
PVCC
IC power positive supply. Connect directly to the drain of the high-side NMOS power FET. Recommend
placing at least a 10-µF ceramic capacitor directly from the drain of the high-side NMOS power FET to
PGND. Up to 40 µF may be needed to prevent resonance filtering inductance. Also, a 0.1-µF
decoupling ceramic capacitor is recommended.
24
31
32
4
DESCRIPTION
NAME
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Copyright © 2005–2009, Texas Instruments Incorporated
Product Folder Link(s): bq24721 bq24721C
bq24721
bq24721C
www.ti.com.................................................................................................................................................. SLUS683F – NOVEMBER 2005 – REVISED APRIL 2009
ABSOLUTE MAXIMUM RATINGS
over operating free-air temperature range (unless otherwise noted) (1)
(2)
PARAMETER
PIN
VALUE / UNIT
ACN, ACP, PVCC, ACDRV, SYNN, SYNP, SRP, SRN,
BATDRV, BAT, BYPASS, SYS, VCC
Supply voltage range
–0.3 V to 30 V
PH
–1 V to 30 V
LODRV, REGN, FBO, EAI, EAO, ISYNSET, CHGEN, TS ,
VREF5, ACDET, IOUT, ALARM, SCL, SDA
–0.3 V to 7 V
BTST, HIDRV (with respect to AGND and PGND)
Maximum differential voltage
AGND-PGND
Maximum difference voltage
ACP–ACN , SRP–SRN, and SYNP–SYNN
–1 V to 36 V
–0.3 V to 0.3 V
0.6 V
Operating ambient temperature range (TA)
–40°C to 85°C
Maximum junction temperature (TJ_MAX)
150°C
Storage temperature range (Tstg)
(1)
(2)
–65°C to 150°C
Stresses beyond those listed under absolute maximum ratings may cause permanent damage to the device. These are stress ratings
only, and functional operation of the device at these or any other conditions beyond those indicated under recommended operating
conditions is not implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
All voltages are with respect to AGND, unless otherwise noted. Currents are positive into, negative out of the specified terminal. Consult
Packaging Section of the Databook for thermal limitations and considerations of packages.
RECOMMENDED OPERATING CONDITIONS
PARAMETER
PIN
ACN, ACP, PVCC, ACDRV, SRP, SRN,
BATDRV, BAT, BYPASS, SYS, VCC, SYNN,
SYNP
PH
Supply voltage range
MIN
0
NOM
MAX
UNIT
24
V
–0.5
30
V
LODRV, REGN, VREF5
0
6.5
V
FBO, EAI, EAO, ISYNSET, CHGEN, TS ,
ACDET, SCL, SDA, ALARM
0
5.5
V
IOUT, ACDET
0
5.5
V
BTST, HIDRV
0
30
V
0
V
Maximum differential voltage
AGND-PGND
Maximum difference voltage
ACP–ACN, SYNN–SYNP, SRP–SRN
Junction temperature Range (TJ)
Storage temperature Range (Tstg)
Copyright © 2005–2009, Texas Instruments Incorporated
Product Folder Link(s): bq24721 bq24721C
0.5
V
0
125
°C
-55
150
°C
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bq24721C
SLUS683F – NOVEMBER 2005 – REVISED APRIL 2009.................................................................................................................................................. www.ti.com
ELECTRICAL CHARACTERISTICS
8 Vdc ≤ V(VCC) ≤ 24 Vdc, 0°C ≤ TJ ≤ 125°C, all voltages with respect to AGND (unless otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
V
BATTERY VOLTAGE REGULATION
V(BAT_ICR)
V(VBATREG)
VBAT Input voltage range
V(BAT)
Battery Regulation Voltage Accuracy
Full valid voltage DAC range,
SMBus DAC register 0×15
0
PVCC
TJ = 0°C – 85°C
–0.4%
0.4%
TJ = 0°C – 125°C
–0.5%
0.5%
9
19.2
1.28
162.56
BAT voltage regulation range
V
PWM AC ADAPTER INPUT CURRENT REGULATION, DPM (Dynamic Power Management),
I(REG_DPM)= V(IREG_DPM)/R(SENSE_DPM)
V(IREG_DPM)
ACP-ACN differential voltage range
for input current regulation
V(IREG_DPM) = V(ACP)- V(ACN)
SMBus DAC register 0×3F, bits b0–b13
I(REG_step_DPM)
Current regulation LSB programming
current step
V(ACP-ACN) / 10mΩ
Using a 10mΩ sense resistor, R(SNS)
Current regulation accuracy
VCC ≥ VCC (min),
VCC ≥ VI(BAT) + V(DO-MAX), (1)
Over differential threshold range, V(IREG),
Does not include error induced by the
tolerance of the sense resistor, R(SNS)
128
mV
mA
V(ACP) – V(ACN) > 40.96 mV
(4096 mA with 10 mΩ)
–3%
3%
V(ACP) – V(ACN) > 20.48 mV
(2048 mA with 10 mΩ)
–5%
5%
V(ACP) – V(ACN) > 5.12 mV
(512 mA with 10 mΩ)
–25%
25%
1.28
162.56
PWM BATTERY CHARGE CURRENT REGULATION, I(REG_CHG)= V(IREG_CHG)/ R(SENSE_CHG)
V(IREG_CHG)
SRP-SRN differential voltage range
for input current regulation
V(IREG_CHG) = V(SRP)- V(SRN)
SMBus DAC register 0×14, bits b0–b13
I(REG_step_CHG)
Current regulation LSB programming
current step
V(SRP-SRN) / 10 mΩ
Using a 10mΩ sense resistor, R(SNS)
Current regulation accuracy
VCC ≥ VCC (min),
VCC ≥ VI(BAT) + V(DO-MAX), (1)
Over differential threshold range, V(IREG),
Does not include error induced by the
tolerance of the sense resistor, R(SNS)
128
mV
mA
V(SRP-SRN) > 40.96 mV
(4096 mA with 10 mΩ)
–3%
3%
V(SRP-SRN) >20.48 mV
(2048 mA with 10 mΩ)
–5%
5%
V(SRP-SRN) > 5.12 mV
(512 mA with 10 mΩ)
–25%
25%
2.5
20
V
0
3.5
V
CURRENT SENSE AMPLIFIERS – IBAT AMPLIFIER and IADAPT AMPLIFIER → MUX TO IOUT
SRP, SRN common-mode input
voltage range
V(IOUT_IBAT)
G(IBAT)
IOUT output voltage range with IBAT
selected
V(IOUT) = V(SRP, SRN) × A(IBAT)
V(BAT) > 2.5 V or V(BAT) > V(IOUT) + V(DO-MAX) (1)
Voltage gain
A(IOUT) = V(IOUT)/ V(SRP, SRN)
Charge current amplifier accuracy
V(BAT) > 2.5 V or V(BAT) >
V(IOUT) + V(DO-MAX) (1)
20
SRN)
= 40 mV and higher
V(SRP,
SRN)
= 20 mV and higher
–3%
3%
V(SRP,
SRN)
= 5 mV and higher
–25%
25%
0
24
ACP, ACN Common-mode input
voltage range
V(IOUT_IADAPT)
G(IADP)
I(OUT_LIM)
IOUT output voltage range with
IADAPT selected
V(IOUT) = V(ACP, ACN) × A(IADP)
V(SRP) > 2.5 V or V(SRP) > V(IOUT) + 1 V
Voltage gain
A(IADP) = V(IOUT)/ V(ACP, ACN)
Adapter current amplifier accuracy,
bq24721
V(BAT) > 2.5 V or V(BAT) >
V(IOUT) + V(DO-MAX) (1)
Adapter current amplifier accuracy,
bq24721C
V(BAT) > 2.5 V or V(BAT) >
V(IOUT) + V(DO-MAX) (1)
IOUT output current limit
IOUT shorted to AGND
–2%
V/V
V(SRP,
2%
0
3.5
20
V
V
V/V
V(ACP,
ACN)
= 40 mV and higher
–2%
2%
V(ACP,
ACN)
= 30 mV and higher
–3%
3%
V(ACP,
ACN)
= 5 mV and higher
–25%
25%
V(ACP,
ACN)
= 40 mV and higher
–2%
2%
V(ACP,
ACN)
= 30 mV and higher
–5%
5%
V(ACP,
ACN)
= 5 mV and higher
–25%
25%
4.5
mA
OPERATING CONDITIONS
V(INOP)
(1)
6
V(VCC), V(PVCC), input voltage operating
range
Selector and charger operational.
8
24
V
V(DO-max) is defined as the maximum drop-out voltage. V(DO-MAX) = 1 V unless other wise specified. In an actual application,
V(DO - MAX) = (R(SNS) × IO) + V(DSON_HIGH_SIDE_FET) + V(DSON_BYPASS_FET).
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www.ti.com.................................................................................................................................................. SLUS683F – NOVEMBER 2005 – REVISED APRIL 2009
ELECTRICAL CHARACTERISTICS (continued)
8 Vdc ≤ V(VCC) ≤ 24 Vdc, 0°C ≤ TJ ≤ 125°C, all voltages with respect to AGND (unless otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
QUIESCENT CURRENT – NO ADAPTER CONNECTED
I(VCC,PVCC)
VCC and PVCC quiescent current
I(VCC,PVCC) = ( I(VCC) + I(PVCC) ) at V(VCC) = V(PVCC) = 16.8 V
I(ACP,ACN)
ACP and ACN quiescent current
I(ACP,ACN) = ( I(ACP) + I(ACN) ) at V(ACP) = V(ACN) = V(VCC) = V(PVCC) = 16.8 V
I(BAT)
BAT quiescent current
I(BAT) at V(BAT) = V(VCC) = V(PVCC) = 16.8 V
I(SRP,SRN)
SRP and SRN quiescent current
I(SRP,SRN) = ( I(SRP) + I(SRN) ) at V(SRP) = V(SRN) = V(VCC) = V(PVCC) = 16.8 V
I(SYNN,SYNP)
SYNN and SYNP quiescent current
I(SYNN,SYNP) = ( I(SYNN) + I(SYNP) ) at V(SYNP) = V(SYNN) = V(VCC) = V(PVCC) = 16.8 V
I(SYS)
SYS quiescent current
I(SYS) at V(SYS) = V(VCC) = V(PVCC) = 16.8 V
I(PH)
PH quiescent current
I(BTST)
BTST quiescent current
254
µA
1
µA
17
µA
1
µA
1
µA
25
µA
I(PH) at V(PH) = V(VCC) = V(PVCC) = 16.8 V
1
µA
I(BTST) at V(BTST) = V(VCC) = V(PVCC) = 16.8 V
1
µA
QUIESCENT CURRENT – ADAPTER CONNECTED AND READY TO CHARGE
I(VCC,PVCC)
VCC and PVCC quiescent current
I(VCC,PVCC) = (I(VCC) + I(PVCC) at V(VCC) = V(PVCC) = 16.8 V
4.45
mA
I(ACP,ACN)
ACP and ACN quiescent current
I(ACP,ACN) = (I(ACP) + I(ACN) ) at V(ACP) = V(ACN) = V(VCC) = V(PVCC) = 16.8 V
815
µA
I(BAT)
BAT quiescent current
I(BAT) at V(BAT) = V(VCC) = V(PVCC) = 16.8 V
500
µA
I(SRP,SRN)
SRP and SRN quiescent current
I(SRP,SRN) = ( I(SRP) + I(SRN) ) at V(SRP) = V(SRN) = V(VCC) = V(PVCC) = 16.8 V
305
µA
I(SYNN,SYNP,SYS)
SYNN, SYNP, and SYS quiescent
current
I(SYNN,SYNP,SYS) = ( I(SYNN) + I(SYNP) + I(SYS) ) at V(SYNP) = V(SYNN) = V(SYS) = V(VCC) =
V(PVCC) = 16.8 V
321
µA
I(PH)
PH quiescent current
I(PH) at V(PH) = V(VCC) = V(PVCC) = 16.8 V
1
µA
I(BTST)
BTST quiescent current
I(BTST) at V(BTST) = V(VCC) = V(PVCC) = 16.8 V
1
µA
VCC Current while converter is
switching including gate drive current
I(VCC_SW) = I(VCC)
FPWM = 300 kHz, charger on (CHGEN = LO) = ENABLED
Q(G) at HIDRV = Q(G) at LODRV = 30 nC, [No Load on VREF5]
Gate drive switching current = Q(G) × FPWM = (30nC + 30nC) × 300kHz =
18mA
I(VCC_SW)
25
mA
5-V REFERENCE LDO VOLTAGE AND AC DETECTION STATUS (VREF5, TURNS ON WHEN AC DETECTED)
V(VREF5)
5V Regulator output voltage
Adapter detected (VACDET >V(ACD)), VCC> 7 V
0 → 10 mA, source current
V(VREF5_SAT)
Saturation voltage when VREF5 is off
Adapter not detected, (VACDET< V(ACD))
0 → –10 mA, ac adapter inserted, CO = 1 µF, discharge Load
I(VREF5_LIM)
Short-circuit current
V(VREF5) = AGND
4.75
5
5.25
V
0.3
V
20
mA
UNDERVOLTAGE LOCKOUT CIRCUIT
Undervoltage lockout threshold
VREF5 rising, POR mode set at VREF5 < V(UVLO)
3.7
V
V(UVLO) hysteresis
VREF5 falling
100
mV
UVLO
SBS-Like SMBus LOGIC LEVELS
VIL
Input low threshold level
2.7 V < V(pull-up) < 5.5 V, SDA and SCL
VIH
Input high threshold level
2.7 V < V(pull-up) < 5.5 V, SDA and SCL
I(bias)
Input bias current
2.7 V < V(pull-up) < 5.5 V, SDA and SCL
0.8
2.1
V
V
1
µA
ALARM OPEN DRAIN OUTPUT
V(ALARM_sat)
ALARM output low saturation level
I(ALARM) = 5mA
Ilkg(ALARM)
ALARM leakage current
V(ALARM) = 5V
0.5
V
1
µA
THERMAL SHUTDOWN, IC OVERTEMPERATURE PROTECTION
T(SHUT)
Thermal shutdown Threshold
TJ rising, Charge disabled at TJ > T(SHUT)
T(SHUTH)
Hysteresis
TJ falling, Charge enabled at TJ < T(SHUT) – T(SHUTH)
Deglitch time, thermal shutdown
TJ rising/falling
Copyright © 2005–2009, Texas Instruments Incorporated
Product Folder Link(s): bq24721 bq24721C
145
°C
15
°C
8
ms
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7
bq24721
bq24721C
SLUS683F – NOVEMBER 2005 – REVISED APRIL 2009.................................................................................................................................................. www.ti.com
ELECTRICAL CHARACTERISTICS (continued)
8 Vdc ≤ V(VCC) ≤ 24 Vdc, 0°C ≤ TJ ≤ 125°C, all voltages with respect to AGND (unless otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
THERMISTOR COMPARATORS, TS
V(LTF)
Cold temperature threshold, TS pin
voltage
V(TS) rising
72.8
73.5
74.2
%VREF5
V(LTFH)
Hysteresis for LTF threshold
V(TS) falling
0.5
1
1.5
%VREF5
V(TCO)
Cutoff temperature threshold, TS pin
voltage
V(TS) rising
28.7
29.3
29.9
%VREF5
V(HTF)
Hot temperature threshold, TS pin
voltage
V(TS) rising/falling
33.7
34.4
35.1
%VREF5
V(TSDET)
Pack thermistor insertion detected
VT(S) rising/falling
82.45
85
87.55
%VREF5
Deglitch time for temperature out of
range detection
V(TS) rising above V(LTF), or V(TS) falling below V(TCO) , or V(TS) falling below V(HTF)
Deglitch time for temperature in valid
range detection
V(TS) falling below (V(LTF) - V(LTFH)), or V(TS) rising above V(TCO) , or V(TS) rising
above V(HTF)
Deglitch time for thermistor removal
detection
V(TS) rising above V(TSDET)
V(TS) > V(TSDET) (pack removed)
Deglitch time for thermistor insertion
detection
V(TS) falling below V(TSDET)
V(TS) < V(TSDET) (pack inserted)
16
µs
8
ms
16
µs
1
s
CHARGE OVERCURRENT COMPARATOR
V(OLP)
Overcurrent protection threshold
V(SYNP-SYNN) rising,
V(BAT) = 12.4 V, V(IREG–CHG) DAC programmed to 81.92 mV
V(OLPH)
Hysteresis
V(SYNP-SYNN) falling
Deglitch time
V(SYNP-SYNN) rising and falling
170
200
250 %I(REG_CHG)
20
%I(REG_CHG)
1
µs
1
V
SYNCHRONOUS to NONSYNCHRONOUS CURRENT COMPARATOR (ISYNSET)
V(SYNSET)
K(SYNSET)
V(SYN_HYS)
ISYNSET pin set voltage
V(SYNP-SYNN) falling,
ISYN_NSYN= (V(SYNSET) × K(SYNSET))/(R(SYNSET) × R(SENSE_CHG))
500
V/A
V(SYNP-SYNN) hysteresis voltage, rising
V(SYNP-SYNN) rising
1.5
mV
Deglitch time, Synch to Non-Synch
V(SYNP-SYNN) rising and falling
1
µs
ISYNSET current set factor
ADAPTER OVERCURRENT COMPARATOR (ACOC)
V(ACOC)
V(ACOCH)
ACOC Input over-current protection
sense resistor voltage threshold
V(ACP-ACN) ≥ V(ACOC) , where SMBus charge mode register (0×12),
b6 = 1 = ACOC_protection_enabled, b8 = 0, b7 = 0
110
130
155
% of
I(REG_DPM)
V(ACP-ACN) ≥ V(ACOC) , where SMBus charge mode register (0×12),
b6 = 1 = ACOC_protection_enabled, b8 = 0, b7 = 1
130
150
175
% of
I(REG_DPM)
V(ACP-ACN) ≥ V(ACOC) , where SMBus charge mode register (0×12),
b6 = 1 = ACOC_protection_enabled, b8 = 1, b7 = 0
150
170
195
% of
I(REG_DPM)
V(ACP-ACN) ≥ V(ACOC) , where SMBus charge mode register (0×12),
b6 1 = ACOC_protection_enabled, b8 = 1, b7 = 1
170
190
215
% of
I(REG_DPM)
ACOC Hysteresis
V(SRP-SRN) falling
7
%I(REG_CHG)
ACOC deglitch time before ACDRV
turns-off
V(ACP-ACN) rising
16
µs
ACOC delay time after
V(ACP-ACN)≤V(ACOC) before ACDRV
turn-on
V(ACP-ACN) falling, V(ACP-ACN)≤V(ACOC)
8
ms
SYSTEM STATUS COMPARATORS INPUT SPECIFICATIONS
Common mode input range at pin:
ACDET
VICR
Common mode input range at TS pin
Common mode input voltage range
at pins: BAT, SYS
I(bias)
0
5
V
0.5
VREF5
V
0
VCC
V
0.2
µA
2.2
2.9
V/cell
-2%
2%
Input bias currents at pins: ACDET,
TS, BATDEP, SYS
BATTERY DEPLETED COMPARATOR (BATDEP)
V(BATDEP)
BAT depleted voltage range
V(BATDEP) = 2.2 V + V(step) × batdep_dac_code, where batdep_dac_code = 0 - 7,
and Vstep = 0.1V, SMBus Charge Mode register 0×12, bits b9, b10, b11
For programmed V(BATREG) = 9 V, then cell = 2
For programmed V(BATREG) = 12 V - 14.4 V, then cell = 3
For programmed V(BATREG) = 16 V - 19.2 V, then cell = 4
BAT depleted accuracy
8
Battery depleted detection deglitch
time
V(BAT) falling
1
s
Battery not depleted detection
deglitch time
V(BAT) rising
1
s
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bq24721C
www.ti.com.................................................................................................................................................. SLUS683F – NOVEMBER 2005 – REVISED APRIL 2009
ELECTRICAL CHARACTERISTICS (continued)
8 Vdc ≤ V(VCC) ≤ 24 Vdc, 0°C ≤ TJ ≤ 125°C, all voltages with respect to AGND (unless otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
1.176
1.2
1.224
UNIT
AC ADAPTER DETECT COMPARATOR (ACDET)
V(ACD)
AC adapter detect threshold
V(ACDET) rising, When adapter detected, VREF5 is enabled, and REGN
regulates to 6 V
V(ACDH)
AC adapter detect hysteresis
V(ACDET) falling
15
mV
AC adapter detected deglitch time
V(ACDET) rising
8
ms
AC adapter not detected deglitch
time
V(ACDET) falling
1
µs
V
AC ADAPTER (ACP) - BATTERY (BAT) COMPARATOR
V(ACP-BAT)
ACP voltage above BAT voltage
threshold
V(ACP) falling with respect to V(BAT)
V(ACP-BAT)
Hysteresis
V(ACP-BAT) rising with respect to V(BAT)
50
mV
V(ACP-BAT) falling below threshold
deglitch time
V(ACP) falling with respect to V(BAT)
16
µs
V(ACP-BAT) rising above threshold
deglitch time
V(ACP-BAT) rising with respect to V(BAT)
8
ms
250
300
mV
SYSTEM (SYS) - BATTERY (BAT) COMPARATOR
V(SYS-BAT)
System voltage above pack voltage
at V(VS,BAT) > V(SYS)
V(SYS) falling with respect to V(BAT)
V(SYS-BAT)
Hysteresis
V(SYS-BAT) rising with respect to V(BAT)
V(SYS-BAT) falling below threshold
deglitch time
V(SYS-BAT) rising above threshold
deglitch time
250
300
mV
50
mV
V(SYS) falling with respect to V(BAT)
1
µs
V(SYS-BAT) rising with respect to V(BAT)
8
ms
BATTERY SHORTED COMPARATOR
V(BATSHORT)
V(SHRT_HYS)
Battery shorted threshold (2)
V(BAT) falling,
Programmed V(BAT) = 9V
3.230
3.4
3.570
V
V(BAT) falling,
Programmed V(BAT) = 12-14.4V
4.845
5.1
5.355
V
V(BAT) falling,
Programmed V(BAT) = 16-19.2V
6.460
6.8
7.140
V
Hysteresis
V(BAT) rising
Battery shorted deglitch time
V(BAT) rising/falling
200
1
mV/cell
BYPASS off-state resistance
Driver output = HI, BYPASS = V(PVCC), V(PVCC) = 18 V
1
2
kΩ
BYPASS on-state resistance
Driver output = LO,
BYPASS = V(PVCC) - V(REGBYPASS), V(PVCC) = 18 V
1
2
kΩ
Drive regulator turn-on voltage for
BYPASS with respect to V(PVCC)
V(VCC, BYPASS), V(VCC) > 13 V, I(BYPASS) = 5 mA
-6
-7.5
100
150
Ω
10
20
kΩ
-6.5
-7.5
V
100
150
Ω
10
20
kΩ
-6.5
-7.5
V
s
BYPASS P-Channel MOSFET DRIVER (BYPASS)
R(DS_BYP)
R(DS_BYP)
Hi
Lo
V(REGBYPASS)
-5
V
AC ADAPTER P-Channel MOSFET DRIVER (ACDRV)
R(DS_AC)
Hi
ACDRV off-state resistance
Driver output = HI, ACDRV = PVCC, V(PVCC) = 18 V
R(DS_AC)
Lo
ACDRV on-state resistance
Driver output = LO, ACDRV =V(PVCC)-V(REGAC), V(PVCC) = 18 V
Drive regulator turn-on voltage for
ACDRV with respect to V(PVCC)
V(VCC, ACDRV), V(VCC) > 13 V, I(ACDRV) = 5 mA
V(REGAC)
-5
BATTERY P-Channel MOSFET DRIVER (BATDRV )
R(DS_BAT)
Hi
BATDRV off-state resistance
Driver output = HI, V(PVCC) = 18 V
R(DS_BAT)
Lo
BATDRV on-state resistance
Driver output = LO, BATDRV = V(SYS)- V(REGBAT), V(SYS) = 18 V
Drive regulator negative turn-on
voltage for BATDRV with respect to
V(SYS)
V(VCC, BATDRV), V(VCC) > 13 V, I(BATDRV) = 5 mA
V(REGBAT)
-5
SYSTEM POWER SELECTOR TIMING
Dead time when switching between
ACDRV and BATDRV
No load at ACDRV and BATDRV
1
µs
1
µs
BYPASS SWITCH TIMING
Delay to turn-off BYPASS
(2)
For the bq24721: When BAT falls below the V(BATSHORT) threshold, the charger continues regulating at the programmed current down to
zero volts on BAT; then after 1 second deglitch time, the charge current automatically changes to 1/8 the programmed charge current.
The charge current automatically changes from 1/8 the programmed charge current to the full programmed charge current when BAT
voltage rises above (V(BATSHORT) + 200 mV hysteresis), after 1 second deglitch time.
Copyright © 2005–2009, Texas Instruments Incorporated
Product Folder Link(s): bq24721 bq24721C
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bq24721
bq24721C
SLUS683F – NOVEMBER 2005 – REVISED APRIL 2009.................................................................................................................................................. www.ti.com
ELECTRICAL CHARACTERISTICS (continued)
8 Vdc ≤ V(VCC) ≤ 24 Vdc, 0°C ≤ TJ ≤ 125°C, all voltages with respect to AGND (unless otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
PWM HIGH-SIDE N-Channel MOSFET DRIVER (HIDRV), bq24721
R(DS_HIDRV)
Hi
High-side on-state resistance
HSD switch on, HIDRV = HI, V(BOOST,PH) = 5.5 V
2.2
3
Ω
R(DS_HIDRV)
Lo
High-side off-state resistance
HSD switch off, HIDRV = LO, V(BOOST,PH) = 5.5 V
1.5
2.5
Ω
PWM HIGH-SIDE N-Channel MOSFET DRIVER (HIDRV), bq24721C
R(DS_HIDRV)
Hi
High-side on-state resistance
HSD switch on, HIDRV = HI, V(BOOST,PH) = 5.5 V
4.5
8.6
Ω
R(DS_HIDRV)
Lo
High-side off-state resistance
HSD switch off, HIDRV = LO, V(BOOST,PH) = 5.5 V
1.5
2.6
Ω
PWM LOW-SIDE N-Channel MOSFET DRIVER (LODRV), bq24721
R(DS_LODRV)
Hi
Low-side on-state resistance
LSD switch on, LODRV = HI, V(PVCC) = 7 V
2.2
3
Ω
R(DS_LODRV)
Lo
Low-side off-state resistance
LSD switch off, LODRV = LO, V(PVCC) = 7 V
1.5
2.5
Ω
PWM LOW-SIDE N-Channel MOSFET DRIVER (LODRV), bq24721C
R(DS_LODRV)
Hi
Low-side on-state resistance
LSD switch on, LODRV = HI, V(PVCC) = 7 V
4.5
8.6
Ω
R(DS_LODRV)
Lo
Low-side off-state resistance
LSD switch off, LODRV = LO, V(PVCC) = 7 V
1.5
2.6
Ω
6
6.5
V
PWM LOW-SIDE DRIVER REGULATOR (REGN)
VO(HREGN)
IO(REGN_SW)
I(REGN_LIM)
REGN output voltage
REGN output current while charger
switching
REGN Current limit
Adapter detected
REGN Current limit
Adapter not detected
V(REGN) at I(REGN) = 10 mA, sourcing,
Adapter detected (V(ACDET) > V(ACD)), V(PVCC) > 7 V
5.5
V(REGN) at I(REGN) = 10 mA, sourcing,
Adapter not detected, (V(ACDET) < V(ACD)), V(PVCC) > 7 V
4.2
V
2 times 25 nC load, fs = 300 kHz
15
mA
2 times 25 nC load, fs = 500 kHz
25
mA
VREGN = 5 V
Adapter detected (V(ACDET) > V(ACD)), V(PVCC)> 7 V
100
mA
VREGN = 0 V, shorted
Adapter detected (V(ACDET) > V(ACD)), V(PVCC)> 7 V
13.3
mA
15
mA
30
ns
0.35
V
VREGN = 4.2 V
Adapter not detected, (V(ACDET) < V(ACD)), V(PVCC)> 7 V
PWM DRIVERS TIMING
Dead time when switching between
LSD and HSD, no load at LSD and
HSD
PWM OSCILLATOR
V(RAMPLO)
PWM oscillator ramp voltage , low
value
0% duty cycle occurs below this threshold
V(RAMPHI)
PWM oscillator ramp voltage , high
value
near 100% duty cycle occurs above this threshold
3
V
VPP(RAMP)
PWM ramp peak-to-peak amplitude
0.1×VCC
V(RAMPCL)
PWM oscillator ramp clamp voltage
3.5
V
FS
PWM oscillator frequency (300 kHz)
265
300
345
kHz
PWM oscillator frequency (500 kHz)
425
500
575
kHz
V
INTERNAL SOFT START (8 steps to Ireg)
SRSET pin voltage number of steps
during soft start.
Eight steps of charge current regulation to get to programmed value
(SRSET = 1 V).
Step Duration.
Eight steps of charge current regulation to get to programmed value
(SRSET = 1 V).
8
0.8
1
step
1.2
ms/step
CHARGER SECTION POWER-UP SEQUENCING
Time delay between power up of
charger block references (first) and
start charge (second)
Time delay from adapter detected
until ACDRV enable and charger
block enable
10
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1
ms
500
ms
Copyright © 2005–2009, Texas Instruments Incorporated
Product Folder Link(s): bq24721 bq24721C
bq24721
bq24721C
www.ti.com.................................................................................................................................................. SLUS683F – NOVEMBER 2005 – REVISED APRIL 2009
TYPICAL CHARACTERISTICS
VREF5 LOAD REGULATION
vs
LOAD CURRENT
VREF5 LINE REGULATION
vs
INPUT VOLTAGE
5
114
VI = 20 V
VI = 20 V
TJ = 0oC
4.98
o
TJ = 25 C
4.97
4.96
TJ = 85oC
4.95
4.94
TJ = 125oC
4.93
o
o
TJ = 0 C
4.984
TJ = 25 C
o
4.982
TJ = 85 C
4.98
4.978
o
TJ = 125 C
4.976
4.92
IL - VREF5 Current Limit - mA
4.986
VREF5 Line Regulation - V
VREF5 Load Regulation - V
4.988
VI = 20 V
4.99
VREF5 CURRENT LIMIT
vs
JUNCTION TEMPERATURE
5
10 15
20 25 30 35 40 45 50
112
111
110
109
108
4.974
0
113
7
9
11
IL − Load Current − mA
13
15
17
19
21
110 120 125
REGN LOAD REGULATION
vs
LOAD CURRENT
REGN LINE REGULATION
vs
INPUT VOLTAGE
REGN CURRENT LIMIT
vs
JUNCTION TEMPERATURE
116
VI = 20 V
VI = 20 V
TJ = 0 C
TJ = 0 C
TJ = 25oC
5.85
5.8
o
TJ = 85 C
5.75
5.91
o
TJ = 25 C
5.9
o
TJ = 85 C
5.89
5.88
o
TJ = 125 C
o
TJ = 125 C
5.7
IL - REGN Current Limit - mA
o
5.9
REGN Loin Regulation - V
10 15
20 25 30 35 40 45 50
7
9
IL − Load Current − mA
11
13
15
17
19
21
115.5
115
114.5
114
113.5
113
0
5.87
5
23 24
20
40
60
80
100
110 120 125
TJ − Junction Temperature − oC
VI − Input Voltage − V
Figure 4.
Figure 5.
Figure 6.
REGN CURRENT LIMIT
vs
FORCED REGN VOLTAGE
QUIESCENT CURRENT, NO ADAPTER
vs
JUNCTION TEMPERATURE
QUIESCENT CURRENT, WITH
ADAPTER
vs
JUNCTION TEMPERATURE
160
280
Quiescent Current, No Adapter - mA
VCC = 24 V
120
100
80
60
40
20
0
270
260
250
VI = 16.8 V
240
VI = 12.6 V
230
220
210
200
1
2
3
4
5
6
0
VI − Forced REGN Voltage − V
20
40
60
80
100
110 120 125
TJ − Junction Temperature − oC
Figure 7.
Figure 8.
Copyright © 2005–2009, Texas Instruments Incorporated
Product Folder Link(s): bq24721 bq24721C
Quescent Current, Adapter Connected - mA
REGN Load Regulation - V
100
Figure 3.
o
IL - REGN Current Limit - mA
80
Figure 2.
5.92
0
60
Figure 1.
VI = 20 V
140
40
TJ − Junction Temperature − oC
VI − Input Voltage − V
5.95
0
20
0
23 24
5.8
5.6
5.4
VI = 16.8 V
5.2
5
VI = 12.6 V
4.8
4.6
4.4
4.2
4
-15
5
25
45
65
85
105 125
TJ − Junction Temperature − oC
Figure 9.
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SLUS683F – NOVEMBER 2005 – REVISED APRIL 2009.................................................................................................................................................. www.ti.com
TYPICAL CHARACTERISTICS (continued)
ACDET THRESHOLD
vs
JUNCTION TEMPERATURE
ACDET HYSTERESIS
vs
JUNCTION TEMPERATURE
1.2012
19.25
VI = 20 V
6
19.2
1.2008
1.2006
1.2004
1.2002
1.2
1.1998
VI = 20 V
19.15
19.1
19.05
19
18.95
18.9
18.85
40
60
80
100
110 120 125
0
o
20
40
60
80
100
110 120 125
0
20
40
60
80
100
120 125
o
TJ − Junction Temperature − C
Figure 11.
Figure 12.
HIDRV, rDS(on), PULL DOWN
vs
JUNCTION TEMPERATURE
LODRV, rDS(on), PULL UP
vs
JUNCTION TEMPERATURE
LODRV, rDS(on), PULL DOWN
vs
JUNCTION TEMPERATURE
1.9
6
VI = 20 V
VI = 20 V
1.7
1.6
1.5
1.4
1.3
1.2
5
bq24721C
4.5
4
3.5
3
2.5
bq24721
2
20
40
60
80
100
110 120 125
1.7
1.6
1.5
1.4
1.3
1.2
1.1
1.5
1
VI = 20 V
1.8
rDS(on) - LODRV Pull Down - W
5.5
rDS(on) - LODRV Pull Up - W
1
0
TJ − Junction Temperature − oC
20
40
60
80
100
0
120 125
20
40
60
80
100
110 120 125
TJ − Junction Temperature − oC
o
TJ − Junction Temperature − C
Figure 13.
Figure 14.
Figure 15.
BATTERY VOLTAGE REGULATION
ACCURACY
vs
BATTERY VOLTAGE
BATTERY VOLTAGE REGULATION
vs
JUNCTION TEMPERATURE
CHARGE CURRENT REGULATION
ACCURACY
vs
SRP-SRN VOLTAGE
Battery Voltage Regulation Accuracy - %
0.1
VI = 24 V
0.08
0.06
o
TJ = 85 C
0.04
0.02
0
-0.02
TJ = 25oC
-0.04
o
TJ = 125 C
-0.06
-0.08
o
TJ = 0 C
0.1
15
Charge Current, Regulation Accuracy - %
rDS(on) - HIDRV Pull Down - W
bq24721
Figure 10.
1.1
Battery Voltage Regulation Accuracy - %
3
2.5
TJ − Junction Temperature − C
1.9
0
3.5
o
TJ − Junction Temperature − C
1.8
bq24721C
4
1.5
18.75
20
0
5
4.5
2
18.8
1.1996
VI = 20 V
5.5
rDS(on) - HIDRV Pull Up - W
Vhys - ACDET Hysteresis - mV
VI - ACDET Threshold - V
1.201
0.08
0.06
0.04
VI = 12.6 V
0.02
0
VI = 16.8 V
-0.02
-0.04
-0.06
-0.08
-0.1
-0.1
9
11
13
15
17
19
V(BAT) − Battery Voltage − V
Figure 16.
12
HIDRV, rDS(on), PULL UP
vs
JUNCTION TEMPERATURE
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0
20
40
60
80
100
110 120 125
o
VI = 20 V
13
11
o
TJ = 125 C
9
7
o
TJ = 85 C
5
o
TJ = 25 C
3
TJ = 0oC
1
-1
5
15
25
35
45
55
65
VI − SPR-SRN Voltage − mV
TJ − Junction Temperature − C
Figure 17.
Figure 18.
Copyright © 2005–2009, Texas Instruments Incorporated
Product Folder Link(s): bq24721 bq24721C
bq24721
bq24721C
www.ti.com.................................................................................................................................................. SLUS683F – NOVEMBER 2005 – REVISED APRIL 2009
TYPICAL CHARACTERISTICS (continued)
13
VI = 20 V
11
9
o
TJ = 125 C
7
TJ = 85oC
5
o
TJ = 25 C
3
o
TJ = 0 C
1
-1
9
13
17
21
25
29
33
37
41
VI = 20 V
4
5.12 mV
3
2
10.24 mV
1
40.96 mV
20.48 mV
0
-1
14
16
24
o
TJ = 85 C
0.2
0.18
0.16
TJ = 125oC
o
TJ = 25 C
0.14
0.12
0.1
0
10
20 30 40 50 60
70 80 90 100
Figure 21.
IO = DPM CURRENT SENSE
AMPLIFIER OFFSET
vs
JUNCTION TEMPERATURE
IO = ADAPTER OUTPUT CURRENT
LIMIT
vs
JUNCTION TEMPERATURE
IO = CHARGE SENSE AMPLIFIER
ACCURACY
vs
SRP-SRN VOLTAGE
VI = 20 V
0.7
0.6
0.5
0.4
0.3
0.2
0.1
0
20
40
60
80
100
110 120 125
5.95
5.9
5.85
5.8
5.75
5.7
5.65
5.6
0
o
TJ − Junction Temperature − C
20
40
60
80
100
110 120 125
o
TJ − Junction Temperature − C
0.2
0
-0.2
TJ = 125oC
-0.4
TJ = 85oC
-0.6
-0.8
o
TJ = 25 C
-1
VI = 20 V
o
TJ = 0 C
-1.2
-1.4
0
10
20 30 40 50 60
70 80 90 100
VI − SRP-SRN Voltage − mV
Figure 22.
Figure 23.
Figure 24.
IO = CHARGE CURRENT LIMIT
vs
JUNCTION TEMPERATURE
300-kHz SWITCHING FREQUENCY
vs
INPUT VOLTAGE
300-kHz SWITCHING FREQUENCY
vs
JUNCTION TEMPERATURE
311
300-kHz Switching Frequency - kHz
5.7
5.65
5.6
5.55
5.5
5.45
5.4
311
o
TJ = 85 C
VI = 20 V
300-kHz Switching Frequency - kHz
5.75
310.5
TJ = 125oC
310
309.5
o
TJ = 25 C
309
308.5
o
TJ = 0 C
20
40
60
80
100
110 120 125
14
o
16
18
20
22
24
VI − Input Voltage − V
TJ − Junction Temperature − C
Figure 25.
VI = 20 V
310.5
310
309.5
309
308.5
308
0
0.22
VI − ACP-ACN Voltage − mV
6
0.8
0.24
Figure 20.
1
0.9
VI = 20 V
o
TJ = 0 C
Figure 19.
IO - Adapter Output Current Limit - mA
IO = DPM Current Sense Amplifier Offset - mV
22
0.28
0.26
VI − Voltage − mV
VI − ACP-ACN Voltage − mV
IO - Charge Output Current Limit - mA
20
18
IO = Charge Current Sense Amplifier Accuracy - %
5
5
IO = DPM CURRENT SENSE
AMPLIFIER ACCURACY
vs
ACP-ACN VOLTAGE
IO = DPM Current Sense Amplifier Accuracy - %
INPUT CURRENT REGULATION (DPM)
ACCURACY
vs
INPUT VOLTAGE
Input Current Regulation (DPM) Accuracy - %
Input Current Regulation (DPM) Accuracy - %
INPUT CURRENT REGULATION (DPM)
ACCURACY
vs
ACP-ACN VOLTAGE
Figure 26.
Copyright © 2005–2009, Texas Instruments Incorporated
Product Folder Link(s): bq24721 bq24721C
0
20
40
60
80
100
110 120 125
o
TJ − Junction Temperature − C
Figure 27.
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bq24721C
SLUS683F – NOVEMBER 2005 – REVISED APRIL 2009.................................................................................................................................................. www.ti.com
TYPICAL CHARACTERISTICS (continued)
500-kHz SWITCHING FREQUENCY
vs
INPUT VOLTAGE
73.65
497.5
497
496.5
o
TJ = 25 C
496
o
TJ = 0 C
495
494.5
VI = 20 V
498
497.5
497
496.5
496
495.5
495
20
22
24
0
VI − Input Voltage − V
73.55
73.5
73.45
73.4
20
40
60
80
100
110 120 125
0
o
20
40
60
80
100
110 120 125
o
TJ − Junction Temperature − C
TJ − Junction Temperature − C
Figure 28.
Figure 29.
Figure 30.
THERMISTOR LTF THRESHOLD
vs
JUNCTION TEMPERATURE
THERMISTOR TCO THRESHOLD
vs
JUNCTION TEMPERATURE
THERMISTOR TSDET THRESHOLD
vs
JUNCTION TEMPERATURE
34.395
VI = 20 V
34.39
34.385
34.38
34.375
34.37
34.365
34.36
34.355
34.35
VREF% - Thermister TSDET Threshold - %
29.3
VREF% - Thermister TCO Threshold - %
34.4
VREF% - Thermister HTF Threshold - %
VI = 20 V
73.6
73.35
494.5
18
16
14
VI = 20 V
29.28
29.26
29.24
29.22
29.2
0
20
40
60
80
100
110 120 125
o
14
VREF% - Thermister LTF Threshold - %
o
TJ = 125 C
495.5
THERMISTOR LTF THRESHOLD
vs
JUNCTION TEMPERATURE
498.5
o
TJ = 85 C
498
500-kHz Switching Frequency - kHz
500-kHz Switching Frequency - kHz
498.5
500-kHz SWITCHING FREQUENCY
vs
JUNCTION TEMPERATURE
0
20
40
60
80
100
110 120 125
o
84.8
VI = 20 V
84.79
84.78
84.77
84.76
84.75
84.74
84.73
84.72
84.71
84.7
0
20
40
60
80
100
110 120 125
o
TJ − Junction Temperature − C
TJ − Junction Temperature − C
TJ − Junction Temperature − C
Figure 31.
Figure 32.
Figure 33.
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bq24721C
www.ti.com.................................................................................................................................................. SLUS683F – NOVEMBER 2005 – REVISED APRIL 2009
TYPICAL CHARACTERISTICS (continued)
EFFICIENCY
vs
BATTERY CHARGE CURRENT
REGULATION CURRENT
vs
SYSTEM CURRENT
4500
100
IDPM
V(BAT) = 16.8 V
4000
3500
Regulation Current − mA
Efficiency - %
95
90
V(BAT) = 12.6 V
85
VI = 19.5 V
80
TA = 20oC
V(BAT) = 12 V
VCC = 20 V
3000
R(sns) = 10 mW
2500
ICHG
2000
DPM Active
1500
1000
ILOOP Active
500
75
0
1
2
3
4
5
6
7
0
8
0
Battery Charge Current − A
500
1 k 1.5 k 2 k 2.5 k 3 k 3.5 k
4k
System Current − mA
Figure 35.
Figure 34.
TRANSIENT SYSTEM LOAD
SYSTEM SELECTOR GATE DRIVES AFTER ACDET
Adapter plugged in (ACDET goes above 1.2 V)
ACDRV has a 500-ms delay before turning on
Ch3
5 V/div
Power Loop Verification (DPM) Transient Response
Constant Current Regulation, Then (DPM) Regulation
BYPASS
Ch1
Ch3
0A
Ch1
20 V
Ch2
20 V
ACDRV
Ch4
20 V
Ch1
2 V/div
Ch3 = Battery Current
Ch2
5 V/div
Ch4
0A
Ch1 = Input Current
Ch4
5 V/div
Ch3
2 A/div
Ch4
2 A/div
Ch4 = System Current
Ch1
2 A/div
4.5 k
BATDRV
ACDET
t − Time = 100 ms/div
t − Time = 1 ms/div
Figure 36.
Ch1
0V
Figure 37.
Copyright © 2005–2009, Texas Instruments Incorporated
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SLUS683F – NOVEMBER 2005 – REVISED APRIL 2009.................................................................................................................................................. www.ti.com
TYPICAL CHARACTERISTICS (continued)
SYSTEM SELECTOR ADAPTER INSERTION
SYSTEM SELECTOR ADAPTER REMOVAL
System Power Selection
System Power Selection
Adapter is unplugged in, System switches over to battery once the adapter
voltage drops below VAIRLINE.
Adapter Voltage
Battery Voltage
System Voltage
Ch4
5 V/div
1-A System Load
Ch1
Ch2
Ch4
0V
Ch2
Ch4
Ch1
5 V/div 5 V/div 5 V/div
Ch2
Ch1
5 V/div 5 V/div
Adapter is plugged in, then 500 ms later the system switches over.
System Voltage
1-A System Load
Battery Voltage
Adapter Voltage
t − Time = 200 ms/div
Figure 38.
t − Time = 1 s/div
Figure 39.
REGN VREF5 POWER UP
SOFTSTART CHARGE CURRENT
Ch1
5 V/div
Ch2
2 V/div
Softstart Operation
Ch3
1 A/div
Ch4
2 V/div
VREF5 and REGN Power-Up
Inductor Current
Ch4
0V
REGN
VREF5
0A
Ch2
0V
AVCC/PVCC
Ch1
0V
t − Time = 1 ms/div
Figure 41.
t − Time = 2 ms/div
Figure 40.
16
Ch1
Ch2
Ch4
0V
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www.ti.com.................................................................................................................................................. SLUS683F – NOVEMBER 2005 – REVISED APRIL 2009
TYPICAL CHARACTERISTICS (continued)
NONSYNCHRONOUS TO SYNCHRONOUS TRANSITION
SYNCHRONOUS TO NONSYNCHRONOUS TRANSITION
Transition from Non-Synchronous to Synchronous
Transition from Non-Synchronous to Synchronous
Inductor Current
Ch4
0V
Ch1
10 V/div
Low side NMOS gate voltage
Ch1
0V
PH
Ch3
1 A/div
Ch3
0A
Ch4
5 V/div
Ch4
5 V/div
Ch3
0A
Low side NMOS gate voltage
Ch1
10 V/div
Ch3
1 A/div
Inductor Current
Ch1
0V
PH
t − Time = 10 ms/div
Figure 43.
t − Time = 10 ms/div
Figure 42.
BATTERY SHORT RESPONSE
Bootstrap Refresh Comparator Operation
BATSHORT Functionality
Inductor Current
Ch2
5 V/div
Ch3
2 A/div
NEAR 100% DUTY CYCLE BTST RECHARGE PULSE
Trefresh = 250 ms
BAT
Ch2
0V
BAT goes from 13 V to 3 V, 1 sec deglitch, charge turns off
BAT goes back to 13 V, 1 sec deglitch, charge turns back on
Ch2
0V
Low side NMOS gate
Ch3
1 A/div
Ch2
5 V/div
Ch3
0A
Ch1
10 V/div
Ch4
0V
PH
Inductor Current
Ch3
0A
Ch1
0V
t − Time = 400 ms/div
Figure 45.
t − Time = 40 ms/div
Figure 44.
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TYPICAL CHARACTERISTICS (continued)
CHARGE OVERCURRENT
ACOC ACDRV TURN OFF (32-µs DEGLITCH)
ACOC Rising Edge (32 ms deglitch)
ACN decreases below ACOC threshold
Regulating 4 A, then add 1 W in parallel
to reduce VBAT
VBAT
Ch1
5 V/div
Ch2
2 V/div
Charge Overcurrent Comparator Functionality
Ch1
20 V
ADCRV
Ch3
4A
Ch2
2 V/div
Ch3
2 A/div
Ch2
0V
ACN (ACP fixed)
Ch2
20 V
Inductor Current
t − Time = 20 ms/div
t − Time = 20 ms/div
Figure 47.
Figure 46.
ACOC ACDRV TURN ON (500-µs DEGLITCH)
SWITCHING CONTINUOUS CURRENT MODE (CCM)
Steady State Operation
ACOC Falling Edge (500 ms deglitch)
Ch1
20 V
ADCRV
Ch2
20 V
ACN (ACP fixed)
Ch2
Ch1
Ch3
Ch4
50 mV/div 2 A/div 10 V/div 10 V/div
Ch2
2 V/div
Ch1
5 V/div
ACN increases above ACOC threshold
Ch2 = VPH
Ch3 = Inductor Current
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Ch2
0V
Ch3
4A
Ch4
12 V
Ch4 = VOUTCAPS_AC Ripple
Max (C1)
Max (C2)
Max (C3)
Max (C4)
10.4 V
20.4 V
5.04 A
42.0 mV
Min (C1)
Min (C2)
Min (C3)
Min (C4)
8.0 V
-1.2 V
3.12 A
32.0 mV
t − Time = 1 ms/div
t − Time = 200 ms/div
Figure 48.
18
Ch1
12 V
Ch1 = VOUTCAPS
Figure 49.
Copyright © 2005–2009, Texas Instruments Incorporated
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www.ti.com.................................................................................................................................................. SLUS683F – NOVEMBER 2005 – REVISED APRIL 2009
TYPICAL CHARACTERISTICS (continued)
SWITCHING DISCONTINUOUS CURRENT MODE
vs
(DCM)
Ch3
Ch4
2 A/div 50 mV/div
Ch1
Ch2
10 V/div 10 V/div
Steady State Operation
Ch2 = VPH
Ch1
12.6 V
Ch1 = VOUTCAPS
Ch2
0V
Ch4 = VOUTCAPS_AC Ripple
Ch4
12.6 V
Max (C1)
Max (C2)
Max (C3)
Max (C4)
14.0 V
20.8 V
1.12 A
30.0 mV
Ch3 = Inductor
Current
Min (C1)
Min (C2)
Min (C3)
Min (C4)
11.6 V
-1.2 V
320.0 mA
22.0 mV
Ch3
4A
t − Time = 1 ms/div
Figure 50.
Copyright © 2005–2009, Texas Instruments Incorporated
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bq24721C
SLUS683F – NOVEMBER 2005 – REVISED APRIL 2009.................................................................................................................................................. www.ti.com
SIMPLIFIED BLOCK DIAGRAM
AGND
VCC
REFERENCE SYSTEM
−
−
−
−
−
ACDRV
BYPASS
VREF5
BYPASS SW
DRIVER
5v ldo when ACPRES
Power−on reset
UVLO
Voltage references
Internal timebase
20X
Select Adaptor current
or Charge current
Always on
AC SW AND BAT SW DRIVERS
− break−before−make LOGIC
Always on
IOUT
BATDRV
Always on
SRP
Always on
BYPON
SYNCHRONOUS SWITCHING
PWM CONVERTER
ACON
SRN
ACOC
LOGIC
ACP
CHGOC
− Power up/down sequencing
− Charge enable logic
− Charge enable sequencing
BDEP, ACCHG, THDET,
− System power selector logic
TCOLD, THOT ,VSHI, TERMDET, − System selector bbm logic
ADPSRC, VCCGTBAT, TCMP
− Deglitch times
CLK, POR
SYSTEM STATUS
COMPARATORS
ACGOOD
−
−
−
−
−
−
−
ACDET
SYS
TS
POWERON
CHARGERON
Always on
Battery depleted (1)
AC detection
(1)
*Thermistor inserted (1)
*Thermistor cold (2)
*Thermistor hot (2)
System voltage / pack voltage (2)
VCC above pack voltage (2)
− current loop
− dpm loop
− voltage loop
− sync/nonsync comparator
− charge overcurrent
− comparator
− charge current reference
− dpm current reference
− break−before−make logic
− duty−cycle limited, 0% to near
100% (99.5%)
− NMOS/NMOS drivers
− Internal soft start
− PWM oscillator 300 kHz/500 kHz
ACN
EAO
EAI
FBO
SYNP
SYNN
ISYNSET
BAT
BTST
HIDRV
(1) always on
(2) enabled when AC is detected
Enabled only when AC is
detected and CHGEN is low
TTL INPUT
BUFFER
ALARM
PH
REGN
LODRV
PVCC
VREG DAC
DECODER
SCL
SMBUS INTERFACE
MEMORY
SBS [b1−b14] to
bq24721 [7bit +1bit]
VREG DAC
1BIT Select Cells
7BIT (6.25mV/bit)
SR IREG DAC
7BIT (1.28mV/bit)
SDA
Always on
Always on
AC IREG DAC
7BIT (1.28mV/bit)
CHGEN
PGND
Figure 51. bq24721 SBS-Like SMBus Controlled Simplified Block Diagram
20
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www.ti.com.................................................................................................................................................. SLUS683F – NOVEMBER 2005 – REVISED APRIL 2009
TYPICAL APPLICATION bq24721
Q1
SI4435
Q2
SI4435
R1
0.010
ADAPTER+
C1
10 uF
bq24721
PACK+
C6
1 uF
ADAPTER-
R3
464 k
1%
SYS
ACN
C8
0.1uF
BATDRV
ACP
C9
0.1uF
C2
10 uF
ACDRV
VCC
D1
BAT54C
C7
0.1 uF
PVCC
C3
4x10uF
BYPASS
ACDET
R4
33.2 k
1%
PACK
THERMISTOR
SENSE
VREF5
VREF5
R5
5.6 k
1%
C10
1 uF
AGND
PH
L1
10 uH
C12 , 0.1uF
D2
BAT54
C13, 1uF
CHGEN
R6
10 k
R7
10 k
R10
20k
R9
7.5k
SMBus
IRQ
C11
0.1uF
IOUT
C18
0.1 uF
OPTIONAL
C16
0.1 uF
C22
130pF
R11
200k
C21
2000pF
ISYNSET
R12
33k
ALARM
A/D
C5
2x10 uF
C19
0.1 uF
EAO
EAI
FBO
SCL
SDA
C14
0.1uF
C15
0.1uF
C20
51pF
R8
10 k
PACK-
C17
0.1 uF
OPTIONAL
SRP
SRN
BAT
VREF5
PACK+
C4
2x10 uF
Q4
FDS6670A
LODRV
PGND
SYNP
SYNN
ACGOOD
R2
0.010
BTST
REGN
R13
118 k
1%
Q5
SI4435
Q3
FDS6670A
HIDRV
TS
EMBEDDED
CONTROLLER
HOST
SYSTEM
PowerPAD
Short PowerPAD to
PGND and AGND
Figure 52. bq24721 SBS-Like SMBus Host Control With System Power Selector
(bq24721 With TS Thermistor Sense Input Pin)
BOM Key Components (For Figure 52, bq24721 Typical Application Circuit)
Reference Designator
Description (1)
Qty
Q3
1
N-channel MOSFET, 30V, 12.5A, SO-8, FDS6680A
Q4
1
N-channel MOSFET, 30V, 13A, SO-8, FDS6670A
Q1, Q2, Q5
3
P-channel MOSFET, -30V,-6A, SO-8, Vishay-Siliconix, Si4435
D1
1
Diode, Dual Schottky, 30V, 200mA, SOT23, Fairchild, BAT54C
D2
1
Diode, Single Schottky, 30V, 200mA, SOT23, Fairchild, BAT54
L1
1
Inductor, 10µH, 7A, 31mΩ, Vishay-Dale, IHLP5050FD-01
R1, R2
2
Sense Resistor, 10 mΩ, 1%, 1W, 2010, Vishay-Dale, WSL2010R0100F
C1, C2, C3, C4, C5
10
Capacitor, Ceramic, 10µF, 35V, 20%, X5R, 1206, Panasonic,
ECJ-3YB1E106M
C6, C10, C13
3
Capacitor, Ceramic, 1µF, 25V, 10%, X7R, 2012, TDK, C2012X7R1E105K
C7, C8, C9, C12, C14, C15, C16,
C19, (C17 and C18 optional)
10
Capacitor, Ceramic, 0.1µF, 50V, 10%, X7R, 0805, Kemet,
C0805C104K5RACTU
C20
1
Capacitor, Ceramic, 51pF, 50V, 5%, NPO, 0603
(1)
The manufacturer's part number are used for test purposes only.
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BOM Key Components (For Figure 52, bq24721 Typical Application Circuit) (continued)
Reference Designator
Description (1)
Qty
C21
1
Capacitor, Ceramic, 2000pF, 50V, 5%, X7R, 0805
C22
1
Capacitor, Ceramic, 130pF, 50V, 5%, NPO, 0603
R3
1
Resistor, Chip, 464kΩ, 1/16W, 1%, 0402
R4
1
Resistor, Chip, 33.2kΩ, 1/16W, 1%, 0402
R12
1
Resistor, Chip, 33kΩ, 1/16W, 5%, 0402
R9
1
Resistor, Chip, 7.54kΩ, 1/16W, 1%, 0402
R10
1
Resistor, Chip, 20kΩ, 1/16W, 1%, 0402
R11
1
Resistor, Chip, 200kΩ, 1/16W, 1%, 0402
R6, R7, R8
3
Resistor, Chip, 10kΩ, 1/16W, 5%, 0402
R5
1
Resistor, Chip, 5.6kΩ, 1/16W, 1%, 0402
R13
1
Resistor, Chip, 118kΩ, 1/16W, 1%, 0402
Typical bq24721 Narrow VDC (NVDC) Application (2 sense resistors)
Q1
SI4435
SYSTEM
ADAPTER +
C1
10 uF
bq24721
PACK+
C6
1uF
ADAPTER -
R3
464k
1%
C8
0.1uF
VCC
SYS
ACN
BATDRV
D1
BAT54C
C7
0.1uF
ACP
C9
0.1uF
C2
10 uF
ACDRV
PVCC
C3
4x10uF
BYPASS
ACDET
PACK
THERMISTOR
SENSE
R4
33.2k
1%
VREF5
VREF5
C10
1uF
R5
5.6 k
1%
AGND
Q3
FDS6670A
HIDRV
PH
C12, 0.1uF
D2
BAT54
EMBEDDED
CONTROLLER
HOST
C13, 1uF
C4
LODRV
PGND
SYNP
SYNN
CHGEN
R6
10k
R7
10k
EAI
FBO
SCL
SDA
SMBus
IRQ
ISYNSET
C11
0.1uF
IOUT
C22
130 pF
R11
200 k
C21
2000 pF
ALARM
A/D
R10
20k
C16
0.1uF
C19
0.1uF
EAO
R9
7.5k
C14
0.1uF
C18
0.1uF
C15
0.1uF
C20
51pF
R8
10 k
C5
2x10uF
C17
0.1uF
SRP
SRN
BAT
VREF5
PACK-
2x10uF
Q4
FDS6670A
TS
ACGOOD
PACK+
BTST
REGN
R1
118k
1%
R13
0.010
R2
0.010
L1
10uH
R12
33k
PowerPAD
Short PowerPAD to
PGND and AGND
Figure 53. bq24721 SBS-Like SMBus Host Control, NVDC (no system power selector) With 2 Sense
Resistors. ACP and ACN Regulating Converter Current Instead of Input Current
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www.ti.com.................................................................................................................................................. SLUS683F – NOVEMBER 2005 – REVISED APRIL 2009
Typical bq24721 Narrow VDC (NVDC) Application (3 sense resistors)
Q1
SI4435
R1
0.010
SYSTEM
ADAPTER+
C1
10 uF
bq24721
PACK+
C6
1 uF
C7
0.1 uF
ADAPTER-
D1
BAT54C
R3
464 k
1%
C8
0.1 uF
VCC
SYS
ACN
BATDRV
ACP
C9
0.1 uF
C2
10 uF
ACDRV
PVCC
C3
4x10uF
BYPASS
R4
33.2 k
1%
PACK
THERMISTOR
SENSE
VREF5
PH
VREF5
R5
5.6 k
1%
C12
0.1 uF
REGN
D2
BAT54
LODRV
PGND
SYNP
SYNN
R13
118 k
1%
C4
2x10 uF
Q4
FDS6670A
ACGOOD
CHGEN
VREF5
R7
10 k
C5
2x10 uF
C17
0.1 uF
EAO
R9
7.5 k
EAI
FBO
R8
10 k
C21
2000 pF
SCL
SDA
IRQ
ISYNSET
ALARM
A/D
IOUT
C11
0.1uF
C14
0.1 uF
C18
0.1 uF
C15
0.1 uF
C20
51 pF
SMBus
PACK+
PACK-
SRP
SRN
BAT
R6
10 k
R13
0.010
C13, 1 uF
TS
EMBEDDED
CONTROLLER
HOST
R2
0.010
L1
10 uH
BTST
AGND
C10
1 uF
Q3
FDS6670A
HIDRV
ACDET
R10
20 k
C16
0.1 uF
C19
0.1 uF
C22
130 pF
R11
200 k
R12
33 k
PowerPAD
Short PowerPAD to
PGND and AGND
Figure 54. bq24721 SBS-Like SMBus Host Control, NVDC (no system power selector)
With 3 Sense Resistors
Copyright © 2005–2009, Texas Instruments Incorporated
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bq24721
bq24721C
SLUS683F – NOVEMBER 2005 – REVISED APRIL 2009.................................................................................................................................................. www.ti.com
5-V LDO
VCC
VREF5
ENA
ACDET
ADAPTER
DETECTED
+
-
BAT
CHGEN
BATDEP
-
SMBUS
BATDEPSET
SYS
+
SYS-BAT
-
250 mV
PVCC-6V
SRP-BAT
+
+
_
1.2 V
ACOC
BAT + 250 mV
PVCC-6V
LDO
PVCC
PVCC
BYPASS
EAO
PVCC-6V
SYSTEM
POWER
SELECTOR
LOGIC
EAI
ACDRV
PVCC-6V
SYS-6V
FBO
ACOC_SET
+
SYS
SYS 6-V
LDO
SYS
ACOC
BATDRV
ACP
+
V(ACP-ACN)
-
IIN
DAC
ACN
250 mV
SYS-6V
ACP-BAT
-
SMBUS
CHGEN
COMP
ERROR
AMPLIFIER
+
BTST
CHGEN
-
+
+
_
BAT
ICH_ER
+
1V
BAT + 250 mV
VBAT
DAC
LEVEL
SHIFTER
BAT_ER
+
HIDRV
20 mA
OCP
PH
SRP
SYNCH
+
V(SRP-SRN)
ICH_ER
-
IBAT
DAC
SRN
ACP-BAT
DC-DC
CONVERTER
PWM LOGIC
+
VCC
20 mA
SUSPEND
REGN
6-V LDO
TSDET
500 kHz
2 x IBAT_DAC
SYNP
+
-
V(SYNP-N)
4.5 V
+
SYNN
300 kHz
REFRESH
CBTST
LODRV
SW FREQ
SELECT
+
_
PH
SYNCH
PGND
-
ISYNSET x 500
VREF5
1V
+
_
+
+
-
ISYNSET
BTST
OCP
+
IC TJ
+
145oC
-
-
SRP
+
SRN
-
20x
20x
IOUT
V(IBAT)
CHARGE STATUS
REGISTER
LTF
HTF
SUSPEND
+
-
TSDET
TS
AGND
ACN
V(IADAPT)
SELECT
CHARGE MODE
REGISTER
+
+
+
TSHUT
-
ACP
VBAT
DAC
ALARM
SDA
SMBus LOGIC
IBAT
DAC
TCO
SCL
IIN (DPM)
DAC
Figure 55. bq24721 Functional Block Diagram
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ADAPTER
POWER
external
SYSTEM & CHARGE POWER
SELECTOR
Vin
Bat
Chrg
SYSTEM
Chrg
Dischrg
Dischrg
bq24721
BATTERY
CHARGER
CONTROLLER
IC
&
CONVERTER
PSID
I/O
SMBus
BATTERY
Primary
BATTERY
Secondary
SMBus
SMBus
I/O
HOST - Embedded Controller (EC)
Figure 56. Host Controller
BLOCK DESCRIPTION
Detail Block Diagram
The bq24721 charge controller can be used to charge Li-Ion, NiMH, or NiCd batteries. The high efficiency
synchronous buck controller uses n-channel power MOSFETs for both the high-side control device and the
low-side synchronous device. The controller offers high regulation accuracy of the charge current, battery
voltage, and input current limits. The low offset of the current loops allow using sense resistors with low-value,
such as 10 mΩ.
An embedded controller host programs the battery voltage, charge current, and input current regulation limit
thresholds through an SMBus interface using SBS-like DAC registers. The embedded host can control the
operation of the charger through a Charge Control (0x12) register, and monitor the status of the charger through
a Charger Status (0x13) register.
The voltage loop regulates the battery voltage to the programmed value, and prevents the voltage from
exceeding that value when the battery is connected. The charge current loop regulates the battery charge current
to the programmed value, and prevents the charge current from exceeding that value. Through the use of
dynamic power management (DPM), the input current loop regulates the battery charge current to the
programmed value, and prevents the input current from exceeding that value. The three regulation loops operate
independently, yet only require a single loop compensation network.
The system power selector function selects the appropriate power source for the system load. If the adapter is
detected, then the adapter is connected to the system load. When the adapter is removed, the battery is selected
to power the system load. A battery learn cycle is performed when the adapter is present by setting the
CONTROL(0x12) register into Learn Mode via SMBus by the embedded host. This disconnects the adapter from
the system; and instead, connect the battery to the system. This is typically done for Ni-based batteries.
SMBus Interface
The bq24721 uses all the SMBus communications protocol, except for packet error correction (PEC). The
charger IC address is (0x12), although it is not 100% SBS compliant. In most applications, the extra functionality
provided by the differing SBS-Like interface enhances the control of the charger application, while simplifying the
interface block, using only the pertinent functions. Five 16-bit registers are used to interface between the
embedded host and the charge control IC. The Charging Voltage (0x15) register is used to set the battery
regulation voltage. The Charging Current (0x14) register is used to set the battery charge regulation current. The
Input Current (0x3F) register is used to set the input regulation current. The Charger Mode (0x12) control register
is used to set the charger operating modes. Finally, the Charger Status (0x13) register is used to monitor the
operating status of the charger.
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ADDRESS
REGISTER
DESCRIPTION
0x15
Charging voltage
Used to set the battery regulation voltage.
0x14
Charging current
Used to set the battery charge regulation current.
0x3F
Input current
Used to set the input regulation current.
0x12
Charger mode
Used to set the charger operating modes.
0x13
Charger status
Used to monitor the operating status of the charger.
The SMBus communications requires only two pins besides the analog ground pin—through the SDA (data) and
SCL (clock) pins. The open-drain SCL and SDA pins require pull-up resistors on the board pulling up to the host
digital output voltage rail. The pins can be pulled up to any rail between 3 V to 5 V.
An alarm is sent to the host through the ALARM open drain pin. This is used to trigger an interrupt request (IRQ).
A low on the ALARM pin indicates there was a change on the Charger Status (0x13) register. The ALARM pin
stays low until the host reads the Charger Status (0x13) register, then the ALARM pin clears and returns to the
HI state. The open-drain ALARM pin requires a pull-up resistor that pulls up to the host digital input voltage rail.
The pin can be pulled up to any rail between 3 V to 5 V. The charge controller continues to operate whether the
host chooses to read or not to read the Charger Status (0x13) register. There is no communications watchdog
timer, so there is no need to continuously poll the Charger Status (0x13) register or have to continuously
reprogram any of the other registers.
Setting Charge Voltage (VBAT DAC register)
The charge voltage can be programmed by setting Charging Voltage(0x15) register. The SBS specification asks
for 16 bits to set the regulation voltage with a 1-mV LSB—giving a maximum possible voltage of 65.535 V. The
bq24721 uses bits 1 through 14 only, and maps them into the closest value of an internal 7-bit DAC on a per cell
basis using a 6.25-mV per cell LSB for three ranges: 9-V precharge voltage; 3 cells Li+ battery pack range; or 4
cells Li+ battery pack range. The 3 cell portion has an LSB of 18.75 mV and a range from 12 V-14.4 V; while the
4 cell portion has an LSB of 25 mV and a range from 16 V-19.2 V. Intermediate programmed voltages between
the internal 7-bit DAC values are truncated to the lower value to avoid an overvoltage on the battery.
Programmed voltages above 19.2 V are automatically set to the maximum 19.2-V limit. The charger is disabled
for programmed voltages below 12 V (except for 9 V), and for programmed voltages between 14.4 V–16 V. This
triggers a voltage-out-of-range condition and the VOR bit of the Charger Status (0x13) register is set, and an
alarm (ALARM pin pulled low) is sent to the host. The default power-up-reset voltage value is 0 V, charger
disabled.
Setting Charge Current (IBAT DAC register)
The charge current can be programmed using the Charging Current (0x14) register. The SBS specification asks
for 16 bits to set the charge current with a 1-mA LSB—giving a maximum possible current of 65.535 A using a
10-mΩ sense resistor. The bq24721 uses bits 7 – 14 only to limit the range within a practical operating range.
The current range is 0 mA to 16.384 A with an LSB of 128 mA using a 10-mΩ sense resistor. The charger is
disabled when the programmed input current is 0 A. The default power-up-reset current value is 0 A, charger
disabled. Other sense resistors can be used to set the charge current—the user needs to transform the DAC
current table values to the new current values by dividing the current by 10 mΩ, then multiplying by the new
sense resistor value used.
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Setting Input (DPM) Current (IDPM DAC register)
The input current can be similarly programmed using the Input Current (0x3F) register. The SBS specification
asks for 16 bits to set the input current with a 1-mA LSB—giving a maximum possible current of 65.535 A using
a 10-mΩ sense resistor. The bq24721 uses bits 7 – 14 only to limit the range within a practical operating range.
The current range is 0 mA to 16.384 A with an LSB of 128 mA using a 10-mΩ sense resistor. The charger is
disabled when the programmed input current is 0 A. The default power-up-reset current value is 0 A, charger
disabled. Other sense resistors can be used to set the input current—the user needs to transform the DAC
current table values to the new current values by dividing the current by 10 mΩ, then multiplying by the new
sense resistor value used.
Power Up
When the adapter is not detected, the REGN output voltage is 4.6 V and the VREF5 LDO regulator is off, to
lower the power consumption from the battery. The VREF5 LDO is pulled-down to AGND when the adapter is
not detected. The REGN LDO regulator begins to regulate at 4.6 V when the input VCC voltage is greater than
6 V. The REGN output voltage is then 6 V when the adapter is detected, and the VCC is greater than 7 V. If
adapter is detected, but VCC is less than 7 V, then the REGN is in dropout, and REGN output voltage depends
on the VCC voltage and REGN load current. The VREF5 LDO is allowed to turn-on and regulate to 5 V, 5 ms
after REGN is 6 V and the adapter is detected. There is a 500-ms delay from the time the adapter is detected,
until the ACFET from the system power selector is allowed to turn-on, and until the charger is allowed to turn-on.
The battery continues to be connected to the system during this 500-ms delay.
Adapter Detect
The adapter detect threshold is programmed by an external voltage divider resistor from the adapter to the
ACDET pin. The internal ACDET comparator has a 1.2-V rising-edge threshold and a 15-mV falling-edge
hysteresis. The adapter detect value is typically programmed to a value greater than the maximum battery
voltage, and lower than the minimum allowed adapter voltage. The ACDETECT divider is placed before the
BYPASS FET in order to sense the true adapter input voltage whether the BYPASS is on or off.
The VREF5 LDO output is also used to indicate when the adapter is detected, for both the bq24721 and the
bq24721, since the VREF5 LDO only comes up when the adapter is detected.
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System Power Selector
The bq24721 can automatically switch between adapter power or battery power to the system load. The battery
is connected to the system when there is no adapter detected. The adapter is connected to the system when the
adapter is detected. An automatic break-before-make logic prevents shoot-through currents when the selector
switches.
When no adapter is detected the ACDRV pin is pulled to the PVCC pin to keep the external ACFET p-channel
power MOSFET off, disconnecting the adapter from system. The break-before-make logic waits until the ACFET
is off and the System to battery voltage comparator indicates the system voltage is within 250 mV of the Battery
(SYS voltage falling-edge, with a 50-mV hysteresis SYS voltage rising-edge). This prevents shoot-through
currents or large discharge currents from going into the battery. The BATDRV pin is then set to the SYS pin
voltage minus 6 V by an internal regulator in order to turn on the external BATFET p-channel power MOSFET,
connecting the battery pack to the system.
When adapter is detected there is a 500-ms delay; then, the BATDRV is set to the SYS pin voltage to turn off the
external BATFET p-channel power MOSFET, in order to disconnect the battery. The break-before-make logic
waits until the BATFET is off to prevent shoot-through currents. The ACDRV pin is then set to the PVCC pin
voltage minus 6 V by an internal regulator in order to turn on the external ACFET p-channel power MOSFET,
connecting the adapter to the system.
The host can override the adapter to system connection when adapter is present, by setting the charger into
Learn Mode. The host can then induce a learn cycle in which the battery is allowed to discharge. A learn cycle is
used to recalibrate the fuel gauge for Ni-based batteries, or for clearing the memory effect of a Ni-cd battery.
After discharging the battery, Learn Mode can be disabled allowing the adapter to be reconnected to the system,
then resuming a normal charge cycle.
Whenever the battery is connected to the system (whether in learn cycle with adapter present, or no adapter
present), there is a Low Battery comparator that monitors the battery voltage, and alerts the host and charge
controller that the battery has been depleted. The BAT_DEP threshold can be programmed through the Charger
Mode control (0x12) register through bits b9-b11. The three bits can program the threshold between 2.2-V per
cell to 2.9-V per cell, in 100-mV increments. The number of cells is determined by the programmed battery
regulation voltage (0x15) register. If the battery voltage falls below the BAT_DEP threshold, then the battery is
disconnected from the system and the adapter is connected to the system. The Battery Voltage Low bit (b11) is
set in the Charger Status (0x13) register, and the ALARM pin is pulled low to alert the host. There is a 1 second
deglitch time to prevent false triggering.
The Charger Status (0x13) register bits b6 and b7 also always indicate whether the battery is connected to the
system (b6), or the adapter is connected to the system (b7). An alarm is triggered whenever these states
change.
Asymmetrical gate drives (100 Ω turn-off; 10 kΩ turn-on) for the ACDRV and BATDRV drivers provide fast
turn-off and slow turn-on of the ACFET and BATFET to help the break-before-make logic and to allow a soft-start
at turn-on of either FET. The soft-start time is further increased by putting a capacitor from gate to source of the
p-channel power MOSFETs.
Input Overcurrent Protection (ACOC)
For solutions using the selector functions, an input overcurrent protection function (ACOC) is provided which
disconnects the ACFET by turning off the ACDRV pin, whenever the sensed input current exceeds the
programmed ACOC threshold. The ACOC threshold is programmed through the SMBus Charge Mode (0x12)
control register, bits b6, b7, b8. The ACOC function is automatically disabled upon power-on-reset. The host
needs to enable it by setting the ACOC bit (b6) HI. The ACOC threshold is set by the SET_ACOC bits (b7, b8),
to thresholds of 130%, 150%, 170%, or 190% of the input current (DPM) regulation limit threshold from the Input
Current (0x3F) register.
The ACFET turns off when the sensed current exceeds the threshold after a 200-µs deglitch time. The ACFET
automatically turns on after 2 ms, to limit the on-time duty cycle, and limit the power dissipation on the ACFET.
Care must be taken to ensure the system load power does not exceed the power-up allowable power when the
ACOC function is used.
The function is not intended for system short-circuits, as usually the adapter self protects. Instead, the function is
intended for long-term overcurrent protection of the selector power devices, and to limit start-up peak current.
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Bypass FET
The BYPASS pin is used to control an input FET that is off to prevent reverse discharge from the battery to the
adapter, and is on during input current draw to the system or battery, to minimize the power dissipation, as
compared to using a Schottky diode. If no adapter is detected, the BYPASS FET is off, by setting the BYPASS
pin to the PVCC pin. When the adapter is detected there is a 500-ms delay, then an ACP-to-BAT voltage
comparator is used to control the BYPASS pin. The BYPASS driver is set to the PVCC pin voltage when the
adapter voltage (ACP pin) is not more than 250 mV (ACP voltage falling-edge) above the battery voltage (BAT
pin), in order to turn off the external BYPASS p-channel power MOSFET. There is a 50 mV (ACP voltage
rising-edge) hysteresis, to protect from noise and prevent chatter. When adapter is detected and the ACP pin
voltage is greater than 300 mV above the BAT pin voltage, the BYPASS pin voltage is set to PVCC pin voltage
minus 6 V, in order to turn on the external BYPASS p-channel power MOSFET.
The ACP-to-BAT comparator also prevents the battery voltage from holding-up the ACDET sensed value and
falsely detecting ACDET when the adapter is removed, this prevents the system power selector from getting
stuck in an adapter always detected state. When ACP gets near to BAT, the external BYPASS p-channel power
MOSFET is turned off. This isolates the ACDET network from the battery, and allows the adapter input node to
discharge to PGND.
The BYPASS driver has a symmetrical gate drive of 1 kΩ turn-on and turn-off and does not need to be slowed
down.
Enabling Charge
Charge is only enabled 500 ms after adapter is detected. To initiate charge the CHGEN pin must be low, and the
Charger Mode (0x12) register END CHARGE bit (b0) must be set LO. The power-on-reset default for the END
CHARGE bit is HI, disabling charge.
The Charger Status (0x13) register NOT READY TO CHARGE bit (b0) indicates whether the charger is ready to
charge. A HI indicates the charger is not ready to charge, while a LO indicates the charger is ready to charge.
The NOT READY TO CHARGE bit (b0) must be LO in order for the charger to be enabled. The conditions that
make the charger not ready to charge are: adapter not detected, 500-ms delay after adapter detected not over,
REGN voltage not up, or VREF5 voltage not up.
The Charger Status (0x13) register CHARGER NOT ON bit (b1) indicates whether the charger is not on (HI), or
the charger is on (LO). For charger to be on, the CHGEN pin must be low, the Charger Mode (0x13) register
END CHARGE bit (b0) must be LO, the IC junction temperature must be below the TSHUT threshold, over
current is not detected, Thermistor Sense (TS) indicated battery pack is within programmed permissible charge
temperature range.
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ACP
VCC I
(adapter) REG
+
-
1 kW
+
-
1V
ACN
+
-
ACSET
Adapter Current
DAC 6BIT
3.2 mV - 200 mV
CHARGE_ENZ
I_ACSET
VCC
SRP
I ( VCC/10)
VCC
RAMP
(VPP = VCC/10)
PWM
To Gate
DriveLogic
Q
1V
S
R
RAMP
EAO
1 kW
EAI
SRN
+
-
SRSET
Charge Current
DAC 6BIT
I_SRSET
3.2 mV - 200 mV
CHARGE_ENZ
Compensation
CLAMP
-
+
-
OSC
Q
I(BAT) REG
+
FBO
+
VCC
VCC
BAT
V(BAT) REG
+
-
1V
Voltage
DAC 7BIT
2 V - 2.4 V
20 mA
BAT
SMBus
Cells
Select
Divider
20 mA
CHARGE_EN_DG
Figure 57. PWM Control Logic
Converter Operation
The synchronous buck PWM converter uses a fixed frequency voltage mode with feed-forward control scheme. A
type III compensation network allows using ceramic output capacitors. The compensation input stage is
connected between the feedback output (FBO) pin and the error amplifier input (EAI) pin. The feedback
compensation stage is connected between the error amplifier input (EAI) pin and error amplifier output (EAO) pin.
An internal saw-tooth ramp is compared to the EAO pin error control signal to vary the duty-cycle of the
converter. The ramp height is one-tenth of the input adapter voltage making it always directly proportional to the
input adapter voltage. This cancels out any loop gain variation due to a change in input voltage, and simplifies
the loop compensation. The ramp is offset by 300 mV in order to allow zero percent duty-cycle, when the EAO
signal is below the ramp. The EAO signal is also allowed to exceed the saw-tooth ramp signal in order to get a
100% duty-cycle PWM request. Internal gate drive logic allows achieving 99.98% duty-cycle while ensuring the
N-channel upper device always has enough voltage to stay fully on. If the BTST pin to PH pin voltage falls below
4.5 V for more than 3 cycles, then the high-set n-channel power MOSFET is turned off and the low-side
n-channel power MOSFET is turned on to pull the PH node down and recharge the BTST capacitor. Then the
high-side driver returns to 100% duty-cycle operation until the voltage is detected to fall low again due to leakage
current discharging the BTST capacitor below the 4.5 V, and the reset pulse is reissued.
The fixed frequency oscillator keeps tight control of the switching frequency under all conditions of input voltage,
battery voltage, charge current, and temperature, simplifying output filter design and keeping it out of the audible
noise region. The switching frequency can be changed from 300 kHz to 500 kHz by the Charger Mode (0x12)
register PWM FS bit (b5) – a HI is 500 kHz, a LO is 300 kHz. The switching frequency is 300 kHs by default after
power-on-reset. Typical chargers use 300 kHz, but 500 kHz allows a smaller inductance value
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The charge current sense resistor should be placed with at least half or more of the total output capacitance
placed before the sense resistor contacting both sense resistor and the output inductor; and the other half or
remaining capacitance placed after the sense resistor. The output capacitance should be divided and placed
onto both sides of the charge current sense resistor. A ratio of 50:50 percent gives the best performance; but the
node in which the output inductor and sense resistor connect should have a minimum of 50% of the total
capacitance. This capacitance provides sufficient filtering to remove the switching noise and give better sense
accuracy. The type III compensation is already providing phase boost near the cross-over frequency, giving
sufficient phase margin.
ISYNSET
The ISYNSET pin is used to program the charge current threshold at which the charger changes from
nonsynchronous operation into synchronous operation. This prevents negative inductor current. Negative
inductor current may cause a boost effect in which the input voltage increases as power is transferred from the
battery to the input capacitors—this can lead to an overvoltage on the PVCC node and potentially cause some
damage to the system.
This programmable value allows setting the current threshold for any inductor current ripple, and avoiding
negative inductor current. The SYNP and SYNN pins are used to sense across the charge current sense resistor.
To program the threshold, a resistor is connected from the ISYNSET pin to AGND. The minimum synchronous
threshold should be set from the inductor current ripple to the full ripple current, where the inductor current ripple
is given by:
I(RIPPLE_MAX)
2
£ I(SYN) £ I(RIPPLE_MAX)
(1)
The ISYNSET pin is internally regulated to 1 V. When the R(SYNSET) resistor is connected to AGND, it sets an
ISYNSET current equal to 1 V/R(SYNSET). The ISYNSET current internally flows through a 500 Ω creating a
voltage at which the voltage across R(SENSE) is compared. The ISYN charge current threshold is the voltage
divided by the R(SENSE) sense resistor value. The R(SYNSET) resistor value is calculated by:
R(SYNSET) =
1 V x 500 W
I(SYN) x R(SENSE)
(2)
where ISYN is the charge current threshold at which the converter changes to synchronous operation.
ISYNC FET Charger Undercurrent
Comparator
SYNP
+
500W
−
ISYNC_LOWD
+
−
1V
CHARGE_ENZ
−
+
t_bo
SYNN
40ns
ISYNSET
R_ISYNSET
NOTE: Patent Pending
Figure 58. Synchronous to Nonsynchronous threshold, ISYNSET, Block – Charger Undercurrent
(prevents negative inductor current)
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Synchronous versus Nonsynchronous Operation
The charger operates in nonsynchronous mode when the sensed charge current is below the ISYNSET
programmed value. When above the ISYNSET programmed value, the charger operates in synchronous mode.
During synchronous mode, the low-side n-channel power MOSFET is on, when the high-side n-channel power
MOSFET is off. The internal gate drive logic ensures there is break-before-make switching to prevent
shoot-through currents. During the dead-time where both FETs are off, the back-diode of the low-side power
MOSFET conducts the inductor current. Having the low-side FET turn-on keeps the power dissipation low, and
allows safely charging at high currents. During Synchronous mode the inductor current is always flowing and
operates in Continuous Conduction Mode (CCM) creating a fixed two-pole system. During nonsynchronous
operation: after the high-side n-channel power MOSFET turns off, and after the break-before-make dead-time,
the low-side n-channel power MOSFET turns on for around 80 ns, then the low-side power MOSFET turns off
and stays off until the beginning of the next cycle, where the high-side power MOSFET is turned on again. The
80 ns low-side MOSFET on-time is done to ensure the bootstrap capacitor is always recharged and able to keep
the high-side power MOSFET on during the next cycle. This is important for battery chargers, where unlike
regular dc-dc converters, there is a battery load that maintains a voltage and can both source and sink current.
The 80 ns low-side pulse pulls the PH node (connection between high and low-side MOSFET) down, allowing
the bootstrap capacitor to recharge up to the REGN LDO value. After the 80 ns, the low-side MOSFET is kept off
to prevent negative inductor current from occurring. The inductor current is blocked by the off low-side MOSFET,
and the inductor current becomes discontinuous. This mode is called Discontinuous Conduction Mode (DCM).
During the DCM mode the loop response automatically changes and has a single pole system at which the pole
is proportional to the load current, because the converter does not sink current, and only the load provides a
current sink. This means at very low currents the loop response is slower, as there is less sinking current
available to discharge the output voltage.
At very low currents during nonsynchronous operation, there may be a small amount of negative inductor current
during the 80 ns recharge pulse. The charge should be low enough to be absorbed by the input capacitance.
Whenever the converter goes into zero percent duty-cycle, the high-side MOSFET does not turn-on, and the
low-side MOSFET does not turn-on (no 80 ns recharge pulse), so there is no discharge from the battery.
Battery Voltage Regulation Loop
The BAT pin is used to sense the battery voltage and should be connected as close to the battery as possible, or
directly to the output capacitor. A 0.1-µF ceramic capacitor from BAT to AGND is recommended—added as close
to the BAT pin as possible to decouple high frequency noise.
The voltage regulation feedback is through the BAT pin. This input is tied directly to the positive side of the
battery pack. The bq24721 monitors the battery-pack voltage between the BAT and VSS pins. The regulation
voltage is programmed through the SBS-like SMBus interface.
The voltage regulation DAC register input is decoded into an internal 7-bit DAC that programs the voltage on a
per-cell basis, then is multiplied by the number of cells. There are a total of 128 voltage steps with a 6.25 mV
step, giving a per cell range of [4 V – (4.8 V–6.25 mV)]. There are 128 steps in the 3-cell voltage range of
[12 V – (14.4 V–18.75 mV)]. There are 128 steps in the 4-cell voltage range of [16 V – (19.2 V–25 mV)].
Valid voltage values are 9 V, 12 V–14.381 V, and 16 V–19.175 V. The internal voltage DAC allows programming
to 9 V which is used for waking-up or closing the battery pack. A 9 V programmed voltage is interpreted as a
2-cell voltage by the BATDEP and BATSHORT thresholds. Step size for 3-cell battery programming voltage is
18.75 mV. The Charger interprets a 3-cell battery for any voltage between 12 V–14.4 V. Step size for 4-cell
battery programming voltage is 25 mV. The charger interprets a 4-cell battery for any voltage programmed
between 16 V–19.2 V.
Invalid DAC voltages are indicated by the VOR (voltage out of range) bit of the status register. Voltages below
12 V (except for 9 V) are out of range and keep the converter disabled. Voltages between 14.4 V–16 V (including
14.4 V, but not including 16 V) are out of range and keep the converter disabled. Voltages above 19.2 V
(including 19.2 V) are out of range and allow the converter to charge, but the voltage is always set to the
maximum allowable voltage of 19.175 V = (19.2 V–25 mV).
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Battery Charge Current Regulation Loop
The battery charge current DAC is set for a 10-mΩ sense resistor; however, resistors of other values can also be
used. The larger the sense resistance, the larger the sensed voltage, and the higher the regulation accuracy, but
at the expense of higher conduction losses. The SRP and SRN pins are used to sense across the sense resistor.
The battery charge current, IO(CHARGE), is established by setting the external sense resistor, R(SNS_CHG), and the
SMBus charge current DAC (0 × 14). In order to set the current, first R(SNS_CHG) should be chosen based on the
regulation threshold V(IREG_CHG), across this resistor. The listed SBS current corresponds to a 10-mΩ sense
resistor.
R(SNS_CHG) = V(IREG_CHG) / IO(CHARGE)
Input Current Regulation Loop (DPM)
The ACP and ACN pins are used to sense across the sense resistor. The input current DAC is set for a 10-mΩ
sense resistor; however, resistors of other values can also be used. The larger the sense resistance, the larger
the sensed voltage, and the higher the regulation accuracy, but at the expense of higher conduction losses.
The input current, II(DPM), is established by setting the external sense resistor, R(SNS_DPM), and the SMBus input
current DAC (0 × 3F). In order to set the current, first R(SNS_DPM) is chosen based on the regulation threshold
V(IREG_DPM), across this resistor. The listed SBS current corresponds to a 10-mΩ sense resistor.
R(SNS_DPM) = V(IREG_DPM) / IO(CHARGE)
Automatic Internal Soft-Start Charger Current
The charger automatically soft-starts the charger regulation current every time the charger is enabled, in order to
ensure there is no overshoot or stress on the output capacitors or the power converter. The soft-start consists of
stepping-up the charge regulation current into eight evenly divided steps up to the programmed charge current of
register (0x14). Duration of each step is around 1 ms, for a typical rise time of 8 ms. No external components are
needed for this function.
The bq24721 current regulation loop reference steps-up whenever charge is enabled, and when returning from
fault/suspend mode into charge where the current regulator is turned on. The loop should take control within a
few hundred micro-seconds with very little overshoot due to the LC output filter and the high compensation loop
bandwidth with 300 kHz or 500 kHz operating frequency; therefore, the reference could ramp up from precharge
to fast-charge within 50 µs to 500 ms. Going into fault/suspend mode, short circuit (V(BAT) < V(UVT)), Sleepmode
(V(ACP) < V(BAT)), or UVLO (VCC < 3.7 V) initiates an immediate shut-off of the high-side PWM FET by setting its
gate to V(PH). The output inductor and battery load determines the ramp-down rate as it freewheels through the
Schottky diode.
High Accuracy Current Sense Amplifiers (CSA), IOUT pin, for Input Current and Charge Current
Industry standard, high accuracy current sense amplifiers (CSA) can be used to monitor the input current or the
charge current by the host or some discrete logic through the analog voltage output of the IOUT pin. The current
sense amplifier from the input current and the current sense amplifier of the charge current (voltage across
SRP-SRN pins) amplifies the input sensed voltage by 20x, through the Iout pin. The IOUT output is selectable
between the input current or the charge current, through a multiplexor that is controlled by the Charge Mode
(0x12) control register, IOUT Select bit (b3). The default setting is LO selecting the input current CSA.
Programming a HI selects the charger current CSA.
The IOUT output is a voltage source 20 times the input differential voltage. If the user wants to lower the voltage,
use a sense resistor from IOUT to AGND, and still achieve accuracy overtemperature as the resistors match their
thermal coefficients.
A 0.1-µF capacitor connected on the output is recommended for decoupling high-frequency noise. An additional
RC filter is optional, if additional filtering is desired. Note that adding filtering also adds additional response delay.
Charger Overcurrent Protection
The charger has a secondary overcurrent protection function that monitors the charge current, and prevents it
from exceeding 200% of the programmed charge current. The high-side gate drive turns off and automatically
resume when the current falls below the overcurrent threshold
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Current Regulation Down to Zero Battery Voltage
The bq24721 charger regulates charge current and input current down to zero volts on the battery BAT voltage.
If there is a drop below 2 V, then the converter immediately turns off both high-side and low-side FETs to stop
current flow, then resumes regulating the current. This ensures there is no overcurrent surge that could cause
damage to the battery, charger, or system.
Thermal Shutdown Protection
The QFN package has low thermal impedance which provides good thermal conduction from the silicon to the
ambient, to keep junctions temperatures low. As added level of protection, the charger converter turns off and
self-protects whenever the junction temperature exceeds the T(SHUT) threshold of 145°C. The charger stays off
until the junction temperature falls below 130°C.
Battery Short-Circuit or Low Condition
The number of cells determines the value for the BATDEP threshold and for the BATSHORT threshold, as these
values are programmed on a per-cell basis. The programmed regulation voltage determines the number of cells.
The battery depleted (BATDEP) threshold is programmed by the control register bits b9-b11. The three bit dac
sets the voltage between 2.2 V to 2.9 V per cell at 100 mV increments. The BATSHORT threshold is 1.7 V/Cell
falling entering a shorted condition, and 1.9 V/Cell rising leaving shorted condition, and entering normal
condition. BATSHORT has a 1 second deglitch on both edge directions to protect from transient conditions, and
to allow closing deeply discharge battery packs.
The PWM duty-cycle immediately resets to zero percent when the battery voltage is sensed to drop below 2 V,
then the regulation loop allows the duty-cycle to settle in the current regulation value, C. After a 1 second
deglitch, the converter regulates battery current to C/8 when the battery voltage falls below 1.7 V/cell. The
converter regulates back at C after a 1 second deglitch from the time the battery voltage rises above 1.9 V/cell.
Charge Termination for Li-Ion or Li-Polymer
The primary termination method for Li-Ion and Li-Polymer is minimum current. Secondary temperature
termination methods is also provided for additional safety. The host controls the charge initiation and the
termination. A battery pack gas gauge assists the hosts on setting the voltages and determining when to
terminate based on the battery pack state of charge.
Temperature Qualification (TS pin)
The bq24721 continuously monitors battery temperature by measuring the voltage between the TS pin and
AGND. A negative temperature coefficient thermistor (NTC) and an external voltage divider typically develop this
voltage. The bq24721 compares this voltage against its internal thresholds to determine if charging is allowed. To
initiate a charge cycle, the battery temperature must be within the V(LTF) to V(HTF) thresholds. If battery
temperature is outside of this range, the bq24721 suspends charge and waits until the battery temperature is
within the V(LTF) to V(HTF) range. During the charge cycle (both precharge and fast charge) the battery
temperature must be within the V(LTF) to V(TCO) thresholds. If battery temperature is outside of this range, the
bq24721 suspends charge and waits until the battery temperature is within the V(LTF) to V(HTF) range. The
bq24721 suspends charge by turning off the PWM charge FETs. Figure 59 summarizes this operation.
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VREF5
VREF5
V(TSDET)
V(TSDET)
Charge Suspend
Charge Suspend
V(LTF)
V(LTFH)
V(LTF)
V(LTFH)
Temperature
range to initiate
charge
Temperature
range during a
charge cycle
Charge Suspend
Charge Suspend
V(HTF)
V(TCO)
AGND
AGND
Figure 59.
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SBS-Like SMBus
SBS-Like interface is not 100% SBS compliant, as control and status registers were changed to simplify and
enhance the charger control device. Address 0x12 is used, but the charger should not to be used on systems
requiring 100% SBS compliance. Effort was taken to use the same commands when possible. Comparison
included on SBS-like vs SBS Spec.
• SBS-Like SMBus
– SBS 1.1 protocol, slave operation only
– No CRC
– Address: 12 (Note SBS-Like is not 100% SBS compliant and has simplified enhancements
• Smart charger commands implemented
– ChargingCurrent (), with modifications on data byte contents, SBS-Like SMBus
– ChargingVoltage (), with modifications on data byte contents, SBS-Like SMBus
– ChargerStatus (), with modifications on status bits
• Smart charger commands not implemented
– ChargerSpecInfo (),
– AlarmWarning ()
SBS-Like SMBus: Commands Implemented: Charge Current DAC Register
Charging Current, (0x14), Write/Read (1)
Table 1.
BIT
(1)
SBS SPEC
bq24721 SBS-Like
0 (LSB)
1 mA
NA
1
2 mA
NA
2
4 mA
NA
3
8 mA
NA
4
16 mA
NA
5
32 mA
NA
6
64 mA
NA
7
128 mA
128 mA (1.28 mV)
8
256 mA
256 mA (2.56 mV)
9
512 mA
512 mA (5.12 mV)
10
1024 mA
1024 mA (10.24 mV)
11
2048 mA
2048 mA (20.48 mV)
12
4096 mA
4096 mA (40.96 mV)
13
8192 mA
8192 mA (81.92 mV)
14
16384 mA
NA
15 (MSB)
32768 mA
NA
NOTES
Charge current limit setting
16384 mA max
(162.56 mV max)
Shunt drop voltage shown at LO/HI bit state;
DAC current value is based on using 10-mΩ sense resistor.
Max charge current with 10-mΩ sense resistor is 16.256 A.
Bit 13 allows using larger sense resistors for increasing accuracy in low charge current applications.
• Valid charge currents are 0 A–16.256 A using a 10-mΩ sense resistor. Minimum step is 128 mA.
• Programmed charge currents above max 16.256 A → Charger is enabled and the current regulation value is
set to the maximum 16.256 A.
• Programmed charge current = 0 A → Charger is disabled.
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SBS-Like SMBus: Commands Implemented: Input Current DAC Register
Input Current, (0x3F), Write/Read (1)
BIT
(1)
SBS SPEC
bq24721 SBS-Like
0 (LSB)
1 mA
NA
1
2 mA
NA
2
4 mA
NA
3
8 mA
NA
4
16 mA
NA
5
32 mA
NA
6
64 mA
NA
7
128 mA
128 mA (1.28 mV)
8
256 mA
256 mA (2.56 mV)
9
512 mA
512 mA (5.12 mV)
10
1024 mA
1024 mA (10.24 mV)
11
2048 mA
2048 mA (20.48 mV)
12
4096 mA
4096 mA (40.96 mV)
13
8192 mA
8192 mA (81.92 mV)
14
16384 mA
NA
15 (MSB)
32768 mA
NA
NOTES
Input DPM current limit setting
16384 mA max
(162.56 mV max)
Shunt drop voltage shown at LO/HI bit state;
DAC current value is based on using 10-mΩ sense resistor.
Max input current with 10-mΩ sense resistor is 16.256 A.
Bit 13 allows using larger sense resistors for increasing accuracy in low input current applications.
• Valid input currents are 0 A–16.256 A using a 10-mΩ sense resistor. Minimum step is 128 mA.
• Programmed input currents above max 16.256 A → Charger is enabled and the current regulation value is set
to the maximum 16.256 A.
• Programmed input current = 0 A → Charger is disabled.
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SBS-Like SMBus: SBS Translator Voltage DAC Commands Implemented
Charging Voltage, (0x15), Write/Read
BIT
(1)
bq24721 SBS-Like
CHARGE VOLTAGE LOOK-UP TABLE
SBS SPEC
0 (LSB)
1 mV
NA
1
2 mV
0 / 2 mV
2
4 mV
0 / 4 mV
3
8 mV
0 / 8 mV
4
16 mV
0 / 16 mV
5
32 mV
0 / 32 mV
6
64 mV
0 / 64 mV
NOTES
Not used
7
128 mV
0 / 128 mV
8
256 mV
0 / 256 mV
9
512 mV
0 / 512 mV
10
1024 mV
0 / 1024 mV
11
2048 mV
0 / 2048 mV
12
4096 mV
0 / 4096 mV
13
8192 mV
0 / 8192 mV
14
16384 mV
0 / 16384 mV
15 (MSB)
32768 mV
NA
See Notes below for valid voltage range
and minimum step size (resolution (1))
Not used
Battery voltage range: 4 V–4.8 V per cell
3 Cell range: 12 V to (12 V–14.4 V); step size = 18.75 mV
4 Cell range: 16 V to (16 V –19.2 V) ; step size = 25 mV
a. Valid voltages are 9 V, 12 V–14.4 V, and 16 V–19.2 V. Intermediate programmed voltages between Internal DAC values are
truncated to the lower value to avoid overvoltage on battery.
b. Programmed Voltages above max 19.2 V → Charger is enabled and the voltage regulation value is set to the maximum 19.2 V.
c. Programmed voltages below 12 V (except 9 V), and between 14.4 V–16 V → Charger is disabled, VOR bit is triggered, and an alarm
is sent to the host.
d. Programmed 9 V (9000 mV = 2328 Hex) → Charger regulates to 9-V output for battery wakeup.
SBS-Like SMBus: IC Internal Voltage DAC for Reference Only
Internal decoder translates the SBS voltage register value into the internal closest value. Values are truncated to the lower
valid setting.
Charging Voltage, IC Internal Only
BIT
0 (LSB)
0/18.75 mV
0/25 mV
1
0 / 37.5 mV
0 / 50 mV
2
0 / 75 mV
0 / 100 mV
3
0 / 150 mV
0 / 200 mV
4
0 / 300 mV
0 / 400 mV
5
0 / 600 mV
0 / 800 mV
0 / 1.2 V
0 / 1.6 V
LO – 12 V
HI – 16 V
6 (MSB)
(1)
38
bq24721 SBS-Like
CHARGE VOLTAGE INTERNAL DECODER DAC
NOTES
Offset that can be added to dc level
See the following notes for valid voltage range and
minimum step size (resolution) (1)
3 Cell or 4 Cell dc level select
Battery voltage range: 4 V–4.8 V per cell
3 Cell range: 12 V to (12 V–14.4 V); step size = 18.75 mV
4 Cell range: 16 V to (16 V –19.2 V) ; step size = 25 mV
a. Valid voltages are 9 V, 12 V–14.4 V, and 16 V–19.2 V. Intermediate programmed voltages between Internal DAC values are
truncated to the lower value to avoid overvoltage on battery.
b. Programmed Voltages above max 19.2 V → Charger is enabled and the voltage regulation value is set to the maximum 19.2 V.
c. Programmed voltages below 12 V (except 9 V), and between 14.4 V–16 V → Charger is disabled, VOR bit is triggered, and an alarm
is sent to the host.
d. Programmed 9 V (9000 mV = 2328 Hex) → Charger regulates to 9 V output for battery wake-up.
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Programmed Voltage Value Difference Between Li-Ion and NiMH Battery Pack Charging
•
Li-Ion based battery packs require both current regulation and voltage regulation loops – where
constant-current and constant voltage is needed to fully charge the battery pack.
NiMH batteries require only a constant current for charging. Thereby, the voltage regulation loop is not
needed. To accomplish this, set the regulation voltage at a value greater than the maximum battery pack
voltage. This does not allow the voltage regulation loop to become active. The maximum value of 19.2 V is
set for most NiMH battery packs.
•
SBS-Like SMBus: Commands Implemented: Control Register
Charger Mode, (0x12), Write-Only [Default POR and Reset Value in BOLD TYPE]
BIT
SBS SPEC
bq24721 SBS-Like
END CHARGE
0 (LSB)
INHIBIT CHARGE
(1)
1: Disable Charger
0: Enable Charger
This affects STATUS register 0x13, bit 1.
1
ENABLE POLLING
RESERVED: Not Used. Bit always resets to zero.
2
POR_RESET
RESET: (Return to POR values) Charger disabled, DPM active, PWM at 300 kHz, VO = 0,
CHARGE CURR = 0, IDPM = 0
1: Reset
0: No Reset
IOUT Select: Current Sense Amplifier Output select
1: Charge current is the 20x amplifier output at IOUT pin
3
RESET_TO_ZERO
0: Adapter current is the 20x amplifier output at IOUT pin
The IOUT pin is multiplexed to either the charge current or the input current sense amplifiers. The
Input current (ac) is the default upon power up.
LEARN CYCLE
4
RESERVED
1: Connect Battery to system (learn cycle) even when adapter connected
0: Connect AC adapter to system (no learn) when adapter connected
This bit affects STATUS register 0x13, bits 6 and 7.
5
RESERVED
6
RESERVED
7
RESERVED
8
RESERVED
PWM FS: Switching Frequency Select
1: PWM at 500 kHz
0: PWM at 300 kHz
ACOC – AC (adapter) Overcurrent
1: ACOC protection enabled
0: ACOC protection disabled
SET_ACOC
00: ACOC=DPM × 1.3
01: ACOC=DPM × 1.5
10: ACOC=DPM × 1.7
11: ACOC=DPM × 1.9
Bit 6 of this Control register 0×12 needs to be high for these bits to take effect.
ACOC threshold is a percentage of the DPM (input current) regulation threshold programmed by
the DPM register 0×3F.
9
RESERVED
SET_BAT_DEPL Battery depleted threshold
10
RESERVED
Value set per cell
11
RESERVED
000: [2.2V], 001:[2.3V], 010:[2.4V], 011:[2.5V]
100:[2.6V], 101:[2.7V], 110:[2.8V], 111:[2.9V]
These bits affect STATUS register 0×13, bits 11, 6 and 7.
12
RESERVED
RESERVED: Not Used. Bit always resets to zero.
13–15(MSB)
RESERVED
RESERVED: Not Used. Bit always resets to zero.
(1)
END_CHARGE bit must be LO AND the CHRGEN pin must be LO to enable the charger. If the END_CHARGE bit is HI or the
CHRGEN pin is HI, then the charger is disabled.
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SBS-Like SMBus: Commands Implemented: Status Register
Charger Status, (0x13), Read-Only
All Status register bits are automatically updated by the IC whenever the system state changes, some bits have
an 8 ms deglitch to prevent false alerts. Whenever a status bit changes, the ALARM open drain output pulls
down to alert the host of a change in the status register. The ALARM stays low until the host performs a
STATUS register (0×13) read, afterward, the ALARM clears and goes high impedance. An external pull-up
resistor is needed at the ALARM pin.
BIT
SBS SPEC
bq24721 SBS-Like
NOT READY TO CHARGE
0 (LSB)
CHARGE_INHIBITED
1=
Charger is not ready to charge (IC comes up in this state)
0=
Charger is ready to charge
(Note: Same logic polarity as SBS spec, but added Feature, Not part of SBS spec.) This bit indicates
all the conditions are met to allow the charger to start charging. Precursers are Power-on-Reset is
completed; adapter is detected, and REGN driver rail is ready.
CHARGER NOT ON
1
MASTER MODE
1=
Charger is Not On (IC comes up in this state)
0=
Charger is On
(Note: same logic polarity and different bit, from SBS spec bit 0) Hi on this bit indicates that the charger
is on . If Low, the charger is off. This bit allows the user to determine if there is a fault that disable the
charger. Bits in this STATUS register 0x13 are used in conjunction with this bit to determine the source
of charger turning off.
CHARGE VOLTAGE LOOP NOT ACTIVE
2
VOLTAGE_NOTREG
1=
Charge Voltage Loop Not Active (IC comes up in this state, and in this state when charger is not
enabled)
0=
Charge Voltage Loop Active
(Note: same logic polarity as SBS spec bit 2). A Lo on this bit indicates the battery voltage regulation
loop is active and is influencing the charger regulation. There is an overlap as one loop transitions to
another loop to allow for a soft transition knee. This overlap alerts that another loop is getting close to
its limit and is influencing the output.
CHARGE CURRENT LOOP NOT ACTIVE
3
CURRENT_NOTREG
1=
Charge Current Loop Not Active (IC comes up in this state, and in this state when charger is not
enabled)
0=
Charge Current Loop Active
(Note: same logic polarity from SBS spec bit 3) A LO on this bit indicates the charge current regulation
loop is active and is influencing the charger regulation. There is an overlap as one loop transitions to
another loop to allow for a soft transition knee. This overlap alerts that another loop is getting close to
its limit and is influencing the output.
DPM INPUT CURRENT LOOP NOT ACTIVE
4
LEVEL2
1=
DPM Input Current Loop Not Active (IC comes up in this state, and in this state when charger is
not enabled)
0=
DPM Input Current Loop Active
(Note: Added Feature, Not part of SBS spec. Same logic polarity from bits 2 and 3 of SBS spec)
Dynamic Power Management (DPM) dynamically reduces the charge current when the system current
increases enough to cause the total input current to reach the input power limit. This gives power
delivery precedence to the system current over the charge current. A LO on this bit indicates the input
current regulation loop is active and is influencing the charger regulation. There is an overlap as one
loop transitions to another loop to allow for a soft transition knee. This overlap can alert that another
loop is getting close to its limit and is influencing the output.
CHARGE OVERCURRENT FAULT
(Note: Added Feature, Not part of SBS spec.)
5
40
LEVEL3
HI indicates the charge current was higher than the charge overcurrent limit (SROC) which is 1.7 times
the charge regulation setting. This is to protect from shorts or large load transients at the output. If
SROC is detected, the high-side FET is immediately turned-off, and an alarm is sent out. The charger
retries again, allowing the high-side FET to turn-on the next cycle. This fault current limit continues until
the fault is removed.
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BIT
SBS SPEC
bq24721 SBS-Like
BATTERY CONNECTED TO SYSTEM (by selector BATDRV)
(Note: Added Feature, Not part of SBS spec.)
HI indicates the battery is connected to the system output. LO indicates battery is not connected.
During this HI state, the BATDRV pin is pulled 6V below the SYS pin voltage to turn on the external
PMOS BAT FET. When in LO state, the BATDRV pin voltage is pulled-up to the SYS pin voltage.
6
CURRENT_OR
Battery is connected to system when ac is not present, or when ac is present and LEARN CYCLE is
set at the control register.
FEATURE: If in learn cycle and the battery falls below the BATDEP threshold (programmed by the
control register), then the IC automatically switches from battery power to ac adapter power.
FEATURE: Also, in all cases if system (SYS) voltage is more than 150 mV above the battery (BAT)
voltage then the BAT FET is kept off until SYS discharges to within 150 mV to prevent the system
capacitors from discharging into the battery. This feature protects the battery from extended high surge
currents.
ADAPTER CONNECTED TO SYSTEM (by selector ACDRV)
(Note: Added Feature, Not part of SBS spec.)
HI indicates the adapter is connected to the system output. LO indicates adapter is not connected.
During this HI state, the ACDRV pin is pulled 6 V below the PVCC pin voltage to turn on the external
PMOS AC FET. When in LO state, the ACDRV pin voltage is pulled-up to the PVCC pin voltage.
7
VOLTAGE_OR
AC is connected to system when adapter is present (ACDET voltage is above threshold) and LEARN
CYCLE is not set at the control register.
FEATURE: If in learn cycle and the battery falls below the BATDEP threshold (programmed by the
control register), then the IC automatically switches from battery power to ac adapter power.
FEATURE: If there is an adapter overcurrent (ACOC) is detected, then the AC FET is turned off
immediately to prevent a short-circuit condition or a surge current. This feature protects the battery
from extended high surge currents.
IC THERMAL SHUTDOWN FAULT
(Note: Added Feature, Not part of SBS spec.)
8
THERMISTOR_OR
A HI indicates the IC temperature is too high, and had to turn-off the charger. LO indicates temperature
is safe.
FAULT DETECTION AND PROTECTION FEATURE: The IC monitors its own junction temperature. If
the IC junction temperature exceeds 145°C, then the IC automatically shuts down the charger and an
ALARM is triggered. SMBus communications continues to function. The charger remains off until the IC
junction temperature falls below 130°C. The charger automatically restarts once the temperature falls
below the hysteresis at 130°C.
[THERMISTOR COLD
9
THERMISTOR_COLD
HI indicates the thermistor temperature is too cold to allow charging. The TS pin voltage threshold is
higher than the TS cold voltage threshold when this bit is HI. A LO means the TS voltage is below the
TS cold voltage threshold. Programmed by a voltage divider pulled up to the VREF5 pin.
Note: VREF5 only comes up when ac is detected to conserve battery power, since no charging would
be required.
THERMISTOR HOT (start) or (operating)
10
THERMISTOR_HOT
HI indicates the thermistor temperature is too hot to allow charging. The TS pin voltage threshold is
lower than the TS hot voltage threshold when this bit is HI. A LO means the TS voltage is above the
TS hot voltage threshold. There is a different voltage threshold before starting charge than during
charging (operating). Programmed by a voltage divider pulled up to the VREF5 pin.
Note: VREF5 only comes up when AC is detected to conserve battery power, since no charging is
required.
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BIT
SBS SPEC
bq24721 SBS-Like
BATTERY VOLTAGE LOW (battery depleted)
(Note: Added Feature, Not part of SBS spec.)
11
THERMISTOR_UR
This bit goes HI when the sensed BAT voltage is below the Battery Depleted threshold programmed by
the CONTROL register 0x12, bits 9, 10, and 11. The programmed voltage is per cell; therefore, multiply
by the number of cells set by the programmed battery voltage regulation DAC, register 0x15. [3 cell for
voltages between 12 V-14.4 V; and 4 cell for voltages between 16 V-19.2 V].
FEATURE: the BATDRV is turned off when the battery voltage falls below the BATDEP threshold, and
the AC DRV is turned on if adapter is connected, and had previously been in LEARN Cycle (bit 4 of
CONTROL register, 0x12).
VOR, VOLTAGE OUT OF RANGE (Note: moved function from bit7 of SBS spec) A HI indicates the
voltage DAC value from register 0x15 is out of the valid DAC voltage range. The IC either disables the
charge if below, or stays at the maximum possible programmed voltage (19.2 V) according to the
following. Valid voltage DAC range and effect.
12
ALARM_INHIBITED
• Valid voltages are 9 V, 12 V –14.4 V, and 16 V–19.2 V. Intermediate programmed voltages
between Internal DAC values are truncated to the lower value to avoid overvoltage on battery.
• Programmed Voltages above max 19.2 V → Charger is enabled and the voltage regulation value
is set to the maximum 19.2 V.
• Programmed voltages below 12 V (except 9 V), and between 14.4 V–16 V → Charger is
disabled, VOR bit is triggered, and an alarm sent to host.
• Programmed 9 V (9000 mV = 2328 Hex) → Charger regulates to 9-V output for battery wake-up.
ADAPTER VOLTAGE BELOW BATTERY PACK VOLTAGE
13
POWER_FAIL
(Note: Added Feature, Not part of SBS spec.) A HI on this bit indicates the battery voltage is above the
adapter voltage (BAT pin > ACP pin).
FEATURE: If the input falls below 150 mV above the Battery voltage, the BYPASS FET immediately
turns off to prevent reverse discharge of the battery into the adapter. There is an 8 ms deglitch and
100 mV to hysteresis before allowing the BYPASS FET to turn-on.
BATTERY DETECTED ( through thermistor pin)
FEATURE: HI indicates a battery is detected through the TS pin. The TS pin voltage threshold is lower
than the TS battery detect threshold when this bit is HI. A LO means the TS voltage is above the TS
battery detect threshold. A resistor to ground inside the battery is used to distinguish a good battery
from a bad battery as well. Programmed by a voltage divider pulled up to the VREF5 pin.
14
BATTERY PRESENT
Note: VREF5 only comes up when ac is detected to conserve battery power, since no charging is
required.
(Note: Added Feature. This is used for detecting proper battery connected by selecting the proper valid
resistance. This is need to be ORed for dual battery pack chargers to make feature optimal. This
feature also is used as a safety feature to stop charge immediately when the battery pack is not
detected – this allows a faster response time by autonomously terminating charge when no battery is
detected, as opposed to waiting until the host detects the battery is removed and then alerting the
charger.
AC ADAPTOR DETECTED
15
(MSB)
AC_PRESENT
This bit is HI when the adapter is detected (ACDET pin voltage is above the acdetect threshold of
2.4V). The voltage can be programmed by an external resistor divider from adapter input to ground.
Charge is not allowed until the ACDET threshold is exceeded. VREF5 LDO output only turns on when
the ACDET threshold is exceeded (adapter detected).
REGN POR and SMBus communication begins soon after the ACDET threshold is above 1.2 V and
the VCC is above 6.5 V.
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ELECTRICAL CHARACTERISTICS
7 Vdc ≤ V(VCC) ≤ 24 Vdc, –20°C < TJ < 125°C, ref = AGND (unless otherwise noted).
PARAMETER
MIN
TYP
MAX
UNIT
SMBus TIMING CHARACTERISTICS
tR
SCLK/SDATA rise time
1
µs
tF
SCLK/SDATA fall time
tW(H)
SCLK pulse width high
4
tW(L)
SCLK Pulse Width Low
4.7
µs
tSU(STA)
Setup time for START condition
4.7
µs
tH(STA)
START condition hold time after which first clock pulse is generated
4
µs
tSU(DAT)
Data setup time
250
ns
tH(DAT)
Data hold time
300
ns
tSU(STOP)
Setup time for STOP condition
4
µs
t(BUF)
Bus free time between START and STOP condition
4.7
FS(CL)
Clock Frequency
10
100
kHz
6
10
ms
35
ms
300
ns
50
µs
µs
ALARM LOGIC TIMING CHARACTERISTICS
Time delay from status register bit toggling to ALERT: LO → HI transition
tALM
HOST COMMUNICATION FAILURE
ttimeout
SMBus bus release timeout
tBOOT
Deglitch for watchdog reset signal
25
tWDI
Watchdog timeout period
10
140
170
ms
200
s
0.4
V
OUTPUT BUFFER CHARACTERISTICS
V(SDAL)
Output LO voltage at SDA, I(SDA) = 3 mA
tw(H)
tsu(STA)
tw(L)
tf
tr
SCL
tf
tr
SDA
START
th(STA)
th(DAT)
th(DAT)
STOP
tsu(DAT)
SCL
1
2
7
3
8
SDA
ACK
START
SCL
9
1
tsu(STOP)
7
3
2
8
SDA
9
ACK
t(BUF)
STOP
Figure 60.
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FUNCTIONAL DESCRIPTION
SMBus Overview
An SMBus communication port provides a simple way for an SMBus compatible host to access system status
information and reset fault modes. Functioning as a SLAVE port enables SMBus compatible hosts to WRITE to
internal registers or READ from internal registers. The bq24721 SMBus port is a 2-wire bidirectional interface
using SCL (clock) and SDA (data) pins; the SDA pin is open drain and requires an external pull-up. The SMBus
is designed to operate at SCL frequencies up to 100 kHz. The standard 8 bit command is supported, the CMD
part of the sequence is the 8 bit register address which is READ from or WRITE to. The bq24721 does not
support packet error correction, PEC, as a mechanism to confirm proper communication between it and the host.
SMBus Address
The SMBus specification contains several global addresses, to which the slaves on the bus are required to
respond. The bq24721 responds to the SBS charger addresses of 0×12 for writes and 0×13 for reads. The
bq24721 only responds (ACK) to the above listed addresses, and does (NACK) not respond to any other
address.
BIT
BYTE
MSB
6
5
4
3
2
1
LSB
bq24721 SMBus WRITE ADDRESS SBS Charger
0
0
0
1
0
0
1
0
bq24721 SMBus READ ADDRESS SBS Charger
0
0
0
1
0
0
1
1
COMMAND
0
0
0
1
C3
C2
C1
C0
D7
D6
D5
D4
D3
D2
D1
D0
I/O DATA BUS
Internal Register Map
The status data and control data is referenced by the following Commands.
COMMAND
R/W
0x12
R/W
REGISTER DESCRIPTION
Charger Mode
0x13
R
Charger Status
0x14
R/W
Charging Current
0x15
R/W
Charging Voltage
0x3F
R/W
Input Current
SMBus Bus Release
The bq24721 SMBus engine does not create START or STOP states on the SMBus bus during normal
operation. However, if a 2 second (typical) SDA low timeout is exceeded the bq24721 releases the bus, thus
creating a stop condition.
SMBus Communication Protocol
The following conventions is used when describing the communication protocol.
CONDITION
CODE
START sent from host
S
STOP sent from host
P
bq24721 SMBus slave address sent from host, bus direction set from host to bq24721 (WRITE)
bq24721 register address sent from bq24721, bus direction is from bq24721 to host (READ)
Non-valid SMBus slave address sent from host
hA0
hA1
hA_N
Valid bq24721 register address sent from host
HCMD
Nonvalid bq24721 register address sent from host
HCMD_N
I/O data byte (8 bits) sent from host to bq24721
hDATA
I/O data byte (8 bits) sent from bq24721 to host
bqDATA
Acknowledge (ACK) from host
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CONDITION
CODE
Not acknowledge (NACK) from host
hN
Acknowledge (ACK) from bq24721
bqA
Not acknowledge (NACK) from bq24721
bqN
Repeated Start
R
STOP
CONDITION
(P)
START
CONDITION
(S)
BIT 7
MSB
BIT 6
BIT 0
LSB
STOP
CONDITION
(P)
START
CONDITION
(S)
BIT 7
MSB
BIT 6
BIT 0
LSB
ACKNOWLEDGE
(hAor bqA)
STOP
CONDITION
(P)
SCL
SDA
NOT
ACKNOWLEDGE
(hN or bqN)
STOP
CONDITION
(P)
SCL
SDA
Figure 61.
SMBus Read/Write Sequences
The bq24721 supports the standard SMBus Word Write, as well as the SMBus Word READ. The basic SMBus
word read protocol has the following steps:
1. Host sends a start and bq24721 SMBus slave write address
2. bq24721 ACKs that this is a valid SMBus address and that the bus is configured for write
3. Host sends bq24721 command
4. bq24721 ACKs that this is a valid command and stores the command for a possible read
5. Host sends a repeated start and bq24721 SMBus slave read address, reconfiguring the bus for read
6. bq24721 ACKs that this is a valid address and that bus is reconfigured
7. Bus is in read mode, bq24721 starts sending 2 bytes of data chosen by the command.
The SMBus write protocol is similar to the read, without the need for a repeated start and bus being set in write
mode. If the address sent by host is not a valid address , the command is NACKed. The host can complete a
READ or a WRITE sequence with either a STOP or a START. In a WRITE it is not necessary to end each word
WRITE command with a STOP, a START has the same effect (repeated start).
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Valid Write Sequences
The bq24721 always ACKs its own address. If the CMD points to an allowable READ or WRITE data bq24721
writes the command into its command register and send an ACK. If the CMD points to an nonallowed command
bq24721 does NOT write the command into its command register and sends a NACK.
S
hA0
S
hA0
bqA
bqA
hCMD_N
bqN
Word Write
The data is written to a control register at the end of the ACK after the second byte in the sequence. If an word
write sequence is intrupted by a STOP or START, no data is written to the device. The host can cancel a WRITE
by sending a STOP or START before the trailing edge of ACK clock pulse.
S
hA0
bqA
hCMD
bqA
hDATA
bqA
hDATA
bqA
P
Valid Read Sequences
The bq24721 only ACKs its READ address if it occurs following a repeated start where the high SCL clock time
is less then the tW(H), MAX of 50 µs. Upon receiving hA1, bq24721 outputs 2 bytes of data as indicated by the
preceding command. The command sequence is terminated by a STOP. The START and the STOP both act as
priority interrupts. If the host has been interrupted and is not sure where it left off, it sends a STOP and resets the
bq24721 state machine to the Idle state. Once in idle state, bq24721 ignores all activity on the SCL and SDA
lines until it receives a START. If a read sequence is terminated early by a START or STOP, the entire sequence
must be restarted by the host for valid data.
S
hA0
bqA
hCMD
bqA
R
hA1
bqA
bqDATA
hA
bqDATA
hN
P
SMBus Word Read
A valid Command is required to write to the bq24721, and a valid Command is required to specify the data to be
read. Once a read command is received the register data for the specified command is output to the host.
Nonvalid Sequences
START and non-hA0 or non-hA1 Address
A START followed by an address which is not bqA0 or bqA1 is NACKED.
Attempt to Specify Nonallowed Command
If the CMD points to a nonallowed command (reserved registers) , bq24721 sends a NACK back to the host.
Note that bq24721 NACKS whether a stop is sent or not.
S
hA0
S
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bqA
hCMD_N
bqN
hA_N
bqN
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Status and Control Registers
The SMBUS-Like communication engine allows easy access to system status data contained in internal registers
that contain information on current status for system power selection, charger mode and fault conditions. Any
change on the state of those bits generates an interrupt request to the host (ALARM pin = HI).
Table 2. Status Register Latched Bits
ACOC/CHGOC LATCHES
SET
RESET
CONTROL LOGIC ACTION AT DETECTION
Charger overcurrent detected
Host read register
Latches data going to status register, disables further updates on status
register CHGOC bit until reset.
AC adapter overcurrent detected
Host read register
Latches data going to status register, disables further updates on ACOC bit
until reset.
DESIGN CALCULATIONS EXAMPLE
System Requirements (See Figure 52)
For a system using a 20 V, ±5%, 80-W adapter, use a 12.6-V (3-cell) or 16.8-V (4-cell) battery, with a charge
current regulation threshold of 4 A, and a precharge/wake-up current of 300 mA. The battery pack voltage varies
from a deeply discharged voltage of 3-V per cell, and up to a regulation voltage of 4.2-V per cell— giving ranges
of 9-V to 12.6-V for 3-cell batteries, and 12-V to 16.8-V for 4-cell batteries.
Select the Sense Resistors
The SBS specification was done based on a 10-mΩ sense resistor. The charger operates with low resistance. If
the thermal dissipation is tolerable, a larger sense resistor (such as 20 mΩ) is used to provide greater accuracy
in the regulation and current sense amplifier.
For this design example, a 10-mΩ sense resistor is selected for both the input current and output charge current
sense resistors.
The power dissipation for each sense resistor is:
I (INPUT_REG) 2 + 10 mW
P (RSENSE_INPUT) + R(SENSE_DPM)
ǒ8019 WV Ǔ
2
+ 177 mW
(3)
where the maximum input current is the input power limit divided by the input voltage.
I (INPUT_REG) 2 + 10 mW
P (RSENSE_CHRG) + R(SENSE_CHRG)
2
(4A) + 160 mW
(4)
A 1 W, 2010 rating provides sufficient margin.
Select Switching Frequency:
Select switching frequency = 300 kHz.
Selecting the Output Inductor
Inductor is designed for current ripple 40% of the 4-A regulation threshold (I(RIPPLE) = 1.6 A).
ǒVI max * V(BAT) minǓ
L out +
40%
ǒ
V
(BAT)
V
I
min
max
Ǔ ǒ Ǔ
1
f
SW
I (CHRG_REG)
(5)
Where the VImax is the maximum input voltage, V(BAT)min is the minimum battery voltage, and fSW is the
switching frequency.
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ǒ219 VVǓ ǒ3001kHzǓ
(21 V * 9 V)
L out +
40%
4A
(6)
LOUT = 11 µH ≈ 10 µH, a standard 10-µH part is selected.
Selecting the Output Capacitor
The rule for selecting ceramic capacitors as the bulk output capacitor is to select 10 µF per 1 A of charge current.
In this case, a 4-A charge regulation current requires 40-µF output capacitance. Four 10-µF X5R 25-V ceramic
capacitors are recommended instead of 22 µF, because of a better trade-off with the voltage rating, size,
capacitance variation, and cost.
The output capacitors is divided evenly. Place the charge current sense resistor between the capacitors. This
provides filtering and a higher phase margin for accurate current sensing and regulation.
With this output capacitance, the steady state output ripple voltage is:
ǒ
I(RIPPLE)
VO(RIPPLE) +
V(BAT) min
VI max
Ǔ ǒ Ǔ
1
fSW
CO
)
ǒ
ESR
2
I (RIPPLE)
2
ǒ
) ESR ) R(SENSE)
2
) ESL
Ǔ
I (RIPPLE)
2
Ǔ
I(RIPPLE)
ǒ
V(BAT) min
VI max
Ǔ
ǒ Ǔ
1
f
SW
(7)
where ESR is the equivalent series resistance of the capacitors (10 mΩ each giving 5 mΩ for two capacitors
before the sense resistor and 5 mΩ for two capacitors after the sense resistor), and ESL is the equivalent series
inductance of the capacitors (0.5 nH).
1.6 A
VO(RIPPLE) +
ǒ219VVǓ ǒ3001kHzǓ
40 mF
) (5 mW
0.7 A ) 15 mW
ȡ
ȧ
Ȣ
0.7 A) ) 0.5 nH
ȣ
ǒ219 VVǓ ǒ3001kHzǓȧ
Ȥ
1.6 A
(8)
V(OUT_RIPPLE) = 57 mV + 14 mV + 0.56 mV = 72 mV ripple voltage (which equals 0.6% of 12.6 V) worst case
peak-to-peak steady state ripple when the battery is removed and the charger is on.
Selecting the Input Capacitor
The input capacitance is at a minimum the same as the output capacitance. Lower capacitance is used directly
at the converter input when the input is tied directly to the system load while charging. The capacitors are placed
as close as possible to the high-side FET drain (PVCC) and low-side FET source (PGND). A rule for selecting
ceramic capacitors as the bulk output capacitor is to select 10 µF per 1 A of charge current. In this case, a 4-A
charge regulation current requires 40-µF output capacitance. Four 10-µF X5R 25-V ceramic capacitors are
recommended instead of 22 µF, because of a better trade-off with the voltage rating, size, capacitance variation,
and cost. The input ripple voltage is usually larger because the input current ripple through the capacitor is the
full charge current.
I(CHRG)
VI(RIPPLE) +
ǒ
V(BAT) min
VI max
CI
Ǔ ǒ Ǔ
1
fSW
ǒ
) ESR
4
Ǔ
I (CHRG) ) ESL
I(CHRG)
ǒ
V(BAT) min
VI max
Ǔ ǒ Ǔ
1
fSW
(9)
where I(CHRG) is the charge regulation current.
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ǒ219 VVǓ ǒ3001kHzǓ
4A
VI(RIPPLE) +
40 mF
ǒ
) 10 mW
4
Ǔ
4 A ) 0.5 nH
4A
ǒ219 VVǓ ǒ3001kHzǓ
(10)
VI(RIPPLE) = 143 mV + 10 mV + 1.4 mV = 154 mV which is 0.77% of the nominal 20-V input voltage.
Note that a single 10-µF capacitor is used from PVCC (close to the drain of the high-side FET) to PGND, when
the system load is connected to PVCC, because the capacitors on the system rail provide the hold-up
capacitance needed.
Selecting the High-Side Power MOSFET, Q3
The high-side power MOSFET should be an NMOS power MOSFET with a standard voltage rating of 30 V to
support the 20-V input voltage. The current carrying capability should be at least 2x the maximum charge current.
Both the rDS(on) and the gate charge contribute to the power dissipation, while the worst case condition occurs at
max duty cycle (minimum input voltage and maximum battery voltage). The gate drive losses are the only
component that directly dissipate heat in the charger IC.
The FDS6680A was selected. The FDS6680A is a NMOS, 30-V, 12-mΩ device in an SO-8 package.
The conduction losses equal:
P (CON) + I (CHRG)
Ǹ
ȡ
ȧ
Ȣ
2
ȣ ǒr
ȧ DS(on)Ǔ
Ȥ
V (BAT) max
VI min
(11)
where the charger’s 6-V gate drive voltage helps reduce the rDS(on) for lower conduction losses.
The first order approximation switching losses equal:
P (SW) + I (CHRG)
VI min
ȡǒQ(GS) ) Q(GD)Ǔȣ
ȧ
ȧ
i(G)
Ȣ
Ȥ
f SW
(12)
and the gate drive losses are:
P(GD) = Q(GTOT) × VImim x fSW.
Where G(GS) is the gate charge from threshold current conducts until full charge current conducts; Q(GD) is the
miller charge where the drain voltage drops, and the Q(GTOT) is the total gate charge form off to fully-enhanced
on. The full input voltage is used for gate drive calculation because the internal gate drive regulator dissipates
the drop from input voltage to the 6-V output voltage which must be added to charging the gates to 6-V every
cycle.
P (CON) + (4A )
P (SW) + 4 A
P (GD) + 18 nC
2
ǒǸ
16.8 V
19 V
19 V
19 V
Ǔ
(12 mW) + 180 mW
ǒ(5 nC )IA 7 nC)Ǔ
300 kHz + 274 mW
300 kHz + 103 mW
(13)
Selecting the Low-Side Power MOSFET, Q4
The low-side power MOSFET should be an NMOS power MOSFET with a standard voltage rating of 30 V to
support the 20-V input voltage. The current carrying capability should be at least 2x the maximum charge current.
By nature of the synchronous rectifier operation, the drain-to-source voltage is always low at full charge current
when the low-side FET turns on—this makes switching losses insignificant, except for the reverse recovery and
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dead-time contributors that arise from letting the body-diode conduct. The conduction losses and gate drive
losses are the dominant power dissipation; while, the worst case condition occurs at minimum duty cycle
(maximum input voltage and minimum battery voltage). The gate drive losses are the only component that
directly dissipate heat in the charger IC. Note that the reverse recovery losses are included in this calculation, but
the dissipation occurs in the high-side power MOSFET.
The FDS6670A was selected. The FDS6670A is an NMOS, 30-V, 12-mΩ device in an SO-8 package.
The conduction losses equal:
P (CON) + I (CHRG)
2
Ǹ
ȡ
ȧ
Ȣ
ȣ ǒr
ȧ DS(on)Ǔ
Ȥ
V (BAT) max
VI min
(14)
where the charger’s 6-V gate drive voltage helps reduce the rDS(on) for lower conduction losses.
The first order approximation switching losses are dominated by reverse recovery losses and dead-time losses
given by:
P(SW) = VImax × Q(RR) × fSW + I(CGRG) × V(F) × 2 × t(dead-time) x fSW
and the gate drive losses are :
P(GD) = Q(GTOT) × VImim x fSW.
Where t(dead-time) is the dead-time where both FETs are off on either edge, Q(RR) is the reverse recovery charge,
Q(GTOT) is the total gate charge form off to fully-enhanced on. The full input voltage is used for gate drive
calculation because the internal gate drive regulator dissipates the drop from input voltage to the 6-V output
voltage which must be added to charging the gates to 6 V every cycle.
ǒǸ
1* 9 V
21 V
P (CON) + (4A )
2
P
21 nC
(SW)
+ 21 V
P (GD) + 26 nC
21 V
Ǔ
300 kHz ) 4 A
(9 mW) + 109 mW
0.8 V
2
30 ns
300 kHz + 132 mW ) 57.6 mW + 190 mW
300 kHz + 164 mW
(15)
Power MOSFET Thermal Limit Verification
The thermal limit verification allows for a 40°C temperature rise from 85°C ambient to 124°C silicon junction
temperatures. For SO-9, the RθJA is 50°C/W.
The low-side power FET expected temperature rise is:
DT (LOW*SIDE) + RthetaJA
o
P(LOSS_LOW_SIDE) + 50 C
W
(109 mW ) 57.6 mW ) 164 mW) + 16 oC
(16)
The high-side power FET expected temperature rise is:
DT(HIGH*SIDE) + R qJA
o
P (LOSS_HIGH_SIDE) + 50 C
W
(180 mW ) 274 mW ) 103 mW ) 134 mW) + 35 oC
(17)
Optional Schottky Diode across Low-Side FET
An optional Schottky diode can be used across the low-side power MOSFET (with cathode to drain and anode to
source) to help reduce both the V(F) losses and reverse recovery losses. The Schottky diode is selected on the
basis of the V(F) at I(CHRG), but a power rating for the full current is not required since the Schottky primarily
conducts during the dead-times, which is a fraction of the total switching period.
Power loss for the Schottky are:
P(SW) = I(CHRG) × VF × 2 × t(dead-time) x fSW = 4 A × 0.5 V × 2 × 30 ns × 300 kHz = 36 mW
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Selecting the System Power Selector Power MOSFETs, Q1, Q2, Q5
The system power selector power MOSFETs (Q1, Q2, Q5) must be PMOS power MOSFETs with a standard
voltage rating of 30 V to support the 20-V input voltage, and must have a low rDS(on) to minimize conduction
losses. The continuous current carrying capability should be at least 2x the maximum charge current. Both the
rDS(on) and the gate charge contribute to the power dissipation, while the worst case condition occurs at max duty
cycle (minimum input voltage and maximum battery voltage). The gate drive losses are the only component that
directly dissipate heat in the charger IC.
The SI4435 was selected for Q1, Q2, and Q5. The SI4435 is a PMOS, 30-V, 35-mΩ device in an SO-8 package.
The conduction losses equal:
P(CON) = I(RMS)2 × rDS(on),
I(RMS_Q1) = I(RMS_Q2) = I(RMS_Q5) = I(SYS)max
where I(RMS) is the RMS current expected for each MOSFET, and I(SYS)max is the maximum continuous system
current.
Note that for Q1, the charge current is not used because the charge current drops to zero as the system current
(I(SYS)max) equals or exceeds the programmed input current DAC threshold (IDPM). This is assuming the typical
case where I(SYS)max > I(DPM).
The system power selector’s -6-V gate drive voltage helps reduce the rDS(on) for lower conduction losses.
Calculate the Bootstrap Capacitor
The minimum bootstrap capacitor is calculated by the high-side turn-on charge requirements per cycle and the
gate drive voltage required. The total gate charge for the FDS6680A high-side power FET is Q(GTOT) = 18 nC,
and the maximum gate drive voltage drop allowed is V(DROP) = 0.5 V. The switching frequency of 300 kHz gives a
cycle period of 3.33 µs.
Q (GTOT)
C (BTST) min +
+ 18 nC + 36 nF
V(DROP) max
0.5 V
(18)
Select C(BTST) = C12 = 100 nF = 0.1 µF. Connect C12 between the PH and the BTST pins. Also connect a
bootstrap Schottky diode, D2 , from the REGN pin to the BTST pin. The current rating for the bootstrap diode,
D2, is determined by:
Q (GTOT)
I (DBTST_RATING) min +
+ 18 nC + 5.4 mA
3.33 ms
Ts
(19)
Use a 100-mA rated Schottky diode.
An optional 4.7-Ω bootstrap resistor is placed between the BTST pin and the node where C12 and D2 are
connected, in order to minimize ringing by lowering slew rate on the PH node voltage. For lower gate charge
FETs, a larger resistor value can be used up to 10 Ω or 15 Ω. Using larger resistor values increase the switching
losses, and lower the efficiency.
Calculate the ACDET Programming Resistors
The adapter is detected when the ACDET pin voltage (sensed adapter input voltage scaled down by resistor
divider) exceeds the 1.2 V. The adapter voltage threshold is set to a value less than the minimum adapter
voltage, and higher than the maximum battery pack voltage. [Important: don’t set the value less than the
maximum battery pack voltage, otherwise adapter removal is never detected when the battery is connected, and
the battery pack could potentially be drained when no adapter is present]. For this design example, VImin = 19 V,
V(BAT) = 16.8 V; therefore, select an adapter detect voltage of V(ADAPT_DET) = 18 V.
V
+ 1.2 V R3 ) R4
R4
Using the equation: (ADAPT_DET)
and setting V(ADAPT_DET) = 18 V, and selecting (R3 + R4) ≈ 500 Ω.
Calculate the standard 1% resistor values of R4 = 33.2 kΩ, and R4 = 464 kΩ.
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Calculate the ISYNSET Programming Resistor
The inductor current ripple is defined by:
ǒ
V
ǒVI max * V(BAT) minǓ
DI L +
(BAT)
V
I
min
Ǔ ǒ Ǔ
1
f
max
SW
L out
(20)
where the VImax is the maximum input voltage, V(BAT)min is the minimum battery voltage, and fSW is the
switching frequency.
9V
1
(21 V * 9 V)
21 V
300 kHz
DI L +
10 mH
(21)
ǒ
Ǔ ǒ
Ǔ
DI L
The worst case inductor current ripple is ΔIL = 1.71 A. To set the I(SYNSET) threshold to a value between
and ΔIL . Set I(SYNSET) = 1.5 A. R(SYNSET) is calculated by the following equation:
1 V 500 W
R12 + R (SYNSET) +
+ 1 V 500 W + 33 kW.
I (SYNSET) R(SENSE)
1.5 A 10 mW
2
:
(22)
Calculate the Thermistor Sense, TS, Programming Resistors
The TS comparator programming resistors need the cold pack temperature to disable charge defined, and the
hot pack temperature to disable charge defined. The battery cell manufacturer defines the limits. For this
example, the pack charging temperature limits are: cold temperature = T(COLD) = 0°C, and hot temperature
T(HOT) = 45°C. The charger is not allowed to charge at these temperatures and outside this range, to protect the
battery pack.
Using an industry standard Semitec 202AT NTC Thermistor, the thermistor cold temperature resistance is
R(TH_C) = 64.88 kΩ; and the thermistor hot temperature resistance of R(TH_H) = 8.716 kΩ. These values are found
in the 202AT thermistor manufacturer’s data sheet.
The comparator voltage cold limit threshold is V(LTF) = 73.5% of VREF5 = 0.735 × 5 V = 3.675 V, where
VREF5 = 5 V. Likewise, the comparator voltage hot limit threshold is V(HTF) = 34.4% of VREF5 = 0.344 × 5 V =
1.72 V.
Since the internal references are with respect to VREF5, R5 is connected from the VREF5 pin to the TS pin, and
connect R13 from the TS pin to the AGND pin. The R5 and R13 resistor values are calculated by solving the
following two equations:
R (VREF_TS) + R5 +
R(TS_AGND) = R13 =
R(TH_C)
R(TH_H)
V (LTF)
ǒV(LTF) * V(HTF)Ǔ
ǒR(TH_C) * R(TH_H)Ǔ
VREF5
V(HTF)
(23)
(
)
(R(TH_H) ´ V(LTF) ´ VREF5 )+ (R(TH_C) ´ V(LTF) ´ V(HTF) )- (R(TH_C) ´ V(HTF) ´ VREF5 )- (R(TH_H) ´ V(LTF) ´ V(HTF) )
R(TH_C) ´ R(TH_H) ´ VREF5 ´ V(HTF) - V(LTF)
(24)
giving standard 1% resistor values of R5 = 15.8 kΩ, and R13 = 130 kΩ.
Using these programming, check that when the battery pack is removed (i.e. the Thermistor is disconnected), the
TS voltage is pulled-above the battery detect threshold (V(TSDET) = 0.85 × 5 V = 4.25 V).
5 V 130 kW
V (NO_PACK) + VREF5 R13 +
+ 4.458 V.
(R5 ) R13)
(15.8 kW ) 130 kW)
(25)
V(NO_PACK) > V(TSDET) allowing for proper battery detect.
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PACKAGE OPTION ADDENDUM
www.ti.com
14-Oct-2022
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
Samples
(4/5)
(6)
BQ24721CRHBR
ACTIVE
VQFN
RHB
32
3000
RoHS & Green
NIPDAU
Level-2-260C-1 YEAR
-40 to 85
BQ
24721C
Samples
BQ24721CRHBT
ACTIVE
VQFN
RHB
32
250
RoHS & Green
NIPDAU
Level-2-260C-1 YEAR
-40 to 85
BQ
24721C
Samples
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of
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