INA301-Q1
SBOS786B – APRIL 2016 – REVISED APRIL 2022
INA301-Q1 36-V, Automotive, High-Speed, Zero-Drift, Voltage-Output
Current-Shunt Monitor With High-Speed, Overcurrent Protection Comparator
1 Features
3 Description
•
•
The INA301-Q1 includes both a high commonmode, current-sensing amplifier and a high-speed
comparator configured to provide overcurrent
protection by measuring the voltage developed
across a current-sensing or current-shunt resistor
and comparing that voltage to a defined threshold
limit. This device features an adjustable limit-threshold
range that is set using a single external limitsetting resistor. This current-shunt monitor measures
differential voltage signals on common-mode voltages
that can vary from 0 V up to 36 V, independent of the
supply voltage.
•
•
•
•
•
•
•
•
•
Qualified for automotive applications
AEC-Q100 qualified with the following results:
– Device temperature grade 1: –40°C to +125°C
ambient operating temperature
– Device HBM ESD classification level 2
– Device CDM ESD classification level C6
Functional Safety-Capable
– Documentation available to aid functional safety
system design
Wide common-mode input range: 0 V to 36 V
Dual output: amplifier and comparator output
High accuracy amplifier:
– Offset voltage: 35 µV (maximum)
– Offset voltage drift: 0.5 µV/°C (maximum)
– Gain error: 0.1% (maximum)
– Gain error drift: 10 ppm/°C
Available amplifier gains:
– INA301A1-Q1: 20 V/V
– INA301A2-Q1: 50 V/V
– INA301A3-Q1: 100 V/V
Programmable alert threshold set through a single
resistor
Total alert response time: 1 µs
Open-drain output with both transparent and
latching modes
Package: VSSOP-8
The open-drain alert output can be configured to
operate in either a transparent mode, where the
output status follows the input state, or in a latched
mode, where the alert output is cleared when the latch
is reset. The device alert response time is under 1 µs,
allowing for quick detection of overcurrent events.
This device operates from a single 2.7-V to 5.5V supply, drawing a maximum supply current of
700 µA. The device is specified over the extended
operating temperature range of –40°C to +125°C, and
is available in an 8-pin VSSOP package.
Device Information(1)
PART NUMBER
2 Applications
INA301-Q1
•
•
•
•
•
•
•
•
•
(1)
Solenoid Control
Low-Side Motor Monitoring
Electronic Power Steering
Power Seats
Power Windows
Body Control Modules
Electronic Control Units
Overcurrent Protection
eFuses
PACKAGE
BODY SIZE (NOM)
VSSOP (8)
3.00 mm × 3.00 mm
For all available packages, see the package option
addendum at the end of the data sheet.
CBYPASS
0.1 F
2.7 V to 5.5 V
Supply
(0 V to 36 V)
VS
IN+
+
INA301-Q1
RPULL-UP
10 k
Microcontroller
OUT
ADC
ALERT
IN±
Load
GPIO
RESET
GPIO
LIMIT
DAC
GND
RLIMIT
Copyright © 2016, Texas Instruments Incorporated
Typical Application
An IMPORTANT NOTICE at the end of this data sheet addresses availability, warranty, changes, use in safety-critical applications,
intellectual property matters and other important disclaimers. PRODUCTION DATA.
INA301-Q1
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SBOS786B – APRIL 2016 – REVISED APRIL 2022
Table of Contents
1 Features............................................................................1
2 Applications..................................................................... 1
3 Description.......................................................................1
4 Revision History.............................................................. 2
5 Pin Configuration and Functions...................................3
6 Specifications.................................................................. 4
6.1 Absolute Maximum Ratings........................................ 4
6.2 ESD Ratings............................................................... 4
6.3 Recommended Operating Conditions.........................4
6.4 Thermal Information....................................................4
6.5 Electrical Characteristics.............................................5
6.6 Typical Characteristics................................................ 7
7 Detailed Description......................................................13
7.1 Overview................................................................... 13
7.2 Functional Block Diagram......................................... 13
7.3 Feature Description...................................................14
7.4 Device Functional Modes..........................................16
8 Applications and Implementation................................ 18
8.1 Application Information............................................. 18
8.2 Typical Application.................................................... 22
9 Power Supply Recommendations................................24
10 Layout...........................................................................24
10.1 Layout Guidelines................................................... 24
10.2 Layout Example...................................................... 25
11 Device and Documentation Support..........................26
11.1 Documentation Support.......................................... 26
11.2 Receiving Notification of Documentation Updates.. 26
11.3 Support Resources................................................. 26
11.4 Trademarks............................................................. 26
11.5 Electrostatic Discharge Caution.............................. 26
11.6 Glossary.................................................................. 26
12 Mechanical, Packaging, and Orderable
Information.................................................................... 26
4 Revision History
NOTE: Page numbers for previous revisions may differ from page numbers in the current version.
Changes from Revision A (June 2016) to Revision B (April 2022)
Page
• Added Functional Safety information..................................................................................................................1
• Changed the Power Supply Recommendations section...................................................................................24
Changes from Revision * (April 2016) to Revision A (June 2016)
Page
• Changed from product preview to production data.............................................................................................1
2
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5 Pin Configuration and Functions
VS
1
8
IN+
OUT
2
7
IN±
LIMIT
3
6
ALERT
GND
4
5
RESET
Not to scale
Figure 5-1. DGK Package 8-Pin VSSOP Top View
Table 5-1. Pin Functions
PIN
NO.
NAME
I/O
1
VS
Analog
2
OUT
Analog output
3
LIMIT
Analog input
DESCRIPTION
Power supply, 2.7 V to 5.5 V
Output voltage
Alert threshold limit input; see the Section 7.3.2 section for details on setting the limit
threshold.
4
GND
Analog
5
RESET
Digital input
Ground
6
ALERT
Digital output
Overlimit alert, active-low, open-drain output
7
IN–
Analog input
Negative voltage input. Connect to load side of the shunt resistor.
8
IN+
Analog input
Positive voltage input. Connect to supply side of the shunt resistor.
Transparent or latch mode selection input
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6 Specifications
6.1 Absolute Maximum Ratings
over operating free-air temperature range (unless otherwise noted)(1)
MIN
Supply voltage, VS
)(2)
UNIT
6
V
–40
40
Common-mode(3)
GND – 0.3
40
Analog input
LIMIT pin
GND – 0.3
(VS) + 0.3
V
Analog output
OUT pin
GND – 0.3
(VS) + 0.3
V
Digital input
RESET pin
GND – 0.3
(VS) + 0.3
V
Digital output
ALERT pin
GND – 0.3
6
V
–55
150
°C
150
°C
150
°C
Analog inputs (IN+, IN–)
Differential (VIN+) – (VIN–
MAX
Operating temperature, TA
Junction temperature, TJ
Storage temperature, Tstg
(1)
(2)
(3)
–65
V
Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. These are stress
ratings only, which do not imply functional operation of the device at these or any other conditions beyond those indicated under
Recommended Operating Conditions. Exposure to absolute-maximum-rated conditions for extended periods may affect device
reliability.
VIN+ and VIN– are the voltages at the IN+ and IN– pins, respectively.
Input voltage can exceed the voltage shown without causing damage to the device if the current at that pin is limited to 5 mA.
6.2 ESD Ratings
VALUE
V(ESD)
(1)
(2)
Electrostatic discharge
Human-body model (HBM), per ANSI/ESDA/JEDEC JS-001(1)
±2000
Charged-device model (CDM), per JEDEC specification JESD22-C101(2)
±1000
UNIT
V
JEDEC document JEP155 states that 500-V HBM allows safe manufacturing with a standard ESD control process.
JEDEC document JEP157 states that 250-V CDM allows safe manufacturing with a standard ESD control process.
6.3 Recommended Operating Conditions
over operating free-air temperature range (unless otherwise noted)
MIN
VCM
Common-mode input voltage
VS
Operating supply voltage
2.7
TA
Operating free-air temperature
–40
NOM
MAX
12
UNIT
V
5
5.5
V
125
°C
6.4 Thermal Information
INA301-Q1
THERMAL
METRIC(1)
DGK (VSSOP)
UNIT
8 PINS
RθJA
Junction-to-ambient thermal resistance
161.5
°C/W
RθJC(top)
Junction-to-case (top) thermal resistance
62.3
°C/W
RθJB
Junction-to-board thermal resistance
81.4
°C/W
ψJT
Junction-to-top characterization parameter
6.8
°C/W
ψJB
Junction-to-board characterization parameter
80
°C/W
RθJC(bot)
Junction-to-case (bottom) thermal resistance
N/A
°C/W
(1)
4
For more information about traditional and new thermal metrics, see the Semiconductor and IC Package Thermal Metrics application
report.
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6.5 Electrical Characteristics
at TA = 25°C, VSENSE = VIN+ – VIN– = 10 mV, VS = 5 V, VIN+ = 12 V, and VLIMIT = 2 V (unless otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
INPUT
VCM
VIN
CMR
VOS
Common-mode input voltage range
Differential input voltage range
Common-mode rejection
Offset voltage, RTI(1)
0
36
VIN = VIN+ – VIN–, INA301A1-Q1
0
250
VIN = VIN+ – VIN–, INA301A2-Q1
0
100
VIN = VIN+ – VIN–, INA301A3-Q1
0
50
INA301A1-Q1, VIN+ = 0 V to 36 V,
TA = –40°C to +125°C
100
110
INA301A2-Q1, VIN+ = 0 V to 36 V,
TA = –40°C to +125°C
106
118
INA301A3-Q1, VIN+ = 0 V to 36 V,
TA = –40°C to +125°C
110
120
V
mV
dB
INA301A1-Q1
±25
±125
INA301A2-Q1
±15
±50
INA301A3-Q1
±10
±35
0.1
0.5
µV/°C
±0.1
±10
µV/V
µV
dVOS/dT
Offset voltage drift, RTI(1)
TA= –40°C to +125°C
PSRR
Power-supply rejection ratio
VS = 2.7 V to 5.5 V, VIN+ = 12 V,
TA = –40°C to +125°C
IB
Input bias current
IB+, IB–
120
µA
IOS
Input offset current
VSENSE = 0 mV
±0.1
µA
OUTPUT
INA301A1-Q1
G
Gain
Gain error
20
INA301A2-Q1
50
INA301A3-Q1
100
V/V
INA301A1-Q1, VOUT = 0.5 V to VS – 0.5 V
±0.03%
±0.1%
INA301A2-Q1, VOUT = 0.5 V to VS – 0.5 V
±0.05%
±0.15%
INA301A3-Q1, VOUT = 0.5 V to VS – 0.5 V
±0.11%
±0.2%
3
10
TA= –40°C to 125°C
Nonlinearity error
VOUT = 0.5 V to VS – 0.5 V
Maximum capacitive load
No sustained oscillation
ppm/°C
±0.01%
500
pF
VOLTAGE OUTPUT
Swing to VS power-supply rail
RL = 10 kΩ to GND,
TA = –40°C to +125°C
VS – 0.05
VS – 0.1
Swing to GND
RL = 10 kΩ to GND,
TA = –40°C to +125°C
VGND + 20
VGND + 30
V
mV
FREQUENCY RESPONSE
BW
SR
Bandwidth
INA301A1-Q1
550
INA301A2-Q1
500
INA301A3-Q1
450
Slew rate
kHz
4
V/µs
30
nV/√ Hz
NOISE, RTI(1)
Voltage noise density
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at TA = 25°C, VSENSE = VIN+ – VIN– = 10 mV, VS = 5 V, VIN+ = 12 V, and VLIMIT = 2 V (unless otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
COMPARATOR
tp
ILIMIT
Total alert propagation delay
Input overdrive = 1 mV
Slew-rate-limited tp
VOUT step = 0.5 V to 4.5 V, VLIMIT = 4 V
Limit threshold output current
TA = 25°C
79.7
TA = –40°C to +125°C
79.2
INA301A1-Q1
VOS
VHYS
Comparator offset voltage
Hysteresis
0.75
1
1
1.5
80
80.3
80.8
1
1
4
INA301A3-Q1
1.5
4.5
INA301A1-Q1
20
50
INA301A3-Q1
100
µA
3.5
INA301A2-Q1
INA301A2-Q1
µs
mV
mV
VIH
High-level input voltage
1.4
6
VIL
Low-level input voltage
0
0.4
V
V
VOL
Alert low-level output voltage
IOL = 3 mA
70
300
mV
ALERT pin leakage input current
VOH = 3.3 V
0.1
1
µA
Digital leakage input current
0 ≤ VIN ≤ VS
1
µA
POWER SUPPLY
IQ
Quiescent current
(1)
RTI = referred-to-input.
6
VSENSE = 0 mV, TA = 25°C
TA = –40°C to +125°C
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500
650
700
µA
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6.6 Typical Characteristics
Input Offset Voltage (PV)
100
80
60
40
20
0
-20
-40
-80
-100
-60
Population
100
80
60
40
20
0
-20
-40
-60
-80
-100
Population
at TA = 25°C, VS = 5 V, VIN+ = 12 V, and alert pullup resistor = 10 kΩ (unless otherwise noted)
Input Offset Voltage (PV)
Figure 6-1. Input Offset Voltage Distribution (INA301A1-Q1)
Figure 6-2. Input Offset Voltage Distribution (INA301A2-Q1)
60
INA301A1-Q1
INA301A2-Q1
INA301A3-Q1
Population
Offset Voltage (µV)
40
20
100
80
60
40
20
0
-20
-40
-60
-80
-100
0
-20
-50
Input Offset Voltage (PV)
0
25
50
75
Temperature (°C)
100
125
150
Figure 6-4. Input Offset Voltage vs. Temperature
Figure 6-3. Input Offset Voltage Distribution (INA301A3-Q1)
CMRR (PV/V)
Figure 6-5. Common-Mode Rejection Ratio Distribution
(INA301A1-Q1)
5
4
3
2
1
0
-1
-2
-3
-4
-5
10
8
6
4
2
0
-2
-4
-6
-8
-10
Population
Population
-25
CMRR (PV/V)
Figure 6-6. Common-Mode Rejection Ratio Distribution
(INA301A2-Q1)
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6.6 Typical Characteristics (continued)
at TA = 25°C, VS = 5 V, VIN+ = 12 V, and alert pullup resistor = 10 kΩ (unless otherwise noted)
2.5
2
INA301A1-Q1
INA301A2-Q1
INA301A3-Q1
1.5
1
0.5
0
-0.5
-1
-50
3
2.5
2
1
1.5
0.5
0
-0.5
-1
-2
-1.5
-2.5
-3
Population
Common-Mode Rejection Ratio (µV/V)
3
CMRR (PV/V)
Figure 6-7. Common-Mode Rejection Ratio Distribution
(INA301A3-Q1)
-25
0
25
50
75
Temperature (°C)
100
125
150
Figure 6-8. Common-Mode Rejection Ratio vs. Temperature
INA301A1-Q1
INA301A2-Q1
INA301A3-Q1
120
Population
100
.
0.1
0.08
0.06
0.04
Gain Error (%)
Figure 6-10. Gain Error Distribution (INA301A1-Q1)
Gain Error (%)
0.2
0.16
0.12
0.08
0.04
0
-0.04
-0.08
-0.12
-0.16
-0.2
Population
0.1
0.08
0.06
0.04
0.02
0
-0.02
-0.04
-0.06
-0.08
-0.1
Population
Figure 6-9. Common-Mode Rejection Ratio vs. Frequency
Gain Error (%)
Figure 6-11. Gain Error Distribution (INA301A2-Q1)
8
0.02
1M
0
100k
-0.02
1k
10k
Frequency (Hz)
-0.04
100
-0.06
60
10
-0.08
80
-0.1
Common-Mode Rejection Ratio (dB)
140
Figure 6-12. Gain Error Distribution (INA301A3-Q1)
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6.6 Typical Characteristics (continued)
at TA = 25°C, VS = 5 V, VIN+ = 12 V, and alert pullup resistor = 10 kΩ (unless otherwise noted)
0.5
50
INA301A1-Q1
INA301A2-Q1
INA301A3-Q1
0.4
0.3
40
30
0.1
Gain (dB)
Gain Error (%)
0.2
0
-0.1
-0.2
20
10
0
-0.3
INA301A1-Q1
INA301A2-Q1
INA301A3-Q1
-10
-0.4
-0.5
-50
-20
-25
0
25
50
75
Temperature (°C)
100
125
150
1
10
1k
10k
Frequency (Hz)
100k
1M
10M
Figure 6-14. Gain vs. Frequency
Figure 6-13. Gain Error vs. Temperature
140
VS
Output Voltage Swing (V)
120
100
PSRR (dB)
100
80
60
VS - 1
VS - 2
GND + 3
GND + 2
125ºC
25ºC
-40ºC
GND + 1
40
GND
0
2
4
20
1
10
100
1k
10k
Frequency (Hz)
100k
1M
10M
8
10
12
14
.
Figure 6-15. Power-Supply Rejection Ratio vs. Frequency
Figure 6-16. Output Voltage Swing vs. Output Current
150
250
200
120
Input Bias Current (PA)
Input Bias Current (PA)
6
Output Current (mA)
150
100
50
90
60
30
0
0
-50
0
5
10
15
20
25
30
Common-Mode Voltage (V)
35
40
0
VS = 5 V
5
10
15
20
25
30
Common-Mode Voltage (V)
35
40
VS = 0 V
Figure 6-17. Input Bias Current vs. Common-Mode Voltage
Figure 6-18. Input Bias Current vs. Common-Mode Voltage
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6.6 Typical Characteristics (continued)
at TA = 25°C, VS = 5 V, VIN+ = 12 V, and alert pullup resistor = 10 kΩ (unless otherwise noted)
145
600
140
Quiescent Current (PA)
Input Bias Current (PA)
550
135
130
125
120
115
110
500
450
400
350
105
100
-50
-25
0
25
50
75
Temperature (qC)
100
125
300
2.7
150
540
3.6
3.9 4.2 4.5 4.8
Supply Voltage (V)
5.1
5.4
5.7
Input-Referred Voltage Noise (nV/Ö Hz)
35
520
Quiescent Current (PA)
3.3
Figure 6-20. Quiescent Current vs. Supply Voltage
Figure 6-19. Input Bias Current vs. Temperature
500
480
460
440
420
-50
3
30
25
20
15
10
INA301A1-Q1
INA301A2-Q1
INA301A3-Q1
5
0
-25
0
25
50
75
Temperature (qC)
100
125
150
10
100
1k
10k
Frequency (Hz)
1M
Input
Output
Output (1 V/div)
Referred-to-Input Voltage Noise
(200 nV/div)
100k
Figure 6-22. Input-Referred Voltage Noise vs. Frequency
Input (200 mV/div)
Figure 6-21. Quiescent Current vs. Temperature
1
Time (1 s/div)
Time (1 Ps/div)
.
Figure 6-23. 0.1-Hz to 10-Hz Referred-to-Input Voltage Noise
10
4-VPP output step
Figure 6-24. Voltage Output Rising Step Response
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6.6 Typical Characteristics (continued)
VCM
VOUT
Time (1 Ps/div)
VOUT (60 mV/div)
Output (1 V/div)
Input
Output
Common-Mode Voltage (10 V/div)
Input (200 mV/div)
at TA = 25°C, VS = 5 V, VIN+ = 12 V, and alert pullup resistor = 10 kΩ (unless otherwise noted)
Time (2 Ps/div)
4-VPP output step
.
Figure 6-25. Voltage Output Falling Step Response
Figure 6-26. Common-Mode Voltage Transient Response
80.8
Voltage (2 V/div)
Limit Current Source (PA)
80.6
VSUPPLY
VOUT
80.4
80.2
80
79.8
79.6
79.4
79.2
-50
Time (5 Ps/div)
.
0
25
50
75
Temperature (qC)
100
125
150
Figure 6-28. Limit Current Source vs. Temperature
VIN * 20 V/V
Alert
VLIMIT
Voltage (0.5 V/div)
Figure 6-27. Start-Up Response
Voltage (0.5 V/div)
-25
Time (200 ns/div)
VIN * 50 V/V
Alert
VLIMIT
Time (200 ns/div)
Figure 6-29. Total Propagation Delay (INA301A1-Q1)
Figure 6-30. Total Propagation Delay (INA301A2-Q1)
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6.6 Typical Characteristics (continued)
at TA = 25°C, VS = 5 V, VIN+ = 12 V, and alert pullup resistor = 10 kΩ (unless otherwise noted)
1,000
Propagation Delay (ns)
Voltage (0.5 V/div)
800
VIN * 100 V/V
Alert
VLIMIT
600
400
200
0
-50
Time (200 ns/div)
-25
0
.
100
125
150
VOD = 1 mV
Figure 6-31. Total Propagation Delay (INA301A3-Q1)
Figure 6-32. Comparator Propagation Delay vs. Temperature
120
120
100
100
80
80
Hysteresis (mV)
Low-Level Output Voltage (mV)
25
50
75
Temperature (qC)
60
40
20
INA301A1-Q1
INA301A2-Q1
INA301A3-Q1
60
40
20
0
0
0.5
1
1.5
2
2.5
3
3.5
4
Low-Level Output Current (mA)
4.5
5
Figure 6-33. Comparator Alert VOL vs. IOL
0
-50
-25
0
25
50
75
Temperature (°C)
100
125
150
Figure 6-34. Hysteresis vs. Temperature
Voltage (2 V/div)
Reset
Alert
Time (2 Ps/div)
Figure 6-35. Comparator Reset Response
12
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7 Detailed Description
7.1 Overview
The INA301-Q1 is a 36-V common-mode, zero-drift topology, current-sensing amplifier that can be used in
both low-side and high-side configurations. These specially-designed, current-sensing amplifiers are able to
accurately measure voltages developed across current-sensing resistors (also known as current-shunt resistors)
on common-mode voltages that far exceed the supply voltage powering the device. Current can be measured
on input voltage rails as high as 36 V, and the device can be powered from supply voltages as low as 2.7 V.
The device can also withstand the full 36-V common-mode voltage at the input pins when the supply voltage is
removed without causing damage.
The zero-drift topology enables high-precision measurements with maximum input offset voltages as low as
35 μV with a temperature contribution of only 0.5 μV/°C over the full temperature range of –40°C to +125°C. The
low total offset voltage of the INA301-Q1 enables smaller current-sense resistor values to be used, and allows
for a more efficient system operation without sacrificing measurement accuracy resulting from the smaller input
signal.
The INA301-Q1 uses a single external resistor to allow for a simple method of setting the corresponding current
threshold level for the device to use for out-of-range comparison. Combining the precision measurement of the
current-sense amplifier and the onboard comparator enables an all-in-one overcurrent detection device. This
combination creates a highly-accurate solution that is capable of fast detection of out-of-range conditions, and
allows the system to take corrective actions to prevent potential component or system-wide damage.
7.2 Functional Block Diagram
CBYPASS
0.1 F
2.7 V to 5.5 V
Power Supply
(0 V to 36 V)
VS
IN+
INA301-Q1
RPULL-UP
10 k
+
OUT
Gain = 20, 50,
100
IN±
Load
ALERT
+
RESET
GND
LIMIT
RSET
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7.3 Feature Description
7.3.1 Alert Output ( ALERT Pin)
The device ALERT pin is an active-low, open-drain output that is designed to be pulled low when the input
conditions are detected to be out-of-range. Add a 10-kΩ pullup resistor from ALERT pin to the supply voltage.
This open-drain pin can be pulled up to a voltage beyond the VS supply voltage, but must not exceed 5.5 V.
Figure 7-1 shows the alert output response of the internal comparator. When the output voltage of the amplifier
is less than the voltage developed at the LIMIT pin, the comparator output is in the default high state. When
the amplifier output voltage exceeds the threshold voltage set at the LIMIT pin, the comparator output becomes
active and pulls low. This active low output indicates that the measured signal at the amplifier input has
exceeded the programmed threshold level, indicating an overcurrent or out-of-range condition has occurred.
6
VOUT
VLIMIT
ALERT
5
Voltage (V)
4
3
2
1
0
±1
Time (5 ms/div)
C001
Figure 7-1. Overcurrent Alert Response
7.3.2 Current-Limit Threshold
The INA301-Q1 determines if an overcurrent event is present by comparing the amplified measured voltage
developed across the current-sensing resistor to the corresponding signal developed at the LIMIT pin. The
threshold voltage for the LIMIT pin is set using a single external resistor, or by connecting an external voltage
source to the LIMIT pin.
7.3.2.1 Resistor-Controlled Current Limit
The typical method for setting the limit threshold voltage is to connect a resistor from the LIMIT pin to ground.
The value of this resistor, RLIMIT, is chosen in order to create a corresponding voltage at the LIMIT pin equivalent
to the output voltage, VOUT, when the maximum desired load current is flowing through the current-sensing
resistor. An internal 80-µA current source is connected to the LIMIT pin to create a corresponding voltage used
to compare to the amplifier output voltage, depending on the value of the RLIMIT resistor.
In the equations from Table 7-1, VTRIP represents the overcurrent threshold that the device is programmed to
monitor, and VLIMIT is the programmed signal set to detect the VTRIP level.
Table 7-1. Calculating the Threshold-Limit-Setting Resistor, RLIMIT
PARAMETER
VTRIP
VOUT at the desired-current trip value
VLIMIT
Threshold limit voltage
RLIMIT
Threshold limit-setting resistor value
EQUATION
ILOAD × RSENSE x Gain
VLIMIT = VTRIP
ILIMIT × RLIMIT
VLIMIT / ILIMIT
VLIMIT / 80 µA
7.3.2.1.1 Resistor-Controlled, Current-Limit Example
If the current level indicating an out-of-range condition is present is 20 A, and the current-sense resistor value is
10 mΩ, then the input threshold signal is 200 mV. The INA301A1-Q1 has a gain of 20, therefore, the resulting
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output voltage at the 20-A input condition is 4 V. The value for RLIMIT is selected to allow the device to detect
to this 20-A threshold, indicating an overcurrent event occurred. When the INA301-Q1 detects this out-of-range
condition, the ALERT pin asserts and pulls low. For this example, Table 7-2 lists the calculated value of RLIMIT
required to detect a 4-V level as 50 kΩ.
Table 7-2. Example of Calculating the Limit Threshold Setting Resistor, RLIMIT
PARAMETER
VTRIP
VOUT at the desired current trip value
VLIMIT
Threshold limit voltage
RLIMIT
Threshold limit-setting resistor value
EQUATION
ILOAD × RSENSE x Gain
↓
20 A x 10 mΩ x 20 V/V = 4 V
VLIMIT = VTRIP
ILIMIT × RLIMIT
VLIMIT / ILIMIT
↓
4 V / 80 µA = 50 kΩ
7.3.2.2 Voltage-Source-Controlled Current Limit
Another method for setting the limit voltage is to connect the LIMIT pin to a programmable digital-to-analog
converter (DAC) or other external voltage source. The benefit of this method is the ability to adjust the currentlimit threshold to account for different threshold voltages that are used for different system operating conditions.
For example, this method can be used in a system that has one current-limit threshold level that must be
monitored during a power-up sequence, but different threshold levels that must be monitored during other
system operating modes.
In Table 7-3, V TRIP represents the overcurrent threshold that the device is programmed to monitor, and VSOURCE
is the programmed signal set to detect the VTRIP level.
Table 7-3. Calculating the Limit Threshold Voltage Source, VSOURCE
PARAMETER
EQUATION
VTRIP
VOUT at the desired current trip value
VSOURCE
Threshold limit voltage
ILOAD × RSENSE × Gain
VSOURCE = VTRIP
7.3.3 Hysteresis
The onboard comparator in the INA301-Q1 reduces the possibility of oscillations in the alert output when the
measured signal level is near the overlimit threshold level because of noise. When the output voltage (VOUT)
exceeds the voltage developed at the LIMIT pin, the ALERT pin is asserted and pulls low. The output voltage
must drop below the LIMIT pin threshold voltage by the gain-dependent hysteresis level for the ALERT pin to
deassert and return to the nominal high state (see Figure 7-2).
ALERT
Alert
Output
VOUT
VLIMIT - Hysteresis
VLIMIT
Figure 7-2. Typical Comparator Hysteresis
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7.4 Device Functional Modes
7.4.1 Alert Mode
The device has two output operating modes, transparent and latched, that are selected based on the RESET pin
setting. These modes change how the ALERT pin responds following an alert when the overcurrent condition is
removed.
7.4.1.1 Transparent Output Mode
The device is set to transparent mode when the RESET pin is pulled low, thus allowing the output alert state
to change and follow the input signal with respect to the programmed alert threshold. For example, when the
differential input signal rises above the alert threshold, the ALERT output pin is pulled low. As soon as the
differential input signal drops below the alert threshold, the output returns to the default high-output state. A
common implementation using the device in transparent mode is to connect the ALERT pin to a hardware
interrupt input on a microcontroller. As soon as an overcurrent condition is detected and the ALERT pin is pulled
low, the hardware interrupt input detects the output-state change, and the microcontroller can begin to make
changes to the system operation required to address the overcurrent condition. Under this configuration, the
ALERT pin transition from high to low is captured by the microcontroller so that the output can return to the
default high state when the overcurrent event is removed.
7.4.1.2 Latch Output Mode
Some applications do not have the functionality available to continuously monitor the state of the output ALERT
pin to detect an overcurrent condition as described in the Transparent Output Mode section. A typical example
of this application is a system that is only able to poll the ALERT pin state periodically to determine if the system
is functioning correctly. If the device is set to transparent mode in this type of application, the state change of
the ALERT pin might be missed when ALERT is pulled low to indicate an out-of-range event, if the out-of-range
condition does not appear during one of these periodic polling events. Latch mode is specifically intended to
accommodate these applications.
The INA301-Q1 is placed into the corresponding output modes based on the signal connected to RESET (see
Table 7-4). The difference between latch mode and transparent mode is how the ALERT pin responds when an
overcurrent event ends. In transparent mode (RESET = low), when the differential input signal drops below the
limit threshold level after the ALERT pin asserts because of an overcurrent event, the ALERT pin state returns to
the default high setting to indicate that the overcurrent event has ended.
Table 7-4. Output Mode Settings
OUTPUT MODE
RESET PIN SETTING
Transparent mode
RESET = low
Latch mode
RESET = high
In latch mode (RESET = high), when an overlimit condition is detected and the ALERT pin is pulled low, the
ALERT pin does not return to the default high state when the differential input signal drops below the alert
threshold level. In order to clear the alert, pull the RESET pin low for at least 100 ns. Pulling the RESET pin low
allows the ALERT pin to return to the default high level, provided that the differential input signal has dropped
below the alert threshold. If the input signal is still greater than the threshold limit when the RESET pin is pulled
low, the ALERT pin remains low. When the alert condition is detected by the system controller, the RESET pin
can be set back to high in order to place the device back in latch mode.
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The latch and transparent modes represented in Figure 7-3 show that when VIN drops back below the VLIMIT
threshold for the first time, the RESET pin is pulled high. With the RESET pin is pulled high, the device is set to
latch mode, so that the ALERT pin output state does not return high when the input signal drops below the VLIMIT
threshold. Only when the RESET pin is pulled low does the ALERT pin return to the default high level, thus
indicating that the input signal is below the limit threshold. When the input signal drops below the limit threshold
for the second time, the RESET pin is already pulled low. The device is set to transparent mode at this point and
the ALERT pin is pulled back high as soon as the input signal drops below the alert threshold.
VLIMIT
VIN
(VIN+ - VIN-)
0V
Latch Mode
RESET
Transparent Mode
Alert Clears
ALERT
Alert Does Not Clear
Figure 7-3. Transparent Mode vs. Latch Mode
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8 Applications and Implementation
Note
Information in the following applications sections is not part of the TI component specification,
and TI does not warrant its accuracy or completeness. TI’s customers are responsible for
determining suitability of components for their purposes, as well as validating and testing their design
implementation to confirm system functionality.
8.1 Application Information
The INA301-Q1 enables easy configuration to detect overcurrent conditions in an application. This device is
individually targeted towards unidirectional overcurrent detection of a single threshold. However, this device can
also be paired with additional INA301-Q1 devices and circuitry to create more complex monitoring functional
blocks.
8.1.1 Selecting a Current-Sensing Resistor
The INA301-Q1 measures the differential voltage developed across a resistor when current flows through
the component in order to determine if the current being monitored exceeds a defined limit. This resistor is
commonly referred to as a current-sensing resistor or a current-shunt resistor, with each term commonly used
interchangeably. The flexible design of this device allows for measuring a wide differential input signal range
across the current-sensing resistor.
Selecting the value of this current-sensing resistor is primarily based on two factors: the required accuracy of the
current measurement, and the allowable power dissipation across the current-sensing resistor. Larger voltages
developed across this resistor allow for more accurate measurements to be made. Amplifiers have fixed internal
errors that are largely dominated by the inherent input offset voltage. When the input signal decreases, these
fixed internal amplifier errors become a larger portion of the measurement and increase the uncertainty in the
measurement accuracy. When the input signal increases, the measurement uncertainty is reduced because
the fixed errors are a smaller percentage of the signal being measured. Therefore, the use of larger-value,
current-sensing resistors inherently improves measurement accuracy.
However, a system design trade-off must be evaluated through the use of larger input signals that improve
measurement accuracy. Increasing the current sense resistor value results in an increase in power dissipation
across the current-sensing resistor, and also increases the differential voltage developed across the resistor
when current passes through the component. This increase in voltage across the resistor increases the power
that the resistor must be able to dissipate. Decreasing the value of the current-shunt resistor reduces the power
dissipation requirements of the resistor, but increases the measurement errors resulting from the decreased
input signal. Selecting the optimal value for the shunt resistor requires factoring both the accuracy requirement
for the specific application, and the allowable power dissipation of this component.
Low-ohmic-value resistors enable large currents to be accurately monitored with the INA301-Q1. An increasing
number of very low-ohmic-value resistors are becoming more widely available, with values of 200 μΩ and less,
and power dissipations of up to 5 W.
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8.1.1.1 Selecting a Current-Sensing Resistor Example
In this example, the trade-offs involved in selecting a current-sensing resistor are described. This example
requires 2.5% accuracy for detecting a 10-A overcurrent event, with only 250 mW of allowable power dissipation
across the current-sensing resistor at the full-scale current level. Although the maximum power dissipation
is defined as 250 mW, a lower dissipation is preferred in order to improve system efficiency. Some initial
assumptions are made that are used in this example:
•
•
the limit-setting resistor (RLIMIT) is a 1% component
the maximum tolerance specification for the internal threshold setting current source (0.5%) is used
Given the total error budget of 2.5%, up to 1% of error is available to be attributed to the measurement error of
the device under these conditions.
As shown in Table 8-1, the maximum value calculated for the current-sensing resistor with these requirements
is 2.5 mΩ. Although this value satisfies the maximum power dissipation requirement of 250 mW, headroom
is available from the 2.5% maximum total overcurrent detection error in order to reduce the value of the
current-sensing resistor, and reduce the power dissipation further. Selecting a 1.5-mΩ, current-sensing resistor
value offers a good tradeoff for reducing the power dissipation in this scenario by approximately 40% while still
remaining within the accuracy region.
Table 8-1. Calculating the Current-Sensing Resistor, RSENSE
PARAMETER
EQUATION
VALUE
UNIT
IMAX
Maximum current
10
A
PD_MAX
Maximum allowable power dissipation
250
mW
RSENSE_MAX
Maximum allowable RSENSE
2.5
mΩ
VOS
Offset voltage
150
µV
VOS_ERROR
Initial offset voltage error
EG
Gain error
ERRORTOTAL
Total measurement error
PD_MAX / IMAX 2
(VOS / (RSENSE_MAX × IMAX ) × 100
0.6%
0.25%
√(VOS_ERROR 2 + EG 2)
Allowable current threshold accuracy
0.65%
2.5%
ERRORINITIAL
Initial threshold error
ILIMIT Tolerance + RLIMIT Tolerance
ERRORAVAILABLE
Maximum allowable measurement error
Maximum Error – ERRORINITIAL
1.5%
VOS_ERROR_MAX
Maximum allowable offset error
√(ERRORAVAILABLE 2 – EG 2)
VDIFF_MIN
Minimum differential voltage
VOS / VOS_ERROR_MAX (1%)
15
mV
RSENSE_MIN
Minimum sense resistor value
VDIFF_MIN / IMAX
1.5
mΩ
PD_MIN
Minimum power dissipation
RSENSE_MIN × IMAX 2
150
mW
1%
0.97%
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8.1.2 Input Filtering
External system noise can significantly affect the ability of a comparator to accurately measure and detect
whether input signals exceed the reference threshold levels and reliably indicate overrange conditions. The most
obvious effect that external noise has on the operation of a comparator is to cause a false-alert condition. If a
comparator detects a large noise transient coupled into the signal, the device can easily interpret this transient
as an overrange condition.
External filtering helps reduce the amount of noise that reaches the comparator, and thus reduce the likelihood
of a false alert from occurring. The tradeoff to adding this noise filter is that the alert response time is increased
because of the input signal being filtered along with the noise. Figure 8-1 shows the implementation of an input
filter for the device.
2.7 V to 5.5 V
CBYPASS
0.1 F
Supply
(0 V to 36 V)
VS
IN+
RFILTER
” 10
CFILTER
INA301-Q1
RPULL-UP
10 k
+
OUT
ALERT
IN±
Load
RESET
LIMIT
GND
RLIMIT
Copyright © 2016, Texas Instruments Incorporated
Figure 8-1. Input Filter
Limiting the input resistance this filter is important because this resistance can have a significant affect on the
input signal that reaches the device input pins because of the device input bias currents. A typical system
implementation involves placing the current-sensing resistor very near the device so that the traces are very
short and the trace impedance is very small. This layout helps reduce the ability of coupling additional noise into
the measurement. Under these conditions, the characteristics of the input bias currents have minimal affect on
device performance.
As illustrated in Figure 8-2, the input bias currents increase in opposite directions when the differential input
voltage increases. This increase results from a device design that allows common-mode input voltages to far
exceed the device supply voltage range. With input filter resistors now placed in series with these unequal input
bias currents, there are unequal voltage drops developed across these input resistors. The difference between
these two voltage drops appears as an added signal that, in this case, subtracts from the voltage developed
across the current-sensing resistor, thus reducing the signal that reaches the device input pins. Smaller-value
input resistors reduce this effect of signal attenuation to allow for a more accurate measurement.
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225
Input Bias Current (µA)
200
175
150
125
100
75
50
25
0
0
50
100
150
200
250
Differential Input Voltage (mV)
C002
Figure 8-2. Input Bias Current vs. Differential Input Voltage
For example, with a differential voltage of 10 mV developed across a current-sensing resistor and using 20-Ω
resistors, the differential signal that actually reaches the device is 9.85 mV. A measurement error of 1.5% is
created as a result of these external input filter resistors. Use 10-Ω input filter resistors instead of the 20-Ω
resistors to reduce this added error from 1.5% down to 0.75%.
8.1.3 INA301-Q1 Operation With Common-Mode Voltage Transients Greater Than 36 V
With a small amount of additional circuitry, the INA301-Q1 can be used in circuits subject to transients greater
than 36 V. Use only Zener diodes or Zener-type transient absorbers (sometimes referred to as transzorbs). Any
other type of transient absorber has an unacceptable time delay. Start by adding a pair of resistors as a working
impedance for the Zener diode, as shown in Figure 8-3. Keep these resistors as small as possible; preferably,
10 Ω or less. Larger values can be used, but with an additional induced error resulting from less signal reaching
the device input pins. Because this circuit limits only short-term transients, many applications are satisfied with a
10-Ω resistor along with conventional Zener diodes of the lowest power rating available. This combination uses
the least amount of board space. These diodes can be found in packages as small as SOT-523 or SOD-523.
2.7 V to 5.5 V
CBYPASS
0.1 F
Supply
(0 V to 36 V)
VS
IN+
INA301-Q1
RPULL-UP
10k
+
OUT
RPROTECT
” 10
ALERT
IN±
Load
RESET
LIMIT
GND
RLIMIT
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Figure 8-3. Transient Protection
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8.2 Typical Application
Although this device is only able to measure current through a current-sensing resistor flowing in one direction, a
second INA301-Q1 can be used to create a bidirectional monitor (see Figure 8-4).
CBYPASS
0.1 F
2.7 V to 5.5 V
RPULL-UP
10 k
VS
IN+
+
OUT
IN±
Power Supply
(0 V to 36 V)
OCP+
ALERT
LIMIT
GND
RLIMIT
Current
CBYPASS
0.1 F
Output
2.7 V to 5.5 V
Load
RPULL-UP
10 k
VS
IN+
+
OUT
IN±
OCP±
ALERT
LIMIT
GND
RLIMIT
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Figure 8-4. Bidirectional Application
8.2.1 Design Requirements
For this design example, use the parameters listed in Table 8-2 as the input parameters.
Table 8-2. Design Parameters
DESIGN PARAMETERS
22
EXAMPLE VALUE
Supply voltage
3.3 V
Common-mode voltage
12 V
Voltage gain
100 V/V
Sense resistance
5 mΩ
Source-current swing
–2 A to +2 A
Voltage trip points
–1 A and +1 A
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8.2.2 Detailed Design Procedure
First, reverse the input pins of the second INA301-Q1 across the current-sensing resistor. The second device is
now able to detect current flowing in the other direction relative to the first device.
Then, select limit resistors to set the voltage trip points by using the equations in Table 7-1. For this application
example, these equations give a value of 6.25 kΩ for both limit resistors.
Connect the outputs of each device to an AND gate in order to detect if either of the limit threshold levels are
exceeded. Table 8-3shows that the output of the AND gate is high if neither overcurrent limit thresholds are
exceeded. A low output state of the AND gate indicates that either the positive overcurrent limit or the negative
overcurrent limit are surpassed.
Table 8-3. Bidirectional Overcurrent Output Status
OCP STATUS
OUTPUT
OCP+
0
OCP–
0
No OCP
1
8.2.3 Application Curve
Input
(5 mV/div)
Alert Output
(1 V/div)
Figure 8-5 shows two INA301-Q1 devices being used in a bidirectional configuration and an output control circuit
to detect if one of the two alerts is exceeded.
Positive Limit
0V
Negtive Limit
Time (5 ms/div)
Figure 8-5. Bidirectional Application Curve
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9 Power Supply Recommendations
The device input circuitry accurately measures signals on common-mode voltages beyond the power-supply
voltage, VS. For example, the voltage applied to the VS power-supply pin can be 5 V, whereas the load
power-supply voltage being monitored (VCM) can be as high as 36 V. At power up, for applications where the
common-mode voltage (VCM) slew rate is greater than 6 V/μs with a final common-mode voltage greater than 20
V, TI recommends that the VS supply be present before VCM. If the use case requires VCM to be present before
VS with VCM under these same slewing conditions, then a 331-Ω resistor must be added between the VS supply
and the VS pin bypass capacitor.
Power-supply bypass capacitors are required for stability and must be placed as close as possible to the supply
and ground pins of the device. A typical value for this supply bypass capacitor is 0.1 µF. Applications with noisy
or high-impedance power supplies may require additional decoupling capacitors to reject power-supply noise.
During slow power-up events, current flow through the sense resistor or voltage applied to the REF pin can
result in the output voltage momentarily exceeding the voltage at the LIMITx pins, resulting in an erroneous
indication of an out-of-range event on the ALERTx output. When powering the device with a slow ramping power
rail where an input signal is already present, all alert indications should be disregarded until the supply voltage
has reached the final value.
10 Layout
10.1 Layout Guidelines
•
•
•
24
Place the power-supply bypass capacitor as close as possible to the supply and ground pins. The
recommended value of this bypass capacitor is 0.1 µF. Add more decoupling capacitance to compensate
for noisy or high-impedance power supplies.
Connect RLIMIT to the ground pin as directly as possible to limit additional capacitance on this node. If
possible, route this connection to the same plane in order to avoid vias to internal planes. If the connection
cannot be routed on the same plane and must pass through vias, make sure that a path is routed from RLIMIT
back to the ground pin, and that RLIMIT is not simply connected directly to a ground plane.
Pull up the open-drain output pin to the supply voltage rail through a 10-kΩ pullup resistor.
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10.2 Layout Example
RSHUNT
Power
Supply
Load
Alert Output
5
ALERT RESET
6
IN±
7
8
IN+
RPULL-UP
VIA to
Ground
Plane
CBYPASS
1
2
3
4
INA301-Q1
VS
OUT
LIMIT
GND
Supply
Voltage
VIA to
Ground
Plane
RLIMIT
Output Voltage
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Connect the limit resistor directly to the GND pin.
Figure 10-1. Recommended Layout
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11 Device and Documentation Support
11.1 Documentation Support
11.1.1 Related Documentation
INA301EVM User Guide (SBOU154)
11.2 Receiving Notification of Documentation Updates
To receive notification of documentation updates, navigate to the device product folder on ti.com. Click on
Subscribe to updates to register and receive a weekly digest of any product information that has changed. For
change details, review the revision history included in any revised document.
11.3 Support Resources
TI E2E™ support forums are an engineer's go-to source for fast, verified answers and design help — straight
from the experts. Search existing answers or ask your own question to get the quick design help you need.
Linked content is provided "AS IS" by the respective contributors. They do not constitute TI specifications and do
not necessarily reflect TI's views; see TI's Terms of Use.
11.4 Trademarks
TI E2E™ is a trademark of Texas Instruments.
All trademarks are the property of their respective owners.
11.5 Electrostatic Discharge Caution
This integrated circuit can be damaged by ESD. Texas Instruments recommends that all integrated circuits be handled
with appropriate precautions. Failure to observe proper handling and installation procedures can cause damage.
ESD damage can range from subtle performance degradation to complete device failure. Precision integrated circuits may
be more susceptible to damage because very small parametric changes could cause the device not to meet its published
specifications.
11.6 Glossary
TI Glossary
This glossary lists and explains terms, acronyms, and definitions.
12 Mechanical, Packaging, and Orderable Information
The following pages include mechanical, packaging, and orderable information. This information is the most
current data available for the designated devices. This data is subject to change without notice and revision of
this document. For browser-based versions of this data sheet, refer to the left-hand navigation.
26
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Copyright © 2022 Texas Instruments Incorporated
Product Folder Links: INA301-Q1
PACKAGE OPTION ADDENDUM
www.ti.com
21-Jan-2022
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
(4/5)
(6)
INA301A1QDGKRQ1
ACTIVE
VSSOP
DGK
8
2500
RoHS & Green
NIPDAUAG
Level-2-260C-1 YEAR
-40 to 125
ZGG6
INA301A1QDGKTQ1
ACTIVE
VSSOP
DGK
8
250
RoHS & Green
NIPDAUAG
Level-2-260C-1 YEAR
-40 to 125
ZGG6
INA301A2QDGKRQ1
ACTIVE
VSSOP
DGK
8
2500
RoHS & Green
NIPDAUAG
Level-2-260C-1 YEAR
-40 to 125
ZGK6
INA301A2QDGKTQ1
ACTIVE
VSSOP
DGK
8
250
RoHS & Green
NIPDAUAG
Level-2-260C-1 YEAR
-40 to 125
ZGK6
INA301A3QDGKRQ1
ACTIVE
VSSOP
DGK
8
2500
RoHS & Green
NIPDAUAG
Level-2-260C-1 YEAR
-40 to 125
ZGJ6
INA301A3QDGKTQ1
ACTIVE
VSSOP
DGK
8
250
RoHS & Green
NIPDAUAG
Level-2-260C-1 YEAR
-40 to 125
ZGJ6
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of
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