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LM2738
SNVS556C – APRIL 2008 – REVISED JANUARY 2016
LM2738 550-kHz/1.6-MHz 1.5-A Step-Down DC-DC Switching Regulator
1 Features
3 Description
•
The LM2738 regulator is a monolithic, highfrequency, PWM step-down DC-DC converter in an 8pin WSON or 8-pin MSOP-PowerPAD package. It
provides all the active functions for local DC-DC
conversion with fast transient response and accurate
regulation in the smallest possible PCB area.
1
•
•
•
•
•
•
•
•
•
•
Space-Saving WSON and MSOP-PowerPAD™
Packages
3-V to 20-V Input Voltage Range
0.8-V to 18-V Output Voltage Range
1.5-A Output Current
550-kHz (LM2738Y) and 1.6-MHz (LM2738X)
Switching Frequencies
250-mΩ NMOS Switch
400-nA Shutdown Current
0.8-V, 2% Internal Voltage Reference
Internal Soft-Start
Current-Mode, PWM Operation
Thermal Shutdown
2 Applications
•
•
•
•
•
•
Local Point of Load Regulation
Core Power in HDDs
Set-Top Boxes
Battery Powered Devices
USB Powered Devices
DSL Modems
With a minimum of external components, the LM2738
is easy to use. The ability to drive 1.5-A loads with an
internal 250-mΩ NMOS switch using state-of-the-art
0.5-µm BiCMOS technology results in the best power
density available. Switching frequency is internally set
to 550 kHz (LM2738Y) or 1.6 MHz (LM2738X),
allowing the use of extremely small surface-mount
inductors and chip capacitors. Even though the
operating frequencies are very high, efficiencies up to
90% are easy to achieve. External enable is included,
featuring an ultralow standby current of 400 nA. The
LM2738 utilizes current-mode control and internal
compensation to provide high-performance regulation
over a wide range of operating conditions. Additional
features include internal soft-start circuitry to reduce
in-rush current, cycle-by-cycle current limit, thermal
shutdown, and output over-voltage protection.
Device Information(1)
PART
NUMBER
LM2738
PACKAGE
BODY SIZE (NOM)
WSON (8)
3.00 mm × 3.00 mm
MSOP-PowerPAD (8)
3.00 mm × 3.00 mm
(1) For all available packages, see the orderable addendum at
the end of the data sheet.
Typical Application Circuit
Efficiency vs Load Current
VIN = 12 V, VOUT = 3.3 V
D2
VIN
BOOST
VIN
C3
C1
L1
SW
LM2738
ON
OFF
VOUT
D1
EN
C2
R1
FB
GND
R2
1
An IMPORTANT NOTICE at the end of this data sheet addresses availability, warranty, changes, use in safety-critical applications,
intellectual property matters and other important disclaimers. PRODUCTION DATA.
LM2738
SNVS556C – APRIL 2008 – REVISED JANUARY 2016
www.ti.com
Table of Contents
1
2
3
4
5
6
7
Features ..................................................................
Applications ...........................................................
Description .............................................................
Revision History.....................................................
Pin Configuration and Functions .........................
Specifications.........................................................
1
1
1
2
3
4
6.1
6.2
6.3
6.4
6.5
6.6
4
4
4
5
5
6
Absolute Maximum Ratings ......................................
ESD Ratings..............................................................
Recommended Operating Conditions.......................
Thermal Information ..................................................
Electrical Characteristics...........................................
Typical Characteristics ..............................................
Detailed Description ............................................ 10
7.1
7.2
7.3
7.4
Overview .................................................................
Functional Block Diagram .......................................
Feature Description.................................................
Device Functional Modes........................................
10
11
11
14
8
Application and Implementation ........................ 15
8.1 Application Information............................................ 15
8.2 Typical Applications ................................................ 15
9 Power Supply Recommendations...................... 30
10 Layout................................................................... 30
10.1 Layout Guidelines ................................................. 30
10.2 Layout Example .................................................... 31
10.3 Thermal Considerations ........................................ 31
11 Device and Documentation Support ................. 33
11.1
11.2
11.3
11.4
11.5
11.6
Device Support......................................................
Documentation Support ........................................
Community Resources..........................................
Trademarks ...........................................................
Electrostatic Discharge Caution ............................
Glossary ................................................................
33
33
33
33
33
33
12 Mechanical, Packaging, and Orderable
Information ........................................................... 33
4 Revision History
NOTE: Page numbers for previous revisions may differ from page numbers in the current version.
Changes from Revision B (April 2013) to Revision C
•
Added Device Information table, ESD Ratings table, Thermal Information table, Feature Description section, Device
Functional Modes, Application and Implementation section, Power Supply Recommendations section, Layout
section, Device and Documentation Support section, and Mechanical, Packaging, and Orderable Information section ...... 1
Changes from Revision A (April 2013) to Revision B
•
2
Page
Page
Changed layout of National Data Sheet to TI format ........................................................................................................... 29
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5 Pin Configuration and Functions
NGQ Package
8-Pin WSON With Exposed Thermal Pad
Top View
DGN Package
8-Pin MSOP-PowerPAD
Top View
Pin Functions
PIN
TYPE (1)
DESCRIPTION
NO.
NAME
1
BOOST
I
2
VIN
PWR
3
VCC
I
Input supply voltage of the device. Connect a bypass capacitor to this pin. Must tie pins 2
and 3 together at the package.
4
EN
I
Enable control input. Logic high enables operation. Do not allow this pin to float or be greater
than VIN + 0.3 V.
GND
PWR
Signal and power ground pins. Place the bottom resistor of the feedback network as close as
possible to these pins.
5, 7
Boost voltage that drives the internal NMOS control switch. A bootstrap capacitor is
connected between the BOOST and SW pins.
Supply voltage for output power stage. Connect a bypass capacitor to this pin. Must tie pins
2 and 3 together at package.
6
FB
I
Feedback pin. Connect FB to the external resistor divider to set output voltage.
8
SW
O
Output switch. Connects to the inductor, catch diode, and bootstrap capacitor.
GND
—
Signal and power ground. Must be connected to GND on the PCB.
DAP
(1)
I = Input, O = Output, and PWR = Power
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6 Specifications
6.1 Absolute Maximum Ratings
over operating free-air temperature range (unless otherwise noted)
(1) (2)
MIN
MAX
UNIT
VIN, VCC
–0.5
24
V
SW voltage
–0.5
24
V
Boost voltage
–0.5
30
V
Boost to SW voltage
–0.5
6
V
FB voltage
–0.5
3
V
EN voltage
–0.5
VIN + 0.3
V
150
°C
Infrared and convection reflow (15 s)
220
°C
Wave soldering lead temperature (10 s)
260
°C
150
°C
Junction temperature
Soldering information
Storage temperature, Tstg
(1)
(2)
–65
Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. These are stress ratings
only, which do not imply functional operation of the device at these or any other conditions beyond those indicated under Recommended
Operating Conditions. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
If Military or Aerospace specified devices are required, contact the Texas Instruments Sales Office or Distributors for availability and
specifications.
6.2 ESD Ratings
V(ESD)
(1)
(2)
Electrostatic discharge
Human-body model (HBM), per ANSI/ESDA/JEDEC JS-001
(1) (2)
VALUE
UNIT
±2000
V
JEDEC document JEP155 states that 500-V HBM allows safe manufacturing with a standard ESD control process.
Human body model, 1.5 kΩ in series with 100 pF.
6.3 Recommended Operating Conditions
over operating free-air temperature range (unless otherwise noted)
MIN
MAX
3
20
V
SW voltage
–0.5
20
V
Boost voltage
V
VIN, VCC
UNIT
–0.5
25.5
Boost to SW voltage
2.5
5.5
V
Junction temperature
−40
125
°C
165
°C
Thermal shutdown
4
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6.4 Thermal Information
LM2738
THERMAL METRIC
(1)
NGQ (WSON)
DGN (MSOP
PowerPAD)
8 PINS
8 PINS
(2)
UNIT
RθJA
Junction-to-ambient thermal resistance
45.9
50.3
°C/W
RθJC(top)
Junction-to-case (top) thermal resistance
44.6
54.2
°C/W
RθJB
Junction-to-board thermal resistance
13.2
31.4
°C/W
ψJT
Junction-to-top characterization parameter
0.5
4.8
°C/W
ψJB
Junction-to-board characterization parameter
13.2
31.2
°C/W
RθJC(bot)
Junction-to-case (bottom) thermal resistance
5.8
4
°C/W
(1)
(2)
For more information about traditional and new thermal metrics, see the Semiconductor and device Package Thermal Metrics
application report, SPRA953.
Typical thermal shutdown occurs if the junction temperature exceeds 165°C. The maximum power dissipation is a function of TJ(MAX) ,
RθJA and TA . The maximum allowable power dissipation at any ambient temperature is PD = (TJ(MAX) – TA) / RθJA. All numbers apply for
packages soldered directly onto a 3 inches × 3 inches PC board with 2 oz. copper on 4 layers in still air in accordance to JEDEC
standards. Thermal resistance varies greatly with layout, copper thickness, number of layers in PCB, power distribution, number of
thermal vias, board size, ambient temperature, and air flow.
6.5 Electrical Characteristics
All typical limits apply over TJ = 25°C, and all maximum and minimum limits apply over the full operating temperature range
(TJ = –40°C to +125°C). VIN = 12 V, VBOOST – VSW = 5 V unless otherwise specified. Data sheet minimum and maximum
specification limits are ensured by design, test, or statistical analysis.
PARAMETER
TEST CONDITIONS
MIN (1)
TYP (2)
MAX (1)
0.784
0.800
0.816
VFB
Feedback voltage
ΔVFB/ΔVIN
Feedback voltage line regulation
VIN = 3 V to 20 V
IFB
Feedback input bias current
Sink or source
0.1
100
Undervoltage lockout
VIN Rising
2.7
2.9
Undervoltage lockout
VIN Falling
UVLO
0.02
2
UVLO hysteresis
FSW
Switching frequency
DMAX
Maximum duty cycle
1.28
1.6
1.92
LM2738Y
0.364
0.55
0.676
LM2738X , Load = 150 mA
95%
LM2738X
7.5%
LM2738Y
2%
VBOOST – VSW = 3 V, Load = 400 mA
250
ICL
Switch current limit
VBOOST – VSW = 3 V, VIN = 3 V
Boost pin current
500
2.9
mΩ
A
1.9
Non-Switching
1.9
mA
VEN = 0 V
400
nA
LM2738X (27% Duty Cycle)
4.5
LM2738Y (27% Duty Cycle)
2.5
VEN Falling
Enable threshold voltage
VEN Rising
IEN
Enable pin current
Sink / Source
ISW
Switch leakage
VIN = 20 V
(1)
(2)
2
Switching
Shutdown threshold voltage
VEN_TH
MHz
92%
LM2738Y, Load = 150 mA
Switch ON resistance
IBOOST
V
LM2738X
RDS(ON)
Quiescent current (shutdown)
nA
0.4
Minimum duty cycle
Quiescent current
V
%/V
2.3
DMIN
IQ
UNIT
3
mA
mA
0.4
1.4
V
10
nA
100
nA
Ensured to average outgoing quality level (AOQL).
Typicals represent the most likely parametric norm.
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6.6 Typical Characteristics
All curves taken at VIN = 12 V, VBOOST – VSW = 5 V, and TA = 25°C, unless specified otherwise.
VOUT = 5 V
VOUT = 5 V
Figure 1. Efficiency vs Load Current – X Version
VOUT = 3.3 V
VOUT = 3.3 V
Figure 3. Efficiency vs Load Current – X Version
VOUT = 1.5 V
Figure 4. Efficiency vs Load Current – Y Version
VOUT = 1.5 V
Figure 5. Efficiency vs Load Current – X Version
6
Figure 2. Efficiency vs Load Current – Y Version
Figure 6. Efficiency vs Load Current – Y Version
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Typical Characteristics (continued)
All curves taken at VIN = 12 V, VBOOST – VSW = 5 V, and TA = 25°C, unless specified otherwise.
Figure 7. Oscillator Frequency vs Temperature – X Version
Figure 8. Oscillator Frequency vs Temperature – Y Version
VIN = 5 V
Figure 9. Current Limit vs Temperature
Figure 10. IQ Non-Switching vs Temperature
Figure 11. VFB vs Temperature
Figure 12. RDSON vs Temperature
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Typical Characteristics (continued)
All curves taken at VIN = 12 V, VBOOST – VSW = 5 V, and TA = 25°C, unless specified otherwise.
VOUT = 1.5 V
IOUT = 750 mA
VOUT = 1.5 V
Figure 13. Line Regulation – X Version
VOUT = 3.3 V
Figure 14. Line Regulation – Y Version
VOUT = 3.3 V
IOUT = 750 mA
VOUT = 1.5 V
Figure 17. Load Regulation – X Version
8
IOUT = 750 mA
Figure 16. Line Regulation – Y Version
Figure 15. Line Regulation – X Version
VOUT = 1.5 V
IOUT = 750 mA
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Figure 18. Load Regulation – Y Version
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Typical Characteristics (continued)
All curves taken at VIN = 12 V, VBOOST – VSW = 5 V, and TA = 25°C, unless specified otherwise.
VOUT = 3.3 V
VOUT = 3.3 V
Figure 19. Load Regulation – X Version
Figure 20. Load Regulation – Y Version
VOUT = 3.3 V
Figure 22. Load Transient – X Version
Figure 21. IQ Switching vs Temperature
VOUT = 3.3 V
VIN = 12 V
IOUT = 1.5 A
Figure 23. Startup – X Version (Resistive Load)
VIN = 12 V
VOUT = 3.3 V
VIN = 12 V
IOUT = 1.5 A
Figure 24. In-Rush Current – X Version (Resistive Load)
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7 Detailed Description
7.1 Overview
The LM2738 is a constant frequency PWM buck regulator device that delivers a 1.5-A load current. The regulator
has a preset switching frequency of either 550 kHz (LM2738Y) or 1.6 MHz (LM2738X). These high frequencies
allow the LM2738 to operate with small surface-mount capacitors and inductors, resulting in DC-DC converters
that require a minimum amount of board space. The LM2738 is internally compensated, so it is simple to use and
requires few external components. The LM2738 uses current-mode control to regulate the output voltage.
The LM2738 supplies a regulated output voltage by switching the internal NMOS control switch at constant
frequency and variable duty cycle. A switching cycle begins at the falling edge of the reset pulse generated by
the internal oscillator. When this pulse goes low, the output control logic turns on the internal NMOS control
switch. During this on time, the SW pin voltage (VSW) swings up to approximately VIN, and the inductor current
(IL) increases with a linear slope. IL is measured by the current-sense amplifier, which generates an output
proportional to the switch current. The sense signal is summed with the regulator’s corrective ramp and
compared to the error amplifier’s output, which is proportional to the difference between the feedback voltage
and VREF. When the PWM comparator output goes high, the output switch turns off until the next switching cycle
begins. During the switch off-time, inductor current discharges through Schottky diode D1, which forces the SW
pin to swing below ground by the forward voltage (VD) of the catch diode. The regulator loop adjusts the duty
cycle (D) to maintain a constant output voltage. See Functional Block Diagram and Figure 25.
VSW
D = TON/TSW
VIN
SW
Voltage
TOFF
TON
0
VD
IL
t
TSW
IPK
Inductor
Current
t
0
Figure 25. LM2738 Waveforms of SW Pin Voltage and Inductor Current
10
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7.2 Functional Block Diagram
VIN
VIN
Current-Sense Amplifier
EN
OFF
RSENSE
+
-
CIN
D2
Thermal
Shutdown
BOOST
VBOOST
Under
Voltage
Lockout
Current
Limit
Oscillator
Output
Control
Logic
Reset
Pulse
+
ISENSE
+
+
Corrective Ramp
0.25:
Switch
Driver
SW
OVP
Comparator
-
ON
Internal
Regulator
and
Enable
Circuit
Error
Signal
D1
+
PWM
Comparator
CBOOST
VSW L
IL
VOUT
COUT
0.93V
+
-
R1
FB
Internal
Compensation
+
Error Amplifier
+
-
VREF
0.8V
R2
GND
7.3 Feature Description
7.3.1 Boost Function
Capacitor CBOOST and diode D2 in Figure 26 are used to generate a voltage VBOOST. VBOOST – VSW is the gatedrive voltage to the internal NMOS control switch. To properly drive the internal NMOS switch during its on time,
VBOOST must be at least 2.5 V greater than VSW. TI recommends that VBOOST be greater than 2.5 V above VSW for
best efficiency. VBOOST – VSW must not exceed the maximum operating limit of 5.5 V. For best performance, see
Equation 1.
5.5 V > VBOOST – VSW > 2.5 V
(1)
When the LM2738 starts up, internal circuitry from the BOOST pin supplies a maximum of 20 mA to CBOOST. This
current charges CBOOST to a voltage sufficient to turn the switch on. The BOOST pin continues to source current
to CBOOST until the voltage at the feedback pin is greater than 0.76 V.
There are various methods to derive VBOOST:
1. From the input voltage (3 V < VIN < 5.5 V)
2. From the output voltage (2.5 V < VOUT < 5.5 V)
3. From an external distributed voltage rail (2.5 V < VEXT < 5.5 V)
4. From a shunt or series Zener diode
As seen on the Functional Block Diagram, capacitor CBOOST and diode D2 supply the gate-drive voltage for the
NMOS switch. Capacitor CBOOST is charged via diode D2 by VIN. During a normal switching cycle, when the
internal NMOS control switch is off (TOFF) (refer to Figure 25), VBOOST equals VIN minus the forward voltage of D2
(VFD2), during which the current in the inductor (L) forward biases the Schottky diode D1 (VFD1). Therefore the
voltage stored across CBOOST is Equation 2:
VBOOST – VSW = VIN – VFD2 + VFD1
(2)
When the NMOS switch turns on (TON), the switch pin rises to Equation 3:
VSW = VIN – (RDSON × IL),
(3)
forcing VBOOST to rise, thus reverse biasing D2. The voltage at VBOOST is then Equation 4:
VBOOST = 2 VIN – (RDSON × IL) – VFD2 + VFD1
(4)
which is approximately 2 VIN – 0.4 V for many applications. Thus the gate-drive voltage of the NMOS switch is
approximately VIN – 0.2 V.
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Feature Description (continued)
An alternate method for charging CBOOST is to connect D2 to the output as shown in Figure 26. The output
voltage must be between 2.5 V and 5.5 V so that proper gate voltage is applied to the internal switch. In this
circuit, CBOOST provides a gate-drive voltage that is slightly less than VOUT.
VBOOST
D2
BOOST
VIN
VIN
LM2738
CIN
CBOOST
L
SW
VOUT
GND
COUT
D1
Figure 26. VOUT Charges CBOOST
In applications where both VIN and VOUT are greater than 5.5 V, or less than 3 V, CBOOST cannot be charged
directly from these voltages. If VIN and VOUT are greater than 5.5 V, CBOOST can be charged from VIN or VOUT
minus a Zener voltage by placing a Zener diode D3 in series with D2, as shown in Figure 27. When using a
series Zener diode from the input, ensure that the regulation of the input supply does not create a voltage that
falls outside the recommended VBOOST voltage.
(VINMAX – VD3) < 5.5 V
(VINMIN – VD3) > 2.5 V
D2
D3
VIN
VIN
CIN
BOOST
VBOOST
CBOOST
LM2738
L
SW
VOUT
GND
D1
COUT
Figure 27. Zener Reduces Boost Voltage from VIN
An alternative method is to place the Zener diode D3 in a shunt configuration as shown in Figure 28. A small
350-mW to 500-mW 5.1-V Zener in a SOT-23 or SOD package can be used for this purpose. A small ceramic
capacitor such as a 6.3-V, 0.1-µF capacitor (C4) must be placed in parallel with the Zener diode. When the
internal NMOS switch turns on, a pulse of current is drawn to charge the internal NMOS gate capacitance. The
0.1-µF parallel shunt capacitor ensures that the VBOOST voltage is maintained during this time.
12
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Feature Description (continued)
VZ
C4
D2
D3
R3
VIN
BOOST
VIN
CIN
VBOOST
CBOOST
LM2738
L
SW
VOUT
GND
D1
COUT
Figure 28. Boost Voltage Supplied from the Shunt Zener on VIN
Resistor R3 must be selected to provide enough RMS current to the Zener diode (D3) and to the BOOST pin. A
recommended choice for the Zener current (IZENER) is 1 mA. The current IBOOST into the BOOST pin supplies the
gate current of the NMOS control switch and varies typically according to the formula in Equation 5 for the X
version:
IBOOST = 0.56 × (D + 0.54) × (VZENER – VD2) mA
(5)
IBOOST can be calculated for the Y version using Equation 6:
IBOOST = 0.22 × (D + 0.54) × (VZENER – VD2) µA
where
•
•
•
•
•
D is the duty cycle
VZENER and VD2 are in volts
IBOOST is in milliamps
VZENER is the voltage applied to the anode of the boost diode (D2)
VD2 is the average forward voltage across D2
(6)
The formula for IBOOST in Equation 6 gives typical current. For the worst case IBOOST, increase the current by
40%. In that case, the worst case boost current is Equation 7:
IBOOST-MAX = 1.4 × IBOOST
(7)
R3 is then given by Equation 8:
R3 = (VIN – VZENER) / (1.4 × IBOOST + IZENER)
(8)
For example, using the X-version let VIN = 10 V, VZENER = 5 V, VD2 = 0.7 V, IZENER = 1 mA, and duty cycle
D = 50%. Then Equation 9 and Equation 10:
IBOOST = 0.56 × (0.5 + 0.54) × (5 – 0.7) mA = 2.5 mA
R3 = (10 V – 5 V) / (1.4 × 2.5 mA + 1 mA) = 1.11 kΩ
(9)
(10)
7.3.2 Soft-Start
This function forces VOUT to increase at a controlled rate during start-up. During soft-start, the error amplifier’s
reference voltage ramps from 0 V to its nominal value of 0.8 V in approximately 600 µs. This forces the regulator
output to ramp up in a more linear and controlled fashion, which helps reduce in-rush current.
7.3.3 Output Overvoltage Protection
The overvoltage comparator compares the FB pin voltage to a voltage that is 16% higher than the internal
reference VREF. Once the FB pin voltage goes 16% above the internal reference, the internal NMOS control
switch is turned off, which allows the output voltage to decrease toward regulation.
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Feature Description (continued)
7.3.4 Undervoltage Lockout
Undervoltage lockout (UVLO) prevents the LM2738 from operating until the input voltage exceeds 2.7 V (typical).
The UVLO threshold has approximately 400 mV of hysteresis, so the part operates until VIN drops below 2.3 V
(typical). Hysteresis prevents the part from turning off during power up if the VIN ramp-up is non-monotonic.
7.3.5 Current Limit
The LM2738 uses cycle-by-cycle current limiting to protect the output switch. During each switching cycle, a
current limit comparator detects if the output switch current exceeds 2.9 A (typical), and turns off the switch until
the next switching cycle begins.
7.3.6 Thermal Shutdown
Thermal shutdown limits total power dissipation by turning off the output switch when the device junction
temperature exceeds 165°C. After thermal shutdown occurs, the output switch doesn’t turn on until the junction
temperature drops to approximately 150°C.
7.4 Device Functional Modes
7.4.1 Enable Pin and Shutdown Mode
The LM2738 has a shutdown mode that is controlled by the enable pin (EN). When a logic low voltage is applied
to EN, the part is in shutdown mode, and its quiescent current drops to typically 400 nA. The voltage at this pin
must never exceed VIN + 0.3 V.
14
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8 Application and Implementation
NOTE
Information in the following applications sections is not part of the TI component
specification, and TI does not warrant its accuracy or completeness. TI’s customers are
responsible for determining suitability of components for their purposes. Customers must
validate and test their design implementation to confirm system functionality.
8.1 Application Information
The LM2738 operates over a wide range of conditions, which is limited by the ON time of the device. Figure 29
shows the recommended operating area for the X version at the full load (1.5 A) and at 25°C ambient
temperature. The Y version of the LM2738 operates at a lower frequency, and therefore operates over the entire
range of operating voltages.
Figure 29. LM2738X – 1.6 MHz (25°C, Load = 1.5 A)
8.2 Typical Applications
8.2.1 LM2738X Circuit Example 1
D2
VIN
BOOST
VIN
C3
C1
R3
LM2738
ON
OFF
L1
SW
VOUT
D1
EN
C2
R1
FB
GND
R2
Figure 30. LM2738X (1.6 MHz)
VBOOST Derived from VIN
5 V to 1.5 V/1.5 A
8.2.1.1 Design Requirements
The device must be able to operate at any voltage within the Recommended Operating Conditions. The load
current must be defined to properly size the inductor, input, and output capacitors. The inductor must be able to
support the full expected load current as well as the peak current generated from load transients and start-up.
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Typical Applications (continued)
8.2.1.2 Detailed Design Procedure
Table 1. Bill of Materials for Figure 30
PART ID
PART VALUE
PART NUMBER
MANUFACTURER
U1
1.5-A Buck Regulator
LM2738X
Texas Instruments
C1, Input Cap
10 µF, 6.3 V, X5R
C3216X5ROJ106M
TDK
C2, Output Cap
22 µF, 6.3 V, X5R
C3216X5ROJ226M
TDK
C3, Boost Cap
0.1 uF, 16 V, X7R
C1005X7R1C104K
TDK
D1, Catch Diode
0.34 VF Schottky 1.5 A, 30 V
CRS08
Toshiba
D2, Boost Diode
1 VF at 100-mA Diode
BAT54WS
Diodes, Inc.
L1
2.2 µH, 1.9 A,
MSS5131-222ML
Coilcraft
R1
8.87 kΩ, 1%
CRCW06038871F
Vishay
R2
10.2 kΩ, 1%
CRCW06031022F
Vishay
R3
100 kΩ, 1%
CRCW06031003F
Vishay
8.2.1.2.1 Inductor Selection
The duty cycle (D) can be approximated quickly using the ratio of output voltage (VO) to input voltage (VIN), using
Equation 11:
VO
D=
VIN
(11)
The catch diode (D1) forward voltage drop and the voltage drop across the internal NMOS switch must be
included to calculate a more accurate duty cycle. Calculate D by using Equation 12:
VO + VD
D=
VIN + VD - VSW
(12)
VSW can be approximated by Equation 13:
VSW = IOUT × RDSON
(13)
The diode forward drop (VD) can range from 0.3 V to 0.7 V depending on the quality of the diode. The lower the
VD, the higher the operating efficiency of the converter. The inductor value determines the output ripple current.
Lower inductor values decrease the size of the inductor, but increase the output ripple current. An increase in the
inductor value decreases the output ripple current.
One must ensure that the minimum current limit (2 A) is not exceeded, so the peak current in the inductor must
be calculated. The peak current (ILPK) in the inductor is calculated by Equation 14 and Equation 15:
ILPK = IOUT + ΔiL
(14)
Figure 31. Inductor Current
VIN - VOUT 2DiL
=
L
DTS
(15)
In general in Equation 16,
16
ΔiL = 0.1 × (IOUT) → 0.2 × (IOUT)
(16)
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Typical Applications (continued)
If ΔiL = 33.3% of 1.5 A, the peak current in the inductor is 2 A. The minimum specified current limit over all
operating conditions is 2 A. One can either reduce ΔiL, or make the engineering judgment that zero margin is
safe enough. The typical current limit is 2.9 A.
The LM2738 operates at frequencies allowing the use of ceramic output capacitors without compromising
transient response. Ceramic capacitors allow higher inductor ripple without significantly increasing output ripple.
See the Output Capacitor section for more details on calculating output voltage ripple. Now that the ripple current
is determined, the inductance is calculated by Equation 17:
2
æ
1 æ Di ö ö
PCOND = (IOUT 2 ´ D) ç 1 + ´ ç L ÷ ÷ RDSON
ç
3 è IOUT ø ÷
è
ø
where
TS =
•
1
fS
(17)
When selecting an inductor, make sure that it is capable of supporting the peak output current without saturating.
Inductor saturation results in a sudden reduction in inductance and prevents the regulator from operating
correctly. Because of the speed of the internal current limit, the peak current of the inductor need only be
specified for the required maximum output current. For example, if the designed maximum output current is 1 A
and the peak current is 1.25 A, the inductor must be specified with a saturation current limit of > 1.25 A. There is
no must specify the saturation or peak current of the inductor at the 2.9-A typical switch current limit. Because of
the operating frequency of the LM2738, ferrite based inductors are preferred to minimize core losses. This
presents little restriction because of the variety of ferrite-based inductors available. Lastly, inductors with lower
series resistance (RDCR) provide better operating efficiency. For recommended inductors see LM2738X Circuit
Example 1.
8.2.1.2.2 Input Capacitor
An input capacitor is necessary to ensure that VIN does not drop excessively during switching transients. The
primary specifications of the input capacitor are capacitance, voltage, RMS current rating, and equivalent series
inductance (ESL). The recommended input capacitance is 10 µF. The input voltage rating is specifically stated by
the capacitor manufacturer. Make sure to check any recommended deratings and also verify if there is any
significant change in capacitance at the operating input voltage and the operating temperature. The input
capacitor maximum RMS input current rating (IRMS-IN) must be greater than Equation 18:
2DiL ù
é
IRMS _ IN D êIOUT 2 (1- D) +
3 úû
ë
(18)
Neglecting inductor ripple simplifies Equation 18 to Equation 19:
IRMS _ IN = IOUT ´ D(1- D)
(19)
Equation 19 shows that maximum RMS capacitor current occurs when D = 0.5. Always calculate the RMS at the
point where the duty cycle D is closest to 0.5. The ESL of an input capacitor is usually determined by the
effective cross-sectional area of the current path. A large leaded capacitor has high ESL and a 0805 ceramicchip capacitor has very low ESL. At the operating frequencies of the LM2738, leaded capacitors may have an
ESL so large that the resulting impedance (2πfL) is higher than that required to provide stable operation. As a
result, surface-mount capacitors are strongly recommended.
Sanyo POSCAP, Tantalum or Niobium, Panasonic SP, and multilayer ceramic capacitors (MLCC) are all good
choices for both input and output capacitors and have very low ESL. For MLCCs, TI recommends using X7R or
X5R type capacitors due to their tolerance and temperature characteristics. Consult the capacitor manufacturer's
data sheets to see how rated capacitance varies over operating conditions.
8.2.1.2.3 Output Capacitor
The output capacitor is selected based upon the desired output ripple and transient response. The initial current
of a load transient is provided mainly by the output capacitor. The output ripple of the converter is Equation 20:
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Typical Applications (continued)
æ
ö
1
DVOUT = DIL ç RESR +
÷
8 ´ FSW ´ COUT ø
è
(20)
When using MLCCs, the equivalent series resistance (ESR) is typically so low that the capacitive ripple may
dominate. When this occurs, the output ripple is approximately sinusoidal and 90° phase shifted from the
switching action. Given the availability and quality of MLCCs and the expected output voltage of designs using
the LM2738, there is really no must review any other capacitor technologies. Another benefit of ceramic
capacitors is the ability to bypass high-frequency noise. A certain amount of switching edge noise couples
through parasitic capacitances in the inductor to the output. A ceramic capacitor bypasses this noise while a
tantalum capacitor does not. Since the output capacitor is one of the two external components that control the
stability of the regulator control loop, most applications require a minimum of 22 µF of output capacitance.
Capacitance, in general, is often increased when operating at lower duty cycles. Refer to the Circuit Examples for
suggested output capacitances of common applications. Like the input capacitor, recommended multilayer
ceramic capacitors are X7R or X5R types.
8.2.1.2.4 Catch Diode
The catch diode (D1) conducts during the switch off time. A Schottky diode is recommended for its fast switching
times and low forward voltage drop. The catch diode must be chosen so that its current rating is greater than
Equation 21:
ID1 = IOUT × (1-D)
(21)
The reverse breakdown rating of the diode must be at least the maximum input voltage plus appropriate margin.
To improve efficiency, choose a Schottky diode with a low forward-voltage drop.
8.2.1.2.5 Output Voltage
The output voltage is set using Equation 22 and Equation 23 where R2 is connected between the FB pin and
GND, and R1 is connected between VO and the FB pin. A good value for R2 is 10 kΩ. When designing a unity
gain converter (VO = 0.8 V), R1 must be between 0 Ω and 100 Ω, and R2 must not be loaded.
æ V
ö
R1 = ç O - 1÷ ´ R2
V
è REF
ø
(22)
(23)
VREF = 0.80 V
8.2.1.2.6 Calculating Efficiency and Junction Temperature
The complete LM2738 DC-DC converter efficiency can be calculated by Equation 24 or Equation 25:
P
h = OUT
PIN
(24)
or,
h=
POUT
POUT + PLOSS
(25)
Calculations for determining the most significant power losses are shown in Equation 26. Other losses totaling
less than 2% are not discussed.
Power loss (PLOSS) is the sum of two basic types of losses in the converter: switching and conduction.
Conduction losses usually dominate at higher output loads, whereas switching losses remain relatively fixed and
dominate at lower output loads. The first step in determining the losses is to calculate the duty cycle (D):
VOUT + VD
D=
VIN + VD - VSW
(26)
VSW is the voltage drop across the internal NFET when it is on, and is equal to Equation 27:
VSW = IOUT × RDSON
18
(27)
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Typical Applications (continued)
VD is the forward voltage drop across the Schottky catch diode. It can be obtained from the diode manufacturer's
data sheet Electrical Characteristics section. If the voltage drop across the inductor (VDCR) is accounted for, the
equation becomes Equation 28:
VOUT + VD + VDCR
D=
VIN + VD + VDCR - VSW
(28)
The conduction losses in the free-wheeling Schottky diode are calculated by Equation 29:
PDIODE = VD × IOUT × (1-D)
(29)
Often this is the single most significant power loss in the circuit. Care must be taken to choose a Schottky diode
that has a low forward-voltage drop.
Another significant external power loss is the conduction loss in the output inductor. The equation can be
simplified to Equation 30:
PIND = IOUT2 × RDCR
(30)
The LM2738 conduction loss is mainly associated with the internal NFET switch in Equation 31:
2
æ
1 æ DiL ö ÷ö
2
ç
PCOND = (IOUT ´ D) 1 + ´ ç
÷ RDSON
ç
3 è IOUT ø ÷
è
ø
(31)
If the inductor ripple current is fairly small, the conduction losses can be simplified to Equation 32:
PCOND = IOUT2 × RDSON × D
(32)
Switching losses are also associated with the internal NFET switch. They occur during the switch on and off
transition periods, where voltages and currents overlap resulting in power loss. The simplest means to determine
this loss is to empirically measure the rise and fall times (10% to 90%) of the switch at the switch node.
Switching Power Loss is calculated as follows in Equation 33, Equation 34, and Equation 35:
PSWR = 1/2(VIN × IOUT × FSW × TRISE)
PSWF = 1/2(VIN × IOUT × FSW × TFALL)
PSW = PSWR + PSWF
(33)
(34)
(35)
Another loss is the power required for operation of the internal circuitry in Equation 36:
PQ = IQ × VIN
(36)
IQ is the quiescent operating current, and is typically around 1.9 mA for the 0.55-MHz frequency option.
Table 2 lists the power losses for a typical application, and in Equation 37, Equation 38, and Equation 39.
Table 2. Typical Configuration and Power Loss Calculation
PARAMETER
VALUE
POWER PARAMETER
VIN
12 V
—
CALCULATED POWER
—
VOUT
3.3 V
POUT
4.125 W
IOUT
1.25 A
—
—
VD
0.34 V
PDIODE
317 mW
FSW
550 kHz
—
—
IQ
1.9 mA
PQ
22.8 mW
TRISE
8 nS
PSWR
33 mW
TFALL
8 nS
PSWF
33 mW
RDS(ON)
275 mΩ
PCOND
118 mW
INDDCR
70 mΩ
PIND
110 mW
D
0.275
PLOSS
634 mW
η
86.7%
PINTERNAL
207 mW
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ΣPCOND + PSW + PDIODE + PIND + PQ = PLOSS
ΣPCOND + PSWF + PSWR + PQ = PINTERNAL
PINTERNAL = 207 mW
(37)
(38)
(39)
8.2.1.3 Application Curve
VOUT = 5 V
Figure 32. Efficiency vs Load Current – X Version
20
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8.2.2 LM2738X Circuit Example 2
Figure 33. LM2738X (1.6 MHz)
VBOOST Derived from VOUT
12 V to 3.3 V / 1.5 A
8.2.2.1 Detailed Design Procedure
Table 3. Bill of Materials for Figure 33
PART ID
PART VALUE
PART NUMBER
MANUFACTURER
U1
1.5-A Buck Regulator
LM2738X
Texas Instruments
C1, Input Cap
10 µF, 25 V, X7R
C3225X7R1E106M
TDK
C2, Output Cap
33 µF, 6.3 V, X5R
C3216X5ROJ336M
TDK
C3, Boost Cap
0.1 µF, 16 V, X7R
C1005X7R1C104K
TDK
D1, Catch Diode
0.34 VF Schottky 1.5 A, 30 V
CRS08
Toshiba
D2, Boost Diode
1 VF at 100-mA Diode
BAT54WS
Diodes, Inc.
L1
5 µH, 2.9 A
MSS7341- 502NL
Coilcraft
R1
31.6 kΩ, 1%
CRCW06033162F
Vishay
R2
10 kΩ, 1%
CRCW06031002F
Vishay
R3
100 kΩ, 1%
CRCW06031003F
Vishay
8.2.2.2 Application Curve
VOUT = 3.3 V
Figure 34. Efficiency vs Load Current – X Version
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8.2.3 LM2738X Circuit Example 3
C4
D3
R4
D2
BOOST
VIN
VIN
C3
C1
R3
L1
SW
VOUT
LM2738
ON
D1
EN
OFF
C2
R1
FB
GND
R2
Figure 35. LM2738X (1.6 MHz)
VBOOST Derived from VSHUNT
18 V to 1.5 V / 1.5 A
8.2.3.1 Detailed Design Procedure
Table 4. Bill of Materials for Figure 35
PART ID
PART VALUE
PART NUMBER
MANUFACTURER
U1
1.5-A Buck Regulator
LM2738X
Texas Instruments
C1, Input Cap
10 µF, 25 V, X7R
C3225X7R1E106M
TDK
C2, Output Cap
47 µF, 6.3 V, X5R
C3216X5ROJ476M
TDK
C3, Boost Cap
0.1 µF, 16 V, X7R
C1005X7R1C104K
TDK
C4, Shunt Cap
0.1 µF, 6.3 V, X5R
C1005X5R0J104K
TDK
D1, Catch Diode
0.34 VF Schottky 1.5 A, 30 V
CRS08
Toshiba
D2, Boost Diode
1 VF at 100-mA Diode
BAT54WS
Diodes, Inc.
D3, Zener Diode
5.1-V 250-Mw SOT-23
BZX84C5V1
Vishay
L1
2.7 µH, 1.76 A
VLCF5020T-2R7N1R7
TDK
R1
8.87 kΩ, 1%
CRCW06038871F
Vishay
R2
10.2 kΩ, 1%
CRCW06031022F
Vishay
R3
100 kΩ, 1%
CRCW06031003F
Vishay
R4
4.12 kΩ, 1%
CRCW06034121F
Vishay
22
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8.2.3.2 Application Curve
VOUT = 1.5 V
Figure 36. Efficiency vs Load Current – X Version
8.2.4 LM2738X Circuit Example 4
D3
D2
BOOST
VIN
VIN
C1
C3
R3
LM2738
ON
VOUT
D1
EN
OFF
L1
SW
C2
R1
FB
GND
R2
Figure 37. LM2738X (1.6 MHz)
VBOOST Derived from Series Zener Diode (VIN)
15 V to 1.5 V / 1.5 A
8.2.4.1 Detailed Design Procedure
Table 5. Bill of Materials for Figure 37
PART ID
PART VALUE
PART NUMBER
MANUFACTURER
U1
1.5-A Buck Regulator
LM2738X
Texas Instruments
C1, Input Cap
10 µF, 25 V, X7R
C3225X7R1E106M
TDK
C2, Output Cap
47 µF, 6.3 V, X5R
C3216X5ROJ476M
TDK
C3, Boost Cap
0.1 µF, 16 V, X7R
C1005X7R1C104K
TDK
D1, Catch Diode
0.34 VF Schottky 1.5 A, 30 V
CRS08
Toshiba
D2, Boost Diode
1 VF at 100-mA Diode
BAT54WS
Diodes, Inc.
D3, Zener Diode
11-V 350-Mw SOT-23
BZX84C11T
Diodes, Inc.
L1
3.3 µH, 3.5 A
MSS7341-332NL
Coilcraft
R1
8.87 kΩ, 1%
CRCW06038871F
Vishay
R2
10.2 kΩ, 1%
CRCW06031022F
Vishay
R3
100 kΩ, 1%
CRCW06031003F
Vishay
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8.2.5 LM2738X Circuit Example 5
D3
D2
VIN
BOOST
VIN
C3
C1
R3
LM2738
ON
VOUT
D1
EN
OFF
L1
SW
C2
R1
FB
GND
R2
Figure 38. LM2738X (1.6 MHz)
VBOOST Derived from Series Zener Diode (VOUT)
15 V to 9 V / 1.5 A
8.2.5.1 Detailed Design Procedure
Table 6. Bill of Materials for Figure 38
PART ID
PART VALUE
PART NUMBER
MANUFACTURER
U1
1.5-A Buck Regulator
LM2738X
Texas Instruments
C1, Input Cap
10 µF, 25 V, X7R
C3225X7R1E106M
TDK
C2, Output Cap
22 µF, 16 V, X5R
C3216X5R1C226M
TDK
C3, Boost Cap
0.1 µF, 16 V, X7R
C1005X7R1C104K
TDK
D1, Catch Diode
0.34 VF Schottky 1.5 A, 30 V
CRS08
Toshiba
D2, Boost Diode
1 VF at 100-mA Diode
BAT54WS
Diodes, Inc.
D3, Zener Diode
4.3-V 350-mw SOT-23
BZX84C4V3
Diodes, Inc.
L1
6.2 µH, 2.5 A
MSS7341-622NL
Coilcraft
R1
102 kΩ, 1%
CRCW06031023F
Vishay
R2
10.2 kΩ, 1%
CRCW06031022F
Vishay
R3
100 kΩ, 1%
CRCW06031003F
Vishay
24
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8.2.6 LM2738Y Circuit Example 6
D2
VIN
BOOST
VIN
C3
C1
R3
L1
SW
LM2738
ON
VOUT
D1
EN
OFF
C2
R1
FB
GND
R2
Figure 39. LM2738Y (550 kHz)
VBOOST Derived from VIN
5 V to 1.5 V / 1.5 A
8.2.6.1 Detailed Design Procedure
Table 7. Bill of Materials for Figure 39
PART ID
PART VALUE
PART NUMBER
MANUFACTURER
U1
1.5-A Buck Regulator
LM2738Y
Texas Instruments
C1, Input Cap
10 µF, 6.3 V, X5R
C3216X5ROJ106M
TDK
C2, Output Cap
47 µF, 6.3 V, X5R
C3216X5ROJ476M
TDK
C3, Boost Cap
0.1 µF, 16 V, X7R
C1005X7R1C104K
TDK
D1, Catch Diode
0.34 VF Schottky 1.5 A, 30 V
CRS08
Toshiba
D2, Boost Diode
1 VF at 100-mA Diode
BAT54WS
Diodes, Inc.
L1
6.2 µH, 2.5 A,
MSS7341-622NL
Coilcraft
R1
8.87 kΩ, 1%
CRCW06038871F
Vishay
R2
10.2 kΩ, 1%
CRCW06031022F
Vishay
R3
100 kΩ, 1%
CRCW06031003F
Vishay
8.2.6.2 Application Curve
VOUT = 1.5 V
Figure 40. Efficiency vs Load Current – Y Version
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8.2.7 LM2738Y Circuit Example 7
Figure 41. LM2738Y (550 kHz)
VBOOST Derived from VOUT
12 V to 3.3 V / 1.5 A
8.2.7.1 Detailed Design Procedure
Table 8. Bill of Materials for Figure 41
PART ID
PART VALUE
PART NUMBER
MANUFACTURER
U1
1.5-A Buck Regulator
LM2738Y
Texas Instruments
C1, Input Cap
10 µF, 25 V, X7R
C3225X7R1E106M
TDK
C2, Output Cap
47 µF, 6.3 V, X5R
C3216X5ROJ476M
TDK
C3, Boost Cap
0.1 µF, 16 V, X7R
C1005X7R1C104K
TDK
D1, Catch Diode
0.34 VF Schottky 1.5 A, 30 V
CRS08
Toshiba
D2, Boost Diode
1 VF at 100-mA Diode
BAT54WS
Vishay
L1
12 µH, 1.7 A,
MSS7341-123NL
Coilcraft
R1
31.6 kΩ, 1%
CRCW06033162F
Vishay
R2
10 kΩ, 1%
CRCW06031002F
Vishay
R3
100 kΩ, 1%
CRCW06031003F
Vishay
8.2.7.2 Application Curve
VOUT = 3.3 V
Figure 42. Efficiency vs Load Current – Y Version
26
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8.2.8 LM2738Y Circuit Example 8
C4
D3
R4
D2
BOOST
VIN
VIN
C3
C1
R3
L1
SW
VOUT
LM2738
ON
D1
EN
OFF
C2
R1
FB
GND
R2
Figure 43. LM2738Y (550 kHz)
VBOOST Derived from VSHUNT
18 V to 1.5 V / 1.5 A
8.2.8.1 Detailed Design Procedure
Table 9. Bill of Materials for Figure 43
PART ID
PART VALUE
PART NUMBER
MANUFACTURER
U1
1.5-A Buck Regulator
LM2738Y
Texas Instruments
C1, Input Cap
10 µF, 25 V, X7R
C3225X7R1E106M
TDK
C2, Output Cap
(47 µF, 6.3 V, X5R) × 2 = 94 µF
C3216X5ROJ476M
TDK
C3, Boost Cap
0.1 µF, 16 V, X7R
C1005X7R1C104K
TDK
C4, Shunt Cap
0.1 µF, 6.3 V, X5R
C1005X5R0J104K
TDK
D1, Catch Diode
0.34 VF Schottky 1.5 A, 30 V
CRS08
Toshiba
D2, Boost Diode
1 VF at 100-mA Diode
BAT54WS
Diodes, Inc.
D3, Zener Diode
5.1-V 250-Mw SOT-23
BZX84C5V1
Vishay
L1
8.7 µH, 2.2 A
MSS7341-872NL
Coilcraft
R1
8.87 kΩ, 1%
CRCW06038871F
Vishay
R2
10.2 kΩ, 1%
CRCW06031022F
Vishay
R3
100 kΩ, 1%
CRCW06031003F
Vishay
R4
4.12 kΩ, 1%
CRCW06034121F
Vishay
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8.2.8.2 Application Curve
VOUT = 1.5 V
Figure 44. Efficiency vs Load Current – Y Version
8.2.9 LM2738Y Circuit Example 9
D3
D2
BOOST
VIN
VIN
C1
C3
R3
LM2738
ON
VOUT
D1
EN
OFF
L1
SW
C2
R1
FB
GND
R2
Figure 45. LM2738Y (550 kHz)
VBOOST Derived from Series Zener Diode (VIN)
15 V to 1.5 V / 1.5 A
8.2.9.1 Detailed Design Procedure
Table 10. Bill of Materials for Figure 45
PART ID
PART VALUE
PART NUMBER
MANUFACTURER
U1
1.5-A Buck Regulator
LM2738Y
Texas Instruments
C1, Input Cap
10 µF, 25 V, X7R
C3225X7R1E106M
TDK
C2, Output Cap
(47 µF, 6.3 V, X5R) × 2 = 94 µF
C3216X5ROJ476M
TDK
C3, Boost Cap
0.1 µF, 16 V, X7R
C1005X7R1C104K
TDK
D1, Catch Diode
0.34 VF Schottky 1.5 A, 30 V
CRS08
Toshiba
D2, Boost Diode
1 VF at 100-mA Diode
BAT54WS
Diodes, Inc.
D3, Zener Diode
11-V 350-Mw SOT-23
BZX84C11T
Diodes, Inc.
L1
8.7 µH, 2.2 A
MSS7341-872NL
Coilcraft
R1
8.87 kΩ, 1%
CRCW06038871F
Vishay
R2
10.2 kΩ, 1%
CRCW06031022F
Vishay
R3
100 kΩ, 1%
CRCW06031003F
Vishay
28
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8.2.9.2 Application Curve
VOUT = 1.5 V
Figure 46. Efficiency vs Load Current – Y Version
8.2.10 LM2738Y Circuit Example 10
D3
D2
VIN
BOOST
VIN
C3
C1
R3
LM2738
ON
VOUT
D1
EN
OFF
L1
SW
C2
R1
FB
GND
R2
Figure 47. LM2738Y (550 kHz)
VBOOST Derived from Series Zener Diode (VOUT)
15 V to 9 V / 1.5 A
8.2.10.1 Detailed Design Procedure
Table 11. Bill of Materials for Figure 47
PART ID
PART VALUE
PART NUMBER
MANUFACTURER
U1
1.5-A Buck Regulator
LM2738Y
Texas Instruments
C1, Input Cap
10 µF, 25 V, X7R
C3225X7R1E106M
TDK
C2, Output Cap
22 µF, 16 V, X5R
C3216X5R1C226M
TDK
C3, Boost Cap
0.1 µF, 16 V, X7R
C1005X7R1C104K
TDK
D1, Catch Diode
0.34 VF Schottky 1.5 A, 30 V
CRS08
Toshiba
D2, Boost Diode
1 VF at 100-mA Diode
BAT54WS
Diodes, Inc.
D3, Zener Diode
4.3-V 350-mw SOT-23
BZX84C4V3
Diodes, Inc.
L1
15 µH, 2.1 A
SLF7055T150M2R1-3PF
TDK
R1
102 kΩ, 1%
CRCW06031023F
Vishay
R2
10.2 kΩ, 1%
CRCW06031022F
Vishay
R3
100 kΩ, 1%
CRCW06031003F
Vishay
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9 Power Supply Recommendations
The input voltage is rated as 3 V to 20 V. Care must be taken in certain circuit configurations, such as when
VBOOST is derived from VIN, where the requirement that VBOOST – VSW is less than 5.5 V must be observed. Also
for best efficiency, VBOOST must be at least 2.5 V above VSW. The voltage on the enable (EN) pin must not
exceed VIN by more than 0.3 V.
10 Layout
10.1 Layout Guidelines
When planning layout there are a few things to consider when trying to achieve a clean, regulated output. The
most important consideration is the close coupling of the GND connections of the input capacitor and the catch
diode D1. These ground ends must be close to one another and be connected to the GND plane with at least
two through-holes. Place these components as close as possible to the device. Next in importance is the location
of the GND connection of the output capacitor, which must be near the GND connections of CIN and D1. There
must be a continuous ground plane on the bottom layer of a two-layer board except under the switching node
island. The FB pin is a high-impedance node, and take care to make the FB trace short to avoid noise pickup
and inaccurate regulation. The feedback resistors must be placed as close to the device as possible, with the
GND of R1 placed as close to the GND of the device as possible. The VOUT trace to R2 must be routed away
from the inductor and any other traces that are switching. High AC currents flow through the VIN, SW, and VOUT
traces, so they must be as short and wide as possible. However, making the traces wide increases radiated
noise, so the designer must make this trade-off. Radiated noise can be decreased by choosing a shielded
inductor. The remaining components must also be placed as close to the device as possible. See AN-1229
SIMPLE SWITCHER® PCB Layout Guidelines (SNVA054) for further considerations, and the LM2738 demo
board as an example of a four-layer layout.
10.1.1 WSON Package
Figure 48. Internal WSON Connection
For certain high power applications, the PCB land may be modified to a dog-bone shape (see Figure 49). By
increasing the size of ground plane, and adding thermal vias, the RθJA for the application can be reduced.
30
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10.2 Layout Example
Figure 49. 8-Lead WSON PCB Dog Bone Layout
10.3 Thermal Considerations
Heat in the LM2738 due to internal power dissipation is removed through conduction and/or convection.
Conduction: Heat transfer occurs through cross sectional areas of material. Depending on the material, the
transfer of heat can be considered to have poor to good thermal conductivity properties (insulator vs. conductor).
Heat Transfer goes as:
Silicon → package → lead frame → PCB
Convection: Heat transfer is by means of airflow. This could be from a fan or natural convection. Natural
convection occurs when air currents rise from the hot device to cooler air.
Thermal impedance is defined as Equation 40:
DT
Rq =
Power
(40)
Thermal impedance from the silicon junction to the ambient air is defined as Equation 41:
T -T
RqJA = J A
Power
(41)
The PCB size, weight of copper used to route traces and ground plane, and number of layers within the PCB can
greatly effect RθJA. The type and number of thermal vias can also make a large difference in the thermal
impedance. Thermal vias are necessary in most applications. They conduct heat from the surface of the PCB to
the ground plane. Four to six thermal vias must be placed under the exposed pad to the ground plane if the
WSON package is used.
Thermal impedance also depends on the thermal properties due to the application's operating conditions (VIN,
VO, IO and so forth), and the surrounding circuitry.
10.3.1 Silicon Junction Temperature Determination Methods
To accurately measure the silicon temperature for a given application, two methods can be used.
10.3.1.1 Method 1
The first method requires the user to know the thermal impedance of the silicon junction to top case temperature.
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Thermal Considerations (continued)
To clarify:
RθJC is the thermal impedance from all six sides of a device package to silicon junction.
In this data sheet RΦJC is used, allowing the user to measure top case temperature with a small thermocouple
attached to the top case.
RΦJC is approximately 30°C/W for the 8-pin WSON package with the exposed pad. With the internal dissipation
from the efficiency calculation given previously, and the case temperature, RΦJC can be empirically measured on
the bench as Equation 42.
T -T
RF JC = J C
Power
(42)
Therefore in Equation 43:
Tj = (RΦJC × PLOSS) + TC
(43)
From the previous example, shows Equation 44 and Equation 45:
Tj = (RΦJC × PINTERNAL) + TC
Tj = 30°C/W × 0.207 W + TC
(44)
(45)
10.3.1.2 Method 2
The second method can give a very accurate silicon junction temperature.
The first step is to determine RθJA of the application. The LM2738 has overtemperature protection circuitry. When
the silicon temperature reaches 165°C, the device stops switching. The protection circuitry has a hysteresis of
about 15°C. Once the silicon temperature has decreased to approximately 150°C, the device starts to switch
again. Knowing this, the RθJA for any application can be characterized during the early stages of the design one
may calculate the RθJA by placing the PCB circuit into a thermal chamber. Raise the ambient temperature in the
given working application until the circuit enters thermal shutdown. If the SW pin is monitored, it is obvious when
the internal NFET stops switching, indicating a junction temperature of 165°C. Knowing the internal power
dissipation from the above equations, the junction temperature and the ambient temperature RθJA can be
determined with Equation 46.
165° - TA
RqJA =
PINTERNAL
(46)
Once RθJA is determined, the maximum ambient temperature allowed for a desired junction temperature can be
calculated.
An example of calculating RθJA for an application using the Texas Instruments LM2738 WSON demonstration
board is shown in Equation 48.
The four-layer PCB is constructed using FR4 with ½ oz copper traces. The copper ground plane is on the bottom
layer. The ground plane is accessed by two vias. The board measures 3 cm × 3 cm. It was placed in an oven
with no forced airflow. The ambient temperature was raised to 144°C, and at that temperature, the device went
into thermal shutdown.
From the previous example, Equation 47 and Equation 48 shows:
PINTERNAL = 207 mW
(47)
165°C - 144°C
RqJA =
= 102°C/W
207 mW
(48)
If the junction temperature is kept below 125°C, then the ambient temperature cannot go above 109°C, seen in
Equation 49 and Equation 50.
Tj – (RθJA × PLOSS) = TA
125°C – (102°C/W × 207 mW) = 104°C
32
(49)
(50)
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11 Device and Documentation Support
11.1 Device Support
11.1.1 Third-Party Products Disclaimer
TI'S PUBLICATION OF INFORMATION REGARDING THIRD-PARTY PRODUCTS OR SERVICES DOES NOT
CONSTITUTE AN ENDORSEMENT REGARDING THE SUITABILITY OF SUCH PRODUCTS OR SERVICES
OR A WARRANTY, REPRESENTATION OR ENDORSEMENT OF SUCH PRODUCTS OR SERVICES, EITHER
ALONE OR IN COMBINATION WITH ANY TI PRODUCT OR SERVICE.
11.2 Documentation Support
11.2.1 Related Documentation
For related documentation see the following:
AN-1229 SIMPLE SWITCHER® PCB Layout Guidelines (SNVA054)
11.3 Community Resources
The following links connect to TI community resources. Linked contents are provided "AS IS" by the respective
contributors. They do not constitute TI specifications and do not necessarily reflect TI's views; see TI's Terms of
Use.
TI E2E™ Online Community TI's Engineer-to-Engineer (E2E) Community. Created to foster collaboration
among engineers. At e2e.ti.com, you can ask questions, share knowledge, explore ideas and help
solve problems with fellow engineers.
Design Support TI's Design Support Quickly find helpful E2E forums along with design support tools and
contact information for technical support.
11.4 Trademarks
PowerPAD, E2E are trademarks of Texas Instruments.
All other trademarks are the property of their respective owners.
11.5 Electrostatic Discharge Caution
These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
11.6 Glossary
SLYZ022 — TI Glossary.
This glossary lists and explains terms, acronyms, and definitions.
12 Mechanical, Packaging, and Orderable Information
The following pages include mechanical, packaging, and orderable information. This information is the most
current data available for the designated devices. This data is subject to change without notice and revision of
this document. For browser-based versions of this data sheet, refer to the left-hand navigation.
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PACKAGE OPTION ADDENDUM
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10-Dec-2020
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
(4/5)
(6)
LM2738XMY/NOPB
ACTIVE
HVSSOP
DGN
8
1000
RoHS & Green
SN
Level-1-260C-UNLIM
-40 to 125
STDB
LM2738XSD/NOPB
ACTIVE
WSON
NGQ
8
1000
RoHS & Green
SN
Level-3-260C-168 HR
-40 to 125
L237B
LM2738YMY/NOPB
ACTIVE
HVSSOP
DGN
8
1000
RoHS & Green
SN
Level-1-260C-UNLIM
-40 to 125
SJBB
LM2738YSD/NOPB
ACTIVE
WSON
NGQ
8
1000
RoHS & Green
SN
Level-3-260C-168 HR
-40 to 125
L174B
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of
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