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LM3405
SNVS429C – OCTOBER 2006 – REVISED DECEMBER 2016
LM3405 1.6-MHz, 1-A Constant Current Buck Regulator For Powering LEDS
1 Features
3 Description
•
•
The LM3405 is a 1-A constant current buck LED
driver designed to provide a simple, high efficiency
solution for driving high power LEDs. With a 0.205-V
reference voltage feedback control to minimize power
dissipation, an external resistor sets the current as
required for driving various types of LEDs. Switching
frequency is internally set to 1.6 MHz, allowing small
surface mount inductors and capacitors to be used.
The LM3405 uses current-mode control and internal
compensation offering ease of use and predictable,
high performance regulation over a wide range of
operating conditions. With a maximum input voltage
of 15 V, the device can drive up to 3 High-Brightness
LEDs in series at 1-A forward current, with the single
LED forward voltage of approximately 3.7 V.
Additional features include user accessible EN/DIM
pin for enabling and PWM dimming of LEDs, thermal
shutdown, cycle-by-cycle current limit and overcurrent
protection.
1
•
•
•
•
•
•
•
•
•
•
VIN Operating Range of 3 V to 15 V
Drives up to 5 High-Brightness LEDs in Series at
1A
Thin SOT-6 Package
1.6-MHz Switching Frequency
EN/DIM Input for Enabling and PWM Dimming of
LEDs
300-mΩ NMOS Switch
40-nA Shutdown Current at VIN = 5 V
Internally Compensated Current-mode Control
Cycle-by-Cycle Current Limit
Input Voltage UVLO
Overcurrent Protection
Thermal Shutdown
2 Applications
•
•
•
•
Device Information(1)
LED Drivers
Constant Current Sources
Industrial Lighting
LED Flashlights
PART NUMBER
LM3405
PACKAGE
SOT (6)
BODY SIZE (NOM)
2.90 mm × 1.60 mm
(1) For all available packages, see the orderable addendum at
the end of the data sheet.
Typical Application Circuit
Efficiency vs LED Current (VIN = 5 V)
D2
VIN
VIN
BOOST
C3
C1
VOUT
L1
SW
ON
OFF
LM3405
D1
EN/DIM
C2
IF
C4
FB
GND
R1
Copyright © 2016, Texas Instruments Incorporated
Copyright © 2016, Texas Instruments Incorporated
1
An IMPORTANT NOTICE at the end of this data sheet addresses availability, warranty, changes, use in safety-critical applications,
intellectual property matters and other important disclaimers. PRODUCTION DATA.
LM3405
SNVS429C – OCTOBER 2006 – REVISED DECEMBER 2016
www.ti.com
Table of Contents
1
2
3
4
5
6
7
Features ..................................................................
Applications ...........................................................
Description .............................................................
Revision History.....................................................
Pin Configuration and Functions .........................
Specifications.........................................................
1
1
1
2
3
4
6.1
6.2
6.3
6.4
6.5
6.6
4
4
4
5
5
6
Absolute Maximum Ratings .....................................
ESD Ratings..............................................................
Recommended Operating Conditions.......................
Thermal Information ..................................................
Electrical Characteristics...........................................
Typical Characteristics ..............................................
Detailed Description .............................................. 8
7.1
7.2
7.3
7.4
Overview ................................................................... 8
Functional Block Diagram ......................................... 9
Feature Description................................................... 9
Device Functional Modes........................................ 14
8
Application and Implementation ........................ 15
8.1 Application Information............................................ 15
8.2 Typical Applications ................................................ 19
8.3 System Examples ................................................... 21
9 Power Supply Recommendations...................... 24
10 Layout................................................................... 24
10.1 Layout Guidelines ................................................. 24
10.2 Layout Example .................................................... 24
11 Device and Documentation Support ................. 25
11.1
11.2
11.3
11.4
11.5
11.6
Documentation Support ........................................
Receiving Notification of Documentation Updates
Community Resource............................................
Trademarks ...........................................................
Electrostatic Discharge Caution ............................
Glossary ................................................................
25
25
25
25
25
25
12 Mechanical, Packaging, and Orderable
Information ........................................................... 25
4 Revision History
NOTE: Page numbers for previous revisions may differ from page numbers in the current version.
Changes from Revision B (April 2013) to Revision C
Page
•
Added ESD Ratings table, Feature Description section, Device Functional Modes, Application and Implementation
section, Power Supply Recommendations section, Layout section, Device and Documentation Support section, and
Mechanical, Packaging, and Orderable Information section ................................................................................................. 1
•
Deleted Soldering information (220°C, maximum) from Absolute Maximum Ratings............................................................ 4
•
Changed Thermal resistance, θJA, in Thermal Information From: 118°C/W To: 182.9°C/W .................................................. 5
Changes from Revision A (May 2013) to Revision B
•
2
Page
Changed layout of National Semiconductor Data Sheet to TI format .................................................................................. 23
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5 Pin Configuration and Functions
DDC Package
6-Pin SOT
Top View
DDC Package
6-Pin SOT
Pin 1 Identification
BOOST
1
6
SW
1
6
GND
2
5
VIN
2
5
FB
3
4
EN/DIM
3
4
Pin Functions
PIN
I/O
DESCRIPTION
NO.
NAME
1
BOOST
O
Boost voltage that drives the NMOS output switch. A bootstrap capacitor is connected between the
BOOST and SW pins.
2
GND
—
Signal and power ground pin. Place the bottom resistor of the feedback network as close as possible
to this pin.
3
FB
I
Feedback pin. Connect FB to the LED string cathode and an external resistor to ground to set the
LED current.
4
EN/DIM
I
Enable control input. Logic high enables operation. Toggling this pin with a periodic logic square
wave of varying duty cycle at different frequencies controls the brightness of LEDs. Do not allow this
pin to float or be greater than VIN + 0.3 V.
5
VIN
I
Input supply voltage. Connect a bypass capacitor locally from this pin to GND.
6
SW
O
Switch pin. Connect this pin to the inductor, catch diode, and bootstrap capacitor.
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6 Specifications
6.1 Absolute Maximum Ratings
over operating free-air temperature range (unless otherwise noted) (1)
MIN
MAX
UNIT
Input voltage, VIN
–0.5
20
V
SW voltage
–0.5
20
V
Boost voltage
–0.5
26
V
Boost to SW voltage
–0.5
6
V
FB voltage
–0.5
3
V
EN/DIM voltage
–0.5
(VIN + 0.3)
V
150
°C
150
°C
Junction temperature, TJ
Storage temperature, Tstg
(1)
–65
Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. These are stress ratings
only, which do not imply functional operation of the device at these or any other conditions beyond those indicated under Recommended
Operating Conditions. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
6.2 ESD Ratings
VALUE
V(ESD)
(1)
(2)
Electrostatic discharge
Human-body model (HBM), per ANSI/ESDA/JEDEC JS-001
(1)
UNIT
±2000
Charged-device model (CDM), per JEDEC specification JESD22-C101 (2)
V
±1000
JEDEC document JEP155 states that 500-V HBM allows safe manufacturing with a standard ESD control process.
JEDEC document JEP157 states that 250-V CDM allows safe manufacturing with a standard ESD control process.
6.3 Recommended Operating Conditions
MIN
MAX
Input voltage, VIN
3
15
UNIT
V
EN/DIM voltage
0
(VIN + 0.3)
V
Boost to SW voltage
2.5
5.5
V
Junction temperature, TJ
–40
125
°C
4
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6.4 Thermal Information
LM3405
THERMAL METRIC
DDC
(SOT)
(1)
UNIT
6 PINS
RθJA
Junction-to-ambient thermal resistance
182.9
°C/W
RθJC(top)
Junction-to-case (top) thermal resistance
53.4
°C/W
RθJB
Junction-to-board thermal resistance
28.1
°C/W
ψJT
Junction-to-top characterization parameter
1.2
°C/W
ψJB
Junction-to-board characterization parameter
27.7
°C/W
RθJC(bot)
Junction-to-case (bottom) thermal resistance
—
°C/W
(1)
For more information about traditional and new thermal metrics, see the Semiconductor and IC Package Thermal Metrics application
report.
6.5 Electrical Characteristics
VIN = 12 V, typical values are for TJ = 25°C only; minimum and maximum limits apply over the junction temperature (TJ) range
of –40°C to 125°C (unless otherwise noted). Typical values represent the most likely parametric norm, and are provided for
reference purposes only.
PARAMETER
VFB
TEST CONDITIONS
Feedback voltage
MIN
TYP
MAX
UNIT
0.188
0.205
0.22
V
ΔVFB/(ΔVIN×VFB) Feedback voltage line regulation VIN = 3 V to 15 V
IFB
UVLO
Feedback input bias current
Undervoltage lockout
0.01%
Sink or source
VIN rising
VIN falling
1.9
UVLO hysteresis
10
250
2.74
2.95
2.3
0.44
fSW
Switching frequency
DMAX
Maximum duty cycle
VFB = 0 V
RDS(ON)
Switch ON resistance
VBOOST – VSW = 3 V
ICL
Switch current limit
VBOOST – VSW = 3 V, VIN = 3 V
Quiescent current
Switching, VFB = 0.195 V
Quiescent current (shutdown)
VEN/DIM = 0 V
0.3
Enable threshold voltage
VEN/DIM rising
Shutdown threshold voltage
VEN/DIM falling
IEN/DIM
EN/DIM pin current
Sink or source
ISW
Switch leakage
VIN = 15 V
IQ
VEN/DIM_TH
V
1.2
1.6
85%
94%
1.2
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V
V
1.9
MHz
300
600
mΩ
2
2.8
A
1.8
2.8
mA
µA
1.8
V
0.4
V
0.01
µA
0.1
µA
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nA
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6.6 Typical Characteristics
VIN = 12 V, VBOOST – VSW = 5 V, and TA = 25°C (unless otherwise noted).
IF = 1 A
Figure 2. Efficiency vs Input Voltage
Figure 1. Efficiency vs LED Current
IF = 0.7 A
6
IF = 0.35 A
Figure 3. Efficiency vs Input Voltage
Figure 4. Efficiency vs Input Voltage
Figure 5. VFB vs Temperature
Figure 6. Oscillator Frequency vs Temperature
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Typical Characteristics (continued)
VIN = 12 V, VBOOST – VSW = 5 V, and TA = 25°C (unless otherwise noted).
VBOOST – VSW = 3 V
Figure 7. Current Limit vs Temperature
Figure 8. SOT RDS(ON) vs Temperature
VIN = 15 V
Figure 9. Quiescent Current vs Temperature
IF = 0.2 A
Figure 10. Start-Up Response to EN/DIM Signal
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7 Detailed Description
7.1 Overview
The LM3405 device is a PWM, current-mode controlled buck switching regulator designed to provide a simple,
high efficiency solution for driving LEDs with a preset switching frequency of 1.6MHz. This high frequency allows
the LM3405 to operate with small surface mount capacitors and inductors, resulting in LED drivers that only
require a minimum amount of board space. The LM3405 is internally compensated, simple to use, and requires
few external components.
The following sections refer to Functional Block Diagram and to the waveforms in Figure 11. The LM3405
supplies a regulated output current by switching the internal NMOS power switch at constant frequency and
variable duty cycle. A switching cycle begins at the falling edge of the reset pulse generated by the internal
oscillator. When this pulse goes low, the output control logic turns on the internal NMOS power switch. During
this on-time, the SW pin voltage (VSW) swings up to approximately VIN, and the inductor current (IL) increases
with a linear slope. IL is measured by the current sense amplifier, which generates an output proportional to the
switch current. The sense signal is summed with the regulator’s corrective ramp and compared to the error
amplifier’s output, which is proportional to the difference between the feedback voltage and VREF. When the
PWM comparator output goes high, the internal power switch turns off until the next switching cycle begins.
During the switch off-time, inductor current discharges through the catch diode D1, which forces the SW pin to
swing below ground by the forward voltage (VD1) of the catch diode. The regulator loop adjusts the duty cycle (D)
to maintain a constant output current (IF) through the LED, by forcing FB pin voltage to be equal to VREF
(0.205 V).
VSW
D = TON/TSW
VIN
SW
Voltage
TOFF
TON
0
-VD1
IL
t
TSW
ILPK
IF
'iL
Inductor
Current
t
0
Figure 11. SW Pin Voltage and Inductor Current Waveforms of LM3405
8
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7.2 Functional Block Diagram
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7.3 Feature Description
7.3.1 Boost Function
Capacitor C3 and diode D2 in the Functional Block Diagram are used to generate a voltage VBOOST. The voltage
across C3, VBOOST – VSW, is the gate drive voltage to the internal NMOS power switch. To properly drive the
internal NMOS switch during its on-time, VBOOST must be at least 2.5-V greater than VSW. TI recommends a large
value of VBOOST – VSW to achieve better efficiency by minimizing both the internal switch ON resistance (RDS(ON))
and the switch rise and fall times. However, VBOOST – VSW must not exceed the maximum operating limit of 5.5 V.
When the LM3405 starts up, internal circuitry from VIN supplies a 20-mA current to the BOOST pin, flowing out of
the BOOST pin into C3. This current charges C3 to a voltage sufficient to turn the switch on. The BOOST pin
continues to source current to C3 until the voltage at the feedback pin is greater than 123 mV.
There are various methods to derive VBOOST:
1. From the input voltage (VIN)
2. From the output voltage (VOUT)
3. From a shunt or series Zener diode
4. From an external distributed voltage rail (VEXT)
The first method is shown in Functional Block Diagram. Capacitor C3 is charged through diode D2 by VIN. During
a normal switching cycle, when the internal NMOS power switch is off, TOFF (see Figure 11), VBOOST equals VIN
minus the forward voltage of D2 (VD2), during which the current in the inductor (L1) forward biases the catch
diode D1 (VD1). Therefore, the gate drive voltage stored across C3 is shown in Equation 1.
VBOOST – VSW = VIN – VD2 + VD1
(1)
When the NMOS switch turns on (TON), the switch pin rises to Equation 2.
VSW = VIN – (RDS(ON) × IL)
(2)
Because the voltage across C3 remains unchanged, VBOOST is forced to rise thus reverse biasing D2. The
voltage at VBOOST is then calculated with Equation 3.
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Feature Description (continued)
VBOOST = 2VIN – (RDS(ON) × IL) – VD2 + VD1
(3)
Depending on the quality of the diodes D1 and D2, the gate drive voltage in this method can be slightly less or
larger than the input voltage VIN. For best performance, ensure that the variation of the input supply does not
cause the gate drive voltage to fall outside the recommended range in Equation 4.
2.5 V < VIN – VD2 + VD1 < 5.5 V
(4)
The second method for deriving the boost voltage is to connect D2 to the output as shown in Figure 12. The gate
drive voltage in this configuration is shown in Equation 5.
VBOOST – VSW = VOUT – VD2 + VD1
(5)
Because the gate drive voltage must be in the range of 2.5 V to 5.5 V, the output voltage VOUT must be limited to
a certain range. For the calculation of VOUT, see Output Voltage.
Copyright © 2016, Texas Instruments Incorporated
Figure 12. VBOOST Derived from VOUT
The third method can be used in the applications where both VIN and VOUT are greater than 5.5 V. In these
cases, C3 cannot be charged directly from these voltages; instead C3 can be charged from VIN or VOUT minus a
Zener voltage (VD3) by placing a Zener diode D3 in series with D2 as shown in Figure 13. When using a series
Zener diode from the input, the gate drive voltage is VIN – VD3 – VD2 + VD1.
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Figure 13. VBOOST Derived from VIN Through a Series Zener
10
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Feature Description (continued)
An alternate method is to place the Zener diode D3 in a shunt configuration as shown in Figure 14. A small,
350-mW to 500-mW, 5.1-V Zener in a SOT or SOD package can be used for this purpose. A small ceramic
capacitor such as a 6.3-V, 0.1-µF capacitor (C5) must be placed in parallel with the Zener diode. When the
internal NMOS switch turns on, a pulse of current is drawn to charge the internal NMOS gate capacitance. The
0.1-µF parallel shunt capacitor ensures that the VBOOST voltage is maintained during this time. Resistor R2 must
be chosen to provide enough RMS current to the Zener diode and to the BOOST pin. TI's recommended choice
for the Zener current (IZENER) is 1 mA. The current IBOOST into the BOOST pin supplies the gate current of the
NMOS power switch. It reaches a maximum of around 3.6 mA at the highest gate drive voltage of 5.5 V over the
LM3405 operating range.
For the worst case IBOOST, increase the current by 50%. In that case, the maximum boost current is Equation 6.
IBOOST-MAX = 1.5 × 3.6 mA = 5.4 mA
(6)
R2 is calculated with Equation 7.
R2 = (VIN – VZENER) / (IBOOST_MAX + IZENER)
(7)
For example, let VIN = 12 V, VZENER = 5V, IZENER = 1 mA, then calculate Equation 8.
R2 = (12 V – 5 V) / (5.4 mA + 1 mA) = 1.09 kΩ
(8)
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Figure 14. VBOOST Derived from VIN Through a Shunt Zener
The fourth method can be used in an application which has an external low voltage rail, VEXT. C3 can be charged
through D2 from VEXT, independent of VIN and VOUT voltage levels. Again for best performance, ensure that the
gate drive voltage, VEXT – VD2 + VD1, falls in the range of 2.5 V to 5.5 V.
7.3.2 Setting the LED Current
LM3405 is a constant current buck regulator. The LEDs are connected between VOUT and the FB pin as shown in
the Typical Applications. The FB pin is at 0.205 V in regulation and therefore the LED current IF is set by VFB and
resistor R1 from FB to ground by Equation 9.
IF = VFB / R1
(9)
IF must not exceed the 1-A current capability of LM3405 and, therefore, R1 minimum must be approximately
0.2 Ω. IF must also be kept above 200 mA for stable operation, and therefore R1 maximum must be
approximately 1 Ω. If average LED currents less than 200 mA are desired, the EN/DIM pin can be used for PWM
dimming. See LED PWM Dimming.
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Feature Description (continued)
7.3.3 Output Voltage
The output voltage is primarily determined by the number of LEDs (n) connected from VOUT to FB pin and
therefore VOUT can be calculated with Equation 10.
VOUT = ((n × VF) + VFB)
where
•
VF is the forward voltage of one LED at the set LED current level (see LED manufacturer data sheet for
forward characteristics curve)
(10)
7.3.4 Enable Mode or Shutdown Mode
The LM3405 has both enable and shutdown modes that are controlled by the EN/DIM pin. Connecting a voltage
source greater than 1.8 V to the EN/DIM pin enables the operation of LM3405, while reducing this voltage below
0.4 V places the part in a low quiescent current (0.3 µA typical) shutdown mode. There is no internal pullup on
EN/DIM pin, therefore an external signal is required to initiate switching. Do not allow this pin to float or rise to
0.3 V above VIN. It must be noted that when the EN/DIM pin voltage rises above 1.8 V while the input voltage is
greater than UVLO, there is a finite delay before switching starts. During this delay, the LM3405 goes through a
power on reset state after which the internal soft-start process commences. The soft-start process limits the
inrush current and brings up the LED current (IF) in a smooth and controlled fashion. The total combined duration
of the power on reset delay, soft-start delay and the delay to fully establish the LED current is in the order of
100 µs (see Figure 19).
The simplest way to enable the operation of LM3405 is to connect the EN/DIM pin to VIN which allows self startup of LM3405 whenever the input voltage is applied. However, when an input voltage of slow rise time is used to
power the application and if both the input voltage and the output voltage are not fully established before the softstart time elapses, the control circuit commands maximum duty cycle operation of the internal power switch to
bring up the output voltage rapidly. When the feedback pin voltage exceeds 0.205 V, the duty cycle has to
reduce from the maximum value accordingly, to maintain regulation. It takes a finite amount of time for this
reduction of duty cycle and this results in a spike in LED current for a short duration as shown in Figure 15. In
applications where this LED current overshoot is undesirable, EN/DIM pin voltage can be separately applied and
delayed such that VIN is fully established before the EN/DIM pin voltage reaches the enable threshold. The effect
of delaying EN/DIM with respect to VIN on the LED current is shown in Figure 16. For a fast rising input voltage
(200 µs for example), there is no need to delay the EN/DIM signal, because soft-start can smoothly bring up the
LED current as shown in Figure 17.
Figure 15. Start-Up Response to VIN With 5-ms Rise Time
12
Figure 16. Start-Up Response to VIN With EN/DIM Delayed
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Feature Description (continued)
Figure 17. Start-Up Response to VIN With 200-µs Rise Time
7.3.5 LED PWM Dimming
The LED brightness can be controlled by applying a periodic pulse signal to the EN/DIM pin and varying its
frequency and/or duty cycle. This so-called PWM dimming method controls the average light output by pulsing
the LED current between the set value and zero. A logic high level at the EN/DIM pin turns on the LED current
whereas a logic low level turns off the LED current. Figure 18 shows a typical LED current waveform in PWM
dimming mode. As explained in the previous section, there is approximately a 100-µs delay from the EN/DIM
signal going high to fully establishing the LED current as shown in Figure 19. This 100-µs delay sets a maximum
frequency limit for the driving signal that can be applied to the EN/DIM pin for PWM dimming. Figure 20 shows
the average LED current versus duty cycle of PWM dimming signal for various frequencies. The applicable
frequency range to drive LM3405 for PWM dimming is from 100 Hz to 5 kHz. The dimming ratio reduces
drastically when the applied PWM dimming frequency is greater than 5 kHz.
Figure 18. PWM Dimming of LEDs
Using the EN/DIM Pin
Figure 19. Start-Up Response to EN/DIM
With IF = 1 A
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Feature Description (continued)
Figure 20. Average LED Current vs
Duty Cycle of PWM Dimming Signal at EN/DIM Pin
7.3.6 Undervoltage Lockout
Undervoltage lockout (UVLO) prevents the LM3405 from operating until the input voltage exceeds 2.74 V
(typical). The UVLO threshold has approximately 440 mV of hysteresis, so the part operates until VIN drops
below 2.3 V (typical). Hysteresis prevents the part from turning off during power up if VIN is non-monotonic.
7.3.7 Current Limit
The LM3405 uses cycle-by-cycle current limit to protect the internal power switch. During each switching cycle, a
current limit comparator detects if the power switch current exceeds 2 A (typical), and turns off the switch until
the next switching cycle begins.
7.3.8 Overcurrent Protection
The LM3405 has a built-in overcurrent comparator that compares the FB pin voltage to a threshold voltage that is
60% higher than the internal reference VREF. Once the FB pin voltage exceeds this threshold level (typically
328 mV), the internal NMOS power switch is turned off, which allows the feedback voltage to decrease towards
regulation. This threshold provides an upper limit for the LED current. LED current overshoot is limited to 328
mV/R1 by this comparator during transients.
7.4 Device Functional Modes
7.4.1 Thermal Shutdown
Thermal shutdown limits total power dissipation by turning off the internal power switch when the IC junction
temperature exceeds 165°C. After thermal shutdown occurs, the power switch does not turn on until the junction
temperature drops below approximately 150°C.
14
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8 Application and Implementation
NOTE
Information in the following applications sections is not part of the TI component
specification, and TI does not warrant its accuracy or completeness. TI’s customers are
responsible for determining suitability of components for their purposes. Customers should
validate and test their design implementation to confirm system functionality.
8.1 Application Information
8.1.1 Inductor (L1)
The duty cycle (D) can be approximated quickly using the ratio of output voltage (VOUT) to input voltage (VIN) in
Equation 11.
D=
VOUT
VIN
(11)
The catch diode (D1) forward voltage drop and the voltage drop across the internal NMOS must be included to
calculate a more accurate duty cycle. Calculate D by using Equation 12.
VOUT + VD1
D=
VIN + VD1 - VSW
(12)
VSW can be approximated by Equation 13.
VSW = IF × RDS(ON)
(13)
The diode forward drop (VD1) can range from 0.3 V to 0.7 V depending on the quality of the diode. The lower VD1
is, the higher the operating efficiency of the converter.
The inductor value determines the output ripple current (ΔiL, as defined in Figure 11). Lower inductor values
decrease the size of the inductor, but increases the output ripple current. An increase in the inductor value
decreases the output ripple current. The ratio of ripple current to LED current is optimized when it is set between
0.3 and 0.4 at 1A LED current. This ratio r is defined as:
r=
'iL
lF
(14)
One must also ensure that the minimum current limit (1.2 A) is not exceeded, so the peak current in the inductor
must be calculated. The peak current (ILPK) in the inductor is calculated with Equation 15.
ILPK = IF + ΔiL/2
(15)
When the designed maximum output current is reduced, the ratio (r) can be increased. At a current of 0.2 A,
r can be made as high as 0.7. The ripple ratio can be increased at lighter loads because the net ripple is actually
quite low, and if r remains constant the inductor value can be made quite large. An equation empirically
developed for the maximum ripple ratio at any current below 2 A is calculated with Equation 16 (note that this is
just a guideline).
r = 0.387 × IOUT–0.3667
(16)
The LM3405 operates at a high frequency allowing the use of ceramic output capacitors without compromising
transient response. Ceramic capacitors allow higher inductor ripple without significantly increasing LED current
ripple. See the output capacitor and feed-forward capacitor sections for more details on LED current ripple.
Now that the ripple current or ripple ratio is determined, the inductance is calculated by Equation 17.
L=
VOUT + VD1
IF x r x fSW
x (1-D)
where
•
•
fSW is the switching frequency
IF is the LED current
(17)
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Application Information (continued)
When selecting an inductor, make sure that it is capable of supporting the peak output current without saturating.
Inductor saturation results in a sudden reduction in inductance and prevent the regulator from operating correctly.
Because of the operating frequency of the LM3405, ferrite based inductors are preferred to minimize core losses.
This presents little restriction, because the variety of ferrite based inductors is huge. Lastly, inductors with lower
series resistance (DCR) provides better operating efficiency. For recommended inductor selection, see Circuit
Examples and Recommended Inductance Range in Table 1.
Table 1. Recommended Inductance Range
IF
INDUCTANCE RANGE AND INDUCTOR CURRENT RIPPLE
4.7 µH TO 10 µH
1A
Inductance
4.7 µH
6.8 µH
10 µH
ΔiL / IF (1)
51%
35%
24%
6.8 µH TO 15 µH
0.6 A
Inductance
6.8 µH
10 µH
15 µH
ΔiL / IF (1)
58%
40%
26%
4.7 µH (2) TO 22 µH
0.2 A
(1)
(2)
Inductance
10 µH
15 µH
22 µH
ΔiL / IF (1)
119%
79%
54%
Maximum over full range of VIN and VOUT.
Small inductance improves stability without causing a significant increase in LED current ripple.
8.1.2 Input Capacitor (C1)
An input capacitor is necessary to ensure that VIN does not drop excessively during switching transients. The
primary specifications of the input capacitor are capacitance, voltage rating, RMS current rating, and ESL
(Equivalent Series Inductance). The input voltage rating is specifically stated by the capacitor manufacturer.
Check any recommended deratings and also verify if there is any significant change in capacitance at the
operating input voltage and the operating temperature. The input capacitor maximum RMS input current rating
(IRMS-IN) must be greater than Equation 18.
IRMS-IN = IF x
r2
Dx 1-D+
12
(18)
Equation 18 shows that maximum RMS capacitor current occurs when D = 0.5. Always calculate the RMS at the
point where the duty cycle D, is closest to 0.5. The ESL of an input capacitor is usually determined by the
effective cross sectional area of the current path. A large-leaded capacitor has high ESL and an 0805 ceramic
chip capacitor has very low ESL. At the operating frequency of the LM3405, certain capacitors may have an ESL
so large that the resulting inductive impedance (2 πfL) is higher than that required to provide stable operation. TI
strongly recommends using ceramic capacitors due to their low ESR and low ESL. A 10-µF multilayer ceramic
capacitor (MLCC) is a good choice for most applications. In cases where large capacitance is required, use
surface mount capacitors such as Tantalum capacitors and place at least a 1-µF ceramic capacitor close to the
VIN pin. For MLCCs, TI recommends using X7R or X5R dielectrics. Consult capacitor manufacturer datasheet to
see how rated capacitance varies over operating conditions.
8.1.3 Output Capacitor (C2)
The output capacitor is selected based upon the desired reduction in LED current ripple. A 1-µF ceramic
capacitor results in very low LED current ripple for most applications. Due to the high switching frequency, the
1-µF capacitor alone (without feed-forward capacitor C4) can filter more than 90% of the inductor current ripple
for most applications where the sum of LED dynamic resistance and R1 is larger than 1 Ω. Because the internal
compensation is tailored for small output capacitance with very low ESR, TI strongly recommends using a
ceramic capacitor with capacitance less than 3.3 µF.
16
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Given the availability and quality of MLCCs and the expected output voltage of designs using the LM3405, there
is really no need to review other capacitor technologies. A benefit of ceramic capacitors is their ability to bypass
high frequency noise. A certain amount of switching edge noise couples through the parasitic capacitances in the
inductor to the output. A ceramic capacitor bypasses this noise. In cases where large capacitance is required,
use Electrolytic or Tantalum capacitors with large ESR, and verify the loop performance on the bench. Like the
input capacitor, multilayer ceramic capacitors are recommended X7R or X5R. Again, verify actual capacitance at
the desired operating voltage and temperature.
Check the RMS current rating of the capacitor. The maximum RMS current rating of the capacitor is calculated
with Equation 19.
IRMS-OUT = IF x
r
12
(19)
One may select a 1206 size ceramic capacitor for C2, because its current rating is typically higher than 1 A,
more than enough for the requirement.
8.1.4 Feed-Forward Capacitor (C4)
The feed-forward capacitor (designated as C4) connected in parallel with the LED string is required to provide
multiple benefits to the LED driver design. It greatly improves the large signal transient response and suppresses
LED current overshoot that may otherwise occur during PWM dimming; it also helps to shape the rise and fall
times of the LED current pulse during PWM dimming thus reducing EMI emission; it reduces LED current ripple
by bypassing some of inductor ripple from flowing through the LED. For most applications, a 1-µF ceramic
capacitor is sufficient. In fact, the combination of a 1-µF feed-forward ceramic capacitor and a 1-µF output
ceramic capacitor leads to less than 1% current ripple flowing through the LED. Lower and higher C4 values can
be used, but bench validation is required to ensure the performance meets the application requirement.
Figure 21 shows a typical LED current waveform during PWM dimming without feed-forward capacitor. At the
beginning of each PWM cycle, overshoot can be seen in the LED current. Adding a 1-µF feed-forward capacitor
can totally remove the overshoot as shown in Figure 22.
Figure 21. PWM Dimming Without Feed-Forward Capacitor
Figure 22. PWM Dimming With a 1-µF Feed-Forward
Capacitor
8.1.5 Catch Diode (D1)
The catch diode (D1) conducts during the switch off-time. A Schottky diode is required for its fast switching time
and low forward voltage drop. The catch diode must be chosen such that its current rating is greater than
Equation 20.
ID1 = IF × (1-D)
(20)
The reverse breakdown rating of the diode must be at least the maximum input voltage plus appropriate margin.
To improve efficiency, choose a Schottky diode with a low forward voltage drop.
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8.1.6 Boost Diode (D2)
TI recommends a standard diode such as the 1N4148 type. For VBOOST circuits derived from voltages less than
3.3 V, a small-signal Schottky diode is recommended for better efficiency. A good choice is the BAT54 small
signal diode.
8.1.7 Boost Capacitor (C3)
A 0.01-µF ceramic capacitor with a voltage rating of at least 6.3 V is sufficient. The X7R and X5R MLCCs
provide the best performance.
8.1.8 Power Loss Estimation
The main power loss in LM3405 includes three basic types of loss in the internal power switch: conduction loss,
switching loss, and gate charge loss. In addition, there is loss associated with the power required for the internal
circuitry of IC.
The conduction loss is calculated with Equation 21.
2
§
1 § 'iL · ¸·
u¨
PCOND
IF2 u D u ¨ 1
¸ u RDS(ON)
¨ 3 © IF ¹ ¸
©
¹
(21)
If the inductor ripple current is fairly small (for example, less than 40%), the conduction loss can be simplified
with Equation 22.
PCOND = IF2 × RDS(ON) × D
(22)
The switching loss occurs during the switch on and off transition periods, where voltage and current overlap
resulting in power loss. The simplest means to determine this loss is to empirically measure the rise and fall
times (10% to 90%) of the voltage at the switch pin.
Switching power loss is calculated with Equation 23.
PSW = 0.5 × VIN × IF × fSW × ( TRISE + TFALL )
(23)
The gate charge loss is associated with the gate charge QG required to drive the switch with Equation 24.
PG = fSW × VIN × QG
(24)
The power loss required for operation of the internal circuitry is calculated with Equation 25.
PQ = IQ × VIN
(25)
IQ is the quiescent operating current, and is typically around 1.8mA for the LM3405.
The total power loss in the IC is Equation 26.
PINTERNAL = PCOND + PSW + PG + PQ
(26)
An example of power losses for a typical application is shown in Table 2, Equation 27, and Equation 28 (D is
calculated to be 0.36).
Table 2. Power Loss Tabulation
CONDITIONS
18
POWER LOSS
VIN
12 V
—
—
VOUT
3.9 V
—
—
IOUT
1A
—
—
VD1
0.45 V
—
—
RDS(ON)
300 mΩ
PCOND
111 mW
fSW
1.6 MHz
—
—
TRISE
18 ns
TFALL
12 ns
PSW
288 mW
IQ
1.8 mA
PQ
22 mW
QG
1.4 nC
PG
27 mW
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Σ ( PCOND + PSW + PQ + PG ) = PINTERNAL
PINTERNAL = 448 mW
(27)
(28)
8.2 Typical Applications
8.2.1 VBOOST Derived from VIN (VIN = 5 V, IF = 1 A)
D2
VIN
VIN
BOOST
C3
C1
L1
VOUT
SW
LM3405
D1
DC or
PWM
C2
EN/DIM
C4
IF
LED1
FB
GND
R1
Copyright © 2016, Texas Instruments Incorporated
Figure 23. VBOOST Derived from VIN
(VIN = 5 V, IF = 1 A) Diagram
8.2.1.1 Design Requirements
The following are the parameter specifications for this design example:
• Input voltage, VIN = 5 V ± 10%
• LED current, IF = 1 A
• LED forward voltage, VLED = 3.4 V
• Output voltage, VOUT = 3.4 V + 0.2 V = 3.6 V
• Ripple ratio = r < 0.6
• PWM dimmable
8.2.1.2 Detailed Design Procedure
8.2.1.2.1 Calculate Duty Cycle (D)
Calculate the nominal duty cycle for calculations and ensure the maximum duty cycle is not exceeded in the
application using Equation 29.
D=
VOUT 3.6V
=
= 0.72
5V
VIN
(29)
Using the same equation DMAX can be calculated for the minimum input voltage of 4.5 V. The duty cycle at 4.5 V
is 0.8 which is less than the minimum DMAX of 0.85 specified in Electrical Characteristics.
8.2.1.2.2 Choose Capacitor Values (C1, C2, C3, and C4)
Low input voltage applications and PWM dimming applications generally require more input capacitance so the
higher value of C1 = 10 µF is chosen for best performance. The other capacitor values chosen are the
recommended values of C2 = C4 = 1 µF and C3 = 0.01 µF. All capacitors chosen are X5R or X7R dielectric
ceramic capacitors of sufficient voltage rating.
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Typical Applications (continued)
8.2.1.2.3 Set the Nominal LED Current (R1)
The nominal LED current at 100% PWM dimming duty cycle is set by the resistor R1. R1 can be calculated using
Equation 30.
R1 =
VFB
0.205V
=
= 0.205
1A
IF
(30)
The standard value of R1 = 0.2 Ω is chosen. R1 must have a power rating of at least 1/4 W.
8.2.1.2.4 Choose Diodes (D1 and D2)
For the boost diode, D2, choose a low current diode with a voltage rating greater than the input voltage to give
some margin. D2 must also be a schottky to minimize the forward voltage drop. For this example a schottky
diode of D2 = 100 mA, 30 V is chosen. The catch diode, D1, must be a schottky diode and must have a voltage
rating greater than the input voltage and a current rating greater than the average current. The average current in
D1 can be calculated with Equation 31.
ID1 = IF × :1 - D; = 1A × :1 - 0.72; = 0.28A
(31)
For this example D1 = 1 A, 10 V is chosen.
8.2.1.2.5 Calculate the Inductor Value (L1)
The inductor value is chosen for a given ripple ratio (r). To calculate L1 the forward voltage of D1 is required. In
this case the chosen diode has a forward voltage drop of VF = 0.37 V. Given the desired ripple ratio L1 is
calculated with Equation 32.
L=
VOUT + VD1
3.6V + 0.37V
=
= 4.1
IF × r × fSW
1A × 0.6 × 1.6MHz
H
(32)
The next larger standard value of L1 = 4.7 µH is chosen. A ripple ratio of 0.6 translates to a ΔiL of 600 mA and a
peak inductor current of 1.3 A (IF + ΔiL/2). Choose an inductor with a saturation current rating of greater than
1.3 A.
Table 3. Bill of Materials for Figure 23
PART ID
PART VALUE
PART NUMBER
MANUFACTURER
U1
1-A LED Driver
LM3405
Texas Instruments
C1, Input capacitor
10 µF, 6.3 V, X5R
C3216X5R0J106M
TDK
C2, Output capacitor
1 µF, 10 V, X7R
GRM319R71A105KC01D
Murata
C3, Boost capacitor
0.01 µF, 16 V, X7R
0805YC103KAT2A
AVX
C4, Feedforward capacitor
1 µF, 10 V, X7R
GRM319R71A105KC01D
Murata
D1, Catch diode
Schottky, 0.37 V at 1A, VR = 10 V
MBRM110LT1G
ON Semiconductor
D2, Boost diode
Schottky, 0.36 V at 15 mA
CMDSH-3
Central Semiconductor
L1
4.7 µH, 1.6 A
SLF6028T-4R7M1R6
TDK
R1
0.2 Ω, 0.5 W, 1%
WSL2010R2000FEA
Vishay
20
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8.2.1.3 Application Curve
Figure 24. Efficiency vs Input Voltage
8.3 System Examples
8.3.1 VBOOST Derived From VOUT (VIN = 12 V, IF = 1 A)
D2
VIN
VIN
BOOST
C3
C1
L1
VOUT
SW
LM3405
C2
D1
DC or
PWM
EN/DIM
C4
IF
LED1
FB
GND
R1
Copyright © 2016, Texas Instruments Incorporated
Figure 25. VBOOST Derived From VOUT
(VIN = 12 V, IF = 1 A) Diagram
8.3.1.1 Bill of Materials
Table 4. Bill of Materials for Figure 25
PART ID
PART VALUE
PART NUMBER
MANUFACTURER
U1
1-A LED Driver
LM3405
Texas Instruments
C1, Input capacitor
10 µF, 25 V, X5R
ECJ-3YB1E106K
Panasonic
C2, Output capacitor
1 µF, 10 V, X7R
GRM319R71A105KC01D
Murata
C3, Boost capacitor
0.01 µF, 16 V, X7R
0805YC103KAT2A
AVX
C4, Feedforward capacitor
1 µF, 10 V, X7R
GRM319R71A105KC01D
Murata
D1, Catch diode
Schottky, 0.5 V at 1 A, VR = 30 V
SS13
Vishay
D2, Boost diode
Schottky, 0.36 V at 15 mA
CMDSH-3
Central Semiconductor
L1
4.7 µH, 1.6 A
SLF6028T-4R7M1R6
TDK
R1
0.2 Ω, 0.5 W, 1%
WSL2010R2000FEA
Vishay
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8.3.2 VBOOST Derived From VIN Through a Series Zener Diode, D3 (VIN = 15 V, IF = 1 A)
D3
D2
BOOST
VIN
VIN
C3
C1
VOUT
L1
SW
LM3405
DC or
PWM
D1
C2
C4
EN/DIM
IF
LED1
FB
GND
R1
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Figure 26. VBOOST Derived From VIN Through a Series Zener Diode, D3
(VIN = 15 V, IF = 1 A) Diagram
8.3.2.1 Bill of Materials
Table 5. Bill of Materials for Figure 26
PART ID
PART VALUE
PART NUMBER
MANUFACTURER
U1
1-A LED Driver
LM3405
Texas Instruments
C1, Input capacitor
10 µF, 25 V, X5R
ECJ-3YB1E106K
Panasonic
C2, Output capacitor
1 µF, 10 V, X7R
GRM319R71A105KC01D
Murata
C3, Boost capacitor
0.01 µF, 16 V, X7R
0805YC103KAT2A
AVX
C4, Feedforward capacitor
1 µF, 10 V, X7R
GRM319R71A105KC01D
Murata
D1, Catch diode
Schottky, 0.5 V at 1A, VR = 30 V
SS13
Vishay
D2, Boost diode
Schottky, 0.36 V at 15 mA
CMDSH-3
Central Semiconductor
D3, Zener diode
11 V, 350 mW, SOT-23
BZX84C11
Fairchild
L1
6.8 µH, 1.5 A
SLF6028T-6R8M1R5
TDK
R1
0.2 Ω, 0.5 W, 1%
WSL2010R2000FEA
Vishay
8.3.3 VBOOST Derived From VIN Through a Shunt Zener Diode, D3 (VIN = 15 V, IF = 1 A)
C5
D3
R2
VIN
D2
VIN
BOOST
C3
C1
VOUT
L1
SW
LM3405
DC or
PWM
D1
EN/DIM
C2
C4
IF
LED1
FB
GND
R1
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Figure 27. VBOOST Derived From VIN Through a Shunt Zener Diode, D3
(VIN = 15 V, IF = 1 A) Diagram
22
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8.3.3.1 Bill of Materials
Table 6. Bill of Materials for Figure 27
PART ID
PART VALUE
PART NUMBER
MANUFACTURER
U1
1-A LED Driver
LM3405
Texas Instruments
C1, Input capacitor
10 µF, 25 V, X5R
ECJ-3YB1E106K
Panasonic
C2, Output capacitor
1 µF, 10 V, X7R
GRM319R71A105KC01D
Murata
C3, Boost capacitor
0.01 µF, 16 V, X7R
0805YC103KAT2A
AVX
C4, Feedforward capacitor 1 µF, 10 V, X7R
GRM319R71A105KC01D
Murata
C5, Shunt capacitor
0.1 µF, 16 V, X7R
GRM219R71C104KA01D
Murata
D1, Catch diode
Schottky, 0.5 V at 1 A, VR = 30 V
SS13
Vishay
D2, Boost diode
Schottky, 0.36 V at 15 mA
CMDSH-3
Central Semiconductor
D3, Zener diode
4.7 V, 350 mW, SOT-23
BZX84C4 V7
Fairchild
L1
6.8 µH, 1.5 A
SLF6028T-6R8M1R5
TDK
R1
0.2 Ω, 0.5 W, 1%
WSL2010R2000FEA
Vishay
R2
1.91 kΩ, 1%
CRCW08051K91FKEA
Vishay
8.3.4 VBOOST Derived from VOUT Through a Series Zener Diode, D3 (VIN = 15 V, IF = 1 A)
D3
D2
BOOST
VIN
VIN
C3
C1
VOUT
L1
SW
LM3405
DC or
PWM
D1
C2
EN/DIM
C4
IF
LED1
FB
GND
R1
Copyright © 2016, Texas Instruments Incorporated
Figure 28. VBOOST Derived from VOUT Through a Series Zener Diode, D3
(VIN = 15 V, IF = 1 A) Diagram
8.3.4.1 Bill of Materials
Table 7. Bill of Materials for Figure 28
PART ID
PART VALUE
PART NUMBER
MANUFACTURER
U1
1-A LED Driver
LM3405
Texas Instruments
C1, Input capacitor
10 µF, 25 V, X5R
ECJ-3YB1E106K
Panasonic
C2, Output capacitor
1 µF, 16 V, X7R
GRM319R71A105KC01D
Murata
C3, Boost capacitor
0.01 µF, 16 V, X7R
0805YC103KAT2A
AVX
C4, Feedforward capacitor
1 µF, 16 V, X7R
GRM319R71A105KC01D
Murata
D1, Catch diode
Schottky, 0.5 V at 1 A, VR = 30 V SS13
Vishay
D2, Boost diode
Schottky, 0.36 V at 15 mA
CMDSH-3
Central Semiconductor
D3, Zener diode
11 V, 350 mW, SOT-23
BZX84C11
Fairchild
L1
6.8 µH, 1.5 A
SLF6028T-6R8M1R5
TDK
R1
0.2 Ω, 0.5 W, 1%
WSL2010R2000FEA
Vishay
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9 Power Supply Recommendations
Any DC output power supply may be used provided it has a high enough voltage and current rating required for
the particular application.
10 Layout
10.1 Layout Guidelines
When planning the layout there are a few things to consider when trying to achieve a clean, regulated output.
The most important consideration when completing the layout is the close coupling of the GND connections of
the input capacitor C1 and the catch diode D1. These ground ends must be close to one another and be
connected to the GND plane with at least two vias. Place these components as close to the IC as possible. The
next consideration is the location of the GND connection of the output capacitor C2, which must be near the
GND connections of C1 and D1.
There must be a continuous ground plane on the bottom layer of a two-layer board.
The FB pin is a high impedance node and take care to make the FB trace short to avoid noise pickup that
causes inaccurate regulation. The LED current setting resistor R1 must be placed as close as possible to the IC,
with the GND of R1 placed as close as possible to the GND of the IC. The VOUT trace to LED anode must be
routed away from the inductor and any other traces that are switching.
High AC currents flow through the VIN, SW and VOUT traces, so they must be as short and wide as possible.
Radiated noise can be decreased by choosing a shielded inductor.
The remaining components must also be placed as close as possible to the IC. See AN-1229 SIMPLE
SWITCHER® PCB Layout Guidelines (SNVA054) for further considerations.
10.2 Layout Example
LED+
GND
LED-
1
BOOST SW
6
2
GND
5
3
FB EN/DIM 4
VIN
VIN
GND
VIA (GND VIAS TIED TO BOTTOM LAYER GROUND PLANE)
Schematic in Figure 23
Figure 29. LM3405 Layout Example
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11 Device and Documentation Support
11.1 Documentation Support
11.1.1 Related Documentation
For related documentation see the following:
• AN-1229 SIMPLE SWITCHER® PCB Layout Guidelines (SNVA054)
• AN-1644 Powering and Dimming High-Brightness LEDs with the LM3405 Constant-Current Buck Regulator
(SNVA247)
• AN-1656 Design Challenges of Switching LED Drivers (SNVA253)
• AN-1685 LM3405A Demo Board (SNVA271)
• AN-1899 LM3405A VSSOP Evaluation Board (SNVA370)
• AN-1982 Small, Wide Input Voltage Range LM2842 Keeps LEDs Cool (SNVA402)
• LM3405A Reference Design for MR16 LED Bulb, 600mA (SNVU101)
11.2 Receiving Notification of Documentation Updates
To receive notification of documentation updates, navigate to the device product folder on ti.com. In the upper
right corner, click on Alert me to register and receive a weekly digest of any product information that has
changed. For change details, review the revision history included in any revised document.
11.3 Community Resource
The following links connect to TI community resources. Linked contents are provided "AS IS" by the respective
contributors. They do not constitute TI specifications and do not necessarily reflect TI's views; see TI's Terms of
Use.
TI E2E™ Online Community TI's Engineer-to-Engineer (E2E) Community. Created to foster collaboration
among engineers. At e2e.ti.com, you can ask questions, share knowledge, explore ideas and help
solve problems with fellow engineers.
Design Support TI's Design Support Quickly find helpful E2E forums along with design support tools and
contact information for technical support.
11.4 Trademarks
E2E is a trademark of Texas Instruments.
SIMPLE SWITCHER is a registered trademark of Texas Instruments.
All other trademarks are the property of their respective owners.
11.5 Electrostatic Discharge Caution
These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
11.6 Glossary
SLYZ022 — TI Glossary.
This glossary lists and explains terms, acronyms, and definitions.
12 Mechanical, Packaging, and Orderable Information
The following pages include mechanical, packaging, and orderable information. This information is the most
current data available for the designated devices. This data is subject to change without notice and revision of
this document. For browser-based versions of this data sheet, refer to the left-hand navigation.
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Copyright © 2006–2016, Texas Instruments Incorporated
Product Folder Links: LM3405
25
PACKAGE OPTION ADDENDUM
www.ti.com
10-Dec-2020
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
(4/5)
(6)
LM3405XMK/NOPB
ACTIVE
SOT-23-THIN
DDC
6
1000
RoHS & Green
SN
Level-1-260C-UNLIM
-40 to 125
SPNB
LM3405XMKX/NOPB
ACTIVE
SOT-23-THIN
DDC
6
3000
RoHS & Green
SN
Level-1-260C-UNLIM
-40 to 125
SPNB
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of
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