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LM43603-Q1
SNVSA82C – APRIL 2015 – REVISED OCTOBER 2017
LM43603-Q1 3.5-V to 36-V, 3-A Synchronous Step-Down Voltage Converter
1 Features
3 Description
•
•
The LM43603-Q1 regulator is an easy-to-use
synchronous step-down DC-DC converter capable of
driving up to 3 A of load current from an input voltage
ranging from 3.5 V to 36 V (42 V absolute maximum).
The LM43603-Q1 provides exceptional efficiency,
output accuracy, and dropout voltage in a very small
solution size. An extended family is available in 0.5-A,
1-A, and 2-A load current options in pin-to-pin
compatible packages. Peak current mode control is
employed
to
achieve
simple
control
loop
compensation and cycle-by-cycle current limiting.
Optional features such as programmable switching
frequency,
synchronization,
power-good
flag,
precision enable, internal soft-start, extendable soft
start, and tracking provide a flexible and easy to use
platform for a wide range of applications.
Discontinuous conduction and automatic frequency
modulation at light loads improve light load efficiency.
The family requires few external components and pin
arrangement allows simple, optimum PCB layout.
Protection features include thermal shutdown, VCC
undervoltage lockout, cycle-by-cycle current limit, and
output short circuit protection. The LM43603-Q1
device is available in the HTSSOP (PWP) 16-pin
leaded package (6.6 mm × 5.1 mm × 1.2 mm). The
LM43603A-Q1 version is optimized for PFM operation
and recommended for new design. The device is pinto-pin compatible with LM4360x and LM4600x family.
1
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Qualified for Automotive Applications
AEC-Q100 Qualified With the Following Results:
– Device Temperature Grade 1: –40°C to
+125°C Operating Junction Temperature
27-µA Quiescent Current in Regulation
High Efficiency at Light Load (DCM and PFM)
Meets EN55022/CISPR 22 EMI Standards
Integrated Synchronous Rectification
Adjustable Frequency Range: 200 kHz to 2.2 MHz
(500 kHz default)
Frequency Synchronization to External Clock
Internal Compensation
Stable With Almost any Combination of Ceramic,
Polymer, Tantalum, and Aluminum Capacitors
Power-Good Flag
Soft Start into a Pre-Biased Load
Internal Soft Start: 4.1 ms
Extendable Soft-Start Time by External Capacitor
Output Voltage Tracking Capability
Precision Enable to a Program System UVLO
Output Short Circuit Protection with Hiccup Mode
Overtemperature Thermal Shutdown Protection
Create a Custom Design Using the LM43603-Q1
With the WEBENCH® Power Designer
Device Information
PART NUMBER
PACKAGE
BODY SIZE
2 Applications
LM43603-Q1
HTSSOP (16)
6.60 mm × 5.10 mm
•
•
•
•
•
LM43603A-Q1
HTTSOP (16)
6.60 mm × 5.10 mm
Sub-AM Band Automotive
Industrial Power Supplies
General Purpose Wide VIN Regulation
High Efficiency Point-Of-Load Regulation
Telecommunications Systems
Radiated Emission Graph
12 VIN to 3.3 VOUT, FS = 500 kHz, IOUT = 3 A
Simplified Schematic
VIN
VIN
CIN
VOUT
SW
LM43603Q
ENABLE
CBOOT
CBOOT
BIAS
PGOOD
CBIAS
SS/TRK
RT
SYNC
AGND
COUT
CFF
RFBT
FB
VCC
CVCC
RFBB
Radiated EMI Emissions (dBµV/m)
80
L
Evaluation Board
70
EN 55022 Class B Limit
EN 55022 Class A Limit
60
50
40
30
20
10
0
PGND
0
C001
200
400
600
Frequency (MHz)
800
1000
C001
1
An IMPORTANT NOTICE at the end of this data sheet addresses availability, warranty, changes, use in safety-critical applications,
intellectual property matters and other important disclaimers. PRODUCTION DATA.
LM43603-Q1
SNVSA82C – APRIL 2015 – REVISED OCTOBER 2017
www.ti.com
Table of Contents
1
2
3
4
5
6
7
Features ..................................................................
Applications ...........................................................
Description .............................................................
Revision History.....................................................
Pin Configuration and Functions .........................
Specifications.........................................................
1
1
1
2
4
5
6.1
6.2
6.3
6.4
6.5
6.6
6.7
6.8
5
5
5
6
6
7
8
9
Absolute Maximum Ratings ......................................
ESD Ratings..............................................................
Recommended Operating Conditions.......................
Thermal Information ..................................................
Electrical Characteristics...........................................
Timing Requirements ................................................
Switching Characteristics ..........................................
Typical Characteristics ..............................................
Detailed Description ............................................ 15
7.1 Overview ................................................................. 15
7.2 Functional Block Diagram ....................................... 15
7.3 Feature Description................................................. 16
7.4 Device Functional Modes........................................ 23
8
Applications and Implementation ...................... 25
8.1 Application Information............................................ 25
8.2 Typical Applications ................................................ 25
9 Power Supply Recommendations...................... 40
10 Layout................................................................... 40
10.1 Layout Guidelines ................................................. 40
10.2 Layout Example .................................................... 43
11 Device and Documentation Support ................. 44
11.1
11.2
11.3
11.4
11.5
11.6
Device Support......................................................
Development Support ...........................................
Receiving Notification of Documentation Updates
Community Resources..........................................
Electrostatic Discharge Caution ............................
Glossary ................................................................
44
44
44
44
44
44
12 Mechanical, Packaging, and Orderable
Information ........................................................... 45
4 Revision History
Changes from Revision B (April 2017) to Revision C
Page
•
Added top navigator icon for TI reference design ................................................................................................................. 1
•
Changed MAX value for VBIAS rising threshold from "3.15" to "3.18" V .............................................................................. 6
Changes from Revision A (May 2015) to Revision B
Page
•
Added links for Webench ....................................................................................................................................................... 1
•
Added LM43603A Device Information.................................................................................................................................... 1
•
Added Maximum Operating Junction Temperature ............................................................................................................... 5
•
Added standard FN1 to Thermal Information ........................................................................................................................ 6
•
Updating the RPGOOD value on EN = 3.3V and EN = 0V ................................................................................................... 7
•
Updating Figure 11 to match Figure 87 ............................................................................................................................... 10
•
Updating EN Falling Threshold Curve .................................................................................................................................. 12
•
Updating EN Rising Threshold Curve ................................................................................................................................. 12
•
Updating EN Hysteresis Curve ............................................................................................................................................ 12
•
Changed Figure 33 into conducted EMI Curve .................................................................................................................... 14
•
Replaced last few sentences of Application Information due to new Webench content ...................................................... 25
•
Added Equation 25 ............................................................................................................................................................... 31
•
Added Equation 26 ............................................................................................................................................................... 31
•
Added Figure 73 to Figure 78. Application Performance Curves for VOUT = 5 V, Fs = 500 kHz. ........................................ 36
•
Changed Figure 86............................................................................................................................................................... 38
•
Changed Figure 87 .............................................................................................................................................................. 38
2
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SNVSA82C – APRIL 2015 – REVISED OCTOBER 2017
Changes from Original (April 2015) to Revision A
Page
•
Changed device from Product Preview to Production Data .................................................................................................. 1
•
Deleted "Super-AM Band" ...................................................................................................................................................... 1
•
Added "BIAS pin voltage should never exceed VIN" to BIAS pin description ......................................................................... 4
•
Added "of 10000 hours" to Ab Max FN 2 ............................................................................................................................... 5
•
Changed info in Vfb rows; in ILKG-FB changed value of from FB=1.011 V to 1.015 V ............................................................. 7
•
Changed VFB = 1.011 V to VFB = 1.015 V............................................................................................................................. 27
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SNVSA82C – APRIL 2015 – REVISED OCTOBER 2017
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5 Pin Configuration and Functions
PWP Package
16-Pin HTSSOP
Top View
SW
1
SW
2
CBOOT
VCC
3
4
BIAS
SYNC
RT
PGOOD
16
15
PGND
PGND
14
VIN
13
VIN
12
EN
6
11
SS/TRK
7
8
10
AGND
5
PAD
9
FB
Pin Functions
PIN
DESCRIPTION
NAME
NO.
TYPE (1)
SW
1, 2
P
Switching output of the regulator. Internally connected to both power MOSFETs. Connect to power
inductor.
CBOOT
3
P
Boot-strap capacitor connection for high-side driver. Connect a high quality 470-nF capacitor from
CBOOT to SW.
VCC
4
P
Internal bias supply output for bypassing. Connect bypass capacitor from this pin to AGND. Do not
connect external loading to this pin. Never short this pin to ground during operation.
BIAS
5
P
Optional internal LDO supply input. To improve efficiency, TI recommends tying to VOUT when 3.3 V ≤
VOUT ≤ 28 V, or tie to an external 3.3 V or 5 V rail if available. When used, place a bypass capacitor (1
to 10 µF) from this pin to ground. Tie to ground when not in use. Do not float. BIAS pin voltage should
never exceed VIN.
SYNC
6
A
Clock input to synchronize switching action to an external clock. Use proper high-speed termination to
prevent ringing. Connect to ground if not used. Do not float.
RT
7
A
Connect a resistor RT from this pin to AGND to program switching frequency. Leave floating for 500
kHz default switching frequency.
PGOOD
8
A
Open drain output for power-good flag. Use a 10-kΩ to 100-kΩ pullup resistor to logic rail or other DC
voltage no higher than 12 V.
FB
9
A
Feedback sense input pin. Connect to the midpoint of feedback divider to set VOUT. Do not short this
pin to ground during operation.
AGND
10
G
Analog ground pin. Ground reference for internal references and logic. Connect to system ground.
SS/TRK
11
A
Soft-start control pin. Leave floating for internal soft-start slew rate. Connect to a capacitor to extend
soft start time. Connect to external voltage ramp for tracking.
EN
12
A
Enable input to the internal LDO and regulator. High = ON and low = OFF. Connect to VIN, or to VIN
through resistor divider,or to an external voltage or logic source. Do not float.
VIN
13,14
P
Supply input pins to internal LDO and high side power FET. Connect to power supply and bypass
capacitors CIN. Path from VIN pin to high frequency bypass CIN and PGND must be as short as
possible.
PGND
15,16
G
Power ground pins, connected internally to the low side power FET. Connect to system ground, PAD,
AGND, ground pins of CIN and COUT. Path to CIN must be as short as possible.
-
-
Low impedance connection to AGND. Connect to PGND on PCB. Major heat dissipation path of the
die. Must be used for heat sinking to ground plane on PCB.
PAD
(1)
4
P = Power, G = Ground, A = Analog
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6 Specifications
6.1 Absolute Maximum Ratings
over the recommended operating junction temperature (TJ) range of -40°C to +125°C (1)
PARAMETER
MIN
MAX
VIN to PGND
–0.3
42 (2)
EN to PGND
–0.3
VIN + 0.3
FB, RT, SS/TRK to AGND
–0.3
3.6
PGOOD to AGND
–0.3
15
SYNC to AGND
–0.3
5.5
BIAS to AGND
–0.3
30 or VIN (3)
AGND to PGND
–0.3
0.3
SW to PGND
–0.3
VIN + 0.3
SW to PGND less than 10-ns transients
–3.5
42
CBOOT to SW
–0.3
5.5
VCC to AGND
–0.3
3.6
Storage temperature, Tstg
–65
150
°C
Operating junction temperature
–40
150
°C
Input voltages
Output voltages
(1)
(2)
(3)
UNIT
V
V
Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. These are stress ratings
only, and functional operation of the device at these or any other conditions beyond those indicated under Recommended Operating
Conditions is not implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
At maximum duty cycle 0.01%
Whichever is lower
6.2 ESD Ratings
VALUE
V(ESD)
(1)
Electrostatic discharge
Human-body model (HBM), per AEC Q100-002
(1)
UNIT
±2000
Charged-device model (CDM), per AEC Q100-011
V
±750
AEC Q100-002 indicates that HBM stressing shall be in accordance with the ANSI/ESDA/JEDEC JS-001 specification.
6.3 Recommended Operating Conditions
over the recommended operating junction temperature (TJ) range of –40°C to +125°C
Input voltages
(1)
PARAMETER
MIN
MAX
VIN to PGND
3.5
36
EN
–0.3
VIN
FB
–0.3
1.1
PGOOD
–0.3
12
BIAS input not used
–0.3
0.3
BIAS input used
3.3
AGND to PGND
28 or VIN
UNIT
V
(2)
–0.1
0.1
Output voltage
VOUT
1
28
V
Output current
IOUT
0
3
A
Temperature
Operating junction temperature, TJ
–40
125
°C
(1)
(2)
Recommended Operating Conditions indicates conditions for which the device is intended to be functional, but do not ensure specific
performance limits. For verified specifications, see Electrical Characteristics.
Whichever is lower.
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6.4 Thermal Information
LM43603-Q1
THERMAL METRIC (1) (2) (3)
PWP (HTSSOP)
UNIT
16 PINS
38.9 (4)
°C/W
Junction-to-case (top) thermal resistance
24.3
°C/W
Junction-to-board thermal resistance
19.9
°C/W
ψJT
Junction-to-top characterization parameter
0.7
°C/W
ψJB
Junction-to-board characterization parameter
19.7
°C/W
RθJC(bot)
Junction-to-case (bottom) thermal resistance
1.7
°C/W
RθJA
Junction-to-ambient thermal resistance
RθJC (Top)
RθJB
(1)
(2)
(3)
(4)
For more information about traditional and new thermal metrics, see the Semiconductor and IC Package Thermal Metrics application
report.
The package thermal impedance is calculated in accordance with JESD 51-7;
Thermal resistances were simulated on a 4 layer, JEDEC board.
See Figure 98 for RθJA vs Copper Area Curve
6.5 Electrical Characteristics
Limits apply over the recommended operating junction temperature (TJ) range of –40°C to +125°C, unless otherwise stated.
Minimum and Maximum limits are specified through test, design or statistical correlation. Typical values represent the most
likely parametric norm at TJ = 25°C, and are provided for reference purposes only. Unless otherwise stated, the following
conditions apply: VIN = 12 V, VOUT = 3.3 V, FS = 500 kHz.
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNIT
SUPPLY VOLTAGE (VIN PIN)
VIN-MIN-ST
Minimum input voltage for start-up
ISHDN
Shutdown quiescent current
VEN = 0 V
3.8
V
1.2
3.1
µA
IQ-NONSW
Operating quiescent current (nonswitching) from VIN
VEN = 3.3 V
VFB = 1.5 V
VBIAS = 3.4 V external
5
10
µA
IBIAS-NONSW
Operating quiescent current (nonswitching) from external VBIAS
VEN = 3.3 V
VFB = 1.5 V
VBIAS = 3.4 V external
85
130
µA
Operating quiescent current
(switching)
VEN = 3.3 V
IOUT = 0 A
RT = open
VBIAS = VOUT = 3.3 V
RFBT = 1 Meg
IQ-SW
27
VEN-VCC-H
Voltage level to enable the internal
LDO output VCC
VENABLE high level
VEN-VCC-L
Voltage level to disable the internal
LDO output VCC
VENABLE low level
VEN-VOUT-H
Precision enable level for switching
and regulator output: VOUT
VENABLE high level
VEN-VOUT-HYS
Hysteresis voltage between VOUT
precision enable and disable
thresholds
VENABLE hysteresis
–290
ILKG-EN
Enable input leakage current
VEN = 3.3 V
0.85
VIN ≥ 3.8 V
3.28
V
3.1
V
µA
ENABLE (EN PIN)
1.2
2
V
2.20
0.525
V
2.42
V
mV
1.75
µA
INTERNAL LDO (VCC and BIAS PINS)
VCC
Internal LDO output voltage VCC
VCC-UVLO
Undervoltage lockout (UVLO)
thresholds for VCC
VBIAS-ON
Internal LDO input change over
threshold to BIAS
6
VCC rising threshold
Hysteresis voltage between rising
and falling thresholds
–520
VBIAS rising threshold
2.94
Hysteresis voltage between rising
and falling thresholds
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-75
mV
3.18
V
mV
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Electrical Characteristics (continued)
Limits apply over the recommended operating junction temperature (TJ) range of –40°C to +125°C, unless otherwise stated.
Minimum and Maximum limits are specified through test, design or statistical correlation. Typical values represent the most
likely parametric norm at TJ = 25°C, and are provided for reference purposes only. Unless otherwise stated, the following
conditions apply: VIN = 12 V, VOUT = 3.3 V, FS = 500 kHz.
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNIT
TJ = 25 ºC
1.012
1.015
1.019
TJ = -40 ºC to 125 ºC
0.999
1.015
1.032
FB = 1.015 V
0.2
65
Shutdown threshold
160
ºC
Recovery threshold
150
ºC
VOLTAGE REFERENCE (FB PIN)
VFB
Feedback voltage
ILKG-FB
Input leakage current at FB pin
V
nA
THERMAL SHUTDOWN
TSD
(1)
Thermal shutdown
CURRENT LIMIT AND HICCUP
IHS-LIMIT
Peak inductor current limit
4.4
5.5
6.4
A
ILS-LIMIT
Inductor current valley limit
2.6
3
3.3
A
2
2.75
µA
SOFT START (SS/TRK PIN)
ISSC
Soft-start charge current
RSSD
Soft-start discharge resistance
1.25
UVLO, TSD, OCP, or EN = 0 V
18
kΩ
POWER GOOD (PGOOD PIN)
VPGOOD-HIGH
Power-good flag over voltage tripping
% of FB voltage
threshold
VPGOOD-LOW
Power-good flag under voltage
tripping threshold
% of FB voltage
VPGOOD-HYS
Power-good flag recovery hysteresis
% of FB voltage
PGOOD pin pulldown resistance
when power bad
VEN = 3.3 V
RPGOOD
MOSFETS
110%
77%
113%
88%
6%
69
150
VEN = 0 V
150
350
Ω
(2)
RDS-ON-HS
High-side MOSFET ON-resistance
IOUT = 1 A
VBIAS = VOUT = 3.3 V
120
mΩ
RDS-ON-LS
Low-side MOSFET ON-resistance
IOUT = 1 A
VBIAS = VOUT = 3.3 V
65
mΩ
(1)
(2)
Ensured by design
Measured at pins
6.6 Timing Requirements
MIN
NOM
MAX
UNIT
CURRENT LIMIT AND HICCUP
NOC
Hiccup wait cycles when LS current limit tripped
32
Cycles
TOC
Hiccup retry delay time
5.5
ms
4.1
ms
TPGOOD-RISE Power-good flag rising transition deglitch delay
220
µs
TPGOOD-FALL Power-good flag falling transition deglitch delay
220
µs
SOFT START (SS/TRK PIN)
TSS
Internal soft-start time when SS pin open circuit
POWER GOOD (PGOOD PIN)
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6.7 Switching Characteristics
Limits apply over the recommended operating junction temperature (TJ) range of –40°C to +125°C, unless otherwise stated.
Minimum and Maximum limits are specified through test, design or statistical correlation. Typical values represent the most
likely parametric norm at TJ = 25°C, and are provided for reference purposes only. Unless otherwise stated, the following
conditions apply: VIN = 12 V, VOUT = 3.3 V, FS = 500 kHz.
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
SW (SW PIN)
tON-MIN (1)
Minimum high side MOSFET ON
time
125
165
ns
tOFF-MIN (1)
Minimum high side MOSFET OFF
time
200
250
ns
500
580
kHz
OSCILLATOR (SW and SYNC PINS)
FOSC-DEFAULT
Oscillator default frequency
RT pin open circuit
425
Minimum adjustable frequency
FADJ
Maximum adjustable frequency
With 1% resistors at RT pin
Frequency adjust accuracy
200
kHz
2200
kHz
10%
VSYNC-HIGH
Sync clock high level threshold
VSYNC-LOW
Sync clock low level threshold
DSYNC-MAX
Sync clock maximum duty cycle
90%
DSYNC-MIN
Sync clock minimum duty cycle
10%
TSYNC-MIN
Mininum sync clock ON and OFF
time
(1)
8
2
V
0.4
80
V
ns
Ensured by design
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6.8 Typical Characteristics
100
100
90
90
80
80
Efficiency (%)
Efficiency (%)
Unless otherwise specified, VIN = 12 V, VOUT = 3.3 V, FS = 500 kHz, L = 6.8 µH and room temperature. See Application
Performance Curves for bill of materials for other VOUT and FS combinations.
70
60
70
60
5VIN
50
12VIN
50
12VIN
24VIN
24VIN
40
0.001
0.01
0.1
40
0.001
1
Current (A)
VOUT = 3.3 V
0.1
1
Current (A)
FS = 500 kHz
VOUT = 5V
Figure 1. Efficiency
C001
FS = 200 kHz
Figure 2. Efficiency
100
100
90
90
80
80
Efficiency (%)
Efficiency (%)
0.01
C001
70
60
70
60
12VIN
12VIN
50
50
24VIN
40
0.001
0.01
0.1
1
Current (A)
VOUT = 5 V
24VIN
40
0.001
FS = 500 kHz
0.1
1
Current (A)
VOUT = 5 V
Figure 3. Efficiency
C001
FS = 1 MHz
Figure 4. Efficiency
100
100
90
90
80
80
Efficiency (%)
Efficiency (%)
0.01
C001
70
60
70
60
12VIN
50
50
24VIN
16VIN
40
0.001
0.01
0.1
Current (A)
VOUT = 5 V
FS = 2.2 MHz
36VIN
40
0.001
1
0.01
0.1
1
Current (A)
C001
VOUT = 12 V
Figure 5. Efficiency
C049
FS = 500 kHz
Figure 6. Efficiency
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Typical Characteristics (continued)
Unless otherwise specified, VIN = 12 V, VOUT = 3.3 V, FS = 500 kHz, L = 6.8 µH and room temperature. See Application
Performance Curves for bill of materials for other VOUT and FS combinations.
3.40
5.25
5VIN
3.38
3.36
VOUT (V)
VOUT (V)
3.30
3.28
5.05
5.00
4.95
3.26
4.90
3.24
4.85
3.22
4.80
0.01
0.1
4.75
0.001
1
Current (A)
Figure 7. VOUT Regulation
VOUT (V)
VOUT (V)
4.95
5.05
5.00
4.95
4.90
4.90
4.85
4.85
4.80
4.80
0.01
0.1
4.75
0.001
1
0.01
0.1
1
Current (A)
C004
FS = 500 kHz
VOUT = 5 V
Figure 9. VOUT Regulation
C005
FS = 1 MHz
Figure 10. VOUT Regulation
5.25
12.5
5.20
12.4
5.15
12.3
5.10
12.2
5.05
12.1
VOUT (V)
VOUT (V)
24VIN
5.10
Current (A)
5.00
4.95
4.90
24VIN
36VIN
12.0
11.9
11.8
4.85
12VIN
11.7
4.80
16VIN
11.6
0.01
0.1
Current (A)
FS = 2.2 MHz
11.5
0.001
1
0.01
0.1
Current (A)
C001
VOUT = 12 V
Figure 11. VOUT Regulation
10
12VIN
5.15
24VIN
5.00
VOUT = 5 V
8VIN
5.20
12VIN
5.05
4.75
0.001
C003
Figure 8. VOUT Regulation
5.10
VOUT = 5 V
1
FS = 200 kHz
5.25
8VIN
5.15
0.1
Current (A)
VOUT = 5 V
5.20
0.01
C001
FS = 500 kHz
5.25
4.75
0.001
24VIN
5.10
3.32
VOUT = 3.3V
12VIN
5.15
24VIN
3.34
3.20
0.001
8VIN
5.20
12VIN
1
C050
FS = 500 kHz
Figure 12. VOUT Regulation
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Typical Characteristics (continued)
Unless otherwise specified, VIN = 12 V, VOUT = 3.3 V, FS = 500 kHz, L = 6.8 µH and room temperature. See Application
Performance Curves for bill of materials for other VOUT and FS combinations.
3.5
5.4
3.4
5.2
5.0
VOUT (V)
VOUT (V)
3.3
3.2
0.1A
0.5A
1A
1.5A
2A
2.5A
3.1
3.0
3.7
3.9
4.1
4.3
4.0
5.00
4.5
VOUT = 5 V
5.2
5.20
5.0
5.00
4.8
4.6
0.1A
0.5A
1A
1.5A
2A
2.5A
4.4
4.2
VIN (V)
VIN (V)
12.0
5.00
11.8
VOUT (V)
VOUT (V)
5.20
4.80
4.60
0.01A
0.1A
0.5A
1A
1.5A
2A
2.5A
VIN (V)
FS = 2.2 MHz
C007
FS = 1 MHz
Figure 16. Dropout Curve
12.2
VOUT = 5 V
0.1A
0.5A
1A
1.5A
2A
2.5A
4.00
5.00 5.20 5.40 5.60 5.80 6.00 6.20 6.40 6.60 6.80 7.00
Figure 15. Dropout Curve
6.00
C007
4.60
VOUT = 5 V
5.80
6.40
4.80
5.40
5.60
6.20
FS = 200 kHz
C007
4.20
6.00
4.20
FS = 500 kHz
4.40
5.80
4.40
4.0
5.00 5.20 5.40 5.60 5.80 6.00 6.20 6.40 6.60 6.80 7.00
5.40
5.60
Figure 14. Dropout Curve
5.40
VOUT (V)
VOUT (V)
Figure 13. Dropout Curve
5.20
5.40
VIN (V)
5.4
4.00
5.00
5.20
C007
FS = 500 kHz
VOUT = 5 V
0.1A
0.5A
1A
1.5A
2A
2.5A
4.2
VIN (V)
VOUT = 3.3 V
4.6
4.4
2.9
3.5
4.8
6.20
11.6
11.4
0.1A
0.5A
1A
1.5A
2A
2.5A
3A
11.2
11.0
6.40
10.8
12.0
12.5
13.0
13.5
14.0
VIN (V)
C007
VOUT = 12 V
Figure 17. Dropout Curve
14.5
C007
FS = 500 kHz
Figure 18. Dropout Curve
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Typical Characteristics (continued)
Unless otherwise specified, VIN = 12 V, VOUT = 3.3 V, FS = 500 kHz, L = 6.8 µH and room temperature. See Application
Performance Curves for bill of materials for other VOUT and FS combinations.
2.200
2.000
1.950
2.150
EN Voltage (V)
EN Voltage (V)
1.900
1.850
1.800
1.750
2.100
2.050
2.000
1.700
1.950
1.650
1.900
1.600
±40 ±25 ±10
5
20
35
50
65
80
95
Temperature (deg C)
110 125
±40 ±25 ±10
5
20
35
50
65
80
95
110 125
Temperature (deg C)
C001
Figure 19. EN Falling Threshold
C001
Figure 20. EN Rising Threshold
1.020
310
300
EN Hysteresis (mV)
1.015
FB Voltage (V)
290
280
270
1.010
1.005
260
1.000
250
±40 ±25 ±10
5
20
35
50
65
80
95
Temperature (deg C)
110 125
±40 ±25 ±10
35
50
65
80
95
110 125
C013
Figure 22. FB Voltage vs Junction Temperature
190
90
170
80
LS RDS-ON (m )
HS RDS-ON (m )
20
Temperature (ƒC)
Figure 21. EN Hysteresis
150
130
110
70
60
50
90
40
±40 ±25 ±10
5
20
35
50
65
Temperature (ƒC)
80
95
110 125
±40 ±25 ±10
5
20
35
50
65
Temperature (ƒC)
C013
Figure 23. High-Side FET On Resistance vs Junction
Temperature
12
5
C001
80
95
110 125
C013
Figure 24. Low-Side FET On Resistance vs Junction
Temperature
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Typical Characteristics (continued)
6.0
3.3
5.8
3.2
LS Current Limit (A)
HS Current Limit (A)
Unless otherwise specified, VIN = 12 V, VOUT = 3.3 V, FS = 500 kHz, L = 6.8 µH and room temperature. See Application
Performance Curves for bill of materials for other VOUT and FS combinations.
5.6
5.4
3.1
3.0
5.2
2.9
5.0
2.8
±40 ±25 ±10
5
20
35
50
65
80
95
Temperature (ƒC)
110 125
±40 ±25 ±10
Figure 25. High-Side Current Limit vs Junction Temperature
35
50
65
80
95
110 125
C013
Figure 26. Low-Side Current Limit vs Junction Temperature
116
Percentage of FB Voltage (%)
Percentage of FB Voltage (%)
20
Temperature (ƒC)
110
108
106
104
102
114
112
110
108
±40 ±25 ±10
5
20
35
50
65
80
95
Temperature (ƒC)
110 125
±40 ±25 ±10
5
20
35
50
65
80
95
110 125
Temperature (ƒC)
C013
Figure 27. PGOOD OVP Falling Threshold vs Junction
Temperature
C013
Figure 28. PGOOD OVP Rising Threshold vs Junction
Temperature
98
Percentage of FB Voltage (%)
92
Percentage of FB Voltage (%)
5
C013
91
90
89
88
87
86
97
96
95
94
93
92
±40 ±25 ±10
5
20
35
50
65
Temperature (ƒC)
80
95
110 125
±40 ±25 ±10
Figure 29. PGOOD UVP Falling Threshold vs Junction
Temperature
5
20
35
50
65
80
95
110 125
Temperature (ƒC)
C013
C013
Figure 30. PGOOD UVP Rising Threshold vs Junction
Temperature
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Typical Characteristics (continued)
Unless otherwise specified, VIN = 12 V, VOUT = 3.3 V, FS = 500 kHz, L = 6.8 µH and room temperature. See Application
Performance Curves for bill of materials for other VOUT and FS combinations.
80
Evaluation Board
70
Radiated EMI Emissions (dBµV/m)
Radiated EMI Emissions (dBµV/m)
80
EN 55022 Class B Limit
EN 55022 Class A Limit
60
50
40
30
20
10
0
Evaluation Board
70
EN 55022 Class B Limit
EN 55022 Class A Limit
60
50
40
30
20
10
0
0
200
400
600
800
1000
Frequency (MHz)
VOUT = 3.3 V
0
FS = 500 kHz
IOUT = 3 A
VOUT = 5 V
Average Limit
Conducted EMI (dBµV)
Conducted EMI (dBµV)
1000
C001
FS = 500 kHz
IOUT = 3 A
Peak Emissions
90
70
60
50
40
30
20
10
Quasi Peak Limit
80
Average Limit
70
60
50
40
30
20
10
0
0
0.1
1
10
100
Frequency (MHz)
VOUT = 3.3 V
Cd = 47 µF
FS = 500 kHz
Lin = 1 µH
0.1
IOUT = 3 A
CIN4 = 68 µF
1
10
100
Frequency (MHz)
C001
VOUT = 5 V
Cd = 47 µF
Figure 33. Conducted EMI Curve
14
800
Figure 32. Radiated EMI Curve
Quasi Peak Limit
80
600
100
Peak Emissions
90
400
Frequency (MHz)
Figure 31. Radiated EMI Curve
100
200
C001
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Lin = 1 µH
C001
IOUT = 3 A
CIN4 = 68 µF
Figure 34. Conducted EMI Curve
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7 Detailed Description
7.1 Overview
The LM43603-Q1 regulator is an easy-to-use synchronous step-down DC-DC converter that operates from 3.5 V
to 36 V supply voltage. It is capable of delivering up to 3-A DC load current with exceptional efficiency and
thermal performance in a very small solution size. An extended family is available in 0.5-A, 1-A, and 2-A load
options in pin-to-pin compatible packages.
The LM43603-Q1 employs fixed frequency peak current mode control with discontinuous conduction mode
(DCM) and pulse frequency modulation (PFM) mode at light load to achieve high efficiency across the load
range. The device is internally compensated, which reduces design time, and requires fewer external
components. The switching frequency is programmable from 200 kHz to 2.2 MHz by an external resistor RT. It is
default at 500 kHz without RT resistor. The LM43603-Q1 is also capable of synchronization to an external clock
within the 200 kHz to 2.2 MHz frequency range. The wide switching frequency range allows the device to be
optimized to fit small board space at higher frequency, or high efficient power conversion at lower frequency.
Optional features are included for more comprehensive system requirements, including power-good (PGOOD)
flag, precision enable, synchronization to external clock, extendable soft-start time, and output voltage tracking.
These features provide a flexible and easy to use platform for a wide range of applications. Protection features
include over temperature shutdown, VCC undervoltage lockout (UVLO), cycle-by-cycle current limit, and shortcircuit protection with hiccup mode.
The LM4360x family requires few external components, and the pin arrangement was designed for simple,
optimum PCB layout. The LM43603-Q1 device is available in the HTSSOP (PWP) 16-pin leaded package.
7.2 Functional Block Diagram
ENABLE
VCC
Enable
Internal
SS
ISSC
BIAS
VCC
LDO
VIN
Precision
Enable
SS/TRK
CBOOT
HS I Sense
+
EA
REF
RC
+
±
+±
TSD
UVLO
CC
PGOOD
AGND
OV/UV
Detector
FB
SW
PWM CONTROL LOGIC
PFM
Detector
PGood
Slope
Comp
Freq
Foldback
Zero
Cross
HICCUP
Detector
Oscillator
LS I Sense
FB
PGood
SYNC
PGND
RT
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7.3 Feature Description
7.3.1 Fixed Frequency Peak Current Mode Controlled Step-Down Regulator
The following operating description of the LM43603-Q1 refers to the Functional Block Diagram and to the
waveforms in Figure 35. The LM43603-Q1 is a step-down buck regulator with both a high-side (HS) and low-side
(LS) switch integrated into the device. The LM43603-Q1 supplies a regulated output voltage by turning on the HS
and LS NMOS switches with controlled ON time. During the HS switch ON time, the SW pin voltage VSW swings
up to approximately VIN, and the inductor current iL increases with a linear slope (VIN – VOUT) / L. When the HS
switch is turned off by the control logic, the LS switch is turned on after a anti-shoot-through dead time. Inductor
current discharges through the LS switch with a slope of –VOUT / L. The control parameter of buck converters are
defined as duty cycle D = tON / TSW, where tON is the HS switch ON time and TSW is the switching period. The
regulator control loop maintains a constant output voltage by adjusting the duty cycle D. In an ideal buck
converter, where losses are ignored, D is proportional to the output voltage and inversely proportional to the input
voltage: D = VOUT / VIN.
VSW
D = tON / TSW
SW Voltage
VIN
tOFF
tON
0
t
-VD1
Inductor Current
iL
TSW
ILPK
IOUT
ûiL
0
t
Figure 35. SW Node and Inductor Current Waveforms in Continuous Conduction Mode (CCM)
The LM43603-Q1 synchronous buck converter employs peak current mode control topology. A voltage feedback
loop is used to get accurate DC voltage regulation by adjusting the peak current command based on voltage
offset. The peak inductor current is sensed from the HS switch and compared to the peak current to control the
ON time of the HS switch. The voltage feedback loop is internally compensated, which allows for fewer external
components, makes it easy to design, and provides stable operation with almost any combination of output
capacitors. The regulator operates with fixed switching frequency in CCM and DCM. At very light load, the
LM43603-Q1 operates in PFM to maintain high efficiency and the switching frequency decreases with reduced
load current.
7.3.2 Light Load Operation
DCM operation is employed in the LM43603-Q1 when the inductor current valley reaches zero. The LM43603-Q1
is in DCM when load current is less than half of the peak-to-peak inductor current ripple in CCM. In DCM, the LS
switch is turned off when the inductor current reaches zero. Switching loss is reduced by turning off the LS FET
at zero current, and the conduction loss is lowered by not allowing negative current conduction. Power
conversion efficiency is higher in DCM than CCM under the same conditions.
In DCM, the HS switch ON time reduces with lower load current. When either the minimum HS switch ON time
(tON-MIN) or the minimum peak inductor current (IPEAK-MIN) is reached, the switching frequency decreases to
maintain regulation. At this point, the LM43603-Q1 operates in PFM. In PFM, switching frequency is decreased
by the control loop when load current reduces to maintain output voltage regulation. Switching loss is further
reduced in PFM operation due to less frequent switching actions.
16
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Feature Description (continued)
In PFM operation, a small positive DC offset is required at the output voltage to activate the PFM detector. The
lower the frequency in PFM, the more DC offset is needed at VOUT. Refer to the Typical Characteristics for typical
DC offset at very light load. If the DC offset on VOUT is not acceptable for a given application, a static load at
output is recommended to reduce or eliminate the offset. Lowering values of the feedback divider RFBT and RFBB
can also serve as a static load. In conditions with low VIN and/or high frequency, the LM43603-Q1 may not enter
PFM mode if the output voltage cannot be charged up to provide the trigger to activate the PFM detector. Once
the LM43603-Q1 is operating in PFM mode at higher VIN, it remains in PFM operation when VIN is reduced. See
Figure 45 for a sample of PFM operation.
7.3.3 Adjustable Output Voltage
The voltage regulation loop in the LM43603-Q1 regulates output voltage by maintaining the voltage on FB pin
(VFB) to be the same as the internal REF voltage (VREF). A resistor divider pair is needed to program the ratio
from output voltage VOUT to VFB. The resistor divider is connected from the VOUT of the LM43603-Q1 to ground
with the mid-point connecting to the FB pin.
VOUT
RFBT
FB
RFBB
Figure 36. Output Voltage Setting
The voltage reference system produces a precise voltage reference over temperature. The internal REF voltage
is 1.011 V typically. To program the output voltage of the LM43603-Q1 to be a certain value VOUT, RFBB can be
calculated with a selected RFBT by Equation 1:
VFB
RFBB
RFBT
VOUT VFB
(1)
The choice of the RFBT depends on the application. TI recommends RFBT in the range from 10 kΩ to 100 kΩ for
most applications. A lower RFBT value can be used if static loading is desired to reduce VOUT offset in PFM
operation. Lower RFBT reduces efficiency at very light load. Less static current goes through a larger RFBT and
might be more desirable when light load efficiency is critical. But RFBT larger than 1 MΩ is not recommended
because it makes the feedback path more susceptible to noise. Larger RFBT value requires more carefully
designed feedback path on the PCB. The tolerance and temperature variation of the resistor dividers affect the
output voltage regulation. TI recommends using divider resistors with 1% tolerance or better and temperature
coefficient of 100 ppm or lower.
If the resistor divider is not connected properly, output voltage cannot be regulated because the feedback loop is
broken. If the FB pin is shorted to ground, the output voltage is driven close to VIN, because the regulator sees
very low voltage on the FB pin and tries to regulator it up. The load connected to the output could be damaged
under such a condition. Do not short FB pin to ground when the LM43603-Q1 is enabled. It is important to route
the feedback trace away from the noisy area of the PCB. For more layout recommendations, see the Layout
section.
7.3.4 Enable (EN)
Voltage on the EN pin (VEN) controls the ON or OFF operation of the LM43603-Q1. Applying a voltage less than
0.4 V to the EN input shuts down the operation of the LM43603-Q1. In shutdown mode the quiescent current
drops to typically 1.2 µA at VIN = 12 V.
The internal LDO output voltage VCC is turned on when VEN is higher than 1.2 V. Switching action and output
regulation are enabled when VEN is greater than 2.1 V (typical). The LM43603-Q1 supplies regulated output
voltage when enabled and output current up to 3 A.
The EN pin is an input and cannot be open circuit or floating. The simplest way to enable the operation of the
LM43603-Q1 is to connect the EN pin to VIN pins directly. This allows self-start-up when VIN is within the
operation range.
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Feature Description (continued)
Many applications benefit from use of an enable divider RENT and RENB in Figure 37 to establish a precision
system UVLO level for the stage. System UVLO can be used for supplies operating from utility power as well as
battery power. It can be used for sequencing, ensuring reliable operation, or supply protection, such as a battery
discharge level. An external logic signal can also be used to drive EN input for system sequencing and
protection.
VIN
RENT
ENABLE
RENB
Figure 37. System UVLO by Enable Dividers
7.3.5 VCC, UVLO, and BIAS
The LM43603-Q1 integrates an internal LDO to generate VCC for control circuitry and MOSFET drivers. The
nominal voltage for VCC is 3.28 V. The VCC pin is the output of the LDO must be properly bypassed. Place a
high-quality ceramic capacitor with 2.2-µF to 10-µF capacitance and 6.3 V or higher rated voltage as possible to
VCC and grounded to the exposed PAD and ground pins. The VCC output pin must not be loaded, left floating,
or shorted to ground during operation. Shorting VCC to ground during operation may cause damage to the
LM43603-Q1.
Undervoltage lockout (UVLO) prevents the LM43603-Q1 from operating until the VCC voltage exceeds 3.1 V
(typical). The VCC UVLO threshold has 520 mV of hysteresis (typically) to prevent undesired shuting down due to
temporary VIN droops.
The internal LDO has two inputs: primary from VIN and secondary from BIAS input. The BIAS input powers the
LDO when VBIAS is higher than the change-over threshold. Power loss of an LDO is calculated by ILDO × (VINLDO – VOUT-LDO). The higher the difference between the input and output voltages of the LDO, the more power
loss occur to supply the same output current. The BIAS input is designed to reduce the difference of the input
and output voltages of the LDO to reduce power loss and improve LM43603-Q1 efficiency, especially at light
load. It is recommended to tie the BIAS pin to VOUT when VOUT ≥ 3.3 V. Ground the BIAS pin in applications with
VOUT less than 3.3 V. BIAS input can also come from an external voltage source, if available, to reduce power
loss. When used, TI recommends a 1-µF to 10-µF high-quality ceramic capacitor to bypass the BIAS pin to
ground.
7.3.6 Soft-Start and Voltage Tracking (SS/TRK)
The LM43603-Q1 has a flexible and easy-to-use start-up rate control pin: SS/TRK. Soft-start feature is to prevent
inrush current impacting the LM43603-Q1 and its supply when power is first applied. Soft start is achieved by
slowly ramping up the target regulation voltage when the device is first enabled or powered up.
The simplest way to use the part is to leave the SS/TRK pin open circuit or floating. The LM43603-Q1 employs
the internal soft-start control ramp and starts up to the regulated output voltage in 4.1 ms typically.
In applications with a large amount of output capacitors, or higher VOUT, or other special requirements the softstart time can be extended by connecting an external capacitor CSS from SS/TRK pin to AGND. Extended softstart time further reduces the supply current needed to charge up output capacitors and supply any output
loading. An internal current source (ISSC = 2 µA) charges CSS and generates a ramp from 0 V to VFB to control
the ramp-up rate of the output voltage. For a desired soft start time tSS, the capacitance for CSS can be found
with Equation 2:
CSS ISSC u t SS
(2)
The LM43603-Q1 is capable of starting up into prebiased output conditions. When the inductor current reaches
zero, the LS switch is turned off to avoid negative current conduction. This operation mode is also called diode
emulation mode. It is built-in by the DCM operation in light loads. With a prebiased output voltage, the LM43603Q1 waits until the soft-start ramp allows regulation above the prebiased voltage and then follows the soft-start
ramp to the regulation level.
18
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Feature Description (continued)
When an external voltage ramp is applied to the SS/TRK pin, the LM43603-Q1 FB voltage follows the ramp if the
ramp magnitude is lower than the internal soft-start ramp. A resistor divider pair can be used on the external
control ramp to the SS/TRK pin to program the tracking rate of the output voltage. The final voltage seen by the
SS/TRK pin should not fall below 1.2 V to avoid abnormal operation.
EXT RAMP
RTRT
SS/TRK
RTRB
Figure 38. Soft Start Tracking External Ramp
VOUT tracked to external voltage ramps has options of ramping up slower or faster than the internal voltage ramp.
VFB always follows the lower potential of the internal voltage ramp and the voltage on the SS/TRK pin. Figure 39
shows the case when VOUT ramps slower than the internal ramp, while Figure 40 shows when VOUT ramps faster
than the internal ramp. Faster start-up time may result in inductor current tripping current protection during startup. Use with special care.
Enable
Internal SS Ramp
Ext Tracking Signal to SS pin
VOUT
Figure 39. Tracking with Longer Start-up Time than the Internal Ramp
Enable
Internal SS Ramp
Ext Tracking Signal to SS pin
VOUT
Figure 40. Tracking with Shorter Start-up Time than the Internal Ramp
7.3.7 Switching Frequency (RT) and Synchronization (SYNC)
The switching frequency of the LM43603-Q1 can be programmed by the impedance RT from the RT pin to
ground. The frequency is inversely proportional to the RT resistance. The RT pin can be left floating, and the
LM43603-Q1 operates at 500-kHz default switching frequency. The RT pin is not designed to be shorted to
ground. For a desired frequency, typical RT resistance can be found by Equation 3. Table 1 gives typical RT
values for a given FS.
RT(kΩ) = 40200 / Freq (kHz) – 0.6
(3)
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Feature Description (continued)
250
RT Resistance (kŸ)
200
150
100
50
0
0
500
1000
1500
2000
Switching Frequency (kHz)
2500
C008
Figure 41. RT vs Frequency Curve
Table 1. Typical Frequency Setting RT Resistance
FS (kHz)
RT (kΩ)
200
200
350
115
500
78.7
750
53.6
1000
39.2
1500
26.1
2000
19.6
2200
17.8
The LM43603-Q1 switching action can also be synchronized to an external clock from 200 kHz to 2.2 MHz.
Connect an external clock to the SYNC pin, with proper high-speed termination, to avoid ringing. Ground the
SYNC pin if not used.
SYNC
EXT CLOCK
RTERM
Figure 42. Frequency Synchronization
The recommendations for the external clock include high level no lower than 2 V, low level no higher than 0.4 V,
duty cycle between 10% and 90%, and both positive and negative pulse width no shorter than 80 ns. When the
external clock fails at logic high or low, the LM43603-Q1 switches at the frequency programmed by the RT
resistor after a time-out period. TI recommends connecting a resistor RT to the RT pin so that the internal
oscillator frequency is the same as the target clock frequency when the LM43603-Q1 is synchronized to an
external clock. This allows the regulator to continue operating at approximately the same switching frequency if
the external clock fails.
The choice of switching frequency is usually a compromise between conversion efficiency and the size of the
circuit. Lower switching frequency implies reduced switching losses (including gate charge losses, switch
transition losses, etc.) and usually results in higher overall efficiency. However, higher switching frequency allows
use of smaller LC output filters and hence a more compact design. Lower inductance also helps transient
response (higher large signal slew rate of inductor current), and reduces the DCR loss. The optimal switching
frequency is usually a trade-off in a given application and thus needs to be determined on a case-by-case basis.
It is related to the input voltage, output voltage, most frequent load current level(s), external component choices,
and circuit size requirement. The choice of switching frequency may also be limited if an operating condition
triggers TON-MIN or TOFF-MIN.
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Feature Description (continued)
7.3.8 Minimum ON Time, Minimum OFF Time and Frequency Foldback at Dropout Conditions
Minimum ON time, TON-MIN, is the smallest duration of time that the HS switch can be on. TON-MIN value is typically
125 ns in the LM43603-Q1. Minimum OFF time, TOFF-MIN, is the smallest duration that the HS switch can be off.
TOFF-MIN value is typically 200 ns in the LM43603-Q1.
In CCM operation, TON-MIN and TOFF-MIN limits the voltage conversion range given a selected switching frequency.
The minimum duty cycle allowed is:
DMIN = TON-MIN × FS
(4)
And the maximum duty cycle allowed is:
DMAX = 1 – TOFF-MIN × FS
(5)
Given fixed TON-MIN and TOFF-MIN, the higher the switching frequency the narrower the range of the allowed duty
cycle. In the LM43603-Q1, frequency foldback scheme is employed to extend the maximum duty cycle when
TOFF-MIN is reached. The switching frequency decreases once longer duty cycle is needed under low VIN
conditions. The switching frequency can be decreased to approximately 1/10 of the programmed frequency by RT
or the synchronization clock. Such wide range of frequency foldback allows the LM43603-Q1 output voltage stay
in regulation with a much lower supply voltage VIN. This leads to a lower effective dropout voltage. See Typical
Characteristics for more details.
Given an output voltage, the choice of the switching frequency affects the allowed input voltage range, solution
size and efficiency. The maximum operatable supply voltage can be found by:
VIN-MAX = VOUT / (FS × TON-MIN)
(6)
At lower supply voltage, the switching frequency decreases once TOFF-MIN is tripped. The minimum VIN without
frequency foldback can be approximated by Equation 7:
VIN-MIN = VOUT / (1 – FS × TOFF-MIN)
(7)
Taking considerations of power losses in the system with heavy load operation, VIN-MIN is higher than the result
calculated in Equation 7. With frequency foldback, VIN-MIN is lowered by decreased FS.
Frequency (Hz)
1000000
100000
0.1A
0.5A
1A
1.5A
2A
2.5A
10000
5.00 5.20 5.40 5.60 5.80 6.00 6.20 6.40 6.60 6.80 7.00
VIN (V)
C007
Figure 43. VOUT = 5 V Fs = 500 kHz
Frequency Foldback at Dropout
7.3.9 Internal Compensation and CFF
The LM43603-Q1 is internally compensated with RC = 400 kΩ and CC = 50 pF as shown in Functional Block
Diagram. The internal compensation is designed such that the loop response is stable over the entire operating
frequency and output voltage range. Depending on the output voltage, the compensation loop phase margin can
be low with all ceramic capacitors. TI recommends an external feed-forward capacitor, CFF, be placed in parallel
with the top resistor divider RFBT for optimum transient performance.
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Feature Description (continued)
VOUT
RFBT
CFF
FB
RFBB
Figure 44. Feed-Forward Capacitor for Loop Compensation
The feed-forward capacitor CFF in parallel with RFBT places an additional zero before the cross over frequency of
the control loop to boost phase margin. The zero frequency can be found with Equation 8:
fZ-CFF = 1 / (2π × RFBT × CFF).
(8)
An additional pole is also introduced with CFF at the frequency of
fP-CFF = 1 / (2π × CFF × (RFBT // RFBB)).
(9)
Select the CFF so that the bandwidth of the control loop without the CFF is centered between fZ-CFF and fP-CFF. The
zero fZ-CFF adds phase boost at the crossover frequency and improves transient response. The pole fP-CFF helps
maintaining proper gain margin at frequency beyond the crossover.
Designs with different combinations of output capacitors need different CFF. Different types of capacitors have
different equivalent series resistance (ESR). Ceramic capacitors have the smallest ESR and need the most CFF.
Electrolytic capacitors have much larger ESR
fZ-ESR = 1 / (2π × ESR × COUT)
(10)
and the ESR zero frequency would be low enough to boost the phase up around the crossover frequency.
Designs using mostly electrolytic capacitors at the output may not need any CFF.
The CFF creates a time constant with RFBT that couples in the attenuated output voltage ripple to the FB node. If
the CFF value is too large, it can couple too much ripple to the FB and affect VOUT regulation. It could also couple
too much transient voltage deviation and falsely trip PGOOD thresholds. Therefore, calculate CFF based on
output capacitors used in the system. At cold temperatures, the value of CFF might change based on the
tolerance of the chosen component. This may reduce its impedance and ease noise coupling on the FB node. To
avoid this, more capacitance can be added to the output or the value of CFF can be reduced. See Detailed
Design Procedure for the calculation of CFF.
7.3.10 Bootstrap Voltage (BOOT)
The driver of the HS switch requires a bias voltage higher than VIN when the HS switch is ON. The capacitor
connected between CBOOT and SW pins works as a charge pump to boost voltage on the CBOOT pin to (VSW +
VCC). The boot diode is integrated on the LM43603-Q1 die to minimize the bill of material (BOM). A synchronous
switch is also integrated in parallel with the boot diode to reduce voltage drop on CBOOT. A high-quality ceramic
0.47 µF, 6.3 V or higher capacitor is recommended for CBOOT.
7.3.11 Power Good (PGOOD)
The LM43603-Q1 has a built-in power-good flag shown on PGOOD pin to indicate whether the output voltage is
within its regulation level. The PGOOD signal can be used for start-up sequencing of multiple rails or fault
protection. The PGOOD pin is an open-drain output that requires a pullup resistor to an appropriate DC voltage.
Voltage detected by the PGOOD pin must never exceed 12 V. A resistor divider pair can be used to divide the
voltage down from a higher potential. A typical range of pullup resistor value is 10 kΩ to 100 kΩ.
When the FB voltage is within the power-good band, +4% above a –7% below the internal reference VREF
typically, the PGOOD switch will be turned off, and the PGOOD voltage will be pulled up to the voltage level
defined by the pullup resistor or divider. When the FB voltage is outside of the tolerance band, +10% above or
–13% below VREF typically, the PGOOD switch turns on, and the PGOOD pin voltage will be pulled low to
indicate power bad. Both rising and falling edges of the power-good flag have a built-in 220 µs (typical) deglitch
delay.
22
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Feature Description (continued)
7.3.12 Overcurrent and Short-Circuit Protection
The LM43603-Q1 is protected from overcurrent conditions by cycle-by-cycle current limiting on both the peak and
valley of the inductor current. Hiccup mode is activated if a fault condition persists to prevent over heating.
High-side MOSFET overcurrent protection is implemented by the nature of the peak current mode control. The
HS switch current is sensed when the HS is turned on after a set blanking time. The HS switch current is
compared to the output of the error amplifier (EA) minus slope compensation every switching cycle. Refer to
Functional Block Diagram for more details. The peak current of the HS switch is limited by the maximum EA
output voltage minus the slope compensation at every switching cycle. The slope compensation magnitude at the
peak current is proportional to the duty cycle.
When the LS switch is turned on, the current going through it is also sensed and monitored. The LS switch is not
turned OFF at the end of a switching cycle if its current is above the LS current limit ILS-LIMIT. The LS switch is
kept ON so that inductor current keeps ramping down, until the inductor current ramps below the LS current limit.
Then the LS switch is turned OFF, and the HS switch is turned on, after a dead time. If the current of the LS
switch is higher than the LS current limit for 32 consecutive cycles, and the power-good flag is low, hiccup
current protection mode is activated. In hiccup mode, the regulator is shut down and kept off for 5.5 ms typically
before the LM43603-Q1 tries to start again. If an overcurrent or short-circuit fault condition still exists, hiccup
repeats until the fault condition is removed. Hiccup mode reduces power dissipation under severe overcurrent
conditions, prevents overheating, and potential damage to the device.
Hiccup is only activated when power-good flag is low. Under non-severe overcurrent conditions when VOUT has
not fallen outside of the PGOOD tolerance band, the LM43603-Q1 reduces the switching frequency and keep the
inductor current valley clamped at the LS current limit level. This operation mode allows slight over current
operation during load transients without tripping hiccup. If the power-good flag becomes low, hiccup operation
starts after LS current limit is tripped 32 consecutive cycles.
7.3.13 Thermal Shutdown
Thermal shutdown is a built-in self protection to limit junction temperature and prevent damage due to
overheating. Thermal shutdown turns off the device when the junction temperature exceeds 160°C typically to
prevent further power dissipation and temperature rise. Junction temperature reduces after thermal shutdown.
The LM43603-Q1 restarts when the junction temperature drops to 150°C.
7.4 Device Functional Modes
7.4.1 Shutdown Mode
The EN pin provides electrical ON and OFF control for the LM43603-Q1. When VEN is below 0.4 V, the device is
in shutdown mode. Both the internal LDO and the switching regulator are off. In shutdown mode the quiescent
current drops to 1.2 µA typically with VIN = 12 V. The LM43603-Q1 also employs undervoltage lockout protection.
If VCC voltage is below the UVLO level, the output of the regulator is turned off.
7.4.2 Stand-by Mode
The internal LDO has a lower enable threshold than the regulator. When VEN is above 1.2 V and below the
precision enable falling threshold (1.8 V typically), the internal LDO regulates the VCC voltage at 3.2 V. The
precision enable circuitry is turned on once VCC is above the UVLO threshold. The switching action and voltage
regulation are not enabled unless VEN rises above the precision enable threshold (2.1 V typically).
7.4.3 Active Mode
The LM43603-Q1 is in active mode when VEN is above the precision enable threshold and VCC is above its UVLO
level. The simplest way to enable the LM43603-Q1 is to connect the EN pin to VIN. This allows self start-up when
the input voltage is in the operation range: 3.5 V to 36 V. See Enable (EN) and VCC, UVLO, and BIAS for details
on setting these operating levels.
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Device Functional Modes (continued)
In active mode, depending on the load current, the LM43603-Q1 will be in one of four modes:
1. Continuous conduction mode (CCM) with fixed switching frequency when load current is above half of the
peak-to-peak inductor current ripple;
2. Discontinuous conduction mode (DCM) with fixed switching frequency when load current is lower than half of
the peak-to-peak inductor current ripple in CCM operation;
3. Pulse frequency modulation (PFM) when switching frequency is decreased at very light load;
4. Fold-back mode when switching frequency is decreased to maintain output regulation at lower supply voltage
VIN.
7.4.4 CCM Mode
CCM operation is employed in the LM43603-Q1 when the load current is higher than half of the peak-to-peak
inductor current. In CCM operation, the frequency of operation is fixed by internal oscillator unless the the
minimum HS switch ON time (TON_MIN) or OFF time (TOFF_MIN) is exceeded. Output voltage ripple is at a minimum
in this mode and the maximum output current of 2 A can be supplied by the LM43603-Q1.
7.4.5 Light Load Operation
When the load current is lower than half of the peak-to-peak inductor current in CCM, the LM43603-Q1 operates
in DCM, also known as diode emulation mode (DEM). In DCM operation, the LS FET is turned off when the
inductor current drops to 0 A to improve efficiency. Both switching losses and conduction losses are reduced in
DCM, comparing to forced PWM operation at light load.
At even lighter current loads, PFM is activated to maintain high efficiency operation. When the HS switch ON
time reduces to TON_MIN or peak inductor current reduces to its minimum IPEAK-MIN, the switching frequency
reduces to maintain proper regulation. Efficiency is greatly improved by reducing switching and gate drive losses.
Frequency (Hz)
1000000
100000
10000
8V
12V
24V
36V
1000
0.001
0.01
0.1
1
Current (A)
C007
Figure 45. VOUT = 5 V, Fs = 500 kHz
Pulse Frequency Mode Operation
7.4.6 Self-Bias Mode
For highest efficiency of operation,TI recommends that the BIAS pin be connected directly to VOUT when VOUT ≥
3.3 V. In this self-bias mode of operation, the difference between the input and output voltages of the internal
LDO are reduced and therefore the total efficiency is improved. These efficiency gains are more evident during
light load operation. During this mode of operation, the LM43603-Q1 operates with a minimum quiescent current
of 27 µA (typical). See VCC, UVLO, and BIAS for more details.
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8 Applications and Implementation
NOTE
Information in the following applications sections is not part of the TI component
specification, and TI does not warrant its accuracy or completeness. TI’s customers are
responsible for determining suitability of components for their purposes. Customers should
validate and test their design implementation to confirm system functionality.
8.1 Application Information
The LM43603-Q1 is a step-down DC-DC regulator. It is typically used to convert a higher DC voltage to a lower
DC voltage with a maximum output current of 3 A. The following design procedure can be used to select
components for the LM43603-Q1.
8.2 Typical Applications
The LM43603-Q1 only requires a few external components to convert from a wide voltage range supply to a
fixed output voltage. Figure 46 shows a basic schematic when BIAS is connected to VOUT and this is
recommended for VOUT ≥ 3.3 V. For VOUT < 3.3 V, connect BIAS to ground, as shown in Figure 47.
L
VIN
VIN
CIN
VOUT
SW
LM43603Q
ENABLE
CBOOT
COUT
CBIAS
CFF
CBOOT
BIAS
PGOOD
SS/TRK
RT
RFBT
FB
VCC
SYNC
AGND
RFBB
CVCC
PGND
C001
Figure 46. LM43603-Q1 Basic Schematic for VOUT ≥ 3.3 V, tie BIAS to VOUT
L
VIN
VIN
CIN
VOUT
SW
LM43603Q
CBOOT
COUT
CBOOT
ENABLE
PGOOD
BIAS
CFF
SS/TRK
RT
SYNC
AGND
RFBT
FB
VCC
CVCC
RFBB
PGND
Figure 47. LM43603-Q1 Basic Schematic for VOUT < 3.3 V, tie BIAS to Ground
The LM43603-Q1 also integrates a full list of optional features to aid system design requirements such as
precision enable, VCC UVLO, programmable soft start, output voltage tracking, programmable switching
frequency, clock synchronization and power-good indication. Each application can select the features for a more
comprehensive design. A schematic with all features utilized is shown in Figure 48.
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Typical Applications (continued)
L
VIN
RENT
VOUT
SW
VIN
COUT
LM43603Q
CIN
ENABLE
RENB
RFBT
CBOOT
CBOOT
FB
VCC
RFBB
CVCC
SS/TRK
CSS
RT
CFF
BIAS
RT
CBIAS
SYNC
PGOOD
AGND
PGND
RPG
RSYNC
Tie BIAS to PGND
when VOUT < 3.3 V
Figure 48. LM43603-Q1 Schematic with All Features
The external components must fulfill the needs of the application, as well as the stability criteria of the device
control loop. The LM43603-Q1 is optimized to work within a range of external components. The inductance and
capacitance of the LC output filter must considered in conjunction, creating a double pole, responsible for the
corner frequency of the converter. Table 2 can be used to simplify the output filter component selection.
Table 2. L, COUT and CFF Typical Values
(1)
(2)
(3)
(4)
(5)
26
FS (kHz)
VOUT (V)
L (µH) (1)
200
1
4.8
600
none
200
100
COUT (µF)
(2)
CFF (pF)
(3) (4)
RT (kΩ)
RFBB (kΩ)
500
1
2.2
400
none
80.6 or open
100
1000
1
1
250
none
39.2
100
2200
1
0.47
150
none
17.8
100
200
3.3
15
300
470
200
43.2
500
3.3
4.7
150
330
80.6 or open
43.2
1000
3.3
3.3
100
220
39.2
43.2
2200
3.3
1
50
180
17.8
43.2
200
5
18
200
680
200
24.9
500
5
6.8
120
440
80.6 or open
24.9
1000
5
3.3
100
330
39.2
24.9
2200
5
1.5
50
220
17.8
24.9
200
12
33
100
See (5)
200
9.09
500
12
15
50
680
80.6 or open
9.09
1000
12
6.8
44
560
39.2
9.09
200
24
44
47
See (5)
200
4.32
500
24
18
47
See (5)
80.6 or open
4.32
1000
24
10
33
See (5)
39.2
4.32
(3) (4)
Inductance value is calculated based on VIN = 12 V, except for VOUT = 12 V and VOUT = 24 V, the VIN value is 24 V and 48 V,
respectively.
All the COUT values are after derating. Add more when using ceramics.
RFBT = 0 Ω for VOUT = 1 V. RFBT = 100 kΩ for all other VOUT settings.
For designs with RFBT other than 100 kΩ, adjust CFF so that (CFF × RFBT) is unchanged and adjust RFBB such that (RFBT / RFBB) is
unchanged.
High ESR COUT will give enough phase boost, and CFF is not needed.
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Typical Applications (continued)
8.2.1 Design Requirements
Detailed design procedure is described based on a design example. For this design example, use the
parameters listed in Table 3 as the input parameters.
Table 3. Design Example Parameters
DESIGN PARAMETER
VALUE
Input voltage VIN
12 V typical, range from 3.5 V to 36 V
Output voltage VOUT
3.3 V
Input ripple voltage
400 mV
Output ripple voltage
30 mV
Output current rating
3A
Operating frequency
500 kHz
Soft-start time
10 ms
8.2.2 Detailed Design Procedure
8.2.2.1 Custom Design With WEBENCH® Tools
Click here to create a custom design using the LM43603-Q1 device with the WEBENCH® Power Designer.
1. Start by entering the input voltage (VIN), output voltage (VOUT), and output current (IOUT) requirements.
2. Optimize the design for key parameters such as efficiency, footprint, and cost using the optimizer dial.
3. Compare the generated design with other possible solutions from Texas Instruments.
The WEBENCH Power Designer provides a customized schematic along with a list of materials with real-time
pricing and component availability.
In most cases, these actions are available:
• Run electrical simulations to see important waveforms and circuit performance
• Run thermal simulations to understand board thermal performance
• Export customized schematic and layout into popular CAD formats
• Print PDF reports for the design, and share the design with colleagues
Get more information about WEBENCH tools at www.ti.com/WEBENCH.
8.2.2.2 Output Voltage Setpoint
The output voltage of the LM43603-Q1 device is externally adjustable using a resistor divider network. The
divider network is comprised of top feedback resistor RFBT and bottom feedback resistor RFBB. Equation 11 is
used to determine the output voltage of the converter:
VFB
RFBB
RFBT
VOUT VFB
(11)
Choose the value of the RFBT to be 100 kΩ to minimize quiescent current to improve light load efficiency in this
application. With the desired output voltage set to be 3.3 V and the VFB = 1.015 V, the RFBB value can then be
calculated using Equation 11. The formula yields a value of 43.478 kΩ. Choose the closest available value of
43.2 kΩ for the RFBB. See Adjustable Output Voltage for more details.
8.2.2.3 Switching Frequency
The default switching frequency of the LM43603-Q1 device is set at 500 kHz when RT pin is open circuit. The
switching frequency is selected to be 500 kHz in this application for one less passive components. If other
frequency is desired, use Equation 12 to calculate the required value for RT.
RT(kΩ) = 40200 / Freq (kHz) – 0.6
(12)
For 500 kHz, the calculated RT is 79.8 kΩ and standard value 80.6 kΩ can also be used to set the switching
frequency at 500 kHz.
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8.2.2.4 Input Capacitors
The LM43603-Q1 device requires high frequency input decoupling capacitor(s) and a bulk input capacitor,
depending on the application. The typical recommended value for the high frequency decoupling capacitor is
between 4.7 µF to 10 µF. TI recommends a high-quality ceramic type X5R or X7R with sufficiency voltage rating.
The voltage rating must be greater than the maximum input voltage. To compensate the derating of ceramic
capactors, TI recommends a voltage rating of twice the maximum input voltage. Additionally, some bulk
capacitance can be required, especially if the LM43603-Q1 circuit is not located within approximately 5 cm from
the input voltage source. This capacitor is used to provide damping to the voltage spiking due to the lead
inductance of the cable or trace. The value for this capacitor is not critical but must be rated to handle the
maximum input voltage including ripple. For this design, a 10-µF, X7R dielectric capacitor rated for 100 V is used
for the input decoupling capacitor. The ESR is approximately 3 mΩ, and the current-rating is 3 A. Include a
capacitor with a value of 0.1 µF for high-frequency filtering and place it as close as possible to the device pins.
NOTE
DC Bias effect: High capacitance ceramic capacitors have a DC bias effect, which will
have a strong influence on the final effective capacitance. Therefore, carefully choose the
correct capacitor value. Package size and voltage rating in combination with dielectric
material are responsible for differences between the rated capacitor value and the
effective capacitance.
8.2.2.5 Inductor Selection
The first criterion for selecting an output inductor is the inductance itself. In most buck converters, this value is
based on the desired peak-to-peak ripple current, ΔiL, that flows in the inductor along with the DC load current.
As with switching frequency, the selection of the inductor is a tradeoff between size and cost. Higher inductance
gives lower ripple current and hence lower output voltage ripple with the same output capacitors. Lower
inductance could result in smaller, less expensive component. An inductance that gives a ripple current of 20% to
40% of the 3 A at the typical supply voltage is a good starting point. ΔiL = (1/5 to 2/5) × IOUT. The peak-to-peak
inductor current ripple can be found by Equation 13 and the range of inductance can be found by Equation 14
with the typical input voltage used as VIN.
'iL
(VIN
VOUT ) u D
L u FS
(13)
(VIN VOUT ) u D
(V
VOUT ) u D
d L d IN
0.4 u FS u IL MAX
0.2 u FS u IL MAX
(14)
D is the duty cycle of the converter where in a buck converter case it can be approximated as D = VOUT / VIN,
assuming no loss power conversion. By calculating in terms of amperes, volts, and megahertz, the inductance
value will come out in micro Henries. The inductor ripple current ratio is defined by:
'iL
r
IOUT
(15)
The second criterion is inductor saturation current rating. The inductor must be rated to handle the maximum
load current plus the ripple current:
IL-PEAK = ILOAD-MAX + ΔiL/ 2
(16)
The LM43603-Q1 has both valley current limit and peak current limit. During an instantaneous short, the peak
inductor current can be high due to a momentary increase in duty cycle. The inductor current rating must be
higher than the HS current limit. It is advised to select an inductor with a larger core saturation margin and
preferably a softer roll off of the inductance value over load current.
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In general, it is preferable to choose lower inductance in switching power supplies, because it usually
corresponds to faster transient response, smaller DCR, and reduced size for more compact designs. But too low
of an inductance can generate too large of an inductor current ripple such that over current protection at the full
load could be falsely triggered. It also generates more conduction loss, because the RMS current is slightly
higher relative that with lower current ripple at the same DC current. Larger inductor current ripple also implies
larger output voltage ripple with the same output capacitors. With peak current mode control, it is not
recommended to have too small of an inductor current ripple. A larger peak current ripple improves the
comparator signal to noise ratio.
Once the inductance is determined, the type of inductor must be selected. Ferrite designs have very low core
losses and are preferred at high switching frequencies, so design goals can concentrate on copper loss and
preventing saturation. Ferrite core material saturates hard, which means that inductance collapses abruptly when
the peak design current is exceeded. The ‘hard’ saturation results in an abrupt increase in inductor ripple current
and consequent output voltage ripple. Do not allow the core to saturate!
For the design example, a standard 6.8 μH inductor from Würth Elektronik, Coilcraft, or Vishay can be used for
the 3.3 V output with plenty of current rating margin.
8.2.2.6 Output Capacitor Selection
The device is designed to be used with a wide variety of LC filters. Use as little output capacitance as possible to
keep cost and size down. Choose the output capacitor(s), COUT, with care because it directly affects the steady
state output voltage ripple, loop stability and the voltage over/undershoot during load current transients.
The output voltage ripple is essentially composed of two parts. One is caused by the inductor current ripple going
through the ESR of the output capacitors:
ΔVOUT-ESR = ΔiL × ESR
(17)
The other is caused by the inductor current ripple charging and discharging the output capacitors:
ΔVOUT-C =ΔiL/ (8 × FS × COUT )
(18)
The two components in the voltage ripple are not in phase, so the actual peak-to-peak ripple is smaller than the
sum of the two peaks.
Output capacitance is usually limited by transient performance specifications if the system requires tight voltage
regulation with presence of large current steps and fast slew rates. When a fast large load transient happens,
output capacitors provide the required charge before the inductor current can slew to the appropriate level. The
initial output voltage step is equal to the load current step multiplied by the ESR. VOUT continues to droop until
the control loop response increases or decreases the inductor current to supply the load. To maintain a small
overshoot or undershoot during a transient, small ESR and large capacitance are desired. But these also come
with higher cost and size. Thus, the motivation is to seek a fast control loop response to reduce the output
voltage deviation.
For a given input and output requirement, Equation 19 gives an approximation for an absolute minimum output
capacitor required:
COUT !
ª§ r 2
·
1
u «¨ u (1 Dc) ¸
¸
(FS u r u 'VOUT / IOUT ) «¬¨© 12
¹
º
Dc u (1 r) »
»¼
(19)
Along with this for the same requirement, calculate the maximum ESR as per Equation 20:
ESR
Dc
1
u ( 0.5)
FS u COUT r
where
•
•
•
•
•
r = Ripple ratio of the inductor ripple current (ΔIL / IOUT)
ΔVOUT = target output voltage undershoot
D’ = 1 – duty cycle
FS = switching frequency
IOUT = load current
(20)
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A general guideline for COUT range is that COUT must be larger than the minimum required output capacitance
calculated by Equation 19, and smaller than 10 times the minimum required output capacitance or 1 mF. In
applications with VOUT less than 3.3 V, it is critical that low ESR output capacitors are selected. This limits
potential output voltage overshoots as the input voltage falls below the device normal operating range. To
optimize the transient behavior a feed-forward capacitor could be added in parallel with the upper feedback
resistor. For this design example, three 47-µF, 10-V, X7R ceramic capacitors are used in parallel.
8.2.2.7 Feed-Forward Capacitor
The LM43603-Q1 is internally compensated and the internal R-C values are 400 kΩ and 50 pF, respectively.
Depending on the VOUT and frequency FS, if the output capacitor COUT is dominated by low ESR (ceramic types)
capacitors, it could result in low phase margin. To improve the phase boost an external feedforward capacitor
CFF can be added in parallel with RFBT. CFF is chosen such that phase margin is boosted at the crossover
frequency without CFF. A simple estimation for the crossover frequency without CFF (fx) is shown in Equation 21,
assuming COUT has very small ESR.
5.3
fx
VOUT u COUT
(21)
Equation 22 was tested for CFF:
CFF
1
1
u
2Sfx
RFBT u (RFBT / /RFBB )
(22)
This equation indicates that the crossover frequency is geometrically centered on the zero and pole frequencies
caused by the CFF capacitor.
For designs with higher ESR, CFF is not needed when COUT has very high ESR, and CFF calculated from
Equation 22 must be reduced with medium ESR. Table 2 can be used as a quick starting point.
For the application in this design example, a 470 pF COG capacitor is selected.
8.2.2.8 Bootstrap Capacitors
Every LM43603-Q1 design requires a bootstrap capacitor, CBOOT. The recommended bootstrap capacitor value is
0.47 μF and rated at 6.3 V or higher. The bootstrap capacitor is located between the SW pin and the CBOOT
pin. The bootstrap capacitor must be a high-quality ceramic type with X7R or X5R grade dielectric for
temperature stability.
8.2.2.9 VCC Capacitor
The VCC pin is the output of an internal LDO for LM43603-Q1. The input for this LDO comes from either VIN or
BIAS (see Functional Block Diagram for LM43603-Q1). To insure stability of the part, place a minimum of 2.2-µF,
10-V capacitor for this pin to ground.
8.2.2.10 BIAS Capacitors
For an output voltage of 3.3 V and greater, the BIAS pin can be connected to the output in order to increase light
load efficiency. This pin is an input for the VCC LDO. When BIAS is not connected, the input for the VCC LDO is
internally connected into VIN. Because this is an LDO, the voltage differences between the input and output
affects the efficiency of the LDO. If necessary, a capacitor with a value of 1 μF can be added close to the BIAS
pin as an input capacitor for the LDO.
30
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8.2.2.11 Soft-Start Capacitors
The user can left the SS/TRK pin floating, and the LM43603-Q1 implements a soft-start time of 4.1 ms typically.
In order to use an external soft-start capacitor, the capacitor must be sized so that the soft start time is longer
than 4.1 ms. Use Equation 23 in order to calculate the soft start capacitor value:
CSS ISSC u t SS
where
•
•
•
CSS = Soft start capacitor value (µF)
ISS = Soft start charging current (µA)
tSS = Desired soft start time (s)
(23)
For the desired soft-start time of 10 ms and soft-start charging current of 2 µA, Equation 23 above yield a soft
start capacitor value of 0.02 µF.
8.2.2.12 Undervoltage Lockout Setpoint
The undervoltage lockout (UVLO) is adjusted using the external voltage divider network of RENT and RENB. RENT
is connected between the VIN pin and the EN pin of the LM43603-Q1. RENB is connected between the EN pin
and the GND pin. The UVLO has two thresholds, one for power up when the input voltage is rising and one for
power down or brownouts when the input voltage is falling. Equation 24 can be used to determine the VIN UVLO
level.
VIN-UVLO-RISING = VENH × (RENB + RENT) / RENB
(24)
The EN rising threshold (VENH) for LM43603-Q1 is set to be 2.2 V (typical). Choose the value of RENB to be 1 MΩ
to minimize input current from the supply. If the desired VIN UVLO level is at 5 V, then the value of RENT can be
calculated using Equation 25:
RENT = (VIN-UVLO-RISING / VENH – 1) × RENB
(25)
Equation 25 yields a value of 1.27 MΩ. The resulting falling UVLO threshold, equals 4.3 V, can be calculated by
Equation 26, where EN falling threshold (VENL) is 1.9 V (typical).
VIN-UVLO-FALLING = VENL × (RENB + RENT) / RENB
(26)
8.2.2.13 PGOOD
A typical pullup resistor value is 10 kΩ to 100 kΩ from PGOOD pin to a voltage no higher than 12 V. If it is
desired to pull up PGOOD pin to a voltage higher than 12 V, a resistor can be added from PGOOD pin to ground
to divide the voltage seen by the PGOOD pin to a value no higher than 12 V.
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8.2.3 Application Performance Curves
Unless otherwise specified, VIN = 12 V, VOUT = 3.3 V, FS = 500 kHz and room temperature. See below for component values
for each VOUT and FS combination.
2.2 µH
VIN
VIN
100
1 VOUT
SW
4.7
µF
LM43603Q
SS/TRK
VCC
0.47 µF
400 µF
100 kΩ
90
Open
Open
2.2 µF
RT
SYNC
AGND
80
Efficiency (%)
ENABLE
CBOOT
FB
BIAS
PGOOD
70
60
3.5VIN
PGND
50
5VIN
12VIN
40
0.001
0.01
0.1
1
Current (A)
VOUT = 1 V
FS = 500 kHz
VOUT = 1 V
Figure 49. Component Values for VOUT= 1 V,
FS = 500 kHz
Figure 50. Efficiency
1.050
1000000
3.5VIN
1.040
5VIN
12VIN
Frequency (Hz)
1.030
VOUT (V)
C001
Fs = 500 kHz
1.020
1.010
1.000
100000
10000
3.5VIN
5VIN
8VIN
12VIN
0.990
0.980
0.001
0.01
0.1
Current (A)
VOUT = 1 V
1000
0.001
1
0.01
0.1
1
Current (A)
C001
FS = 500 kHz
VOUT = 1 V
Figure 51. Output Voltage Regulation
C007
FS = 500 kHz
Figure 52. Frequency vs Load
3.5
3.0
IOUT (500 mA/DIV)
Curremt (A)
2.5
VOUT (50 mV/DIV)
2.0
1.5
12VIN
1.0
18VIN
0.5
24VIN
0.0
65
Time (100µs/DIV)
75
VIN = 12 V
VOUT = 1 V
VOUT = 1 V
Figure 53. Load Transient 0.1 A to 1 A
32
85
95
105
Ambient Temperature (ƒC)
C001
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FS = 500 kHz
115
125
C050
RθJA = 20°C/W
Figure 54. Derating Curve
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Unless otherwise specified, VIN = 12 V, VOUT = 3.3 V, FS = 500 kHz and room temperature. See below for component values
for each VOUT and FS combination.
6.8 µH
VIN
VIN
4.7
µF
100
3.3 VOUT
SW
LM43603Q
0.47 µF
120 µF
VCC
SS/TRK
100 kΩ
43.2 kΩ
2.2 µF
RT
BIAS
SYNC
PGOOD
AGND
90
330 pF
Efficiency (%)
ENABLE
CBOOT
FB
80
70
60
5VIN
PGND
50
12VIN
24VIN
40
0
0.5
1
1.5
2
2.5
3
Current (A)
VOUT = 3.3 V
FS = 500 kHz
VOUT = 3.3 V
FS = 500 kHz
Figure 56. Efficiency at Room Temperature
100
100
90
90
80
80
Efficiency (%)
Efficiency (%)
Figure 55. Component Values for VOUT = 3.3 V,
FS = 500 kHz
C001
70
60
70
60
5VIN
50
5VIN
50
12VIN
12VIN
24VIN
40
0.001
0.01
0.1
1
Current (A)
VOUT = 3.3 V
24VIN
40
0.001
FS = 500 kHz
VOUT = 3.3 V
3
12VIN
C001
FS = 500 kHz
5VIN
12VIN
2.5
24VIN
Power Dissipation (W)
Power Dissipation (W)
1
Figure 58. Efficiency at 85ºC Ambient Temperature
5VIN
2.5
0.1
Current (A)
Figure 57. Efficiency at Room Temperature
3
0.01
C001
2
1.5
1
0.5
24VIN
2
1.5
1
0.5
0
0
0
0.5
1
1.5
2
2.5
Current (A)
VOUT = 3.3 V
3
0
0.5
FS = 500 kHz
VOUT = 3.3 V
Figure 59. Power Loss at Room Temperature
1
1.5
2
2.5
Current (A)
C001
3
C001
FS = 500 kHz
Figure 60. Power Loss at 85°C Ambient Temperature
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Unless otherwise specified, VIN = 12 V, VOUT = 3.3 V, FS = 500 kHz and room temperature. See below for component values
for each VOUT and FS combination.
1000000
3.5
FREQUENCY (Hz)
3.4
VOUT (V)
3.3
3.2
0.1A
0.5A
1A
1.5A
2A
2.5A
3.1
3.0
100000
0.1A
0.5A
1A
1.5A
2A
2.5A
10000
1000
2.9
3.5
3.7
3.9
4.1
4.3
VIN (V)
VOUT = 3.3 V
3.5
4.5
3.7
3.9
4.1
FS = 500 kHz
VOUT = 3.3 V
Figure 61. Dropout Curve
4.5
C007
FS = 500 kHz
Figure 62. Frequency vs VIN
1000000
3.40
5VIN
3.38
12VIN
3.36
24VIN
3.34
100000
VOUT (V)
FREQUENCY (Hz)
4.3
VIN (V)
C007
5VIN
8VIN
12VIN
24VIN
10000
1000
0.001
0.01
0.1
VOUT = 3.3 V
3.30
3.28
3.26
3.24
3.22
3.20
0.001
1
Current (A)
3.32
0.01
0.1
1
Current (A)
C007
FS = 500 kHz
VOUT = 3.3 V
Figure 63. Frequency vs Load
C001
FS = 500 kHz
Figure 64. Output Voltage Regulation
3.5
3.0
IOUT (1A/DIV)
Curremt (A)
2.5
2.0
1.5
12VIN
1.0
VOUT (200 mV/DIV)
18VIN
0.5
24VIN
0.0
65
Time (100µs/DIV)
75
VOUT = 3.3 V
FS = 500 kHz
VOUT = 3.3 V
Figure 65. Load Transient 0.1 A to 2 A
34
85
95
105
Ambient Temperature (ƒC)
C001
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FS = 500 kHz
115
125
C050
RθJA = 20°C/W
Figure 66. Derating Curve
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Unless otherwise specified, VIN = 12 V, VOUT = 3.3 V, FS = 500 kHz and room temperature. See below for component values
for each VOUT and FS combination.
22 µH
VIN
VIN
4.7
µF
100
5 VOUT
SW
LM43603Q
0.47 µF
VCC
SS/TRK
2.2 µF
150 µF
100 kΩ
24.9 kΩ
RT
200 kΩ
SYNC
BIAS
PGOOD
AGND
90
680 pF
Efficiency (%)
ENABLE
CBOOT
FB
80
70
60
PGND
12VIN
50
24VIN
40
0.001
0.01
0.1
1
Current (A)
VOUT = 5 V
FS = 200 kHz
VOUT = 5 V
Figure 67. Component Values for VOUT = 5 V,
FS = 200 kHz
5.25
Figure 68. Efficiency at Room Temperature
5.4
8VIN
5.20
12VIN
5.15
5.2
24VIN
5.0
VOUT (V)
5.10
VOUT (V)
C001
FS = 200 kHz
5.05
5.00
4.95
4.90
4.8
4.6
0.1A
0.5A
1A
1.5A
2A
2.5A
4.4
4.85
4.2
4.80
4.75
0.001
0.01
0.1
Current (A)
VOUT = 5 V
4.0
5.00
1
5.20
5.40
5.60
5.80
6.00
6.20
VIN (V)
C003
FS = 200 kHz
VOUT = 5 V
Figure 69. Output Voltage Regulation
6.40
C007
FS = 200 kHz
Figure 70. Dropout Curve
3.5
3.0
IOUT (1A/DIV)
Curremt (A)
2.5
2.0
1.5
12VIN
18VIN
24VIN
28VIN
1.0
VOUT (200 mV/DIV)
0.5
0.0
65
Time (100µs/DIV)
75
VOUT = 5 V
FS = 200 kHz
85
95
105
115
Ambient Temperature (ƒC)
C001
VOUT = 5 V
Figure 71. Load Transient 0.1 A to 2 A
FS = 200 kHz
C050
RθJA = 20°C/W
Figure 72. Derating Curve
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Unless otherwise specified, VIN = 12 V, VOUT = 3.3 V, FS = 500 kHz and room temperature. See below for component values
for each VOUT and FS combination.
10 µH
VIN
VIN
4.7
µF
100
5 VOUT
SW
LM43603Q
0.47 µF
VCC
SS/TRK
2.2 µF
100 µF
100 kΩ
24.9 kΩ
RT
BIAS
SYNC
PGOOD
AGND
90
440 pF
Efficiency (%)
ENABLE
CBOOT
FB
80
70
60
PGND
12VIN
50
24VIN
40
0.001
0.01
0.1
1
Current (A)
VOUT = 5 V
FS = 500 kHz
VOUT = 5 V
Figure 73. Component Values for VOUT = 5 V,
FS = 500 kHz
5.25
Figure 74. Efficiency at Room Temperature
5.4
8VIN
5.20
12VIN
5.15
5.2
24VIN
5.0
VOUT (V)
5.10
VOUT (V)
C001
FS = 500 kHz
5.05
5.00
4.95
4.90
4.8
4.6
0.1A
0.5A
1A
1.5A
2A
2.5A
4.4
4.85
4.2
4.80
4.75
0.001
0.01
0.1
Current (A)
VOUT = 5 V
4.0
5.00 5.20 5.40 5.60 5.80 6.00 6.20 6.40 6.60 6.80 7.00
1
VIN (V)
C004
FS = 500 kHz
VOUT = 5 V
Figure 75. Output Voltage Regulation
C007
FS = 500 kHz
Figure 76. Dropout Curve
3.5
3.0
IOUT (1A/DIV)
Curremt (A)
2.5
2.0
1.5
12VIN
18VIN
24VIN
28VIN
1.0
VOUT (200 mV/DIV)
0.5
0.0
65
Time (100µs/DIV)
75
VOUT = 5 V
FS = 500 kHz
VOUT = 5 V
Figure 77. Load Transient 0.1 A to 2 A
36
85
95
105
Ambient Temperature (ƒC)
C001
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FS = 500 kHz
115
125
C050
RθJA = 20°C/W
Figure 78. Derating Curve
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SNVSA82C – APRIL 2015 – REVISED OCTOBER 2017
Unless otherwise specified, VIN = 12 V, VOUT = 3.3 V, FS = 500 kHz and room temperature. See below for component values
for each VOUT and FS combination.
4.7 µH
VIN
VIN
4.7
µF
100
5 VOUT
SW
LM43603Q
0.47 µF
68 µF
VCC
SS/TRK
100 kΩ
24.9 kΩ
2.2 µF
RT
39.2 kΩ
BIAS
SYNC
PGOOD
AGND
90
330 pF
80
Efficiency (%)
ENABLE
CBOOT
FB
70
60
PGND
12VIN
50
24VIN
40
0.001
0.01
0.1
1
Current (A)
VOUT = 5 V
FS = 1 MHz
VOUT = 5 V
Figure 79. Component Values for VOUT = 5 V,
FS = 1 MHz
5.25
Figure 80. Efficiency
5.40
8VIN
5.20
12VIN
5.15
5.20
24VIN
5.00
VOUT (V)
5.10
VOUT (V)
C001
FS = 1 MHz
5.05
5.00
4.95
4.90
4.80
4.60
0.1A
0.5A
1A
1.5A
2A
2.5A
4.40
4.85
4.20
4.80
4.75
0.001
0.01
0.1
Current (A)
VOUT = 5 V
4.00
5.00 5.20 5.40 5.60 5.80 6.00 6.20 6.40 6.60 6.80 7.00
1
VIN (V)
C005
FS = 1 MHz
VOUT = 5 V
Figure 81. Output Voltage Regulation
C007
FS = 1 MHz
Figure 82. Dropout Curve
3.5
3.0
IOUT (1A/DIV)
Curremt (A)
2.5
2.0
1.5
12VIN
18VIN
24VIN
28VIN
1.0
VOUT (200 mV/DIV)
0.5
0.0
65
Time (100µs/DIV)
75
VOUT = 5 V
FS = 1 MHz
85
95
105
115
Ambient Temperature (ƒC)
C001
VOUT = 5 V
Figure 83. Load Transient
FS = 1 MHz
125
C050
RθJA = 20°C/W
Figure 84. Derating Curve
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Unless otherwise specified, VIN = 12 V, VOUT = 3.3 V, FS = 500 kHz and room temperature. See below for component values
for each VOUT and FS combination.
2.2 µH
VIN
VIN
4.7
µF
100
5 VOUT
SW
LM43603Q
0.47 µF
68 µF
VCC
SS/TRK
100 kΩ
24.9 kΩ
2.2 µF
RT
17.8 kΩ
BIAS
SYNC
PGOOD
AGND
90
220 pF
80
Efficiency (%)
ENABLE
CBOOT
FB
70
60
PGND
12VIN
50
16VIN
40
0.001
0.01
0.1
1
Current (A)
VOUT = 5 V
FS = 2.2 MHz
VOUT = 5 V
C001
FS = 2.2 MHz
Figure 85. Component Values for VOUT = 5 V,
FS = 2.2 MHz
Figure 86. Efficiency
5.25
5.40
5.20
5.20
5.15
5.00
5.05
VOUT (V)
VOUT (V)
5.10
5.00
4.95
4.90
4.80
4.60
0.01A
0.1A
0.5A
1A
1.5A
2A
2.5A
4.40
4.85
12VIN
4.80
16VIN
4.75
0.001
0.01
0.1
4.00
5.00
1
Current (A)
VOUT = 5 V
4.20
5.20
5.40
5.60
5.80
6.00
6.20
VIN (V)
C001
FS = 2.2 MHz
VOUT = 5 V
6.40
C007
FS = 2.2 MHz
Figure 87. Output Voltage Regulation
Figure 88. Dropout Curve
3.5
3.0
IOUT (1A/DIV)
Curremt (A)
2.5
VOUT (200 mV/DIV)
2.0
1.5
1.0
0.5
12VIN
0.0
65
Time (100µs/DIV)
75
VOUT = 5 V
FS = 2.2 MHz
VOUT = 5 V
Figure 89. Load Transient
38
85
95
105
Ambient Temperature (ƒC)
C001
FS = 2.2 MHz
115
125
C050
RθJA = 20°C/W
Figure 90. Derating Curve
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Unless otherwise specified, VIN = 12 V, VOUT = 3.3 V, FS = 500 kHz and room temperature. See below for component values
for each VOUT and FS combination.
16 µH
VIN
VIN
100
12 VOUT
SW
4.7
µF
LM43603Q
ENABLE
0.47 µF
VCC
SS/TRK
2.2 µF
68 µF
100 kΩ
9.09 kΩ
RT
BIAS
SYNC
PGOOD
AGND
90
680 pF
80
Efficiency (%)
CBOOT
FB
70
60
PGND
50
24VIN
36VIN
40
0.001
0.01
0.1
1
Current (A)
VOUT = 12 V
VOUT = 12 V
FS = 500 kHz
C049
FS = 500 kHz
Figure 92. Efficiency
Figure 91. Component Values for VOUT = 12 V,
FS = 500 kHz
12.2
12.5
24VIN
12.4
12.0
36VIN
12.3
11.8
VOUT (V)
VOUT (V)
12.2
12.1
12.0
11.9
11.8
11.6
11.4
0.1A
0.5A
1A
1.5A
2A
2.5A
3A
11.2
11.7
11.0
11.6
11.5
0.001
0.01
0.1
Current (A)
VOUT = 12 V
10.8
12.0
1
12.5
13.0
13.5
14.0
VIN (V)
C050
VOUT = 12 V
FS = 500 kHz
14.5
C007
FS = 500 kHz
Figure 94. Dropout Curve
Figure 93. Output Voltage Regulation
3.5
3.0
IOUT (1A/DIV)
Curremt (A)
2.5
2.0
1.5
24VIN
1.0
VOUT (200 mV/DIV)
28VIN
0.5
36VIN
0.0
65
Time (100µs/DIV)
75
VOUT = 12 V
FS = 500 kHz
VIN = 24 V
85
95
105
115
Ambient Temperature (ƒC)
C001
VOUT = 12 V
Figure 95. Load Transient 0.1 A to 2 A
FS = 500 kHz
C050
RθJA = 20°C/W
Figure 96. Derating Curve
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9 Power Supply Recommendations
The LM43603-Q1 is designed to operate from an input voltage supply range between 3.5 V and 36 V. This input
supply must be well regulated and able to withstand maximum input current and maintain a stable voltage. The
resistance of the input supply rail must be low enough that an input current transient does not cause a high
enough drop at the LM43603-Q1 supply voltage that can cause a false UVLO fault triggering and system reset.
If the input supply is located more than a few inches from the LM43603-Q1 additional bulk capacitance may be
required in addition to the ceramic bypass capacitors. The amount of bulk capacitance is not critical, but a 47 µF
or 100 µF electrolytic capacitor is a typical choice.
10 Layout
The performance of any switching converter depends as much upon the layout of the PCB as the component
selection. The following guidelines will help users design a PCB with the best power conversion performance,
thermal performance, and minimized generation of unwanted EMI.
10.1 Layout Guidelines
1. Place ceramic high frequency bypass CIN as close as possible to the LM43603-Q1 VIN and PGND pins.
Grounding for both the input and output capacitors should consist of localized top side planes that connect to
the PGND pins and PAD.
2. Place bypass capacitors for VCC and BIAS close to the pins and ground the bypass capacitors to device
ground.
3. Minimize trace length to the FB pin net. Locate both feedback resistors, RFBT and RFBB close to the FB pin.
Place CFF directly in parallel with RFBT. If VOUT accuracy at the load is important, make sure VOUT sense is
made at the load. Route VOUT sense path away from noisy nodes and preferably through a layer on the other
side of a shieldig layer.
4. Use ground plane in one of the middle layers as noise shielding and heat dissipation path.
5. Have a single point ground connection to the plane. Route the ground connections for the feedback, softstart, and enable components to the ground plane. This prevents any switched or load currents from flowing
in the analog ground traces. If not properly handled, poor grounding can result in degraded load regulation or
erratic output voltage ripple behavior.
6. Make VIN, VOUT and ground bus connections as wide as possible. This reduces any voltage drops on the
input or output paths of the converter and maximizes efficiency.
7. Provide adequate device heat sinking. Use an array of heat-sinking vias to connect the exposed pad to the
ground plane on the bottom PCB layer. If the PCB has multiple copper layers, these thermal vias can also be
connected to inner layer heat-spreading ground planes. Ensure enough copper area is used for heat-sinking
to keep the junction temperature below 125°C.
10.1.1 Compact Layout for EMI Reduction
Radiated EMI is generated by the high di/dt components in pulsing currents in switching converters. The larger
area covered by the path of a pulsing current, the more EMI is generated. The key to minimize radiated EMI is to
identify pulsing current path and minimize the area of the path. In Buck converters,the pulsing current path is
from the VIN side of the input capacitors to HS switch, to the LS switch, and then return to the ground of the input
capacitors, as shown in Figure 97.
BUCK
CONVERTER
VIN
VIN
SW
L
CIN
VOUT
COUT
PGND
High di/dt
current
PGND
Figure 97. Buck Converter High Δi/Δt Path
40
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Layout Guidelines (continued)
High-frequency ceramic bypass capacitors at the input side provide primary path for the high di/dt components of
the pulsing current. Placing ceramic bypass capacitor(s) as close as possible to the VIN and PGND pins is the
key to EMI reduction.
The SW pin connecting to the inductor must be as short as possible, and just wide enough to carry the load
current without excessive heating. Use short, thick traces or copper pours (shapes) for high current condution
path to minimize parasitic resistance. The output capacitors must be place close to the VOUT end of the inductor
and closely grounded to PGND pin and exposed PAD.
Place the bypass capacitors on VCC and BIAS pins as close as possible to the pins respectively and closely
grounded to PGND and the exposed PAD.
10.1.2 Ground Plane and Thermal Considerations
TI recommends using one of the middle layers as a solid ground plane. Ground plane provides shielding for
sensitive circuits and traces. It also provides a quiet reference potential for the control circuitry. The AGND and
PGND pins must be connected to the ground plane using vias right next to the bypass capacitors. PGND pins
are connected to the source of the internal LS switch. They must be connected directly to the grounds of the
input and output capacitors. The PGND net contains noise at switching frequency and may bounce due to load
variations. Constrain PGND trace, as well as PVIN and SW traces, to one side of the ground plane. The other
side of the ground plane contains much less noise and should be used for sensitive routes.
TI recommends providing adequate device heat sinking by utilizing the PAD of the device as the primary thermal
path. Use a recommended 4 by 3 array of 10-mil thermal vias to connect the PAD to the system ground plane
heat sink. The vias must be evenly distributed under the PAD. Use as much copper as possible, for system
ground plane, on the top and bottom layers for the best heat dissipation. Use a four-layer board with the copper
thickness for the four layers, starting from the top of, 2 oz / 1 oz / 1 oz / 2 oz. Four layer boards with enough
copper thickness provides low current conduction impedance, proper shielding and lower thermal resistance.
The thermal characteristics of the LM43603-Q1 are specified using the parameter RθJA, which characterize the
junction temperature of silicon to the abient temperature in a specific system. Although the value of RθJA is
dependant on manhy variables, it still can be used to approximate the operating junction temperature of the
device. To obtain an estimate of the device junction temperature, one may use Equation 27:
TJ = PD x RθJA+ TA
where
•
•
•
•
•
TJ = Junction temperature in °C
PD = VIN x IIN × (1 - Efficiency) – 1.1 x IOUT × DCR
RθJA = junction-to-ambient thermal resistance of the device in °C/W
DCR = inductor DC parasitic resistance in Ω
TA = ambient temperature in °C
(27)
The maximum operating junction temperature of the LM43603-Q1 is 125°C. RθJA is highly related to PCB size
and layout, as well as enviromental factors such as heat sinking and air flow. Figure 98 shows measured results
of RθJA with different copper area on a 2-layer board and 4-layer board.
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Layout Guidelines (continued)
50.0
1W @ 0fpm - 2 layer
2W @ 0fpm - 2 layer
45.0
1W @ 0fpm - 4 layer
2W @ 0fpm - 4 layer
,JA (ƒC/W)
40.0
35.0
30.0
25.0
20.0
20mm x 20mm
30mm x 30mm
40mm x 40mm
Copper Area
50mm x 50mm
C007
Figure 98. RθJAvs Copper Area
2 oz Copper on Outer Layers and 1 oz Copper on Inner Layers
10.1.3 Feedback Resistors
To reduce noise sensitivity of the output voltage feedback path, it is important to place the resistor divider and
CFF close to the FB pin, rather than close to the load. The FB pin is the input to the error amplifier, so it is a high
impedance node and very sensitive to noise. Placing the resistor divider and CFF closer to the FB pin reduces the
trace length of FB signal and reduces noise coupling. The output node is a low impedance node, so the trace
from VOUT to the resistor divider can be long if short path is not available.
If voltage accuracy at the load is important, make sure voltage sense is made at the load. Doing so corrects for
voltage drops along the traces and provide the best output accuracy. The voltage sense trace from the load to
the feedback resistor divider must be routed away from the SW node path and the inductor to avoid
contaminating the feedback signal with switch noise, while also minimizing the trace length. This is most
important when high value resistors are used to set the output voltage. TI recommends routing the voltage sense
trace and place the resistor divider on a different layer than the inductor and SW node path, such that there is a
ground plane in between the feedback trace and inductor/SW node polygon. This provides further shielding for
the voltage feedback path from EMI noises.
42
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10.2 Layout Example
TO LOAD
+
VOUT sense point
is away from
inductor and
past COUT
VOUT
VOUT distribution
point is away
from inductor
and past COUT
COUT
As much copper area as possible, for
better thermal performance
L
GND
Place
bypass caps
close to
terminals
CVCC
Ground
bypass caps
to DAP
1
SW
2
CBOOT
3
14
VIN
VCC
4
13
VIN
BIAS
5
12
EN
SYNC
6
11
SS/TRK
RT
7
10
AGND
PGOOD
8
9
Thermal Vias under DAP
PAD
(17)
CBIAS
16
PGND
15
PGND
+
CBOOT
SW
CIN
VIN
Place ceramic
bypass caps close
to VIN and PGND
terminals
RFBB
FB
RFBT
CFF
Route VOUT
sense trace
away from SW
and VIN
nodes.
Preferably
shielded in an
alternative
layer
GND Plane
As much copper area as possible, for better thermal performance
Figure 99. LM43603-Q1 Board Layout Recommendations
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11 Device and Documentation Support
11.1 Device Support
11.1.1 Third-Party Products Disclaimer
TI'S PUBLICATION OF INFORMATION REGARDING THIRD-PARTY PRODUCTS OR SERVICES DOES NOT
CONSTITUTE AN ENDORSEMENT REGARDING THE SUITABILITY OF SUCH PRODUCTS OR SERVICES
OR A WARRANTY, REPRESENTATION OR ENDORSEMENT OF SUCH PRODUCTS OR SERVICES, EITHER
ALONE OR IN COMBINATION WITH ANY TI PRODUCT OR SERVICE.
11.2 Development Support
11.2.1 Custom Design With WEBENCH® Tools
Click here to create a custom design using the LM43603-Q1 device with the WEBENCH® Power Designer.
1. Start by entering the input voltage (VIN), output voltage (VOUT), and output current (IOUT) requirements.
2. Optimize the design for key parameters such as efficiency, footprint, and cost using the optimizer dial.
3. Compare the generated design with other possible solutions from Texas Instruments.
The WEBENCH Power Designer provides a customized schematic along with a list of materials with real-time
pricing and component availability.
In most cases, these actions are available:
• Run electrical simulations to see important waveforms and circuit performance
• Run thermal simulations to understand board thermal performance
• Export customized schematic and layout into popular CAD formats
• Print PDF reports for the design, and share the design with colleagues
Get more information about WEBENCH tools at www.ti.com/WEBENCH.
11.3 Receiving Notification of Documentation Updates
To receive notification of documentation updates, navigate to the device product folder on ti.com. In the upper
right corner, click on Alert me to register and receive a weekly digest of any product information that has
changed. For change details, review the revision history included in any revised document.
11.4 Community Resources
The following links connect to TI community resources. Linked contents are provided "AS IS" by the respective
contributors. They do not constitute TI specifications and do not necessarily reflect TI's views; see TI's Terms of
Use.
TI E2E™ Online Community TI's Engineer-to-Engineer (E2E) Community. Created to foster collaboration
among engineers. At e2e.ti.com, you can ask questions, share knowledge, explore ideas and help
solve problems with fellow engineers.
Design Support TI's Design Support Quickly find helpful E2E forums along with design support tools and
contact information for technical support.
11.5 Electrostatic Discharge Caution
These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
11.6 Glossary
SLYZ022 — TI Glossary.
This glossary lists and explains terms, acronyms, and definitions.
44
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12 Mechanical, Packaging, and Orderable Information
The following pages include mechanical packaging and orderable information. This information is the most
current data available for the designated devices. This data is subject to change without notice and revision of
this document. For browser-based versions of this data sheet, refer to the left-hand navigation.
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PACKAGE OPTION ADDENDUM
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10-Dec-2020
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
(4/5)
(6)
LM43603AQPWPRQ1
ACTIVE
HTSSOP
PWP
16
2000
RoHS & Green
NIPDAU
Level-3-260C-168 HR
-40 to 125
43603AQ
LM43603AQPWPTQ1
ACTIVE
HTSSOP
PWP
16
250
RoHS & Green
NIPDAU
Level-3-260C-168 HR
-40 to 125
43603AQ
LM43603QPWPRQ1
NRND
HTSSOP
PWP
16
2000
RoHS & Green
NIPDAU
Level-3-260C-168 HR
-40 to 125
43603Q1
LM43603QPWPTQ1
NRND
HTSSOP
PWP
16
250
RoHS & Green
NIPDAU
Level-3-260C-168 HR
-40 to 125
43603Q1
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of
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