LMR23615
LMR23615
SNVSAV8B – JUNE 2017 – REVISED AUGUST
2020
SNVSAV8B – JUNE 2017 – REVISED AUGUST 2020
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LMR23615 SIMPLE SWITCHER® 36-V, 1.5-A Synchronous Step-Down Converter
1 Features
3 Description
•
•
•
•
•
•
•
•
•
•
•
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•
•
The LMR23615 SIMPLE SWITCHER® is an easy-touse 36-V, 1.5-A synchronous step-down regulator.
With a wide input range from 4 V to 36 V, the device is
suitable for various industrial applications for power
conditioning from unregulated sources. Peak current
mode control is employed to achieve simple controlloop compensation and cycle-by-cycle current limiting.
A quiescent current of 75 µA makes the device
suitable for battery-powered systems. An ultra-low 2µA shutdown current can further prolong battery life.
Internal loop compensation means that the user is
free from the tedious task of loop-compensation
design and also minimizes the external components
needed. An extended family is available in 2.5-A
(LMR23625) and 3-A (LMR23630) load-current
options in pin-to-pin compatible packages, allowing
simple, optimum PCB layout. A precision enable input
allows simplification of regulator control and system
power sequencing. Protection features include cycleby-cycle current limit, hiccup-mode short-circuit
protection, and thermal shutdown due to excessive
power dissipation.
•
•
4-V to 36-V input range
1.5-A Continuous output current
Integrated synchronous rectification
Current-mode control with internal compensation
Minimum switch ON-time: 60 ns
Adjustable switching frequency
PFM mode at light load
Frequency synchronization to external clock
75-µA Quiescent current
Soft start into a prebiased load
High-duty-cycle operation supported
Output short-circuit protection with hiccup mode
Thermal protection
12-Pin WSON wettable flanks package with
PowerPAD™
Use the LMZM22602 module for faster time to
market
Create a custom design using the LMR23615 with
the WEBENCH® Power Designer
2 Applications
•
•
•
Device Information
Factory and building automation systems: PLC
CPU, HVAC control, elevator control
Asset tracking
General purpose wide VIN regulation
PART NUMBER (1)
PACKAGE
LMR23615
(1)
WSON (12)
BODY SIZE (NOM)
3.00 mm × 3.00 mm
For all available packages, see the orderable addendum at
the end of the data sheet.
space
space
space
100
VIN up to 36 V
90
CIN
VIN
CBOOT
AGND
L
VOUT
SW
RFBT
Efficiency (%)
EN/SYNC
BOOT
80
70
60
COUT
VCC
FB
RFBB
50
VOUT = 5 V
VOUT = 3.3 V
CVCC
PGND
Copyright © 2017, Texas Instruments Incorporated
Simplified Schematic
40
1E-5
0.0001
0.001
0.01
IOUT (A)
0.1
1
10
LMR2
Efficiency vs Load, VIN = 12 V
An IMPORTANT NOTICE at the end of this data sheet addresses availability, warranty, changes, use in safety-critical applications,
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Incorporated
intellectual
property
matters
and other important disclaimers. PRODUCTION DATA.
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Table of Contents
1 Features............................................................................1
2 Applications..................................................................... 1
3 Description.......................................................................1
4 Revision History.............................................................. 2
5 Pin Configuration and Functions...................................3
Pin Functions.................................................................... 3
6 Specifications.................................................................. 4
6.1 Absolute Maximum Ratings........................................ 4
6.2 ESD Ratings............................................................... 4
6.3 Recommended Operating Conditions.........................4
6.4 Thermal Information....................................................5
6.5 Electrical Characteristics.............................................5
6.6 Timing Characteristics.................................................6
6.7 Switching Characteristics............................................6
6.8 Typical Characteristics................................................ 7
7 Detailed Description........................................................9
7.1 Overview..................................................................... 9
7.2 Functional Block Diagram........................................... 9
7.3 Feature Description...................................................10
7.4 Device Functional Modes..........................................16
8 Application and Implementation.................................. 17
8.1 Application Information............................................. 17
8.2 Typical Applications.................................................. 17
9 Power Supply Recommendations................................23
10 Layout...........................................................................24
10.1 Layout Guidelines................................................... 24
10.2 Layout Example...................................................... 26
11 Device and Documentation Support..........................27
11.1 Device Support........................................................27
11.2 Receiving Notification of Documentation Updates.. 27
11.3 Support Resources................................................. 27
11.4 Trademarks............................................................. 27
11.5 Electrostatic Discharge Caution.............................. 27
11.6 Glossary.................................................................. 27
12 Mechanical, Packaging, and Orderable
Information.................................................................... 27
4 Revision History
NOTE: Page numbers for previous revisions may differ from page numbers in the current version.
Changes from Revision A (February 2018) to Revision B (July 2020)
Page
• Added LMZM33602 bullet to Section 1 ..............................................................................................................1
• Updated the numbering format for tables, figures and cross-references throughout the document...................1
Changes from Revision * (June 2017) to Revision A (February 2018)
Page
• First release of production-data data sheet; added WEBENCH content ........................................................... 1
• Changed Programmable Logic Controller Power Supply to Factory and Building Automation System... in
Applications ....................................................................................................................................................... 1
• Deleted Multi-Function Printers and Industrial Power Supplies and reworded Applications ............................. 1
• Changed HVAC Systems from Applications to General Purpose Wide VIN Regulation ................................... 1
• Changed the BOOT Capacitor value on Pin Functions to indicate value from 470nF to 100nF or higher......... 3
• Change the Abs Max Rating for EN/SYNC to AGND to VIN + 0.3 from 42V...................................................... 4
• Changed Typical Value for VIN_UVLO Rising threshold typical from 3.6-V to 3.7-V and minimum Falling
threshold from 3-V to 2.9-V ................................................................................................................................5
• Change Figure 7-8from VOUT = 5 V, fSW = 1600 kHz to VOUT = 5 V, fSW = 2100 kHz.......................................13
• Changed from VOUT = 7 V to 36 V to VIN = 7 V to 36 V on Figure 8-7 .............................................................22
2
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5 Pin Configuration and Functions
SW
1
12
PGND
SW
2
11
NC
BOOT
3
10
VIN
PAD
13
VCC
4
9
VIN
FB
5
8
EN/SYNC
RT
6
7
AGND
Figure 5-1. 12-Pin WSON DRR Package With Thermal Pad (Top View)
Pin Functions
PIN
NUMBER
NAME
I/O
DESCRIPTION
SW
P
Switching output of the regulator. Internally connected to both power MOSFETs. Connect to
power inductor.
3
BOOT
P
Boot-strap capacitor connection for high-side driver. Connect a high-quality 100nF to 470nF
capacitor from BOOT to SW.
4
VCC
P
Internal bias supply output for bypassing. Connect bypass capacitor from this pin to AGND. Do
not connect external loading to this pin. Never short this pin to ground during operation.
5
FB
A
Feedback input to regulator, connect the feedback resistor divider tap to this pin.
1, 2
6
RT
A
Connect a resistor RT from this pin to AGND to program switching frequency. Leave floating for
400-kHz default switching frequency.
7
AGND
G
Analog ground pin. Ground reference for internal references and logic. Connect to system
ground.
A
Enable input to regulator. High=On, Low=Off. Can be connected to VIN. Do not float. Adjust the
input under voltage lockout with two resistors. The internal oscillator can be synchronized to an
external clock by coupling a positive pulse into this pin through a small coupling capacitor. See
Section 7.3.4 for detail.
Input supply voltage.
8
EN/SYNC
9, 10
VIN
P
11
NC
N/A
12
PGND
G
Power ground pin, connected internally to the low side power FET. Connect to system ground,
PAD, AGND, ground pins of CIN and COUT. Path to CIN must be as short as possible.
13
PAD
G
Low impedance connection to AGND. Connect to PGND on PCB. Major heat dissipation path
of the die. Must be used for heat sinking to ground plane on PCB.
Not for use. Leave this pin floating.
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6 Specifications
6.1 Absolute Maximum Ratings
Over the recommended operating junction temperature range of –40°C to 125°C (unless otherwise noted)(1)
PARAMETER
MIN
MAX
UNIT
VIN to PGND
–0.3
42
EN/SYNC to AGND
–5.5
VIN + 0.3
FB to AGND
–0.3
4.5
RT to AGND
–0.3
4.5
AGND to PGND
–0.3
0.3
SW to PGND
–1
VIN + 0.3
SW to PGND less than 10-ns transients
–5
42
BOOT to SW
–0.3
5.5
VCC to AGND
–0.3
4.5(2)
Junction temperature, TJ
–40
150
°C
Storage temperature, Tstg
–65
150
°C
Input voltages
Output voltages
(1)
(2)
V
V
Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. These are stress ratings
only, which do not imply functional operation of the device at these or any other conditions beyond those indicated under
Recommended Operating Conditions. Exposure to absolute-maximum-rated conditions for extended periods may affect device
reliability.
In shutdown mode, the VCC to AGND maximum value is 5.25 V.
6.2 ESD Ratings
VALUE
V(ESD)
(1)
(2)
Electrostatic discharge
Human-body model
(HBM)(1)
UNIT
±2500
Charged-device model (CDM)(2)
V
±1000
JEDEC document JEP155 states that 500-V HBM allows safe manufacturing with a standard ESD control process.
JEDEC document JEP157 states that 250-V CDM allows safe manufacturing with a standard ESD control process.
6.3 Recommended Operating Conditions
Over the recommended operating junction temperature range of –40°C to 125°C (unless otherwise noted)
MIN
VIN
Input voltage
(1)
4
EN/SYNC
–5
FB
–0.3
36
36 V
1.2
Output voltage, VOUT
1
28 V
Output current, IOUT
0
1.5 A
Operating junction temperature, TJ
(1)
4
MAX UNIT
–40
125 °C
Recommended Operating Ratings indicate conditions for which the device is intended to be functional, but do not ensure specific
performance limits. For ensured specifications, see Section 6.5.
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6.4 Thermal Information
LMR23615
THERMAL
METRIC(1) (2)
DRR (WSON)
UNIT
(12 PINS)
RθJA
Junction-to-ambient thermal resistance
41.5
°C/W
ψJT
Junction-to-top characterization parameter
0.3
°C/W
ψJB
Junction-to-board characterization parameter
16.5
°C/W
RθJC(top)
Junction-to-case (top) thermal resistance
39.1
°C/W
RθJC(bot)
Junction-to-case (bottom) thermal resistance
3.4
°C/W
RθJB
Junction-to-board thermal resistance
16.3
°C/W
(1)
(2)
For more information about traditional and new thermal metrics, see the Semiconductor and IC Package Thermal Metrics application
report.
Determine power rating at a specific ambient temperature (TA) with a maximum junction temperature (TJ) of 125°C (see Section 6.3).
6.5 Electrical Characteristics
Limits apply over the recommended operating junction temperature (TJ) range of –40°C to +125°C, unless
otherwise stated. Minimum and Maximum limits are specified through test, design or statistical correlation.
Typical values represent the most likely parametric norm at TJ = 25°C, and are provided for reference purposes
only.
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX UNIT
POWER SUPPLY (VIN PIN)
VIN
Operation input voltage
4
36
Rising threshold
3.3
3.7
3.9
Falling threshold
2.9
3.3
3.5
2
4
VIN_UVLO
Undervoltage lockout thresholds
ISHDN
Shutdown supply current
VEN = 0 V, VIN = 12 V, TJ = –40 °C to 125°C
IQ
Operating quiescent current
(non- switching)
VIN =12 V, VFB = 1.2 V, TJ = –40 °C to 125°C,
PFM mode
75
V
V
μA
μA
ENABLE (EN/SYNC PIN)
VEN_H
Enable rising threshold voltage
VEN_HYS
Enable hysteresis voltage
VWAKE
Wake-up threshold
IEN
Input leakage current at EN pin
1.4
1.55
1.7
0.4
0.4
VIN = 4 V to 36 V, VEN= 2 V
V
V
V
10
VIN = 4 V to 36 V, VEN= 36 V
100
nA
1
μA
VOLTAGE REFERENCE (FB PIN)
VREF
Reference voltage
ILKG_FB
Input leakage current at FB pin
VIN = 4 V to 36 V, TJ = 25 °C
0.985
1
1.015
VIN = 4 V to 36 V, TJ = –40 °C to 125°C
0.980
1
1.020
VFB= 1 V
10
V
nA
INTERNAL LDO (VCC PIN)
VCC
VCC_UVLO
Internal LDO output voltage
VCC undervoltage lockout
thresholds
4.1
V
Rising threshold
2.8
3.2
3.6
Falling threshold
2.4
2.8
3.2
V
CURRENT LIMIT
IHS_LIMIT
Peak inductor current limit
2.9
3.9
4.9
A
ILS_LIMIT
Valley inductor current limit
1.9
2.5
3.2
A
IL_ZC
Zero cross current limit
–0.04
A
INTEGRATED MOSFETS
RDS_ON_HS
High-side MOSFET ONresistance
VIN = 12 V, IOUT = 1 A
160
mΩ
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Limits apply over the recommended operating junction temperature (TJ) range of –40°C to +125°C, unless
otherwise stated. Minimum and Maximum limits are specified through test, design or statistical correlation.
Typical values represent the most likely parametric norm at TJ = 25°C, and are provided for reference purposes
only.
PARAMETER
RDS_ON_LS
Low-side MOSFET ONresistance
TEST CONDITIONS
MIN
VIN = 12 V, IOUT = 1 A
TYP
MAX UNIT
95
mΩ
THERMAL SHUTDOWN
TSHDN
Thermal shutdown threshold
THYS
Hysteresis
162
170
178
°C
15
°C
6.6 Timing Characteristics
Over the recommended operating junction temperature range of –40°C to +125°C (unless otherwise noted)
MIN
NOM
MAX UNIT
HICCUP MODE
NOC (1)
Number of cycles that LS current limit is tripped to enter hiccup mode
64
Cycles
TOC
Hiccup retry delay time
10
ms
6
ms
SOFT START
Internal soft-start time. The time of internal reference to increase from 0 V
to 1 V
TSS
(1)
Specified by design.
6.7 Switching Characteristics
Over the recommended operating junction temperature range of –40°C to +125°C (unless otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX UNIT
SW (SW PIN)
TON_MIN
Minimum turnon time
60
TOFF_MIN (1)
Minimum turnoff time
100
90
ns
ns
SYNC (EN/SYNC PIN)
fSW_DEFAULT
FADJ
RT pin open circuit
340
Minimum adjustable frequency
RT = 198 kΩ with 1% accuracy
150
Maximum adjustable frequency
RT = 17.8 kΩ with 1% accuracy
1750
400
460
kHz
200
250
kHz
2150
2425
kHz
kHz
fSYNC
SYNC frequency range
200
2200
VSYNC
Amplitude of SYNC clock AC
signal (measured at SYNC pin)
2.8
5.5
TSYNC_MIN
Minimum sync clock ON and OFF
time
(1)
6
Oscillator default frequency
100
V
ns
Ensured by design.
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6.8 Typical Characteristics
100
100
90
90
80
80
70
70
Efficiency (%)
Efficiency (%)
Unless otherwise specified the following conditions apply: VIN = 12 V, fSW = 1600 kHz, L = 4.7 µH, COUT = 47 µF,
TA = 25 °C.
60
50
40
30
50
40
30
20
20
VIN = 12 V
VIN = 24 V
VIN = 36 V
10
0
1E-5
0.0001
0.001
0.01
IOUT (A)
0.1
1
VIN = 8 V
VIN = 12 V
VIN = 24 V
10
0
1E-5
10
0.0001
0.001
LMR2
fSW = 1000 kHz
VOUT = 5 V
Figure 6-1. Efficiency vs Load Current
100
90
90
80
80
70
70
50
40
0.1
1
30
LMR2
VOUT = 3.3 V
60
50
40
30
20
20
VIN = 8 V
VIN = 12 V
VIN = 24 V
10
0
1E-5
0.0001
0.001
0.01
IOUT (A)
0.1
1
VIN = 8 V
VIN = 12 V
VIN = 20 V
10
0
1E-5
10
0.0001
0.001
LMR2
fSW = 2200 kHz
VOUT = 5 V
Figure 6-3. Efficiency vs Load Current
0.01
IOUT (A)
0.1
1
10
LMR2
fSW = 2200 kHz (Sync)
VOUT = 3.3 V
Figure 6-4. Efficiency vs Load Current
5.1
5.12
VIN = 12 V
VIN = 24 V
VIN = 36 V
5.11
5.1
5.09
5.08
IOUT = 1.5 A
IOUT = 0.2 A
IOUT = 0 A
5.09
5.07
VOUT (V)
5.08
VOUT (V)
10
Figure 6-2. Efficiency vs Load Current
100
60
0.01
IOUT (A)
fSW = 1000 kHz
Efficiency (%)
Efficiency (%)
60
5.07
5.06
5.05
5.06
5.05
5.04
5.04
5.03
5.03
5.02
5.02
5.01
5.01
0
0.2
0.4
0.6
0.8
IOUT (A)
1
1.2
1.4
fSW = 1000 kHz
1.6
5
10
15
LMR2
VOUT = 5 V
Figure 6-5. Load Regulation
20
25
VIN (V)
30
35
fSW = 1000 kHz
40
LMR2
VOUT = 5 V
Figure 6-6. Line Regulation
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5.5
5.5
5.3
5.3
5.1
5.1
4.9
4.9
4.7
4.7
VOUT (V)
VOUT (V)
SNVSAV8B – JUNE 2017 – REVISED AUGUST 2020
4.5
4.3
4.1
4.5
4.3
4.1
IOUT = 0 A
IOUT = 0.2 A
IOUT = 0.8 A
IOUT = 1.5 A
3.9
3.7
IOUT = 0 A
IOUT = 0.2 A
IOUT = 0.8 A
IOUT = 1.5 A
3.9
3.7
3.5
3.5
4
4.5
5
VIN (V)
5.5
6
4
fSW = 1000 kHz
VOUT = 5 V
5
VIN (V)
5.5
6
LMR2
fSW = 2200 kHz
Figure 6-7. Dropout Curve
VOUT = 5 V
Figure 6-8. Dropout Curve
80
VIN UVLO Rising Threshold (V)
3.67
75
IQ (µA)
4.5
LMR2
70
65
60
-50
0
50
Temperature (°C)
100
150
3.66
3.65
3.64
3.63
3.62
3.61
-50
0
50
Temperature (°C)
100
150
D009
D008
VIN = 12 V
VFB = 1.1 V
Figure 6-10. VIN UVLO Rising Threshold vs
Junction Temperature
Figure 6-9. IQ vs Junction Temperature
0.425
4.5
LS Limit
HS Limit
Current Limit (A)
VIN UVLO Hysteresis (V)
4
0.42
3.5
3
2.5
0.415
2
-50
0.41
-50
0
50
Temperature (°C)
100
50
Temperature (qC)
100
150
LMR2
VIN = 12 V
150
D010
Figure 6-11. VIN UVLO Hysteresis vs Junction
Temperature
8
0
Figure 6-12. HS and LS Current Limit vs Junction
Temperature
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7 Detailed Description
7.1 Overview
The LMR23615 SIMPLE SWITCHER® regulator is an easy-to-use synchronous step-down DC-DC converter
operating from a 4-V to 36-V supply voltage. It is capable of delivering up to 1.5-A DC load current with good
thermal performance in a small solution size. An extended family is available in multiple current options from 1.5
A to 3 A in pin-to-pin compatible packages.
The LMR23615 employs constant frequency peak-current-mode control. The device enters PFM mode at light
load to achieve high efficiency. The device is internally compensated, which reduces design time, and requires
few external components. The switching frequency is adjustable from 200 kHz to 2.2 MHz, leaving the RT pin
open for 400-kHz default switching frequency. The LMR23615 is also capable of synchronization to an external
clock within the range of 200 kHz to 2.2 MHz.
Additional features such as precision enable and internal soft start provide a flexible and easy-to-use solution for
a wide range of applications. Protection features include thermal shutdown, VIN and VCC undervoltage lockout,
cycle-by-cycle current limit, and hiccup-mode short-circuit protection.
The LMR236xx family requires very few external components and has a pinout designed for simple, optimum
PCB layout.
7.2 Functional Block Diagram
EN/SYNC
SYNC Signal
VCC
SYNC
Detector
VCC
Enable
LDO
VIN
Precision
Enable
Internal
SS
CBOOT
HS I Sense
EA
REF
Rc
TSD
UVLO
Cc
PWM CONTROL LOGIC
PFM
Detector
OV/UV
Detector
SW
FB
Slope
Comp
Freq
Foldback
Zero
Cross
HICCUP
Detector
SYNC Signal
RT
LS I
Sense
Oscillator
FB
AGND
PGND
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7.3 Feature Description
7.3.1 Fixed-Frequency, Peak-Current-Mode Control
The following operating description of the LMR23615 refers to Section 7.2 and to the waveforms in Figure 7-1.
The LMR23615 device is a step-down, synchronous buck regulator with integrated high-side (HS) and low-side
(LS) switches (synchronous rectifier). The LMR23615 supplies a regulated output voltage by turning on the HS
and LS NMOS switches with controlled duty cycle. During high-side switch ON-time, the SW pin voltage swings
up to approximately VIN, and the inductor current IL increase with linear slope (VIN – VOUT) / L. When the HS
switch is turned off by the control logic, the LS switch is turned on after an anti-shoot-through dead time. Inductor
current discharges through the LS switch with a slope of –VOUT / L. The control parameter of a buck converter is
defined as duty cycle D = tON / TSW, where tON is the high-side switch ON time and TSW is the switching period.
The regulator control loop maintains a constant output voltage by adjusting the duty cycle D. In an ideal buck
converter, where losses are ignored, D is proportional to the output voltage and inversely proportional to the
input voltage: D = VOUT / VIN.
VSW
SW Voltage
D = tON/ TSW
VIN
tON
tOFF
t
0
-VD
Inductor Current
iL
TSW
ILPK
IOUT
'iL
t
0
Figure 7-1. SW Node and Inductor Current Waveforms in Continuous Conduction Mode (CCM)
The LMR23615 employs fixed-frequency peak-current-mode control. A voltage-feedback loop is used to get
accurate DC voltage regulation by adjusting the peak current command based on voltage offset. The peak
inductor current is sensed from the high-side switch and compared to the peak current threshold to control the
on-time of the high-side switch. The voltage feedback loop is internally compensated, which allows for fewer
external components, makes it easy to design, and provides stable operation with almost any combination of
output capacitors. The regulator operates with fixed switching frequency at normal load condition. At light load
condition, the LMR23615 operates in PFM mode to maintain high efficiency.
7.3.2 Adjustable Frequency
The switching frequency can be programmed by the resistor from the RT pin to ground. The frequency is
inversely proportional to the RT resistance. The RT pin can be left floating, and the LMR23615 operates at 400kHz default switching frequency. The RT pin is not designed to be shorted to ground. For a desired frequency,
typical RT resistance can be found by Equation 1. Table 7-1 gives typical RT values for a given switching
frequency (fSW ).
RT(kΩ) = 40200 / fSW(kHz) – 0.6
10
(1)
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250
RT Resistance (kŸ)
200
150
100
50
0
0
500
1000
1500
2000
Switching Frequency (kHz)
2500
C008
Figure 7-2. RT vs Frequency Curve
Table 7-1. Typical Frequency Setting RT Resistance
fSW (kHz)
RT (kΩ)
200
200
350
115
500
78.7
750
53.6
1000
39.2
1500
26.1
2000
19.6
2200
17.8
7.3.3 Adjustable Output Voltage
A precision 1-V reference voltage is used to maintain a tightly regulated output voltage over the entire operating
temperature range. The output voltage is set by a resistor divider from output voltage to the FB pin. TI
recommends using 1% tolerance resistors with a low temperature coefficient for the FB divider. Select the lowside resistor RFBB for the desired divider current and use Equation 2 to calculate high-side RFBT. RFBT in the
range from 10 kΩ to 100 kΩ is recommended for most applications. A lower RFBT value can be used if static
loading is desired to reduce VOUT offset in PFM operation. Lower RFBT reduces efficiency at very light load. Less
static current goes through a larger RFBT and might be more desirable when light load efficiency is critical.
However, RFBT larger than 1 MΩ is not recommended because it makes the feedback path more susceptible to
noise. Larger RFBT value requires more carefully designed feedback path on the PCB. The tolerance and
temperature variation of the resistor dividers affect the output voltage regulation.
VOUT
RFBT
FB
RFBB
Figure 7-3. Output Voltage Setting
RFBT
VOUT VREF
u RFBB
VREF
(2)
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7.3.4 Enable/Sync
The voltage on the EN pin controls the ON or OFF operation of LMR23615 device. A voltage less than 1 V
(typical) shuts down the device while a voltage higher than 1.6 V (typical) is required to start the regulator. The
EN pin is an input and cannot be left open or floating. The simplest way to enable the operation of the
LMR23615 is to connect the EN to VIN. This allows self-start-up of the LMR23615 when VIN is within the
operation range.
Many applications benefit from the employment of an enable divider RENT and RENB (Figure 7-4) to establish a
precision system UVLO level for the converter. System UVLO can be used for supplies operating from utility
power as well as battery power. It can be used for sequencing, ensuring reliable operation, or supply protection,
such as a battery discharge level. An external logic signal can also be used to drive EN input for system
sequencing and protection.
VIN
RENT
EN/SYNC
RENB
Figure 7-4. System UVLO by Enable Divider
The EN pin also can be used to synchronize the internal oscillator to an external clock. The internal oscillator can
be synchronized by AC coupling a positive edge into the EN pin. The AC coupled peak-to-peak voltage at the
EN pin must exceed the SYNC amplitude threshold of 2.8 V (typical) to trip the internal synchronization pulse
detector, and the minimum SYNC clock ON and OFF time must be longer than 100 ns (typical). A 3.3-V or a
higher amplitude pulse signal coupled through a 1-nF capacitor CSYNC is a good starting point. Keeping RENT //
RENB (RENT parallel with RENB) in the 100-kΩ range is a good choice. RENT is required for this synchronization
circuit, but RENB can be left unmounted if system UVLO is not needed. Switching action of the LMR23615 device
can be synchronized to an external clock from 200 kHz to 2.2 MHz. Figure 7-6 and Figure 7-7 show the device
synchronized to an external system clock.
VIN
CSYNC
RENT
EN/SYNC
RENB
Clock
Source
Figure 7-5. Synchronizing to External Clock
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Figure 7-6. Synchronizing in PWM Mode
Figure 7-7. Synchronizing in PFM Mode
7.3.5 VCC, UVLO
The LMR23615 integrates an internal LDO to generate VCC for control circuitry and MOSFET drivers. The
nominal voltage for VCC is 4.1 V. The VCC pin is the output of an LDO and must be properly bypassed. Place a
high-quality ceramic capacitor with a value of 2.2 µF to 10 µF, 16 V or higher rated voltage as close as possible
to VCC, grounded to the exposed PAD and ground pins. The VCC output pin must not be loaded, or shorted to
ground during operation. Shorting VCC to ground during operation may cause damage to the LMR23615 device.
VCC undervoltage lockout (UVLO) prevents the LMR23615 from operating until the VCC voltage exceeds 3.2 V
(typical). The VCC_UVLO threshold has 400 mV (typical) of hysteresis to prevent undesired shutdown due to
temporary VIN drops.
7.3.6 Minimum ON-Time, Minimum-OFF Time, and Frequency Foldback at Dropout Conditions
Minimum ON-time, TON_MIN, is the smallest duration of time that the HS switch can be on. TON_MIN is typically 60
ns in the LMR23615. Minimum OFF-time, TOFF_MIN, is the smallest duration that the HS switch can be off.
TOFF_MIN is typically 100 ns in the LMR23615. In CCM operation, TON_MIN and TOFF_MIN limit the voltage
conversion range given a selected switching frequency.
The minimum duty cycle allowed is:
DMIN = TON_MIN × fSW
(3)
And the maximum duty cycle allowed is:
DMAX = 1 – TOFF_MIN × fSW
(4)
Given fixed TON_MIN and TOFF_MIN, the higher the switching frequency the narrower the range of the allowed duty
cycle. In the LMR23615 device, a frequency foldback scheme is employed to extend the maximum duty cycle
when TOFF_MIN is reached. The switching frequency decreases once longer duty cycle is needed under low VIN
conditions. Wide range of frequency foldback allows the LMR23615 output voltage stay in regulation with a
much lower supply voltage VIN. This leads to a lower effective drop-out voltage.
Given an output voltage, the choice of the switching frequency affects the allowed input voltage range, solution
size, and efficiency. The maximum operation supply voltage can be found by:
VIN _ MAX
VOUT
fSW u TON _ MIN
(5)
At lower supply voltage, the switching frequency decreases once TOFF_MIN is tripped. The minimum VIN without
frequency foldback can be approximated by:
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VIN _ MIN
VOUT
1 fSW u TOFF _ MIN
(6)
Taking considerations of power losses in the system with heavy load operation, VIN_MAX is higher than the result
calculated in Equation 5. With frequency foldback, VIN_MIN is lowered by decreased fSW.
Switching Frequency (kHz)
2500
2000
1500
1000
500
0.5 A
1.0 A
1.5 A
0
5
5.5
6
6.5
7
Input Voltage (V)
7.5
8
LMR2
Figure 7-8. Frequency Foldback at Dropout (VOUT = 5 V, fSW = 2100 kHz)
7.3.7 Internal Compensation and CFF
The LMR23615 is internally compensated as shown in Section 7.2. The internal compensation is designed such
that the loop response is stable over the entire operating frequency and output voltage range. Depending on the
output voltage, the compensation loop phase margin can be low with all ceramic capacitors. An external
feedforward capacitor CFF is recommended to be placed in parallel with the top resistor divider RFBT for optimum
transient performance.
VOUT
RFBT
CFF
FB
RFBB
Figure 7-9. Feedforward Capacitor for Loop Compensation
The feedforward capacitor CFF in parallel with RFBT places an additional zero before the crossover frequency of
the control loop to boost phase margin. The zero frequency can be found by
fZ _ CFF
1
2S u CFF u RFBT
(7)
An additional pole is also introduced with CFF at the frequency of
fP _ CFF
14
1
2S u CFF u RFBT //RFBB
(8)
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The zero fZ_CFF adds phase boost at the crossover frequency and improves transient response. The pole fP-CFF
helps maintaining proper gain margin at frequency beyond the crossover. Table 8-1 lists the combination of
COUT, CFF and RFBT for typical applications, designs with similar COUT but RFBT other than recommended value,
adjust CFF such that (CFF × RFBT) is unchanged and adjust RFBB such that (RFBT / RFBB) is unchanged.
Designs with different combinations of output capacitors need different CFF. Different types of capacitors have
different equivalent series resistance (ESR). Ceramic capacitors have the smallest ESR and need the most CFF.
Electrolytic capacitors have much larger ESR than ceramic, and the ESR zero frequency location would be low
enough to boost the phase up around the crossover frequency. Designs that use mostly electrolytic capacitors at
the output may not need any CFF. The location of this ESR zero frequency can be calculated with Equation 9:
fZ _ESR
1
2S u COUT u ESR
(9)
The CFF creates a time constant with RFBT that couples in the attenuate output voltage ripple to the FB node. If
the CFF value is too large, it can couple too much ripple to the FB and affect VOUT regulation. Therefore,
calculate CFF based on output capacitors used in the system. At cold temperatures, the value of CFF might
change based on the tolerance of the chosen component. This may reduce its impedance and ease noise
coupling on the FB node. To avoid this, more capacitance can be added to the output or the value of CFF can be
reduced.
7.3.8 Bootstrap Voltage (BOOT)
The LMR23615 device provides an integrated bootstrap voltage regulator. A small capacitor between the BOOT
and SW pins provides the gate-drive voltage for the high-side MOSFET. The BOOT capacitor is refreshed when
the high-side MOSFET is off and the low-side switch conducts. The recommended value of the BOOT capacitor
is 0.1 μF to 0.47 μF . TI recommends a ceramic capacitor with an X7R or X5R grade dielectric with a voltage
rating of 16 V or higher for stable performance over temperature and voltage.
7.3.9 Overcurrent and Short-Circuit Protection
The LMR23615 is protected from overcurrent conditions by cycle-by-cycle current limit on both the peak and
valley of the inductor current. Hiccup mode is activated if a fault condition persists to prevent overheating.
High-side MOSFET overcurrent protection is implemented by the nature of the peak-current-mode control. The
HS switch current is sensed when the HS is turned on after a set blanking time. The HS switch current is
compared to the output of the error amplifier (EA) minus slope compensation every switching cycle. See Section
7.2 for more details. The peak current of HS switch is limited by a clamped maximum peak current threshold
IHS_LIMIT, which is constant. Thus the peak current limit of the high-side switch is not affected by the slope
compensation and remains constant over the full duty-cycle range.
The current going through LS MOSFET is also sensed and monitored. When the LS switch turns on, the inductor
current begins to ramp down. The LS switch does not turn OFF at the end of a switching cycle if its current is
above the LS current limit ILS_LIMIT. The LS switch is kept ON so that inductor current keeps ramping down, until
the inductor current ramps below the LS current limit ILS_LIMIT. Then the LS switch turns OFF, and the HS
switches on, after a dead time. This is somewhat different than the more typical peak-current limit and results in
Equation 10 for the maximum load current.
IOUT _ MAX
ILS _ LIMIT
VIN
VOUT
2 u fSW u L
u
VOUT
VIN
(10)
If the current of the LS switch is higher than the LS current limit for 64 consecutive cycles, hiccup-currentprotection mode is activated. In hiccup mode, the regulator is shut down and kept off for 5 ms, typically, before
the LMR23615 tries to start again. If an overcurrent or short-circuit fault condition still exist, hiccup repeats until
the fault condition is removed. Hiccup mode reduces power dissipation under severe overcurrent conditions,
prevents over-heating and potential damage to the device.
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7.3.10 Thermal Shutdown
The LMR23615 provides an internal thermal shutdown to protect the device when the junction temperature
exceeds 170°C (typical). The device is turned off when thermal shutdown activates. Once the die temperature
falls below 155°C (typical), the device reinitiates the power up sequence controlled by the internal soft-start
circuitry.
7.4 Device Functional Modes
7.4.1 Shutdown Mode
The EN pin provides electrical on- and off-control for the LMR23615. When VEN is below 1 V (typical), the device
is in shutdown mode. The LMR23615 also employs VIN and VCC UVLO protection. If VIN or VCC voltage is
below their respective UVLO level, the regulator is turned off.
7.4.2 Active Mode
The LMR23615 is in active mode when VEN is above the precision enable threshold, and VIN and VCC are above
their respective UVLO level. The simplest way to enable the LMR23615 is to connect the EN pin to VIN pin. This
allows self start-up when the input voltage is in the operating range: 4 V to 36 V. See Section 7.3.5 and Section
7.3.4 for details on setting these operating levels.
In active mode, depending on the load current, the LMR23615 will be in one of three modes:
1. Continuous conduction mode (CCM) with fixed switching frequency when load current is above half of the
peak-to-peak inductor current ripple.
2. Discontinuous conduction mode (DCM) with fixed switching frequency when load current is lower than half of
the peak-to-peak inductor current ripple in CCM operation.
3. Pulse frequency modulation mode (PFM) when switching frequency is decreased at very light load.
7.4.3 CCM Mode
CCM operation is employed in the LMR23615 device when the load current is higher than half of the peak-topeak inductor current. In CCM operation, the frequency of operation is fixed, output voltage ripple is at a
minimum in this mode, and the maximum output current of 1.5 A can be supplied by the device.
7.4.4 Light Load Operation
When the load current is lower than half of the peak-to-peak inductor current in CCM, the LMR23615 operate in
DCM , also known as diode emulation mode (DEM). In DCM, the LS switch is turned off when the inductor
current drops to IL_ZC (–40 mA typical). Both switching losses and conduction losses are reduced in DCM,
compared to forced PWM operation at light load.
At even lighter current loads, PFM is activated to maintain high efficiency operation. When either the minimum
HS switch ON time (tON_MIN ) or the minimum peak inductor current IPEAK_MIN (300 mA typical) is reached, the
switching frequency decreasse to maintain regulation. In PFM, switching frequency is decreased by the control
loop when load current reduces to maintain output voltage regulation. Switching loss is further reduced in PFM
operation due to less frequent switching actions. The external clock synchronizing is not valid when the
LMR23615 device enters into PFM mode.
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8 Application and Implementation
Note
Information in the following applications sections is not part of the TI component specification, and TI
does not warrant its accuracy or completeness. TI’s customers are responsible for determining
suitability of components for their purposes. Customers should validate and test their design
implementation to confirm system functionality.
8.1 Application Information
The LMR23615 is a step-down DC-to-DC regulator. It is typically used to convert a higher DC voltage to a lower
DC voltage with a maximum output current of 1.5 A. The following design procedure can be used to select
components for the LMR23615. Alternately, the WEBENCH® software may be used to generate complete
designs. When generating a design, the WEBENCH software utilizes iterative design procedure and accesses
comprehensive databases of components. See Section 8.2.2.1 and ti.com for more details.
8.2 Typical Applications
The LMR23615 only requires a few external components to convert from a wide voltage range supply to a fixed
output voltage. Figure 8-1 shows a basic schematic.
VIN 12 V
BOOT
VIN
CBOOT
0.1 F
L
4.7 H
CIN
10 F
VOUT
5 V/1.5 A
SW
EN/
SYNC
PAD
CFF
22 pF
FB
CVCC
2.2 F
RFBB
22.1 NŸ
VCC
RT
PGND
AGND
RFBT
88.7 NŸ
COUT
33 F
RT
24.3 NŸ
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Figure 8-1. Application Circuit
The external components must fulfill the needs of the application, but also the stability criteria of the device
control loop. Table 8-1 can be used to simplify the output filter component selection.
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Table 8-1. L, COUT, and CFF Typical Values
fSW (kHz)
VOUT (V)
L (µH) (1)
COUT (µF) (2)
CFF (pF)(4)
RFBT (kΩ)(3)
3.3
22
200
220
51
5
33
150
120
88.7
12
56
68
See note(5)
243
note(5)
510
200
400
1000
2200
(1)
(2)
(3)
(4)
(5)
24
56
33
3.3
10
120
See
100
51
5
15
90
68
88.7
12
33
47
See note(5)
243
24
33
22
See note(5)
510
3.3
4.7
68
47
51
5
5.6
47
22
88.7
12
10
33
See note(5)
243
3.3
2.2
33
22
51
5
3.3
22
15
88.7
Inductance value is calculated based on VIN = 36 V.
All the COUT values are after derating. Add more when using ceramic capacitors.
RFBT = 0 Ω for VOUT = 1 V. RFBB = 22.1 kΩ for all other VOUT settings.
For designs with RFBT other than recommended value, adjust CFF so that (CFF × RFBT) is unchanged and adjust RFBB such that (RFBT /
RFBB) is unchanged.
High ESR COUT gives enough phase boost and CFF not needed.
8.2.1 Design Requirements
Detailed design procedure is described based on a design example. For this design example, use the
parameters listed in Table 8-2 as the input parameters.
Table 8-2. Design Example Parameters
DESIGN PARAMETER
EXAMPLE VALUE
Input voltage, VIN
12 V typical, range from 8 V to 28 V
Output voltage, VOUT
5V
Maximum output current IO_MAX
1.5 A
Transient response 0.2 A to 1.5 A
5%
Output voltage ripple
50 mV
Input voltage ripple
400 mV
Switching frequency, fSW
1600 kHz
8.2.2 Detailed Design Procedure
8.2.2.1 Custom Design With WEBENCH® Tools
Click here to create a custom design using the LMR23615 device with the WEBENCH® Power Designer.
1. Start by entering the input voltage (VIN), output voltage (VOUT), and output current (IOUT) requirements.
2. Optimize the design for key parameters such as efficiency, footprint, and cost using the optimizer dial.
3. Compare the generated design with other possible solutions from Texas Instruments.
The WEBENCH Power Designer provides a customized schematic along with a list of materials with real-time
pricing and component availability.
In most cases, these actions are available:
• Run electrical simulations to see important waveforms and circuit performance
• Run thermal simulations to understand board thermal performance
• Export customized schematic and layout into popular CAD formats
• Print PDF reports for the design, and share the design with colleagues
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Get more information about WEBENCH tools at www.ti.com/WEBENCH.
8.2.2.2 Output Voltage Setpoint
The output voltage of LMR23615 is externally adjustable using a resistor divider network. The divider network is
comprised of top feedback resistor RFBT and bottom feedback resistor RFBB. Equation 11 is used to determine
the output voltage:
RFBT
VOUT VREF
u RFBB
VREF
(11)
For example, choosing the value of RFBB as 22.1 kΩ, the desired output voltage set to 5 V, and the VREF = 1 V,
the RFBB value is calculated using Equation 11. The formula yields to a value 88.7 kΩ.
8.2.2.3 Switching Frequency
The switching frequency can be adjusted by RT resistance from RT pin to ground. Use Equation 1 to calculate
the required value of RT. The device can also be synchronized to an external clock for a desired frequency. See
Section 7.3.4 for more details.
For 1600 kHz frequency, the calculated RT is 24.5 kΩ, and standard value 24.3 kΩ is selected to set the
frequency approximate to 1600 kHz.
8.2.2.4 Inductor Selection
The most critical parameters for the inductor are the inductance, saturation current, and the rated current. The
inductance is based on the desired peak-to-peak ripple current ΔiL. Because the ripple current increases with the
input voltage, the maximum input voltage is always used to calculate the minimum inductance LMIN. Use
Equation 13 to calculate the minimum value of the output inductor. KIND is a coefficient that represents the
amount of inductor ripple current relative to the maximum output current of the device. A reasonable value of
KIND would be 20% to 40%. During an instantaneous short or overcurrent operation event, the RMS and peak
inductor current can be high. The inductor current rating must be higher than the current limit of the device.
'iL
LMIN
VOUT u VIN _ MAX
VOUT
VIN _ MAX u L u fSW
VIN _ MAX
VOUT
IOUT u KIND
u
(12)
VOUT
VIN _ MAX u fSW
(13)
In general, it is preferable to choose lower inductance in switching power supplies, because lower inductance
usually corresponds to faster transient response, smaller DCR, and reduced size for more compact designs. But
inductance that is too low can generate an inductor current ripple that is too large such that overcurrent
protection at the full load could be falsely triggered. It also generates more conduction loss and inductor core
loss. Larger inductor current ripple also implies larger output voltage ripple with same output capacitors. With
peak-current-mode control, TI does not recommend having an inductor current ripple that is too small. A larger
peak-current ripple improves the comparator signal-to-noise ratio.
For this design example, choose KIND = 0.4, the minimum inductor value is calculated to be 4.3 µH. Choose the
nearest standard 4.7-μH ferrite inductor with a capability of 2-A RMS current and 4-A saturation current.
8.2.2.5 Output Capacitor Selection
Choose the output capacitor(s), COUT with care because it directly affects the steady-state output-voltage ripple,
loop stability, and the voltage over/undershoot during load current transients.
The output ripple is essentially composed of two parts. One is caused by the inductor current ripple going
through the ESR of the output capacitors:
'VOUT_ESR
'iL u ESR
KIND u IOUT u ESR
(14)
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The other is caused by the inductor current ripple charging and discharging the output capacitors:
'VOUT _ C
'iL
8 u fSW u COUT
KIND u IOUT
8 u fSW u COUT
(15)
where
•
KIND = Ripple ratio of the inductor ripple current (ΔiL / IOUT)
The two components in the voltage ripple are not in phase, so the actual peak-to-peak ripple is smaller than the
sum of two peaks.
Output capacitance is usually limited by transient performance specifications if the system requires tight voltage
regulation with presence of large current steps and fast slew rate. When a fast large load increase happens,
output capacitors provide the required charge before the inductor current can slew up to the appropriate level.
The control loop of the regulator usually needs four or more clock cycles to respond to the output voltage droop.
The output capacitance must be large enough to supply the current difference for four clock cycles to maintain
the output voltage within the specified range. Equation 16 shows the minimum output capacitance needed for
specified output undershoot. When a sudden large load decrease happens, the output capacitors absorb energy
stored in the inductor, which causes an output voltage overshoot. Equation 17 calculates the minimum
capacitance required to keep the voltage overshoot within a specified range.
COUT !
4 u IOH IOL
fSW u VUS
(16)
2
2
IOH
IOL
COUT !
VOUT
VOS
2
2
VOUT
uL
(17)
where
•
•
•
•
IOL = Low level output current during load transient
IOH = High level output current during load transient
VUS = Target output voltage undershoot
VOS = Target output voltage overshoot
For this design example, the target output ripple is 50 mV. Presuppose ΔVOUT_ESR = ΔVOUT_C = 50 mV, and
choose KIND = 0.4. Equation 14 yields ESR no larger than 83.3 mΩ, and Equation 15 yields COUT no smaller
than 0.9 μF. For the target over/undershoot range of this design, VUS = VOS = 5% × VOUT = 250 mV. The COUT
can be calculated to be no smaller than 14 μF and 4.1 μF by Equation 16 and Equation 17, respectively. Taking
into account the derating factor of ceramic capacitor over temperature and voltage, one 33-μF, 16-V ceramic
capacitor with 5-mΩ ESR is selected.
8.2.2.6 Feedforward Capacitor
The LMR23615 device is internally compensated. Depending on the VOUT and frequency fSW, if the output
capacitor COUT is dominated by low ESR (ceramic types) capacitors, it could result in low phase margin. To
improve the phase boost an external feedforward capacitor CFF can be added in parallel with RFBT. CFF is
chosen such that phase margin is boosted at the crossover frequency without CFF. A simple estimation for the
crossover frequency (fX) without CFF is shown in Equation 18, assuming COUT has very small ESR, and COUT
value is after derating.
fX
8.32
VOUT u COUT
(18)
Equation 19 for CFF was tested:
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1
4S u fX u RFBT
(19)
For designs with higher ESR, CFF is not needed when COUT has very high ESR, and CFF calculated from
Equation 19 should be reduced with medium ESR. Table 8-1 can be used as a quick starting point.
For the application in this design example, a 18-pF, 50-V, COG capacitor is selected.
8.2.2.7 Input Capacitor Selection
The LMR23615 device requires high-frequency input decoupling capacitor(s) and a bulk input capacitor,
depending on the application. The typical recommended value for the high-frequency decoupling capacitor is 4.7
μF to 10 μF. TI recommends a high-quality ceramic capacitor type X5R or X7R with sufficiency voltage rating. To
compensate the derating of ceramic capacitors, a voltage rating twice the maximum input voltage is
recommended. Additionally, some bulk capacitance can be required, especially if the LMR23615 circuit is not
located within approximately 5 cm from the input voltage source. This capacitor is used to provide damping to
the voltage spike due to the lead inductance of the cable or the trace. For this design, two 4.7-μF, 50-V, X7R
ceramic capacitors are used. A 0.1-μF for high-frequency filtering and place it as close as possible to the device
pins.
8.2.2.8 Bootstrap Capacitor Selection
Every LMR23615 design requires a bootstrap capacitor (CBOOT). The recommended capacitor is 0.1 μF and
rated 16 V or higher. The bootstrap capacitor is located between the SW pin and the BOOT pin. The bootstrap
capacitor must be a high-quality ceramic type with an X7R or X5R grade dielectric for temperature stability.
8.2.2.9 VCC Capacitor Selection
The VCC pin is the output of an internal LDO for the LMR23615 device. To insure stability of the device, place a
minimum of 2.2-μF, 16-V, X7R capacitor from this pin to ground.
8.2.2.10 Undervoltage Lockout Setpoint
The system undervoltage lockout (UVLO) is adjusted using the external voltage divider network of RENT and
RENB. The UVLO has two thresholds, one for power up when the input voltage is rising and one for power down
or brownouts when the input voltage is falling. Equation 20 can be used to determine the VIN UVLO level.
VIN _ RISING
VENH u
RENT RENB
RENB
(20)
The EN rising threshold (VENH) for LMR23615 is set to be 1.55 V (typical). Choose the value of RENB to be 287
kΩ to minimize input current from the supply. If the desired VIN UVLO level is at 6 V, then the value of RENT can
be calculated using Equation 21:
RENT
§ VIN _ RISING
¨¨
© VENH
·
1¸¸ u RENB
¹
(21)
Equation 21 yields a value of 820 kΩ. The resulting falling UVLO threshold, equals 4.4 V, can be calculated by
Equation 22, where EN hysteresis (VEN_HYS) is 0.4 V (typical).
VIN _ FALLING
VENH
VEN _ HYS u
RENT RENB
RENB
(22)
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8.2.3 Application Curves
Unless otherwise specified the following conditions apply: VIN = 12 V, fSW = 1600 kHz, L = 4.7 µH, COUT = 47 µF,
TA = 25 °C.
VOUT = 5 V
IOUT = 1.5 A
fSW = 1600 kHz
VOUT = 5 V
IOUT = 0 mA
Figure 8-2. CCM Mode
VIN = 12 V
VOUT = 5 V
Figure 8-3. PFM Mode
IOUT = 1.5 A
VIN = 12 V
Figure 8-4. Start-Up by VIN
VIN = 12 V
VIN = 7 V to 36 V,
2 V / μs
IOUT = 1.5 A
VOUT = 5 V
IOUT = 1.5 A
Figure 8-7. Line Transient
Figure 8-6. Load Transient
22
VOUT = 5 V
Figure 8-5. Start-Up by EN
VOUT = 5 V
IOUT = 0.2 A to 1.5 A, 100 mA / μs
fSW = 1600 kHz
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VOUT = 5 V
IOUT = 1 A to short
VOUT = 5 V
Figure 8-8. Short Protection
IOUT = short to 1 A
Figure 8-9. Short Recovery
9 Power Supply Recommendations
The LMR23615 is designed to operate from an input voltage supply range between 4 V and 36 V. This input
supply must be able to withstand the maximum input current and maintain a stable voltage. The resistance of the
input supply rail must be low enough that an input current transient does not cause a high enough drop at the
LMR23615 supply voltage that can cause a false UVLO fault triggering and system reset. If the input supply is
located more than a few inches from the LMR23615, additional bulk capacitance may be required in addition to
the ceramic input capacitors. The amount of bulk capacitance is not critical, but a 47-μF or 100-μF electrolytic
capacitor is a typical choice.
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10 Layout
10.1 Layout Guidelines
Layout is a critical portion of good power supply design. The following guidelines will help users design a PCB
with the best power-conversion performance, thermal performance, and minimized generation of unwanted EMI.
1. The input bypass capacitor CIN must be placed as close as possible to the VIN and PGND pins. Grounding
for both the input and output capacitors should consist of localized top side planes that connect to the PGND
pin and PAD.
2. Place bypass capacitors for VCC close to the VCC pin and ground the bypass capacitor to device ground.
3. Minimize trace length to the FB pin net. Both feedback resistors, RFBT and RFBB must be located close to the
FB pin. Place CFF directly in parallel with RFBT. If VOUT accuracy at the load is important, ensure that the VOUT
sense is made at the load. Route VOUT sense path away from noisy nodes and preferably through a layer on
the other side of a shielded layer.
4. Use ground plane in one of the middle layers as noise shielding and heat dissipation path.
5. Have a single point ground connection to the plane. Route the ground connections for the feedback and
enable components to the ground plane. This prevents any switched or load currents from flowing in the
analog ground traces. If not properly handled, poor grounding can result in degraded load regulation or erratic
output voltage ripple behavior.
6. Make VIN, VOUT and ground bus connections as wide as possible. This reduces any voltage drops on the
input or output paths of the converter and maximizes efficiency.
7. Provide adequate device heat sinking. Use an array of heat-sinking vias to connect the exposed pad to the
ground plane on the bottom PCB layer. If the PCB has multiple copper layers, these thermal vias can also be
connected to inner layer heat-spreading ground planes. Ensure enough copper area is used for heat sinking
to keep the junction temperature below 125°C.
10.1.1 Compact Layout for EMI Reduction
Radiated EMI is generated by the high di/dt components in pulsing currents in switching converters. The larger
area covered by the path of a pulsing current, the more EMI is generated. High frequency ceramic bypass
capacitors at the input side provide primary path for the high di/dt components of the pulsing current. Placing
ceramic bypass capacitor(s) as close as possible to the VIN and PGND pins is the key to EMI reduction.
The SW pin connecting to the inductor must be as short as possible, and just wide enough to carry the load
current without excessive heating. Use short, thick traces or copper pours (shapes) for high-current conduction
path to minimize parasitic resistance. The output capacitors must be placed close to the VOUT end of the inductor
and closely grounded to PGND pin and exposed PAD.
Place the bypass capacitors on VCC as close as possible to the pin and closely grounded to PGND and the
exposed PAD.
10.1.2 Ground Plane and Thermal Considerations
TI recommends using one of the middle layers as a solid ground plane. Ground plane provides shielding for
sensitive circuits and traces. It also provides a quiet reference potential for the control circuitry. Connect the
AGND and PGND pins to the ground plane using vias right next to the bypass capacitors. PGND pin is
connected to the source of the internal LS switch. They must be connected directly to the grounds of the input
and output capacitors. The PGND net contains noise at switching frequency and may bounce due to load
variations. PGND trace, as well as VIN and SW traces, must be constrained to one side of the ground plane. The
other side of the ground plane contains much less noise and should be used for sensitive routes.
TI recommends providing adequate device heat sinking by utilizing the PAD of the device as the primary thermal
path. Use a minimum 4 by 2 array of 12 mil thermal vias to connect the PAD to the system ground plane heat
sink. The vias should be evenly distributed under the PAD. Use as much copper as possible, for system ground
plane, on the top and bottom layers for the best heat dissipation. Use a four-layer board with the copper
thickness for the four layers, starting from the top of, 2 oz / 1 oz / 1 oz / 2 oz. Four-layer boards with enough
copper thickness provides low current conduction impedance, proper shielding, and lower thermal resistance.
24
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The thermal characteristics of the LMR23615 are specified using the parameter RθJA, which characterize the
junction temperature of silicon to the ambient temperature in a specific system. Although the value of RθJA is
dependent on many variables, it still can be used to approximate the operating junction temperature of the
device. To obtain an estimate of the device junction temperature, one may use the following relationship:
TJ = PD × RθJA + TA
(23)
PD = VIN x IIN × (1 – Efficiency) – 1.1 × IOUT 2 × DCR in watt
(24)
where
•
•
•
•
•
TJ = junction temperature in °C
PD = device power dissipation in watt
RθJA = junction-to-ambient thermal resistance of the device in °C/W
TA = ambient temperature in °C
DCR = inductor DC parasitic resistance in ohm
The recommended operating junction temperature of the LMR23615 is 125°C. RθJA is highly related to PCB size
and layout, as well as environmental factors such as heat sinking and air flow.
10.1.3 Feedback Resistors
To reduce noise sensitivity of the output voltage feedback path, it is important to place the resistor divider and
CFF close to the FB pin, rather than close to the load. The FB pin is the input to the error amplifier, so it is a high
impedance node and very sensitive to noise. Placing the resistor divider and CFF closer to the FB pin reduces
the trace length of FB signal and reduces noise coupling. The output node is a low impedance node, so the trace
from VOUT to the resistor divider can be long if short path is not available.
If voltage accuracy at the load is important, make sure voltage sense is made at the load. Doing so corrects for
voltage drops along the traces and provide the best output accuracy. Route the voltage sense trace from the
load to the feedback resistor divider away from the SW node path and the inductor to avoid contaminating the
feedback signal with switch noise, while also minimizing the trace length. This is most important when high-value
resistors are used to set the output voltage. TI recommends routing the voltage sense trace and place the
resistor divider on a different layer than the inductor and SW node path, such that there is a ground plane in
between the feedback trace and inductor/SW node polygon. This provides further shielding for the voltage
feedback path from EMI noises.
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10.2 Layout Example
Output
Inductor
Output Bypass
Capacitor
BOOT
Capacitor
VCC
Capacitor
SW
PGND
SW
NC
BOOT
VIN
VCC
VIN
FB
EN/SYNC
RT
AGND
Input Bypass
Capacitor
UVLO Adjust
Resistor
RT
Thermal VIA
Output Voltage
Set Resistor
VIA (Connect to GND Plane)
Figure 10-1. LMR23615 Layout
26
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11 Device and Documentation Support
11.1 Device Support
11.1.1 Development Support
11.1.1.1 Custom Design With WEBENCH® Tools
Click here to create a custom design using the LMR23615 device with the WEBENCH® Power Designer.
1. Start by entering the input voltage (VIN), output voltage (VOUT), and output current (IOUT) requirements.
2. Optimize the design for key parameters such as efficiency, footprint, and cost using the optimizer dial.
3. Compare the generated design with other possible solutions from Texas Instruments.
The WEBENCH Power Designer provides a customized schematic along with a list of materials with real-time
pricing and component availability.
In most cases, these actions are available:
• Run electrical simulations to see important waveforms and circuit performance
• Run thermal simulations to understand board thermal performance
• Export customized schematic and layout into popular CAD formats
• Print PDF reports for the design, and share the design with colleagues
Get more information about WEBENCH tools at www.ti.com/WEBENCH.
11.2 Receiving Notification of Documentation Updates
To receive notification of documentation updates, navigate to the device product folder on ti.com. Click on
Subscribe to updates to register and receive a weekly digest of any product information that has changed. For
change details, review the revision history included in any revised document.
11.3 Support Resources
TI E2E™ support forums are an engineer's go-to source for fast, verified answers and design help — straight
from the experts. Search existing answers or ask your own question to get the quick design help you need.
Linked content is provided "AS IS" by the respective contributors. They do not constitute TI specifications and do
not necessarily reflect TI's views; see TI's Terms of Use.
11.4 Trademarks
PowerPAD™ is a trademark of TI.
TI E2E™ is a trademark of Texas Instruments.
WEBENCH® is a registered trademark of Texas Instruments.
SIMPLE SWITCHER® and are registered trademarks of TI.
All other trademarks are the property of their respective owners.
11.5 Electrostatic Discharge Caution
This integrated circuit can be damaged by ESD. Texas Instruments recommends that all integrated circuits be handled
with appropriate precautions. Failure to observe proper handling and installation procedures can cause damage.
ESD damage can range from subtle performance degradation to complete device failure. Precision integrated circuits may
be more susceptible to damage because very small parametric changes could cause the device not to meet its published
specifications.
11.6 Glossary
TI Glossary
This glossary lists and explains terms, acronyms, and definitions.
12 Mechanical, Packaging, and Orderable Information
The following pages include mechanical, packaging, and orderable information. This information is the most
current data available for the designated devices. This data is subject to change without notice and revision of
this document. For browser-based versions of this data sheet, refer to the left-hand navigation.
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PACKAGE OPTION ADDENDUM
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25-Jan-2021
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
(4/5)
(6)
LMR23615DRRR
ACTIVE
WSON
DRR
12
3000
RoHS & Green
SN
Level-2-260C-1 YEAR
-40 to 125
23615
LMR23615DRRT
ACTIVE
WSON
DRR
12
250
RoHS & Green
SN
Level-2-260C-1 YEAR
-40 to 125
23615
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of
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