OPA211-HT
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SBOS481D – JULY 2009 – REVISED OCTOBER 2013
1.1 nV/√Hz Noise, Low Power, Precision Operational Amplifier
Check for Samples: OPA211-HT
FEATURES
1
•
•
2
•
•
•
•
•
•
•
•
•
•
•
Low Voltage Noise: 1.1 nV/√Hz at 1 kHz
Input Voltage Noise:
80 nVPP (0.1 Hz to 10 Hz)
THD+N: –136dB (G = 1, f = 1 kHz)
Offset Voltage: 240 μV (max)
Offset Voltage Drift: 0.35 μV/°C (typ)
Low Supply Current: 6 mA/Ch (typ)
Unity-Gain Stable
Gain Bandwidth Product:
80 MHz (G = 100)
45 MHz (G = 1)
Slew Rate: 27 V/μs
16-Bit Settling: 700 ns
Wide Supply Range:
±2.25 V to ±18 V, 4.5 V to 36 V
Rail-to-rail output
Output current: 30 mA
SUPPORTS EXTREME TEMPERATURE
APPLICATIONS
•
•
•
•
•
•
•
•
Controlled Baseline
One Assembly/Test Site
One Fabrication Site
Available in Extreme (–55°C/210°C)
Temperature Range (1)
Extended Product Life Cycle
Extended Product-Change Notification
Product Traceability
Texas Instruments high temperature products
utilize highly optimized silicon (die) solutions
with design and process enhancements to
maximize performance over extended
temperatures.
HKJ PACKAGE
(TOP VIEW)
NC
-IN
+IN
V-
APPLICATIONS
•
•
Down-Hole Drilling
High Temperature Environments
1
8
2
7
3
6
4
5
NC
V+
OUT
NC
NC denotes no internal connection
HKQ PACKAGE
(TOP VIEW)
NC
8
1
NC
V+
-IN
OUT
+IN
NC
5
4
V-
HKQ as formed or HKJ mounted dead bug
(1)
Custom temperature ranges available
DESCRIPTION
The OPA211 series of precision operational amplifiers achieves very low 1.1 nV/√Hz noise density with a supply
current of only 3.6 mA. This series also offers rail-to-rail output swing, which maximizes dynamic range.
The extremely low voltage and low current noise, high speed, and wide output swing of the OPA211 series make
these devices an excellent choice as a loop filter amplifier in PLL applications.
1
2
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
All trademarks are the property of their respective owners.
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of the Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
Copyright © 2009–2013, Texas Instruments Incorporated
OPA211-HT
SBOS481D – JULY 2009 – REVISED OCTOBER 2013
www.ti.com
In precision data acquisition applications, the OPA211 series of op amps provides 700-ns settling time to 16-bit
accuracy throughout 10-V output swings. This ac performance, combined with only 240-μV of offset and 0.35μV/°C of drift over temperature, makes the OPA211 ideal for driving high-precision 16-bit analog-to-digital
converters (ADCs) or buffering the output of high-resolution digital-to-analog converters (DACs).
The OPA211 series is specified over a wide dual-power supply range of ±2.25 V to ±18 V, or for single-supply
operation from 4.5 V to 36 V.
This series of op amps is specified from TA = –55°C to 210°C.
INPUT VOLTAGE NOISE DENSITY vs FREQUENCY
Voltage Noise Density (nV/ÖHz)
100
10
1
0.1
1
10
100
1k
10k
100k
Frequency (Hz)
Table 1. ORDERING INFORMATION (1)
TA
–55°C to 210°C
(1)
PACKAGE
ORDERABLE PART NUMBER
TOP-SIDE MARKING
HKJ
OPA211SHKJ
OPA211SHKJ
HKQ
OPA211SHKQ
OPA211SHKQ
KGD
OPA211SKGD1
NA
For the most current package and ordering information, see the Package Option Addendum at the end of this document, or see the TI
Web site at www.ti.com.
This integrated circuit can be damaged by ESD. Texas Instruments recommends that all integrated circuits be handled with
appropriate precautions. Failure to observe proper handling and installation procedures can cause damage.
ESD damage can range from subtle performance degradation to complete device failure. Precision integrated circuits may be more
susceptible to damage because very small parametric changes could cause the device not to meet its published specifications.
2
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SBOS481D – JULY 2009 – REVISED OCTOBER 2013
BARE DIE INFORMATION
DIE THICKNESS
BACKSIDE FINISH
BACKSIDE
POTENTIAL
BOND PAD
METALLIZATION COMPOSITION
15 mils.
Silicon with backgrind
V-
Al-Si-Cu (0.5%)
Origin
a
c
b
d
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Table 2. BOND PAD COORDINATES
4
DESCRIPTION
PAD NUMBER
a
b
c
d
-IN
1
34.4000
792.000
109.400
867.000
+IN
2
34.4000
33.000
109.400
108.000
NC
3
461.850
33.000
536.850
108.000
V-
4
692.650
54.600
767.650
129.600
OUT
5
920.400
33.000
995.400
108.000
V+
6
920.400
720.150
995.400
795.150
NC
7
388.050
792.000
463.050
867.000
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SBOS481D – JULY 2009 – REVISED OCTOBER 2013
ABSOLUTE MAXIMUM RATINGS (1)
Over operating free-air temperature range (unless otherwise noted).
VS = (V=) – (V-)
Supply Voltage
VIN
Input Voltage
IIN
Input Current (Any pin except power-supply pins)
VALUE
UNIT
40
V
(V–) – 0.5 to (V+) + 0.5
V
±10
mA
Output Short-Circuit (2)
Continuous
TA
Operating Temperature
–55 to 210
°C
TSTG
Storage Temperature
–65 to 210
°C
TJ
Junction Temperature
210
°C
Human Body Model (HBM)
3000
V
Charged Device Model
(CDM)
1000
V
ESD Ratings
(1)
(2)
Stresses above these ratings may cause permanent damage. Exposure to absolute maximum conditions for extended periods may
degrade device reliability. These are stress ratings only, and functional operation of the device at these or any other conditions beyond
those specified is not supported.
Short-circuit to VS/2 (ground in symmetrical dual supply setups), one amplifier per package.
THERMAL CHARACTERISTICS FOR HKJ OR HKQ PACKAGE
over operating free-air temperature range (unless otherwise noted)
PARAMETER
θJC
Junction-to-case thermal resistance
to ceramic side of case
to top of case lid (metal side of case)
MIN
TYP
MAX
5.7
13.7
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UNIT
°C/W
5
OPA211-HT
SBOS481D – JULY 2009 – REVISED OCTOBER 2013
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ELECTRICAL CHARACTERISTICS: VS = ±2.25 V to ±18 V
At TA = 25°C, RL = 10 kΩ connected to midsupply, VCM = VOUT = midsupply, unless otherwise noted.
TA = –55 to 125°C
PARAMETER
CONDITIONS
MIN
TA = 210°C
TYP
MAX
MIN
TYP
MAX
UNIT
OFFSET VOLTAGE
Input Offset Voltage
Drift
VOS
±30
±180
±70
±260
μV
0.35
1.5
0.35
2.0
μV/°C
VS = ±2.25V to ±18V
0.1
3
0.1
3
μV/V
VS = ±15V
dVOS/dT
vs Power Supply
PSRR
INPUT BIAS CURRENT
Input Bias Current
Offset Current
IB
VCM = 0V
±60
±200
±60
±250
nA
IOS
VCM = 0V
±25
±150
±25
±150
nA
en
f = 0.1Hz to 10Hz
80
80
nVPP
f = 10Hz
2
2
nV/√Hz
f = 100Hz
1.4
1.4
nV/√Hz
f = 1kHz
1.1
1.1
nV/√Hz
f = 10Hz
3.2
3.2
pA/√Hz
f = 1kHz
1.7
1.7
pA/√Hz
NOISE
Input Voltage Noise
Input Voltage Noise Density
Input Current Noise
Density
In
INPUT VOLTAGE RANGE (1)
Common-Mode Voltage
Range
Common-Mode Rejection
Ratio
VCM
CMRR
VS ≥ ±5V
(V–) + 1.8
(V+) – 1.4
(V–) + 1.8
(V+) – 1.4
(V–) + 2
(V+) – 1.4
(V+) – 1.4
V
VS < ±5V
(V–) + 2
VS ≥ ±5V, (V–) + 2V ≤ VCM ≤ (V+) – 2V
114
120
113
120
dB
V
VS < ±5V, (V–) + 2V ≤ VCM ≤ (V+) – 2V
108
120
93
100
dB
INPUT IMPEDANCE
Differential
20k || 8
20k || 8
Ω || pF
Common-Mode
109 || 2
109 || 2
Ω || pF
OPEN-LOOP GAIN
Open-Loop Voltage Gain
AOL
(V–) + 0.2V ≤ VO ≤ (V+) – 0.2V,
RL = 10kΩ
114
130
112
118
dB
AOL
(V–) + 0.6V ≤ VO ≤ (V+) – 0.6V,
RL = 600Ω
110
114
90
93
dB
FREQUENCY RESPONSE
Gain-Bandwidth Product
Slew Rate
Settling Time, 0.01%
GBW
G = 100
80
80
MHz
G=1
45
45
MHz
27
27
V/μs
VS = ±15V, G = –1, 10V Step, CL =
100pF
490
580
ns
VS = ±15V, G = –1, 10V Step, CL =
100pF
700
750
ns
G = –10
500
500
ns
G = 1, f = 1kHz,
VO = 3VRMS, RL = 600Ω
0.00001
5
0.000015
%
–136
–136
dB
SR
tS
0.0015% (16-bit)
Overload Recovery Time
Total Harmonic Distortion +
Noise
(1)
6
THD+N
The OPA211-HT is not intended to be used as a comparator due to its limited differential input range capability.
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ELECTRICAL CHARACTERISTICS: VS = ±2.25 V to ±18 V (continued)
At TA = 25°C, RL = 10 kΩ connected to midsupply, VCM = VOUT = midsupply, unless otherwise noted.
TA = –55 to 125°C
PARAMETER
CONDITIONS
MIN
RL = 10kΩ, AOL ≥ 114dB
RL = 600Ω, AOL ≥ 110dB, ±18V
TYP
TA = 210°C
MAX
MIN
(V–) + 0.2
(V+) – 0.2
(V–) + 0.6
(V+) – 0.6
TYP
MAX
UNIT
(V–) + 0.2
(V+) – 0.2
V
(V–) + 1.2
(V+) – 0.6
OUTPUT
Voltage Output
VOUT
Short-Circuit Current
ISC
Capacitive Load Drive
+35/–50
CLOAD
Open-Loop Output
Impedance
+30/–45
V
mA
See Typical Characteristics
ZO
f = 1MHz
Ω
5
POWER SUPPLY
Specified Voltage
VS
Quiescent Current
(per channel)
IQ
±2.25
IOUT = 0A
±18
3.6
±2.25
6
6.0
±18
V
7.5
mA
TEMPERATURE RANGE
Specified range
–55°C to 210°C
Operating range
–55°C to 210°C
1000000
100000
Estimated Life (Hours)
10000
1000
Electromigration Fail Mode
100
10
1
110
130
150
170
190
210
230
250
Continous T J (°C)
Figure 1. OPA211SKGD1 Operating Life Derating Chart
Notes:
1. See datasheet for absolute maximum and minimum recommended operating conditions.
2. Silicon operating life design goal is 10 years at 105°C junction temperature (does not include package
interconnect life).
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TYPICAL CHARACTERISTICS
At TA = 25°C, VS = ±18 V, and RL = 10 kΩ, unless otherwise noted.
INPUT VOLTAGE NOISE DENSITY
vs FREQUENCY
INPUT CURRENT NOISE DENSITY
vs FREQUENCY
100
Current Noise Density (pA/ÖHz)
Voltage Noise Density (nV/ÖHz)
100
10
10
1
1
0.1
1
10
100
1k
10k
0.1
100k
1
10
100
Frequency (Hz)
Figure 2.
-140
0.00001
100
1k
10k 20k
Total Harmonic Distortion + Noise (%)
Total Harmonic Distortion + Noise (%)
G=1
VOUT = 3VRMS
0.1
-60
0.01
-80
G = 11
-100
0.001
-120
0.0001
G=1
0.00001
VS = ±15V
RL = 600W
1kHz Signal
0.000001
0.01
-140
G = -1
0.1
1
10
Total Harmonic Distortion + Noise (dB)
-120
0.0001
Total Harmonic Distortion + Noise (dB)
G = 11
VOUT = 3VRMS
10
100k
THD+N RATIO vs OUTPUT VOLTAGE AMPLITUDE
-100
VS = ±15V
RL = 600W
G = -1
VOUT = 3VRMS
10k
Figure 3.
THD+N RATIO vs FREQUENCY
0.001
1k
Frequency (Hz)
-160
100
Output Voltage Amplitude (VRMS)
Frequency (Hz)
Figure 4.
Figure 5.
20nV/div
0.1-Hz TO 10-Hz NOISE
Time (1s/div)
Figure 6.
8
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TYPICAL CHARACTERISTICS (continued)
At TA = 25°C, VS = ±18 V, and RL = 10 kΩ, unless otherwise noted.
POWER-SUPPLY REJECTION RATIO
vs FREQUENCY (Referred to Input)
COMMON-MODE REJECTION RATIO
vs FREQUENCY
160
140
140
120
100
100
CMRR (dB)
PSRR (dB)
120
-PSRR
80
+PSRR
60
80
60
40
40
20
20
0
0
1
10
100
1k
10k
1M
100k
10M
10k
100M
100k
10M
1M
100M
Frequency (Hz)
Frequency (Hz)
Figure 7.
Figure 8.
OPEN-LOOP OUTPUT IMPEDANCE
vs FREQUENCY
GAIN AND PHASE vs FREQUENCY
180
140
10k
120
Gain (dB)
100
100
10
80
90
60
40
Gain
20
1
135
Phase
Phase (°)
ZO (W)
1k
45
0
-20
0.1
10
100
1k
10k
1M
100k
10M
100
100M
1k
10k
100k
1M
10M
0
100M
Frequency (Hz)
Frequency (Hz)
Figure 9.
Figure 10.
OPEN-LOOP GAIN vs TEMPERATURE
5
Open-Loop Gain (mV/V)
4
RL = 10kW
3
2
300mV Swing From Rails
1
0
-1
200mV Swing From Rails
-2
-3
-4
-5
-75 -50 -25
0
25
50
75 100 125 150 175 200
Temperature (°C)
Figure 11.
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TYPICAL CHARACTERISTICS (continued)
At TA = 25°C, VS = ±18 V, and RL = 10 kΩ, unless otherwise noted.
OFFSET VOLTAGE DRIFT PRODUCTION DISTRIBUTION
112.5
125.0
87.5
100.0
62.5
75.0
37.5
50.0
25.0
0
12.5
-12.5
-37.5
-25.0
-62.5
-50.0
-87.5
-75.0
-112.5
-100.0
-125.0
Population
Population
OFFSET VOLTAGE PRODUCTION DISTRIBUTION
0
0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1.0 1.1 1.2 1.3 1.4 1.5
Offset Voltage Drift (mV/°C)
Offset Voltage (mV)
Figure 12.
Figure 13.
OFFSET VOLTAGE vs COMMON-MODE VOLTAGE
200
2000
150
1500
100
1000
+IB
50
IOS
0
-50
-IB
-100
500
VOS (mV)
IB and IOS Bias Current (nA)
IB AND IOS CURRENT
vs
TEMPERATURE
0
-500
-1000
-150
-1500
-200
-2000
-50
-25
0
25
50
75
100
125
150
(V-)+1.0 (V-)+1.5 (V-)+2.0
(V+)-1.5 (V+)-1.0 (V+)-0.5
Ambient Temperature (°C)
VCM (V)
Figure 14.
Figure 15.
VOS WARMUP
12
10
INPUT OFFSET CURRENT vs SUPPLY VOLTAGE
100
20 Typical Units Shown
80
40
2
0
-2
-4
-6
20
0
-20
-40
-60
-8
-80
-10
-12
0
10
5 Typical Units Shown
60
6
4
IOS (nA)
VOS Shift (mV)
8
10
20
30
40
50
60
-100
2.25
4
6
8
10
Time (s)
VS (±V)
Figure 16.
Figure 17.
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14
16
18
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TYPICAL CHARACTERISTICS (continued)
At TA = 25°C, VS = ±18 V, and RL = 10 kΩ, unless otherwise noted.
INPUT OFFSET CURRENT vs COMMON-MODE VOLTAGE
INPUT BIAS CURRENT vs SUPPLY VOLTAGE
100
150
VS = 36V
3 Typical Units Shown
75
3 Typical Units Shown
100
Unit 1
Unit 2
50
25
IB (nA)
IOS (nA)
50
0
0
Unit 3
-25
-50
Common-Mode Range
-50
-100
-75
-IB
+IB
-100
1
5
10
15
20
25
30
-150
2.25
35
6
8
10
12
VS (±V)
Figure 18.
Figure 19.
INPUT BIAS CURRENT vs COMMON-MODE VOLTAGE
14
16
18
QUIESCENT CURRENT vs TEMPERATURE
6
150
-IB
VS = 36V
3 Typical Units Shown
50
+IB
5
4
Unit 2
Unit 1
IQ (mA)
100
IB (nA)
4
VCM (V)
0
3
2
-50
Unit 3
-100
1
Common-Mode Range
-150
0
1
5
10
15
20
25
30
-75 -50 -25
35
25
50
75 100 125 150 175 200
Figure 20.
Figure 21.
QUIESCENT CURRENT vs
SUPPLY VOLTAGE
NORMALIZED QUIESCENT CURRENT
vs TIME
0.05
3.5
0
3.0
-0.05
IQ Shift (mA)
4.0
2.5
IQ (mA)
0
Temperature (°C)
VCM (V)
2.0
1.5
-0.10
-0.15
-0.20
1.0
0.5
-0.25
0
-0.30
Average of 10 Typical Units
0
4
8
12
16
20
24
28
32
36
0
60
120 180 240 300 360 420 480 540
600
Time (s)
VS (V)
Figure 22.
Figure 23.
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TYPICAL CHARACTERISTICS (continued)
At TA = 25°C, VS = ±18 V, and RL = 10 kΩ, unless otherwise noted.
60
50
40
30
20
10
0
-10
-20
-30
-40
-50
SMALL-SIGNAL STEP RESPONSE
(100 mV)
G = -1
CL = 10pF
Sourcing
CF
5.6pF
20mV/div
ISC (mA)
SHORT-CIRCUIT CURRENT
vs TEMPERATURE
RI
604W
RF
604W
+18V
OPA211
CL
Sinking
-18V
-60
-75 -50 -25
0
25
50
75
Time (0.1ms/div)
100 125 150 175 200
Temperature (°C)
Figure 24.
Figure 25.
SMALL-SIGNAL STEP RESPONSE
(100 mV)
SMALL-SIGNAL STEP RESPONSE
(100 mV)
G = +1
RL = 600W
CL = 10pF
G = -1
CL = 100pF
RI
604W
20mV/div
20mV/div
CF
5.6pF
RF
604W
+18V
OPA211
+18V
OPA211
-18V
RL
CL
CL
-18V
Time (0.1ms/div)
Time (0.1ms/div)
Figure 26.
Figure 27.
SMALL-SIGNAL STEP RESPONSE
(100 mV)
SMALL-SIGNAL OVERSHOOT
vs CAPACITIVE LOAD (100-mV Output Step)
60
+18V
OPA211
-18V
G = +1
50
Overshoot (%)
20mV/div
G = +1
RL = 600W
CL = 100pF
RL
40
G = -1
30
G = 10
20
CL
10
Time (0.1ms/div)
0
0
200
400
600
800
1000
1200
1400
Capacitive Load (pF)
Figure 28.
12
Figure 29.
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TYPICAL CHARACTERISTICS (continued)
At TA = 25°C, VS = ±18 V, and RL = 10 kΩ, unless otherwise noted.
LARGE-SIGNAL STEP RESPONSE
LARGE-SIGNAL STEP RESPONSE
G = -1
CL = 100pF
RL = 600W
G = +1
CL = 100pF
RL = 600W
RF = 100W
2V/div
2V/div
RF = 0W
Note: See the
Applications Information
section, Input Protection.
Time (0.5ms/div)
Figure 31.
LARGE-SIGNAL POSITIVE SETTLING TIME
(10 VPP, CL = 100 pF)
LARGE-SIGNAL POSITIVE SETTLING TIME
(10 VPP, CL = 10 pF)
0.8
0.008
0.6
0.006
0.6
0.006
0.4
0.004
0.002
0
0
-0.002
-0.2
(±1/2 LSB = ±0.00075%)
-0.4
-0.004
0.4
0
(±1/2 LSB = ±0.00075%)
-0.004
-0.006
-0.008
-0.8
-1.0
-0.010
700 800 900 1000
-1.0
0
100
200 300
Figure 32.
-0.010
700 800 900 1000
LARGE-SIGNAL NEGATIVE SETTLING TIME
(10 VPP, CL = 10 pF)
0.8
0.008
0.8
0.008
0.6
0.006
0.6
0.006
0.4
0.004
0.002
0
0
-0.2
(±1/2 LSB = ±0.00075%)
-0.4
-0.002
-0.004
-0.6
-0.006
-0.8
-1.0
0
100
200 300
400 500 600
Time (ns)
0.010
0.4
16-Bit Settling
0.2
0
0.004
0.002
0
-0.2
(±1/2 LSB = ±0.00075%)
-0.4
-0.002
-0.004
-0.6
-0.006
-0.008
-0.8
-0.008
-0.010
700 800 900 1000
-1.0
0
100
Figure 34.
200 300
400 500 600
Time (ns)
D From Final Value (%)
16-Bit Settling
0.2
D From Final Value (mV)
0.010
1.0
D From Final Value (%)
D From Final Value (mV)
400 500 600
Time (ns)
Figure 33.
LARGE-SIGNAL NEGATIVE SETTLING TIME
(10 VPP, CL = 100 pF)
1.0
-0.002
-0.6
-0.008
400 500 600
Time (ns)
0.002
-0.4
-0.8
200 300
0.004
0
-0.2
-0.006
100
16-Bit Settling
0.2
-0.6
0
0.010
D From Final Value (%)
16-Bit Settling
0.2
D From Final Value (mV)
1.0
0.008
D From Final Value (%)
0.010
0.8
1.0
D From Final Value (mV)
Time (0.5ms/div)
Figure 30.
-0.010
700 800 900 1000
Figure 35.
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TYPICAL CHARACTERISTICS (continued)
At TA = 25°C, VS = ±18 V, and RL = 10 kΩ, unless otherwise noted.
NEGATIVE OVERLOAD RECOVERY
POSITIVE OVERLOAD RECOVERY
G = -10
VIN
G = -10
10kW
VOUT
1kW
0V
VOUT
OPA211
VIN
5V/div
5V/div
10kW
1kW
OPA211
VOUT
VIN
0V
VOUT
VIN
Time (0.5ms/div)
Time (0.5ms/div)
Figure 36.
Figure 37.
OUTPUT VOLTAGE vs OUTPUT CURRENT
NO PHASE REVERSAL
20
0°C
15
5
5V/div
VOUT (V)
Output
+85°C
+125°C
10
+125°C
0
-55°C
0°C
+150°C
-5
+18V
-10
Output
+85°C
37VPP
(±18.5V)
-15
-20
0
10
20
30
40
IOUT (mA)
50
60
-18V
0.5ms/div
70
Figure 38.
Figure 39.
TURN-OFF TRANSIENT
TURN-ON TRANSIENT
20
20
15
15
10
10
Output Signal
Shutdown Signal
5
5V/div
5
5V/div
OPA211
0
-5
0
Output Signal
-5
-10
-10
Shutdown Signal
-15
VS = ±15V
-20
-15
VS = ±15V
-20
Time (2ms/div)
Time (2ms/div)
Figure 40.
14
Figure 41.
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TYPICAL CHARACTERISTICS (continued)
At TA = 25°C, VS = ±18 V, and RL = 10 kΩ, unless otherwise noted.
TURN-ON/TURN-OFF TRANSIENT
20
1.6
15
1.2
10
0.8
5
0.4
0
-5
0
Output
-0.4
-10
-0.8
-15
-20
Output Voltage (V)
Shutdown Pin Voltage (V)
Shutdown Signal
-1.2
VS = ±15V
-1.6
Time (100ms/div)
Figure 42.
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APPLICATION INFORMATION
The OPA211 is a unity-gain stable, precision op amp
with very low noise. Applications with noisy or highimpedance power supplies require decoupling
capacitors close to the device pins. In most cases,
0.1-μF capacitors are adequate. Figure 43 shows a
simplified schematic of the OPA211. This die uses a
SiGe bipolar process and contains 180 transistors.
OPERATING VOLTAGE
OPA211 series op amps operate from ±2.25-V to
±18-V
supplies
while
maintaining
excellent
performance. The OPA211 series can operate with as
little as 4.5 V between the supplies and with up to 36
V between the supplies. However, some applications
do not require equal positive and negative output
voltage swing. With the OPA211 series, power-supply
voltages do not need to be equal. For example, the
positive supply could be set to 25 V with the negative
supply at –5 V or vice-versa.
The common-mode voltage must be maintained
within the specified range. In addition, key
parameters are assured over the specified
temperature range, TA = –55°C to 210°C. Parameters
that vary significantly with operating voltage or
temperature are shown in the Typical Characteristics.
V+
Pre-Output Driver
IN-
OUT
IN+
V-
Figure 43. OPA211 Simplified Schematic
16
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INPUT PROTECTION
-
OPA211
Input
Output
+
(1)
Figure 45 shows total circuit noise for varying source
impedances with the op amp in a unity-gain
configuration (no feedback resistor network, and
therefore no additional noise contributions). Two
different op amps are shown with total circuit noise
calculated. The OPA211 has very low voltage noise,
making it ideal for low source impedances (less than
2 kΩ). A similar precision op amp, the OPA227, has
somewhat higher voltage noise but lower current
noise. It provides excellent noise performance at
moderate source impedance (10 kΩ to 100 kΩ).
Above 100 kΩ, a FET-input op amp such as the
OPA132 (very low current noise) may provide
improved performance. The equation in Figure 45 is
shown for the calculation of the total circuit noise.
Note that en = voltage noise, In = current noise,
RS = source impedance, k = Boltzmann’s constant =
1.38 × 10–23 J/K, and T is temperature in K.
(1)
RS
OPA227
OPA211
100
Resistor Noise
10
2
2
2
EO = en + (in RS) + 4kTRS
1
100
1k
10k
100k
1M
Source Resistance, RS (W)
Figure 45. Noise Performance of the OPA211 and
OPA227 in Unity-Gain Buffer Configuration
Design of low-noise op amp circuits requires careful
consideration of a variety of possible noise
contributors: noise from the signal source, noise
generated in the op amp, and noise from the
feedback network resistors. The total noise of the
circuit is the root-sum-square combination of all noise
components.
The resistive portion of the source impedance
produces thermal noise proportional to the square
root of the resistance. This function is plotted in
Figure 45. The source impedance is usually fixed;
consequently, select the op amp and the feedback
resistors to minimize the respective contributions to
the total noise.
Figure 44. Pulsed Operation
NOISE PERFORMANCE
EO
1k
BASIC NOISE CALCULATIONS
RF
RI
10k
Votlage Noise Spectral Density, EO
The input terminals of the OPA211 are protected from
excessive differential voltage with back-to-back
diodes, as shown in Figure 44. In most circuit
applications, the input protection circuitry has no
consequence. However, in low-gain or G = 1 circuits,
fast ramping input signals can forward bias these
diodes because the output of the amplifier cannot
respond rapidly enough to the input ramp. This effect
is illustrated in Figure 31 of the Typical
Characteristics. If the input signal is fast enough to
create this forward bias condition, the input signal
current must be limited to 10mA or less. If the input
signal current is not inherently limited, an input series
resistor can be used to limit the signal input current.
This input series resistor degrades the low-noise
performance of the OPA211, and is discussed in the
Noise Performance section of this data sheet.
Figure 44 shows an example implementing a currentlimiting feedback resistor.
VOLTAGE NOISE SPECTRAL DENSITY
vs SOURCE RESISTANCE
Figure 45 depicts total noise for varying source
impedances with the op amp in a unity-gain
configuration (no feedback resistor network, and
therefore no additional noise contributions). The
operational amplifier itself contributes both a voltage
noise component and a current noise component.
The voltage noise is commonly modeled as a timevarying component of the offset voltage. The current
noise is modeled as the time-varying component of
the input bias current and reacts with the source
resistance to create a voltage component of noise.
Therefore, the lowest noise op amp for a given
application depends on the source impedance. For
low source impedance, current noise is negligible and
voltage noise generally dominates. For high source
impedance, current noise may dominate.
OPA227 and OPA132 have not been characterized or tested
at 210°C.
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Figure 46 illustrates both inverting and noninverting
op amp circuit configurations with gain. In circuit
configurations with gain, the feedback network
resistors also contribute noise. The current noise of
the op amp reacts with the feedback resistors to
create additional noise components. The feedback
resistor values can generally be chosen to make
these noise sources negligible. The equations for
total noise are shown for both configurations.
TOTAL HARMONIC DISTORTION
MEASUREMENTS
OPA211 series op amps have excellent distortion
characteristics. THD + Noise is below 0.0001%
(G = 1, VO = 3 VRMS) throughout the audio frequency
range, 20 Hz to 20 kHz, with a 600-Ω load.
The distortion produced by OPA211 series op amps
is below the measurement limit of many commercially
available distortion analyzers. However, a special test
circuit illustrated in Figure 47 can be used to extend
the measurement capabilities.
Op amp distortion can be considered an internal error
source that can be referred to the input. Figure 47
shows a circuit that causes the op amp distortion to
be 101 times greater than that normally produced by
the op amp. The addition of R3 to the otherwise
standard noninverting amplifier configuration alters
the feedback factor or noise gain of the circuit. The
closed-loop gain is unchanged, but the feedback
available for error correction is reduced by a factor of
18
101, thus extending the resolution by 101. Note that
the input signal and load applied to the op amp are
the same as with conventional feedback without R3.
The value of R3 should be kept small to minimize its
effect on the distortion measurements.
Validity of this technique can be verified by
duplicating measurements at high gain and/or high
frequency where the distortion is within the
measurement capability of the test equipment.
Measurements for this data sheet were made with an
Audio Precision System Two distortion/noise
analyzer, which greatly simplifies such repetitive
measurements. The measurement technique can,
however, be performed with manual distortion
measurement instruments.
SHUTDOWN
The shutdown (enable) function of the OPA211 is
referenced to the positive supply voltage of the
operational amplifier. A valid high disables the op
amp. A valid high is defined as (V+) – 0.35 V of the
positive supply applied to the shutdown pin. A valid
low is defined as (V+) – 3 V below the positive supply
pin. For example, with VCC at ±15 V, the device is
enabled at or below 12 V. The device is disabled at
or above 14.65 V. If dual or split power supplies are
used, care should be taken to ensure the valid high
or valid low input signals are properly referred to the
positive supply voltage. This pin must be connected
to a valid high or low voltage or driven, and not left
open-circuit. The enable and disable times are
provided in the Typical Characteristics section (see
Figure 40 through Figure 42). When disabled, the
output assumes a high-impedance state.
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Noise in Noninverting Gain Configuration
Noise at the output:
R2
2
2
EO
R1
= 1+
R2
R1
2
2
2
2
2
2
en + e1 + e2 + (inR2) + eS + (inRS)
EO
R2
Where eS = Ö4kTRS ´ 1 +
R1
2
1+
R2
R1
= thermal noise of RS
RS
e1 = Ö4kTR1 ´
VS
R2
R1
= thermal noise of R1
e2 = Ö4kTR2 = thermal noise of R2
Noise in Inverting Gain Configuration
Noise at the output:
R2
2
2
EO = 1 +
R1
R2
R1 + RS
2
EO
RS
Where eS = Ö4kTRS ´
2
2
2
2
en + e1 + e2 + (inR2) + eS
R2
R1 + RS
= thermal noise of RS
VS
e1 = Ö4kTR1 ´
R2
R1 + RS
= thermal noise of R1
e2 = Ö4kTR2 = thermal noise of R2
For the OPA211 series op amps at 1kHz, en = 1.1nV/ÖHz and in = 1.7pA/ÖHz.
Figure 46. Noise Calculation in Gain Configurations
R1
R2
SIG. DIST.
GAIN GAIN
R3
Signal Gain = 1+
OPA211
VOUT
R2
R1
Distortion Gain = 1+
R2
R1 II R3
Generator
Output
R1
R2
R3
1
101
¥
1kW
10W
11
101
100W
1kW
11W
Analyzer
Input
Audio Precision
System Two(1)
with PC Controller
Load
Figure 47. Distortion Test Circuit
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ELECTRICAL OVERSTRESS
An ESD event produces a short duration, highvoltage pulse that is transformed into a short
duration, high-current pulse as it discharges through
a semiconductor device. The ESD protection circuits
are designed to provide a current path around the
operational amplifier core to prevent it from being
damaged. The energy absorbed by the protection
circuitry is then dissipated as heat.
Designers often ask questions about the capability of
an operational amplifier to withstand electrical
overstress. These questions tend to focus on the
device inputs, but may involve the supply voltage pins
or even the output pin. Each of these different pin
functions have electrical stress limits determined by
the voltage breakdown characteristics of the
particular semiconductor fabrication process and
specific circuits connected to the pin. Additionally,
internal electrostatic discharge (ESD) protection is
built into these circuits to protect them from
accidental ESD events both before and during
product assembly.
When an ESD voltage develops across two or more
of the amplifier device pins, current flows through one
or more of the steering diodes. Depending on the
path that the current takes, the absorption device
may activate. The absorption device has a trigger, or
threshold voltage, that is above the normal operating
voltage of the OPA211 but below the device
breakdown voltage level. Once this threshold is
exceeded, the absorption device quickly activates
and clamps the voltage across the supply rails to a
safe level.
It is helpful to have a good understanding of this
basic ESD circuitry and its relevance to an electrical
overstress event. Figure 48 illustrates the ESD
circuits contained in the OPA211 (indicated by the
dashed line area). The ESD protection circuitry
involves several current-steering diodes connected
from the input and output pins and routed back to the
internal power-supply lines, where they meet at an
absorption device internal to the operational amplifier.
This protection circuitry is intended to remain inactive
during normal circuit operation.
When the operational amplifier connects into a circuit
such as that illustrated in Figure 48, the ESD
protection components are intended to remain
inactive and not become involved in the application
circuit operation. However, circumstances may arise
where an applied voltage exceeds the operating
voltage range of a given pin. Should this condition
occur, there is a risk that some of the internal ESD
protection circuits may be biased on, and conduct
current. Any such current flow occurs through
steering diode paths and rarely involves the
absorption device.
RF
+V
+VS
OPA211
RI
ESD CurrentSteering Diodes
-In
+In
Op-Amp
Core
Edge-Triggered ESD
Absorption Circuit
ID
VIN
Out
RL
(1)
-V
-VS
(1) VIN = +VS + 500mV.
Figure 48. Equivalent Internal ESD Circuitry and Its Relation to a Typical Circuit Application
20
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Figure 48 depicts a specific example where the input
voltage, VIN, exceeds the positive supply voltage
(+VS) by 500 mV or more. Much of what happens in
the circuit depends on the supply characteristics. If VS
can sink the current, one of the upper input steering
diodes conducts and directs current to VS.
Excessively high current levels can flow with
increasingly higher VIN. As a result, the datasheet
specifications recommend that applications limit the
input current to 10 mA.
If the supply is not capable of sinking the current, VIN
may begin sourcing current to the operational
amplifier, and then take over as the source of positive
supply voltage. The danger in this case is that the
voltage can rise to levels that exceed the operational
amplifier absolute maximum ratings. In extreme but
rare cases, the absorption device triggers on while VS
and –VS are applied. If this event happens, a direct
current path is established between the VS and –VS
supplies. The power dissipation of the absorption
device is quickly exceeded, and the extreme internal
heating destroys the operational amplifier.
Another common question involves what happens to
the amplifier if an input signal is applied to the input
while the power supplies VS and/or –VS are at 0 V.
Again, it depends on the supply characteristic while at
0 V, or at a level below the input signal amplitude. If
the supplies appear as high impedance, then the
operational amplifier supply current may be supplied
by the input source via the current steering diodes.
This state is not a normal bias condition; the amplifier
most likely will not operate normally. If the supplies
are low impedance, then the current through the
steering diodes can become quite high. The current
level depends on the ability of the input source to
deliver current, and any resistance in the input path.
DFN PACKAGE
The OPA211 is offered in an DFN-8 package (also
known as SON). The DFN package is a QFN
package with lead contacts on only two sides of the
bottom of the package. This leadless package
maximizes board space and enhances thermal and
electrical characteristics through an exposed pad.
DFN packages are physically small, and have a
smaller routing area, improved thermal performance,
and improved electrical parasitics. Additionally, the
absence of external leads eliminates bent-lead
issues.
The DFN package can be easily mounted using
standard printed circuit board (PCB) assembly
techniques. See Application Note QFN/SON PCB
Attachment (SLUA271) and Application Report Quad
Flatpack No-Lead Logic Packages (SCBA017), both
available for download at www.ti.com.
The exposed leadframe die pad on the bottom of
the package must be connected to V–. Soldering
the thermal pad improves heat dissipation and
enables specified device performance.
DFN LAYOUT GUIDELINES
The exposed leadframe die pad on the DFN package
should be soldered to a thermal pad on the PCB. A
mechanical drawing showing an example layout is
attached at the end of this data sheet. Refinements to
this layout may be necessary based on assembly
process requirements. Mechanical drawings located
at the end of this data sheet list the physical
dimensions for the package and pad. The five holes
in the landing pattern are optional, and are intended
for use with thermal vias that connect the leadframe
die pad to the heatsink area on the PCB.
Soldering the exposed pad significantly improves
board-level reliability during temperature cycling, key
push, package shear, and similar board-level tests.
Even with applications that have low-power
dissipation, the exposed pad must be soldered to the
PCB to provide structural integrity and long-term
reliability.
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PACKAGE OPTION ADDENDUM
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15-Jun-2022
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
Samples
(4/5)
(6)
OPA211SHKJ
ACTIVE
CFP
HKJ
8
25
RoHS & Green
Call TI
N / A for Pkg Type
-55 to 210
OPA211S
HKJ
Samples
OPA211SHKQ
ACTIVE
CFP
HKQ
8
25
RoHS & Green
AU
N / A for Pkg Type
-55 to 210
OPA211S
HKQ
Samples
OPA211SKGD1
ACTIVE
XCEPT
KGD
0
400
RoHS & Green
Call TI
N / A for Pkg Type
-55 to 210
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of
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