OPA2822
OPA
282
2
SBOS188E – MARCH 2001 – REVISED AUGUST 2008
www.ti.com
Dual, Wideband, Low-Noise
Operational Amplifier
FEATURES
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DESCRIPTION
LOW INPUT NOISE VOLTAGE: 2.0nV/√Hz
HIGH UNITY GAIN BANDWIDTH: 500MHz
HIGH GAIN BANDWIDTH PRODUCT: 240MHz
HIGH OUTPUT CURRENT: 90mA
SINGLE +5V TO +12V OPERATION
LOW SUPPLY CURRENT: 4.8mA/ch
The OPA2822 offers very low 2.0nV/√Hz input noise in a
wideband, unity-gain stable, voltage-feedback architecture.
Intended for xDSL receiver applications, the OPA2822 also
supports this low input noise with exceptionally low harmonic
distortion, particularly in differential configurations. Adequate
output current is provided to drive the potentially heavy load
of a passive filter between this amplifier and the codec.
Harmonic distortion for a 2VPP differential output operating
from +5V to +12V supplies is ≤ –100dBc through 1MHz input
frequencies. Operating on a low 4.8mA/ch supply current,
the OPA2822 can satisfy all xDSL receiver requirements
over a wide range of possible supply voltages—from a single
+5V condition, to ±5V, up to a single +12V design.
APPLICATIONS
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xDSL DIFFERENTIAL LINE RECEIVERS
HIGH DYNAMIC RANGE ADC DRIVERS
LOW NOISE PLL INTEGRATORS
TRANSIMPEDANCE AMPLIFIERS
PRECISION BASEBAND I/Q AMPLIFIERS
ACTIVE FILTERS
General-purpose applications on a single +5V supply will
benefit from the high input and output voltage swing available
on this reduced supply voltage. Low-cost precision integrators for PLLs will also benefit from the low voltage noise and
offset voltage. Baseband I/Q receiver channels can achieve
almost perfect channel match with noise and distortion to
support signals through 5MHz with > 14-bit dynamic range.
OPA2677
RO
n:1
OPA2822 RELATED PRODUCTS
xDSL Driver
RO
500Ω
FEATURES
High Slew Rate
1kΩ
SINGLES
DUALS
TRIPLES
OPA690
OPA2690
OPA3690
R/R Input/Output
OPA353
OPA2353
—
1.3nV Input Noise
OPA846
OPA2686
—
1.5nV Input Noise
—
THS6062
—
500Ω
OPA2822
500Ω
1kΩ
xDSL Receiver
OPA2822
500Ω
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
All trademarks are the property of their respective owners.
Copyright © 2001-2008, Texas Instruments Incorporated
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of Texas Instruments
standard warranty. Production processing does not necessarily include
testing of all parameters.
www.ti.com
PACKAGE/ORDERING INFORMATION(1)
SPECIFIED
TEMPERATURE
RANGE
PACKAGE
MARKING
PRODUCT
PACKAGE-LEAD
PACKAGE
DESIGNATOR
OPA2822U
SO-8 Surface-Mount
D
–40°C to +85°C
OPA2822U
OPA2822U
Rails, 100
"
"
"
"
OPA2822U/2K5
Tape and Reel, 2500
MSOP-8 Surface-Mount
DGK
–40°C to +85°C
D22
OPA2822E/250
Tape and Reel, 250
"
"
"
"
OPA2822E/2K5
Tape and Reel, 2500
"
OPA2822E
"
ORDERING
NUMBER
TRANSPORT
MEDIA, QUANTITY
NOTE: (1) For the most current package and ordering information, see the Package Option Addendum located at the end of this data sheet.
ABSOLUTE MAXIMUM RATINGS(1)
Supply Voltage ................................................................................. ±6.5V
Internal Power Dissipation ........................... See Thermal Characteristics
Differential Input Voltage .................................................................. ±1.2V
Input Voltage Range ............................................................................ ±VS
Storage Temperature Range ......................................... –65°C to +125°C
Lead Temperature (SO-8) ............................................................. +260°C
Junction Temperature (TJ ) ........................................................... +150°C
ESD Rating (Human Body Model) .................................................. 2000V
(Machine Model) ........................................................... 200V
NOTE: (1) Stresses above these ratings may cause permanent damage.
Exposure to absolute maximum conditions for extended periods may degrade
device reliability. These are stress ratings only, and functional operation of the
device at these or any other conditions beyond those specified is not implied.
ELECTROSTATIC
DISCHARGE SENSITIVITY
Electrostatic discharge can cause damage ranging from performance degradation to complete device failure. Texas Instruments recommends that all integrated circuits be handled
and stored using appropriate ESD protection methods.
ESD damage can range from subtle performance degradation
to complete device failure. Precision integrated circuits may be
more susceptible to damage because very small parametric
changes could cause the device not to meet published specifications.
PIN CONFIGURATION / MSOP PACKING MARKING
Top View
SO
MSOP PACKAGE MARKING
OPA2822
8
Out A
1
8
+VS
–In A
2
7
Out B
+In A
3
6
–In B
–VS
4
5
+In B
6
5
D22
1
2
7
2
3
4
OPA2822
www.ti.com
SBOS188E
ELECTRICAL CHARACTERISTICS: VS = ±6V
Boldface limits are tested at +25°C.
RF = 402Ω, RL = 100Ω, and G = +2, (see Figure 1 for AC performance only), unless otherwise noted.
OPA2822U, E
TYP
PARAMETER
AC PERFORMANCE (see Figure 1)
Small-Signal Bandwidth
Gain-Bandwidth Product
Bandwidth for 0.1dB Gain Flatness
Peaking at a Gain of +1
Large-Signal Bandwidth
Slew Rate
Rise-and-Fall Time
Settling Time to 0.02%
0.1%
Harmonic Distortion
2nd-Harmonic
3rd-Harmonic
Input Voltage Noise
Input Current Noise
Differential Gain
Differential Phase
Channel-to-Channel Crosstalk
DC PERFORMANCE(4)
Open-Loop Voltage Gain (AOL)
Input Offset Voltage
Average Offset Voltage Drift
Input Bias Current
Average Bias Current Drift (magnitude)
Input Offset Current
Average Offset Current Drift
INPUT
Common-Mode Input Range (CMIR)(5)
Common-Mode Rejection Ratio (CMRR)
Input Impedance
Differential-Mode
Common-Mode
OUTPUT
Voltage Output Swing
Current Output, Sourcing
Current Output, Sinking
Short-Circuit Current
Closed-Loop Output Impedance
POWER SUPPLY
Specified Operating Voltage
Maximum Operating Voltage Range
Max Quiescent Current
Min Quiescent Current
Power-Supply Rejection Ratio (–PSRR)
THERMAL CHARACTERISTICS
Specified Operating Range U, E Package
Thermal Resistance, θJA
U SO-8
E MSOP
CONDITIONS
+25°C
G = +1, VO = 0.1VPP, RF = 0Ω
G = +2, VO = 0.1VPP
G = +10, VO = 0.1VPP
G ≥ 20
G = +2, VO < 0.1VPP
VO < 0.1VPP
G = +2, VO = 2VPP
G = +2, 4V Step
G = +2, VO = 0.2V Step
G = +2, VO = 2V Step
G = +2, VO = 2V Step
400
200
24
240
16
5
27
170
1.5
35
32
G = +2, f = 1MHz, VO = 2VPP
RL = 200Ω
RL ≥ 500Ω
RL = 200Ω
RL ≥ 500Ω
f > 10kHz
f > 10kHz
G = +2, PAL, VO = 1.4Vp, RL = 150
G = +2, PAL, VO = 1.4Vp, RL = 150
f = 1MHz, Input Referred
–91
–95
–100
–105
2.0
1.6
0.02
0.03
–95
VO = 0V, RL = 100Ω
VCM = 0V
VCM = 0V
VCM = 0V
VCM = 0V
VCM = 0V
VCM = 0V
MIN/MAX OVER TEMPERATURE
+25°C(1)
0°C to
70°C(2)
–40°C to
+85°C(2)
120
15
150
110
13
130
105
12
125
110
105
100
–88
–91
–95
–99
2.2
2.0
–87
–90
–92
–96
2.3
2.1
85
MIN/ TEST
MAX LEVEL(3)
MHz
MHz
MHz
MHz
MHz
dB
MHz
V/µs
ns
ns
ns
typ
min
min
min
typ
typ
typ
min
typ
typ
typ
C
B
B
B
C
C
C
B
C
C
C
–86
–89
–91
–95
2.5
2.3
dBc
dBc
dBc
dBc
nV/√Hz
pA/√Hz
%
deg
dBc
max
max
max
max
max
max
typ
typ
typ
B
B
B
B
B
B
C
C
C
82
±1.4
5
–19
50
±600
5
80
±1.5
5
–21
50
±700
5
dB
mV
µV/°C
µA
nA/°C
nA
nA/°C
min
max
max
max
max
max
max
A
A
B
A
B
A
B
±4.4
82
±4.4
80
V
dB
min
min
A
A
kΩ || pF
MΩ || pF
typ
typ
C
C
V
V
mA
mA
mA
Ω
min
min
min
min
typ
typ
A
A
A
A
C
C
V
V
mA
mA
dB
typ
max
max
min
min
C
A
A
A
A
–40 to +85
°C
typ
C
125
150
°C/W
°C/W
typ
typ
C
C
100
±0.2
±1.2
–9
–18
±100
±400
VCM = ±1V
±4.8
110
±4.5
VCM = 0
VCM = 0
18 0.6
7 1
No Load
100Ω Load
VO = 0, Linear Operation
VO = 0, Linear Operation
Output Shorted to Ground
G = +2, f = 100kHz
±4.9
±4.7
+150
–150
220
0.01
±6
VS = ±6V, both channels
VS = ±6V, both channels
Input Referred
UNITS
9.6
9.6
95
85
±4.7
±4.5
+90
–90
±6.3
11.8
8.2
85
±4.6
±4.4
+85
–85
±4.6
±4.4
+80
–80
±6.3
11.9
8.1
82
±6.3
12.0
8.0
80
Junction-to-Ambient
NOTES: (1) Junction temperature = ambient for +25°C tested specifications.
(2) Junction temperature = ambient at low temperature limit: junction temperature = ambient +23°C at high temperature limit for over temperature tested
specifications.
(3) Test Levels: (A) 100% tested at +25°C. Over temperature limits by characterization and simulation. (B) Limits set by characterization and simulation. (C)
Typical value only for information.
(4) Current is considered positive-out-of node. VCM is the input common-mode voltage.
(5) Tested < 3dB below minimum CMRR specification at ± CMIR limits.
OPA2822
SBOS188E
www.ti.com
3
ELECTRICAL CHARACTERISTICS: VS = +5V
Boldface limits are tested at +25°C.
RF = 402Ω, RL = 100Ω to VS / 2, and G = +2, (see Figure 3 for AC performance only), unless otherwise noted.
OPA2822U, E
TYP
+25°C(1)
0°C to
70°C(2)
–40°C to
+85°C(2)
105
13
130
102
11
110
100
10
105
90
2.7
85
3.2
80
3.3
–85
–87
–99
–103
2.1
1.5
–82
–83
–94
–98
2.3
1.9
–81
–82
–91
–95
2.4
2.0
90
±0.3
±1.3
81
–8
–16
±100
±400
VCM = +2.5V
1.2
3.8
110
1.5
3.5
85
VCM = +2.5V
VCM = +2.5V
15 1
5 1.3
No Load
RL = 100Ω to 2.5V
No Load
RL = 100Ω to 2.5V
3.9
3.7
1.3
1.4
+150
–150
200
0.01
CONDITIONS
+25°C
G = +1, VO = 0.1VPP, RF = 0Ω
G = +2, VO = 0.1VPP
G = +10, VO = 0.1VPP
G > 20
VO < 0.1VPP
G = +2, VO = 2VPP
G = +2, 2V Step
G = +2, VO = 0.2V Step
G = +2, VO = 2V Step
G = +2, VO = 2V Step
350
180
20
200
6
20
120
2.0
40
38
Input Voltage Noise
Input Current Noise
G = +2, f = 1MHz, VO = 2VPP
RL = 200Ω to VS /2
RL = 500Ω to VS /2
RL = 100Ω to VS /2
RL = 1500Ω to VS /2
f > 1MHz
f > 1MHz
DC PERFORMANCE(4)
Open-Loop Voltage Gain
Input Offset Voltage
Average Offset Voltage Drift
Input Bias Current
Average Bias Current Drift
Input Offset Current
Average Offset Current Drift
VO = 0V, RL = 200Ω to 2.5V
VCM = 2.5V
VCM = 2.5V
VCM = 2.5V
VCM = 2.5V
VCM = 2.5V
VCM = 2.5V
PARAMETER
AC PERFORMANCE (see Figure 3)
Small-Signal Bandwidth
Gain-Bandwidth Product
Peaking at a Gain of +1
Large-Signal Bandwidth
Slew Rate
Rise-and-Fall Time
Settling Time to 0.02%
0.1%
Harmonic Distortion
2nd-Harmonic
3rd-Harmonic
INPUT
Least Positive Input Voltage
Most Positive Input Voltage
Common-Mode Rejection Ratio (CMRR)
Input Impedance
Differential-Mode
Common-Mode
OUTPUT
Most Positive Output Voltage
Least Positive Output Voltage
Current Output, Sourcing
Current Output, Sinking
Short-Circuit Current
Closed-Loop Output Impedance
POWER SUPPLY
Specified Single-Supply Operating Voltage
Maximum Single-Supply Operating Voltage
Max Quiescent Current
Min Quiescent Current
Power-Supply Rejection Ratio
THERMAL CHARACTERISTICS
Specified Operating Range U, E Package
Thermal Resistance, θJA
U SO-8
E MSOP
MIN/MAX OVER TEMPERATURE
Output Shorted to Either Supply
G = +1, f = 100kHz
MHz
MHz
MHz
MHz
dB
MHz
V/µs
ns
ns
ns
typ
min
min
min
typ
typ
min
max
typ
typ
C
B
B
B
C
C
B
B
C
C
–80
–81
–90
–94
2.6
2.1
dBc
dBc
dBc
dBc
nV/√Hz
pA/√Hz
max
max
max
max
max
max
B
B
B
B
B
B
78
±1.5
5.5
–19
50
±600
5
76
±1.6
5.5
–20
50
±700
5
dB
mV
µV/°C
µA
nA/°C
nA
nA/°C
min
max
max
max
max
max
max
A
A
B
A
B
A
B
1.6
3.4
82
1.65
3.35
80
V
V
dB
min
max
min
A
A
A
kΩ || pF
MΩ || pF
typ
typ
C
C
V
V
V
V
mA
mA
mA
Ω
min
min
min
min
min
min
typ
typ
A
A
A
A
A
A
C
C
V
V
mA
mA
dB
typ
max
max
min
typ
C
A
A
A
C
–40 to +85
°C
typ
C
125
150
°C/W
°C/W
typ
typ
C
C
3.8
3.5
1.4
1.5
+90
–90
3.6
3.4
1.5
1.6
+85
–85
3.5
3.35
1.55
1.65
+80
–80
12.6
10
7.2
12.6
10.2
7.0
12.6
10.4
6.9
5
VS = +5V, both channels
VS = +5V, both channels
Input Referred
MIN/ TEST
MAX LEVEL(3)
UNITS
8
8
90
Junction-to-Ambient
NOTES: (1) Junction temperature = ambient for +25°C tested specifications.
(2) Junction temperature = ambient at low temperature limit: junction temperature = ambient +23°C at high temperature limit for over temperature tested
specifications.
(3) Test Levels: (A) 100% tested at +25°C. Over temperature limits by characterization and simulation. (B) Limits set by characterization and simulation.
(C) Typical value only for information.
(4) Current is considered positive-out-of node. VCM is the input common-mode voltage.
4
OPA2822
www.ti.com
SBOS188E
TYPICAL CHARACTERISTICS: VS = ±6V
TA = +25°C, G = +2, RF = 402Ω, and RL = 100Ω, unless otherwise noted.
NONINVERTING SMALL-SIGNAL
FREQUENCY RESPONSE
3
6
G = +1
RF = 0Ω
VO = 0.1VPP
–3
G = +2
–6
–9
G = +5
–12
–15
G = +10
–18
–6
G = –5
–9
–12
G = –10
–15
See Figure 2
–24
0.5
1
10
100
500
0.5
1
10
Frequency (MHz)
NONINVERTING LARGE-SIGNAL
FREQUENCY RESPONSE
INVERTING LARGE-SIGNAL
FREQUENCY RESPONSE
G = +2
6
VO = 0.1VPP
0
3
–3
Gain (dB)
VO = 0.5VPP
0
VO = 1VPP
–3
–6
VO = 2VPP
500
VO = 0.1VPP
G = –1
RF = 604Ω
3
6
–9
VO = 0.5VPP
–6
VO = 1VPP
–9
–12
VO = 2VPP
–15
–18
–12
–21
See Figure 1
0.5
1
See Figure 2
–24
10
100
0.5
500
1
10
NONINVERTING PULSE RESPONSE
Large-Signal Right Scale
200
1.5
1.0
100
0.5
Small-Signal Left Scale
0
0
–100
–0.5
–200
–1.0
–300
–1.5
See Figure 1
–400
400
Small-Signal Output Voltage (100mv/div)
G = +2
300
–2.0
Time (20ns/div)
2.0
G = –1
300
1.5
200
1.0
100
0.5
0
0
Small-Signal Left Scale
–100
–0.5
–200
–1.0
Large-Signal Right Scale
–300
–1.5
See Figure 2
–400
–2.0
Time (20ns/div)
OPA2822
SBOS188E
500
INVERTING PULSE RESPONSE
2.0
Larege-Signal Output Voltage (500mv/div)
400
100
Frequency (MHz)
Frequency (MHz)
Small-Signal Output Voltage (100mv/div)
100
Frequency (MHz)
9
Gain (dB)
–3
–21
See Figure 1
–24
–18
G = –2
–18
–21
–15
G = –1
0
Normalized Gain (dB)
Normalized Gain (dB)
0
12
VO = 0.1VPP
RF = 604Ω
3
Larege-Signal Output Voltage (500mv/div)
6
INVERTING SMALL-SIGNAL
FREQUENCY RESPONSE
www.ti.com
5
TYPICAL CHARACTERISTICS: VS = ±6V (Cont.)
TA = +25°C, G = +2, RF = 402Ω, and RL = 100Ω, unless otherwise noted.
HARMONIC DISTORTION vs LOAD RESISTANCE
1MHz HARMONIC DISTORTION vs SUPPLY VOLTAGE
–85
VO = 2VPP
f = 1MHz
VO = 2VPP
RL = 200Ω
Harmonic Distortion (dBc)
Harmonic Distortion (dBc)
–85
–90
2nd-Harmonic
–95
–100
3rd-Harmonic
See Figure 1
2nd-Harmonic
–95
3rd-Harmonic
–100
See Figure 1
–105
±2.5
±3.0
±3.5
–105
100
–90
1k
Load Resistance (Ω)
–85
Harmonic Distortion (dBc)
Harmonic Distortion (dBc)
±5.0
±5.5
±6.0
RL = 200Ω
f = 1MHz
VO = 2VPP
RL = 200Ω
–75
2nd-Harmonic
–85
3rd-Harmonic
–95
2nd-Harmonic
–90
–95
3rd-Harmonic
–100
See Figure 1
See Figure 1
–105
–105
1
0.1
10
1
10
Output Voltage Swing (VPP)
Frequency (MHz)
HARMONIC DISTORTION vs NONINVERTING GAIN
HARMONIC DISTORTION vs INVERTING GAIN
–70
–70
VO = 2VPP
RL = 200Ω
f = 1MHz
–80
Harmonic Distortion (dBc)
Harmonic Distortion (dBc)
±4.5
HARMONIC DISTORTION vs OUTPUT VOLTAGE
HARMONIC DISTORTION vs FREQUENCY
–65
2nd-Harmonic
–90
3rd-Harmonic
–100
VO = 2VPP
RL = 200Ω
RF = 604Ω
f = 1MHz
–80
2nd-Harmonic
–90
3rd-Harmonic
–100
See Figure 2
See Figure 1
–110
–110
1
10
1
Gain (V/V)
6
±4.0
Supply Voltage (V)
10
Gain (V/V)
OPA2822
www.ti.com
SBOS188E
TYPICAL CHARACTERISTICS: VS = ±6V (Cont.)
TA = +25°C, G = +2, RF = 402Ω, and RL = 100Ω, unless otherwise noted.
2-TONE, 3rd-ORDER
INTERMODULATION INTERCEPT
INPUT VOLTAGE AND CURRENT NOISE DENSITY
60
10
Intercept Point (+dBm)
Voltage Noise nV/√Hz
Current Noise pA/√Hz
55
Voltage Noise
2nV/√Hz
45
40
PI
35
50Ω
1/2
OPA2822
102
50Ω
402Ω
103
104
105
106
1
107
10
20
Frequency (MHz)
Frequency (Hz)
CHANNEL-TO-CHANNEL CROSSTALK
GAIN FLATNESS
0.50
Deviation from 6dB Gain (0.1dB/div)
–40
Cross-Talk Input Referred (dB)
PO
20
1
–50
50Ω
402Ω
30
25
1.6pA/√Hz
Current Noise
50
Input Referred
RL = 100Ω
G = +2
–60
–70
–80
–90
–100
0.40
NG = 2
RNG = ∞
0.30
NG = 2.5
RNG = 904Ω
0.20
0.10
G=2
Noise Gain
Adjusted
0.00
–0.10
NG = 3.0
RNG = 452Ω
–0.20
–0.30
NG = 3.5
RNG = 301Ω
–0.40
–0.50
0.1
1
10
100
500
0
See Figure 12
50
Frequency (MHz)
Normalized Gain to Capacitive Load (dB)
RS (Ω)
100
10
For Maximally Flat Response,
See Figure 12
10
100
200
9
6
CL = 10pF
CL = 100pF
3
CL = 22pF
VI
0
RS
1/2
OPA2822
–3
VO
CL
CL = 47pF
1kΩ
402Ω
–6
1kΩ is optional.
402Ω
–9
–12
1
1k
10
100
500
Frequency (MHz)
Capacitive Load (pF)
OPA2822
SBOS188E
150
FREQUENCY RESPONSE vs CAPACITIVE LOAD
RECOMMENDED RS vs CAPACITIVE LOAD
1000
1
100
Frequency (MHz)
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7
TYPICAL CHARACTERISTICS: VS = ±6V (Cont.)
TA = +25°C, G = +2, RF = 402Ω, and RL = 100Ω, unless otherwise noted.
OPEN-LOOP GAIN AND PHASE
CMRR AND PSRR vs FREQUENCY
CMRR
+PSRR
100
100
–PSRR
80
60
40
–30
20 log(AOL)
80
–60
∠ AOL
60
20
104
105
106
107
40
–120
20
–150
0
–180
–210
102
108
103
104
106
107
108
Frequency (Hz)
OUTPUT VOLTAGE AND CURRENT LIMITATIONS
CLOSED-LOOP OUTPUT IMPEDANCE
vs FREQUENCY
6
1W Internal
5
Power Limit
4
Single-Channel
RL = 100Ω
3
2
1
0
–1
–2
–3
–4
–5
–6
–200 –150 –100 –50
0
109
100
RL = 25Ω
RL = 50Ω
1/2
OPA2822
10
ZO
402Ω
1
402Ω
0.1
0.01
1W Internal
Power Limit
Single-Channel
0.001
50
100
150
0.1
200
1
100
INVERTING OVERDRIVE RECOVERY
NONINVERTING OVERDRIVE RECOVERY
8
8
4
RL = 100Ω
G = +2
See Figure 1
2
Output
Left Scale
1
0
0
–2
–1
–4
–2
Input Voltage
4
Input/Output Voltage
3
2
10
Frequency (MHz)
IO (mA)
6
105
Frequency (Hz)
Output Impedance (Ω)
103
Output Voltage
–90
–20
0
VO (V)
0
RL = 100Ω
6 RF = 604Ω
G = –1
4
Output
2
0
–2
–4
Input
–6
Input Right Scale
–8
–3
–6
–4
–8
Time (40ns/div)
8
See Figure 2
Time (40ns/div)
OPA2822
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SBOS188E
Open-Loop Phase (30°/div)
120
Open-Loop Gain (dB)
Common-Mode Rejection Ratio (dB)
Power-Supply Rejection Ratio (dB)
120
TYPICAL CHARACTERISTICS: VS = ±6V (Cont.)
TA = +25°C, G = +2, RF = 402Ω, and RL = 100Ω, unless otherwise noted.
SETTLING TIME
VIDEO DIFFERENTIAL GAIN/DIFFERENTIAL PHASE
0.25
RL = 100Ω
VO = 2V step
G = +2
0.25
Differential Phase (°)
0.15
0.10
0.05
0
–0.05
–0.10
Differential Gain (%)
0.20
–0.15
0.15
dP
0.10
0.05
–0.20
dG
See Figure 1
0
–0.25
0
5
10
15
20
25
30
35
40
45
50
55 60
1
3
6
7
SUPPLY AND OUTPUT CURRENT
vs TEMPERATURE
0
10x Input Offset Current
–0.5
–5
Input Bias Current
–1
–25
0
25
50
75
100
Output Current (25mA/div)
Input Bias and Offset Current (µA)
5
0
12
Sourcing Output Current
Left Scale
225
200
175
9
Supply Current
(both channels)
Right Scale
150
Sinking Output
Current
Left Scale
125
8
7
Current Limited Output
–10
125
100
6
–50
–25
0
25
50
75
100
125
Ambient Temperature (°C)
COMMON-MODE AND DIFFERENTIAL
INPUT IMPEDANCE
COMMON-MODE INPUT RANGE AND
OUTPUT SWING vs SUPPLY VOLTAGE
Input Impedance Magnitude 20Log (Ω)
6
Positive Input
and Output
2
0
–2
Negative Input
and Output
–4
11
10
Ambient Temperature (°C)
4
8
250
Input Offset Voltage
Voltage Range (V)
5
TYPICAL DC DRIFT OVER TEMPERATURE
0.5
–6
4
Video Loads
10
–50
2
Time (ns)
1
Input Offset Voltage (mV)
0.20
107
Common-Mode
106
105
Differential
104
103
102
±2
±3
±4
±5
±6
103
105
106
107
108
Frequency (Hz)
Supply Voltage (±V)
OPA2822
SBOS188E
104
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9
Supply Current (1mA/div)
Percent of Final Value (%)
0.30
TYPICAL CHARACTERISTICS: VS = ±6V
TA = +25°C, Differential Gain = 2, RF = 604Ω, and RL = 400Ω, unless otherwise noted.
DIFFERENTIAL PERFORMANCE
TEST CIRCUIT
DIFFERENTIAL SMALL-SIGNAL
FREQUENCY RESPONSE
6
+6V
VO = 200mVPP
3
GD = +1
GD = +2
GD = 604Ω
RG
1/2
OPA2822
RG
VI
Normalized Gain (dB)
0
604Ω
RG
RL
604Ω
VO
–3
GD = +5
–6
–9
GD = +10
–12
–15
–18
–21
–24
1/2
OPA2822
0.5
1
10
100
500
Frequency (MHz)
–6V
DIFFERENTIAL LARGE-SIGNAL
FREQUENCY RESPONSE
DIFFERENTIAL DISTORTION vs LOAD RESISTANCE
12
6
VO = 1VPP
Gain (dB)
3
0
–3
VO = 2VPP
–6
–9
VO = 5VPP
–12
Harmonic Distortion (dBc)
GD = 2
RL = 400Ω
9
–85
VO = 200mVPP
VO = 4VPP
GD = 2
f = 1MHz
–90
3rd-Harmonic
–95
2nd-Harmonic
–100
–15
–18
–105
0.5
1
10
100
500
10
100
Frequency (MHz)
DIFFERENTIAL DISTORTION vs FREQUENCY
VO = 4VPP
GD = 2
RL = 400Ω
–75
DIFFERENTIAL DISTORTION vs OUTPUT VOLTAGE
–95
3rd-Harmonic
Harmonic Distortion (dBc)
Harmonic Distortion (dBc)
–65
–85
2nd-Harmonic
–95
–105
f = 1MHz
GD = 2
RL = 400Ω
–100
3rd-Harmonic
–105
2nd-Harmonic
–110
–115
1
10
1
Frequency (MHz)
10
1k
Load Resistance (Ω)
10
Differential Output Voltage Swing (VPP)
OPA2822
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SBOS188E
TYPICAL CHARACTERISTICS: VS = +5V
TA = +25°C, G = +2, RF = 402Ω, and RL = 100Ω, unless otherwise noted.
NONINVERTING SMALL-SIGNAL
FREQUENCY RESPONSE
6
VO = 0.1VPP
6
G = +1
RF = 0Ω
0
G = +2
–3
–6
G = +5
–9
–12
G = +10
–15
–3
–6
G = –5
–9
–12
G = –10
–15
–18
–18
–21
–21
See Figure 3
0.5
1
See Figure 4
–24
–24
10
100
500
0.5
1
10
Frequency (MHz)
1.0
0.1
0.5
Small-Signal Left Scale
0
0
–0.1
–0.5
–0.2
–1.0
–0.3
–1.5
See Figure 3
–0.4
Small-Signal Output Voltage (100mv/div)
0.2
1.5
Large-Signal Output Voltage (500mv/div)
Small-Signal Output Voltage (100mv/div)
Large-Signal Right Scale
–2.0
0.4
2.0
0.3
1.5
0.2
1.0
0.1
0.5
0
0
Small-Signal Left Scale
–0.1
–0.5
–0.2
–1.0
Large-Signal Right Scale
–0.3
–2.0
Time (20ns/div)
FREQUENCY RESPONSE vs CAPACITIVE LOAD
RECOMMENDED RS vs CAPACITIVE LOAD
Normalized Gain to Capacitive Load (dB)
1000
100
10
For Maximally Flat Response,
See Figure 12
1
100
9
CL = 10pF
6
+5V
0
VI
0.01µF
CL = 47pF
804Ω
804Ω
–3
CL = 22pF
CL = 100pF
3
1/2
OPA2822
RS
VO
CL
1kΩ
402Ω
–6
1kΩ is optional.
402Ω
–9
0.01µF
–12
1
1000
10
100
500
Frequency (MHz)
Capacitive Load (pF)
OPA2822
SBOS188E
–1.5
See Figure 4
–0.4
Time (20ns/div)
10
500
INVERTING PULSE RESPONSE
2.0
0.3
100
Frequency (MHz)
NONINVERTING PULSE RESPONSE
0.4
Input Impedance Magnitude 20Log (Ω)
G = –2
0
Normalized Gain (dB)
Normalized Gain (dB)
3
G = –1
VO = 0.1VPP
RF = 604Ω
3
Large-Signal Output Voltage (500mv/div)
9
INVERTING SMALL-SIGNAL
FREQUENCY RESPONSE
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11
TYPICAL CHARACTERISTICS: VS = +5V (Cont.)
TA = +25°C, G = +2, RF = 402Ω, and RL = 100Ω, unless otherwise noted.
HARMONIC DISTORTION vs LOAD RESISTANCE
HARMONIC DISTORTION vs FREQUENCY
–60
VO = 2VPP
f = 1MHz
–80
VO = 2VPP
RL = 200Ω
Harmonic Distortion (dBc)
2nd-Harmonic
–85
–90
–95
3rd-Harmonic
–100
–70
2nd-Harmonic
–80
–90
3rd-Harmonic
See Figure 3
See Figure 3
–105
–100
100
1
10
Load Resistance (Ω)
Frequency (MHz)
HARMONIC DISTORTION vs OUTPUT VOLTAGE
2-TONE, 3rd-ORDER
INTERMODULATION INTERCEPT
50
RL = 200Ω
f = 1MHz
2nd-Harmonic
45
Intercept Point (+dBm)
Harmonic Distortion (dBc)
–85
1k
–90
–95
3rd-Harmonic
–100
+5V
40
35
PI
0.1µF
57.6Ω
50Ω
1/2
OPA2822
PO
50Ω
402Ω
25
402Ω
0.1µF
20
–105
0.1
1
1
10
Frequency (MHz)
TYPICAL DC DRIFT OVER TEMPERATURE
SUPPLY AND OUTPUT CURRENT
vs TEMPERATURE
Input Offset Voltage
0
0
10x Input Offset Current
–0.5
–5
Input Bias Current
–1
–25
0
25
50
75
100
Output Current (25mA/div)
5
Input Bias and Offset Current (µA)
0.5
20
200
10
–50
10
Output Voltage Swing (VPP)
1
Input Offset Voltage (mV)
804Ω
30
See Figure 3
–10
125
12
11
Sourcing Output Current
Left Scale
175
Supply Current
(both channels)
Right Scale
150
10
9
8
125
Sinking Output Current
Left Scale
7
Current Limited Output
100
–50
–25
0
25
50
75
100
6
125
Ambient Temperature (°C)
Ambient Temperature (°C)
12
804Ω
Supply Current (1mA/div)
Harmonic Distortion (dBc)
–75
OPA2822
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SBOS188E
TYPICAL CHARACTERISTICS: VS = +5V
TA = +25°C, Differential Gain = +2, RF = 604Ω, and RL = 400Ω, unless otherwise noted.
DIFFERENTIAL PERFORMANCE
TEST CIRCUIT
DIFFERENTIAL SMALL-SIGNAL
FREQUENCY RESPONSE
6
+5V
0.01µF
VI
0.01µF
1/2
OPA2822
RG
0
GD = 604Ω
RG
Normalized Gain (dB)
+2.5V
604Ω
RL
604Ω
RG
GD = +1
VO = 200mVPP
RL = 400Ω
3
VO
GD = +2
–3
–6
GD = +5
–9
–12
GD = +10
–15
–18
–21
–24
1/2
OPA2822
0.5
+2.5V
10
100
500
Frequency (MHz)
DIFFERENTIAL LARGE-SIGNAL
FREQUENCY RESPONSE
DIFFERENTIAL DISTORTION vs LOAD RESISTANCE
–85
12
VO = 4VPP
GD = 2
f = 1MHz
6
Harmonic Distortion (dBc)
VO = 200mVPP
9
VO = 1VPP
3
Gain (dB)
1
0
–3
VO = 2VPP
–6
–9
VO = 5VPP
–12
–90
3rd-Harmonic
–95
2nd-Harmonic
–100
–15
–105
–18
0.5
1
10
100
10
500
100
DIFFERENTIAL DISTORTION vs FREQUENCY
DIFFERENTIAL DISTORTION vs OUTPUT VOLTAGE
–55
–95
f = 1MHz
VO = 2VPP
–65
Harmonic Distortion (dBc)
Harmonic Distortion (dBc)
1k
Resistance (Ω)
Frequency (MHz)
–75
3rd-Harmonic
–85
–95
2nd-Harmonic
–105
–100
–105
2nd-Harmonic
–110
3rd-Harmonic
–115
–115
1
10
1
Frequency (MHz)
OPA2822
SBOS188E
10
Output Voltage Swing (VPP)
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13
APPLICATIONS INFORMATION
WIDEBAND NONINVERTING OPERATION
The OPA2822 provides a unique combination of features in
a wideband dual, unity-gain stable, voltage-feedback amplifier to support the extremely high dynamic range requirements of emerging communications technologies. Combining low 2nV/√Hz input voltage noise with harmonic distortion
performance that can exceed 100dBc SFDR through 2MHz,
the OPA2822 provides the highest dynamic range input
interface for emerging high speed 14-bit (and higher) converters. To achieve this level of performance, careful attention to circuit design and board layout is required.
Figure 1 shows the gain of +2 configuration used as the basis
for the Electrical Characteristics table and most of the Typical
Characteristics at ±6V operation. While the characteristics are
given using split ±6V supplies, most of the electrical and typical
characteristics also apply to a single-supply +12V design where
the input and output operating voltages are centered at the
midpoint of the +12V supply. Operation at ±5V will very nearly
match that shown for the ±6V operating point. Most of the
reference curves were characterized using signal sources with
50Ω driving impedance, and with measurement equipment
presenting a 50Ω load impedance. In Figure 1, the 50Ω shunt
resistor at the VI terminal matches the source impedance of the
test signal generator, while the 50Ω series resistor at the VO
terminal provides a matching resistor for the measurement
equipment load. Generally, data sheet voltage swing specifications are at the output pin (VO in Figure 1), while output power
(dBm) specifications are at the matched 50Ω load. The total
100Ω load at the output, combined with the total 804Ω total
feedback network load for the noninverting configuration of
Figure 1, presents the OPA2822 with an effective output load of
89Ω. While this is a good load value for frequency response
measurements, distortion will improve rapidly with lighter output
loads. Keeping the same feedback network and increasing the
load to 200Ω will result in a total load of 160Ω for the distortion
performance reported in the Electrical Characteristics table.
For higher gains, the feedback resistor (RF) was held at 402Ω
and the gain resistor (RG) adjusted to develop the Typical
Characteristics.
Voltage-feedback op amps, unlike current-feedback designs,
can use a wide range of resistor values to set their gains. A lownoise part like the OPA2822 will deliver low total output noise
only if the resistor values are kept relatively low. For the circuit
of Figure 1, the resistors contribute an input-referred voltage
noise component of 1.8nV/√Hz, which is approaching the value
of the amplifier’s intrinsic 2nV/√Hz. For a more complete
description of the feedback network’s impact on noise, see the
Setting Resistor Values to Minimize Noise section later in this
data sheet. In general, the parallel combination of RF and RG
should be < 300Ω to retain the low-noise performance of the
OPA2822. However, setting these values too low can impair
distortion performance due to output loading, as shown in the
distortion versus load data in the Typical Characteristics.
WIDEBAND INVERTING OPERATION
Operating the OPA2822 as an inverting amplifier has several
benefits and is particularly appropriate as part of the hybrid
design in an xDSL receiver application. Figure 2 shows the
inverting gain of –1 circuit used as the basis of the inverting
mode Typical Characteristics.
+5V
+VS
0.1µF
0.1µF
50Ω Source
RG
604Ω
RS
309Ω
6.8µF
+
1/2
OPA2822
VO
VI
0.1µF
+5V
+VS
6.8µF
+
6.8µF
FIGURE 2. Inverting G = –1 Specification and Test
Circuit.
VI
1/2
OPA2822
VO
50Ω
50Ω Load
RF
402Ω
RG
402Ω
0.1µF
+
6.8µF
–VS
–5V
FIGURE 1. Noninverting G = +2 Specification and Test
Circuit.
14
+
–VS
–5V
50Ω Source
50Ω
50Ω Load
RF
604Ω
RM
54.9Ω
0.1µF
50Ω
In the inverting case, only the RF element of the feedback
network appears as part of the total output load in parallel
with the actual load. For the 100Ω load used in the Typical
Characteristics, this gives an effective load of 86Ω in this
inverting configuration. Gain resistor RG is set to achieve the
desired inverting gain (in this case 604Ω for a gain of –1),
while an additional input matching resistor (RM) can be used
to set the total input impedance equal to the source if
desired. In this case, RM = 54.9Ω in parallel with the 604Ω
gain setting resistor yields a matched input impedance of
50Ω. RM is needed only when the input must be matched to
a source impedance, as in the characterization testing done
using the circuit of Figure 2.
OPA2822
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SBOS188E
To take full advantage of the OPA2822’s excellent DC input
accuracy, the total DC impedance seen at of each of the
input terminals must be matched to get bias current cancellation. For the circuit of Figure 2, this requires the grounded
309Ω resistor on the noninverting input. The calculation for
this resistor value assumes a DC-coupled 50Ω source
impedance along with RG and RM. While this resistor will
provide cancellation for the input bias current, it must be
well decoupled (0.1µF in Figure 2) to filter the noise contribution of the resistor itself and of the amplifier’s input
current noise.
As the required RG resistor approaches 50Ω at higher gains,
the bandwidth for the circuit in Figure 2 will far exceed the
bandwidth at the same gain magnitude for the noninverting
circuit of Figure 1. This occurs due to the lower noise gain for
the circuit of Figure 2 when the 50Ω source impedance is
included in the analysis. For example, at a signal gain of
–12 (RG = 50Ω, RM = open, RF = 604Ω) the noise gain for the
circuit of Figure 2 will be 1 + 604Ω/(50Ω + 50Ω) = 7, due to
the addition of the 50Ω source in the noise gain equation.
This will give considerably higher bandwidth than the noninverting gain of +12.
SINGLE-SUPPLY NONINVERTING OPERATION
The key requirement of broadband single-supply operation is
to maintain input and output signal swings within the usable
voltage range at both input and output. The circuit of Figure 3
establishes an input midpoint bias using a simple resistive
divider from the +5V supply (two 804Ω resistors). These two
resistors are selected to provide DC bias current cancellation
because their parallel combination matches the DC impedance looking out of the inverting node, which equals RF. The
gain setting resistor is not part of the DC impedance looking
out of the inverting node, due to the blocking capacitor in
series with it. The input signal is then AC-coupled into the
midpoint voltage bias. The input impedance matching resistor
(57.6Ω) is selected for testing to give a 50Ω input match (at
high frequencies) when the parallel combination of the biasing
divider network is included. The gain resistor (RG) is ACcoupled, giving a DC gain of +1. This centers the output also
at the input midpoint bias voltage (VS/2). While this circuit is
shown using a +5V supply, this same circuit may be applied
for single-supply operation as high as +12V.
SINGLE-SUPPLY INVERTING OPERATION
For those single +5V Typical Characteristics that require
inverting gain of –1 operation, the test circuit in Figure 4 was
used.
The OPA2822 can also support single +5V operation with
its exceptional input and output voltage swing capability.
While not a rail-to-rail input/output design, both inputs and
outputs can swing to within 1.2V of either supply rail. For a
single amplifier channel, this gives a very clean 2VPP output
capability on a single +5V supply, or 4VPP output for a
differential configuration using both channels together. Figure 3 shows the AC-coupled noninverting gain of +2 used
as the basis of the Electrical Characteristics table and most
of the Typical Characteristics for single +5V supply operation.
+5V
+VS
0.1µF
RB
1.21kΩ
VS/2
0.1µF
50Ω Source
1/2
RB
1.21kΩ OPA2822
RG
0.1µF 604Ω
VO
+
6.8µF
RL
100Ω
VS/2
RF
604Ω
VI
+5V
+VS
0.1µF
VI
57.6Ω
RM
54.9Ω
0.1µF
RB
804Ω
VS/2
RB
804Ω
1/2
OPA2822
VO
+
6.8µF
FIGURE 4. AC-Coupled, G = –1, Single-Supply
Operation: Specification and Test Circuit.
RL
100Ω
VS/2
RF
402Ω
RG
402Ω
0.1µF
FIGURE 3. AC-Coupled, G = +2, Single-Supply
Operation: Specification and Test Circuit.
As with the circuit of Figure 2, the feedback resistor (RF) has
been increased to 604Ω to reduce the loading effect it has
in parallel with the 100Ω actual load. The noninverting input
is biased at VS/2 (2.5V in this case) using the two 1.21kΩ
resistors for RB. The parallel combination of these two
resistors (605Ω) provides input bias current cancellation by
matching the DC impedance looking out of the inverting
input node. The noninverting input bias is also well decoupled using the 0.1µF capacitor to both reduce both
power-supply noise and the resistor and bias current noise
at this input.
OPA2822
SBOS188E
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15
The gain resistor (RG) is set to equal the feedback resistor (RF)
at 604Ω to achieve the desired gain of –1 from VI to VO. A DC
blocking capacitor is included in series with RG to reduce the DC
gain for the noninverting input bias and offset voltages to +1.
This places the VS/2 bias voltage at the output pin and reduces
the output DC offset error terms. The signal input impedance is
matched to the 50Ω source using the additional RM resistor set
to 54.9Ω. At higher frequencies, the parallel combination of RM
and RG provides the input impedance match at 50Ω. This is
principally used for test and characterization purposes—system
applications do not necessarily require this input impedance
match, particularly if the source device is physically near the
OPA2822 and/or does not require a 50Ω input impedance
match. At higher gains, the signal source impedance will start to
materially impact the apparent noise gain (and hence, bandwidth) of the OPA2822.
ADSL RECEIVE AMPLIFIER
One of the principal applications for the OPA2822 is as a lowpower, low-noise receive amplifier in ADSL modem designs.
Applications ranging from single +5V, ±5V, and up to single +12V
supplies can be well supported by the OPA2822. For higher
supplies, consider the dual, low-noise THS6062 ADSL receive
amplifier that can support up to ±15V supplies. Figure 5 shows a
typical ADSL receiver design where the OPA2822 is used as an
inverting summing amplifier to provide both driver output signal
cancellation and receive channel gain. In the circuit of Figure 5,
the driver differential output voltage is shown as VD, while the
receiver channel output is shown as VR.
+5V
1/2
OPA2822
Driver
RS
The two sets of resistors, R1 and R2, are set to provide the
desired gain from the transformer windings for the signal
arriving on the line side of the transformer, and also to provide
nominal cancellation for the driver output signal (VD) to the
receiver output. Typically, the two RS resistors are set to
provide impedance matching through the transformer. This is
accomplished by setting RS = 0.5 • (RL/N2), where N is the
turns ratio used for the line driver design. If RS is set in this
fashion, and the actual twisted pair line shows the expected RL
impedance value, the voltage swing produced at VD will be cut
in half at the transformer input. In this case, setting R1 = 2 • R2
will achieve cancellation of the driver output signal at the
output of the receiver. Essentially, the driver output voltage
produces a current in R1 that is exactly matched by the current
pulled out of R2 due to the attenuated and inverted version of
the output signal at the transformer input. In actual practice, R1
and R2 are usually RC networks to achieve cancellation over
the frequency varying line impedance.
As the transformer turns ratio changes to support different line
driver and supply voltage combinations, the impact of receiver
amplifier noise changes. Typically, DSL systems incur a line
referred noise contribution for the receiver that can be computed for the circuit of Figure 5. For example, targeting an
overall gain of 1 from the line to the receiver output, and
picking the input resistor R2, the remaining resistors will be set
by the driver cancellation and gain requirements. With the
resistor values set, a line referred noise contribution due to the
OPA2822 can be computed. R1 will be set to 2x the value of
R2, and the feedback resistor will be set to recover the gain
loss through the transformer. Table I shows the total line
referred noise floor (in dBm/Hz) using three different values for
R2 over a range of transformer turns ratio (where the amplifier
gain is adjusted at each turns ratio).
TABLE I. Line Referred Noise dBm/Hz, Due to Receiver
Op Amp.
RF
R2
R1
1:n
RL
VD
Line
VR
R1
RS
R2
RF
1/2
OPA2822
N
R2 = 200
R2 = 500
R2 = 1000
1
1.5
2
2.5
3
3.5
4
4.5
5
–151.5
–149.1
–147.2
–145.6
–144.3
–143.2
–142.2
–141.3
–140.4
–150.2
–147.6
–145.6
–144.0
–142.7
–141.5
–140.5
–139.5
–138.7
–148.5
–145.8
–143.7
–142.1
–140.7
–139.5
–138.4
–137.5
–136.6
Table I shows that a lower transformer turns ratio results in
reduced line referred noise, and that the resistor noise will
start to degrade the noise at higher values—particularly in
going from 500Ω to 1kΩ. In general, line referred noise floor
due to the receiver channel will not be the limit to ADSL
modem performance, if it is lower than –145dBm.
–5V
FIGURE 5. Example ADSL Receiver Amplifier.
16
OPA2822
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ACTIVE FILTER APPLICATIONS
As a low-noise, low-distortion, unity-gain stable, voltagefeedback amplifier, the OPA2822 provides an ideal building
block for high-performance active filters. With two channels
available, it can be used either as a cascaded 2-stage active
filter or as a differential filter. Figure 6 shows a 6th-order
bandpass filter cascaded with two 2nd-order Sallen-Key
sections, with transmission zeroes along with a passive post
filter made up of a high-pass and a low-pass section. The first
amplifier provides a 2nd-order high-pass stage while the
second amplifier provides the 2nd-order low-pass stage.
Figure 7 shows the frequency response for this example
filter.
A differential active filter is shown in Figure 8. This circuit
shows a single-supply, 2nd-order high-pass filter with the
corner frequencies set to provide the required high-pass
function for an ADSL CPE modem application. To use this
circuit, the hybrid would be implemented as a passive summing circuit at the input to this filter. For +5V only ADSL
designs, it is preferable to implement a portion of the filtering
prior to the amplifier, thus limiting the amplitude of the
uncancelled line driver signals. This type of receiver stage
would typically then drive a low-pass filter prior to the codec
setting the high-frequency cutoff of the ADC (Analog-toDigital Converter) input signal. Figure 9 shows the frequency
response for the high-pass circuit of Figure 8.
+VS
+5V
365Ω
2.2µF
1/2
OPA2822
2kΩ
730Ω
1µF
VS
2
VI
VI
730Ω
2.2µF
1.0nF
158Ω
1.0nF
1/2
OPA2822
2.2µF
365Ω
FIGURE 8. Single-Supply, 2nd-Order High-Pass Active
Filter with Differential I/O.
180pF
2.1kΩ
1.3kΩ
VO
2kΩ
2.2pF
140Ω
2.2µF
1/2
OPA2822
+5V
225Ω
18pF
150pF
12pF
1.8nF
1/2
OPA2822
300Ω
VO
150Ω
143Ω
107Ω
66pF
–5V
FIGURE 6. 6th-Order Bandpass Filter.
10
3
0
0
–3
–6
Gain (dB)
Gain (dB)
–10
–20
–30
–9
–12
–15
–18
–40
–21
–24
–50
–27
–60
1.0E+04
1.0E+05
1.0E+06
1.0E+07
–30
1.0E+04
1.0E+08
Frequency (Hz)
1.0E+06
1.0E+07
Frequency (Hz)
FIGURE 7. Frequency Response for the Filter in Figure 6.
FIGURE 9. Frequency Response for the Filter in Figure 8.
OPA2822
SBOS188E
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17
HIGH DYNAMIC RANGE ADC DRIVER
Numerous circuit approaches exist to provide the last stage
of amplification before the ADC in high-performance applications. For very high dynamic range applications where the
signal channel can be AC-coupled, the circuit shown in
Figure 10 provides exceptional performance. Most very high
performance ADCs > 12-bit performance require differential
inputs to achieve the dynamic range. The circuit of Figure 10
converts a single-ended source to differential via a 1:2 turns
ratio transformer, which then drives the inverting gain setting
resistors (RG). These resistors are fixed at 100Ω to provide
input matching to a 50Ω source on the transformer primary
side. The gain can then be adjusted by setting the feedback
resistor values. For best performance, this circuit operates
with a ground centered output on ±5V supplies, although a
+12V supply can also provide excellent results. Since most
high-performance converters operate on a single +5V supply, the output is level shifted through an AC blocking
capacitor to the common-mode input voltage (VCM) for the
converter input, and then low-pass filtered prior to the input
of the converter. This circuit is intended for inputs from 10kHz
to 10MHz, so the output high-pass corner is set to 1.6kHz,
while the low-pass cutoff is set to 20MHz. These are example
cutoff frequencies; the actual filtering requirements would be
set by the specific application.
transformer) and then RF is set to get the desired overall
gain. With these constraints (and 0Ω on the noninverting
inputs), the noise figure equation simplifies considerably.
1 1 2 1
2 en + / n + (in nR S )2
2 α
2
4
NF = 10 log2 + +
(1)
α
kTRS
where RG = 1/2 n2RS
n = Transformer Turns Ratio
α = RF/RG
en = Op Amp Input Voltage Noise
in = Inverting Input Current Noise
kT = 4E – 21J[T = 290°K]
Gain (dB) = 20 log[nα]
TABLE II. Noise Figure versus Gain with n = 2 Transformer.
TOTAL GAIN
(V/V)
LOG GAIN
(dB)
REQUIRED
AMPLIFIER GAIN
(α)
NOISE FIGURE
(dB)
4
5
6
7
8
9
10
12.0
14.0
15.6
16.9
18.1
19.1
20.0
2
2.5
3
3.5
4
4.5
5
11.2
10.4
9.9
9.5
9.1
8.9
8.6
The 1:2 turns ratio transformer also provides an improvement
in input referred noise figure. Equation 1 shows the Noise
Figure (NF) calculation for this circuit, where RG has been
constrained to provide an input match to RS (through the
+5V
+5V
1/2
OPA2822
RS = 50Ω
RG
100Ω
VI
0.1µF
80Ω
VI
100pF
RF
1kΩ
1:2
500Ω
VO
RG
100Ω
VI
14-Bit
ADC
RF
1µF
Noise
Figure
Defined
Here
VO
VCM
1kΩ
0.1µF
=2
RF
1/2
OPA2822
RG
80Ω
VI
100pF
–5V
FIGURE 10. Single-Ended to Differential High Dynamic Range ADC Driver.
18
OPA2822
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SBOS188E
DESIGN-IN TOOLS
ENI
DEMONSTRATION BOARDS
Two printed circuit boards (PCBs) are available to assist in
the initial evaluation of circuit performance using the OPA2822
in its two package options. Both of these are offered free of
charge as unpopulated PCBs, delivered with a user’s guide.
The summary information for these fixtures is shown in
Table III.
1/2
OPA2822
RS
ERS
RF
√ 4kTRS
TABLE III. Demonstration Fixtures by Package.
PRODUCT
PACKAGE
ORDERING
NUMBER
LITERATURE
NUMBER
OPA2822U
OPA2822E
SO-8
MSOP-8
DEM-OPA-SO-2A
DEM-OPA-MSOP-2A
SBOU003
SBOU004
EO
IBN
4kT
RG
RG
IBI
√ 4kTRF
4kT = 1.6E –20J
at 290°K
FIGURE 11. Op Amp Noise Analysis Model.
The demonstration fixtures can be requested at the Texas
Instruments web site (www.ti.com) through the OPA2822
product folder.
Dividing this expression by the noise gain (NG = 1 = RF/RG)
will give the total equivalent spot noise voltage referred to the
noninverting input, as shown in Equation 3:
MACROMODELS AND APPLICATIONS SUPPORT
Computer simulation of circuit performance using SPICE is
often a quick way to analyze the performance of the OPA2822
in its intended application. This is particularly true for video
and RF amplifier circuits where parasitic capacitance and
inductance can play a major role in circuit performance. A
SPICE model for the OPA2822 is available through the TI
web site (www.ti.com). These models do a good job of
predicting small-signal AC and transient performance under
a wide variety of operating conditions. They do not do as well
in predicting the harmonic distortion characteristics. These
models do not attempt to distinguish between the package
types in their small-signal AC performance.
OPERATING SUGGESTIONS
SETTING RESISTOR VALUES TO MINIMIZE NOISE
Getting the full advantage of the OPA2822’s low input noise
requires careful attention to the external gain setting and DC
biasing networks. The feedback resistor is part of the overall
output load (which can begin to degrade distortion if set too
low). With this in mind, a good starting point for design is to
select the feedback resistor as low as possible (consistent
with loading distortion concerns), then continue with the
design, and set the other resistors as needed. To retain full
performance, setting the feedback resistor in the range of
200Ω to 750Ω can provide a good start to the design.
Figure 11 shows the full output noise analysis model for any
op amp.
The total output spot noise voltage can be computed as the
square root of the sum of all squared output noise voltage
terms. Equation 2 shows the general form of this output noise
voltage expression using the terms shown in Figure 11.
EO =
(E
NI
2
)
+ (IBN RS )2 + 4kTRS NG2 + (IBI RF )2 +
4kTRF
NG
I R 2 4kTRF
EN = ENI 2 + (IBN RS )2 + 4kTRS + BI F +
(3)
NG
NG
Inserting high resistor values into Equation 3 can quickly
dominate the total equivalent input referred voltage noise. A
250Ω source impedance on the noninverting input will add as
much noise as the amplifier itself. If the noninverting input is
a DC bias path (as in inverting or in some single-supply
applications), it is critical to include a noise shunting capacitor with that resistor to limit the added noise impact of those
resistors (see the example in Figure 2).
FREQUENCY RESPONSE CONTROL
Voltage-feedback op amps such as the OPA2822 exhibit
decreasing closed-loop bandwidth as the signal gain is
increased. In theory, this relationship is described by the
Gain Bandwidth Product (GBP) shown in the Electrical Characteristics. Ideally, dividing GBP by the noninverting signal
gain (also called the Noise Gain, NG) will predict the closedloop bandwidth. In practice, this principle holds true only
when the phase margin approaches 90°, as it does in higher
gain configurations. At low gains, most high-speed amplifiers
will show a more complex response with lower phase margin
and higher bandwidth than predicted by the GBP. The
OPA2822 is compensated to give a slightly peaked frequency response at a gain of +2 (see the circuit in Figure 1).
The 200MHz typical bandwidth at a gain of +2 far exceeds
that predicted by dividing the GBP of 240MHz by a gain of 2.
The bandwidth predicted by the GBP is more closely correct
as the gain increases. As shown in the Typical Characteristics, at a gain of +10, the –3dB bandwidth of 24MHz matches
that predicted by dividing the GBP by 10.
(2)
OPA2822
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19
Inverting operation offers some interesting opportunities to
increase the available signal bandwidth. When the source
impedance is matched by the gain resistor (Figure 10 for
example), the signal gain is (1 + RF/RG) while the noise gain
is (1 + RF/2RG). This reduces the noise gain almost by half,
extending the signal bandwidth and increasing the loop gain.
For instance, setting RF = 500Ω in Figure 10 will give a signal
gain for the amplifier of 5V/V. However, including the 50Ω
source impedance reflected through the 1:2 transformer will
give an additional 100Ω source impedance for the noise gain
analysis for each of the amplifiers. This reduces the noise gain
to 1 + 500Ω/200Ω = 3.5V/V and results in an amplifier
bandwidth of at least 240MHz/3.5 = 68MHz.
The resistor across the two inputs, RNG, can be used to
increase the noise gain while retaining the desired signal
gain. This can be used either to improve flatness at low gains
or to reduce the required value of RS in capacitive load
driving applications. This circuit was used with RNG adjusted
to produce the gain flatness curve in the Typical Characteristics. As shown in that curve, an RNG of 452Ω will give an NG
of 3 giving exceptional frequency response flatness at a
signal gain of +2. Equation 4 shows the calculation for RNG
given a target noise gain (NG) and signal gain (G):
RNG =
RF + RSG
NG − G
(4)
where RS = Total Source Impedance on the Noninverting
Input [25Ω in Figure 12]
One of the most demanding and yet very common load
conditions for an op amp is capacitive loading. Often, the
capacitive load is the input of an ADC, including additional
external capacitance which may be recommended to improve ADC linearity. A high-speed, high open-loop gain
amplifier like the OPA2822 can be very susceptible to decreased stability and closed-loop frequency response peaking when a capacitive load is placed directly on the output
pin. When the amplifier’s open-loop output resistance is
considered, this capacitive load introduces an additional pole
in the signal path that can decrease the phase margin.
Several external solutions to this problem have been suggested. When the primary considerations are frequency
response flatness with low noise and distortion, the simplest
and most effective solution is to isolate the capacitive load
from the feedback loop by inserting a series isolation resistor
between the amplifier output and the capacitive load. This
does not eliminate the pole from the loop response, but
instead shifts it and adds a zero at a higher frequency. The
additional zero acts to cancel the phase lag from the capacitive load pole, thus increasing the phase margin and improving stability.
G = Signal Gain [1 + (RF/RG)]
NG = Noise Gain Target
Using this technique to get initial frequency response flatness will significantly reduce the required series resistor
value to get a flat response at the capacitive load. Using the
best-case noise gain of 3 with a signal gain of 2 allows the
required RS to be reduced, as shown in Figure 13. Here, the
required RS versus Capacitive Load is replotted along with
data from the Typical Characteristics. This demonstrates that
the use of RNG = 452Ω across the inputs results in much
lower required RS values to achieve a flat response.
100
NG = 2, RNG = ∞
RS (Ω)
DRIVING CAPACITIVE LOADS
10
NG = 3, RNG = 452Ω
The Typical Characteristics show the recommended RS versus capacitive load and the resulting frequency response at
the load. For the OPA2822 operating at a gain of +2, the
frequency response at the output pin is already slightly
peaked without the capacitive load, requiring relatively high
values of RS to flatten the response at the load. One way to
reduce the required RS value is to use the noise gain
adjustment circuit of Figure 12.
1
10
100
1000
Capacitive Load (pF)
FIGURE 13. Required RS vs Noise Gain.
DISTORTION PERFORMANCE
50Ω Source
50Ω
RNG
The OPA2822 is capable of delivering exceptionally low
distortion through approximately 5MHz signal frequency.
While principally intended to provide very low noise and
distortion through the maximum ADSL frequency of 1.1MHz,
the OPA2822 in a differential configuration can deliver lower
than –85dBc distortions for a 4VPP swing through 5MHz. For
applications requiring extremely low distortion through higher
frequencies, consider higher slew rate amplifiers such as the
OPA687 or OPA2681.
1/2
OPA2822
RF
402Ω
RG
402Ω
FIGURE 12. Noise Gain Tuning for Noninverting Circuit.
20
OPA2822
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SBOS188E
As the Typical Characteristics show, until the fundamental
signal reaches very high frequencies or power levels, the
limit to SFDR will be 2nd-harmonic distortion rather than the
negligible 3rd-harmonic component. Focusing then on the
second harmonic, increasing the load impedance improves
distortion directly. However, operating differentially offers
the most significant improvement in even-order distortion
terms. For example, the Electrical Characteristics show that
a single channel of the OPA2822, delivering 2VPP at 1MHz
into a 200Ω load, will typically show a 2nd-harmonic product
at –92dBc versus the 3rd-harmonic at –102dBc. Changing
the configuration to a differential driver where each output
still drives 2VPP results in a 4VPP total differential output into
a 400Ω differential load, giving the same single-ended load
of 200Ω for each amplifier. This configuration drops the
2nd-harmonic to –103dBc and the 3rd-harmonic to approximately –105dBc—an overall dynamic range improvement
of more than 10dB.
For general distortion analysis, remember that the total
loading on the amplifier includes the feedback network; in the
noninverting configuration, this is the sum of RF + RG, while
in the inverting configuration this additional loading is simply
RF. Increasing the output voltage swing increases the harmonic distortion directly. A 6dB increase in the output swing
will generally increase the 2nd-harmonic 12dB and the 3rdharmonic 18dB. Increasing the signal gain will also generally
increase both the 2nd- and 3rd-harmonics because the loop
gain decreases at higher gains. Again, a 6dB increase in
voltage gain will increase the 2nd-harmonic distortion by
approximately 6dB. The distortion characteristic curves for
the OPA2822 show little change in the 3rd-harmonic distortion versus gain. Finally, the overall distortion generally
increases as the fundamental frequency increases due to the
rolloff in the loop gain with frequency. Conversely, the distortion will improve going to lower frequencies, down to the
dominant open-loop pole at approximately 50kHz. This will
give essentially unmeasurable levels of harmonic distortion
in the audio band.
The OPA2822 exhibits an extremely low 3rd-order harmonic
distortion. This also gives exceptionally good 2-tone 3rdorder intermodulation intercept as shown in the Typical
Characteristics. This intercept curve is defined at the 50Ω
load when driven through a 50Ω matching resistor to allow
direct comparisons to RF MMIC devices. This network attenuates the voltage swing from the output pin to the load by
6dB. If the OPA2822 drives directly into the input of a highimpedance device, such as an ADC, this 6dB attenuation
does not occur. Under these conditions, the intercept will
improve by at least 6dBm. The intercept is used to predict the
intermodulation spurs for two closely spaced frequencies. If
the two test frequencies, f1 and f2, are specified in terms of
average and delta frequency, fO = (f1 + f2)/2 and ∆F = |f2 – f1|,
the two, 3rd-order, close-in spurious tones will appear at
fO ± 3 • ∆F. The difference between two equal test-tone power
levels and the spurious intermodulation power levels is given
by ∆dBc = 2 • (IM3 – PO), where IM3 is the intercept taken
from the Typical Specification and PO is the power level in
dBm at the 50Ω load for either one of the two closely spaced
test frequencies. For example, at 1MHz in a gain of +2
configuration, the OPA2822 exhibits an intercept of 57dBm
at a matched 50Ω load. If the full envelope of the two
frequencies needs to be 2VPP, each tone will be set to 4dBm.
The 3rd-order intermodulation spurious tones will then be
2 • (57 – 4) = 106dBc below the test-tone power level
(–102dBm). If this same 2VPP 2-tone envelope were delivered directly into the input of an ADC without the matching
loss or loading of the 50Ω network, the intercept would
increase to at least 63dBm. With the same signal and gain
conditions but now driving directly into a light load, the
spurious tones would then be at least 2 • (63 – 4) = 118dBc
below the test-tone power levels.
DC ACCURACY AND OFFSET CONTROL
The OPA2822 can provide excellent DC signal accuracy due
to its high open-loop gain, high common-mode rejection, high
power-supply rejection, and low input offset voltage and bias
current offset errors. To take full advantage of the low input
offset voltage (±1.2mV maximum at 25°C), careful attention
to input bias current cancellation is also required. The highspeed input stage for the OPA2822 has relatively high input
bias current (8µA typical into the pins) but with a very close
match between the two input currents, typically 100nA input
offset current. The total output offset voltage may be reduced
considerably by matching the source impedances looking out
of the two inputs. For example, one way to add bias current
cancellation to the circuit of Figure 1 would be to insert a
175Ω series resistor into the noninverting input from the 50Ω
terminating resistor. If the 50Ω source resistor is DC coupled,
this will increase the source impedance for the noninverting
input bias current to 200Ω. Since this is now equal to the
impedance looking out of the inverting input (RF || RG), the
circuit will cancel the bias current effects, leaving only the
offset current times the feedback resistor as a residual DC
error term at the output. Using a 402Ω feedback resistor, the
output DC error due to the input bias currents will now be less
than 0.7µA • 402Ω = 0.28mV over the full temperature range.
This is significantly lower than the contribution due to the
input offset voltage. At a gain of +2, the maximum input offset
voltage is 1.5mV, giving a total maximum output offset of
(±3mV ± 0.28mV) = ±3.3mV over the –40°C to +85°C
temperature range (for the circuit of Figure 1, including the
additional 175Ω resistor at the noninverting input).
THERMAL ANALYSIS
The OPA2822 will not require heatsinking or airflow under
most operating conditions. Maximum desired junction temperature will limit the maximum allowed internal power dissipation as described below. In no case should the maximum
junction temperature be allowed to exceed +150°C.
Operating junction temperature (TJ) is given by TA + PDθJA.
The total internal power dissipation (PD) is the sum of the
quiescent power (PDO) and additional power dissipated in the
output stage (PDL) to deliver load power. Quiescent power is
simply the specified no-load supply current times the total
supply voltage across the part. PDL will depend on the required
OPA2822
SBOS188E
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21
output signal and load but would, for a grounded resistive load,
be at a maximum when the output is fixed at a voltage equal
to half of either supply voltage (assuming equal bipolar supplies). Under this condition PDL = VS2/(4 • RL) where RL
includes feedback network loading.
Note that it is the power dissipated in the output stage and not in
the load that determines internal power dissipation. As a worstcase example, compute the maximum TJ for the OPA2822E with
both channels operating at AV = +2, RL = 100Ω, RF = 400Ω,
±VS = ±5V, and at the specified maximum TA = 85°C.
PD = 10V • 11.4mA + 2 • (52)/(4 • (100 || 804)) = 255mW
Maximum TJ = 85°C + 0.255W • 150°C/W = 123°C
This calculation represents a worst-case combination of
conditions to reach a maximum possible operating junction
temperature. Under most operating conditions, the junction
temperature will be far lower than the 123°C calculated here.
The output current is limited in the OPA2822 to protect
against damage under short-circuit conditions. This currentlimited output of approximately 220mA exceeds the rated
typical output current of 150mA. The typical and minimum
output current limits are set for linear operation while the
maximum output shown in the Typical Characteristics is
nonlinear limited performance.
BOARD LAYOUT
Achieving optimum performance with a high-frequency amplifier like the OPA2822 requires careful attention to board
layout parasitics and external component types. Recommendations that will optimize performance include:
a) Minimize parasitic capacitance to any AC ground for all
of the signal I/O pins. Parasitic capacitance on the output and
inverting input pins can cause instability: on the noninverting
input, it can react with the source impedance to cause
unintentional bandlimiting. To reduce unwanted capacitance,
a window around the signal I/O pins should be opened in all
of the ground and power planes around those pins. Otherwise, ground and power planes should be unbroken elsewhere on the board.
b) Minimize the distance (< 0.25") from the power-supply
pins to high-frequency 0.1µF decoupling capacitors. At the
device pins, the ground and power plane layout should not
be in close proximity to the signal I/O pins. Avoid narrow
power and ground traces to minimize inductance between
the device pins and the decoupling capacitors. The primary
power-supply connections (on pins 4 and 8) should always
be decoupled with these capacitors. Larger (2.2µF to 6.8µF)
decoupling capacitors, effective at lower frequencies, should
also be used on the main supply pins. These may be placed
somewhat farther from the device and may be shared among
several devices in the same area of the PCB.
22
c) Careful selection and placement of external components will preserve the high-frequency performance of
the OPA2822. Resistors should be a very low reactance
type. Surface-mount resistors work best and allow a tighter
overall layout. Metal film and carbon composition axially
leaded resistors can also provide good high-frequency performance. Again, keep their leads and PCB trace length as
short as possible. Never use wire-wound type resistors in a
high-frequency application. Since the output pin and inverting input pin are the most sensitive to parasitic capacitance,
always position the feedback and series output resistor, if
any, as close as possible to the output pin. Other network
components, such as noninverting input termination resistors, should also be placed close to the package. Even with
a low parasitic capacitance shunting the external resistors,
excessively high resistor values can create significant time
constants that can degrade performance. Good axial metal
film or surface-mount resistors have approximately 0.2pF in
shunt with the resistor. For resistor values > 1.5kΩ, this
parasitic capacitance can add a pole and/or zero below
500MHz that can effect circuit operation. Keep resistor values as low as possible consistent with parasitic load, distortion, and noise considerations. The 402Ω feedback used in
the Typical Characteristics is a good starting point for design.
d) Connections to other wideband devices on the board may
be made with short direct traces or through onboard transmission
lines. For short connections, consider the trace and the input to
the next device as a lumped capacitive load. Relatively wide
traces (50mils to 100mils) should be used, preferably with ground
and power planes opened up around them. Estimate the total
capacitive load and set RS from the plot of recommended RS
versus capacitive load. If a long trace is required, and the 6dB
signal loss intrinsic to a doubly-terminated transmission line is
acceptable, implement a matched impedance transmission line
using microstrip or stripline techniques (consult an ECL design
handbook for microstrip and stripline layout techniques). A 50Ω
environment is normally not necessary onboard, and in fact a
higher impedance environment will improve distortion as shown
in the distortion versus load plots. With a characteristic board
trace impedance defined based on board material and trace
dimensions, a matching series resistor into the trace from the
output of the OPA2822 is used as well as a terminating shunt
resistor at the input of the destination device. Remember also that
the terminating impedance will be the parallel combination of the
shunt resistor and the input impedance of the destination device;
this total effective impedance should be set to match the trace
impedance. Multiple destination devices are best handled as
separate transmission lines, each with their own series and shunt
terminations. If the 6dB attenuation of a doubly-terminated transmission line is unacceptable, a long trace can be series-terminated at the source end only. Treat the trace as a capacitive load
in this case and set the series resistor value as shown in the plot
of RS vs Capacitive Load. This will not preserve signal integrity as
OPA2822
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SBOS188E
well as a doubly-terminated line. If the input impedance of the
destination device is low, there will be some signal attenuation
due to the voltage divider formed by the series output into the
terminating impedance.
+V CC
External
Pin
e) Socketing a high-speed part like the OPA2822 is not
recommended. The additional lead length and pin-to-pin capacitance introduced by the socket can create an extremely
troublesome parasitic network, which can make it almost impossible to achieve a smooth, stable frequency response. Best
results are obtained by soldering the OPA2822 onto the board.
INPUT AND ESD PROTECTION
The OPA2822 is built using a very high-speed complementary
bipolar process. The internal junction breakdown voltages are
relatively low due to these very small geometry devices. These
breakdowns are reflected in the Absolute Maximum Rating
table. All device pins are protected with internal ESD protection
diodes to the power supplies, as shown in Figure 14.
–V CC
FIGURE 14. Internl ESD Protection.
These diodes provide moderate protection to input overdrive
voltages above the supplies as well. The protection diodes can
typically support 30mA continuous current. Where higher currents are possible (for example, in systems with ±15V supply
parts driving into the OPA2822), current-limiting series resistors
should be added into the two inputs. Keep these resistor values
as low as possible since high values degrade both noise
performance and frequency response.
OPA2822
SBOS188E
Internal
Circuitry
www.ti.com
23
Revision History
DATE
REVISION
PAGE
SECTION
8/08
E
2
Abs Max Ratings
5/06
D
8
Typical Characteristics
19
Design-In Tools
DESCRIPTION
Changed Storage Temperature Range from −40°C to +125°C to
−65°C to +125°C.
Axis text change on, Closed-Loop Output Impedance vs Frequency.
Demonstration fixture numbers changed.
NOTE: Page numbers for previous revisions may differ from page numbers in the current version.
24
OPA2822
www.ti.com
SBOS188E
PACKAGE OPTION ADDENDUM
www.ti.com
6-Feb-2020
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
Lead/Ball Finish
MSL Peak Temp
(2)
(6)
(3)
Op Temp (°C)
Device Marking
(4/5)
OPA2822E/250
ACTIVE
VSSOP
DGK
8
250
Green (RoHS
& no Sb/Br)
NIPDAUAG
Level-2-260C-1 YEAR
-40 to 85
D22
OPA2822E/2K5
ACTIVE
VSSOP
DGK
8
2500
Green (RoHS
& no Sb/Br)
NIPDAUAG
Level-2-260C-1 YEAR
-40 to 85
D22
OPA2822U
ACTIVE
SOIC
D
8
75
Green (RoHS
& no Sb/Br)
NIPDAU
Level-2-260C-1 YEAR
-40 to 85
OPA
2822U
OPA2822U/2K5
ACTIVE
SOIC
D
8
2500
Green (RoHS
& no Sb/Br)
NIPDAU
Level-2-260C-1 YEAR
-40 to 85
OPA
2822U
OPA2822UG4
ACTIVE
SOIC
D
8
75
Green (RoHS
& no Sb/Br)
NIPDAU
Level-2-260C-1 YEAR
-40 to 85
OPA
2822U
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of
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