OPA3684IDBQT

OPA3684 OPA 368 4 SBOS241A – MAY 2002 – REVISED SEPTEMBER 2002 Low-Power, Triple Current-Feedback OPERATIONAL AMPLIFIER With Disable FEATURES q q q q q q q q q MINIMAL BANDWIDTH CHANGE VERSUS GAIN 170MHz BANDWIDTH: G = +2 > 120MHz BANDWIDTH TO GAIN > +10 LOW DISTORTION: < –82dBc at 5MHz HIGH OUTPUT CURRENT: 120mA SINGLE +5V TO +12V SUPPLY OPERATION DUAL ±2.5V TO ±6.0V SUPPLY OPERATION LOW SUPPLY CURRENT: 1.7mA/ch LOW SHUTDOWN CURRENT: 100µA/ch APPLICATIONS q q q q q q RGB LINE DRIVERS LOW-POWER BROADCAST VIDEO DRIVERS EQUALIZING FILTERS MULTICHANNEL SUMMING AMPLIFIERS PROFESSIONAL CAMERAS ADC INPUT DRIVERS DESCRIPTION The OPA3684 provides a new level of performance in low-power, wideband, current-feedback (CFB) amplifiers. This CFBPLUS amplifier among the first to use an internally closed-loop input buffer stage that enhances performance significantly over earlier lowpower CFB amplifiers. While retaining the benefits of very low power operation, this new architecture provides many of the benefits of a more ideal CFB amplifier. The closed-loop input stage buffer gives a very low and linearized impedance path at the inverting input to sense the feedback error current. This improved inverting input impedance retains exceptional bandwidth to much higher gains and improves harmonic distortion over earlier solutions limited by inverting input linearity. Beyond simple high-gain applications, the OPA3684 CFBPLUS amplifier permits the gain setting element to be set with considerable freedom from amplifier bandwidth interaction. This allows frequency response peaking elements to be added, multiple input inverting summing circuits to have greater bandwidth, and low-power line drivers to meet the demanding requirements of studio cameras and broadcast video. The output capability of the OPA3684 also sets a new mark in performance for low-power current-feedback amplifiers. Delivering a full ±4Vp-p swing on ±5V supplies, the OPA3684 also has the output current to support > ±3Vp-p into 50Ω. This minimal output headroom requirement is complemented by a similar 1.2V input stage headroom giving exceptional capability for single +5V operation. The OPA3684’s low 1.7mA/ch supply current is precisely trimmed at 25°C. This trim, along with low shift over temperature and supply voltage, gives a very robust design over a wide range of operating conditions. System power may be further reduced by using the optional disable control pin. Leaving this disable pin open, or holding it HIGH, gives normal operation. If pulled LOW, the OPA3684 supply current drops to less than 100µA/ch while the I/O pins go to a high impedance state. 1 of 3 Channels V+ BW (MHz) vs GAIN 6 3 Normalized Gain (3dB/div) G=1 G=2 0 –3 –6 –9 –12 –15 –18 –21 –24 10 G = 50 RF = 800Ω G = 100 100 MHz 200 G = 10 G = 20 G=5 + VO V– IERR RF Z(S) IERR RG Low-Power Patent Pending Amplifier Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet. PRODUCTION DATA information is current as of publication date. Products conform to specifications per the terms of Texas Instruments standard warranty. Production processing does not necessarily include testing of all parameters. Copyright © 2002, Texas Instruments Incorporated www.ti.com ABSOLUTE MAXIMUM RATINGS(1) Power Supply ............................................................................... ±6.5VDC Internal Power Dissipation ................................. See Thermal Information Differential Input Voltage .................................................................. ±1.2V Input Voltage Range ............................................................................ ±VS Storage Temperature Range: ID, IDBQ ........................ –40°C to +125°C Lead Temperature (soldering, 10s) .............................................. +300°C Junction Temperature (TJ ) ........................................................... +175°C ESD Rating: HBM ............................................................................ 1900V CDM ........................................................................... 1500V NOTE: (1) Stresses above these ratings may cause permanent damage. Exposure to absolute maximum conditions for extended periods may degrade device reliability. ELECTROSTATIC DISCHARGE SENSITIVITY This integrated circuit can be damaged by ESD. Texas Instruments recommends that all integrated circuits be handled with appropriate precautions. Failure to observe proper handling and installation procedures can cause damage. ESD damage can range from subtle performance degradation to complete device failure. Precision integrated circuits may be more susceptible to damage because very small parametric changes could cause the device not to meet its published specifications. OPA3684 RELATED PRODUCTS SINGLES OPA684 OPA691 OPA685 OPA692 DUALS OPA2684 OPA2691 — — TRIPLES — OPA3691 — OPA3692 QUADS OPA4684 — — — FEATURES Low-Power CFBplus High Slew Rate CFB > 500MHz CFB Fixed-Gain Video Buffers PACKAGE/ORDERING INFORMATION PRODUCT OPA3684 PACKAGE-LEAD SO-14 PACKAGE DESIGNATOR(1) D SPECIFIED TEMPERATURE RANGE –40°C to +85°C PACKAGE MARKING OPA3684 ORDERING NUMBER OPA3684ID OPA3684IDR OPA3684IDBQT OPA3684IDBQR TRANSPORT MEDIA, QUANTITY Rails, 58 Tape and Reel, 2500 Tape and Reel, 250 Tape and Reel, 2500 " OPA3684 " SSOP-16 " DBQ " –40°C to +85°C " OPA3684 " " " " " NOTE: (1) For the most current specifications, and package information, refer to our web site at www.ti.com. PIN CONFIGURATION Top View SO Top View SSOP DIS A DIS B DIS C +VS +Input A –Input A Output A 1 2 C 3 4 5 A 6 7 B 14 Output C 13 –Input C 12 +Input C 11 –VS 10 +Input B 9 8 –Input B Output B DIS A DIS B DIS C +VS +Input A –Input A Output A NC 1 2 C 3 4 5 A 6 7 8 B 16 Output C 15 –Input C 14 +Input C 13 –VS 12 +Input B 11 –Input B 10 Output B 9 NC 2 OPA3684 www.ti.com SBOS241A ELECTRICAL CHARACTERISTICS: VS = ±5V Boldface limits are tested at +25°C. RF = 800Ω, RL = 100Ω, and G = +2, unless otherwise noted. OPA3684ID, IDBQ TYP PARAMETER AC PERFORMANCE (see Figure 1) Small-Signal Bandwidth (VO = 0.5Vp-p) CONDITIONS G = +1, RF = 800Ω G = +2, RF = 800Ω G = +5, RF = 800Ω G = +10, RF = 800Ω G = +20, RF = 800Ω G = +2, VO = 0.5Vp-p, RF = 800Ω RF = 800Ω, VO = 0.5Vp-p G = +2, VO = 4Vp-p G = –1, VO = 4V Step G = +2,VO = 4V Step G = +2, VO = 0.5V Step G = +2, VO = 4VStep G = +2, f = 5MHz, VO = 2Vp-p RL = 100Ω RL ≥ 1kΩ RL = 100Ω RL ≥ 1kΩ f > 1MHz f > 1MHz f > 1MHz G = +2, NTSC, VO = 1.4Vp, RL = 150Ω G = +2, NTSC, VO = 1.4Vp, RL = 150Ω 2 Channels, f = 5MHz 3rd-Channel Measured VO = 0V, RL = 1kΩ VCM = 0V VCM = 0V VCM = 0V VCM = 0V VCM = 0V VCM = 0V +25°C 250 170 138 120 95 19 1.4 90 780 750 3 6.8 –67 –82 –70 –84 3.7 9.4 17 0.04 0.02 70 MIN/MAX OVER TEMPERATURE +25°C(1) 0°C to 70°C(2) –40°C to +85°C(2) UNITS MHz MHz MHz MHz MHz MHz dB MHz V/µs V/µs ns ns dBc dBc dBc dBc nV/√Hz pA/√Hz pA/√Hz % deg dB MIN/ TEST MAX LEVEL(3) typ min typ typ typ min max typ min min typ typ max max max max max max max typ typ typ C B C C C B B C B B C C B B B B B B B C C C 120 118 117 Bandwidth for 0.1dB Gain Flatness Peaking at a Gain of +1 Large-Signal Bandwidth Slew Rate Rise-and-Fall Time Harmonic Distortion 2nd-Harmonic 3rd-Harmonic Input Voltage Noise Noninverting Input Current Noise Inverting Input Current Noise Differential Gain Differential Phase All Hostile Crosstalk DC PERFORMANCE(4) Open-Loop Transimpedance Gain (ZOL) Input Offset Voltage Average Offset Voltage Drift Noninverting Input Bias Current Average Noninverting Input Bias Current Drift Inverting Input Bias Current Average Inverting Input Bias Current Drift INPUT Common-Mode Input Range(5) (CMIR) Common-Mode Rejection Ratio (CMRR) Noninverting Input Impedance Inverting Input Resistance (RI) OUTPUT Voltage Output Swing Current Output, Sourcing Current Output, Sinking Closed-Loop Output Impedance DISABLE (Disabled LOW) Power-Down Supply Current (+VS) Disable Time Enable Time Off Isolation Output Capacitance in Disable Enable Voltage Disable Voltage Control Pin Input Bias Current (DIS) POWER SUPPLY Specified Operating Voltage Maximum Operating Voltage Range Max Quiescent Current Min Quiescent Current Power-Supply Rejection Ratio (–PSRR) TEMPERATURE RANGE Specification: D, DBQ Thermal Resistance, θJA D SO-14 DBQ SSOP-16 16 4.8 675 680 14 5.9 650 660 14 6.3 575 650 –59 –66 –66 –82 4.1 11 18 –59 –65 –65 –81 4.2 12 18.5 –58 –65 –65 –81 4.4 12.5 19 355 ±1.5 ±5.0 ±5.0 ±3.9 ±12 ±17 ±3.65 53 160 155 ±4.5 ±12 ±13.5 ±25 ±18.5 ±35 ±3.65 52 153 ±4.7 ±12 ±14 ±30 ±19.5 ±40 ±3.6 52 kΩ mV µV/°C µA nA/°C µA nA°/C V dB kΩ || pF Ω V mA mA Ω µA ms ns dB pF V V µA V V mA mA dB °C °C/W °C/W min max max max max max max min min typ typ min min min typ max typ typ typ typ min max max typ max max min typ typ typ typ A A B A B A B A A C C A A A C A C C C C A A A C A A A A C C C VCM = 0V Open-Loop, DC 1kΩ Load VO = 0 VO = 0 G = +2, f = 100kHz VDIS = 0 (all channels) VIN = +1V, G = +2 VIN = +1V, G = +2 G = +2, 5MHz ±3.75 60 50 || 2 4.0 ±4.1 160 –120 0.006 –300 4 40 70 1.7 3.4 1.8 80 ±5 ±3.9 120 –100 ±3.9 115 –95 ±3.8 110 –90 –500 –580 –600 VDIS = 0V/Channel 3.5 1.7 120 3.6 1.6 130 3.7 1.5 135 VS = ±5V/per Channel VS = ±5V/per Channel Input Referred ±6 1.8 1.6 54 1.7 1.7 60 –40 to +85 ±6 1.85 1.55 53 ±6 1.85 1.45 53 Junction-to-Ambient 100 100 NOTES: (1) Junction temperature = ambient for +25°C tested specifications. (2) Junction temperature = ambient at low temperature limit, junction temperature = ambient +2°C at high temperature limit for over temperature tested specifications. (3) Test levels: (A) 100% tested at +25°C. Over-temperature limits by characterization and simulation. (B) Limits set by characterization and simulation. (C) Typical value only for information. (4) Current is considered positive out-of-node. VCM is the input common-mode voltage. (5) Tested < 3dB below minimum specified CMR at ± CMIR limits. OPA3684 SBOS241A www.ti.com 3 ELECTRICAL CHARACTERISTICS: VS = +5V Boldface limits are tested at +25°C. RF = 1.0kΩ, RL = 100Ω, and G = +2, unless otherwise noted. OPA3684ID, IDBQ TYP PARAMETER AC PERFORMANCE (see Figure 3) Small-Signal Bandwidth (VO = 0.5Vp-p) CONDITIONS G = +1, RF = 1.0kΩ G = +2, RF = 1.0kΩ G = +5, RF = 1.0kΩ G = +10, RF = 1.0kΩ G = +20, RF = 1.0kΩ G = +2, VO < 0.5Vp-p, RF = 1.0kΩ RF = 1.0kΩ, VO < 0.5Vp-p G = 2, VO = 2Vp-p G = 2, VO = 2V Step G = 2, VO = 0.5V Step G = 2, VO = 2VStep G = 2, f = 5MHz, VO = 2Vp-p RL = 100Ω to VS/2 RL ≥ 1kΩ to VS/2 RL = 100Ω to VS/2 RL ≥ 1kΩ to VS/2 f > 1MHz f > 1MHz f > 1MHz G = +2, NTSC, VO = 1.4Vp, RL = 150Ω G = +2, NTSC, VO = 1.4Vp, RL = 150Ω 2 Channels, f = 5MHz 3rd-Channel Measured VO = VS/2, RL = 100Ω to VS/2 VCM = VS/2 VCM = VS/2 VCM = VS/2 VCM = VS/2 VCM = VS/2 VCM = VS/2 +25°C 140 110 100 90 75 21 0.5 86 380 4.3 4.8 –65 –84 –65 –74 3.7 9.4 17 0.04 0.07 70 MIN/MAX OVER TEMPERATURE +25°C(1) 0°C to 70°C(2) –40°C to +85°C(2) UNITS MHz MHz MHz MHz MHz MHz dB MHz V/µs ns ns dBc dBc dBc dBc nV/√Hz pA/√Hz pA/√Hz % deg dB MIN/ TEST MAX LEVEL(3) typ min min typ typ min max typ min typ typ max max max max max max max typ typ typ C B C C C B B C B C C B B B B B B B C C C 86 85 82 Bandwidth for 0.1dB Gain Flatness Peaking at a Gain of +1 Large-Signal Bandwidth Slew Rate Rise-and-Fall Time Harmonic Distortion 2nd-Harmonic 3rd-Harmonic Input Voltage Noise Noninverting Input Current Noise Inverting Input Current Noise Differential Gain Differential Phase All Hostile Crosstalk DC PERFORMANCE(4) Open-Loop Transimpedance Gain (ZOL) Input Offset Voltage Average Offset Voltage Drift Noninverting Input Bias Current Average Noninverting Input Bias Current Drift Inverting Input Bias Current Average Inverting Input Bias Current Drift INPUT Least Positive Input Voltage(5) Most Positive Input Voltage(5) Common-Mode Refection Ratio (CMRR) Noninverting Input Impedance Inverting Input Resistance (RI) OUTPUT Most Positive Output Voltage Least Positive Output Voltage Current Output, Sourcing Current Output, Sinking Closed-Loop Output Impedance DISABLE (Disabled LOW) Power-Down Supply Current (+VS) Off Isolation Output Capacitance in Disable Turn-On Glitch Turn-Off Glitch Enable Voltage Disable Voltage Control Pin Input Bias Current (DIS) POWER SUPPLY Specified Single-Supply Operating Voltage Max Single-Supply Operating Voltage Range Max Quiescent Current Min Quiescent Current Power-Supply Rejection Ratio (+PSRR) TEMPERATURE RANGE Specification: D, DBQ Thermal Resistance, θJA Junction-to-Ambient D SO-14 DBQ SSOP-16 12 2.6 300 11 3.4 290 10 3.7 285 –60 –62 –64 –70 4.1 11 18 –59 –61 –63 –70 4.2 12 18.5 –59 –61 –63 –69 4.4 12.5 19 355 ±1.0 ±5 ±5 ±3.4 ±12 ±13 1.32 3.68 51 160 155 ±4.0 ±12 ±13.5 ±25 ±14.5 ±25 1.35 3.65 50 153 ±4.2 ±12 ±14 ±30 ±16 ±30 1.38 3.62 50 kΩ mV µV/°C µA nA/°C µA nA°/C V V dB kΩ || pF Ω V V mA mA Ω µA dB pF mV mV V V µA V V mA mA dB °C °C/W °C/W min max max max max max max max min min typ typ min max min min typ typ typ typ typ typ min max max typ max max min typ typ typ typ A A B A B A B A A A C C A A A A C C C C C C A A A C A A A C C C C VCM = VS/2 Open-Loop RL = 1kΩ to VS/2 RL = 1kΩ to VS/2 VO = VS/2 VO = VS/2 G = +2, f = 100kHz VDIS = 0 (all channels) F = 5.0MHz G = +2, RL = 150Ω, VIN = VS/2 G = +2, RL = 150Ω, VIN = VS/2 1.25 3.75 58 50 || 1 4.5 4.10 0.9 80 70 3.9 1.1 65 55 3.9 1.1 60 50 3.8 1.2 55 45 –300 70 1.7 VDIS = 0V/Channel 3.4 1.8 80 5 3.5 1.7 120 3.6 1.6 130 3.7 1.5 135 VS = +5V/Channel VS = +5V/Channel Input Referred 1.44 1.44 65 –40 to +85 100 100 12 1.55 1.30 12 1.55 1.20 12 1.55 1.15 NOTES: (1) Junction temperature = ambient for +25°C tested specifications. (2) Junction temperature = ambient at low temperature limit, junction temperature = ambient +1°C at high temperature limit for over temperature tested specifications. (3) Test levels: (A) 100% tested at +25°C. Over-temperature limits by characterization and simulation. (B) Limits set by characterization and simulation. (C) Typical value only for information. (4) Current is considered positive out-of-node. VCM is the input common-mode voltage. (5) Tested < 3dB below minimum specified CMR at ± CMIR limits. 4 OPA3684 www.ti.com SBOS241A TYPICAL CHARACTERISTICS: VS = ±5V At TA = +25°C, G = +2, RF = 800Ω, and RL = 100Ω, unless otherwise noted. NONINVERTING SMALL-SIGNAL FREQUENCY RESPONSE 6 3 VO = 0.5Vp-p RF = 800Ω 3 INVERTING SMALL-SIGNAL FREQUENCY RESPONSE VO = 0.5Vp-p RF = 800Ω Normalized Gain (3dB/div) Normalized Gain (3dB/div) 0 –3 –6 –9 –12 –15 See Figure 1 –18 1 10 Frequency (MHz) G = 50 G = 100 G=5 G = 10 G = 20 G=1 G=2 0 –3 –6 G = –1 G = –2 G = –5 G = –10 G = –16 10 Frequency (MHz) 100 200 –9 See Figure 2 1 –12 100 200 NONINVERTING LARGE-SIGNAL FREQUENCY RESPONSE 9 G = +2 RL = 100Ω 3 INVERTING LARGE-SIGNAL FREQUENCY RESPONSE G = –1 RL = 100Ω VO = 0.5Vp-p 1Vp-p VO = 0.5Vp-p 6 0 Gain (dB) Gain (dB) VO = 1Vp-p 3 –3 2Vp-p 5Vp-p –6 0 VO = 2Vp-p VO = 5Vp-p –9 See Figure 2 1 10 Frequency (MHz) 100 200 –3 1 See Figure 1 –12 10 Frequency (MHz) 100 200 NONINVERTING PULSE RESPONSE 0.8 G = +2 1.6 0.8 INVERTING PULSE RESPONSE 1.6 G = –1 Output Voltage (200mV/div) Output Voltage (400mV/div) Output Voltage (200mV/div) 0.4 0.2 Large-Signal Right Scale Small-Signal Left Scale 0.8 0.4 0 –0.4 –0.8 –1.2 0.4 0.2 0 Small-Signal Left Scale –0.2 –0.4 –0.6 See Figure 2 –0.8 Time (10ns/div) Large-Signal Right Scale 0.8 0.4 0 –0.4 –0.8 –1.2 –1.6 0 –0.2 –0.4 –0.6 See Figure 1 –0.8 Time (10ns/div) –1.6 OPA3684 SBOS241A www.ti.com 5 Output Voltage (400mV/div) 0.6 1.2 0.6 1.2 TYPICAL CHARACTERISTICS: VS = ±5V (Cont.) At TA = +25°C, G = +2, RF = 800Ω, and RL = 100Ω, unless otherwise noted. HARMONIC DISTORTION vs LOAD RESISTANCE –50 –55 VO = 2Vp-p f = 5MHz G = +2 HARMONIC DISTORTION vs FREQUENCY –50 VO = 2Vp-p RL = 100Ω Harmonic Distortion (dBc) Harmonic Distortion (dBc) –60 –65 –70 –75 –80 –85 –60 2nd-Harmonic –70 3rd-Harmonic –80 2nd-Harmonic 3rd-Harmonic See Figure 1 –90 100 Load Resistance (Ω) 1k –90 See Figure 1 0.1 1 Frequency (MHz) 10 20 HARMONIC DISTORTION vs OUTPUT VOLTAGE –50 f = 5MHz RL = 100Ω –60 2nd-Harmonic –50 5MHz HARMONIC DISTORTION vs SUPPLY VOLTAGE VO = 2Vp-p RL = 100Ω –60 Harmonic Distortion (dBc) –70 3rd-Harmonic –80 Harmonic Distortion (dBc) 2nd-Harmonic –70 3rd-Harmonic –80 –90 0.5 1 Output Voltage (Vp-p) 5 –90 ±2.5 ±3 ±3.5 ±4 ±4.5 ±5 Supply Voltage (±V) ±5.5 ±6 HARMONIC DISTORTION vs NONINVERTING GAIN –50 –55 2nd-Harmonic –60 –65 –70 –75 3rd-Harmonic –80 –85 –90 1 Noninverting Gain (V/V) 10 20 –50 –55 HARMONIC DISTORTION vs INVERTING GAIN Harmonic Distortion (dBc) Harmonic Distortion (dBc) 2nd-Harmonic –60 –65 –70 –75 –80 –85 –90 1 Inverting Gain (V/V) 10 20 3rd-Harmonic 6 OPA3684 www.ti.com SBOS241A TYPICAL CHARACTERISTICS: VS = ±5V (Cont.) At TA = +25°C, G = +2, RF = 800Ω, and RL = 100Ω, unless otherwise noted. INPUT VOLTAGE AND CURRENT NOISE DENSITY 100 Noninverting Current Noise 9.4pA/√Hz 3rd-Order Spurious Level (dBc) 2-TONE, 3RD-ORDER INTERMODULATION DISTORTION –50 20MHz +5V Inverting Current Noise 17pA/√Hz Voltage Noise (nV/√Hz) Current Noise (pA/√Hz) –60 PI 50Ω OPA3684 50Ω PO 50Ω –5V 800Ω 10MHz 10 –70 800Ω 5MHz –80 1MHz –90 Voltage Noise 3.7nV/√Hz 1 100 1k 10k 100k 1M 10M Frequency (Hz) –8 –7 –6 –5 –4 –3 –2 –1 0 1 2 3 4 5 Power at Load (each tone, dBm) 6 7 8 DISABLE TIME 6 5 VOUT and VDIS (V) –40 DISABLED FEEDTHROUGH G = +2 VDIS = 0 VDIS Feedthrough (dB) –50 –60 –70 –80 –90 4 3 2 1 0 0 2 4 VOUT VIN = 1VDC See Figure 1 See Figure 1 –100 6 8 Time (ms) 10 12 14 16 0.1 1 Frequency (MHz) 10 100 50 RS vs CLOAD 0.5dB Peaking SMALL-SIGNAL BANDWIDTH vs CLOAD 9 12pF 5pF 40 Normalized Gain (dB) 6 100pF +5V VI RS 50Ω OPA3684 CL –5V 800Ω 800Ω 1kΩ VO RS (Ω) 30 3 75pF 20 0 50pF 33pF 20pF 10 –3 0 1 10 CLOAD (pF) 100 –6 1 10 Frequency (MHz) 100 300 OPA3684 SBOS241A www.ti.com 7 TYPICAL CHARACTERISTICS: VS = ±5V (Cont.) At TA = +25°C, G = +2, RF = 800Ω, and RL = 100Ω, unless otherwise noted. OPEN-LOOP TRANSIMPEDANCE GAIN AND PHASE vs FREQUENCY CMRR and PSRR vs FREQUENCY Common-Mode Rejection Ratio (dB) Power-Supply Rejection Ratio (dB) CMRR 60 50 +PSRR 40 –PSRR 30 20 10 0 102 103 104 105 106 Frequency (Hz) 107 108 Open-Loop Transimpedance Gain (dBΩ) 70 120 20log (ZOL) 100 80 60 40 20 0 102 103 104 105 106 Frequency (Hz) 107 108 0 –30 –60 –90 –120 –150 –180 109 ∠ ZOL COMPOSITE VIDEO DIFFERENTIAL GAIN/PHASE 0.10 0.09 0.08 Gain = +2 NTSC, Positive Video OUTPUT CURRENT AND VOLTAGE LIMITATIONS 5 4 3 2 1W Power Limit =1 00Ω Differential Gain (%) Differential Phase (°) 0.07 VO (V) 0.06 0.05 0.04 0.03 0.02 0.01 0 1 dG 1 0 –1 dP –3 –4 –5 2 3 4 Each Channel RL = 500Ω –2 R = RL 50 Ω L 1W Power Limit –150 –100 –50 Number of 150Ω Video Loads 0 IO (MA) 50 100 150 TYPICAL DC DRIFT OVER AMBIENT TEMPERATURE 4 3 Input Bias Currents (µA) and Offset Voltage (mV) SUPPLY AND OUTPUT CURRENT vs AMBIENT TEMPERATURE 200 Sourcing Output Current Output Current (mA) 1.9 Supply Current per Channel (mA) 2 1 Noninverting Input Bias Current 0 –1 –2 Inverting Input Bias Current –3 –4 –50 –25 0 25 50 75 100 125 Ambient Temperature (°C) Input Offset Voltage 175 Supply Current 150 1.8 1.7 125 Sinking Output Current 1.6 100 –25 0 25 50 75 Ambient Temperature (°C) 100 1.5 125 8 OPA3684 www.ti.com SBOS241A Open-Loop Phase (°) TYPICAL CHARACTERISTICS: VS = ±5V (Cont.) At TA = +25°C, G = +2, RF = 800Ω, and RL = 100Ω, unless otherwise noted. SETTLING TIME 0.05 –20 ALL HOSTILE CROSSTALK –25 –30 –35 –40 –45 –50 –55 –60 –65 –70 1Vp-p Output 2-Channels, 100Ω Load % Error to Final Value 0.03 0.02 0.01 0 –0.01 –0.02 –0.03 –0.04 –0.05 0 10 20 30 Time (ns) 40 50 60 Crosstalk (Input referred) (dB) 0.04 2V Step See Figure 1 0.1 1 Frequency (MHz) 10 100 NONINVERTING OVERDRIVE RECOVERY 4.0 3.2 8.0 6.4 8.0 6.4 INVERTING OVERDRIVE RECOVERY 8.0 6.4 Output Voltage (1.6V/div) 1.6 0.8 0 –0.8 –1.6 –2.4 –3.2 –4.0 Time (100ns/div) Input Voltage Left Scale Output Voltage Right Scale See Figure 1 3.2 1.6 0 –1.6 –3.2 –4.8 –6.4 –8.0 3.2 1.6 0 –1.6 –3.2 –4.8 –6.4 –8.0 Time (100ns/div) Input Voltage Left Scale Output Voltage Right Scale 3.2 1.6 0 –1.6 –3.2 –4.8 See Figure 2 –6.4 –8.0 INPUT AND OUTPUT VOLTAGE RANGE vs SUPPLY VOLTAGE 6 5 4 3 2 1 0 –1 –2 –3 –4 –5 –6 ±2 ±3 100 CLOSED-LOOP OUTPUT IMPEDANCE vs FREQUENCY Input and Output Voltage Range Output Impedance (Ω) 10 1/3 OPA3684 800Ω 800Ω ZO Input Voltage Range Output Voltage Range 1 0.01 0.001 ±4 Supply Voltage (±V) ±5 ±6 100 1k 10k 100k Frequency (Hz) 1M 10M 100M OPA3684 SBOS241A www.ti.com 9 Output Voltage (1.6V/div) Input Voltage (0.8V/div) Input Voltage (1.6V/div) 2.4 4.8 4.8 4.8 TYPICAL CHARACTERISTICS: VS = +5V At TA = +25°C, G = +2, RF = 1kΩ, and RL = 100Ω, unless otherwise noted. NONINVERTING SMALL-SIGNAL FREQUENCY RESPONSE 6 3 Normalized Gain (3dB/div) INVERTING SMALL-SIGNAL FREQUENCY RESPONSE 3 RF = 1.0kΩ RF = 1kΩ G = 100 G = 50 Normalized Gain (3dB/div) G=1 G=2 0 –3 –6 –9 –12 –15 See Figure 3 –18 1 10 Frequency (MHz) 100 200 G=5 G = 20 G = 10 0 –3 –6 G = –1 G = –2 G = –5 G = –10 G = –20 10 Frequency (MHz) 100 200 –9 See Figure 4 1 –12 NONINVERTING LARGE-SIGNAL FREQUENCY RESPONSE 9 0.2Vp-p 6 0.5Vp-p INVERTING LARGE-SIGNAL FREQUENCY RESPONSE 3 VO = 0.2Vp-p VO = 0.5Vp-p 0 VO = 1Vp-p VO = 2Vp-p –6 Gain (dB) Gain (dB) 100 200 1Vp-p 3 2Vp-p 0 –3 –9 –3 1 10 Frequency (MHz) –12 1 10 Frequency (MHz) 100 200 NONINVERTING PULSE RESPONSE 0.4 1.6 0.4 INVERTING PULSE RESPONSE 1.6 Output Voltage (200mV/div) Output Voltage (400mV/div) Output Voltage (200mV/div) 0.2 0.1 Large-Signal Right Scale Small-Signal Left Scale 0.8 0.4 0 –0.4 –0.8 –1.2 0.2 0.1 0 Small-Signal Left Scale –0.1 –0.2 –0.3 See Figure 4 –0.4 Time (10ns/div) Large-Signal Right Scale 0.8 0.4 0 –0.4 –0.8 –1.2 –1.6 0 –0.1 –0.2 –0.3 See Figure 3. –0.4 Time (10ns/div) –1.6 10 OPA3684 www.ti.com SBOS241A Output Voltage (400mV/div) 0.3 1.2 0.3 1.2 TYPICAL CHARACTERISTICS: VS = +5V (Cont.) At TA = +25°C, G = +2, RF = 1kΩ, and RL = 100Ω, unless otherwise noted. HARMONIC DISTORTION vs LOAD RESISTANCE –50 –55 VO = 2Vp-p f = 5MHz HARMONIC DISTORTION vs FREQUENCY –50 VO = 2Vp-p RL = 100Ω Harmonic Distortion (dBc) Harmonic Distortion (dBc) –60 3rd-Harmonic –65 –70 –75 –80 2nd-Harmonic –85 See Figure 3 –90 100 Load Resistance (Ω) 1k –60 2nd-Harmonic –70 3rd-Harmonic –80 –90 See Figure 3 0.1 1 Frequency (MHz) 10 20 HARMONIC DISTORTION vs OUTPUT VOLTAGE –50 2nd-Harmonic –60 2-TONE, 3RD-ORDER INTERMODULATION DISTORTION –50 3rd-Order Spurious Level (dBc) 20MHz –60 10MHz Harmonic Distortion (dBc) –70 3rd-Harmonic –70 5MHz –80 –80 –90 See Figure 3 0.5 1 Output Voltage (Vp-p) 2 3 –90 See Figure 3 –15 –14 –13 –12 –11 –10 –9 –8 –7 –6 –5 –4 –3 Power at Load (each tone, dBm) SUPPLY AND OUTPUT CURRENT vs AMBIENT TEMPERATURE 100 Supply and Output Current (mA) COMPOSITE VIDEO DIFFERENTIAL GAIN/PHASE 1.5 Supply Current per Channel (nA) 0.16 0.14 G = +2 NTSC, Positive Video 90 Right-Scale Supply Current 1.4 Differential Gain (%) Differential Phase (°) 0.12 0.10 0.08 0.06 0.04 0.02 0 1 2 3 4 dG dP 80 Left-Scale Sourcing Output Current 1.3 70 Left-Scale Sinking Output Current 60 1.2 1.1 50 –50 –25 0 25 50 75 Ambient Temperature (°C) 100 125 1.0 Number of 150Ω Video Loads OPA3684 SBOS241A www.ti.com 11 APPLICATIONS INFORMATION LOW-POWER, CURRENT-FEEDBACK OPERATION The triple-channel OPA3684 gives a new level of performance in low-power, current-feedback op amps. Using a new input stage buffer architecture, the OPA3684 CFBPLUS amplifier holds nearly constant AC performance over a wide gain range. This closed-loop internal buffer gives a very low and linearized impedance at the inverting node, isolating the amplifier’s AC performance from gain element variations. This allows both the bandwidth and distortion to remain nearly constant over gain, moving closer to the ideal currentfeedback performance of gain bandwidth independence. This low-power amplifier also delivers exceptional output power—it’s ±4V swing on ±5V supplies with > 100mA output drive gives excellent performance into standard video loads or doubly-terminated 50Ω cables. Single +5V supply operation is also supported with similar bandwidths but with reduced output power capability. For lower quiescent power in a CFBPLUS amplifier, consider the OPA683 family; while for higher output power, consider the OPA691 family. Figure 1 shows the DC-coupled, gain of +2, dual powersupply circuit used as the basis of the ±5V Electrical and Typical Characteristics for each channel. For test purposes, the input impedance is set to 50Ω with a resistor to ground and the output impedance is set to 50Ω with a series output resistor. Voltage swings reported in the Electrical Characteristics are taken directly at the input and output pins while load powers (dBm) are defined at a matched 50Ω load. For the circuit of Figure 1, the total effective load will be 100Ω || 1600Ω = 94Ω. Gain changes are most easily accomplished by simply resetting the RG value, holding RF constant at its recommended value of 800Ω. mode signal across the input stage, the slew rate for inverting operation is typically higher and the distortion performance is slightly improved. An additional input resistor, RM, is included in Figure 2 to set the input impedance equal to 50Ω. The parallel combination of RM and RG set the input impedance. As the desired gain increases for the inverting configuration, RG is adjusted to achieved the desired gain, while RM is also adjusted to hold a 50Ω input match. A point will be reached where RG will equal 50Ω, RM is removed, and the input match is set by RG only. With RG fixed to achieve an input match to 50Ω, increasing RF will increase the gain. This will, however, quickly reduce the achievable bandwidth as the feedback resistor increases from its recommended value of 800Ω. If the source does not require an input match to 50Ω, either adjust RM to get the desired load, or remove it and let the RG resistor alone provide the input load. +5V 0.1µF + 6.8µF DIS 1/3 OPA3684 50Ω 50Ω Load 50Ω Source VI RG 800Ω RM 53.6Ω –5V RF 800Ω 0.1µF + 6.8µF +5V FIGURE 2. DC-Coupled, G = –1V/V, Bipolar Supply Specifications and Test Circuit. + 0.1µF 6.8µF VI 50Ω Source RM 50Ω DIS 1/3 OPA3684 50Ω 50Ω Load These circuits show ±5V operation. The same circuits can be applied with bipolar supplies from ±2.5V to ±6V. Internal supply independent biasing gives nearly the same performance for the OPA3684 over this wide range of supplies. Generally, the optimum feedback resistor value (for nominally flat frequency response at G = +2) will increase in value as the total supply voltage across the OPA3684 is reduced. See Figure 3 for the AC-coupled, single +5V supply, gain of +2V/V circuit configuration used as a basis for the +5V only Electrical and Typical Characteristics for each channel. The key requirement of broadband single-supply operation is to maintain input and output signal swings within the usable voltage ranges at both the input and the output. The circuit of Figure 3 establishes an input midpoint bias using a simple resistive divider from the +5V supply (two 10kΩ resistors) to the noninverting input. The input signal is then AC-coupled into this midpoint voltage bias. The input voltage can swing to within 1.25V of either supply pin, giving a 2.5Vp-p input signal range centered between the supply pins. The input impedance of Figure 3 is set to give a 50Ω input match. If the source does not require a 50Ω match, remove this and drive RF 800Ω RG 800Ω –5V 0.1µF + 6.8µF FIGURE 1. DC-Coupled, G = +2V/V, Bipolar Supply Specifications and Test Circuit. Figure 2 shows the DC-coupled, gain of –1V/V, dual powersupply circuit used as the basis of the Inverting Typical Characteristics for each channel. Inverting operation offers several performance benefits. Since there is no common- 12 OPA3684 www.ti.com SBOS241A directly into the blocking capacitor. The source will then see the 5kΩ load of the biasing network as a load. The gain resistor (RG) is AC-coupled, giving the circuit a DC gain of +1, which puts the noninverting input DC bias voltage (2.5V) on the output as well. The feedback resistor value has been adjusted from the bipolar ±5V supply condition to re-optimize for a flat frequency response in +5V only, gain of +2, operation. On a single +5V supply, the output voltage can swing to within 1.0V of either supply pin while delivering more than 70mA output current—easily giving a 3Vp-p output swing into 100Ω (8dBm maximum at the matched 50Ω load). The circuit of Figure 3 shows a blocking capacitor driving into a 50Ω output resistor, then into a 50Ω load. Alternatively, the blocking capacitor could be removed if the load is tied to a supply midpoint or to ground if the DC current then required by the load is acceptable. The circuits of Figure 3 and 4 show single-supply operation at +5V. These same circuits may be used up to single supplies of +12V with minimal change in the performance of the OPA3684. +5V 10kΩ 0.1µF + 6.8µF DIS 0.1µF 10kΩ OPA3684 1/3 0.1µF 50Ω 50Ω Load 50Ω Source VI RG 0.1µF 1.0kΩ RM 52.3Ω RF 1.0kΩ +5V 50Ω Source VI 0.1µF RM 50Ω 10kΩ 0.1µF + 6.8µF FIGURE 4. AC-Coupled, G = –1V/V, Single-Supply Specifications and Test Circuit. DIS 1/3 10kΩ OPA3684 0.1µF 50Ω 50Ω Load LOW-POWER, VIDEO LINE DRIVER APPLICATIONS For low-power, video line driving, the OPA3684 provides the output current and linearity to support 3 channels of either single video lines, or up to 4 video lines in parallel on each output. Figure 5 shows a typical ±5V supply video line driver application where only one channel is shown and only a single line is being driven. The improved 2nd-harmonic distortion of the CFBPLUS architecture, along with the OPA3684’s high output current and voltage, gives exceptional differential gain and phase performance for a lowpower solution. As the Typical Characteristics show, a single video load shows a dG/dP of 0.04%/0.02°. Multiple loads may be driven on each output, with minimal x-talk, while the dG/dP is still < 0.1%/0.1° for up to 4 parallel video loads. The slew rate and gain of 2 bandwidth are also suitable to moderate resolution RGB applications. RF 1kΩ RG 1kΩ 0.1µF FIGURE 3. AC-Coupled, G = +2V/V, Single-Supply Specifications and Test Circuit. Figure 4 shows the AC-coupled, single +5V supply, gain of –1V/V circuit configuration used as a basis for the inverting +5V only Typical Characteristics for each channel. In this case, the midpoint DC bias on the noninverting input is also decoupled with an additional 0.1µF capacitor. This reduces the source impedance at higher frequencies for the noninverting input bias current noise. This 2.5V bias on the noninverting input pin appears on the inverting input pin and, since RG is DC-blocked by the input capacitor, will also appear at the output pin. One advantage to inverting operation is that since there is no signal swing across the input stage, higher slew rates and operation to even lower supply voltages is possible. To retain a 1Vp-p output capability, operation down to a 3V supply is allowed. At a +3V supply, the input stage is saturated, but for the inverting configuration of a current-feedback amplifier, wideband operation is retained even under this condition. +5V VIDEOIN DIS Supply decoupling not shown. 75Ω Coax 75Ω Load 75Ω OPA3684 1kΩ 1kΩ –5V FIGURE 5. Noninverting Differential I/O Amplifier. OPA3684 SBOS241A www.ti.com 13 LOW-POWER RGB MUX/LINE DRIVER Using the shutdown feature, two OPA3684’s can provide an easy low-power way to select one of two possible RGB sources for moderate resolution monitors. Figure 6 shows a recommended circuit where each of the color outputs are combined in a way that provides a net gain of 1 to the matched 75Ω load with a 75Ω output impedance. This brings the two outputs for each color together through a 78.7Ω resistor with a slightly > 2 gain provided by the amplifiers. When one channel is shutdown, the feedback network is still present, slightly attenuating the signal and combining in parallel with the 78.7Ω to give a 75Ω source impedance. Since the OPA3684 does not disable quickly, this approach is not suitable for pixel-by-pixel multiplexing—however, it does provide an easy way to switch between two possible RGB sources. The output swing provided by the active channel will divide back through the inactive channel feedback to appear at the inverting input of the OFF channel. To retain good pulse fidelity, or low distortion, this divided down output signal at the inverting inputs of the OFF channels, plus the OFF channel input signals, should not exceed 0.7Vp-p. As the signal across the buffers of the inactive channels exceeds 0.7Vp-p, diodes across the inputs begin to turn on causing a nonlinear load to the active channel. This will degrade signal purity under those conditions. +5V VDIS +5V U1 R1 75Ω 1/3 OPA3684 78.7Ω VOUT Red 75Ω Line 681Ω 806Ω Power Supply De-Coupling Not Shown LOW-POWER, FLEXIBLE GAIN, DIFFERENTIAL RECEIVER The 3 channels available in the OPA3684 can be applied to a very flexible differential to single-ended receiver. Since the bandwidth does not depend on the gain setting, the gain setting element of Figure 7 (RG) can be adjusted over a wide range with minimal impact on resulting bandwidth. Frequency-response shaping elements may be included in RG as well to provide line equalization or filtering in the final output signal. G1 75Ω 1/3 OPA3684 78.7Ω VOUT Green 75Ω Line 681Ω 806Ω B1 75Ω 1/3 OPA3684 78.7Ω VOUT Blue +5 75Ω Line 681Ω 806Ω V1 1/3 OPA3684 –5V +5V U2 R2 75Ω 1/3 OPA3684 78.7Ω –5 806Ω 402Ω +5 1/3 OPA3684 (1 + 2(806Ω)/RG) (V1 – V2) 402Ω RG 806Ω –5 806Ω 681Ω 806Ω +5 1/3 OPA3684 806Ω G2 75Ω 1/3 OPA3684 V2 78.7Ω –5 High-Speed INA (>120MHz) 681Ω 806Ω FIGURE 7. Low-Power, Wide Gain Range, Differential Receiver. The first two amplifiers provide the differential gain function with a common-mode gain of 1. The second amplifier performs the differencing function to remove the common-mode (referencing the output to ground if the 402Ω resistor is grounded) and providing a differential gain of 1. The resistors have been scaled to provide the same output loading on each first stage amplifier. Typical bandwidths for the circuit of Figure 7 exceed 120MHz. B2 75Ω 1/3 OPA3684 78.7Ω 681Ω 806Ω –5V FIGURE 6. Wideband 2x1 RGB Multiplexer. 14 OPA3684 www.ti.com SBOS241A WIDEBAND PGA FOR ADC DRIVING Using the 3 channels of the OPA3684, and the shutdown feature, can give an easy to use PGA function—which can be applied to driving an ADC. Since the bandwidth does not vary with gain for the CFBPLUS OPA3684, each channel can be set up to a desired gain setting, with each of the noninverting inputs driven with the same input signal. Selecting one of the 3 channels passes on the input with the gain setting provided by the selected channel. Figure 8 shows an example where the channels are set to gains of 2, 5, and 10. Again, the output signal will be divided down back to the inverting inputs of the inactive channels. To retain good pulse fidelity, or low distortion, this divided down output signal at the inverting inputs of the OFF channels, plus the OFF channel input signals, should not exceed 0.7Vp-p. As the signal across the buffers of the inactive channels exceeds +5V 0.7Vp-p, diodes across the inputs begin to turn on causing a nonlinear load to the active channel. This will degrade signal fidelity under those conditions. VIDEO DAC RECONSTRUCTION FILTER Wideband current-feedback op amps make ideal elements for implementing high-speed active filters where the amplifier is used as a fixed gain block inside a passive RC circuit network. The triple channel OPA3684 can be used as a very effective video Digital-to-Analog Converter (DAC) reconstruction filter and line driver. Figure 9 shows an example of this where the delay-equalized filter compensates for the DAC’s sin(x)/x response, and minimizes aliasing artifacts. It is shown here as a single +5V design expecting a 13.5MSPS DAC sampling rate, and giving a 5.5MHz cutoff frequency. 74HC238 D1 D2 Y0 Y1 Y2 20Ω +5V Power-supply decoupling not shown. G = +2 1/3 OPA3684 100Ω 4.99kΩ 4.99kΩ 0.1µF 0.1µF 806Ω 806Ω 200Ω VIN 50Ω 200Ω G = +5 1/3 OPA3684 100Ω 0.1µF +In REFT +3.5V REFB +1.5V 806Ω 100pF –In CM ADS826 10-Bit 60MSPS 20Ω 100Ω G = +10 1/3 OPA3684 0.1µF 90.9Ω 806Ω –5V FIGURE 8. Wideband PGA for ADC Driving. 100pF Video 100µF In +5V 806Ω 806Ω 97.6Ω 237Ω 220pF 402Ω 56pF 1/3 OPA3684 100pF +5V 82.5Ω 243Ω 220pF 412Ω 56pF 1/3 OPA3684 +5V 75.5Ω VO 120pF 1/3 OPA3684 806Ω 806Ω 806Ω +5V 100µF 953Ω 953Ω FIGURE 9. Composite Video Filter. OPA3684 SBOS241A www.ti.com 15 The first stage buffers the video DAC output to the first 3rd-order filter section. This stage also provides group delay equalization while the 2nd and 3rd stages each give a 3rdorder low-pass response with sin(x)/x equalization. Figure 10 shows the frequency response for the filter of Figure 9. OPERATING SUGGESTIONS SETTING RESISTOR VALUES TO OPTIMIZE BANDWIDTH Any current-feedback op amp like the OPA3684 can hold high bandwidth over signal-gain settings with the proper adjustment of the external resistor values. A low-power part like the OPA3684 typically shows a larger change in bandwidth due to the significant contribution of the inverting input impedance to loop-gain changes as the signal gain is changed. Figure 11 shows a simplified analysis circuit for any currentfeedback amplifier. 20 10 0 –10 (dB) f–3dB –20 –30 –40 –50 0 1 Frequency (MHz) 10 100 VI α VO RI iERR Z(S) iERR RF FIGURE 10. Video Filter Frequency Response. RG DESIGN-IN TOOLS DEMONSTRATION BOARDS Two PC boards are available to assist in the initial evaluation of circuit performance using the OPA3684 in its two package styles. Both of these are available, free, as an unpopulated PC board delivered with descriptive documentation. The summary information for these boards is shown in Table I. BOARD PART NUMBER DEM-OPA368xD DEM-OPA368xDBQ LITERATURE REQUEST NUMBER SBOU018 SBOU019 FIGURE 11. Current-Feedback Transfer Function Analysis Circuit. The key elements of this current-feedback op amp model are: α ⇒ Buffer gain from the noninverting input to the inverting input RI ⇒ Buffer output impedance iERR ⇒ Feedback error current signal Z(S) ⇒ Frequency-dependent open-loop transimpedance gain from iERR to VO The buffer gain is typically very close to 1.00 and is normally neglected from signal gain considerations. It will, however, set the CMRR for a single op amp differential amplifier configuration. For the buffer gain α < 1.0 and CMRR = –20 • log(1 – α). The closed-loop input stage buffer used in the OPA3684 gives a buffer gain more closely approaching 1.00 and this shows up in a slightly higher CMRR than previous current-feedback op amps. RI, the buffer output impedance, is a critical portion of the bandwidth control equation. The OPA3684 reduces this element to approximately 4.0Ω using the local loop gain of the input buffer stage. This significant reduction in output impedance, on very low power, contributes significantly to extending the bandwidth at higher gains. A current-feedback op amp senses an error current in the inverting node (as opposed to a differential input error voltage for a voltage-feedback op amp) and passes this on to the output through an internal frequency-dependent PRODUCT OPA3684ID OPA3684IDBQ PACKAGE SO-14 SSOP-16 TABLE I. Demo Board Ordering Information. MACROMODELS Computer simulation of circuit performance using SPICE is often useful in predicting the performance of analog circuits and systems. This is particularly true for Video and RF amplifier circuits where parasitic capacitance and inductance can have a major effect on circuit performance. Check the TI web site (www.ti.com) for SPICE macromodels within the OPA3684 product folder. These models do a good job of predicting small-signal AC and transient performance under a wide variety of operating conditions. They do not do as well in predicting distortion or dG/dP characteristics. Most of these models do not attempt to distinguish between the package types in their small-signal AC performance. 16 OPA3684 www.ti.com SBOS241A transimpedance gain. The Typical Characteristics show this open-loop transimpedance response. This is analogous to the open-loop voltage gain curve for a voltage-feedback op amp. Developing the transfer function for the circuit of Figure 14 gives Equation 1: (1)  R α 1 + F  RG   VO α NG = = VI  RF  1 + RF + RI NG RF + RI 1 +  Z (S )  RG  1+ Z (S )   R  NG = 1 + F    RG      inverting node voltage. While it is always important to keep the inverting node capacitance low for any current-feedback op amp, it is critically important for the OPA3684. External layout capacitance in excess of 2pF will start to peak the frequency response. This peaking can be easily reduced by increasing the feedback resistor value—but it is preferable, from a noise and dynamic range standpoint, to keep that capacitance low, allowing a close to nominal 800Ω feedback resistor for flat frequency response. Very high parasitic capacitance values on the inverting node (> 5pF) can possibly cause input stage oscillation that cannot be filtered by a feedback element adjustment. At very high gains, 2nd-order effects in the inverting output impedance cause the overall response to peak up. If desired, it is possible to retain a flat frequency response at higher gains by adjusting the feedback resistor to higher values as the gain is increased. Since the exact value of feedback that will give a flat frequency response depends strongly in inverting and output node parasitic capacitance values, it is best to experiment in the specific board with increasing values until the desired flatness (or pulse response shape) is obtained. In general, increasing RF (and adjusting RG to the desired gain) will move towards flattening the response, while decreasing it will extend the bandwidth at the cost of some peaking. This is written in a loop-gain analysis format where the errors arising from a non-infinite open-loop gain are shown in the denominator. If Z(S) were infinite over all frequencies, the denominator of Equation 1 would reduce to 1 and the ideal desired signal gain shown in the numerator would be achieved. The fraction in the denominator of Equation 1 determines the frequency response. Equation 2 shows this as the loop-gain equation. Z (S ) RF + RI NG = Loop Gain (2) OUTPUT CURRENT AND VOLTAGE The OPA3684 provides output voltage and current capabilities that can support the needs of driving doubly-terminated 50Ω lines. For a 100Ω load at the gain of +2 (see Figure 1), the total load is the parallel combination of the 100Ω load and the 1.6kΩ total feedback network impedance. This 94Ω load will require no more than 40mA output current to support the ±3.8V minimum output voltage swing specified for 100Ω loads. This is well under the specified minimum +110mA/–90mA output current specifications over the full temperature range. The specifications described above, though familiar in the industry, consider voltage and current limits separately. In many applications, it is the voltage • current, or V-I product, which is more relevant to circuit operation. Refer to the “Output Voltage and Current Limitations” curve in the Typical Characteristics. The X- and Y-axes of this graph show the zero-voltage output current limit and the zero-current output voltage limit, respectively. The four quadrants give a more detailed view of the OPA3684’s output drive capabilities. Superimposing resistor load lines onto the plot shows the available output voltage and current for specific loads. The minimum specified output voltage and current over temperature are set by worst-case simulations at the cold temperature extreme. Only at cold startup will the output current and voltage decrease to the numbers shown in the Electrical Characteristic tables. As the output transistors deliver power, their junction temperatures will increase, decreasing their VBE’s (increasing the available output voltage swing) and increasing their current gains (increasing the available output current). In steady-state operation, the If 20 • log(RF + NG • RI) were drawn on top of the open-loop transimpedance plot, the difference between the two would be the loop gain at a given frequency. Eventually, Z(S) rolls off to equal the denominator of Equation 2 at which point the loop gain has reduced to 1 (and the curves have intersected). This point of equality is where the amplifier’s closed-loop frequency response given by Equation 1 will start to roll off, and is exactly analogous to the frequency at which the noise gain equals the open-loop voltage gain for a voltage-feedback op amp. The difference here is that the total impedance in the denominator of Equation 2 may be controlled somewhat separately from the desired signal gain (or NG). The OPA3684 is internally compensated to give a maximally flat frequency response for RF = 800Ω at NG = 2 on ±5V supplies. That optimum value goes to 1.0kΩ on a single +5V supply. Normally, with a current-feedback amplifier, it is possible to adjust the feedback resistor to hold this bandwidth up as the gain is increased. The CFBPLUS architecture has reduced the contribution of the inverting input impedance to provide exceptional bandwidth to higher gains without adjusting the feedback resistor value. The Typical Characteristics show the small-signal bandwidth over gain with a fixed feedback resistor. Putting a closed-loop buffer between the noninverting and inverting inputs does bring some added considerations. Since the voltage at the inverting output node is now the output of a locally closed-loop buffer, parasitic external capacitance on this node can cause frequency response peaking for the transfer function from the noninverting input voltage to the OPA3684 SBOS241A www.ti.com 17 available output voltage and current will always be greater than that shown in the over temperature specifications since the output stage junction temperatures will be higher than the minimum specified operating ambient. To maintain maximum output stage linearity, no output shortcircuit protection is provided. This will not normally be a problem since most applications include a series-matching resistor at the output that will limit the internal power dissipation if the output side of this resistor is shorted to ground. However, shorting the output pin directly to a power-supply pin will, in most cases, destroy the amplifier. If additional short-circuit protection is required, consider a small-series resistor in the power-supply leads. This will, under heavy output loads, reduce the available output voltage swing. A 5Ω series resistor in each power-supply lead will limit the internal power dissipation to less than 1W for an output short-circuit while decreasing the available output voltage swing only 0.25V for up to 50mA desired load currents. This slight drop in available swing is more if multiple channels are driving heavy loads simultaneously. Always place the 0.1µF powersupply decoupling capacitors after these supply current limiting resistors directly on the supply pins. An alternative approach is to place the 5Ω inside the loop at each output of the amplifiers. This will provide some short-circuit protection, but hurts the phase margin under capacitive load conditions. and add the recommended series resistor as close as possible to the OPA3684 output pin (see Board Layout Guidelines). DISTORTION PERFORMANCE The OPA3684 provides very low distortion in a low-power part. The CFBPLUS architecture also gives two significant areas of distortion improvement. First, in operating regions where the 2nd-harmonic distortion due to output stage nonlinearities is very low (frequencies < 1MHz, low output swings into light loads) the linearization at the inverting node provided by the CFBPLUS design gives 2nd-harmonic distortions that extend into the –90dBc region. Previous currentfeedback amplifiers have been limited to approximately –85dBc due to the nonlinearities at the inverting input. The second area of distortion improvement comes in a distortion performance that is largely gain independent. To the extent that the distortion at a particular output power is output-stage dependent, 3rd-harmonics particularly (and to a lesser extend 2nd-harmonic distortion) are constant as the gain is increased. This is due to the constant loop-gain versus signal gain provided by the CFBPLUS design. As shown in the Typical Characteristic curves, while the 3rd-harmonic is constant with gain, the 2nd-harmonic degrades at higher gains. This is largely due to board parasitic issues. Slightly imbalanced load return currents through the ground plane will couple into the gain resistor to cause a portion of the 2ndharmonic distortion. At high gains, this imbalance has more gain to the output giving reduced 2nd-harmonic distortion. Differential stages using two of the channels together can reduce this 2nd-harmonic issue enormously by getting back to an essentially gain independent distortion. Relative to alternative amplifiers with < 2mA/ch supply current, the OPA3684 holds much lower distortion at higher frequencies (> 5MHz) and to higher gains. Generally, until the fundamental signal reaches very high frequency or power levels, the 2nd-harmonic will dominate the distortion with a lower 3rd-harmonic component. Focusing then on the 2ndharmonic, increasing the load impedance improves distortion directly. Remember that the total load includes the feedback network—in the noninverting configuration (see Figure 1) this is the sum of RF + RG, while in the inverting configuration it is just RF. Also, providing an additional supply decoupling capacitor (0.1µF) between the supply pins (for bipolar operation) improves the 2nd-order distortion slightly (3dB to 6dB). In most op amps, increasing the output voltage swing increases harmonic distortion directly. A low-power part like the OPA3684 includes quiescent boost circuits to provide the large-signal bandwidth in the Electrical Characteristics. These act to increase the bias in a very linear fashion only when high slew rate or output power is required. This also acts to actually reduce the distortion slightly at higher output power levels. The Typical Characteristic curves show the 2ndharmonic holding constant from 500mVp-p to 5Vp-p outputs while the 3rd-harmonics actually decrease with increasing output power. DRIVING CAPACITIVE LOADS One of the most demanding, and yet very common load conditions, for an op amp is capacitive loading. Often, the capacitive load is the input of an ADC—including additional external capacitance which may be recommended to improve ADC linearity. A high-speed, high open-loop gain amplifier like the OPA3684 can be very susceptible to decreased stability and closed-loop response peaking when a capacitive load is placed directly on the output pin. When the amplifier’s open-loop output resistance is considered, this capacitive load introduces an additional pole in the signal path that can decrease the phase margin. Several external solutions to this problem have been suggested. When the primary considerations are frequency response flatness, pulse response fidelity, and/or distortion, the simplest and most effective solution is to isolate the capacitive load from the feedback loop by inserting a series isolation resistor between the amplifier output and the capacitive load. This does not eliminate the pole from the loop response, but rather shifts it and adds a zero at a higher frequency. The additional zero acts to cancel the phase lag from the capacitive load pole, thus increasing the phase margin and improving stability. The Typical Characteristics show the recommended “RS vs CLOAD” and the resulting frequency response at the load. The 1kΩ resistor shown in parallel with the load capacitor is a measurement path and may be omitted. Parasitic capacitive loads greater than 5pF can begin to degrade the performance of the OPA3684. Long PC board traces, unmatched cables, and connections to multiple devices can easily cause this value to be exceeded. Always consider this effect carefully, 18 OPA3684 www.ti.com SBOS241A The OPA3684 has an extremely low 3rd-order harmonic distortion, particularly for light loads and at lower frequencies. This also gives low 2-tone, 3rd-order intermodulation distortion as shown in the Typical Characteristic curves. Since the OPA3684 includes internal power boost circuits to retain good full-power performance at high frequencies and outputs, it does not show a classical 2-tone, 3rd-order intermodulation intercept characteristic. Instead, it holds relatively low and constant 3rd-order intermodulation spurious levels over power. The Typical Characteristic curves show this spurious level as a dBc below the carrier at fixed center frequencies swept over single-tone power at a matched 50Ω load. These spurious levels drop significantly (> 12dB) for lighter loads than the 100Ω used in the “2-Tone, 3rd-Order Intermodulation Distortion” curve. Converter inputs for instance will see < –82dBc 3rd-order spurious to 10MHz for full-scale inputs. For even lower 3rd-order intermodulation distortion to much higher frequencies, consider the OPA3691 triple or OPA691 and OPA685 single-channel current-feedback amplifiers. The total output spot noise voltage can be computed as the square root of the sum of all squared output noise voltage contributors. Equation 3 shows the general form for the output noise voltage using the terms presented in Figure 12. (3) 2 2 EO =  ENI2 + (IBNR S ) + 4kTRS  NG2 + (IBIRF ) + 4kTRFNG   Dividing this expression by the noise gain (NG = (1+RF/RG)) will give the equivalent input referred spot noise voltage at the noninverting input, as shown in Equation 4. (4) 4kTRF 2 I R  EN = ENI2 + (IBNR S ) + 4kTRS +  BI F  +  NG  NG 2 NOISE PERFORMANCE Wideband current-feedback op amps generally have a higher output noise than comparable voltage-feedback op amps. The OPA3684 offers an excellent balance between voltage and current noise terms to achieve low output noise in a lowpower amplifier. The inverting current noise (17pA/√Hz ) is comparable to most other current-feedback op amps while the input voltage noise (3.7nV/√Hz ) is lower than any unitygain stable, comparable slew rate, voltage-feedback op amp. This low input voltage noise was achieved at the price of higher noninverting input current noise (9.4pA/√Hz ). As long as the AC source impedance looking out of the noninverting node is less than 200Ω, this current noise will not contribute significantly to the total output noise. The op amp input voltage noise and the two input current noise terms combine to give low output noise under a wide variety of operating conditions. Figure 12 shows the op amp noise analysis model with all the noise terms included. In this model, all noise terms are taken to be noise voltage or current density terms in either nV/√Hz or pA/√Hz . Evaluating these two equations for the OPA3684 circuit and component values presented in Figure 1 will give a total output spot noise voltage of 16.3nV/√Hz and a total equivalent input spot noise voltage of 8.1nV/√Hz . This total input referred spot noise voltage is higher than the 3.7nV/√Hz specification for the op amp voltage noise alone. This reflects the noise added to the output by the inverting current noise times the feedback resistor. As the gain is increased, this fixed output noise power term contributes less to the total output noise and the total input referred voltage noise given by Equation 3 will approach just the 3.7nV/√Hz of the op amp itself. For example, going to a gain of +20 in the circuit of Figure 1, adjusting only the gain resistor to 42.1Ω, will give a total input referred noise of 3.9nV/√Hz . A more complete description of op amp noise analysis can be found in the Texas Instruments application note, AB-103, “Noise Analysis for High-Speed Op Amps” (SBOA066), located at www.ti.com. DC ACCURACY AND OFFSET CONTROL A current-feedback op amp like the OPA3684 provides exceptional bandwidth in high gains, giving fast pulse settling but only moderate DC accuracy. The Electrical Specifications show an input offset voltage comparable to high slew rate voltage-feedback amplifiers. The two input bias currents, however, are somewhat higher and are unmatched. Whereas bias current cancellation techniques are very effective with most voltage-feedback op amps, they do not generally reduce the output DC offset for wideband current-feedback op amps. Since the two input bias currents are unrelated in both magnitude and polarity, matching the source impedance looking out of each input to reduce their error contribution to the output is ineffective. Evaluating the configuration of Figure 1, using worst-case +25°C input offset voltage and the two input bias currents, gives a worst-case output offset range equal to: ±(NG • VOS(MAX)) + (IBN • RS/2 • NG) ± (IBI • RF) where NG = noninverting signal gain = ±(2 • 3.9mV) ± (12µA • 25Ω • 2) ± (800Ω • 17µA) ENI 1/3 OPA3684 IBN RF √ 4kTRS RG 4kT RG IBI √ 4kTRF 4kT = 1.6E –20J at 290°K RS EO ERS FIGURE 12. Op Amp Noise Analysis Model. = ±7.8mV + 0.6mV ± 13.6mV = ±22mV OPA3684 SBOS241A www.ti.com 19 While the last term, the inverting bias current error, is dominant in this low-gain circuit, the input offset voltage will become the dominant DC error term as the gain exceeds 5V/V. Where improved DC precision is required in a highspeed amplifier, consider the OPA656 unity gain stable and OPA657 high-gain bandwidth JFET input op amps. DISABLE OPERATION The OPA3684 provides an optional disable feature on each channel that may be used to reduce system power when channel operation is not required. If the V DIS control pin is left unconnected, each channel of the OPA3684 will operate normally. To disable, the control pin must be asserted low. Figure 13 shows a simplified internal circuit for the disable control feature. appear as the impedance looking back into the output, but the circuit will still show very high forward and reverse isolation. If configured as an inverting amplifier, the input and output will be connected through the feedback network resistance (RF + RG) giving relatively poor input to output isolation. Each channel of the OPA3684 provides very high power gain on low quiescent current levels. When disabled, internal high impedance nodes discharge slowly which, with the exceptional power gain provided, give a self powering characteristic that leads to a slow turn off characteristic. Typical full turnoff times to rated 100µA disabled supply current are 4ms. Turn-on times are very fast—less than 40ns. The circuit of Figure 13 will control the disable feature using standard 5V CMOS or TTL level signals when the OPA3684 is operated on ±5V or single +5V supplies. Since this circuit is really a current mode control, disable operation for a single +12V supply should be implemented using an open collector logic family. THERMAL ANALYSIS +VS 40kΩ Q1 The OPA3684 will not require external heatsinking for most applications. Maximum desired junction temperature will set the maximum allowed internal power dissipation as described below. In no case should the maximum junction temperature be allowed to exceed 175°C. 250kΩ 25kΩ VDIS IS Control –VS FIGURE 13. Simplified Disable Control Circuit. In normal operation, base current to Q1 is provided through the 250kΩ resistor while the emitter current through the 40kΩ resistor sets up a voltage drop that is inadequate to turn on the two diodes in Q1’s emitter. As V DIS is pulled low, additional current is pulled through the 40kΩ resistor eventually turning on these two diodes (≈ 30µA). At this point, any further current pulled out of V DIS goes through those diodes holding the emitter-base voltage of Q1 at approximately 0V. This shuts off the collector current out of Q1, turning the amplifier off. The supply current in the disable mode are only those required to operate the circuit of Figure 13. When disabled, the output and input nodes go to a high impedance state. If the OPA3684 is operating in a gain of +1 (with a 800Ω feedback resistor still required for stability), this will show a very high impedance (1.7pF || 1MΩ) at the output and exceptional signal isolation. If operating at a gain greater than +1, the total feedback network resistance (RF + RG) will Operating junction temperature (TJ) is given by TA + PD • θJA. The total internal power dissipation (PD) is the sum of quiescent power (PDQ) and additional power dissipated in the output stage (PDL) to deliver load power. Quiescent power is simply the specified no-load supply current times the total supply voltage across the part. PDL will depend on the required output signal and load but would, for a grounded resistive load, be at a maximum when the output is fixed at a voltage equal to 1/2 either supply voltage (for equal bipolar supplies). Under this condition PDL = VS2/(4 • RL) where RL includes feedback network loading. Note that it is the power in the output stage and not into the load that determines internal power dissipation. As an absolute worst-case example, compute the maximum TJ using an OPA3684IDBQ (SSOP-16 package) in the circuit of Figure 1 operating at the maximum specified ambient temperature of +85°C with all channels driving a grounded 100Ω load. PD = 10V • 5.6mA + 3 • (52 /(4 • (100Ω  1.6kΩ))) = 255mW Maximum TJ = +85°C + (0.255W • 100°C/W) = 111°C. This maximum operating junction temperature is well below most system level targets. Most applications will be lower than this since an absolute worst-case output stage power was assumed in this calculation with all 3 channels running maximum output power simultaneously. 20 OPA3684 www.ti.com SBOS241A BOARD LAYOUT GUIDELINES Achieving optimum performance with a high-frequency amplifier like the OPA3684 requires careful attention to board layout parasitics and external component types. Recommendations that will optimize performance include: a) Minimize parasitic capacitance to any AC ground for all of the signal I/O pins. Parasitic capacitance on the output and inverting input pins can cause instability; on the noninverting input, it can react with the source impedance to cause unintentional bandlimiting. To reduce unwanted capacitance, a window around the signal I/O pins should be opened in all of the ground and power planes around those pins. Otherwise, ground and power planes should be unbroken elsewhere on the board. Minimize the distance (< 0.25") from the power-supply pins to high-frequency 0.1µF decoupling capacitors. At the device pins, the ground and power-plane layout should not be in close proximity to the signal I/O pins. Avoid narrow power and ground traces to minimize inductance between the pins and the decoupling capacitors. The power-supply connections should always be decoupled with these capacitors. An optional supply decoupling capacitor (0.01µF) across the two power supplies (for bipolar operation) will improve 2nd-harmonic distortion performance. Larger (2.2 µ F to 6.8 µ F) decoupling capacitors, effective at lower frequencies, should also be used on the main supply pins. These may be placed somewhat farther from the device and may be shared among several devices in the same area of the PC board. Careful selection and placement of external components will preserve the high-frequency performance of the OPA3684. Resistors should be a very low reactance type. Surface-mount resistors work best and allow a tighter overall layout. Metal film and carbon composition axially-leaded resistors can also provide good highfrequency performance. Again, keep their leads and PCboard trace length as short as possible. Never use wirewound type resistors in a high-frequency application. Since the output pin and inverting input pin are the most sensitive to parasitic capacitance, always position the feedback and series output resistor, if any, as close as possible to the output pin. The quad amplifier pinout allows each output and inverting input to be connected by the feedback element with virtually no trace length. Other network components, such as noninverting input termination resistors, should also be placed close to the package. The frequency response is primarily determined by the feedback resistor value as described previously. Increasing its value will reduce the peaking at higher gains, while decreasing it will give a more peaked frequency response at lower gains. The 800Ω feedback resistor used in the Typical Characteristics at a gain of +2 on ±5V supplies is a good starting point for d) design. Note that a 800Ω feedback resistor, rather than a direct short, is required for the unity-gain follower application. A current-feedback op amp requires a feedback resistor even in the unity-gain follower configuration to control stability. Connections to other wideband devices on the board may be made with short direct traces or through onboard transmission lines. For short connections, consider the trace and the input to the next device as a lumped capacitive load. Relatively wide traces (50mils to 100mils) should be used, preferably with ground and power planes opened up around them. Estimate the total capacitive load and set RS from the plot of recommended “ R S v s C LOAD ” . Low parasitic capacitive loads (< 5pF) may not need an RS since the OPA3684 is nominally compensated to operate with a 2pF parasitic load. If a long trace is required, and the 6dB signal loss intrinsic to a doubly-terminated transmission line is acceptable, implement a matched impedance transmission line using microstrip or stripline techniques (consult an ECL design handbook for microstrip and stripline layout techniques). A 50Ω environment is normally not necessary on board, and in fact a higher impedance environment will improve distortion, see the distortion versus load plots. With a characteristic board trace impedance defined based on board material and trace dimensions, a matching series resistor into the trace from the output of the OPA3684 is used, as well as a terminating shunt resistor at the input of the destination device. Remember also that the terminating impedance will be the parallel combination of the shunt resistor and the input impedance of the destination device; this total effective impedance should be set to match the trace impedance. The high output voltage and current capability of the OPA3684 allows multiple destination devices to be handled as separate transmission lines, each with their own series and shunt terminations. If the 6dB attenuation of a doubly-terminated transmission line is unacceptable, a long trace can be series-terminated at the source end only. Treat the trace as a capacitive load in this case and set the series resistor value as shown in the plot of “RS vs CLOAD”. This will not preserve signal integrity as well as a doubly-terminated line. If the input impedance of the destination device is LOW, there will be some signal attenuation due to the voltage divider formed by the series output into the terminating impedance. Socketing a high-speed part like the OPA3684 is not recommended. The additional lead length and pin-topin capacitance introduced by the socket can create an extremely troublesome parasitic network which can make it almost impossible to achieve a smooth, stable frequency response. Best results are obtained by soldering the OPA3684 onto the board. b) c) e) OPA3684 SBOS241A www.ti.com 21 INPUT AND ESD PROTECTION The OPA3684 is built using a very high-speed complementary bipolar process. The internal junction breakdown voltages are relatively low for these very small geometry devices. These breakdowns are reflected in the Absolute Maximum Ratings table where an absolute maximum 13V across the supply pins is reported. All device pins have limited ESD protection using internal diodes to the power supplies, as shown in Figure 14. These diodes provide moderate protection to input overdrive voltages above the supplies as well. The protection diodes can typically support 30mA continuous current. Where higher currents are possible (e.g., in systems with ±15V supply parts driving into the OPA3684), current-limiting series resistors should be added into the two inputs. Keep these resistor values as low as possible since high values degrade both noise performance and frequency response. +V CC External Pin Internal Circuitry –V CC FIGURE 14. Internal ESD Protection. 22 OPA3684 www.ti.com SBOS241A PACKAGE DRAWINGS D (R-PDSO-G**) 8 PINS SHOWN 0.050 (1,27) 8 5 0.020 (0,51) 0.014 (0,35) 0.010 (0,25) PLASTIC SMALL-OUTLINE PACKAGE 0.244 (6,20) 0.228 (5,80) 0.157 (4,00) 0.150 (3,81) 0.008 (0,20) NOM Gage Plane 1 A 4 0°– 8° 0.044 (1,12) 0.016 (0,40) 0.010 (0,25) Seating Plane 0.069 (1,75) MAX 0.010 (0,25) 0.004 (0,10) 0.004 (0,10) PINS ** DIM A MAX A MIN 8 0.197 (5,00) 0.189 (4,80) 14 0.344 (8,75) 0.337 (8,55) 16 0.394 (10,00) 0.386 (9,80) 4040047/E 09/01 NOTES: A. B. C. D. All linear dimensions are in inches (millimeters). This drawing is subject to change without notice. Body dimensions do not include mold flash or protrusion, not to exceed 0.006 (0,15). Falls within JEDEC MS-012 OPA3684 SBOS241A www.ti.com 23 PACKAGE DRAWINGS (Cont.) DBQ (R-PDSO-G**) 24 PINS SHOWN 0.012 (0,30) 0.008 (0,20) 13 PLASTIC SMALL-OUTLINE 0.025 (0,64) 24 0.005 (0,13) M 0.244 (6,20) 0.228 (5,80) 0.157 (3,99) 0.150 (3,81) 0.008 (0,20) NOM 1 A 12 Gage Plane 0.010 (0,25) 0°– 8° 0.069 (1,75) MAX 0.035 (0,89) 0.016 (0,40) Seating Plane 0.010 (0,25) 0.004 (0,10) PINS ** DIM A MAX 0.004 (0,10) 16 0.197 (5,00) 0.188 (4,78) 20 0.344 (8,74) 0.337 (8,56) 24 0.344 (8,74) 0.337 (8,56) 28 0.394 (10,01) 0.386 (9,80) 4073301/E 10/00 A MIN NOTES: A. B. C. D. All linear dimensions are in inches (millimeters). This drawing is subject to change without notice. Body dimensions do not include mold flash or protrusion not to exceed 0.006 (0,15). Falls within JEDEC MO-137 24 OPA3684 www.ti.com SBOS241A PACKAGE OPTION ADDENDUM www.ti.com 22-Feb-2005 PACKAGING INFORMATION Orderable Device OPA3684ID OPA3684IDBQ OPA3684IDBQR OPA3684IDBQT OPA3684IDR (1) Status (1) ACTIVE PREVIEW ACTIVE ACTIVE ACTIVE Package Type SOIC SSOP/ QSOP SSOP/ QSOP SSOP/ QSOP SOIC Package Drawing D DBQ DBQ DBQ D Pins Package Eco Plan (2) Qty 14 16 16 16 14 58 1 2500 250 2500 None None None None None Lead/Ball Finish CU NIPDAU Call TI CU NIPDAU CU NIPDAU CU NIPDAU MSL Peak Temp (3) Level-3-235C-168 HR Call TI Level-3-235C-168 HR Level-3-235C-168 HR Level-3-235C-168 HR The marketing status values are defined as follows: ACTIVE: Product device recommended for new designs. LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect. NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design. PREVIEW: Device has been announced but is not in production. Samples may or may not be available. OBSOLETE: TI has discontinued the production of the device. (2) Eco Plan - May not be currently available - please check http://www.ti.com/productcontent for the latest availability information and additional product content details. None: Not yet available Lead (Pb-Free). Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements for all 6 substances, including the requirement that lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes. Green (RoHS & no Sb/Br): TI defines "Green" to mean "Pb-Free" and in addition, uses package materials that do not contain halogens, including bromine (Br) or antimony (Sb) above 0.1% of total product weight. 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