OPA658N

® OPA 658 OPA658 OPA 658 Wideband, Low Power Current Feedback OPERATIONAL AMPLIFIER FEATURES q UNITY GAIN STABLE BANDWIDTH: 900MHz q LOW POWER: 50mW q LOW DIFFERENTIAL GAIN/PHASE ERRORS: 0.025%/0.02° q HIGH SLEW RATE: 1700V/µs q GAIN FLATNESS: 0.1dB to 135MHz q HIGH OUTPUT CURRENT (80mA) APPLICATIONS q MEDICAL IMAGING q HIGH-RESOLUTION VIDEO q HIGH-SPEED SIGNAL PROCESSING q COMMUNICATIONS q PULSE AMPLIFIERS q ADC/DAC GAIN AMPLIFIER q MONITOR PREAMPLIFIER q CCD IMAGING AMPLIFIER DESCRIPTION The OPA658 is an ultra-wideband, low power current feedback video operational amplifier featuring high slew rate and low differential gain/phase error. The current feedback design allows for superior large signal bandwidth, even at high gains. The low differential gain/phase errors, wide bandwidth and low quiescent current make the OPA658 a perfect choice for numerous video, imaging and communications applications. The OPA658 is optimized for low gain operation and is also available in dual (OPA2658) and quad (OPA4658) configurations. +VS Current Mirror IBIAS In+ In– CCOMP Buffer VOUT IBIAS Current Mirror –VS International Airport Industrial Park • Mailing Address: PO Box 11400, Tucson, AZ 85734 • Street Address: 6730 S. Tucson Blvd., Tucson, AZ 85706 • Tel: (520) 746-1111 • Twx: 910-952-1111 Internet: http://www.burr-brown.com/ • FAXLine: (800) 548-6133 (US/Canada Only) • Cable: BBRCORP • Telex: 066-6491 • FAX: (520) 889-1510 • Immediate Product Info: (800) 548-6132 ® © 1994 Burr-Brown Corporation PDS-1268F 1 Printed in U.S.A. OPA658 March, 1998 SPECIFICATIONS At TA = +25°C, VS = ±5V, RL = 100Ω, and RFB = 402Ω, unless otherwise noted. OPA658P, U, N PARAMETER FREQUENCY RESPONSE Closed-Loop Bandwidth(2) CONDITION G = +1(4) G = +2 G = +5 G = +10 G = +2, 2V Step G = +2, 2V Step G = +2, 2V Step G = +2, 2V Step f = 5MHz, G = +2, VO = 2Vp-p f = 20MHz, G= +2, VO = 2Vp-p f = 10MHz, 4dBm Each Tone G = +2, NTSC, VO = 1.4Vp-p, RL = 150Ω G = +2, NTSC, VO = 1.4Vp-p, RL = 150Ω G = +2 VCM = 0V VS = ±4.7 to ±5.5V VCM = 0V VCM = 0V 55 MIN TYP 900 680 370 200 1700 1500 15 11.5 6 68 56 40 0.025 0.02 135(5) ±3 ±5 64 ±5.7 ±10 ±1.1 ±30 ±5.5 ±8 58 ±30 ±80 ±35 ±75 MAX OPA658UB, NB MIN TYP T(1) T T T T T T T T T T T T T T ±2 ±4 67 T T T T ±4.5 ±7 MAX UNITS MHz MHz MHz MHz V/µs V/µs ns ns ns dBc dBc dBm % degrees MHz mV mV dB µA µA µA µA 400 Slew Rate(3) At Minimum Specified Temperature Settling Time: 0.01% 0.1% 1% Spurious Free Dynamic Range Third Order Intercept Point Differential Gain Differential Phase Bandwidth for 0.1dB Flatness OFFSET VOLTAGE Input Offset Voltage Over Temperature Range Power Supply Rejection Ratio INPUT BIAS CURRENT Non-Inverting Over Temperature Range Inverting Over Temperature Range NOISE Input Voltage Noise Density f = 100Hz f = 2kHz f = 10kHz f = 1MHz fB = 100Hz to 200MHz Input Bias Current Noise Density Inverting: f = 1MHz Non-Inverting: f = 1MHz INPUT VOLTAGE RANGE Common-Mode Input Range Over Temperature Range Common-Mode Rejection INPUT IMPEDANCE Non-Inverting Inverting OPEN-LOOP TRANSRESISTANCE Open-Loop Transresistance Over Temperature Range OUTPUT Voltage Output Over Temperature Range Voltage Output Over Temperature Range Voltage Output Over Temperature Range Output Current, Sourcing Over Temperature Output Current, Sinking Over Temperature Short Circuit Current Output Resistance POWER SUPPLY Specified Operating Voltage Operating Voltage Range Quiescent Current Over Temperature Range TEMPERATURE RANGE Specification: P, U, N, UB, NB Thermal Resistance, θJA P 8-Pin DIP U SO-8 N SOT23-5 1000 900 ±18 ±35 T T 16 4.9 3.2 3.2 45.3 32 11.9 T T T T T T T nV/√Hz nV/√Hz nV/√Hz nV/√Hz µVrms pA/√Hz pA/√Hz VCM = ±1V ±2.5 45 ±2.9 50 500 || 1 50 T T T T T T V dB kΩ || pF Ω kΩ kΩ V V V V V V mA mA mA mA mA Ω V V mA mA °C VO = ±2V, RL = 100Ω VO = ±2V, RL = 100Ω No Load RL = 250Ω RL = 100Ω 150 100 ±2.7 ±2.5 ±2.7 ±2.5 ±2.2 ±2.0 80 70 60 35 190 200 150 T T T T T T T T T T 250 ±2.9 ±2.75 ±2.9 ±2.7 ±2.8 ±2.5 120 80 150 0.02 ±5 ±5 ±5.5 T T T T T T T T T T T 0.1MHz, G = +2 VS = ±5V ±4.5 ±5.5 ±7.75 ±8.5 +85 T ±4.5 ±4.7 T T T T T ±5.75 ±6.5 T °C/W °C/W °C/W –40 100 125 150 NOTES: (1) An asterisk (T) specifies the same value as the grade to the left. (2) Frequency response can be strongly influenced by PC board parasitics. The demonstration boards show low parasitic layouts for this part. Refer to the demonstration board layout for details. (3) Slew rate is rate of change from 10% to 90% of output voltage step. (4) At G = +1, RFB = 560Ω for PDIP and 402Ω for SO-8. (5) This specification is PC board layout dependent. ® OPA658 2 ABSOLUTE MAXIMUM RATINGS Supply ............................................................................................... ±5.5V Internal Power Dissipation .......................... See Thermal Considerations Differential Input Voltage .................................................................. ±1.2V Input Voltage Range ............................................................................ ±VS Storage Temperature Range: P, U, UB, N, NB ............ –40°C to +125°C Lead Temperature (soldering, 10s) .............................................. +300°C (soldering, SOIC 3s) ...................................................................... +260°C Junction Temperature (TJ ) ............................................................ +175°C PIN CONFIGURATION Top View DIP/SO-8 NC –Input +Input –VS 1 2 3 4 8 7 6 5 NC +VS Output NC ELECTROSTATIC DISCHARGE SENSITIVITY Electrostatic discharge can cause damage ranging from performance degradation to complete device failure. Burr-Brown Corporation recommends that all integrated circuits be handled and stored using appropriate ESD protection methods. ESD damage can range from subtle performance degradation to complete device failure. Precision integrated circuits may be more susceptible to damage because very small parametric changes could cause the device not to meet published specifications. SOT23-5 Output 1 5 +VS –VS +Input 2 3 4 –Input PACKAGE/ORDERING INFORMATION PACKAGE DRAWING NUMBER(1) 182 182 331 331 006 TEMPERATURE RANGE –40°C to +85°C –40°C to +85°C –40°C to +85°C –40°C to +85°C –40°C to +85°C PACKAGE MARKING(2) OPA658U OPA658UB A58 A58B OPA658P ORDERING NUMBER(3) OPA658U OPA658UB OPA658N-250 OPA658N-3k OPA658NB-250 OPA658NB-3k OPA658P PRODUCT OPA658U OPA658UB OPA658N OPA658NB OPA658P PACKAGE SO-8 Surface Mount SO-8 Surface Mount 5-pin SOT23-5 5-pin SOT23-5 8-Pin Plastic DIP NOTE: (1) For detailed drawing and dimension table, please see end of data sheet, or Appendix C of Burr-Brown IC Data Book. (2) The “B” grade of the SO-8 will be marked with a “B” by pin 8. The “B” grade of the SOT23-5 will be marked with a “B” near pins 3 and 4. (3) The SOT23-5 is only available on a 7" tape and reel (e.g. ordering 250 pieces of “OPA658N-250” will get a single 250 piece tape and reel. Ordering 3000 pieces of “OPA658N-3k” will get a single 3000 piece tape and reel). Please refer to Appendix B of Burr-Brown IC Data Book for detailed Tape and Reel Mechanical information. The information provided herein is believed to be reliable; however, BURR-BROWN assumes no responsibility for inaccuracies or omissions. BURR-BROWN assumes no responsibility for the use of this information, and all use of such information shall be entirely at the user’s own risk. Prices and specifications are subject to change without notice. No patent rights or licenses to any of the circuits described herein are implied or granted to any third party. BURR-BROWN does not authorize or warrant any BURR-BROWN product for use in life support devices and/or systems. ® 3 OPA658 TYPICAL PERFORMANCE CURVES At TA = +25°C, VS = ±5V, RL = 100Ω, and RFB = 402Ω, unless otherwise noted. PSRR AND CMR vs TEMPERATURE 75 55 COMMON-MODE REJECTION vs INPUT COMMON-MODE VOLTAGE Common-Mode Rejection (dB) 70 PSRR , CMR (dB) PSRR 50 45 40 35 30 25 65 60 55 50 45 –75 PSR+ PSR– CMR –50 –25 0 25 50 75 100 125 –4 –3 –2 –1 0 1 2 3 4 Temperature (°C) Common-Mode Voltage (V) SUPPLY CURRENT vs TEMPERATURE 120 OUTPUT CURRENT vs TEMPERATURE IO+ Supply Current (±mA) 5 Output Current (±mA) 110 100 90 4 80 IO– 70 –75 –50 –25 0 25 50 75 100 125 –75 –50 –25 0 25 50 75 100 125 Ambient Temperature (°C) Ambient Temperature (°C) OUTPUT SWING vs TEMPERATURE 3.10 3.0 Output Swing (V) NON-INVERTING INPUT BIAS CURRENT vs TEMPERATURE RL = 250Ω +VO –VO Non-Inverting Input Bias Current IB+ (µA) 100 3.20 10 8 2.90 2.80 2.70 2.60 2.50 2.40 2.30 –60 –40 –20 0 20 6 –VO +VO RL = 100Ω 4 2 –75 –50 –25 0 25 50 75 100 125 Ambient Temperature (°C) 40 60 80 Temperature (°C) ® OPA658 4 TYPICAL PERFORMANCE CURVES (CONT) At TA = +25°C, VS = ±5V, RL = 100Ω, and RFB = 402Ω, unless otherwise noted. INVERTING INPUT BIAS CURRENT vs TEMPERATURE 2.0 OPEN-LOOP TRANSIMPEDANCE AND PHASE vs FREQUENCY 106 Transimpedance 105 0 –45 Phase 103 102 101 1 –90 –135 –180 –225 1k 10k 100k 1M 10M Frequency (Hz) 100M 1G Inverting Input Bias Current IB– (µA) 1.8 Transimpedance (Ω) 1.6 1.4 1.2 1.0 0.8 0.6 0.4 –75 –50 –25 0 25 50 75 100 125 Temperature (°C) 104 OPEN-LOOP GAIN AND PHASE vs FREQUENCY 60 40 Open-Loop Gain (dB) CLOSED-LOOP BANDWIDTH 6 SO-8 Bandwidth = 881MHz, RFB = 402Ω 3 Open-Loop Phase (°) Gain Phase 0 –45 –90 –135 –180 –225 20 0 –20 –40 –60 1k 10k 100k 1M 10M 100M 1G Frequency (Hz) G = +1 Gain (dB) 0 –3 DIP Bandwidth = 949MHz, RFB = 560Ω –6 –9 1M 10M 100M Frequency (Hz) 1G CLOSED-LOOP BANDWIDTH 9 G = +2 6 DIP Bandwidth = 682MHz 20 CLOSED-LOOP BANDWIDTH G = +5 17 SO-8/DIP Bandwidth= 372MHz 14 Gain (dB) 3 Gain (dB) 11 8 5 2 0 SO-8 Bandwidth = 680MHz –3 –6 1M 10M 100M Frequency (Hz) 1G 1M 10M 100M Frequency (Hz) 1G Open-Loop Phase (°) ® 5 OPA658 TYPICAL PERFORMANCE CURVES At TA = +25°C, VS = ±5V, RL = 100Ω, and RFB = 402Ω, unless otherwise noted. (CONT) CLOSED-LOOP BANDWIDTH 26 23 G = +10 20 SMALL SIGNAL TRANSIENT RESPONSE 160 120 G = +2 SO-8/DIP Bandwidth = 200MHz Output Voltage (mV) 80 40 0 –40 –80 –120 –160 Gain (dB) 17 14 11 8 1M 10M 100M Frequency (Hz) 1G Time (5ns/div) RECOMMENDED ISOLATION RESISTANCE vs CAPACITIVE LOAD 40 G = +2 35 LARGE SIGNAL TRANSIENT RESPONSE 1.6 1.2 G = +2 Output Voltage (V) Isolation Resistance 0.8 0.4 0 –0.4 –0.8 –1.2 –1.6 30 25 OPA658 RISO 20 15 10 10 402Ω 402Ω CL 1kΩ 20 30 40 50 60 70 80 90 100 Capacitive Load (pf) Time (5ns/div) HARMONIC DISTORTION vs FREQUENCY –50 5MHz HARMONIC DISTORTION vs OUTPUT SWING –60 –65 3fO 2fO Harmonic Distortion (dBc) –60 Harmonic Distortion (dBc) –70 –75 –80 –85 –90 –95 –100 –70 G = +2 –80 2fO –90 3fO –100 100k 1M 10M Frequency (Hz) 100M 0 1 2 Output Swing (Vp-p) 3 4 ® OPA658 6 TYPICAL PERFORMANCE CURVES At TA = +25°C, VS = ±5V, RL = 100Ω, and RFB = 402Ω, unless otherwise noted. (CONT) 10MHz HARMONIC DISTORTION vs OUTPUT SWING –60 –60 HARMONIC DISTORTION vs TEMPERATURE (VO = 2Vp-p, G = +2) Harmonic Distortion (dBc) Harmonic Distortion (dBc) –70 2fO –80 3fO –90 –65 3fO –70 –75 2fO –80 –100 0.01 –85 0.1 1 4V 10 –75 –50 –25 0 25 50 75 100 125 Output Swing (Vp-p) Temperature (°C) HARMONIC DISTORTION vs GAIN (fO = 5MHz, VO = 2Vp-p) –50 INPUT VOLTAGE AND CURRENT NOISE vs FREQUENCY 100 Harmonic Distortion (dBc) Voltage Noise (nV/√Hz) Current Noise (pA/√Hz) –55 2fO Inverting Current Noise 10 Non-Inverting Noise –60 –65 3fO –70 Voltage Noise 1 0 1 2 3 4 5 6 7 8 9 10 –75 Non-Inverting Gain (V/V) 102 103 104 105 106 107 Frequency (Hz) ® 7 OPA658 APPLICATIONS INFORMATION THEORY OF OPERATION Conventional op amps depend on feedback to drive their inputs to the same potential, however the current feedback op amp’s inverting and non-inverting inputs are connected by a unity gain buffer, thus enabling the inverting input to automatically assume the same potential as the non-inverting input. This results in very low impedance at the inverting input to sense the feedback as an error current signal. DISCUSSION OF PERFORMANCE The OPA658 is a low-power, unity gain stable, current feedback operational amplifier which operates on ±5V power supply. The current feedback architecture offers the following important advantages over voltage feedback architectures: (1) the high slew rate allows the large signal performance to approach the small signal performance, and (2) there is very little bandwidth degradation at higher gain settings. The current feedback architecture of the OPA658 provides the traditional strength of excellent large signal response plus wide bandwidth, making it a good choice for use in high resolution video, medical imaging and DAC I/V Conversion. The low power requirements make it an excellent choice for numerous portable applications. DC GAIN TRANSFER CHARACTERISTICS The circuit in Figure 1 shows the equivalent circuit for calculating the DC gain. When operating the device in the inverting mode, the input signal error current (IE) is amplified by the open loop transimpedance gain (TO). The output signal generated is equal to TO x IE. Negative feedback is applied through RFB such that the device operates at a gain equal to –RFB/RFF. For non-inverting operation, the input signal is applied to the non-inverting (high impedance buffer) input. The output (buffer) error current (IE) is generated at the low impedance inverting input. The signal generated at the output is fed back to the inverting input such that the overall gain is (1 + RFB/RFF). Where a voltage-feedback amplifier has two symmetrical high impedance inputs, a current feedback amplifier has a low inverting (buffer output) impedance and a high non-inverting (buffer input) impedance. The closed-loop gain for the OPA658 can be calculated using the following equations: R  –  FB   R FF  Inverting Gain = 1 (1) 1+ Loop Gain  R FB  1 +  R FF  Non−Inverting Gain =  1 1+ Loop Gain (2)     TO  where Loop Gain =    R FB    R FB + R S  1 +  R FF       At higher gains the small value inverting input impedance causes an apparent loss in bandwidth. This can be seen from the equation: ƒ ( A = +2 ) BW x (1. 25) V (3) ƒ ACTUAL BW ≈   RS    R FB 1 +   × 1 +  R FF     R FB     [ ] This loss in bandwidth at high gains can be corrected without affecting stability by lowering the value of the feedback resistor from the specified value of 402Ω. OFFSET VOLTAGE AND NOISE The output offset is the algebraic sum of the input offset voltage and bias current errors. The output offset for noninverting operation is calculated by the following equation: + RFF VN VI C1 IE RS (50Ω) LS CC VO R  (4) Output Offset Voltage = ± Ib N × R N  1 + F B  ± R FF    R FB  V IO  1 +  ± Ib I × R FB R FF   If all terms are divided by the gain (1 + RFB/RFF) it can be observed that input referred offsets improve as gain increases. The effective noise at the output can be determined by taking – TO RFB RFB RFF IbI IbN RN VIO FIGURE 1. Equivalent Circuit. FIGURE 2. Output Offset Voltage Equivalent Circuit. ® OPA658 8 the root sum of the squares of equation (4) and applying the spectral noise values found in the Typical Performance Curve graph section. This applies to noise from the op amp only. Note that both the noise figure (NF) and the equivalent input offset voltages improve as the closed loop gain increases (by keeping RFB fixed and reducing RFF with RN = 0Ω). INCREASING BANDWIDTH AT HIGH GAINS The closed-loop bandwidth can be extended at high gains by reducing the value of the feedback resistor RFB. This bandwidth reduction is caused by the feedback current being split between RS and RFF (refer to Figure 1). As the gain increases (for a fixed RFB), more feedback current is shunted through RFF, which reduces closed-loop bandwidth. CIRCUIT LAYOUT AND BASIC OPERATION Achieving optimum performance with a high frequency amplifier like the OPA658 requires careful attention to layout parasitics and selection of external components. Recommendations for PC board layout and component selection include: a) Minimize parasitic capacitance to any ac ground for all of the signal I/O pins. Parasitic capacitance on the output and inverting input pins can cause instability; on the noninverting input it can react with the source impedance to cause unintentional bandlimiting. To reduce unwanted capacitance, a window around the signal I/O pins should be opened in all of the ground and power planes. Otherwise, ground and power planes should be unbroken elsewhere on the board. b) Minimize the distance (< 0.25") from the two power pins to high frequency 0.1µF decoupling capacitors. At the pins, the ground and power plane layout should not be in close proximity to the signal I/O pins. Avoid narrow power and ground traces to minimize inductance between the pins and the decoupling capacitors. Larger (2.2µF to 6.8µF) decoupling capacitors, effective at lower frequencies, should also be used. These may be placed somewhat farther from the device and may be shared among several devices in the same area of the PC board. c) Careful selection and placement of external components will preserve the high frequency performance of the OPA658. Resistors should be a very low reactance type. Surface mount resistors work best and allow a tighter overall layout. Metal film or carbon composition axially-leaded resistors can also provide good high frequency performance. Again, keep their leads as short as possible. Never use wirewound type resistors in a high frequency application. Since the output pin and the inverting input pin are most sensitive to parasitic capacitance, always position the feedback and series output resistor, if any, as close as possible to the package pins. Other network components, such as noninverting input termination resistors, should also be placed close to the package. The feedback resistor value acts as the frequency response compensation element for a current feedback type amplifier. The 402Ω used in setting the specification achieves a nominal maximally flat butterworth response while assuming a 2pF output pin parasitic. Increasing the feedback resistor will over compensate the amplifier, rolling off the frequency response, while decreasing it will decrease phase margin, peaking up the frequency response. Note that a non-inverting, unity gain buffer application still requires a feedback resistor for stability (560Ω for SO-8, 402Ω for PDIP, and 324Ω for SOT23). d) Connections to other wideband devices on the board may be made with short direct traces or through on-board transmission lines. For short connections, consider the trace and the input to the next device as a lumped capacitive load. Relatively wide traces (50 to 100 mils) should be used, preferably with ground and power planes opened up around them. Estimate the total capacitive load and set RISO from the plot of recommended RISO vs capacitive load. Low parasitic loads may not need an RISO since the OPA658 is nominally compensated to operate with a 2pF parasitic load. If a long trace is required and the 6dB signal loss intrinsic to doubly terminated transmission lines is acceptable, implement a matched impedance transmission line using microstrip or stripline techniques (consult an ECL design handbook for microstrip and stripline layout techniques). A 50Ω environment is not necessary on board, and in fact a higher impedance environment will improve distortion as shown in the distortion vs load plot. With a characteristic impedance defined based on board material and desired trace dimensions, a matching series resistor into the trace from the output of the amplifier is used as well as a terminating shunt resistor at the input of the destination device. Remember also that the terminating impedance will be the parallel combination of the shunt resistor and the input impedance of the destination device; the total effective impedance should match the trace impedance. Multiple destination devices are best handled as separate transmission lines, each with their own series and shunt terminations. If the 6dB attenuation loss of a doubly terminated line is unacceptable, a long trace can be series-terminated at the source end only. This will help isolate the line capacitance from the op amp output, but will not preserve signal integrity as well as a doubly terminated line. If the shunt impedance at the destination end is finite, there will be some signal attenuation due to the voltage divider formed by the series and shunt impedances. e) Socketing a high speed part like the OPA658 is not recommended. The additional lead length and pin-to-pin capacitance introduced by the socket creates an extremely troublesome parasitic network which can make it almost impossible to achieve a smooth, stable response. Best results are obtained by soldering the part onto the board. If socketing for the DIP package is desired, high frequency flush mount pins (e.g., McKenzie Technology #710C) can give good results. The OPA658 is nominally specified for operation using ±5V power supplies. A 10% tolerance on the supplies, or an ECL –5.2V for the negative supply, is within the maximum ® 9 OPA658 Output Impedance (Ω) specified total supply voltage of 11V. Higher supply voltages can break down internal junctions possibly leading to catastrophic failure. Single supply operation is possible as long as common mode voltage constraints are observed. The common mode input and output voltage specifications can be interpreted as a required headroom to the supply voltage. Observing this input and output headroom requirement will allow non-standard or single supply operation. Figure 3 shows one approach to single-supply operation. +VS VS 2 VAC OPA658 RL +VS V VOUT = S + AV VAC 2 ROUT 100 10 1 0.1 G = +2 0.01 0.001 10k 100k 1M Frequency (Hz) 10M 100M FIGURE 4. Closed-Loop Output Impedance vs Frequency. THERMAL CONSIDERATIONS The OPA658 will not require heatsinking under most operating conditions. Maximum desired junction temperature will set a maximum allowed internal power dissipation as described below. In no case should the maximum junction temperature be allowed to exceed 175°C. Operating junction temperature (T J ) is given by TA + PD • θJA. The total internal power dissipation (PD) is the sum of quiescent power (PDQ) and additional power dissipated in the output stage (PDL) to deliver load power. Quiescent power is simply the specified no-load supply current times the total supply voltage across the part. PDL will depend on the required output signal and load but would, for a grounded resistive load, be at a maximum when the output is fixed at a voltage equal to 1/2 either supply voltage (for equal bipolar supplies). Under this condition PDL = VS2/(4 • RL) where RL includes feedback network loading. Note that it is the power in the output stage and not into the load that determines internal power dissipation. As an example, compute the maximum TJ for an OPA658N at AV = +2, RL = 100Ω, RFB = 402Ω, ±VS = ±5V, and the specified maximum TA = +85°C. PD = 10V • 8.5mA + 52/ [4 • (100Ω || 804Ω)] = 155mW. Maximum TJ = 85°C + 0.155W • 150°C/W = 108°C. DRIVING CAPACITIVE LOADS The OPA658’s output stage has been optimized to drive low resistive loads. Capacitive loads, however, will decrease the amplifier’s phase margin which may cause high frequency peaking or oscillations. Capacitive loads greater than 5pF should be buffered by connecting a small resistance, usually 10Ω to 35Ω, in series with the output as shown in Figure 5. This is particularly important when driving high capacitance loads such as flash A/D converters. In general, capacitive loads should be minimized for optimum high frequency performance. Coax lines can be driven if the cable is properly terminated. The capacitance of coax cable (29pF/foot for RG-58) will not load the amplifier when the coaxial cable or transmission line is terminated with its characteristic impedance. 10 402Ω AV = +2 402Ω FIGURE 3. Single Supply Operation. ESD PROTECTION ESD static damage has been well recognized for MOSFET devices, but any semiconductor device deserves protection from this potentially damaging source. This is particularly true for very high speed, fine geometry processes. ESD static damage can cause subtle changes in amplifier input characteristics without necessarily destroying the device. In precision operational amplifiers, this may cause a noticeable degradation of offset voltage and drift. Therefore, static protection is strongly recommended when handling the OPA658. OUTPUT DRIVE CAPABILITY The OPA658 has been optimized to drive 75Ω and 100Ω resistive loads. The device can drive 2Vp-p into a 75Ω load. This high-output drive capability makes the OPA658 an ideal choice for a wide range of RF, IF, and video applications. In many cases, additional buffer amplifiers are unneeded. Many demanding high-speed applications such as ADC/DAC buffers require op amps with low wideband output impedance. For example, low output impedance is essential when driving the signal-dependent capacitances at the inputs of flash A/D converters. As shown in Figure 4, the OPA658 maintains very low closed-loop output impedance over frequency. Closed-loop output impedance increases with frequency since loop gain is decreasing with frequency. ® OPA658 402Ω 402Ω 10Ω to 35Ω RISO OPA658 50Ω RL CL close-in spurious tones will appear at fO ±3 • ∆f. The two tone, third-order spurious plot shown in Figure 7 indicates how far below these two equal power, closely spaced, tones the intermodulation spurious will be. The single tone power is at a matched 50Ω load. The unique design of the OPA658 provides much greater spurious free range than what a twotone third-order intermodulation intercept specification would predict. This can be seen in Figure 7 as the spurious free range actually increases at the higher output power levels. TWO TONE, THIRD-ORDER SPURIOUS LEVELS Third-Order Spurious Level (dBc) FIGURE 5. Driving Capacitive Loads. COMPENSATION The OPA658 is internally compensated and is stable in unity gain with a phase margin of approximately 62°, and approximately 64° in a gain of +2V/V when used with the recommended feedback resistor value. Frequency response for other gains are shown in the Typical Performance Curves. The high-frequency response of the OPA658 in a good layout is very flat with frequency. DISTORTION The OPA658’s Harmonic Distortion characteristics into a 100Ω load are shown versus frequency and power output in the Typical Performance Curves. Distortion can be further improved by increasing the load resistance as illustrated in Figure 6. Remember to include the contribution of the feedback resistance when calculating the effective load resistance seen by the amplifier. 5MHz HARMONIC DISTORTION vs LOAD RESISTANCE (G = +2) –55 –60 G = +2, VO = 2Vp-p, fO = 5MHz –65 3fO –70 –75 –80 –85 10 100 Load Resistance (Ω) 1k 2fO –65 20MHz –70 –75 10MHz 5MHz –80 –85 –90 –18 –16 –14 –12 –10 –8 –6 –4 –2 0 2 4 Single Tone Power (dBm) FIGURE 7. Third-Order Spurious Level vs Frequency. DIFFERENTIAL GAIN AND PHASE Differential Gain (dG) and Differential Phase (dP) are among the more important specifications for video applications. dG is defined as the percent change in closed-loop gain over a specified change in output voltage level. dP is defined as the change in degrees of the closed-loop phase over the same output voltage change. Both dG and dP are specified at the NTSC sub-carrier frequency of 3.58MHz and the PAL subcarrier of 4.43MHz. All NTSC measurements were performed using a Tektronix model VM700A Video Measurement Set. dG/dP of the OPA658 were measured with the amplifier in a gain of +2V/V with 75Ω input impedance and the output back-terminated in 75Ω. The input signal selected from the generator was a 0V to 1.4V modulated ramp with sync pulse. With these conditions the test circuit shown in Figure 8 delivered a 100IRE modulated ramp to the 75Ω input of the videoanalyzer. The signal averaging feature of the analyzer FIGURE 6. 5MHz Harmonic Distortion vs Load Resistance. 75Ω Harmonic Distortion (dBc) Narrowband communication channel requirements will benefit from the OPA658’s wide bandwidth and low intermodulation distortion on low quiescent power. If output signal power at two closely spaced frequencies is required, third-order nonlinearities in any amplifier will cause spurious power at frequencies very near the two fundamental frequencies. If the two test frequencies, f1 and f2, are specified in terms of average and delta frequency, fO = (f1 + f2)/2 and ∆f =  f2 – f1, the two, third-order, 75Ω OPA658 75Ω 402Ω 75Ω 402Ω TEK TSG 130A TEK VM700A FIGURE 8. Configuration for Testing Differential Gain/Phase. ® 11 OPA658 was used to establish a reference against which the performance of the amplifier was measured. Signal averaging was also used to measure the dg and dp of the test signal in order to eliminate the generator’s contribution to measured amplifier performance. Typical performance of the OPA658 is 0.025% differential gain and 0.02° differential phase to both NTSC and PAL standards. SPICE MODELS AND EVALUATION BOARDS Computer simulation of circuit performance using SPICE is often useful when analyzing the performance of analog circuits and systems. This is particularly true for Video and RF amplifier circuits where parasitic capacitance and inductance can have a major effect on circuit performance. SPICE models are available on a disk from the Burr-Brown Applications Department. Demonstration boards are available for each OPA658 package style. These boards implement a very low parasitic layout that will produce the excellent frequency and pulse responses shown in the Typical Performance Curves. For each package style, the recommended demonstration board is: DEM-OPA65xP DEM-OPA65xU DEM-OPA6xxN 8-Pin DIP for the OPA658P SO-8 for the OPA658U SOT23 for the OPA658N Contact your local Burr-Brown sales office or distributor to order demonstration boards. J2 –In R3 R4 402Ω R5 C1 2.2µF + C3 0.1µF 1 2 P1 +5V GND 2 J1 +In R7 R6 3 7 OPA658 4 C2 0.1µF C4 2.2µF 6 R1 J1 Out 1 GND –5V P2 R5 2 + FIGURE 9. Layout Detail For DEM-OPA65xP Demonstration Board. ® OPA658 12 DEM-OPA65xP Demonstration Board Layout (A) (B) (C) (D) FIGURE 10a. Evaluation Board Silkscreen (Bottom). 10b. Evaluation Board Silkscreen (Top). 10c. Evaluation Board Layout (Solder Side). 10d. Evaluation Board Layout (Layout Side). TYPICAL APPLICATION 402Ω 402Ω 75Ω Transmission Line 75Ω OPA658 Video Input 75Ω 75Ω V OUT FIGURE 11. Low Distortion Video Amplifier. ® 13 OPA658