OPA837IDBVT

Product Folder Order Now Support & Community Tools & Software Technical Documents Reference Design OPA837, OPA2837 SBOS673D – SEPTEMBER 2017 – REVISED DECEMBER 2018 OPAx837 Low-Power, Precision, 105-MHz, Voltage-Feedback Op Amp 1 Features 3 Description • • • • • • • • The OPA837 and OPA2837 are single- and dualchannel unity-gain stable, voltage-feedback amplifiers that provide among the highest MHz per mW of bandwidth versus power among precision op amps. Using only 600 µA per channel on a single 5-V supply, these 3.0-mW devices deliver 105-MHz bandwidth at a gain of 1 V/V. The very low trimmed offset voltage of ±130 µV (maximum) comes with a typical (±1σ) drift of ±0.4 µV/°C. Bandwidth: 105 MHz (AV = 1 V/V) Very Low (Trimmed) Supply Current: 600 µA Gain Bandwidth Product: 50 MHz Slew Rate: 105 V/µs Negative Rail Input, Rail-to-Rail Output Single-Supply Operating Range: 2.7 V to 5.4 V 25°C Input Offset: ±130 µV (Max) Input Offset Voltage Drift (DCK Package): < ±1.6 µV/°C (Maximum) Input Voltage Noise: 4.7 nV/√Hz (> 100 Hz) HD2: –120 dBc at 2 VPP, 100 kHz HD3: –145 dBc at 2 VPP, 100 kHz Settling Time: 35 ns, 0.5-V Step to 0.1% 5-µA Shutdown Current With Fast Recovery From Shutdown for Power Scaling Applications 1 • • • • • 2 Applications • • • • • • • 12-Bit to 16-Bit, Low-Power SAR Drivers Precision ADC Reference Buffers Very-Low-Power Active Filters Low-Power Transimpedance Amplifiers Sensor Signal Conditioning Wearable Devices Low-Side Current Sensing The OPAx837 is characterized for operation over the a wide temperature range of –40°C to +125°C. Device Information(1) PART NUMBER Low-Power, Low-Noise, Precision, Single-Ended SAR ADC Driver With True Ground Input and Output Range + 5.0 V REF5040 Ideally suited to single-ended, successiveapproximation register (SAR) analog-to-digitalconverter (ADC) driving applications, the OPAx837 provides one of the lowest input spot noise levels at 4.7 nV/√Hz for the 3-mW quiescent power. The very high 50-MHz gain bandwidth product provides the low output impedance to high frequencies required to supply the fast charging currents in SAR ADC driver applications. This low dynamic output impedance is also suitable for reference buffer applications with precision ADCs. The single-channel OPA837 is available in a 6-pin SOT-23 package (that includes a power shutdown feature) and a 5-pin SC70 package whereas the dual-channel OPA2837 is available in an 8-pin VSSOP package and a 10-pin WQFN package. 4.096 V OPA837 OPA2837 PACKAGE BODY SIZE (NOM) SOT-23 (6) 2.90 mm × 1.60 mm SC70 (5) 2.00 mm × 1.25 mm VSSOP (8) 3.00 mm × 3.00 mm WQFN (10) 2.00 mm × 2.00 mm (1) For all available packages, see the package option addendum at the end of the data sheet. GND 3.3 V +VCC 0V 0V 4.0 V PD + OPA837 ± 3.8 V Gain = 1.05 V/V ±VCC 22 Ÿ 2.2 nF 499 Ÿ ADS8860 16-Bit SAR 1 MSPS 22 Ÿ 10.0 kŸ GND GND GND ± 0.23 V LM7705 + 3.3 V 1 An IMPORTANT NOTICE at the end of this data sheet addresses availability, warranty, changes, use in safety-critical applications, intellectual property matters and other important disclaimers. PRODUCTION DATA. OPA837, OPA2837 SBOS673D – SEPTEMBER 2017 – REVISED DECEMBER 2018 www.ti.com Table of Contents 1 2 3 4 5 6 Features .................................................................. Applications ........................................................... Description ............................................................. Revision History..................................................... Pin Configuration and Functions ......................... Specifications......................................................... 6.1 6.2 6.3 6.4 6.5 6.6 6.7 6.8 6.9 6.10 7 1 1 1 2 4 5 Absolute Maximum Ratings ...................................... 5 ESD Ratings.............................................................. 5 Recommended Operating Conditions....................... 5 Thermal Information: OPA837 .................................. 6 Thermal Information: OPA2837 ................................ 6 Electrical Characteristics: VS = 5 V........................... 7 Electrical Characteristics: VS = 3 V........................... 9 Typical Characteristics: VS = 5.0 V ......................... 11 Typical Characteristics: VS = 3.0 V ......................... 14 Typical Characteristics: ±2.5-V to ±1.5-V Split Supply ...................................................................... 17 Detailed Description ............................................ 22 7.1 Overview ................................................................. 22 7.2 Functional Block Diagrams ..................................... 22 7.3 Feature Description................................................. 23 7.4 Device Functional Modes........................................ 26 8 Application and Implementation ........................ 30 8.1 Application Information............................................ 30 8.2 Typical Applications ................................................ 39 9 Power Supply Recommendations...................... 43 10 Layout................................................................... 43 10.1 Layout Guidelines ................................................. 43 10.2 Layout Example .................................................... 44 11 Device and Documentation Support ................. 45 11.1 11.2 11.3 11.4 11.5 11.6 11.7 Documentation Support ........................................ Related Links ........................................................ Receiving Notification of Documentation Updates Community Resources.......................................... Trademarks ........................................................... Electrostatic Discharge Caution ............................ Glossary ................................................................ 45 45 45 45 45 45 46 12 Mechanical, Packaging, and Orderable Information ........................................................... 46 4 Revision History NOTE: Page numbers for previous revisions may differ from page numbers in the current version. Changes from Revision C (August 2018) to Revision D Page • Added reference to WQFN (RUN) package in Description section ....................................................................................... 1 • Added OPA2837 RUN package thermal information to document. ....................................................................................... 6 • Changed values of maximum and minimum input-referred offset voltage at 25℃ and across temperature in 5 V and 3 V Electrical Characteristics tables. ...................................................................................................................................... 7 • Changed value of maximum input offset current drift for OPA2837 in 5 V and 3 V Electrical Characteristics tables. ......... 8 • Changed minimum value of CMRR in 5 V Electrical Characteristics table ............................................................................ 8 • Added reference to TIDA-01565 reference design in Power-Down Operation section........................................................ 24 • Added reference to TIDA-01565 reference design in1-Bit PGA Operation section ............................................................. 42 Changes from Revision B (July 2018) to Revision C • Added OPA2837 RUN package to document ........................................................................................................................ 1 Changes from Revision A (April 2018) to Revision B • Page Page Changed input common-mode impedance in 5V and 3V electrical characteristics tables..................................................... 8 Changes from Original (September 2017) to Revision A Page • Added OPA2837 to document ............................................................................................................................................... 1 • Added Single-Supply Operating Range Features bullet......................................................................................................... 1 • Changed 1 SPS to 1 MSPS in front page figure ................................................................................................................... 1 • Added footnote to Pin Functions table .................................................................................................................................. 4 • Changed footnote describing method of computation of slew rate in Electrical Characteristics: VS = 5 V table ................... 7 2 Submit Documentation Feedback Copyright © 2017–2018, Texas Instruments Incorporated Product Folder Links: OPA837 OPA2837 OPA837, OPA2837 www.ti.com SBOS673D – SEPTEMBER 2017 – REVISED DECEMBER 2018 • Changed default test condition in Electrical Characteristics: VS = 3 V table ......................................................................... 9 • Changed footnote describing method of computation of slew rate in Electrical Characteristics: VS = 3 V table .................. 9 • Changed values for common-mode input range, high in Electrical Characteristics: VS = 3 V table .................................... 10 • Changed values for VOH in Electrical Characteristics: VS = 3 V table .................................................................................. 10 • Changed VO = 20 mVPP to VOUT = 200 mVPP in conditions of Noninverting Response Flatness vs Gain and Inverting Response Flatness vs Gain figures...................................................................................................................................... 11 • Changed gain –1 V/V to gain –2 V/V, swapped legend colors in Inverting Overdrive Recovery figure............................... 12 • Changed VOUT = 2 VPP to VOUT = 1 VPP in conditions of Typical Characteristics: VS = 3.0 V section ................................... 14 • Changed VO = 20 mVPP to VOUT = 200 mVPP in Noninverting Response Flatness vs Gain and Inverting Response Flatness vs Gain figure conditions........................................................................................................................................ 14 • Changed VIN to VIN × –1 gain, swapped legend colors in Inverting Overdrive Recovery figure .......................................... 15 • Changed VO = 2 VPP to VOUT = 1 VPP in Harmonic Distortion vs RLOAD figure conditions..................................................... 16 • Changed VOUT = 2 VPP to VOUT = 1 VPP in Harmonic Distortion vs Gain Magnitude figure conditions ................................. 16 • Changed y-axis caption in Turn-On Time to Sinusoidal Input and Turn-Off Time to Sinusoidal Input figures..................... 19 • Added OPA838 row to Device Family Comparison table..................................................................................................... 23 • Changed EVM link in Split-Supply Operation section from OPA837DBV to OPA835DBV .................................................. 26 • Changed V2 value from 2.5 to –2.5 V in Characterization Test Circuit for Network, Spectrum Analyzer figure ................. 30 • Changed VEE value from 2.5 V to –2.5 V in Inverting Characterization Circuit for Network Analyzer figure ....................... 32 • Changed 1 SPS to 1 MSPS in OPA837 and ADS8860 Example Circuit figure ................................................................... 38 Submit Documentation Feedback Copyright © 2017–2018, Texas Instruments Incorporated Product Folder Links: OPA837 OPA2837 3 OPA837, OPA2837 SBOS673D – SEPTEMBER 2017 – REVISED DECEMBER 2018 www.ti.com 5 Pin Configuration and Functions OPA837 DBV Package 6-Pin SOT-23 Top View OPA837 DCK Package 5-Pin SC70 Top View VOUT 1 6 VS+ VOUT 1 VS- 2 5 PD VS- 2 +IN 3 4 VIN- VIN+ 3 OPA2837 DGK Package 8-Pin VSSOP Top View 5 VS+ 4 VIN- OPA2837 RUN Package 10-Pin WQFN Top View VS+ VOUT1 VIN1- 1 8 2 VS+ 7 VOUT2 6 VIN2- 10 VOUT1 1 A VIN1+ 3 VIN1- 2 B VS- 4 5 VIN1+ 3 VIN2+ 9 VOUT2 8 VIN2- B A 7 VIN2+ PD 1 4 5 6 PD 2 VS- Pin Functions PIN NAME OPA837 FUNCTION (1) OPA2837 DESCRIPTION SOT-23 SC-70 VSSOP WQFN PD 5 — — — I Amplifier power down. Low = disabled, high = normal operation (pin must be driven). PD1 — — — 4 I Amplifier 1 power down. Low = disabled, high = normal operation (pin must be driven). PD2 — — — 6 I Amplifier 2 power down. Low = disabled, high = normal operation (pin must be driven). VIN– 4 4 — — I Inverting input pin VIN+ 3 3 — — I Noninverting input pin VIN1– — — 2 2 I Amplifier 1 inverting input pin VIN1+ — — 3 3 I Amplifier 1 noninverting input pin VIN2– — — 6 8 I Amplifier 2 inverting input pin VIN2+ — — 5 7 I Amplifier 2 noninverting input pin VOUT 1 1 — — O Output pin VOUT1 — — 1 1 O Amplifier 1 output pin VOUT2 — — 7 9 O Amplifier 2 output pin VS– 2 2 4 5 P Negative power-supply pin VS+ 6 5 8 10 P Positive power-supply input (1) 4 I = input, O = output, and P = power. Submit Documentation Feedback Copyright © 2017–2018, Texas Instruments Incorporated Product Folder Links: OPA837 OPA2837 OPA837, OPA2837 www.ti.com SBOS673D – SEPTEMBER 2017 – REVISED DECEMBER 2018 6 Specifications 6.1 Absolute Maximum Ratings over operating free-air temperature range (unless otherwise noted) (1) MIN VS– to VS+ Supply voltage Input voltage VID UNIT 5.5 Supply turn-on/off maximum dV/dT VI MAX (2) V 1 VS– – 0.5 V/µs VS+ + 0.5 V Differential input voltage ±1 V II Continuous input current ±10 mA IO Continuous output current (3) ±20 mA See Thermal Information: OPA837 Continuous power dissipation TJ Maximum junction temperature 150 °C TA Operating free-air temperature –40 125 °C Tstg Storage temperature –65 150 °C (1) (2) (3) Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. These are stress ratings only, which do not imply functional operation of the device at these or any other conditions beyond those indicated under Recommended Operating Conditions. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability. Staying below this ± supply turn-on edge rate prevents the edge-triggered ESD absorption device across the supply pins from turning on. Long-term continuous output current for electromigration limits. 6.2 ESD Ratings VALUE V(ESD) (1) (2) Electrostatic discharge Human body model (HBM), per ANSI/ESDA/JEDEC JS-001 (1) ±1500 Charged-device model (CDM), per JEDEC specification JESD22-C101 (2) ±1000 UNIT V JEDEC document JEP155 states that 500-V HBM allows safe manufacturing with a standard ESD control process. JEDEC document JEP157 states that 250-V CDM allows safe manufacturing with a standard ESD control process. 6.3 Recommended Operating Conditions over operating free-air temperature range (unless otherwise noted) MIN NOM MAX VS+ Single-supply voltage 2.7 5 5.4 UNIT V TA Ambient temperature –40 25 125 °C Submit Documentation Feedback Copyright © 2017–2018, Texas Instruments Incorporated Product Folder Links: OPA837 OPA2837 5 OPA837, OPA2837 SBOS673D – SEPTEMBER 2017 – REVISED DECEMBER 2018 www.ti.com 6.4 Thermal Information: OPA837 OPA837 THERMAL METRIC (1) DBV (SOT23-6) DCK (SC70) UNIT 6 PINS 5 PINS RθJA Junction-to-ambient thermal resistance 194 203 °C/W RθJCtop Junction-to-case (top) thermal resistance 129 152 °C/W RθJB Junction-to-board thermal resistance 39 76 °C/W ψJT Junction-to-top characterization parameter 26 58 °C/W ψJB Junction-to-board characterization parameter 39 76 °C/W (1) For more information about traditional and new thermal metrics, see the Semiconductor and IC Package Thermal Metrics application report. 6.5 Thermal Information: OPA2837 OPA2837 THERMAL METRIC (1) RUN (WQFN-10) DGK (VSSOP-8) 10 PINS 8 PINS UNIT RθJA Junction-to-ambient thermal resistance 124.9 182 °C/W RθJCtop Junction-to-case (top) thermal resistance 72.0 63.5 °C/W RθJB Junction-to-board thermal resistance 63.2 103.6 °C/W ψJT Junction-to-top characterization parameter 4.3 7.9 °C/W ψJB Junction-to-board characterization parameter 63.0 101.8 °C/W (1) 6 For more information about traditional and new thermal metrics, see the Semiconductor and IC Package Thermal Metrics application report. Submit Documentation Feedback Copyright © 2017–2018, Texas Instruments Incorporated Product Folder Links: OPA837 OPA2837 OPA837, OPA2837 www.ti.com SBOS673D – SEPTEMBER 2017 – REVISED DECEMBER 2018 6.6 Electrical Characteristics: VS = 5 V at VS+ = 5 V, VS– = 0 V, RF = 0 Ω, RL = 2 kΩ, G = 1 V/V, input and output referenced to mid-supply, and TA ≈ 25°C (unless otherwise noted) PARAMETER TEST CONDITIONS MIN TYP 90 105 MAX UNIT TEST LEVEL (1) AC PERFORMANCE VOUT = 20 mVPP, G = 1 SSBW Small-signal bandwidth VOUT = 20 mVPP, G = 2 45 VOUT = 20 mVPP, G = 10 5 45 C MHz C C GBP Gain-bandwidth product VOUT = 20 mVPP, G = 10 50 MHz LSBW Large-signal bandwidth VOUT = 2 VPP, G = 2 C 26 MHz Bandwidth for 0.1-dB flatness VOUT = 200 mVPP, G = 2 C 6 MHz SR Slew rate From LSBW (2) C 105 V/µs tR, tF Rise, fall time VOUT = 0.5-V step, G = 2, input tR = 10 ns C ns Overshoot VOUT = 2-V step, G = 2, input tR = 40 ns 7.0% C Settling time to 0.1% VOUT = 2.0-V step, G = 1, input tR = 4 ns 25 ns C Settling time to 0.01% VOUT = 2.0-V step, G = 1, input tR = 4 ns 40 ns C HD2 Second-order harmonic distortion f = 100 kHz, VO = 2 VPP, G = 1 (see Figure 73) –120 dBc C HD3 Third-order harmonic distortion f = 100 kHz, VO = 2 VPP, G = 1 (see Figure 73) –145 dBc C Input voltage noise f = 500 Hz nV/√Hz C 10 11 C 4.7 Voltage noise 1/f corner frequency See Figure 39 35 Hz C Input current noise 0.4 pA/√Hz C 5 kHz C 75 ns C f = 20 kHz Current noise 1/f corner frequency See Figure 39 Overdrive recovery time G = 2, 2x output overdrive (see Figure 30) Closed-loop output impedance f = 1 MHz, G = 1 (see Figure 38) 0.14 Ω C Channel-to-channel crosstalk (OPA2837) f = 10 kHz -126 dBc C dB A DC PERFORMANCE AOL Open-loop voltage gain Input-referred offset voltage Input offset voltage drift (3) VO = ±2 V, RL = 2 kΩ 120 135 TA ≈ 25°C –165 ±30 165 TA = 0°C to +70°C (DCK package) –205 ±30 235 TA = –40°C to +85°C (DCK package) –269 ±30 261 TA = –40°C to +125°C (DCK package) –269 ±30 325 DCK package, TA = –40°C to +125°C –1.6 ±0.4 1.6 DBV, RUN package, TA = –40°C to +125°C –2.0 ±0.4 2.0 DGK package, TA = –40°C to +125°C TA ≈ 25°C Input bias current (4) Input bias current drift (3) Input offset current (1) (2) (3) (4) A µV B B B B µV/°C ±0.67 B B 150 340 520 TA = 0°C to +70°C 50 340 664 A TA = –40°C to +85°C 50 340 718 TA = –40°C to +125°C 50 340 850 TA = –40°C to +125°C 0.8 1.5 3.3 TA ≈ 25°C (OPA837) –40 ±6 40 TA = 0°C to +70°C –46 ±6 52 TA = –40°C to +85°C –56 ±6 55 TA = –40°C to +125°C –56 ±6 65 B TA ≈ 25°C (OPA2837) -60 ±8 60 A nA B B B nA/°C B A B nA B Test levels (all values set by characterization and simulation): (A) 100% tested at 25°C, overtemperature limits by characterization and simulation; (B) Not tested in production, limits set by characterization and simulation; (C) Typical value only for information. This slew rate is the average of the rising and falling time estimated from the large-signal bandwidth as: (0.8 × VPEAK / √2) × 2π × f–3dB where this f–3dB is the typical measured 2-VPP bandwidth at gains of 1 V/V. Input offset voltage drift, input bias current drift, and input offset current drift are average values calculated by taking data at the end points, computing the difference, and dividing by the temperature range. Typical drift specifications are ±1sigma. Maximum drift specifications are set by min/max sample packaged test data using a wafer-level screened drift. Min/Max drift is not specified by final automated test equipment (ATE) nor by QA sample testing. Current is considered positive out of the pin. Submit Documentation Feedback Copyright © 2017–2018, Texas Instruments Incorporated Product Folder Links: OPA837 OPA2837 7 OPA837, OPA2837 SBOS673D – SEPTEMBER 2017 – REVISED DECEMBER 2018 www.ti.com Electrical Characteristics: VS = 5 V (continued) at VS+ = 5 V, VS– = 0 V, RF = 0 Ω, RL = 2 kΩ, G = 1 V/V, input and output referenced to mid-supply, and TA ≈ 25°C (unless otherwise noted) PARAMETER Input offset current drift (3) Input-referred offset voltage mismatch TEST CONDITIONS MIN TYP MAX TA = –40°C to +125°C –250 ±40 250 TA = –40°C to +125°C (OPA2837) –270 ±80 330 TA ≈ 25°C (OPA2837) -220 50 220 –0.2 0 UNIT pA/°C µV TEST LEVEL (1) B B A INPUT TA ≈ 25°C, < 3-dB degradation in CMRR limit Common-mode input range, low TA = –40°C to +125°C, < 3-dB degradation in CMRR limit TA ≈ 25°C, < 3-dB degradation in CMRR limit Common-mode input range, high CMRR TA = –40°C to +125°C, < 3-dB degradation in CMRR limit Common-mode rejection ratio –0.2 3.7 3.7 91 0 A V 3.8 B A V 3.8 110 B dB A Input impedance common-mode 175 || 1.5 MΩ || pF C Input impedance differential mode 180 || 0.5 kΩ || pF C OUTPUT VOL Output voltage, low VOH TA ≈ 25°C, G = 2 TA = –40°C to +125°C, G = 5 0.05 0.1 0.05 0.1 V A B TA ≈ 25°C, G = 2 4.9 4.95 TA = –40°C to +125°C, G = 5 4.8 4.9 Maximum current into a resistive load TA ≈ 25°C, ±1.6 V into 27 Ω, VIO < 2 mV ±58 ±70 mA A Linear current into a resistive load TA ≈ 25°C, ±1.7 V into 37.4 Ω, AOL > 80 dB ±45 ±50 mA A Linear current into a resistive load overtemperature TA = –40°C to +125°C, ±1.31 V into 37.4 Ω, AOL > 80 dB ±35 ±45 mA C Closed-loop output impedance Gain of 1 V/V, ±30-mA DC 0.6 mΩ C V B Output voltage, high V A B POWER SUPPLY Specified operating voltage Quiescent operating current per amplifier (5-V supply) Supply current temperature coefficient per amplifier 2.7 5.4 TA ≈ 25°C (5) 564 592 625 TA = –40°C to +125°C 408 592 865 TA = –40°C to +125°C (see Figure 57) 1.1 1.9 2.4 µA A B µA/°C B +PSRR Positive power-supply rejection ratio 95 110 dB A –PSRR Negative power-supply rejection ratio 92 108 dB A V A V A 50 nA A 10 µA A 10 µA B POWER DOWN (Pin Must be Driven) (5) 8 Enable voltage threshold Specified on above VS– + 1.5 V Disable voltage threshold Specified off below VS– + 0.55 V 0.55 1.5 Power-down pin bias current PD = 0 V to VS+ –50 Power-down quiescent current PD ≤ 0.55 V Power-down quiescent current over temperature PD ≤ 0.55 V, TA = –40°C to +125°C Turnon time delay Time from PD = high to VOUT = 90% of final value 300 ns C Turnoff time delay Time from PD = low to VOUT = 10% of original value 100 ns C 4 5 The typical specification is at 25°C TJ. The min, max limits are expanded for the automated test equipment (ATE) to account for an ambient range from 22°C to 32°C with a 2-µA/°C temperature coefficient on the supply current. Submit Documentation Feedback Copyright © 2017–2018, Texas Instruments Incorporated Product Folder Links: OPA837 OPA2837 OPA837, OPA2837 www.ti.com SBOS673D – SEPTEMBER 2017 – REVISED DECEMBER 2018 6.7 Electrical Characteristics: VS = 3 V at VS+ = 3 V, VS– = 0 V, RF = 0 Ω, RL = 2 kΩ, G = 1 V/V, input and output referenced to mid-supply, and TA ≈ 25°C (unless otherwise noted) PARAMETER TEST CONDITIONS MIN TYP 85 105 MAX UNIT TEST LEVEL (1) AC PERFORMANCE VOUT = 20 mVPP, G = 1 SSBW Small-signal bandwidth VOUT = 20 mVPP, G = 2 45 VOUT = 20 mVPP, G = 10 5 GBP Gain-bandwidth product VOUT = 20 mVPP, G = 10 LSBW Large-signal bandwidth VOUT = 1 VPP, G = 2 Bandwidth for 0.1-dB flatness VOUT = 200 mVPP, G = 2 SR Slew rate tR, tF 40 C MHz C C 50 MHz C 30 MHz C 6 MHz C From LSBW (2) 65 V/µs C Rise, fall time VOUT = 0.5-V step, G = 2, input tR = 10 ns 10 ns C Overshoot VOUT = 2-V step, G = 2, input tR = 40 ns 7% Settling time to 0.1% VOUT = 0.5-V step, G = 1, input tR = 4 ns 35 ns C Settling time to 0.01% VOUT = 0.5-V step, G = 1, input tR = 4 ns 50 ns C HD2 Second-order harmonic distortion f = 100 kHz, VO = 1 VPP, G = 1 (see Figure 73) –125 dBc C HD3 Third-order harmonic distortion f = 100 kHz, VO = 1 VPP, G = 1 (see Figure 73) –138 dBc C Input voltage noise f = 500 Hz nV/√Hz C 11 C 4.9 Voltage noise 1/f corner frequency See Figure 39 35 Hz C Input current noise 0.4 pA/√Hz C 5 kHz C 65 ns C f = 10 kHz Current noise 1/f corner frequency See Figure 39 Overdrive recovery time G = 2, 2x output overdrive (see Figure 29) Closed-loop output impedance f = 1 MHz, G = 1 (see Figure 38) 0.14 Ω C Channel-to-channel crosstalk (OPA2837) f = 10 kHz -126 dBc C DC PERFORMANCE AOL Open-loop voltage gain Input-referred offset voltage Input offset voltage drift (3) VO = ±1 V, RL = 2 kΩ 120 133 VO = ±1 V, RL = 2 kΩ (OPA2837) 110 133 TA ≈ 25°C –165 ±30 165 TA = 0°C to +70°C –205 ±30 235 TA = –40°C to +85°C –269 ±30 261 TA = –40°C to +125°C –269 ±30 325 DCK package, TA = –40°C to +125°C –1.6 ±0.4 1.6 DBV, RUN package, TA = –40°C to +125°C –2.0 ±0.4 2.0 DGK package, TA = –40°C to +125°C TA ≈ 25°C Input bias current (4) Input bias current drift Input offset current (1) (2) (3) (4) (3) dB A A A µV B B B B µV/°C ±0.67 B B 145 320 510 TA = 0°C to +70°C 50 320 659 A TA = –40°C to +85°C 50 320 708 TA = –40°C to +125°C 50 320 840 TA = –40°C to +125°C 0.8 1.5 3.3 TA ≈ 25°C (OPA837) –40 ±6 40 TA = 0°C to +70°C –46 ±6 52 TA = –40°C to +85°C –56 ±6 55 TA = –40°C to +125°C –56 ±6 65 B TA ≈ 25°C (OPA2837) –60 ±8 60 A nA B B B nA/°C B A B nA B Test levels (all values set by characterization and simulation): (A) 100% tested at 25°C, overtemperature limits by characterization and simulation; (B) Not tested in production, limits set by characterization and simulation; (C) Typical value only for information. This slew rate is the average of the rising and falling time estimated from the large-signal bandwidth as: (0.8 x VPEAK / √2) × 2π × f–3dB where this f-3dB is the typical measured 2-Vpp bandwidth at gains of 1V/V. Input offset voltage drift, input bias current drift, and input offset current drift are average values calculated by taking data at the end points, computing the difference, and dividing by the temperature range. Typical drift specifications are ±1sigma. Maximum drift specifications are set by min/max sample packaged test data using a wafer-level screened drift. Min/Max drift is not specified by final automated test equipment (ATE) nor by QA sample testing. Current is considered positive out of the pin. Submit Documentation Feedback Copyright © 2017–2018, Texas Instruments Incorporated Product Folder Links: OPA837 OPA2837 9 OPA837, OPA2837 SBOS673D – SEPTEMBER 2017 – REVISED DECEMBER 2018 www.ti.com Electrical Characteristics: VS = 3 V (continued) at VS+ = 3 V, VS– = 0 V, RF = 0 Ω, RL = 2 kΩ, G = 1 V/V, input and output referenced to mid-supply, and TA ≈ 25°C (unless otherwise noted) PARAMETER Input offset current drift (3) Input-referred offset voltage mismatch TEST CONDITIONS MIN TYP MAX TA = –40°C to +125°C –250 ±40 250 TA = –40°C to +125°C (OPA2837) –250 ±80 330 TA ≈ 25°C (OPA2837) -220 50 220 –0.2 0 UNIT pA/°C µV TEST LEVEL (1) B B A INPUT TA ≈ 25°C, < 3-dB degradation in CMRR limit Common-mode input range, low TA = –40°C to +125°C, < 3-dB degradation in CMRR limit TA ≈ 25°C, < 3-dB degradation in CMRR limit Common-mode input range, high CMRR TA = –40°C to +125°C, < 3-dB degradation in CMRR limit Common-mode rejection ratio –0.2 1.7 1.7 90 0 A V 1.8 B A V 1.8 105 B dB A Input impedance common-mode 300 || 1.5 MΩ || pF C Input impedance differential mode 180 || 0.5 kΩ || pF C OUTPUT VOL Output voltage, low VOH TA ≈ 25°C, G = 2 TA = –40°C to +125°C, G = 2 0.05 0.1 0.10 0.2 V A B TA ≈ 25°C, G = 2 2.9 2.95 TA = –40°C to +125°C, G = 2 2.8 2.9 Maximum current into a resistive load TA ≈ 25°C, ±0.8 V into 17.5 Ω, VIO < 2 mV ±45 ±55 mA A Linear current into a resistive load TA ≈ 25°C, ±0.9 V into 21.5 Ω, AOL > 80 dB ±40 ±45 mA A Linear current into a resistive load overtemperature TA = –40°C to 125°C, ±0.7 V into 21.5 Ω, AOL > 80 dB ±32 ±40 mA C V B TA ≈ 25°C (5) 547 570 607 TA = –40°C to +125°C 404 570 817 (5) 540 570 607 µA A 0.8 1.7 2.2 µA/°C B Output voltage, high V A B POWER SUPPLY Specified operating voltage Quiescent operating current per amplifier (OPA837, 3-V supply) 2.7 Quiescent operating current per amplifier (OPA2837, 3-V supply) TA ≈ 25°C Supply current temperature coefficient per amplifier TA = –40°C to +125°C (see Figure 57) 5.4 µA A B +PSRR Positive power-supply rejection ratio 90 110 dB A –PSRR Negative power-supply rejection ratio 88 105 dB A V A V A 50 nA A 8 µA A 8 µA B POWER DOWN (Pin Must be Driven) (5) 10 Enable voltage threshold Specified on above VS– + 1.5 V Disable voltage threshold Specified off below VS– + 0.55 V 0.55 1.5 Power-down pin bias current PD = 0 V to VS+ –50 Power-down quiescent current PD ≤ 0.55 V Power-down quiescent current over temperature PD ≤ 0.55 V, TA = –40°C to +125°C Turnon time delay Time from PD = high to VOUT = 90% of final value 300 ns C Turnoff time delay Time from PD = low to VOUT = 10% of original value 100 ns C 1 3 The typical spec is at 25oC Tj. The min, max limits are expanded for ATE to account for ambient range from 22oC to 32oC with a +4uA/oC temperature coefficient on the supply current. Submit Documentation Feedback Copyright © 2017–2018, Texas Instruments Incorporated Product Folder Links: OPA837 OPA2837 OPA837, OPA2837 www.ti.com SBOS673D – SEPTEMBER 2017 – REVISED DECEMBER 2018 6.8 Typical Characteristics: VS = 5.0 V 3 3 0 0 Normalized Gain (dB) Normalized Gain (dB) at VS+ = 5.0 V, VS– = 0 V, VOUT = 2 VPP, RF = 0 Ω, RL = 2 kΩ, G = 1 V/V, input and output referenced to mid-supply, and TA ≈ 25°C (unless otherwise noted) -3 -6 -9 Gain = 1 V/V Gain = 2 V/V Gain = 5 V/V Gain = 10 V/V -12 -15 0.1 -6 -9 Gain = -1 V/V Gain = -2 V/V Gain = -5 V/V Gain = -10 V/V -12 1 10 Frequency (MHz) -15 0.1 100 1 D002 Figure 2. Inverting Small-Signal Frequency Response vs Gain 3 6 0 3 -3 Gain (dB) 9 0 100 See Figure 75 and Table 3, VOUT = 20 mVPP, RLOAD = 2 kΩ Figure 1. Noninverting Small-Signal Frequency Response vs Gain -3 -6 -9 VO = 200 mVPP VO = 500 mVPP VO = 1 VPP VO 2 VPP -6 -9 0.1 1 VO = 200 mVPP VO = 500 mVPP VO = 1 VPP VO = 2 VPP -12 10 Frequency (MHz) -15 0.1 100 1 0.8 0.8 0.6 0.6 Normalized Gain (dB) 1 0.4 0.2 0 -0.2 -0.4 Gain = 1 V/V Gain = 2 V/V Gain = 5 V/V Gain = 10 V/V -1 0.1 1 0.4 0.2 0 -0.2 -0.4 -0.6 -0.8 10 Frequency (MHz) D004 Figure 4. Inverting Large-Signal Bandwidth vs VOPP 1 -0.8 100 Gain = –1 V/V, RLOAD = 2 kΩ Figure 3. Noninverting Large-Signal Bandwidth vs VOPP -0.6 10 Frequency (MHz) D003 Gain = 2 V/V, RLOAD = 2 kΩ Normalized Gain (dB) 10 Frequency (MHz) D001 See Figure 74 and Table 2, VOUT = 20 mVPP, RLOAD = 2 kΩ Gain (dB) -3 Gain = -1 V/V Gain = -2 V/V Gain = -5 V/V Gain = -10 V/V -1 0.1 100 1 D005 See Figure 74 and Table 2, VOUT = 200 mVPP, RLOAD = 2 kΩ Figure 5. Noninverting Response Flatness vs Gain 10 Frequency (MHz) 100 D006 See Figure 75 and Table 3, VOUT = 200 mVPP, RLOAD = 2 kΩ Figure 6. Inverting Response Flatness vs Gain Submit Documentation Feedback Copyright © 2017–2018, Texas Instruments Incorporated Product Folder Links: OPA837 OPA2837 11 OPA837, OPA2837 SBOS673D – SEPTEMBER 2017 – REVISED DECEMBER 2018 www.ti.com Typical Characteristics: VS = 5.0 V (continued) 1.2 1 1 0.8 0.8 0.6 0.6 0.4 Output Voltage (V) Output Voltage (V) at VS+ = 5.0 V, VS– = 0 V, VOUT = 2 VPP, RF = 0 Ω, RL = 2 kΩ, G = 1 V/V, input and output referenced to mid-supply, and TA ≈ 25°C (unless otherwise noted) 0.4 0.2 0 -0.2 -0.4 -0.6 VO = ±0.125 V VO = ±0.25 V VO = ±0.5 V VO = ±1 V -0.8 -1 0.2 0 -0.2 -0.4 -0.6 VO = ±0.125 V VO = ±0.25 V VO = ±0.5 V VO = ±1 V -0.8 -1 -1.2 -1.2 0 100 200 300 400 500 600 Time (ns) 700 800 900 1000 0 See Figure 74, gain = 2 V/V, input edge rate set to stay below slew limiting 0.08 0.08 0.06 0.06 0.04 0.04 0.02 0 -0.02 -0.04 AV = 1, 500-mV Step, TR = 10 ns AV = 1, 2-V Step, TR = 40 ns AV = 2, 500-mV Step, TR = 10 ns AV = 2, 2-V Step, TR = 40 ns 700 800 900 1000 D008 AV = -1 , 500-mV Step, TR = 10 ns AV = -1 , 2-V Step, TR = 40 ns AV = -2 , 500-mV Step, TR = 10 ns AV = -2, 2-V Step, TR = 40 ns 0.02 0 -0.02 -0.04 -0.08 -0.1 0 25 50 75 100 125 150 Time (ns) 175 200 225 250 0 25 50 75 D009 See Figure 74 and Table 2 100 125 150 Time (ns) 175 200 225 250 D010 See Figure 75 and Table 3 Figure 9. Simulated Noninverting Settling Time Figure 10. Simulated Inverting Settling Time 5 5 VIN x 2 gain VOUT (AV = 2) 3 2 1 0 -1 -2 -3 3 2 1 0 -1 -2 -3 -4 -4 -5 50 -5 50 250 450 650 850 Time (ns) 1050 1250 VIN x 2 gain VOUT (AV = 2) 4 Input and Output Voltage (V) 4 Input and Output (V) 400 500 600 Time (ns) -0.06 -0.1 1450 250 D011 See Figure 74 and Table 2, gain = 2 V/V Figure 11. Noninverting Overdrive Recovery 12 300 Figure 8. Inverting Step Response vs Time and VOPP 0.1 Settling Value (%) Settling Value (%) Figure 7. Noninverting Step Response vs Time and VOPP -0.08 200 See Figure 75, gain = –1 V/V, input edge rate set to stay below slew limiting 0.1 -0.06 100 D007 450 650 850 Time (ns) 1050 1250 1450 D012 See Figure 75 and Table 3, gain –2 V/V Figure 12. Inverting Overdrive Recovery Submit Documentation Feedback Copyright © 2017–2018, Texas Instruments Incorporated Product Folder Links: OPA837 OPA2837 OPA837, OPA2837 www.ti.com SBOS673D – SEPTEMBER 2017 – REVISED DECEMBER 2018 Typical Characteristics: VS = 5.0 V (continued) at VS+ = 5.0 V, VS– = 0 V, VOUT = 2 VPP, RF = 0 Ω, RL = 2 kΩ, G = 1 V/V, input and output referenced to mid-supply, and TA ≈ 25°C (unless otherwise noted) -80 -100 HD2, Gain = 1 V/V HD3, Gain = 1 V/V HD2, Gain = 1 V/V HD3, Gain = 1 V/V -100 -105 Harmonic Distortion (dBc) Harmonic Distortion (dBc) -90 -110 -120 -130 -140 -110 -115 -120 -125 -130 -135 -140 -145 -150 10k 100k Frequency (Hz) HD2, Gain = 1 V/V HD3, Gain = 1 V/V HD2, Gain = 1 V/V HD3, Gain = 1 V/V -150 100 1M 1k RLOAD (:) D013 See Figure 74, Figure 75, Table 2, and Table 3, VOUT = 2 VPP D014 See Figure 74, Figure 75, Table 2, and Table 3, VOUT = 2 VPP, f = 100 kHz Figure 14. Harmonic Distortion vs RLOAD Figure 13. Harmonic Distortion vs Frequency -100 -110 -105 Harmonic Distortion (dBc) Harmonic Distortion (dBc) -115 -120 -125 -130 HD2, Gain = 2 V/V HD3, Gain = 2 V/V HD2, Gain = 1 V/V HD3, Gain = 1 V/V -135 -140 0.4 -120 -125 -130 -135 HD2, +Gain HD3, +Gain HD2, Gain HD3, Gain -140 -150 0.8 1.2 1.6 2 2.4 2.8 Output Voltage (V) 3.2 3.6 4 1 10 Gain Magnitude (V/V) D015 Figure 16. Harmonic Distortion vs Gain Magnitude Harmonic Distortion (dBc) HD2 HD3 100k Frequency (Hz) 1M -80 -85 -90 -95 -100 -105 -110 -115 -120 -125 -130 -135 -140 -145 -150 10k HD2 (PGA 1) HD3 (PGA 1) HD2 (PGA 2) HD3 (PGA 2) 100k Frequency (Hz) D017 See Figure 87, VOUT = 2 VPP, f = 100 kHz Figure 17. Harmonic Distortion as Active Mux D016 See Figure 74, Figure 75, Table 2, and Table 3, VOUT = 2 VPP, f = 100 kHz Figure 15. Harmonic Distortion vs Output Voltage Harmonic Distortion (dBc) -115 -145 See Figure 74, Figure 75, Table 2, and Table 3, VOUT = 2 VPP, f = 100 kHz -80 -85 -90 -95 -100 -105 -110 -115 -120 -125 -130 -135 -140 -145 -150 10k -110 1M D018 See Figure 87, gain of 1 V/V or 2 V/V, VOUT = 2 VPP, f = 100 kHz Figure 18. Harmonic Distortion as 1-Bit PGA Submit Documentation Feedback Copyright © 2017–2018, Texas Instruments Incorporated Product Folder Links: OPA837 OPA2837 13 OPA837, OPA2837 SBOS673D – SEPTEMBER 2017 – REVISED DECEMBER 2018 www.ti.com 6.9 Typical Characteristics: VS = 3.0 V 3 3 0 0 -3 -3 Normalized Gain (dB) Normalized Gain (dB) at VS+ = 3.0 V, VS– = 0 V, VOUT = 1 VPP, RF = 0 Ω, RL = 2 kΩ, G = 1 V/V, input and output referenced to mid-supply, and TA ≈ 25°C (unless otherwise noted) -6 -9 -12 -15 Gain = 1 V/V Gain = 2 V/V Gain = 5 V/V Gain = 10 V/V -18 -21 0.1 1 -6 -9 -12 -15 -18 10 Frequency (MHz) -21 0.1 100 1 3 0 0 -3 -3 -6 -6 Gain (dB) 3 -9 -12 100 D020 Figure 20. Inverting Small-Signal Response vs Gain -15 -9 -12 -15 VO = 200 mVPP VO = 500 mVPP VO = 1 VPP -18 -21 0.1 1 VO = 200 mVPP VO = 500 mVPP VO = 1 VPP -18 10 Frequency (MHz) -21 0.1 100 1 10 Frequency (MHz) D021 See Figure 74, gain = 2 V/V 100 D022 See Figure 75, gain = –1 V/V Figure 21. Noninverting Large-Signal Bandwidth vs VOPP Figure 22. Inverting Large-Signal Bandwidth vs VOPP 1.2 1 1 0.8 0.8 Normalized Gain (dB) 0.6 Normalized Gain (dB) 10 Frequency (MHz) See Figure 75 and Table 3, VOUT = 20 mVPP, RLOAD = 2 kΩ Figure 19. Noninverting Small-Signal Response vs Gain 0.4 0.2 0 -0.2 -0.4 Gain = 1 V/V Gain = 2 V/V Gain = 5 V/V Gain = 10 V/V -0.6 -0.8 -1 0.1 0.6 0.4 0.2 0 -0.2 -0.4 -0.6 -0.8 -1 1 10 Frequency (MHz) 100 D023 See Figure 74 and Table 2, VOUT = 200 mVPP, RLOAD = 2 kΩ Figure 23. Noninverting Response Flatness vs Gain 14 1 V/V 2 V/V 5 V/V 10 V/V D019 See Figure 74 and Table 2, VOUT = 20 mVPP, RLOAD = 2 kΩ Gain (dB) Gain = Gain = Gain = Gain = Gain = Gain = Gain = Gain = -1.2 0.1 1 V/V 2 V/V 5 V/V 10 V/V 1 10 100 Frequency (MHz) D024 See Figure 75 and Table 3, VOUT = 200 mVPP, RLOAD = 2 kΩ Figure 24. Inverting Response Flatness vs Gain Submit Documentation Feedback Copyright © 2017–2018, Texas Instruments Incorporated Product Folder Links: OPA837 OPA2837 OPA837, OPA2837 www.ti.com SBOS673D – SEPTEMBER 2017 – REVISED DECEMBER 2018 Typical Characteristics: VS = 3.0 V (continued) at VS+ = 3.0 V, VS– = 0 V, VOUT = 1 VPP, RF = 0 Ω, RL = 2 kΩ, G = 1 V/V, input and output referenced to mid-supply, and TA ≈ 25°C (unless otherwise noted) 0.6 1.2 0.4 0.8 1 Output Voltage (V) Output Voltage (V) 0.6 0.2 0 -0.2 0.4 0.2 0 -0.2 -0.4 -0.6 VO = ±0.125 V VO = ±0.25 V VO = ±0.5 V -0.4 VO = ±0.125 V VO = ±0.25 V VO = ±0.5 V VO = ±1 V -0.8 -1 -0.6 -1.2 0 100 200 300 400 500 600 Time (ns) 700 800 900 1000 0 See Figure 74 and Table 2, gain = 2 V/V, input edge rate set to stay below slew limiting 0.08 0.08 0.06 0.06 0.04 0.04 0.02 0 -0.02 -0.04 400 500 600 Time (ns) 700 800 900 1000 D026 AV = AV = AV = AV = 1, 500-mV Step, TR = 10 ns 1 , 2-V Step, TR = 40 ns 2 , 500-mV Step, TR = 10 ns 2 , 1-V Step, TR = 40 ns 0.02 0 -0.02 -0.04 -0.06 AV = 1, 500-mV Step, TR = 10 ns AV = 2 , 1-V Step, TR = 20 ns AV = 2 , 500-mV Step, TR = 10 ns -0.08 -0.1 -0.1 0 25 50 75 100 125 Time (ns) 150 175 200 0 25 50 100 125 150 175 200 225 250 275 Time (ns) D028 See Figure 75 and Table 3 Figure 27. Simulated Noninverting Settling Time Figure 28. Simulated Inverting Settling Time 3 3 VIN x 2 gain VOUT (AV = 2) Input and Output Voltage (V) 2 1 0 -1 -2 -3 50 75 D027 See Figure 74 and Table 2 Input and Output Voltage (V) 300 Figure 26. Inverting Step Response vs VOPP 0.1 Settling Value (%) Settling Value (%) Figure 25. Noninverting Step Response vs VOPP -0.08 200 See Figure 75 and Table 3, gain = –1 V/V, input edge rate set to stay below slew limiting 0.1 -0.06 100 D025 250 450 650 850 Time (ns) 1050 1250 1450 VIN x 1 gain VOUT (AV = 1) 2 1 0 -1 -2 -3 50 250 D029 See Figure 74 and Table 2, gain = 2 V/V Figure 29. Noninverting Overdrive Recovery 450 650 850 Time (ns) 1050 1250 1450 D030 See Figure 75 and Table 3, gain = –1 V/V Figure 30. Inverting Overdrive Recovery Submit Documentation Feedback Copyright © 2017–2018, Texas Instruments Incorporated Product Folder Links: OPA837 OPA2837 15 OPA837, OPA2837 SBOS673D – SEPTEMBER 2017 – REVISED DECEMBER 2018 www.ti.com Typical Characteristics: VS = 3.0 V (continued) at VS+ = 3.0 V, VS– = 0 V, VOUT = 1 VPP, RF = 0 Ω, RL = 2 kΩ, G = 1 V/V, input and output referenced to mid-supply, and TA ≈ 25°C (unless otherwise noted) -80 -100 HD2, Gain = 1 V/V HD3, Gain = 1 V/V HD2, Gain = 1 V/V HD3, Gain = 1 V/V -100 -105 -110 Distortion (dBc) Harmonic Distortion (dBc) -90 -110 -120 -130 -115 -120 -125 -130 -140 HD2, Gain = 1 V/V HD3, Gain = 1 V/V HD2, Gain = 1 V/V HD3, Gain = 1 V/V -135 -150 10k 100k Frequency (Hz) -140 100 1M See Figure 74, Figure 75, Table 2, and Table 3, VOUT = 1 VPP, RLOAD = 2 kΩ Figure 32. Harmonic Distortion vs RLOAD -100 -118 -105 -120 Harmonic Distortion (dBc) Harmonic Distortion (dBc) Figure 31. Harmonic Distortion vs Frequency HD2, Gain = 2 V/V HD3, Gain = 2 V/V HD2, Gain = 1 V/V HD3, Gain = 1 V/V -124 -126 -128 -130 -132 HD2, +Gain HD3, +Gain HD2, Gain HD3, Gain -110 -115 -120 -125 -130 -135 -134 -136 0.4 -140 0.5 0.6 0.7 0.8 VOPP (V) 0.9 1 1.1 1 Figure 33. Harmonic Distortion vs Output Swing Figure 34. Harmonic Distortion vs Gain Magnitude -80 HD2 HD3 -90 Harmonic Distortion (dBc) -100 D034 See Figure 74, Figure 75, Table 2, and Table 3, RLOAD = 2 kΩ, f = 100 kHz, VOUT = 1 VPP -90 -95 10 Gain Magnitude (V/V) D033 See Figure 74, Figure 75, Table 2, and Table 3, RLOAD = 2 kΩ, f = 100 kHz Harmonic Distortion (dBc) D032 See Figure 74, Figure 75, Table 2, and Table 3, VOUT = 1 VPP, f = 100 kHz, RLOAD = 2 kΩ -116 -122 1k Load (:) D031 -105 -110 -115 -120 -125 -130 -135 -140 -100 HD2 (PGA 1) HD3 (PGA 1) HD2 (PGA 2) HD3 (PGA 2) -110 -120 -130 -140 -145 -150 10k 100k Frequency (Hz) 1M 100k Frequency (Hz) D035 See Figure 87, gain = 1 V/V, VOUT = 1 VPP, RLOAD = 2 kΩ Figure 35. Harmonic Distortion as Active Mux 16 -150 10k 1M D036 See Figure 88, gain of 1 V/V and 2 V/V, VOUT = 1 VPP, RLOAD = 2 kΩ Figure 36. Harmonic Distortion as 1-Bit PGA Submit Documentation Feedback Copyright © 2017–2018, Texas Instruments Incorporated Product Folder Links: OPA837 OPA2837 OPA837, OPA2837 www.ti.com SBOS673D – SEPTEMBER 2017 – REVISED DECEMBER 2018 6.10 Typical Characteristics: ±2.5-V to ±1.5-V Split Supply with PD = VCC and TA ≈ 25°C (unless otherwise noted) 90 -90 70 -120 50 -150 30 -180 10 -210 -10 Output Impedance (ohms) 110 130 Open-Loop Gain (dB) 20 10 0 5-V AOL (dB) 3-V AOL (dB) -30 5-V AOL phase (q) 3-V AOL phase (q) -60 Open-Loop Phase (q) 150 -240 1 10 100 1k 10k 100k Frequency (Hz) 1M G = 1, 3-V supply G = 1, 5-V supply G = 2, 3-V supply G = 2, 5-V supply G = 5, 3-V supply G = 5, 5-V supply 1 0.1 0.01 0.001 10k 10M 100M 100k No load simulation 10M D038 Figure 74 and Table 2 (simulation) Figure 37. Open-Loop Gain and Phase vs Frequency Figure 38. Closed-Loop Output Impedance vs Frequency 10 250 5-V supply 3-V supply 200 Input Voltage Noise (nV) Input Spot Voltage (nV/—Hz) and Spot Current (pA/—Hz) noise 1M Frequency (Hz) D037 +5-V En +5-V In +3-V En +3-V In 1 150 100 50 0 -50 -100 -150 0.1 10 -200 100 1k 10k 100k Frequency (Hz) 1M 10M 0 1 D039 Measured then fit to ideal 1/f model Figure 39. Input Spot Noise Density vs Frequency -75 Off-Channel Isolation (dBm) 110 Rejection Ratio (dB) 100 90 80 70 40 30 20 100 CMRR 5 V CMRR 3 V PSRR VCC 5 V PSRR VEE 5 V PSRR VCC 3 V PSRR VEE 3 V 1k 1M 5 6 Time (s) 7 8 9 10 D040 -80 -85 -90 -95 -100 -105 -110 -115 10k 100k Frequency (Hz) 4 Figure 40. Low-Frequency Voltage Noise vs Time -70 50 3 Input-referred voltage noise RS = 0 Ω 120 60 2 10M -120 100k D041 5-V 200-mVPP (Output) 5-V 2-VPP (Output) 3-V 200-mVPP (Output) 3-V 2-VPP (Output) 1M Frequency (Hz) 10M D042 Simulated curves Figure 41. CMRR and PSRR vs Frequency Figure 42. Disabled Isolation Noninverting Input to Output vs Frequency Submit Documentation Feedback Copyright © 2017–2018, Texas Instruments Incorporated Product Folder Links: OPA837 OPA2837 17 OPA837, OPA2837 SBOS673D – SEPTEMBER 2017 – REVISED DECEMBER 2018 www.ti.com Typical Characteristics: ±2.5-V to ±1.5-V Split Supply (continued) with PD = VCC and TA ≈ 25°C (unless otherwise noted) 250 500 5-V Supply 3-V Supply 5-V Supply 3-V Supply 450 200 No. of Units in Each Bin No. of Units in Each Bin 225 175 150 125 100 75 400 350 300 250 200 150 50 100 25 50 0 0 -130-110 -90 -70 -50 -30 -10 10 30 50 70 90 110 130 Input Offset Voltage (PV) D043 -40 -35 -30 -25 -20 -15 -10 -5 0 5 10 15 20 25 30 35 40 Input Offset Current (nA) D044 830 units at each supply voltage 830 units at each supply voltage Figure 43. Input Offset Voltage Distribution Figure 44. Input Offset Current Distribution 30 200 25 20 Input Offset Current (nA) Input Offset Voltage (PV) 150 100 50 0 -50 -100 10 5 0 -5 -10 -15 -20 -150 -200 -40 15 -25 -20 0 20 40 60 80 Ambient Temperature (qC) 100 -30 -40 120 50 units at 5-V and 3-V supply 100 120 D046 Figure 46. Input Offset Current vs Ambient Temperature 5-V Supply 3-V Supply 5-V Supply 3-V Supply No. of Units in 25-pA/qC Bins 55 50 45 40 35 30 25 20 15 10 5 0 -2 5 -2 0 2 -2 5 0 -1 0 7 -1 5 5 -1 0 2 -1 5 00 -7 5 -5 0 -2 5 0 25 50 75 10 0 12 5 15 0 17 5 20 0 22 5 25 0 -1 2 . -1 8 . -1 6 . -1 4 .2 -0 1 . -0 8 . -0 6 . -0 4 .2 0 0. 2 0. 4 0. 6 0. 8 1 1. 2 1. 4 1. 6 1. 8 2 No. of Units in 0.2-PV/qC Bins 20 40 60 80 Ambient Temperature (qC) 60 Input Offset Voltage Drift (PV/qC) D047 –40°C to +125°C fit, 82 units, DBV package Figure 47. Input Offset Voltage Drift Distribution 18 0 50 units at 5-V and 3-V supply Figure 45. Input Offset Voltage vs Ambient Temperature 16 15 14 13 12 11 10 9 8 7 6 5 4 3 2 1 0 -20 D045 Input Offset Current Drift (pA/qC) D048 –40°C to +125°C fit, 82 units, DBV package Figure 48. Input Offset Current Drift Distribution Submit Documentation Feedback Copyright © 2017–2018, Texas Instruments Incorporated Product Folder Links: OPA837 OPA2837 OPA837, OPA2837 www.ti.com SBOS673D – SEPTEMBER 2017 – REVISED DECEMBER 2018 Typical Characteristics: ±2.5-V to ±1.5-V Split Supply (continued) with PD = VCC and TA ≈ 25°C (unless otherwise noted) 21 AV = 1 V/V AV = 2 V/V AV = 5 V/V AV = 10 V/V Small Signal Frequency Response (dB) ROUT (:) 140 130 120 110 100 90 80 70 60 50 40 30 20 10 0 Gain 1, 100 pF Gain 1, 1000 pF Gain 2, 100 pF Gain 2, 1000 pF Gain 5, 100 pF Gain 5, 1000 pF Gain 10, 100 pF Gain 10, 1000 pF 18 15 12 9 6 3 0 -3 -6 1 10 100 CLOAD (pF) 1k 1M 10k 10M 100M Frequency (Hz) 1G D050 D049 See Figure 66 and Table 2, small signal, targeting 30° phase margin Figure 50. Small-Signal Frequency Response vs CLOAD With Recommended ROUT Figure 49. Output Resistor vs CLOAD 3 Disable and VOUT Bipolar Supplies (V) 2 1 0 -1 Power-Down Voltage (5 V) Output Voltage (5 V) Power-Down Voltage (3 V) Output Voltage (3 V) -2 -3 1 1.1 1.2 1.3 Time (Ps} 1.4 2 1 0 -1 -2 -3 0.8 1.5 Figure 51. Turn-On Time to Sinusoidal Input 3 5.5 0.005 5 0.004 4.5 0.003 0.002 0.001 2.5 0 2 -0.001 1.5 -0.002 Disable and VOUT (V) Disable and VOUT (V) 4 3.5 1 1.1 Time (Ps) 1.2 1.3 D052 Figure 52. Turn-Off Time to Sinusoidal Input 0.006 Error to Final Value 5-V Disable 5-V VOUT 5-V %Err 3-V Disable 3-V VOUT 3-V %Err 4.5 0.9 D051 5.5 5 Power-Down Voltage (5 V) Output Voltage (5 V) Power-Down Voltage (3 V) Output Voltage (3 V) 0.006 5-V Disable 5-V VOUT 5-V %Err 3-V Disable 3-V VOUT 3-V %Err 4 3.5 3 0.005 0.004 0.003 0.002 0.001 2.5 0 2 -0.001 1.5 -0.002 1 -0.003 1 -0.003 0.5 -0.004 0.5 -0.004 -0.005 0.4 0 0 0 0.1 0.2 0.3 Time From Turn-On (Ps) 0 0.1 D053 Figure 53. Gain of 1 Turn-On Time to Final DC Value at Midscale (Simulated) 0.2 0.3 Time from Turn-On (Ps) 0.4 Error to Final Value Disable and VOUT Bipolar supplies (V) 3 -0.005 0.5 D054 Figure 54. Gain of 2 Turn-On Time to Final DC Value at Midscale (Simulated) Submit Documentation Feedback Copyright © 2017–2018, Texas Instruments Incorporated Product Folder Links: OPA837 OPA2837 19 OPA837, OPA2837 SBOS673D – SEPTEMBER 2017 – REVISED DECEMBER 2018 www.ti.com Typical Characteristics: ±2.5-V to ±1.5-V Split Supply (continued) with PD = VCC and TA ≈ 25°C (unless otherwise noted) 3 3 2.5 2 2 Output Voltage (V) Output Voltage (V) 1.5 1 VOUT +5 V VOUT +3 V VOUT 5 V VOUT 3 V 0.5 0 -0.5 -1 1 VOUT +5 V VOUT +3 V VOUT 5 V VOUT 3 V 0 -1 -1.5 -2 -2 -2.5 -3 100 -3 100m 0.1 1k RLOAD (:) Figure 55. Output Voltage Swing vs Load Resistor 10 D056 Figure 56. Output Saturation Voltage vs Load Current 840 800 800 700 Supply Current (PA) 760 Supply Current (PA) 1 IOUT (mA) D055 720 680 640 600 560 IQ 5 V IQ 3 V 600 500 400 300 200 520 100 480 440 -40 0 0.5 -20 0 20 40 60 80 Ambient Temperature (qC) 100 0.6 0.7 0.8 120 0.9 1 1.1 PD-VS (V) 1.2 1.3 1.4 1.5 D058 D057 50 units at 5-V and 3-V supply Figure 58. Supply Current vs Power-Down Voltage (Turn-On Higher Than Turn-Off) 800 150 700 60 30 0 -30 -60 -90 -120 -150 -0.4 20 20 5-V IB 5-V IB+ 5-V IOS 90 Input Bias Current (nA) Input Offset Voltage (PV) 120 0.1 0.6 1.1 1.6 2.1 2.6 3.1 3.6 Input Common-Mode Voltage (Single Supply, V) 4.1 3-V IB 3-V IB+ 3-V IOS 15 600 10 500 5 400 0 300 -5 200 -10 100 -15 0 -0.5 0 0.5 D059 1 1.5 2 2.5 3 3.5 Input Common-Mode Voltage (V) 4 -20 4.5 D060 12 units, 5-V and 3-V supplies Measured single device, 5-V and 3-V supplies Figure 59. Input Offset Voltage vs Input Common-Mode Voltage Figure 60. Input Bias and Offset Current vs VICM Submit Documentation Feedback Input Offset Current (nA) Figure 57. Supply Current vs Ambient Temperature Copyright © 2017–2018, Texas Instruments Incorporated Product Folder Links: OPA837 OPA2837 OPA837, OPA2837 www.ti.com SBOS673D – SEPTEMBER 2017 – REVISED DECEMBER 2018 Typical Characteristics: ±2.5-V to ±1.5-V Split Supply (continued) Crosstalk (dBc) with PD = VCC and TA ≈ 25°C (unless otherwise noted) -65 -70 -75 -80 -85 -90 -95 -100 -105 -110 -115 -120 -125 -130 -135 10k Ch-A to Ch-B Ch-B to Ch-A 100k 1M 10M Frequency (Hz) 100M D061 Crosstalk for the OPA2837 only Figure 61. Crosstalk vs Frequency Submit Documentation Feedback Copyright © 2017–2018, Texas Instruments Incorporated Product Folder Links: OPA837 OPA2837 21 OPA837, OPA2837 SBOS673D – SEPTEMBER 2017 – REVISED DECEMBER 2018 www.ti.com 7 Detailed Description 7.1 Overview The OPA837 and OPA2837 are single- and dual-channel, power efficient, unity-gain stable, voltage-feedback amplifiers (VFAs). Combining a negative rail input stage and a rail-to-rail output (RRO) stage, the OPAx837 provides a flexible solution where exceptional precision and wide bandwidth at low power are required. This 50-MHz gain bandwidth product (GBP) amplifier requires less than 0.65 mA of supply current per channel over a 2.7-V to 5.4-V total supply operating range. A shutdown feature on the OPA837 6-pin package version provides power savings where the system requires less than 10 µA when shut down. Offering a unity-gain bandwidth greater than 100 MHz, the OPAx837 provides less than –118-dBc THD at 100 kHz and a 2-VPP output. 7.2 Functional Block Diagrams The OPAx837 is a standard voltage-feedback op amp with two high-impedance inputs and a low-impedance output. Figure 62 and Figure 63 show the supported standard applications circuits. These application circuits are shown with a DC VREF on the inputs that set the DC operating points for single-supply designs. The VREF is often ground, especially for split-supply applications. VSIG VS+ VREF VIN OPA837 RG VOUT GVSIG VREF VREF VSRF Figure 62. Noninverting Amplifier VS+ VREF VSIG VREF RG OPA837 VOUT GVSIG V IN VREF VSRF Figure 63. Inverting Amplifier 22 Submit Documentation Feedback Copyright © 2017–2018, Texas Instruments Incorporated Product Folder Links: OPA837 OPA2837 OPA837, OPA2837 www.ti.com SBOS673D – SEPTEMBER 2017 – REVISED DECEMBER 2018 7.3 Feature Description 7.3.1 OPA837 Comparison Table 1 lists several members of the device family that includes the OPA837. Table 1. Device Family Comparison (1) (1) PART NUMBER Av = +1 BANDWIDTH (MHz) 5-V IQ (mA, Max 25°C) INPUT NOISE VOLTAGE (nV/√Hz) 2-VPP THD (dBc, 100 kHz) RAIL-TO-RAIL INPUT/OUTPUT DUALS OPA837 105 0.63 4.7 –118 VS–, output OPA2837 OPA838 — 0.99 1.9 –110 VS–, output — LMV118 45 0.9 40 — VS–, output — LMH6647 55 1.6 17 –75 Input, output LMH6646 OPA835 56 0.35 9.4 –104 VS–, output OPA2835 OPA625 120 2.2 2.5 –120 VS–, output OPA2625 OPA836 205 1.0 4.6 –118 VS–, output OPA2836 For a complete selection of TI high speed amplifiers, visit www.ti.com. 7.3.2 Input Common-Mode Voltage Range When the primary design goal is a linear amplifier with high CMRR, the design must remain within the input common-mode voltage range (VICR) of an op amp. These ranges are referenced off of each supply as an input headroom requirement. Ensured operation at 25°C is maintained to the negative supply voltage and to within 1.3 V of the positive supply voltage. The common-mode input range specifications in the Electrical Characteristics table use CMRR to set the limit. The limits are selected to ensure CMRR does not degrade more than 3 dB below the minimum CMRR value if the input voltage is within the specified range. Assuming the op amp is in linear operation, the voltage difference between the input pins is small (ideally 0 V); and the input common-mode voltage is analyzed at either input pin with the other input pin assumed to be at the same potential. The voltage at VIN+ is simple to evaluate. In the noninverting configuration of Figure 62, the input signal, VIN, must not violate the VICR. In the inverting configuration of Figure 63, the reference voltage, VREF, must be within the VICR. The input voltage limits have fixed headroom to the power rails and track the power-supply voltages. For one 5-V supply, the typical linear input voltage ranges from –0.2 V to 3.8 V and –0.2 V to 1.5 V for a 2.7-V supply. The delta headroom from each power-supply rail is the same in either case: –0.2 V and 1.2 V, respectively. 7.3.3 Output Voltage Range The OPAx837 is a rail-to-rail output op amp. Rail-to-rail output typically means that the output voltage swings to within 100 mV of the supply rails. There are two different ways to specify this feature: one is with the output still in linear operation and another is with the output saturated. Saturated output voltages are closer to the powersupply rails than the linear outputs, but the signal is not a linear representation of the input. Saturation and linear operation limits are affected by the output current, where higher currents lead to more voltage loss in the output transistors; see Figure 55. The Electrical Characteristics tables list saturated output voltage specifications with a 2-kΩ load. Figure 55 illustrates the saturated voltage-swing limits versus output load resistance, and Figure 56 illustrates the output saturation voltage versus load current. Given a light load, the output voltage limits have nearly constant headroom to the power rails and track the power-supply voltages. For example, with a 2-kΩ load and a single 5-V supply, the linear output voltage ranges from 0.10 V to 4.9 V and ranges from 0.1 V to 2.6 V for a 2.7-V supply. The delta from each power-supply rail is the same in either case: 0.1 V. With devices like the OPA837 and OPA2837 where the input range is lower than the output range, typically the input limits the available signal swing only in a noninverting gain of 1 V/V. Signal swing in noninverting configurations in gains greater than +1 V/V and inverting configurations in any gain are typically limited by the output voltage limits of the op amp. Submit Documentation Feedback Copyright © 2017–2018, Texas Instruments Incorporated Product Folder Links: OPA837 OPA2837 23 OPA837, OPA2837 SBOS673D – SEPTEMBER 2017 – REVISED DECEMBER 2018 www.ti.com 7.3.4 Power-Down Operation The OPA837 includes a power-down mode in the 6-pin SOT23-6 package. Under logic control, the amplifier can switch from normal operation to a standby current of less than 10 µA. When the PD pin is connected high, the amplifier is active. Connecting the PD pin low disables the amplifier and places the output in a high-impedance state. When the amplifier is configured as a unity-gain buffer, the output stage is in a high DC-impedance state. A new feature in the OPA837 is a switch from the external inverting input pin to the internal active transistors. This switch operates with the disable pin function to open up the connection to the internal devices when powered down. Operating in unity gain provides a high-impedance voltage into both the output and inverting input pins. This feature allows direct active multiplexer operation to be implemented; see Figure 87. The TIDA01565 Wired OR MUX and PGA Reference Design demonstrates the use of the OPAx837 in wired-OR multiplexer and programmable gain amplifier applications. When disabled, the internal input devices on the inverting input approximately follow the noninverting input on the other side of the open switch through the backto-back protection diodes across the inputs. When powered up, these diodes (two in each direction) act to limit overdrive currents into the active transistors. The PD pin must be actively driven high or low and must not be left floating. If the power-down mode is not used, PD must be tied to the positive supply rail. PD logic states are referenced relatively low to the negative supply rail, VS–. When the op amp is powered from a single-supply and ground, and the disable line is driven from logic devices with similar VDD voltages to the op amp, the disable operation does not require any special consideration. The OPA837 is specified to be off with PD driven to within 0.55 V of the negative supply and specified to be on when driven more than 1.5 V above the negative supply. Slight hysteresis is provided around a nominal 1-V switch point; see Figure 58. When the op amp is powered from a split supply with VS– below ground, a level shift logic swing below ground is required to operate the disable function. 7.3.5 Low-Power Applications and the Effects of Resistor Values on Bandwidth The OPAx837 can use a direct short in the feedback for a gain of 1 V/V. Table 2 gives a list of recommended values over gain for an increasing noninverting gain target. This table was produced by increasing the R values until they added 50% of the total output noise power. Higher values can be used to reduce power at the cost of higher noise. Lower values can be used to reduce the total output noise at the cost of more load power in the feedback network. Stability is also impaired going to very high values because of the pole introduced into the feedback path with the inverting input capacitance (1.5-pF common-mode). In low-power applications, reducing the current in the feedback path is preferable by increasing the resistor values. Using larger value resistors has two primary side effects (other than lower power) because of the interactions with the inverting input parasitic capacitance. Using large value resistors lowers the bandwidth and lowers the phase margin. When the phase margin is lowered, peaking in the frequency response and overshoot and ringing in the pulse response results. Figure 64 shows the gain = 2 V/V (6 dB) small-signal frequency response with RF and RG equal to 1 kΩ, 2 kΩ, 5 kΩ, 10 kΩ, and 20 kΩ. This test was done with RL = 2 kΩ. Lower RL values can reduce the peaking because of RL loading effects, but higher values do not have a significant effect. 18 15 Gain (dB) 12 Rf = 1 kOhm Rf = 2 kOhm Rf = 5 kOhm Rf = 10 kOhm Rf = 20 kOhm Rf = 20 kOhm || 1.5 pF 9 6 3 0 10k 100k 1M Frequency (Hz) 10M 100M D063 Figure 64. Frequency Response With Various RF = RG Resistor Values 24 Submit Documentation Feedback Copyright © 2017–2018, Texas Instruments Incorporated Product Folder Links: OPA837 OPA2837 OPA837, OPA2837 www.ti.com SBOS673D – SEPTEMBER 2017 – REVISED DECEMBER 2018 As expected, larger value resistors cause lower bandwidth and peaking in the response (peaking in frequency response is synonymous with overshoot and ringing in pulse response). Adding a 1.5-pF capacitor in parallel with RF (equal to the input common-mode capacitance) helps compensate the phase margin loss and restores flat frequency response. Figure 65 shows the test circuit. VIN RG VOUT OPA837 2 kW RF Optional CF Figure 65. G = 2 Test Circuit for Various Gain-Setting Resistor Values 7.3.6 Driving Capacitive Loads The OPAx837 can drive a parasitic load capacitance up through 4 pF on the output with no special considerations. When driving capacitive loads greater than 4 pF, TI recommends using a small resistor (RO) in series with the output as close to the device as possible. Without RO, output capacitance interacts with the output impedance of the amplifier causing phase shift in the loop gain of the amplifier that reduces the phase margin. This reduction causes peaking in the frequency response and overshoot and ringing in the pulse response. Inserting RO isolates the phase shift from the loop-gain path and restores the phase margin; however RO can also limit bandwidth to the capacitive load. Figure 66 shows the test and Figure 49 illustrates the recommended values of RO versus capacitive loads, CL using a 30° phase margin target for the op amp. See Figure 50 for the frequency responses with various values of CL and ROUT parametric on gain. RO VIN VOUT OPA837 CL 2 kΩ Figure 66. ROUT versus CL Test Circuit Submit Documentation Feedback Copyright © 2017–2018, Texas Instruments Incorporated Product Folder Links: OPA837 OPA2837 25 OPA837, OPA2837 SBOS673D – SEPTEMBER 2017 – REVISED DECEMBER 2018 www.ti.com 7.4 Device Functional Modes 7.4.1 Split-Supply Operation (±1.35 V to ±2.7 V) To facilitate testing with common lab equipment, the OPA837EVM (see the OPA835DBV and OPA836DBV EVM User's Guide) allows split-supply operation. This configuration eases lab testing because the mid-point between the power rails is ground, and most signal generators, network analyzers, oscilloscopes, spectrum analyzers, and other lab equipment have inputs and outputs that prefer a ground reference for DC-coupled testing. Figure 67 shows a simple noninverting configuration analogous to Figure 62 with a ±2.5-V supply and VREF equal to ground. The input and output swing symmetrically around ground. For ease of use, split supplies are preferred in systems where signals swing around ground. In this example, an optional bias current cancellation resistor is used in series with the noninverting input. For DC-coupled applications, set this resistor to be equal to the parallel combination of RF and RG. This resistor increases the noise contribution at the input because of that resistor noise (see the Output Noise Calculations section). +2.5 V RF // RG RG VSIG VOUT OPA837 Load -2.5 V RF Figure 67. Split-Supply Operation Figure 68 shows the step response for this gain of 2-V/V circuit with a ±1-V input to a ±2-V output. For a 4-V output step, the input edge rate is set to 40 ns to avoid slew limiting. 2.5 Input and Output Voltage (V) 2 1.5 1 0.5 0 -0.5 -1 -1.5 Input Output -2 -2.5 0 200 400 600 Time (ns) 800 1000 1200 D064 Figure 68. VIN and VOUT vs Time 26 Submit Documentation Feedback Copyright © 2017–2018, Texas Instruments Incorporated Product Folder Links: OPA837 OPA2837 OPA837, OPA2837 www.ti.com SBOS673D – SEPTEMBER 2017 – REVISED DECEMBER 2018 Device Functional Modes (continued) 7.4.2 Single-Supply Operation (2.7 V to 5.4 V) Most newer systems use a single power supply to improve efficiency and to simplify power-supply design. The OPAx837 can be used with single-supply power (ground for the negative supply) with no change in performance from split supply, as long as the input and output pins are biased within the linear operating region of the device. The outputs nominally swing rail-to-rail with approximately a 100-mV headroom required for linear operation. The inputs can typically swing 0.2 V below the negative rail (typically ground) and to within 1.2 V of the positive supply. For DC-coupled single-supply operation, the input swing is below the available output swing range for noninverting gains greater than 1.30 V/V. Typically, the 1.2-V input headroom required to the positive supply only limits output swing range for a unity-gain buffer. To change the circuit from split supply to single-supply, level shift all voltages by half the difference between the power-supply rails. For example, Figure 69 depicts changing from a ±2.5-V split supply to a 5-V single-supply. The load is shown as mid-supply referenced but can be grounded as well. 5V RG VOUT OPA837 VSIG Load RF 2.5 V Figure 69. Single-Supply Concept A practical circuit has an amplifier or other circuit providing the bias voltage for the input, and the output of this amplifier stage provides the bias for the next stage. Figure 70 shows a typical noninverting amplifier circuit. With 5-V single-supply, a mid-supply reference generator is needed to bias the negative side through RG. To cancel the voltage offset that is otherwise caused by the input bias currents, R1 is selected to be equal to RF in parallel with RG. For example, if a gain of 2 V/V is required and RF = 2 kΩ, select RG = 2 kΩ to set the gain, and R1 = 1 kΩ for bias current cancellation which reduces the output DC error to IOS × RF. The value for C is dependent on the reference, and TI recommends a value of at least 0.1 µF to limit noise. The frequency response flatness is impacted by the AC impedance, including the reference and capacitor added to the RG element. Signal and bias from previous stage VSIG 2.5 V 5V R1 RO VOUT OPA837 5V GVSIG RG 2.5 V REF 2.5 V C RF Signal and bias to next stage Figure 70. Noninverting Single-Supply Operation With Reference Submit Documentation Feedback Copyright © 2017–2018, Texas Instruments Incorporated Product Folder Links: OPA837 OPA2837 27 OPA837, OPA2837 SBOS673D – SEPTEMBER 2017 – REVISED DECEMBER 2018 www.ti.com Device Functional Modes (continued) Figure 71 shows a similar noninverting single-supply scenario with the reference generator replaced by the Thevenin equivalent using resistors and the positive supply. RG’ and RG” form a resistor divider from the 5-V supply and are used to bias the negative side with the parallel sum equal to the equivalent RG to set the gain. To cancel the voltage offset that is otherwise caused by the input bias currents, R1 is selected to be equal to RF in parallel with RG’ in parallel with RG” (R1= RF || RG’ || RG”). For example, if a gain of 2 V/V is required and RF = 2 kΩ, selecting RG’ = RG” = 4 kΩ gives an quivalent parallel sum of 2 kΩ, sets the gain to 2, and references the input to mid-supply (2.5 V). R1 is set to 1 kΩ for bias current cancellation. The resistor divider costs less than the 2.5-V reference in Figure 70 but increases the current from the 5-V supply. Any noise or variation on the 5-V supply now also comes into the circuit as an input through the biasing path. Signal and bias from previous stage VSIG 2.5 V 5V R1 RO RG’ VOUT OPA837 GVSIG 5V 2.5 V RG” Signal and bias to next stage RF Figure 71. Noninverting Single-Supply Operation With Resistor Mid-Supply Biasing Figure 72 shows a typical inverting amplifier circuit. With a 5-V single supply, a mid-supply reference generator is needed to bias the positive side through R1. To cancel the voltage offset that is otherwise caused by the input bias currents, R1 is selected to be equal to RF in parallel with RG. For example, if a gain of –2 V/V is required and RF = 2 kΩ, select RG = 1 kΩ to set the gain and R1 = 667 Ω for bias current cancellation. The value for C is dependent on the reference, but TI recommends a value of at least 0.1 µF to limit noise into the op amp. 5V R1 5V 2.5 V REF RO C V OUT OPA837 GVSIG 2.5 V RG VSIG RF Signal and bias to next stage 2.5 V Signal and bias from previous stage Figure 72. Inverting Single-Supply Operation With Reference 28 Submit Documentation Feedback Copyright © 2017–2018, Texas Instruments Incorporated Product Folder Links: OPA837 OPA2837 OPA837, OPA2837 www.ti.com SBOS673D – SEPTEMBER 2017 – REVISED DECEMBER 2018 Device Functional Modes (continued) Figure 73 shows a similar inverting single-supply scenario with the reference generator replaced by the Thevenin equivalent using resistors and the positive supply. R1 and R2 form a resistor divider from the 5-V supply and are used to bias the positive side. To cancel the voltage offset that is otherwise caused by the input bias currents, set the parallel value of R1 and R2 equal to the parallel value of RF and RG. C must be added to limit coupling of noise into the positive input. For example, if gain of –2 V/V is required and RF = 2 kΩ, select RG = 1 kΩ to set the gain. R1 = R2 = 2 × 667 Ω = 1.33 kΩ for the mid-supply voltage bias and for op-amp input-bias current cancellation. A good value for C is 0.1 µF. The resistor divider costs less than the 2.5-V reference in Figure 72 but increases the current from the 5-V supply. Any noise or variation in the 5-V supply also comes into the circuit through this bias setup but be band-limited by the pole formed with R1 || R2 and C. 5V 5V R1 RO R2 C VOUT OPA837 GVSIG 2.5 V RG VSIG RF Signal and bias to next stage 2.5 V Signal and bias from previous stage Figure 73. Inverting Single-Supply Operation With Resistor Midsupply Biasing These examples are only a few of the ways to implement a single-supply design. Many other designs exist that can often be simpler if AC-coupled inputs are allowed. A good compilation of options can be found in the SingleSupply Op Amp Design Techniques application report. Submit Documentation Feedback Copyright © 2017–2018, Texas Instruments Incorporated Product Folder Links: OPA837 OPA2837 29 OPA837, OPA2837 SBOS673D – SEPTEMBER 2017 – REVISED DECEMBER 2018 www.ti.com 8 Application and Implementation NOTE Information in the following applications sections is not part of the TI component specification, and TI does not warrant its accuracy or completeness. TI’s customers are responsible for determining suitability of components for their purposes. Customers should validate and test their design implementation to confirm system functionality. 8.1 Application Information 8.1.1 Noninverting Amplifier The OPAx837 can be used as a noninverting amplifier with a signal input to the noninverting input, VIN+. A basic block diagram of the circuit is illustrated in Figure 62. VREF is often ground when split supplies are used. Calculate the amplifier output according to Equation 1 if VIN = VREF + VSIG. æ RF ö V = VSIG ç 1 + ÷ + VREF OUT RG ø è (1) The signal gain of the circuit is set by Equation 2, and VREF provides a reference around which the input and output signals swing. Output signals are in-phase with the input signals within the flat portion of the frequency response. For a high-speed, low-noise device such as the OPAx837, the values selected for RF (and RG for the desired gain) can strongly influence the operation of the circuit. For the characteristic curves, the noninverting circuit of Figure 74 shows the test configuration set for a gain of 2 V/V. Table 2 lists the recommended resistor values over gain. RF G= 1 + RG (2) RF 2 NŸ 50-Ÿ source Network Analyzer RT 50 Ÿ RS 50 Ÿ U1 OPA837 ± 50-Ÿ Cable + + R3 1.96 NŸ 2-kŸ load PD 50-Ÿ Cable Network Analyzer RLOAD 50 Ÿ VEE R6 51.1 Ÿ RG 2 NŸ V VM1 VCC VCC + VEE + V1 2.5 V2 -2.5 V Figure 74. Characterization Test Circuit for Network, Spectrum Analyzer 30 Submit Documentation Feedback Copyright © 2017–2018, Texas Instruments Incorporated Product Folder Links: OPA837 OPA2837 OPA837, OPA2837 www.ti.com SBOS673D – SEPTEMBER 2017 – REVISED DECEMBER 2018 Application Information (continued) Table 2 lists the recommended resistor values from target gains of 1 V/V to 10 V/V where standard E96 values are shown. This table controls the RF and RG values to set the resistor noise contribution at approximately 50% of the total output noise power. These values increase the spot noise at the output over what the op amp voltage noise produces by 41%. Lower values reduce the output noise of any design at the cost of more power in the feedback circuit. Using the TINA model and simulation tool shows the impact of different resistor value choices on response shape and noise. Table 2. Noninverting Recommended Resistor Values TARGET GAIN (V/V) RF (Ω) RG (Ω) ACTUAL GAIN (V/V) GAIN (dB) 1 0 Open 1.00 0.00 1.5 1190 2370 1.50 3.53 2 2000 2000 2.00 6.02 3 2260 1130 3.00 9.54 4 2370 787 4.01 12.07 5 2490 619 5.02 14.02 6 2550 511 5.99 15.55 7 2610 432 7.04 16.95 8 2670 383 7.97 18.03 9 2670 332 9.04 19.13 10 2670 294 10.08 20.07 8.1.2 Inverting Amplifier The OPAx837 can be used as an inverting amplifier with a signal input to the inverting input, VIN–, through the gain-setting resistor RG. A basic block diagram of the circuit is illustrated in Figure 63. The output of the amplifier can be calculated according to Equation 3 if VIN = VREF + VSIG and the noninverting input is biased to VREF. æ -R VOUT = VSIG ç F è RG ö ÷ + VREF ø (3) Submit Documentation Feedback Copyright © 2017–2018, Texas Instruments Incorporated Product Folder Links: OPA837 OPA2837 31 OPA837, OPA2837 SBOS673D – SEPTEMBER 2017 – REVISED DECEMBER 2018 www.ti.com The signal gain of the circuit is set by Equation 4 and VREF provides a reference point around which the input and output signals swing. For bipolar-supply operation, VREF is often ground. The output signal is 180˚ out-of-phase with the input signal in the pass band of the application. Figure 75 shows the 50-Ω input matched configuration used for the inverting characterization plots set up for a gain of –1 V/V. In this case, an added termination resistor, RT, is placed in parallel with the input RG resistor to provide an impedance match to 50-Ω test equipment. The output network appears as a 2-kΩ load but with a 50-Ω source to the network analyzer. This output interface network does add a 37.9-dB insertion loss that is normalized out in the characterization curves. Table 3 lists the suggested values for RF, RG, and RT for inverting gains from –0.5 V/V to –10 V/V. If a 50-Ω input match is not required, eliminate the RT element. -RF G= RG (4) + VCC RF 2 NŸ ± + VEE Å 50-Ÿ source VEE 50-Ÿ Cable R1 1.96 NŸ + PD 2-kŸ load Æ + + 2.5 V -2.5 V Network Analyzer RLOAD 50 Ÿ RG 2 NŸ RT 51.1 Ÿ RS 50 Ÿ Gain of -1 V/V 50-Ÿ Cable R2 51.1 Ÿ Network Analyzer V VCC Figure 75. Inverting Characterization Circuit for Network Analyzer Table 3. Inverting Recommended Resistor Values INVERTING GAIN (V/V) RF (Ω) RG (Ω) STANDARD RT (Ω) INPUT ZI (Ω) ACTUAL (V/V) GAIN (dB) –0.5 1190 2370 51.1 50.02 –0.50 –5.98 –1 2000 2000 51.1 49.83 –1.00 0.00 –2 2260 1130 52.3 49.99 –2.00 6.02 –3 2370 787 53.6 50.18 –3.01 9.58 –4 2490 619 54.9 50.43 –4.02 12.09 –5 2550 511 54.9 49.57 –4.99 13.96 –6 2610 432 56.2 49.73 –6.04 15.62 –7 2670 383 57.6 50.07 –6.97 16.87 –8 2670 332 59 50.10 –8.04 18.11 –9 2670 294 60.4 50.11 –9.08 19.16 –10 2670 267 61.9 50.25 –10.00 20.00 32 Submit Documentation Feedback Copyright © 2017–2018, Texas Instruments Incorporated Product Folder Links: OPA837 OPA2837 OPA837, OPA2837 www.ti.com SBOS673D – SEPTEMBER 2017 – REVISED DECEMBER 2018 8.1.3 Output DC Error Calculations The OPAx837 can provide excellent DC signal accuracy because of its high open-loop gain, high common-mode rejection, high power-supply rejection, and low input offset voltage and bias current offset errors. To take full advantage of this low input offset voltage, pay careful attention to input bias current cancellation. The low-noise input stage for the OPAx837 has a relatively high input bias current (0.34 µA typical out the pins) but with a close match between the two input currents. The OPAx837 is a negative rail input device using PNP input devices where the base current flows out of the device pins. A large resistor to ground on the V+ input shifts the pin voltage positively because of the input bias current. The mismatch between the two input bias currents is very low, typically only ±10 nA of input offset current. Match the DC source impedances out of the two inputs to reduce the total output offset voltage. Figure 67 illustrates an example of resistor matching for bias current cancellation. Analyzing the simple circuit of Figure 67 (using a gain of 2-V/V target with RF = RG = 2 kΩ) illustrates that the noise gain for the input offset voltage drift is 1 + 2 kΩ / 2 kΩ = 2 V/V. This value results in an output drift term of ±1.6 µV/°C × 2 = ±3.2 µV/°C (DCK package). Because the two impedances out of the inputs are matched, the residual error from the maximum ±250 pA/°C offset current drift is this maximum IOS drift times the 2-kΩ feedback resistor value, or ±50 µV/°C. The total output DC error drift band is ±53.2 µV/°C. If the output DC drift is more important than reduced feedback currents, lower the resistor values to reduce the dominant drift term resulting from the IOS term. 8.1.4 Output Noise Calculations The unity-gain stable, voltage-feedback OPAx837 op amp offers among the lowest input voltage and current noise terms for any device with a supply current less than 0.7 mA. Figure 76 shows the op amp noise analysis model that includes all noise terms. In this model, all noise terms are shown as noise voltage or current density terms in nV/√Hz or pA/√Hz. ENI + OPA837 RS EO IBN ERS RF 4kTRS RG 4kTRF IBI 4kT RG 4kT = 1.6E ± 20J at 290° K Figure 76. Op Amp Noise Analysis Model The total output spot noise voltage is computed as the square root of the squared contributing terms to the output noise voltage. This computation is adding all the contributing noise powers at the output by superposition, then taking the square root to return to a spot noise voltage. The last term includes the noise for both the RG and RF resistors. Equation 5 shows the general form for this output noise voltage using the terms presented in Figure 76. EO 2 ªENI ¬ IBNRS 4kTRS º NG2 ¼ IBIRF 2 4kTRFNG (5) Dividing this expression by the noise gain (NG = 1 + RF / RG), as shown in Equation 6, gives the equivalent input referred spot noise voltage at the noninverting input. EN 2 ENI IBNRS 2 4kTRS § IBIRF · ¨ NG ¸ © ¹ 2 4kTRF NG (6) Submit Documentation Feedback Copyright © 2017–2018, Texas Instruments Incorporated Product Folder Links: OPA837 OPA2837 33 OPA837, OPA2837 SBOS673D – SEPTEMBER 2017 – REVISED DECEMBER 2018 www.ti.com Using the resistor values listed in Table 2 with RS = 0 Ω results in a constant input-referred voltage noise of < 7 nV/√Hz. Reducing the resistor values can reduce this noise value towards the 4.7 nV/√Hz intrinsic to the OPA837. As shown in Equation 5, adding the RS for bias current cancellation in noninverting mode adds the noise from the RS to the total output noise. In inverting mode, bypass the RS bias current cancellation resistor with a capacitor for the best noise performance. For more details on op amp noise analysis, see the Noise Analysis for High-Speed Op Amps application report. 8.1.5 Instrumentation Amplifier Figure 77 is an instrumentation amplifier that combines the high input impedance of the differential-to-differential amplifier circuit and the common-mode rejection of the differential-to-single-ended amplifier circuit. This circuit is often used in applications where high input impedance is required (such as taps from a differential line) or in cases where the signal source is a high impedance. VIN½OPA837 VSIG- RF2 VCM RG2 RF1 RG1 RG2 VSIG+ G[(VSIG+)-(VSIG-)] RF2 VCM ½OPA837 VIN+ VOUT OPA837 RF1 VREF VREF Figure 77. Instrumentation Amplifier (INA) The output of the amplifier can be calculated according to Equation 7 if VIN+ = VCM + VSIG+ and VIN– = VCM + VSIG–. VOUT = (VIN+ - VIN- ) æ 2RF1 ö æ RF2 ö ´ ç1 + ÷ ç ÷ + VREF RG1 ø è RG2 ø è (7) Equation 8 shows the signal gain of the circuit. The input VCM is rejected, and VREF provides a reference voltage or level shift around which the output signal swings. The single-ended output signal is in-phase to the lower input signal polarity. æ 2R F1 ö æ R F2 ö G = ç1 + ÷ ç ÷ R G 1 ø è R G2 ø è (8) Integrated INA solutions are available, but the OPAx837 device provides a high-frequency solution at relatively low power (< 1.8 mA for the three op-amp solution). For best CMRR performance, resistors must be matched. A good rule of thumb is CMRR ≈ the resistor tolerance; so a 0.1% tolerance provides approximately 60-dB CMRR. For higher gain INA implementations with higher bandwidths, apply the OPA838 to the circuit of Figure 77. 8.1.6 Attenuators The noninverting circuit of Figure 62 has a minimum gain of 1. To implement attenuation, a resistor divider can be placed in series with the positive input, and the amplifier set for a gain of 1 V/V by shorting VOUT to VIN– and removing RG. Because the op amp input is high impedance, the resistor divider sets the attenuation. The inverting circuit of Figure 63 is used as an attenuator by making RG larger than RF. The attenuation is the resistor ratio. For example, a 10:1 attenuator can be implemented with RF = 2 kΩ and RG = 20 kΩ. 34 Submit Documentation Feedback Copyright © 2017–2018, Texas Instruments Incorporated Product Folder Links: OPA837 OPA2837 OPA837, OPA2837 www.ti.com SBOS673D – SEPTEMBER 2017 – REVISED DECEMBER 2018 8.1.7 Differential to Single-Ended Amplifier Figure 78 shows a differential amplifier that converts differential signals to single-ended in a single stage and provides gain (or attenuation) and level shifting. This circuit can be used in applications such as a line receiver for converting a differential signal from a Cat5 cable to a single-ended output signal. VSIG- RF VCM RG VINVIN+ RG VOUT OPA837 G[(VSIG+)-(VSIG-)] VSIG+ VCM RF VREF VREF Figure 78. Differential to Single-Ended Amplifier The output of the amplifier can be calculated according to Equation 9 if VIN+ = VCM + VSIG+ and VIN– = VCM + VSIG–. æR ö VOUT = (VIN+ - VIN - ) ´ ç F ÷ + VREF è RG ø (9) The signal gain of the circuit is shown in Equation 10, VCM is rejected, and VREF provides a level shift or reference voltage around which the output signal swings. The single-ended output signal is in-phase with the noninverting input signal. VREF is often ground when split supplies are used on the op amp. RF G= RG (10) Line termination can be accomplished by adding a shunt resistor across the VIN+ and VIN– inputs. The differential impedance is the shunt resistance in parallel with the input impedance of the amplifier circuit, which is usually much higher. For low gain and low line impedance, the resistor value to add is approximately the impedance of the line. For example, if a 100-Ω Cat5 cable is used with a gain of 1 V/V amplifier and RF = RG = 2 kΩ, adding a 100-Ω shunt across the input gives a differential impedance of 99 Ω, which is an adequate match for most applications. For best CMRR performance, resistors must be matched. Assuming CMRR ≈ the resistor tolerance, a 0.1% tolerance provides approximately 60-dB CMRR. Submit Documentation Feedback Copyright © 2017–2018, Texas Instruments Incorporated Product Folder Links: OPA837 OPA2837 35 OPA837, OPA2837 SBOS673D – SEPTEMBER 2017 – REVISED DECEMBER 2018 www.ti.com 8.1.8 Differential-to-Differential Amplifier Figure 79 shows a differential amplifier that is used to amplify differential signals to a differential output. This circuit has high input impedance and is used in differential line driver applications where the signal source is a high-impedance driver (for example, a differential DAC) that must drive a line. VIN½ OPA837 VOUT- VSIGGVSIG- VCM VCM RF RG RF VSIG+ GVSIG+ VCM VCM ½ OPA837 VOUT+ VIN+ Figure 79. Differential-to-Differential Amplifier The output of the amplifier can be calculated according to Equation 11 if VIN± is set to VCM + VSIG±. æ 2RF ö V = VIN ± ´ ç 1 + ÷ + VCM OUT ± RG ø è (11) The signal gain of the circuit is shown in Equation 12, and VCM passes with unity gain. The amplifier combines two noninverting amplifiers into one differential amplifier that shares the RG resistor, which makes RG effectively half its value when calculating the gain. The output signals are in-phase with the input signals. 2RF G= 1 + RG (12) 36 Submit Documentation Feedback Copyright © 2017–2018, Texas Instruments Incorporated Product Folder Links: OPA837 OPA2837 OPA837, OPA2837 www.ti.com SBOS673D – SEPTEMBER 2017 – REVISED DECEMBER 2018 8.1.9 Pulse Application With Single-Supply Circuit For pulsed applications where the signal is at ground and pulses to a positive or negative voltage, the circuit bias-voltage considerations differ from those in an application with a signal that swings symmetrically around a reference point. Figure 80 shows a circuit where the signal is at ground (0 V) and pulses to a positive value. The waveforms are shown slightly above ground because the output stage requires approximately 100 mV headroom to the supplies. To operate with the I/O swing truly to ground on a single-supply setup, consider using the fixed –0.23-V output LM7705. Signal and bias from previous stage VSIG 0V 5V R1 RO VOUT OPA837 GVSIG RG 0V RF Signal and bias to next stage Figure 80. Noninverting Single-Supply Circuit With Pulse As shown in Figure 81, an inverting amplifier is more appropriate if the input signal pulses negative from ground. A key consideration in noninverting and inverting cases is that the input and output voltages are kept within the limits of the amplifier. Because the VICR of the OPA837 includes the negative supply rail, the OPA837 op amp is well-suited for this application. 5V R1 RO VOUT OPA837 GVSIG RG Signal and bias from previous stage 0V 0V RF Signal and bias to next stage VSIG Figure 81. Inverting Single-Supply Circuit With Pulse Submit Documentation Feedback Copyright © 2017–2018, Texas Instruments Incorporated Product Folder Links: OPA837 OPA2837 37 OPA837, OPA2837 SBOS673D – SEPTEMBER 2017 – REVISED DECEMBER 2018 www.ti.com 8.1.10 ADC Driver Performance The OPAx837 provides excellent performance when driving high-performance delta-sigma (ΔΣ) or successiveapproximation-register (SAR) ADCs in low-power audio and industrial applications. Figure 82 repeats the front page diagram. Many designs prefer to work with a true 0-V input range to 0-V output at the ADC. The 100-mV output headroom requirement for the OPAx837 then requires a small negative supply to hold the output linearity to ground. This supply is provided in this example using the low-cost LM7705 fixed negative, –0.23-V output regulator. On a 5-V supply, the input headroom requires at least a 1.2-V headroom to that supply. As shown in Figure 82, this requirement limits the maximum input to 3.8 V. The SAR operates with a precision 4.096-V reference provided by the REF5040, where the gain of 1.05 V/V takes the 3.8-V maximum input to a 4.0-V maximum output. The RC values have been set to limit the overshoot at the OPAx837 output pin to reduce clipping on fast (50 ns) transitions. + 5.0 V REF5040 4.096 V GND 3.3 V +VCC 0V 0V 4.0 V PD + OPA837 ± 3.8 V Gain = 1.05 V/V ±VCC 22 Ÿ 2.2 nF 499 Ÿ ADS8860 16-Bit SAR 1 MSPS 22 Ÿ 10.0 kŸ GND GND GND ± 0.23 V LM7705 + 3.3 V Figure 82. OPA837 and ADS8860 Example Circuit 38 Submit Documentation Feedback Copyright © 2017–2018, Texas Instruments Incorporated Product Folder Links: OPA837 OPA2837 OPA837, OPA2837 www.ti.com SBOS673D – SEPTEMBER 2017 – REVISED DECEMBER 2018 8.2 Typical Applications 8.2.1 Active Filters The OPAx837 is a good choice for active filters. Figure 83 and Figure 84 show MFB and Sallen-Key circuits designed implementing second-order, low-pass Butterworth filter circuits. Figure 85 illustrates the frequency response. The main difference is that the MFB active filter provides an inverting amplifier in the pass band and the SallenKey active filter is noninverting. The primary advantage for each active filter is that the Sallen-Key filter in unity gain has no resistor gain error term or feedback resistor noise contribution. The MFB active filter has better attenuation properties beyond the bandwidth of the op amp. The example circuits are assuming a split-supply operation but single-supply operation is possible with midscale biasing. 1.4 kW 430 pF 1.4 kW 1.91 kW OPA837 2.2 nF Figure 83. MFB Active Filter, 100-kHz, Second-Order, Low-Pass Butterworth Filter Circuit 12 nF 147 W 255 W OPA837 5.6 nF Figure 84. Sallen-Key Active Filter, 100-kHz, Second-Order, Low-Pass Butterworth Filter Circuit 8.2.1.1 Design Requirements For both designs, target the following filter shape characteristic: • Gain of 1 V/V • 100-kHz Butterworth response • Q = 0.707 gives a flat Butterworth design Scale the resistors down to reduce their noise contribution. In the MFB design, the input resistor is the in-band load to the prior stage. Use values slightly below the gain of –1 V/V in Table 3. The Sallen-Key filter shows a high impedance input in-band, so scale those resistors down further to improve noise. The output DC error and drift can be improved by adding bias current cancellation resistors. For the MFB filter that is a resistor (and a noise filter capacitor) on the noninverting input to ground equal to the resistor inside the loop times the noise gain. For the Sallen-Key design, add a feedback resistor equal to the sum of the two input resistors. 8.2.1.2 Detailed Design Procedure The filter designs shown in this section used an improved design flow that reduces the resistor noise and noise gain peaking. For the MFB filter, the design was based on the information in the Design Methodology for MFB Filters in ADC Interface Applications application note. For the Sallen-Key design, the solution is based on the information in the Component Pre-Distortion for Sallen Key Filters application note. Submit Documentation Feedback Copyright © 2017–2018, Texas Instruments Incorporated Product Folder Links: OPA837 OPA2837 39 OPA837, OPA2837 SBOS673D – SEPTEMBER 2017 – REVISED DECEMBER 2018 www.ti.com Typical Applications (continued) 8.2.1.3 Application Curves Figure 85 shows the comparative response curves for each of the filter design examples. Both filters hit the desired response shape exactly. However, notice the loss of stop-band rejection in the Sallen-Key design. This loss results from the op amp output impedance increasing at higher frequencies and allowing the signal to feed through the feedback capacitor to the output. Figure 86 shows a comparison of the output spot noise for the two designs. The Sallen-Key is much lower because of the lower resistor values used. Also, the MFB shows a noise gain of 2 V/V versus the Sallen-Key gain of 1 V/V. This difference immediately increases the MFB output noise by at least twice the input voltage noise from the op amp. The higher resistor values also increase the total output noise for the MFB. 3 20 MFB SKF -3 16 Output Noise (nV/—Hz) Gain Magnitude (dB) -9 -15 -21 -27 -33 -39 14 12 10 8 6 4 -45 -51 1k MFB SKF 18 2 10k 100k Frequency (Hz) 1M 10M 0 100 1k D083 Figure 85. MFB and Sallen-Key Active Filters, SecondOrder, Low-Pass Butterworth Filter Response 10k Frequency (Hz) 100k 1M D084 Figure 86. Output Spot Noise Comparison 8.2.2 Implementing a 2:1 Active Multiplexer The OPA837 includes a unique feature that enables a much improved wired-or mux operation. When disabled, an internal switch opens from the inverting input to the active transistors isolating those nonlinear loads from the signal being driven back into the inverting input through the active channel. Figure 87 illustrates a simple example of this multiplexer. In this figure, one of two signals are selected to be passed on to a shared output. The logic control turns both amplifiers off (logic low) prior to turning one of them on. This control eliminates both outputs being active at the same time. If both amplifiers must be on, as in the simple switch illustrated in Figure 87, adding 100-Ω isolating resistors inside the loop at the outputs limits the current flow when both amplifiers are turned on. This solution offers a very high input impedance to both inputs, very low buffered output drive, and nearly perfect channel-to-channel isolation. The example of Figure 87 also includes a –0.23-V supply generator to allow true swing to ground on the output pins. This negative supply generator is optional if the outputs are more than 0.1 V above ground or intended to be AC-coupled. Testing with a single channel active and an off channel attached to the output showed no degradation in harmonic distortion; see Figure 17 and Figure 35. This approach can be expanded to more than two channels or to operate with gain in the channels. Adding more than two select channels in parallel should add 100-Ω feedback resistors to isolate the inverting input capacitance from the active output channel. 40 Submit Documentation Feedback Copyright © 2017–2018, Texas Instruments Incorporated Product Folder Links: OPA837 OPA2837 OPA837, OPA2837 www.ti.com SBOS673D – SEPTEMBER 2017 – REVISED DECEMBER 2018 +VS Typical Applications (continued) PD R6 100 k +VS Å U1 OPA837 + This line is actually a logic line + + VIN 1 R3 2 k VOUT Å9S +VS + +VS + Å9S 2X1 Active Mux LM7705 ± 230 mV VS1 5 PD +VS Å U2 OPA837 + R7 100 k + + Å9S VIN 2 Figure 87. 2:1 Active Multiplexer 8.2.2.1 Design Requirements To implement a 2:1 active mux, connect the outputs of two OPA837 devices together with separate input signals. If termination is required for the input signals, add this termination as a resistor to ground on the noninverting inputs. The inputs accept an input range from 0 V to 3.8 V by using a negative 0.23-V supply generator, such as the LM7705. 8.2.2.2 Detailed Design Procedure Aside from simply connecting the two outputs together as shown in Figure 87, there are several other considerations as well: • If the source impedance is not 0 Ω, consider adding a resistor in the feedback networks equal to that source impedance to reduce the output DC error resulting from bias currents • If the logic control can place both channels on at the same time, place 100-Ω resistors inside the feedback loop to limit supply currents when both outputs are active • If a matched gain is desired for the two inputs, configure the op amps for that gain instead of gain of 1 V/V • If the load is capacitive, add the required ROUT before the summing point on each op amp output Submit Documentation Feedback Copyright © 2017–2018, Texas Instruments Incorporated Product Folder Links: OPA837 OPA2837 41 OPA837, OPA2837 SBOS673D – SEPTEMBER 2017 – REVISED DECEMBER 2018 www.ti.com Typical Applications (continued) 8.2.3 1-Bit PGA Operation Using the internal inverting input switch that operates along with the power disable function can also allow a simple gain selection on a single input signal. Figure 88 shows an example gain select of either 1 V/V or 2 V/V from a single input to a single output. The logic disables both channels before turning one of them on to avoid high currents in both outputs to be active at the same time. If this approach is not possible, as in the simple switch shown in Figure 88, insert 100-Ω resistors inside the loop of each op amp output. A bipolar supply is shown in Figure 88, but any of the single-supply options are also possible. Any combination of gains can be implemented, but wide gain ranges show a larger change in signal bandwidth. This approach can be expanded to more than two gain settings. Testing with the circuit of Figure 88 showed no change in harmonic distortion; see Figure 18 and Figure 36. R4 2 k R5 2 k Gain of 2 or Gain of 1 Å9S 1-Bit PGA OPA837 VIN ++ VOUT PD R1 100 k +VS R3 1 k +VS Å9S This line is actually a logic line OPA837 Å9S +VS ++ R2 100 k +VS VS1 2.5 Å9S + + PD VS1 Å2.5 +VS Figure 88. 1-Bit PGA 8.2.3.1 Design Requirements Configure two OPA837 device outputs in different gains when driving the noninverting input with the same input signal. Select one the two channels using the disable control. Set one channel to a gain of 1 V/V and the second channel to a gain of 2 V/V using the recommended 2-kΩ values from Table 2. 8.2.3.2 Detailed Design Procedure The simple design of Figure 88 has several options and details to consider, which include: • For split-supply operation, the disable control line must operate to within 0.55 V of the negative supply to disable a channel. A logic level shift is required. • Any combination of gains can be implemented. However, the signal bandwidths may vary widely through the gain bandwidth product effect between the two channels if the gains are widely separated. If a more constant bandwidth between gains is desired, consider adding a fixed RC filter after the combined outputs at a lower cutoff frequency than the slowest gain setting. The TIDA-01565 Wired OR MUX and PGA Reference Design demonstrates the use of the OPAx837 in wired-OR multiplexer and programmable gain amplifier applications. 42 Submit Documentation Feedback Copyright © 2017–2018, Texas Instruments Incorporated Product Folder Links: OPA837 OPA2837 OPA837, OPA2837 www.ti.com SBOS673D – SEPTEMBER 2017 – REVISED DECEMBER 2018 9 Power Supply Recommendations The OPAx837 is intended to work in a nominal supply range of 3.0 V to 5 V. Supply-voltage tolerances are supported with the specified operating range of 2.7 V (–10% on a 3-V supply) and 5.4 V (+8% on a 5-V supply). Good power-supply bypassing is required. Minimize the distance (< 0.1 inch) from the power-supply pins to highfrequency, 0.1-µF decoupling capacitors. A larger capacitor (2.2 µF is typical) is used along with a highfrequency, 0.1-µF supply-decoupling capacitor at the device supply pins. For single-supply operation, only the positive supply has these capacitors. When a split supply is used, use these capacitors for each supply to ground. If necessary, place the larger capacitors further from the device and share these capacitors among several devices in the same area of the printed circuit board (PCB). Avoid narrow power and ground traces to minimize inductance between the pins and the decoupling capacitors. An optional supply decoupling capacitor across the two power supplies (for bipolar operation) reduces second harmonic distortion. The OPA837 has a positive supply current temperature coefficient; see Figure 57. This coefficient helps improve the input offset voltage drift. Supply current requirements in the system design must account for this effect using the maximum intended ambient and Figure 57 to size the supply required. The very low power dissipation for the OPA837 typically does not require any special thermal design considerations. For the extreme case of 125°C operating ambient, use the approximate maximum 200°C/W for the two packages, and a maximum internal power of 5.4-V supply × 0.8-mA 125°C supply current from Figure 57 gives a maximum internal power of 4.3 mW. This power only gives a 0.86°C rise from ambient to junction temperature, which is well below the maximum 150°C junction temperature. Load power adds to this value, but also increases the junction temperature only slightly over ambient temperature. 10 Layout 10.1 Layout Guidelines The OPA837EVM can be used as a reference when designing the circuit board. TI recommends following the EVM layout of the external components near to the amplifier, ground plane construction, and power routing as closely as possible. General guidelines are listed below: 1. Signal routing must be direct and as short as possible into and out of the op amp. 2. The feedback path must be short and direct avoiding vias if possible, especially with G = 1 V/V. 3. Ground or power planes must be removed from directly under the negative input and output pins of the amplifier. 4. TI recommends placing a series output resistor as close to the output pin as possible. See Figure 49 for recommended values for the expected capacitive load. These values are derived targeting a 30° phase margin to the output of the op amp. 5. A 2.2-µF power-supply decoupling capacitor must be placed within two inches of the device and can be shared with other op amps. For split supply, a capacitor is required for both supplies. 6. A 0.1-µF power-supply decoupling capacitor must be placed as close to the supply pins as possible, preferably within 0.1 inch. For split supply, a capacitor is required for both supplies. 7. The PD pin uses low logic swing levels. If the pin is not used, PD must be tied to the positive supply to enable the amplifier. If the pin is used, PD must be actively driven. A bypass capacitor is not necessary, but can be used for robustness in noisy environments. Submit Documentation Feedback Copyright © 2017–2018, Texas Instruments Incorporated Product Folder Links: OPA837 OPA2837 43 OPA837, OPA2837 SBOS673D – SEPTEMBER 2017 – REVISED DECEMBER 2018 www.ti.com 10.2 Layout Example Ground and power plane exist on inner layers Non-inverting input terminated in 50 Ÿ 1 6 Place bypass capacitors close to power pins 2 3 ± Place bypass capacitors close to power pins Ground and power plane removed from inner layers + Place output resistors close to output pins to minimize parasitic capacitance 5 Power control (disable) pin Must be driven 4 Place input resistor close to pin 4 to minimize stray capacitance Place feedback resistor on the bottom of PCB between pins 4 and 6 Remove GND and Power plane under pins 1 and 4 to minimize stray PCB capacitance Figure 89. EVM Layout Example 44 Submit Documentation Feedback Copyright © 2017–2018, Texas Instruments Incorporated Product Folder Links: OPA837 OPA2837 OPA837, OPA2837 www.ti.com SBOS673D – SEPTEMBER 2017 – REVISED DECEMBER 2018 11 Device and Documentation Support 11.1 Documentation Support 11.1.1 Related Documentation For related documentation see the following: • Texas Instruments, ADS8860 16-Bit, 1-MSPS, Serial Interface, microPower, Miniature, Single-Ended Input, SAR Analog-to-Digital Converter data sheet • Texas Instruments, LM7705 Low-Noise Negative Bias Generator data sheet • Texas Instruments, OPA838 1-mA, 300-MHz Gain Bandwidth, Voltage-Feedback Op Amp data sheet • Texas Instruments, REF50xx Low-Noise, Very Low Drift, Precision Voltage Reference data sheet • Texas Instruments, OPA837DBV, OPA836DBV EVM user's guide • Texas Instruments, Single-Supply Op Amp Design Techniques application report • Texas Instruments, Noise Analysis for High-Speed Op Amps application report • Texas Instruments, Design Methodology for MFB Filters in ADC Interface Applications application note • Texas Instruments, Component Pre-Distortion for Sallen Key Filters application note • Texas Instruments, TIDA-01565 Wired OR MUX and PGA Reference Design design guide • Texas Instruments, TINA model and simulation tool 11.2 Related Links The table below lists quick access links. Categories include technical documents, support and community resources, tools and software, and quick access to order now. Table 4. Related Links PARTS PRODUCT FOLDER ORDER NOW TECHNICAL DOCUMENTS TOOLS & SOFTWARE SUPPORT & COMMUNITY OPA837 Click here Click here Click here Click here Click here OPA2837 Click here Click here Click here Click here Click here 11.3 Receiving Notification of Documentation Updates To receive notification of documentation updates, navigate to the device product folder on ti.com. In the upper right corner, click on Alert me to register and receive a weekly digest of any product information that has changed. For change details, review the revision history included in any revised document. 11.4 Community Resources The following links connect to TI community resources. Linked contents are provided "AS IS" by the respective contributors. They do not constitute TI specifications and do not necessarily reflect TI's views; see TI's Terms of Use. TI E2E™ Online Community TI's Engineer-to-Engineer (E2E) Community. Created to foster collaboration among engineers. At e2e.ti.com, you can ask questions, share knowledge, explore ideas and help solve problems with fellow engineers. Design Support TI's Design Support Quickly find helpful E2E forums along with design support tools and contact information for technical support. 11.5 Trademarks E2E is a trademark of Texas Instruments. All other trademarks are the property of their respective owners. 11.6 Electrostatic Discharge Caution These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam during storage or handling to prevent electrostatic damage to the MOS gates. Submit Documentation Feedback Copyright © 2017–2018, Texas Instruments Incorporated Product Folder Links: OPA837 OPA2837 45 OPA837, OPA2837 SBOS673D – SEPTEMBER 2017 – REVISED DECEMBER 2018 www.ti.com 11.7 Glossary SLYZ022 — TI Glossary. This glossary lists and explains terms, acronyms, and definitions. 12 Mechanical, Packaging, and Orderable Information The following pages include mechanical, packaging, and orderable information. This information is the most current data available for the designated devices. This data is subject to change without notice and revision of this document. For browser-based versions of this data sheet, refer to the left-hand navigation. 46 Submit Documentation Feedback Copyright © 2017–2018, Texas Instruments Incorporated Product Folder Links: OPA837 OPA2837 PACKAGE OPTION ADDENDUM www.ti.com 10-Dec-2020 PACKAGING INFORMATION Orderable Device Status (1) Package Type Package Pins Package Drawing Qty Eco Plan (2) Lead finish/ Ball material MSL Peak Temp Op Temp (°C) Device Marking (3) (4/5) (6) OPA2837IDGKR ACTIVE VSSOP DGK 8 2500 RoHS & Green NIPDAUAG Level-2-260C-1 YEAR -40 to 125 2837 OPA2837IDGKT ACTIVE VSSOP DGK 8 250 RoHS & Green NIPDAUAG Level-2-260C-1 YEAR -40 to 125 2837 OPA2837IRUNR ACTIVE QFN RUN 10 3000 RoHS & Green NIPDAU Level-2-260C-1 YEAR -40 to 125 2837 OPA2837IRUNT ACTIVE QFN RUN 10 250 RoHS & Green NIPDAU Level-2-260C-1 YEAR -40 to 125 2837 OPA837IDBVR ACTIVE SOT-23 DBV 6 3000 RoHS & Green NIPDAU Level-2-260C-1 YEAR -40 to 125 19FF OPA837IDBVT ACTIVE SOT-23 DBV 6 250 RoHS & Green NIPDAU Level-2-260C-1 YEAR -40 to 125 19FF OPA837IDCKR ACTIVE SC70 DCK 5 3000 RoHS & Green NIPDAU Level-2-260C-1 YEAR -40 to 125 16K OPA837IDCKT ACTIVE SC70 DCK 5 250 RoHS & Green NIPDAU Level-2-260C-1 YEAR -40 to 125 16K (1) The marketing status values are defined as follows: ACTIVE: Product device recommended for new designs. LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect. NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design. PREVIEW: Device has been announced but is not in production. Samples may or may not be available. OBSOLETE: TI has discontinued the production of the device. (2) RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may reference these types of products as "Pb-Free". RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption. Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of