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THS4022CD

THS4022CD

  • 厂商:

    BURR-BROWN(德州仪器)

  • 封装:

    SOIC-8

  • 描述:

    IC OPAMP VFB 2 CIRCUIT 8SOIC

  • 数据手册
  • 价格&库存
THS4022CD 数据手册
THS4021 THS4022 www.ti.com SLOS265C – SEPTEMBER 1999 – REVISED JULY 2007 350-MHz LOW-NOISE HIGH-SPEED AMPLIFIERS FEATURES 1 THS4021 D and DGN Package (Top View) • Ultralow 1.5-nV/√Hz Voltage Noise • High Speed: – 350-MHz Bandwidth (G = 10, –3 dB) – 470-V/μs Slew Rate – 40-ns Settling Time (0.1%) • Stable at a Gain of 10 (–9) or Greater • High Output Drive, IO = 100 mA (typ) • Excellent Video Performance: – 17-MHz Bandwidth (0.1 dB, G = 10) – 0.02% Differential Gain – 0.08° Differential Phase • Very Low Distortion: – THD = –68 dBc (f = 1 MHz, RL = 150 Ω) • Wide Range of Power Supplies: – VCC = ±5 V to ±15 V • Available in Standard SOIC or MSOP PowerPAD™ Package • Evaluation Module Available 23 THS4022 D and DGN Package (Top View) NULL 1 8 NULL 1OUT 1 8 VCC+ IN– 2 7 VCC+ 1IN– 2 7 2OUT IN+ VCC– 3 6 6 5 1IN+ –VCC 3 4 OUT NC 4 5 2IN– 2IN+ NC - No internal connection Cross Section View Showing PowerPAD Option (DGN) VOLTAGE AND CURRENT NOISE vs FREQUENCY The THS4021 and THS4022 are ultralow voltage noise, high-speed voltage feedback amplifiers that are ideal for applications requiring low voltage noise, including communication and imaging. The single-amplifier THS4021 and the dual-amplifier THS4022 offer very good ac performance with 350-MHz bandwidth, 470-V/μs slew rate, and 40-ns settling time (0.1%). The THS4021 and THS4022 are stable at gains of 10 (–9) or greater. These amplifiers have a high drive capability of 100 mA and draw only 7.8-mA supply current per channel. With total harmonic distortion (THD) of –68 dBc at f = 1 MHz, the THS4021 and THS4022 are ideally suited for applications requiring low distortion. In − Current Noise − pA//Hz DESCRIPTION Vn − Voltage Noise − nV//Hz 100 VCC = ± 15 V and ± 5 V TA = 25°C 10 Vn In 1 10 100 1k 10k 100k f − Frequency − Hz G001 Figure 1. RELATED DEVICES DEVICE DESCRIPTION THS4011/4012 290-MHz Low-Distortion High-Speed Amplifiers THS4031/4032 100-MHz Low-Noise High-Speed Amplifiers THS4061/4062 180-MHz High-Speed Amplifiers 1 2 3 Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet. PowerPAD is a trademark of Texas Instruments. All other trademarks are the property of their respective owners. PRODUCTION DATA information is current as of publication date. Products conform to specifications per the terms of the Texas Instruments standard warranty. Production processing does not necessarily include testing of all parameters. Copyright © 1999–2007, Texas Instruments Incorporated THS4021 THS4022 www.ti.com SLOS265C – SEPTEMBER 1999 – REVISED JULY 2007 CAUTION: The THS4021 and THS4022 provide ESD protection circuitry. However, permanent damage can still occur if this device is subjected to high-energy electrostatic discharges. Proper ESD precautions are recommended to avoid any performance degradation or loss of functionality. AVAILABLE OPTIONS (1) PACKAGED DEVICES TA 0°C to 70°C –40°C to 85°C (1) (2) NUMBER OF CHANNELS PLASTIC SMALL OUTLINE (2) (D) PLASTIC MSOP (2) (DGN) MSOP SYMBOL EVALUATION MODULE 1 THS4021CD THS4021CDGN ACK THS4021EVM 2 THS4022CD THS4022CDGN ACA THS4022EVM 1 THS4021ID THS4021CIDGN ACL – 2 THS4022ID THS4022CIDGN ACB – For the most current package and ordering information, see the Package Option Addendum at the end of this document, or see the TI Web site at www.ti.com. The D and DGN packages are available taped and reeled. Add an R suffix to the device type (for example, THS4021CDGN). FUNCTIONAL BLOCK DIAGRAMS Null 2 1 8 IN– 6 OUT 3 IN+ S0273-01 Figure 2. THS4021—Single Channel VCC 1IN– 1OUT 1IN+ 2IN– 2OUT 2IN+ –VCC S0274-01 Figure 3. THS4022—Dual Channel 2 Submit Documentation Feedback Copyright © 1999–2007, Texas Instruments Incorporated Product Folder Link(s): THS4021 THS4022 THS4021 THS4022 www.ti.com SLOS265C – SEPTEMBER 1999 – REVISED JULY 2007 ABSOLUTE MAXIMUM RATINGS (1) over operating free-air temperature range (unless otherwise noted) VALUE UNIT VCC Supply voltage ±16.5 V VI Input voltage ±VCC V IO Output current 150 mA VIO Differential input voltage ±4 V Continuous total power dissipation TJ TA Tstg See Dissipation Ratings table Maximum junction temperature 0 to 70 I-suffix °C –40 to 85 Storage temperature Lead temperature 1,6 mm (1/16 inch) from case for 10 seconds (1) °C 150 Operating free-air temperature: C-suffix –65 to 150 °C 300 °C Stresses beyond those listed under "absolute maximum ratings" may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated under "recommended operating conditions" is not implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability. DISSIPATION RATINGS θJA (°C/W) θJC (°C/W) TA = 25°C POWER RATING D (1) 167 38.3 740 mW DGN (2) 58.4 4.7 2.14 W PACKAGE (1) (2) This data was taken using the JEDEC standard low-K test PCB. For the JEDEC proposed high-K test PCB, the θJA is 95°C/W with a power rating at TA = 25°C of 1.32 W. This data was taken using 2-oz. (0.071-mm thick) trace and copper pad on a 3-in. × 3-in. (7.62-cm × 7.62-cm) PCB, with the device soldered directly to the board. For further information, see the Application Information section of this data sheet. RECOMMENDED OPERATING CONDITIONS MIN Dual supply VCC+ and VCC– Supply voltage TA Operating free-air temperature NOM MAX ±4.5 ±16 Single supply 9 32 C-suffix 0 70 –40 85 I-suffix UNIT V °C ELECTRICAL CHARACTERISTICS at TA = 25°C, VCC = ±15 V, RL = 150 Ω (unless otherwise noted) PARAMETER TEST CONDITIONS MIN TYP MAX UNIT Dynamic Performance VCC = ±15 V Small-signal bandwidth (–3 dB) Bandwidth for 0 1-dB flatness Full power bandwidth (1) (1) (2) VCC = ±15 V VCC = ±5 V BW SR VCC = ±5 V Slew rate (2) VCC = ±15 V VCC = ±5 V Gain = 10 Gain = 20 Gain = 10 VO(pp) = 20 V, VCC = ±15 V VCC = ±5 V, 5-V step 280 80 70 17 MHz 17 3.7 VO(pp) = 5 V, VCC = ±5 V VCC = ±15 V, 10-V step 350 11.8 Gain = 10 470 370 V/μs Full-power bandwidth = slew rate / 2π VO(Peak). Slew rate is measured from an output level range of 25% to 75%. Copyright © 1999–2007, Texas Instruments Incorporated Product Folder Link(s): THS4021 THS4022 Submit Documentation Feedback 3 THS4021 THS4022 www.ti.com SLOS265C – SEPTEMBER 1999 – REVISED JULY 2007 ELECTRICAL CHARACTERISTICS (continued) at TA = 25°C, VCC = ±15 V, RL = 150 Ω (unless otherwise noted) PARAMETER Settling time to 0.1% ts Settling time to 0.01% TEST CONDITIONS VCC = ±15 V, 5-V step VCC = ±5 V, 2-V step VCC = ±15 V, 5-V step VCC = ±5 V, 2-V step MIN TYP MAX UNIT 40 Gain = –10 50 ns 145 Gain = –10 150 Noise/Distortion Performance THD VO(pp) = 2 V, f = 1 MHz, gain = 2, VCC = ±15 V RL = 150 Ω –68 RL = 1 kΩ –77 VO(pp) = 2 V, f = 1 MHz, gain = 2, VCC = ±5 V RL = 150 Ω –69 RL = 1 kΩ –78 Total harmonic distortion dBc Vn Input voltage noise VCC = ±5 V or ±15 V, f > 10 kHz 1.5 nV/√Hz In Input current noise VCC = ±5 V or ±15 V, f > 10 kHz 2 pA/√Hz XT Differential gain error Gain = 2, NTSC, 40 IRE modulation, ±100 IRE ramp VCC = ±15 0.02% VCC = ±5 V 0.02% Differential phase error Gain = 2, NTSC, 40 IRE modulation, ±100 IRE ramp VCC = ±15 0.08 VCC = ±5 V 0.06 Channel-to-channel crosstalk (THS4022 only) VCC = ±5 V or ±15 V, f = 1 MHz ° –60 dB DC Performance VCC = ±15 V, VO = ±10 V, RL = 1 kΩ TA = 25°C 40 TA = full range 35 VCC = ±5 V, VO = ±2.5 V, RL = 250 Ω TA = 25°C 20 TA = full range 15 Open-loop gain VOS TA = 25°C Input offset voltage Input bias current IOS Input offset current Offset current drift 0.5 2 3 TA = full range VCC = ±5 V or ±15 V V/mV 35 TA = full range Offset voltage drift IIB 60 μV/°C 15 TA = 25°C 3 TA = full range 6 6 TA = 25°C 30 TA = full range 250 400 TA = full range 0.3 mV μA nA nA/°C Input Characteristics VCC = ±15 V ±13.8 ±14.3 VCC = ±5 V ±3.8 ±4.3 74 95 dB 1 MΩ 1.5 pF VICR Common-mode input voltage range CMRR Common-mode rejection ratio VCC = ±15 V, VICR = ±12 V, TA = full range ri Input resistance Ci Input capacitance V Output Characteristics VO Output voltage swing VCC = ±15 V RL = 250 Ω ±12 ±12.5 VCC = ±5 V RL = 150 Ω ±3 ±3.3 ±13 ±13.5 ±3.4 ±3.8 80 100 50 75 VCC = ±15 V VCC = ±5 V VCC = ±15 V IO Output current ISC Short-circuit current (3) VCC = ±15 V RO Output resistance (3) Open loop (3) 4 VCC = ±5 V RL = 1 kΩ RL = 20 Ω V mA 150 mA 13 Ω Observe power dissipation ratings to keep the junction temperature below the absolute maximum rating when the output is heavily loaded or shorted. See the Absolute Maximum Ratings table of this data sheet for more information. Submit Documentation Feedback Copyright © 1999–2007, Texas Instruments Incorporated Product Folder Link(s): THS4021 THS4022 THS4021 THS4022 www.ti.com SLOS265C – SEPTEMBER 1999 – REVISED JULY 2007 ELECTRICAL CHARACTERISTICS (continued) at TA = 25°C, VCC = ±15 V, RL = 150 Ω (unless otherwise noted) PARAMETER TEST CONDITIONS MIN TYP MAX UNIT Power Supply Dual supply Supply voltage operating range VCC 33 V 10 11 TA = 25°C VCC = ±5 V Power-supply rejection ratio 7.8 TA = full range Supply current (per amplifier) PSRR ±16.5 9 TA = 25°C VCC = ±15 V ICC ±4.5 Single supply 6.7 mA 9 TA = full range 10.5 VCC = ±5 V or ±15 V, TA = full range 80 95 dB TYPICAL CHARACTERISTICS CROSSTALK vs FREQUENCY OPEN LOOP GAIN AND PHASE RESPONSE vs FREQUENCY 10 VCC = ± 5 V & ±15 V Gain −20 −30 −40 −50 −60 −70 −80 1M 10M 0 100 100M 80 −30 −60 60 Phase 40 −90 20 −120 0 −150 −20 1k 1G 10k 1M 100k f − Frequency − Hz Figure 5. TOTAL HARMONIC DISTORTION vs FREQUENCY DISTORTION vs OUTPUT VOLTAGE −10 RL = 1 kΩ −80 −10 VCC = ± 15 V RL = 1 kΩ G = 10 f = 1 MHz −30 RL = 150 Ω DISTORTION vs OUTPUT VOLTAGE VCC = ± 15 V RL = 150 Ω G = 10 f = 1 MHz −30 Distortion − dBc VCC = ± 15 V Gain = 10 VO(PP) = 2 V Distortion − dBc THD − Total Harmonic Distortion − dBc −40 −70 −180 1G G003 Figure 4. −60 100M f − Frequency − Hz G002 −50 10M Phase − 5 Crosstalk − dB −10 30 120 VCC = ± 15 V Gain = 10 RF = 220 Ω RL = 150 Ω Open Loop Gain − dB 0 −50 2nd Harmonic −70 −50 2nd Harmonic −70 3rd Harmonic −90 −90 −90 3rd Harmonic −100 100k −110 1M −110 0 10M 5 10 15 VO − Output Voltage − V f − Frequency − Hz G004 Figure 6. 20 G005 Figure 7. Copyright © 1999–2007, Texas Instruments Incorporated Product Folder Link(s): THS4021 THS4022 0 5 10 15 VO − Output Voltage − V 20 G006 Figure 8. Submit Documentation Feedback 5 THS4021 THS4022 www.ti.com SLOS265C – SEPTEMBER 1999 – REVISED JULY 2007 TYPICAL CHARACTERISTICS (continued) DISTORTION vs FREQUENCY −70 2nd Harmonic −80 3rd Harmonic −90 −100 100k 1M −70 −80 −90 1M f − Frequency − Hz G008 DISTORTION vs FREQUENCY OUTPUT AMPLITUDE vs FREQUENCY OUTPUT AMPLITUDE vs FREQUENCY −70 3rd Harmonic RF = 220 Ω 1M 20 RF = 150 Ω 15 VCC = ± 15 V Gain = 10 RL = 150 Ω VO(PP) = 400 mV 10 10k 10M f − Frequency − Hz 1M 100k 10M 100M Figure 13. OUTPUT AMPLITUDE vs FREQUENCY OUTPUT AMPLITUDE vs FREQUENCY 30 VCC = ± 5 V Gain = 10 RL = 150 Ω VO(PP) = 400 mV 100k Output Amplitude − dB RF = 220 Ω 20 VCC = ±15 V RF = 1 kΩ Gain = 20 RL = 150 Ω VO(PP) = 400 mV 1G f − Frequency − Hz 25 20 15 10 100k 1G G012 0.6 RF = 220 Ω VCC = ±5 V Gain = 20 RL = 150 Ω VO(PP) = 400 mV 1M RF = 1 kΩ 0.4 0.2 0.0 VCC = ± 5 V Gain = 10 RF = 220 Ω RL = 150 Ω −0.2 −0.4 −0.6 10M G013 Submit Documentation Feedback 100M 1-V STEP RESPONSE 100M 0 1G 50 100 150 200 250 300 350 400 t − Time − ns f − Frequency − Hz Figure 15. 10M Figure 14. RF = 6.2 kΩ 25 1M f − Frequency − Hz 0.8 RF = 6.2 kΩ 100M 15 10 10k 1G 30 10M RF = 150 Ω G011 Figure 12. 1M RF = 220 Ω 20 f − Frequency − Hz G010 Output Amplitude − dB Output Amplitude − dB Output Amplitude − dB Distortion − dBc 25 2nd Harmonic −90 6 G009 Figure 11. −60 10 100k 10M Figure 10. 25 15 1M f − Frequency − Hz Figure 9. VCC = ± 5 V RL = 150 Ω G = 10 VO(PP) = 2 V −100 100k 3rd Harmonic f − Frequency − Hz −40 −80 −80 −100 100k 10M G007 −50 2nd Harmonic −70 −90 3rd Harmonic −100 100k 10M VCC = ± 15 V RL = 150 Ω G = 10 VO(PP) = 2 V −60 2nd Harmonic Distortion − dBc −60 −50 VCC = ± 5 V RL = 1 kΩ G = 10 VO(PP) = 2 V VO − Output Voltage − V Distortion − dBc −60 DISTORTION vs FREQUENCY −50 VCC = ± 15 V RL = 1 kΩ G = 10 VO(PP) = 2 V Distortion − dBc −50 DISTORTION vs FREQUENCY G015 G014 Figure 16. Figure 17. Copyright © 1999–2007, Texas Instruments Incorporated Product Folder Link(s): THS4021 THS4022 THS4021 THS4022 www.ti.com SLOS265C – SEPTEMBER 1999 – REVISED JULY 2007 TYPICAL CHARACTERISTICS (continued) 5-V STEP RESPONSE 1-V STEP RESPONSE 6 0.6 1 0 −1 VCC = ± 5 V Gain = −10 RF = 220 Ω RL = 150 Ω −2 50 0.4 0.2 0.0 VCC = ± 15 V Gain = 10 RF = 220 Ω RL = 150 Ω −0.2 −0.4 0 −2 VCC = ± 15 V Gain = 10 RF = 220 Ω RL = 150 −6 0 100 150 200 250 300 350 400 2 −4 −0.6 −3 0 4 VO − Output Voltage − V VO − Output Voltage − V 2 VO − Output Voltage − V 10-V STEP RESPONSE 0.8 3 50 100 150 200 250 300 350 400 0 400 Figure 20. INPUT OFFSET VOLTAGE vs FREE-AIR TEMPERATURE INPUT BIAS CURRENT vs FREE-AIR TEMPERATURE OUTPUT VOLTAGE vs SUPPLY VOLTAGE 3.30 14 −0.15 −0.20 VCC = ± 15 V −0.25 −20 0 20 40 60 80 TA − Free-Air Temperature − °C 12 3.20 3.15 3.10 3.05 3.00 −40 100 TA = 25°C 3.25 |VO| − Output Voltage − |V| IIB − Input Bias Current − µA VCC = ± 5 V −0.10 RL = 1 kΩ 10 8 RL = 150 Ω 6 4 2 −20 0 20 40 60 80 TA − Free-Air Temperature − °C G019 100 5 7 9 11 13 +VCC − Supply Voltage − V G020 Figure 21. Figure 22. Figure 23. COMMON-MODE INPUT VOLTAGE vs SUPPLY VOLTAGE OUTPUT VOLTAGE vs FREE-AIR TEMPERATURE SUPPLY CURRENT vs SUPPLY VOLTAGE 15 500 G018 Figure 19. VCC = ± 5 V & ±15 V 15 G021 11 14 TA = 25°C VCC = ± 15 V RL = 250 Ω 12 |VO| − Output Voltage − |V| 13 11 9 7 5 10 VCC = ± 15 V RL = 1 kΩ 8 6 VCC = ± 5 V RL = 1 kΩ 4 VCC = ± 5 V RL = 150 Ω 2 3 5 7 9 11 13 +VCC − Supply Voltage − V 0 −40 15 G022 10 TA=85°C 9 8 TA=25°C 7 6 TA=−40°C 5 −20 0 20 40 60 80 TA − Free-Air Temperature − °C Figure 24. ICC − Supply Current − mA VIO − Input Offset Voltage − mV 300 Figure 18. −0.05 VICR − Common−Mode Input Voltage − +V 200 t − Time − ns G017 G016 −0.30 −40 100 t − Time − ns t − Time − ns 100 G023 Figure 25. Copyright © 1999–2007, Texas Instruments Incorporated Product Folder Link(s): THS4021 THS4022 5 7 9 11 13 +VCC − Supply Voltage − V 15 G024 Figure 26. Submit Documentation Feedback 7 THS4021 THS4022 www.ti.com SLOS265C – SEPTEMBER 1999 – REVISED JULY 2007 TYPICAL CHARACTERISTICS (continued) PSRR − Power Supply Rejection Ratio − dB Vn − Voltage Noise − nV//Hz In − Current Noise − pA//Hz 100 VCC = ± 15 V and ± 5 V TA = 25°C 10 Vn In 1 10 100 1k POWER SUPPLY REJECTION RATIO vs FREQUENCY 10k 100k 0 VCC = ±15 V & ±5 V −10 −20 −VCC −30 −40 +VCC −50 −60 −70 −80 100k 1M 10M 1G f − Frequency − Hz f − Frequency − Hz G026 G025 Figure 27. 8 100M Submit Documentation Feedback COMMON MODE REJECTION RATIO vs FREQUENCY CMRR − Common-Mode Rejection Ratio − dB VOLTAGE AND CURRENT NOISE vs FREQUENCY Figure 28. 0 −10 VCC = ±15 V or ±5 V RF = 20 kΩ VI(PP) = 2 V −20 −30 −40 −50 −60 100k 1M 10M 100M 1G f − Frequency − Hz G027 Figure 29. Copyright © 1999–2007, Texas Instruments Incorporated Product Folder Link(s): THS4021 THS4022 THS4021 THS4022 www.ti.com SLOS265C – SEPTEMBER 1999 – REVISED JULY 2007 APPLICATION INFORMATION Theory of Operation The THS402x is a high-speed operational amplifier configured in a voltage feedback architecture. It is built using a 30-V, dielectrically isolated, complementary bipolar process with NPN and PNP transistors possessing fT of several GHz. This results in an exceptionally high-performance amplifier that has a wide bandwidth, high slew rate, fast settling time, and low distortion. A simplified schematic is shown in Figure 30. (7) VCC+ (6) OUT IN– (2) IN+ (3) (4) VCC– NULL (1) NULL (8) S0276-01 Figure 30. THS4021 Simplified Schematic Noise Calculations and Noise Figure Noise can cause errors on very small signals. This is especially true when amplifying small signals, where signal-to-noise ratio (SNR) is very important. The noise model for the THS402x is shown in Figure 31. This model includes all of the noise sources as follows: • en = Amplifier internal voltage noise (nV/√Hz) • IN+ = Noninverting current noise (pA/√Hz) • IN– = Inverting current noise (pA/√Hz) • eRx = Thermal voltage noise associated with each resistor (eRx = 4 kTRx) Copyright © 1999–2007, Texas Instruments Incorporated Product Folder Link(s): THS4021 THS4022 Submit Documentation Feedback 9 THS4021 THS4022 www.ti.com SLOS265C – SEPTEMBER 1999 – REVISED JULY 2007 eRs RS en Noiseless eni + _ eno IN+ eRf RF eRg IN– RG S0277-01 Figure 31. Noise Model The total equivalent input noise density (eni) is calculated by using the following equation: e ni + Ǹ ǒenǓ ) ǒIN+ 2 R Ǔ S 2 ǒ ) IN− ǒR F ø R ǓǓ G 2 ǒ ) 4 kTR s ) 4 kT R ø R F G Ǔ where: k = Boltzmann’s constant = 1.380658 × 10–23 T = Temperature in degrees Kelvin (273 + °C) RF || RG = Parallel resistance of RF and RG To get the equivalent output noise of the amplifier, just multiply the equivalent input noise density (eni) by the overall amplifier gain (AV). e no + e ni A V ǒ + e ni 1 ) Ǔ RF (noninverting case) RG As the previous equations show, to keep noise at a minimum, small value resistors should be used. As the closed-loop gain is increased (by reducing RG), the input noise is reduced considerably because of the parallel resistance term. This leads to the general conclusion that the most dominant noise sources are the source resistor (RS) and the internal amplifier noise voltage (en). Because noise is summed in a root-mean-squares method, noise sources smaller than 25% of the largest noise source can be effectively ignored. This can greatly simplify the formula and make noise calculations much easier to calculate. For more information on noise analysis, see the Noise Analysis in Operational Amplifier Circuits application report (SLVA043). This brings up another noise measurement usually preferred in RF applications, the noise figure (NF). Noise figure is a measure of noise degradation caused by the amplifier. The value of the source resistance must be defined and is typically 50 Ω in RF applications. NF + 10 Submit Documentation Feedback ȱ e 2ȳ 10logȧ ni ȧ ȧ 2ȧ ȲǒeRsǓ ȴ Copyright © 1999–2007, Texas Instruments Incorporated Product Folder Link(s): THS4021 THS4022 THS4021 THS4022 www.ti.com SLOS265C – SEPTEMBER 1999 – REVISED JULY 2007 Because the dominant noise components are generally the source resistance and the internal amplifier noise voltage, we can approximate noise figure as: ǒ 2 ȱ ǒen Ǔ ) ǒIN+ ȧ NF + 10logȧ1 ) 4 kTR ȧ S ȧ Ȳ R Ǔ 2 ȳ Ǔ S ȧ ȧ ȧ ȧ ȴ Figure 32 shows the noise figure graph for the THS402x. NOISE FIGURE vs SOURCE RESISTANCE 16 14 f = 10 kHz TA = 25°C Noise Figure − dB 12 10 8 6 4 2 0 10 100 1k 10k Source Resistance − Ω G028 Figure 32. Noise Figure vs Source Resistance Copyright © 1999–2007, Texas Instruments Incorporated Product Folder Link(s): THS4021 THS4022 Submit Documentation Feedback 11 THS4021 THS4022 www.ti.com SLOS265C – SEPTEMBER 1999 – REVISED JULY 2007 Driving a Capacitive Load Driving capacitive loads with high-performance amplifiers is not a problem as long as certain precautions are taken. The first is to realize that the THS402x has been internally compensated to maximize its bandwidth and slew-rate performance. When the amplifier is compensated in this manner, capacitive loading directly on the output decreases the device phase margin, leading to high-frequency ringing or oscillations. Therefore, for capacitive loads of greater than 10 pF, it is recommended that a resistor be placed in series with the output of the amplifier, as shown in Figure 33. A minimum value of 20 Ω should work well for most applications. For example, in 75-Ω transmission systems, setting the series resistor value to 75 Ω both isolates any capacitance loading and provides the proper line impedance matching at the source end. 1 kW 50 W Input _ 20 W Output THS402x CLOAD + S0278-01 Figure 33. Driving a Capacitive Load Offset Nulling The THS402x has very low input offset voltage for a high-speed amplifier. However, if additional correction is required, an offset nulling function has been provided on the THS4021. The input offset can be adjusted by placing a potentiometer between terminals 1 and 8 of the device and tying the wiper to the negative supply. This is shown in Figure 34. VCC+ 0.1 mF + THS402x _ 10 kW 0.1 mF VCC– S0279-01 Figure 34. Offset Nulling Schematic 12 Submit Documentation Feedback Copyright © 1999–2007, Texas Instruments Incorporated Product Folder Link(s): THS4021 THS4022 THS4021 THS4022 www.ti.com SLOS265C – SEPTEMBER 1999 – REVISED JULY 2007 Offset Voltage The output offset voltage (VOO) is the sum of the input offset voltage (VIO) and both input bias currents (IIB) times the corresponding gains. The schematic and formula of Figure 35 can be used to calculate the output offset voltage. RG RF IIB– – VOS + RS + – VIO IIB+ æ R ö VOS = (± VIO ± IIB + ´ RS )ç 1 + F ÷ ± IIB - ´ RF ç RG ÷ø è S0280-01 Figure 35. Output Offset Voltage Model General Configurations When receiving low-level signals, limiting the bandwidth of the incoming signals into the system is often required. The simplest way to accomplish this is to place an RC filter at the noninverting terminal of the amplifier (see Figure 36). RG RF – VO VI + R1 C1 f-3dB = 1 2pR1C1 VO æ R öæ 1 ö = ç 1 + F ÷÷ ç ÷ VI çè RG ø è 1 + sR1C1 ø S0281-01 Figure 36. Single-Pole Low-Pass Filter Copyright © 1999–2007, Texas Instruments Incorporated Product Folder Link(s): THS4021 THS4022 Submit Documentation Feedback 13 THS4021 THS4022 www.ti.com SLOS265C – SEPTEMBER 1999 – REVISED JULY 2007 Circuit Layout Considerations To achieve the levels of high-frequency performance of the THS402x, follow proper printed-circuit board high-frequency design techniques. A general set of guidelines is given as follows. In addition, a THS402x evaluation board is available to use as a guide for layout or for evaluating the device performance. • Ground planes—It is highly recommended that a ground plane be used on the board to provide all components with a low-inducance ground connection. However, in the areas of the amplifier inputs and output, the ground plane can be removed to minimize the stray capacitance. • Proper power-supply decoupling—Use a 6.8-μF tantalum capacitor in parallel with a 0.1-μF ceramic capacitor on each supply terminal. It may be possible to share the tantalum among several amplifiers depending on the application, but a 0.1-μF ceramic capacitor should always be used on the supply terminal of every amplifier. In addition, the 0.1-μF capacitor should be placed as close as possible to the supply terminal. As this distance increases, the inductance in the connecting trace makes the capacitor less effective. The designer should strive for distances of less than 0.1 inch (2.54 mm) between the device power terminals and the ceramic capacitors. • Sockets—Sockets are not recommended for high-speed operational amplifiers. The additional lead inductance in the socket pins often produces stability problems. Surface-mount packages soldered directly to the PCB is the best implementation. • Short trace runs/compact part placements—Optimum high-frequency performance is achieved when stray series inductance has been minimized. To realize this, the circuit layout should be made as compact as possible, thereby minimizing the length of all trace runs. Particular attention should be paid to the inverting input of the amplifier. Its length should be kept as short as possible. This helps to minimize stray capacitance at the input of the amplifier. • Surface-mount passive components—Using surface-mount passive components is recommended for high-frequency amplifier circuits for several reasons. First, because of the extremely low lead inductance of surface-mount components, the problem with stray series inductance is greatly reduced. Second, the small size of surface-mount components naturally leads to a more compact layout, thereby minimizing both stray inductance and capacitance. If leaded components are used, it is recommended that the lead lengths be kept as short as possible. 14 Submit Documentation Feedback Copyright © 1999–2007, Texas Instruments Incorporated Product Folder Link(s): THS4021 THS4022 THS4021 THS4022 www.ti.com SLOS265C – SEPTEMBER 1999 – REVISED JULY 2007 General Thermal Pad Design Considerations The THS402x is available packaged in a thermally-enhanced DGN package, which is a member of the PowerPAD family of packages. This package is constructed using a downset leadframe upon which the die is mounted [see Figure 37(a) and Figure 37(b)]. This arrangement results in the lead frame being exposed as a thermal pad on the underside of the package [see Figure 37(c)]. Because this thermal pad has direct thermal contact with the die, excellent thermal performance can be achieved by providing a good thermal path away from the thermal pad. The PowerPAD package allows for both assembly and thermal management in one manufacturing operation. During the surface-mount solder operation (when the leads are being soldered), the thermal pad can also be soldered to a copper area underneath the package. Through the use of thermal paths within this copper area, heat can be conducted away from the package into either a ground plane or other heat dissipating device. The PowerPAD package represents a design breakthrough, combining the small area and ease of the surface mount assembly method to eliminate the previously difficult mechanical methods of heatsinking. DIE Side View (a) Thermal Pad DIE End View (b) Bottom View (c) M0031-01 NOTE: The thermal pad is electrically isolated from all terminals in the package. Figure 37. Views of Thermally Enhanced DGN Package Although there are many ways to heatsink this device properly, the following steps illustrate the recommended approach. Thermal pad area = 68 mils ´ 70 mils (1.73 mm ´1.78 mm) with 5 vias. Via diameter = 13 mils (0.33 mm). M0032-02 Figure 38. Thermal Pad PCB Etch and Via Pattern 1. Prepare the PCB with a top side etch pattern as shown in Figure 38. There should be etch for the leads as well as etch for the thermal pad. 2. Place five holes in the area of the thermal pad. These holes should be 13 mils (0.33 mm) in diameter. Keep them small so that solder wicking through the holes is not a problem during reflow. 3. Additional vias may be placed anywhere along the thermal plane outside of the thermal pad area. These vias help dissipate the heat generated by the THS402xDGN IC. These additional vias may be larger than the 13-mil (0.33-mm) diameter vias directly under the thermal pad. They can be larger because they are not in the thermal pad area to be soldered, so wicking is not a problem. 4. Connect all holes to the internal ground plane. 5. When connecting these holes to the ground plane, do not use the typical web or spoke via connection methodology. Web connections have a high thermal resistance connection that is useful for slowing the heat transfer during soldering operations. This makes the soldering of vias that have plane connections easier. In this application, however, low thermal resistance is desired for the most efficient heat transfer. Therefore, the holes under the THS402xDGN package should connect to the internal ground plane with a complete connection around the entire circumference of the plated-through hole. 6. The top-side solder mask should leave the terminals of the package and the thermal pad area with its five holes exposed. The bottom-side solder mask should cover the five holes of the thermal pad area. This prevents solder from being pulled away from the thermal pad area during the reflow process. Copyright © 1999–2007, Texas Instruments Incorporated Product Folder Link(s): THS4021 THS4022 Submit Documentation Feedback 15 THS4021 THS4022 www.ti.com SLOS265C – SEPTEMBER 1999 – REVISED JULY 2007 7. Apply solder paste to the exposed thermal pad area and all of the IC terminals. 8. With these preparatory steps in place, the THS402xDGN IC is simply placed in position and run through the solder reflow operation as any standard surface-mount component. This results in a part that is properly installed. The actual thermal performance achieved with the THS402xDGN in its PowerPAD package depends on the application. In the example above, if the size of the internal ground plane is approximately 3 inches × 3 inches (7.62 cm × 7.62 cm), then the expected thermal coefficient, θJA, is about 58.4°C/W. For comparison, the non-PowerPAD version of the THS402x IC (SOIC) is shown. For a given θJA, the maximum power dissipation is shown in Figure 39 and is calculated by the following formula: ǒ T P D + –T MAX A q JA Ǔ where: PD = Maximum power dissipation of THS402x IC (watts) TMAX = Absolute maximum junction temperature (150°C) TA = Free-ambient air temperature (°C) θJA = θJC + θCA θJC = Thermal coefficient from junction to case θCA = Thermal coefficient from case to ambient air (°C/W) MAXIMUM POWER DISSIPATION vs FREE-AIR TEMPERATURE 3.5 DGN Package θJA = 58.4°C/W 2 oz. Trace And Copper Pad With Solder Maximum Power Dissipation − W 3.0 DGN Package θJA = 158°C/W 2 oz. Trace And Copper Pad Without Solder 2.5 SOIC Package High-K Test PCB θJA = 98°C/W 2.0 TJ = 150°C 1.5 1.0 0.5 SOIC Package Low-K Test PCB θJA = 167°C/W 0.0 −40 −20 0 20 40 60 80 TA − Free-Air Temperature − °C 100 G029 NOTE: Results are with no air flow and PCB size = 3 in. × 3 in. (7.62 cm × 7.62 cm). Figure 39. Maximum Power Dissipation vs Free-Air Temperature More-complete details of the thermal pad installation process and thermal management techniques can be found in the PowerPAD Thermally Enhanced Package application report (SLMA002). 16 Submit Documentation Feedback Copyright © 1999–2007, Texas Instruments Incorporated Product Folder Link(s): THS4021 THS4022 THS4021 THS4022 www.ti.com SLOS265C – SEPTEMBER 1999 – REVISED JULY 2007 The next consideration is the package constraints. The two sources of heat within an amplifier are quiescent power and output power. The designer should never forget about the quiescent heat generated within the device, especially with multiamplifier devices. Because these devices have linear output stages (Class A-B), most of the heat dissipation is at low output voltages with high output currents. Figure 40 through Figure 43 show this effect, along with the quiescent heat, with an ambient air temperature of 50°C. Obviously, as the ambient temperature increases, the limit lines shown drop accordingly. The area under each respective limit line is considered the safe operating area. Any condition above this line exceeds the amplifier limits and failure may result. When using VCC = ±5 V, there is generally not a heat problem, even with SOIC packages. But, when using VCC = ±15 V, the SOIC package is severely limited in the amount of heat it can dissipate. The other key factor when looking at these graphs is how the devices are mounted on the PCB. The PowerPAD devices are extremely useful for heat dissipation. But the device should always be soldered to a copper plane to use fully the heat dissipation properties of the thermal pad. The SOIC package, on the other hand, is highly dependent on how it is mounted on the PCB. As more trace and copper area is placed around the device, θJA decreases and the heat dissipation capability increases. The currents and voltages shown in these graphs are for the total package. For the dual-amplifier package (THS4022), the sum of the RMS output currents and voltages should be used to choose the proper package. The graphs shown assume that both amplifier outputs are identical. THS4021 MAXIMUM RMS OUTPUT CURRENT vs RMS OUTPUT VOLTAGE DUE TO THERMAL LIMITS VCC = ± 5 V Tj = 150°C TA = 50°C 180 1k Maximum Output Current Limit Line |IO| − Maximum RMS Output Current − mA |IO| − Maximum RMS Output Current − mA 200 THS4021 MAXIMUM RMS OUTPUT CURRENT vs RMS OUTPUT VOLTAGE DUE TO THERMAL LIMITS 160 140 Package With θJA < = 120°C/W 120 100 SO-8 Package θJA = 167°C/W Low-K Test PCB 80 60 40 20 Safe Operating Area 0 VCC = ± 15 V TJ = 150°C TA = 50°C Maximum Output Current Limit Line DGN Package θJA = 58.4°C/W 100 SO-8 Package θJA = 98°C/W High-K Test PCB SO-8 Package θJA = 167°C/W Low-K Test PCB Safe Operating Area 10 0 1 2 3 4 |VO| − RMS Output Voltage − V 5 0 G030 Figure 40. 3 6 9 12 |VO| − RMS Output Voltage − V 15 G031 Figure 41. Copyright © 1999–2007, Texas Instruments Incorporated Product Folder Link(s): THS4021 THS4022 Submit Documentation Feedback 17 THS4021 THS4022 www.ti.com SLOS265C – SEPTEMBER 1999 – REVISED JULY 2007 THS4022 MAXIMUM RMS OUTPUT CURRENT vs RMS OUTPUT VOLTAGE DUE TO THERMAL LIMITS 180 1k Maximum Output Current Limit Line Package With θJA ≤ 60°C/W |IO| − Maximum RMS Output Current − mA |IO| − Maximum RMS Output Current − mA 200 THS4022 MAXIMUM RMS OUTPUT CURRENT vs RMS OUTPUT VOLTAGE DUE TO THERMAL LIMITS 160 140 120 100 SO-8 Package θJA = 167°C/W Low-K Test PCB 80 60 Safe Operating Area 40 VCC = ± 5 V TJ = 150°C TA = 50°C Both Channels SO-8 Package θJA = 98°C/W High-K Test PCB 20 VCC = ± 15 V TJ = 150°C TA = 50°C Both Channels 100 SO-8 Package θJA = 98°C/W High-K Test PCB 10 DGN Package θJA = 58.4°C/W Safe Operating Area 0 SO-8 Package θJA = 167°C/W Low-K Test PCB 1 0 1 2 3 4 |VO| − RMS Output Voltage − V 5 0 3 Submit Documentation Feedback 6 9 12 |VO| − RMS Output Voltage − V G032 Figure 42. 18 Maximum Output Current Limit Line 15 G033 Figure 43. Copyright © 1999–2007, Texas Instruments Incorporated Product Folder Link(s): THS4021 THS4022 THS4021 THS4022 www.ti.com SLOS265C – SEPTEMBER 1999 – REVISED JULY 2007 Evaluation Board Evaluation boards are available for the THS4021 (literature number SLOP129) and THS4022 (literature number SLOP231). These boards have been configured for very low parasitic capacitance in order to realize the full performance of the amplifier. A schematic of the THS4021 evaluation board is shown in Figure 44. The circuitry has been designed so that the amplifier may be used in either an inverting or noninverting configuration. For more information, see the THS4021 High-Speed Operational Amplifier Evaluation Module user’s guide (SLOU063) or the THS4022 Dual High-Speed Operational Amplifier Evaluation Module user’s guide (SLOU064). To order the evaluation board, contact your local TI sales office or distributor or visit the Texas Instruments Web site at www.ti.com. VCC+ + C3 0.1 mF R4 1 kW C2 6.8 mF NULL + IN+ R5 49.9 W THS4021 R3 49.9 W OUT _ NULL R2 49.9 W C4 0.1 mF + C1 6.8 mF IN– VCC– S0282-01 Figure 44. THS4021 Evaluation Board Copyright © 1999–2007, Texas Instruments Incorporated Product Folder Link(s): THS4021 THS4022 Submit Documentation Feedback 19 PACKAGE OPTION ADDENDUM www.ti.com 14-Oct-2022 PACKAGING INFORMATION Orderable Device Status (1) Package Type Package Pins Package Drawing Qty Eco Plan (2) Lead finish/ Ball material MSL Peak Temp Op Temp (°C) Device Marking (3) Samples (4/5) (6) THS4021CD ACTIVE SOIC D 8 75 RoHS & Green NIPDAU Level-1-260C-UNLIM 0 to 70 4021C Samples THS4021CDGN ACTIVE HVSSOP DGN 8 80 RoHS & Green NIPDAU Level-1-260C-UNLIM 0 to 70 ACK Samples THS4021CDGNR ACTIVE HVSSOP DGN 8 2500 RoHS & Green NIPDAU Level-1-260C-UNLIM 0 to 70 ACK Samples THS4021ID ACTIVE SOIC D 8 75 RoHS & Green NIPDAU Level-1-260C-UNLIM -40 to 85 4021I Samples THS4021IDGN ACTIVE HVSSOP DGN 8 80 RoHS & Green NIPDAU Level-1-260C-UNLIM -40 to 85 ACL Samples THS4021IDGNR ACTIVE HVSSOP DGN 8 2500 RoHS & Green NIPDAU Level-1-260C-UNLIM -40 to 85 ACL Samples THS4021IDR ACTIVE SOIC D 8 2500 RoHS & Green NIPDAU Level-1-260C-UNLIM -40 to 85 4021I Samples THS4022CD ACTIVE SOIC D 8 75 RoHS & Green NIPDAU Level-1-260C-UNLIM 0 to 70 4022C Samples THS4022CDGN ACTIVE HVSSOP DGN 8 80 RoHS & Green NIPDAU Level-1-260C-UNLIM 0 to 70 ACA Samples THS4022CDGNR ACTIVE HVSSOP DGN 8 2500 RoHS & Green NIPDAU Level-1-260C-UNLIM 0 to 70 ACA Samples THS4022ID ACTIVE SOIC D 8 75 RoHS & Green NIPDAU Level-1-260C-UNLIM -40 to 85 4022I Samples THS4022IDGN ACTIVE HVSSOP DGN 8 80 RoHS & Green Call TI | NIPDAU Level-1-260C-UNLIM -40 to 85 ACB Samples THS4022IDGNR ACTIVE HVSSOP DGN 8 2500 RoHS & Green Call TI | NIPDAU Level-1-260C-UNLIM -40 to 85 ACB Samples (1) The marketing status values are defined as follows: ACTIVE: Product device recommended for new designs. LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect. NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design. PREVIEW: Device has been announced but is not in production. Samples may or may not be available. OBSOLETE: TI has discontinued the production of the device. (2) RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may reference these types of products as "Pb-Free". RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption. Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of
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