DGN-8
DGK-8
THS4500
THS4501
D-8
SLOS350F – APRIL 2002 – REVISED OCTOBER 2011
www.ti.com
WIDEBAND, LOW-DISTORTION, FULLY DIFFERENTIAL AMPLIFIERS
Check for Samples: THS4500, THS4501
FEATURES
APPLICATIONS
•
•
•
•
•
•
•
•
1
23
•
•
•
Fully Differential Architecture
Bandwidth: 370 MHz
Slew Rate: 2800 V/μs
IMD3: –90 dBc at 30 MHz
OIP3: 49 dBm at 30 MHz
Output Common-Mode Control
Wide Power-Supply Voltage Range: 5 V, ±5 V,
12 V, 15 V
Input Common-Mode Range Shifted to Include
Negative Power-Supply Rail
Power-Down Capability (THS4500)
Evaluation Module Available
High Linearity Analog-to-Digital Converter
Preamplifier
Wireless Communication Receiver Chains
Single-Ended to Differential Conversion
Differential Line Driver
Active Filtering of Differential Signals
•
•
•
•
VIN−
1
8
VIN+
VOCM
2
7
PD
VS+
3
6
VS−
VOUT+
4
5
VOUT−
DESCRIPTION
RELATED DEVICES
DEVICE (1)
The THS4500 and THS4501 are high-performance
fully differential amplifiers from Texas Instruments.
The THS4500, featuring power-down capability, and
the THS4501, without power-down capability, set new
performance standards for fully differential amplifiers
with unsurpassed linearity, supporting 14-bit
operation through 40 MHz. Package options include
the SOIC-8 and the MSOP-8 with PowerPAD™ for a
smaller footprint, enhanced ac performance, and
improved thermal dissipation capability.
(1)
0.1 µF
5V
10 µF
24.9 Ω
+
−
−
ADC
12 Bit/80 MSps
IN
VOCM
1 µF
3.3 V, 100 MHz, 43 V/μs, 3.7 nV/√Hz
THS4130/1
±15 V, 150 MHz, 51 V/μs, 1.3 nV/√Hz
THS4140/1
±15 V, 160 MHz, 450 V/μs, 6.5 nV/√Hz
THS4150/1
±15 V, 150 MHz, 650 V/μs, 7.6 nV/√Hz
IN
+
24.9 Ω
402 Ω
392 Ω
10 pF
Vref
THIRD-ORDER INTERMODULATION
DISTORTION
10
−62
VS = 5 V
−68
VS = ±5 V
−74
12
Bits
VS
370 MHz, 2800 V/μs, Centered VICR
THS4120/1
IMD 3 − Third-Order Intermodulation Distortion − dBc
5V
374 Ω
THS4502/3
Even-numbered devices feature power-down capability.
392 Ω
56.2 Ω
370 MHz, 2800 V/μs, VICR Includes VS–
10 pF
APPLICATION CIRCUIT DIAGRAM
50 Ω
DESCRIPTION
THS4500/1
−80
392 Ω
50 Ω
−86
374 Ω
2.5 V
56.2 Ω
VS
−92
402 Ω
VS+
VOUT
+−
−+
VS−
392 Ω
−98
10
20
30
40
50
60
14
800 Ω
VOCM
70
80
90
16
100
f − Frequency − MHz
1
2
3
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas
Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
PowerPAD is a trademark of Texas Instruments, Incorporated.
All other trademarks are the property of their respective owners.
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of the Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
Copyright © 2002–2011, Texas Instruments Incorporated
THS4500
THS4501
SLOS350F – APRIL 2002 – REVISED OCTOBER 2011
www.ti.com
This integrated circuit can be damaged by ESD. Texas Instruments recommends that all integrated circuits be handled with
appropriate precautions. Failure to observe proper handling and installation procedures can cause damage.
ESD damage can range from subtle performance degradation to complete device failure. Precision integrated circuits may be more
susceptible to damage because very small parametric changes could cause the device not to meet its published specifications.
ABSOLUTE MAXIMUM RATINGS (1)
Over operating free-air temperature range, unless otherwise noted.
UNIT
Supply voltage, VS
16.5 V
±VS
Input voltage, VI
Output current, IO
(2)
150 mA
Differential input voltage, VID
4V
Continuous power dissipation
See Dissipation Rating Table
Maximum junction temperature, TJ
(3)
+150°C
Maximum junction temperature, continuous operation, long-term reliability, TJ
Operating free-air temperature range, TA
+125°C
(4)
C suffix
0°C to +70°C
I suffix
–40°C to +85°C
–65°C to +150°C
Storage temperature range, Tstg
Lead temperature 1,6 mm (1/16 inch) from case for 10 seconds
+300°C
ESD rating:
(1)
(2)
(3)
(4)
HBM
4000 V
CDM
1000 V
MM
100 V
Stresses above these ratings may cause permanent damage. Exposure to absolute maximum conditions for extended periods may
degrade device reliability. These are stress ratings only, and functional operation of the device at these or any other conditions beyond
those specified is not implied.
The THS4500/1 may incorporate a PowerPAD on the underside of the chip. This acts as a heat sink and must be connected to a
thermally dissipative plane for proper power dissipation. Failure to do so may result in exceeding the maximum junction temperature
which could permanently damage the device. See TI technical briefs SLMA002 and SLMA004 for more information about utilizing the
PowerPAD thermally-enhanced package.
The absolute maximum temperature under any condition is limited by the constraints of the silicon process.
The maximum junction temperature for continuous operation is limited by package constraints. Operation above this temperature may
result in reduced reliability and/or lifetime of the device.
DISSIPATION RATINGS TABLE
(1)
(2)
POWER RATING (2)
PACKAGE
θJC
(°C/W)
θJA (1)
(°C/W)
TA ≤ +25°C
TA = +85°C
D (8-pin)
38.3
97.5
1.02 W
410 mW
DGN (8-pin)
4.7
58.4
1.71 W
685 mW
DGK (8-pin)
54.2
260
385 mW
154 mW
This data was taken using the JEDEC standard High-K test PCB.
Power rating is determined with a junction temperature of +125°C. This is the point where distortion starts to substantially increase.
Thermal management of the final PCB should strive to keep the junction temperature at or below +125°C for best performance and
long-term reliability.
RECOMMENDED OPERATING CONDITIONS
MIN
Supply voltage
Operating free- air temperature, TA
2
Submit Documentation Feedback
Dual supply
Single supply
C suffix
I suffix
4.5
NOM
MAX
±5
±7.5
5
15
0
+70
–40
+85
UNIT
V
°C
Copyright © 2002–2011, Texas Instruments Incorporated
Product Folder Link(s): THS4500 THS4501
THS4500
THS4501
SLOS350F – APRIL 2002 – REVISED OCTOBER 2011
www.ti.com
PACKAGE/ORDERING INFORMATION (1)
ORDERABLE PACKAGE AND NUMBER
TEMPERATURE
0°C to +70°C
–40°C to +85°C
(1)
(2)
PLASTIC MSOP (2)
PowerPAD
PLASTIC SMALL
OUTLINE (D)
PLASTIC MSOP (2)
DGN
SYMBOL
DGK
SYMBOL
THS4500CD
THS4500CDGN
BFB
THS4500CDGK
ATVB
THS4501CD
THS4501CDGN
BFD
THS4501CDGK
ATW
THS4500ID
THS4500IDGN
BFC
THS4500IDGK
ASV
THS4501ID
THS4501IDGN
BFE
THS4501IDGK
ASW
For the most current package and ordering information see the Package Option Addendum at the end of this document, or see the TI
web site at www.ti.com.
All packages are available taped and reeled. The R suffix standard quantity is 2500. The T suffix standard quantity is 250 (for example,
THS4501DT).
PIN ASSIGNMENTS
THS4500
(TOP VIEW)
D, DGN, DGK
THS4501
(TOP VIEW)
D, DGN, DGK
VIN- 1
8 VIN+
VIN- 1
8 VIN+
VOCM 2
7 PD
VOCM 2
7 NC
VS+ 3
6 VS-
VS+ 3
6 VS-
VOUT+ 4
5 VOUT-
VOUT+ 4
Copyright © 2002–2011, Texas Instruments Incorporated
Product Folder Link(s): THS4500 THS4501
5 VOUT-
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3
THS4500
THS4501
SLOS350F – APRIL 2002 – REVISED OCTOBER 2011
www.ti.com
ELECTRICAL CHARACTERISTICS: VS = ±5 V
RF = RG = 392 Ω, RL = 800 Ω, G = +1, and single-ended input, unless otherwise noted.
THS4500 AND THS4501
PARAMETER
TEST CONDITIONS
TYP
+25°C
OVER TEMPERATURE
+25°C
0°C to
+70°C
–40°C to
+85°C
UNITS
MIN/
TYP/
MAX
AC PERFORMANCE
G = +1, PIN = –20 dBm, RF = 392 Ω
370
MHz
Typ
G = +2, PIN = –30 dBm, RF = 1 kΩ
175
MHz
Typ
G = +5, PIN = –30 dBm, RF = 2.4 kΩ
70
MHz
Typ
G = +10, PIN = –30 dBm, RF = 5.1 kΩ
30
MHz
Typ
G > +10
300
MHz
Typ
PIN = –20 dBm
150
MHz
Typ
VP = 2 V
220
MHz
Typ
Slew rate
4 VPP Step
2800
V/μs
Typ
Rise time
2 VPP Step
0.4
ns
Typ
Fall time
2 VPP Step
0.5
ns
Typ
VO = 4 VPP
8.3
ns
Typ
VO = 4 VPP
6.3
ns
Typ
Small-signal bandwidth
Gain-bandwidth product
Bandwidth for 0.1-dB flatness
Large-signal bandwidth
Settling time to 0.01%
0.1%
Harmonic distortion
G = +1, VO = 2 VPP
f = 8 MHz
–82
dBc
Typ
f = 30 MHz
–71
dBc
Typ
f = 8 MHz
–97
dBc
Typ
f = 30 MHz
–74
dBc
Typ
VO= 2 VPP, fC= 30 MHz, RF = 392 Ω,
200 kHz tone spacing
–90
dBc
Typ
fC = 30 MHz, RF = 392 Ω,
Referenced to 50 Ω
49
dBm
Typ
2nd harmonic
3rd harmonic
Third-order intermodulation
distortion
Typ
Third-order output intercept point
Input voltage noise
f > 1 MHz
7
nV/√Hz
Typ
Input current noise
f > 100 kHz
1.7
pA/√Hz
Typ
Overdrive = 5.5 V
60
ns
Typ
Overdrive recovery time
DC PERFORMANCE
Open-loop voltage gain
55
52
50
50
dB
Min
Input offset voltage
–4
–7/–1
–8/0
–9/+1
mV
Max
±10
±10
μV/°C
Typ
4
4.6
5
5.2
μA
Max
±10
±10
nA/°C
Typ
Average offset voltage drift
Input bias current
Average bias current drift
Input offset current
0.5
1
Average offset current drift
2
2
μA
Max
±40
±40
nA/°C
Typ
V
Min
INPUT
Common-mode input range
Common-mode rejection ratio
–5.7/2.6
–5.4/2.3
–5.1/2
–5.1/2
80
74
70
70
107 || 1
Input impedance
dB
Min
Ω || pF
Typ
Min
OUTPUT
Differential output voltage swing
RL = 1 kΩ
±8
±7.6
±7.4
±7.4
V
Differential output current drive
RL = 20 Ω
120
110
100
100
mA
Min
PIN = –20 dBm, f = 100 kHz
–58
dB
Typ
f = 1 MHz
0.1
Ω
Typ
Output balance error
Closed-loop output impedance
(single-ended)
4
Submit Documentation Feedback
Copyright © 2002–2011, Texas Instruments Incorporated
Product Folder Link(s): THS4500 THS4501
THS4500
THS4501
SLOS350F – APRIL 2002 – REVISED OCTOBER 2011
www.ti.com
ELECTRICAL CHARACTERISTICS: VS = ±5 V (continued)
RF = RG = 392 Ω, RL = 800 Ω, G = +1, and single-ended input, unless otherwise noted.
THS4500 AND THS4501
PARAMETER
TEST CONDITIONS
TYP
+25°C
OVER TEMPERATURE
+25°C
0°C to
+70°C
–40°C to
+85°C
UNITS
MIN/
TYP/
MAX
OUTPUT COMMON-MODE VOLTAGE CONTROL
Small-signal bandwidth
RL = 400 Ω
180
MHz
Typ
Slew rate
2 VPP Step
92
V/μs
Typ
0.98
V/V
Min
Minimum gain
1
Maximum gain
Common-mode offset voltage
Input bias current
VOCM = 2.5 V
Input voltage range
Input impedance
0.98
0.98
1
1.02
1.02
1.02
V/V
Max
–0.4
–4.6/+3.8
–6.6/+5.8
–7.6/+6.8
mV
Max
100
150
170
170
μA
Max
±4
±3.7
±3.4
±3.4
V
Min
kΩ || pF
Typ
Maximum default voltage
VOCM left floating
25 || 1
0
0.05
0.10
0.10
V
Max
Minimum default voltage
VOCM left floating
0
–0.05
–0.10
–0.10
V
Min
Specified operating voltage
±5
7.5
7.5
7.5
V
Max
Maximum quiescent current
23
28
32
34
mA
Max
Minimum quiescent current
23
18
14
12
mA
Min
Power-supply rejection (±PSRR)
80
76
73
70
dB
Min
V
Min
POWER SUPPLY
POWER-DOWN (THS4500 ONLY)
Enable voltage threshold
Device enabled ON above –2.9 V
Disable voltage threshold
Device disabled OFF below –4.3 V
–2.9
–4.3
V
Max
Power-down quiescent current
800
1000
1200
1200
μA
Max
Input bias current
200
240
260
260
μA
Max
Input impedance
50 || 1
kΩ || pF
Typ
Turn-on time delay
1000
ns
Typ
Turn-off time delay
800
ns
Typ
Copyright © 2002–2011, Texas Instruments Incorporated
Product Folder Link(s): THS4500 THS4501
Submit Documentation Feedback
5
THS4500
THS4501
SLOS350F – APRIL 2002 – REVISED OCTOBER 2011
www.ti.com
ELECTRICAL CHARACTERISTICS: VS = 5 V
RF = RG = 392 Ω, RL = 800 Ω, G = +1, and single-ended input, unless otherwise noted.
THS4500 AND THS4501
PARAMETER
TEST CONDITIONS
TYP
+25°C
OVER TEMPERATURE
+25°C
0°C to
+70°C
–40°C to
+85°C
UNITS
MIN/T
YP/M
AX
AC PERFORMANCE
G = +1, PIN = –20 dBm, RF = 392 Ω
320
MHz
Typ
G = +2, PIN = –30 dBm, RF = 1 kΩ
160
MHz
Typ
G = +5, PIN = –30 dBm, RF = 2.4 kΩ
60
MHz
Typ
G = +10, PIN = –30 dBm, RF = 5.1 kΩ
30
MHz
Typ
G > +10
300
MHz
Typ
PIN = –20 dBm
180
MHz
Typ
VP = 1 V
200
MHz
Typ
Slew rate
2 VPP Step
1300
V/μs
Typ
Rise time
2 VPP Step
0.5
ns
Typ
Fall time
2 VPP Step
0.6
ns
Typ
VO = 2 V Step
13.1
ns
Typ
VO = 2 V Step
8.3
ns
Typ
Small-signal bandwidth
Gain-bandwidth product
Bandwidth for 0.1-dB flatness
Large-signal bandwidth
Settling time to 0.01%
0.1%
Harmonic distortion
G = +1, VO = 2 VPP
2nd harmonic
3rd harmonic
Typ
f = 8 MHz,
–80
dBc
Typ
f = 30 MHz
–55
dBc
Typ
f = 8 MHz
–76
dBc
Typ
f = 30 MHz
–60
dBc
Typ
Input voltage noise
f > 1 MHz
7
nV/√Hz
Typ
Input current noise
f > 100 kHz
1.7
pA/√Hz
Typ
Overdrive = 5.5 V
60
ns
Typ
Overdrive recovery time
DC PERFORMANCE
Open-loop voltage gain
54
51
49
49
dB
Min
Input offset voltage
–4
–7/–1
–8/0
–9/+1
mV
Max
±10
±10
μV/°C
Typ
4
4.6
5
5.2
μA
Max
±10
±10
nA/°C
Typ
0.5
0.7
1.2
1.2
μA
Max
±20
±20
nA/°C
Typ
V
Min
Average offset voltage drift
Input bias current
Average bias current drift
Input offset current
Average offset current drift
INPUT
Common-mode input range
Common-mode rejection ratio
–0.7/2.6
–0.4/2.3
–0.1/2
–0.1/2
80
74
70
70
107 || 1
Input Impedance
dB
Min
Ω || pF
Typ
Min
OUTPUT
RL = 1 kΩ, Referenced to 2.5 V
±3.3
±3
±2.8
±2.8
V
Output current drive
RL = 20 Ω
100
90
80
80
mA
Min
Output balance error
PIN = –20 dBm, f = 100 kHz
–58
dB
Typ
f = 1 MHz
0.1
Ω
Typ
Differential output voltage swing
Closed-loop output impedance
(single-ended)
6
Submit Documentation Feedback
Copyright © 2002–2011, Texas Instruments Incorporated
Product Folder Link(s): THS4500 THS4501
THS4500
THS4501
SLOS350F – APRIL 2002 – REVISED OCTOBER 2011
www.ti.com
ELECTRICAL CHARACTERISTICS: VS = 5 V (continued)
RF = RG = 392 Ω, RL = 800 Ω, G = +1, and single-ended input, unless otherwise noted.
THS4500 AND THS4501
PARAMETER
TEST CONDITIONS
TYP
+25°C
OVER TEMPERATURE
+25°C
0°C to
+70°C
–40°C to
+85°C
UNITS
MIN/T
YP/M
AX
OUTPUT COMMON-MODE VOLTAGE CONTROL
Small-signal bandwidth
RL = 400 Ω
180
MHz
Typ
Slew rate
2 VPP Step
80
V/μs
Typ
0.98
V/V
Min
Minimum gain
1
Maximum gain
Common-mode offset voltage
Input bias current
VOCM = 2.5 V
Input voltage range
Input impedance
0.98
0.98
1
1.02
1.02
1.02
V/V
Max
0.4
–2.6/3.4
–4.2/5.4
–5.6/6.4
mV
Max
1
2
3
3
μA
Max
1/4
1.2/3.8
1.3/3.7
1.3/3.7
V
Min
kΩ || pF
Typ
Maximum default voltage
VOCM left floating
25 || 1
2.5
2.55
2.6
2.6
V
Max
Minimum default voltage
VOCM left floating
2.5
2.45
2.4
2.4
V
Min
Specified operating voltage
5
15
15
15
V
Max
Maximum quiescent current
20
25
29
31
mA
Max
Minimum quiescent current
20
16
12
10
mA
Min
Power-supply rejection (+PSRR)
75
72
69
66
dB
Min
V
Min
POWER SUPPLY
POWER -DOWN (THS4500 ONLY)
Enable voltage threshold
Device enabled ON above 2.1 V
Disable voltage threshold
Device disabled OFF below 0.7 V
2.1
V
Max
Power-down quiescent current
600
800
0.7
1200
1200
μA
Max
Input bias current
100
125
140
140
μA
Max
Input impedance
50 || 1
kΩ || pF
Typ
Turn-on time delay
1000
ns
Typ
Turn-off time delay
800
ns
Typ
Copyright © 2002–2011, Texas Instruments Incorporated
Product Folder Link(s): THS4500 THS4501
Submit Documentation Feedback
7
THS4500
THS4501
SLOS350F – APRIL 2002 – REVISED OCTOBER 2011
www.ti.com
TYPICAL CHARACTERISTICS
Table of Graphs (±5 V)
FIGURE
Small-signal unity-gain frequency response
1
Small-signal frequency response
2
0.1-dB gain flatness frequency response
3
Large-signal frequency response
4
Harmonic distortion (single-ended input to differential output)
vs Frequency
5, 7, 13, 15
Harmonic distortion (differential input to differential output)
vs Frequency
6, 8, 14, 16
Harmonic distortion (single-ended input to differential output)
vs Output voltage swing
9, 11, 17, 19
Harmonic distortion (differential input to differential output)
vs Output voltage swing
10, 12, 18, 20
Harmonic distortion (single-ended input to differential output)
vs Load resistance
21
Harmonic distortion (differential input to differential output)
vs Load resistance
22
Third-order intermodulation distortion
(single-ended input to differential output)
vs Frequency
23
Third-order output intercept point
vs Frequency
24
Slew rate
vs Differential output voltage step
Settling time
25
26, 27
Large-signal transient response
28
Small-signal transient response
29
Overdrive recovery
30, 31
Voltage and current noise
vs Frequency
32
Rejection ratios
vs Frequency
33
Rejection ratios
vs Case temperature
34
Output balance error
vs Frequency
35
Open-loop gain and phase
vs Frequency
36
Open-loop gain
vs Case temperature
37
Input bias offset current
vs Case temperature
38
Quiescent current
vs Supply voltage
39
Input offset voltage
vs Case temperature
40
Common-mode rejection ratio
vs Input common-mode range
41
Output drive
vs Case temperature
42
Harmonic distortion
(single-ended and differential input to differential output)
vs Output common-mode voltage
43
Small-signal frequency response at VOCM
44
Output offset voltage at VOCM
vs Output common-mode voltage
45
Quiescent current
vs Power-down voltage
46
Turn-on and turn-off delay times
Single-ended output impedance in power-down
Power-down quiescent current
8
Submit Documentation Feedback
47
vs Frequency
48
vs Case temperature
49
vs Supply voltage
50
Copyright © 2002–2011, Texas Instruments Incorporated
Product Folder Link(s): THS4500 THS4501
THS4500
THS4501
SLOS350F – APRIL 2002 – REVISED OCTOBER 2011
www.ti.com
Table of Graphs (5 V)
FIGURE
Small-signal unity-gain frequency response
51
Small-signal frequency response
52
0.1-dB gain flatness frequency response
53
Large-signal frequency response
54
Harmonic distortion (single-ended input to differential output)
vs Frequency
55, 57, 63, 65
Harmonic distortion (differential input to differential output)
vs Frequency
56, 58, 64, 66
Harmonic distortion (single-ended input to differential output)
vs Output voltage swing
59, 61, 67, 69
Harmonic distortion (differential input to differential output)
vs Output voltage swing
60, 62, 68, 70
Harmonic distortion (single-ended input to differential output)
vs Load resistance
71
Harmonic distortion (differential input to differential output)
vs Load resistance
72
Third-order intermodulation distortion
vs Frequency
73
Third-order intercept point
vs Frequency
74
Slew rate
vs Differential output voltage step
75
Large-signal transient response
76
Small-signal transient response
77
Voltage and current noise
vs Frequency
78
Rejection ratios
vs Frequency
79
Rejection ratios
vs Case temperature
80
Output balance error
vs Frequency
81
Open-loop gain and phase
vs Frequency
82
Open-loop gain
vs Case temperature
83
Input bias offset current
vs Case temperature
84
Quiescent current
vs Supply voltage
85
Input offset voltage
vs Case temperature
86
Common-mode rejection ratio
vs Input common-mode range
87
Output drive
vs Case temperature
88
Harmonic distortion (single-ended and differential input)
vs Output common-mode voltage
89
Small-signal frequency response at VOCM
90
Output offset voltage
vs Output common-mode voltage
91
Quiescent current
vs Power-down voltage
92
Turn-on and turn-off delay times
93
Single-ended output impedance in power-down
Power-down quiescent current
vs Frequency
94
vs Case temperature
95
vs Supply voltage
96
Copyright © 2002–2011, Texas Instruments Incorporated
Product Folder Link(s): THS4500 THS4501
Submit Documentation Feedback
9
THS4500
THS4501
SLOS350F – APRIL 2002 – REVISED OCTOBER 2011
www.ti.com
TYPICAL CHARACTERISTICS: ±5 V
SMALL-SIGNAL FREQUENCY
RESPONSE
1
22
0.5
20
0.3
−1
−1.5
−2
Gain = 1
RL = 800 Ω
Rf = 392 Ω
PIN = −20 dBm
VS = ±5 V
−2.5
−3
0.2
1
14
Gain = 5, Rf = 2.4 kΩ
12
10
8
4
RL = 800 Ω
PIN = −30 dBm
0
VS = ±5 V
−2
0.1
1
10
100
1000
100
HARMONIC DISTORTION
vs
FREQUENCY
HARMONIC DISTORTION
vs
FREQUENCY
1
−30
−40
−50
−60
−70
HD2
−80
10
100
−100
0.1
1000
Differential Input to
Differential Output
Gain = 1
RL = 800 Ω
Rf = 392 Ω
VO = 1 VPP
VS = ±5 V
−10
−90
−4
0.1
−20
0
Single-Ended Input to
Differential Output
Gain = 1
RL = 800 Ω
Rf = 392 Ω
VO = 1 VPP
VS = ±5 V
Harmonic Distortion − dBc
−3
10
−30
−40
−50
−60
−70
HD2
−80
−90
HD3
1
−20
HD3
−100
0.1
100
1
10
100
f − Frequency − MHz
f − Frequency − MHz
f − Frequency − MHz
Figure 4.
Figure 5.
Figure 6.
HARMONIC DISTORTION
vs
FREQUENCY
HARMONIC DISTORTION
vs
FREQUENCY
HARMONIC DISTORTION
vs
OUTPUT VOLTAGE SWING
0
0
Single-Ended Input to
Differential Output
Gain = 1
RL = 800 Ω
Rf = 392 Ω
VO = 2 VPP
VS = ±5 V
Harmonic Distortion − dBc
−10
−60
−70
HD2
HD3
−90
1
10
f − Frequency − MHz
Figure 7.
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−20
−30
−40
−50
Differential Input to
Differential Output
Gain = 1
RL = 800 Ω
Rf = 392 Ω
VO = 2 VPP
VS = ±5 V
−60
−70
−80
HD2
−90
100
−100
0.1
Single-Ended Input to
Differential Output
Gain = 1
RL = 800 Ω
Rf = 392 Ω
f= 8 MHz
VS = ±5 V
−10
Harmonic Distortion − dBc
0
−100
0.1
1000
LARGE-SIGNAL FREQUENCY
RESPONSE
Gain = 1
RL = 800 Ω
Rf = 392 Ω
PIN = 10 dBm
VS = ±5 V
−80
100
Figure 3.
−2
−50
10
1
1000
Figure 2.
Harmonic Distortion − dBc
Large Signal Gain − dB
−0.3
10
f − Frequency − MHz
−1
−40
Rf = 392 Ω
−0.1
Figure 1.
0
−30
0
−0.2
0
−20
Rf = 499 Ω
f − Frequency − MHz
−10
−10
0.1
f − Frequency − MHz
1
Harmonic Distortion − dBc
Gain = 2, Rf = 1 kΩ
6
2
−4
0.1
16
Gain = 1
RL = 800 Ω
PIN = −20 dBm
VS = ±5 V
Gain = 10, Rf = 5.1 kΩ
0.1 dB Gain Flatness − dB
0
−0.5
−3.5
10
0.1-dB GAIN FLATNESS
FREQUENCY RESPONSE
18
Small Signal Gain − dB
Small Signal Unity Gain − dB
SMALL-SIGNAL UNITY-GAIN
FREQUENCY RESPONSE
−20
−30
−40
−50
−60
−70
HD2
−80
−90
HD3
HD3
−100
1
10
f − Frequency − MHz
100
0
0.5
1
1.5 2
2.5
3
3.5 4
4.5
5
VO − Output Voltage Swing − V
Figure 8.
Figure 9.
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TYPICAL CHARACTERISTICS: ±5 V (continued)
HARMONIC DISTORTION
vs
OUTPUT VOLTAGE SWING
HARMONIC DISTORTION
vs
OUTPUT VOLTAGE SWING
0
Differential Input to
Differential Output
Gain = 1
RL = 800 Ω
Rf = 499 Ω
f= 8 MHz
VS = ±5 V
−20
−30
−40
−50
−60
−70
HD2
−80
−90
HD3
−30
−40
−50
−60
HD2
−70
−80
HD3
0.5
1
1.5 2
2.5
3
3.5 4
4.5
5
0.5
1
1.5 2
2.5
3
3.5 4
4.5
2.5
3
3.5 4
4.5
HARMONIC DISTORTION
vs
FREQUENCY
HARMONIC DISTORTION
vs
FREQUENCY
1
10
f − Frequency − MHz
−30
−40
Harmonic Distortion − dBc
−20
−50
−60
HD2
−70
HD3
−80
−20
−30
−40
−50
−60
HD2
−70
−80
HD3
−90
−100
−100
0.1
100
Single-Ended Input to
Differential Output
Gain = 2
RL = 800 Ω
Rf = 1 kΩ
VO = 2 VPP
VS = ±5 V
−10
−90
−100
1
10
f − Frequency − MHz
0.1
100
1
10
f − Frequency − MHz
100
Figure 13.
Figure 14.
Figure 15.
HARMONIC DISTORTION
vs
FREQUENCY
HARMONIC DISTORTION
vs
OUTPUT VOLTAGE SWING
HARMONIC DISTORTION
vs
OUTPUT VOLTAGE SWING
0
0
Differential Input to
Differential Output
Gain = 2
RL = 800 Ω
Rf = 1 kΩ
VO = 2 VPP
VS = ±5 V
Harmonic Distortion − dBc
HD2
−60
0
Single-Ended Input to
Differential Output
Gain = 2
RL = 800 Ω
Rf = 1 kΩ
f= 8 MHz
VS = ±5 V
−10
−50
−70
−80
HD3
−20
−30
−40
−50
−60
−70
HD2
−80
HD3
Figure 16.
−30
−40
−50
−60
−70
−80
HD2
HD3
−100
0
100
−20
−90
−100
1
10
f − Frequency − MHz
Single-Ended Input to
Differential Output
Gain = 2
RL = 800 Ω
Rf = 1 kΩ
f= 8 MHz
VS = ±5 V
−10
−90
−90
5
0
Differential Input to
Differential Output
Gain = 2
RL = 800 Ω
Rf = 1 kΩ
VO = 1 VPP
VS = ±5 V
−10
HD3
−100
0.1
1.5 2
HARMONIC DISTORTION
vs
FREQUENCY
−90
−40
1
Figure 12.
HD2
−30
0.5
Figure 11.
−80
−20
HD3
Figure 10.
−70
−10
−80
VO − Output Voltage Swing − V
−60
0.1
HD2
−70
0
Harmonic Distortion − dBc
−50
−60
VO − Output Voltage Swing − V
Harmonic Distortion − dBc
−40
−50
5
0
−30
−40
VO − Output Voltage Swing − V
Single-Ended Input to
Differential Output
Gain = 2
RL = 800 Ω
Rf = 1 kΩ
VO = 1 VPP
VS = ±5 V
−20
−30
−100
0
0
−10
−20
−90
−100
0
Harmonic Distortion − dBc
−20
Differentia Input to
Differential Output
Gain = 1
RL = 800 Ω
Rf = 392 Ω
f= 30 MHz
VS = ±5 V
−10
−90
−100
Harmonic Distortion − dBc
0
Single-Ended Input to
Differential Output
Gain = 1
RL = 800 Ω
Rf = 392 Ω
f= 30 MHz
VS = ±5 V
−10
Harmonic Distortion − dBc
Harmonic Distortion − dBc
−10
Harmonic Distortion − dBc
0
HARMONIC DISTORTION
vs
OUTPUT VOLTAGE SWING
0.5
1
1.5 2
2.5
3
3.5 4
4.5
5
0
0.5
1
1.5 2
2.5
3
3.5 4
VO − Output Voltage Swing − V
VO − Output Voltage Swing − V
Figure 17.
Figure 18.
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5
11
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TYPICAL CHARACTERISTICS: ±5 V (continued)
HARMONIC DISTORTION
vs
OUTPUT VOLTAGE SWING
HARMONIC DISTORTION
vs
OUTPUT VOLTAGE SWING
0
Single-Ended Input to
Differential Output
Gain = 2
RL = 800 Ω
Rf = 1 kΩ
f= 30 MHz
VS = ±5 V
−20
−30
−40
−50
HD2
−60
−70
−80
−20
HD3
−30
−40
−50
−60
HD2
−80
HD3
−30
−40
−50
−60
0.5
1
1.5 2
2.5
3
3.5 4
4.5
5
−80
0
0.5
1
1.5 2
2.5
3
3.5 4
4.5
−100
5
0
400
800
1200
1600
RL − Load Resistance − Ω
Figure 19.
Figure 20.
Figure 21.
HARMONIC DISTORTION
vs
LOAD RESISTANCE
THIRD-ORDER INTERMODULATION
DISTORTION
vs
FREQUENCY
THIRD-ORDER OUTPUT INTERCEPT
POINT
vs
FREQUENCY
Differential Input to
Differential Output
Gain = 1
VO = 2 VPP
Rf = 392 Ω
f= 30 MHz
VS = ±5 V
−20
−30
−40
−50
−60
HD2
−70
−80
HD3
−90
−100
400
800
1200
1600
−50
Third-Order Output Intercept Point - dBm
Third-Order Intermodulation Distortion − dBc
VO − Output Voltage Swing − V
0
Single-Ended Input to
Differential Output
Gain = 1
RL = 800 Ω
Rf = 392 Ω
VO = 2 VPP
VS = ±5 V
−60
−70
−80
−90
−100
10
RL − Load Resistance − Ω
100
SLEW RATE
vs
DIFFERENTIAL OUTPUT VOLTAGE
STEP
Gain = 1
RF = 392 W
VO = 2 VPP
VS = ± 5 V
50
45
40
35
30
0
20
40
60
80
Figure 23.
Figure 24.
SETTLING TIME
Rising Edge
Rising Edge
1000
1.0
0.4
VO - Output Voltage - V
VO − Output Voltage − V
0.6
1500
120
1.5
0.8
2000
100
f - Frequency - MHz
SETTLING TIME
3000
Gain = 1
RL = 800 Ω
Rf = 392 Ω
VS = ±5 V
55
f − Frequency − MHz
Figure 22.
2500
HD3
VO − Output Voltage Swing − V
−10
0
HD2
−70
−90
−100
0
SR − Slew Rate − V/ µ s
−20
−90
−100
Single-Ended Input to
Differential Output
Gain = 1
VO = 2 VPP
Rf = 392 Ω
f= 30 MHz
VS = ±5 V
−10
−70
−90
Harmonic Distortion − dBc
0
Differentia Input to
Differential Output
Gain = 2
RL = 800 Ω
Rf = 1 kΩ
f= 8 MHz
VS = ±5 V
−10
Harmonic Distortion − dBc
Harmonic Distortion − dBc
−10
Harmonic Distortion − dBc
0
HARMONIC DISTORTION
vs
LOAD RESISTANCE
Gain = 1
RL = 800 Ω
Rf = 499 Ω
f= 1 MHz
VS = ±5 V
0.2
0
−0.2
−0.4
Falling Edge
Gain = 1
RL = 800 W
RF = 499 W
f = 1 MHz
VS = ±5 V
0.5
0
-0.5
Falling Edge
-1.0
500
−0.6
0
−0.8
0
0.5 1
1.5 2
2.5
3 3.5
4
4.5
VO − Differential Output Voltage Step − V
Figure 25.
12
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5
0
5
10
15
-1.5
20
0
2
4
6
8
t − Time − ns
t - Time - ns
Figure 26.
Figure 27.
10
12
14
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TYPICAL CHARACTERISTICS: ±5 V (continued)
SMALL-SIGNAL TRANSIENT
RESPONSE
1.5
0.3
Gain = 1
RL = 800 Ω
Rf = 499 Ω
tr/tf = 300 ps
VS = ±5 V
0.5
0
−0.5
−1
0
100
200
−0.1
−0.2
300
400
500
1.5
1
1
0.5
0
0
−1
−0.5
−2
−1
−3
−1.5
−4
−2
0
100
200
300
400
−2.5
0
500
0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1
t − Time − µs
t − Time − ns
Figure 29.
Figure 30.
OVERDRIVE RECOVERY
VOLTAGE AND CURRENT NOISE
vs
FREQUENCY
REJECTION RATIOS
vs
FREQUENCY
90
0
−50
Vn
10
PSRR+
80
70
Rejection Ratios − dB
50
Hz
Source
I n − Current Noise − pA/
100
VS = ±5 V
60
50
CMMR
PSRR−
40
30
20
In
10
RL = 800 Ω
VS = ±5 V
Sink
0
1
0.01
−150
−40−30−20−10 0 10 20 30 40 50 60 70 80 90
Case Temperature − °C
0.1
1
10
100
1000
−10
10 k
0.1
1
10
f − Frequency − MHz
f − Frequency − kHz
100
Figure 31.
Figure 32.
Figure 33.
REJECTION RATIOS
vs
CASE TEMPERATURE
OUTPUT BALANCE ERROR
vs
FREQUENCY
OPEN-LOOP GAIN AND PHASE
vs
FREQUENCY
0
−10
CMMR
Output Balance Error − dB
100
PSRR+
80
60
40
60
PIN = 10 dBm
RL = 800 Ω
Rf = 392 Ω
VS = ±5 V
RL = 800 Ω
VS = ±5 V
−20
−30
−40
−50
−60
−80
Case Temperature − °C
0.1
1
10
100
f − Frequency − MHz
Figure 34.
PIN = −30 dBm
RL = 800 Ω
VS = ±5 V
50
−70
0
−40−30−20−10 0 10 20 30 40 50 60 70 80 90
30
Gain
Open-Loop Gain − dB
120
Rejection Ratios − dB
2
2
t − Time − ns
100
20
3
−5
−0.4
−100
Hz
Output Drive − mA
0
4
Figure 28.
150
−100
Gain = 1
RL = 800 Ω
Rf = 499 Ω
tr/tf = 300 ps
VS = ±5 V
0.1
Vn − Voltage Noise − nV/
−2
−100
200
0.2
−0.3
−1.5
2.5
Gain = 4
RL = 800 Ω
Rf = 499 Ω
Overdrive = 4.5 V
VS = ±5 V
Figure 35.
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40
0
−30
−60
30
Phase
Phase − °
1
OVERDRIVE RECOVERY
5
VI − Input Voltage − V
0.4
Single-Ended Output Voltage − V
2
VO − Output Voltage − V
VO − Output Voltage − V
LARGE-SIGNAL TRANSIENT
RESPONSE
20
−90
10
−120
0
0.01
0.1
1
10
100
−150
1000
f − Frequency − MHz
Figure 36.
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TYPICAL CHARACTERISTICS: ±5 V (continued)
INPUT BIAS AND OFFSET
CURRENT
vs
CASE TEMPERATURE
RL = 800 Ω
VS = ±5 V
Open-Loop Gain − dB
56
55
54
53
52
51
50
35
0
VS = ±5 V
3.3
I IB − Input Bias Current − µ A
57
IIB−
−0.01
−0.02
3.2
IIB+
3.1
−0.03
3
−0.04
2.9
−0.05
2.8
−0.06
IOS
2.7
−0.07
2.6
−0.08
TA = 85°C
30
Quiescent Current − mA
3.4
58
QUIESCENT CURRENT
vs
SUPPLY VOLTAGE
I OS − Input Offset Current − µ A
OPEN-LOOP GAIN
vs
CASE TEMPERATURE
25
TA = −40°C
20
15
10
5
−0.09
2.5
−40−30−20−10 0 10 20 30 40 50 60 70 80 90
Case Temperature − °C
49
−40−30−20−10 0 10 20 30 40 50 60 70 80 90
0
Case Temperature − °C
200
110
VS = ±5 V
100
80
60
50
40
30
10
HD2-SE
HD2
-Diff
HD3-SE
HD3-Diff
−80
−90
−2.5 −1.5 −0.5
0.5
1.5
2.5
VOC − Output Common-Mode Voltage − V
Figure 43.
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3.5
Small Signal Frequency Response at VOCM − dB
−40
Source
0
−50
−100
0
−10
−6 −5 −4 −3 −2 −1 0
HARMONIC DISTORTION
vs
OUTPUT COMMON-MODE
VOLTAGE
−50
5
50
20
Figure 40.
Harmonic Distortion − dBc
4 4.5
100
Output Drive − mA
70
Case Temperature − °C
Single-Ended and Differential
Input to Differential Output
Gain = 1, VO = 2 VPP
f= 8 MHz, Rf = 392 Ω
VS = ±5 V
VS = ±5 V
150
90
1
2
3
4
5
Sink
−150
−40−30−20−10 0 10 20 30 40 50 60 70 80 90
6
Input Common-Mode Voltage Range − V
Case Temperature − °C
Figure 41.
Figure 42.
SMALL-SIGNAL FREQUENCY
RESPONSE AT VOCM
OUTPUT OFFSET VOLTAGE AT
VOCM
vs
OUTPUT COMMON-MODE
VOLTAGE
600
3
Gain = 1
RL = 800 Ω
Rf = 392 Ω
PIN= −20 dBm
VS = ±5 V
2
1
VOS − Output Offset Voltage − mV
CMRR − Common-Mode Rejection Ratio − dB
VOS − Input Offset Voltage − mV
1
0
−40 −30−20−10 0 10 20 30 40 50 60 70 80 90
14
3.5
OUTPUT DRIVE
vs
CASE TEMPERATURE
2
−100
−3.5
3
COMMON-MODE REJECTION RATIO
vs
INPUT COMMON-MODE RANGE
3
−70
2.5
INPUT OFFSET VOLTAGE
vs
CASE TEMPERATURE
4
−60
2
Figure 39.
5
−30
1.5
Figure 38.
6
−20
1
Figure 37.
VS = ±5 V
0
0 0.5
VS − Supply Voltage − ±V
7
−10
TA = 25°C
0
400
200
0
−200
−1
−400
−2
−600
−3
1
10
100
1000
f − Frequency − MHz
Figure 44.
−5 −4 −3 −2 −1
0
1
2
3
4
5
VOC − Output Common-Mode Voltage − V
Figure 45.
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TYPICAL CHARACTERISTICS: ±5 V (continued)
QUIESCENT CURRENT
vs
POWER-DOWN VOLTAGE
TURN-ON AND TURN-OFF DELAY
TIME
30
800
15
10
5
0
0.01
Current
0
0
−1
−2
−3
Quiescent Current − mA
Powerdown Voltage Signal − V
20
0.02
−4
−5
−5
0 0.5 1 1.5 2 2.5 3 100.5 101
t − Time − ms
Power-Down Voltage − V
Figure 46.
102
700
600
500
400
300
Gain = 1
RL = 800 Ω
Rf = 392 Ω
PIN = −1 dBm
VS = ±5 V
200
100
0
0.1
−6
−5 −4.5 −4 −3.5 −3 −2.5 −2 −1.5 −1 −0.5 0
103
1
10
100
Figure 48.
POWER-DOWN QUIESCENT CURRENT
vs
CASE TEMPERATURE
POWER-DOWN QUIESCENT CURRENT
vs
SUPPLY VOLTAGE
1000
1000
900
RL = 800 Ω
VS = ±5 V
800
700
600
500
400
300
200
100
0
−40 −30−20−10 0 10 20 30 40 50 60 70 80 90
1000
f − Frequency − MHz
Figure 47.
Power-Down Quiescent Current − µ A
Power-Down Quiescent Current − µ A
ZO− Single-Ended Output Impedance
in Powerdown − Ω
0.03
25
Quiescent Current − mA
SINGLE-ENDED OUTPUT
IMPEDANCE IN POWER-DOWN
vs
FREQUENCY
RL = 800 Ω
900
800
700
600
500
400
300
200
100
0
0
0.5 1
1.5 2
2.5 3
3.5 4
4.5 5
VS − Supply Voltage − ±V
Case Temperature − °C
Figure 49.
Figure 50.
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TYPICAL CHARACTERISTICS: 5 V
SMALL-SIGNAL UNITY-GAIN
FREQUENCY RESPONSE
SMALL-SIGNAL FREQUENCY
RESPONSE
1
0.2
22
Gain = 10, Rf = 5.1 kΩ
−2
Gain = 1
RL = 800 Ω
Rf = 392 Ω
PIN = −20 dBm
VS = 5 V
−3
0.1
1
16
Gain = 5, Rf = 2.4 kΩ
14
12
10
8
Gain = 2, Rf = 1 kΩ
6
4
2
10
100
0
−2
0.1
1000
f − Frequency − MHz
RL = 800 Ω
PIN = −30 dBm
VS = 5 V
1
0
Rf = 392 Ω
−0.1
−0.2
−0.3
Gain = 1
RL = 800 Ω
PIN = −20 dBm
VS = 5 V
−0.4
−0.5
10
100
10
1
1000
100
1000
f − Frequency − MHz
f − Frequency − MHz
Figure 51.
Figure 52.
Figure 53.
LARGE-SIGNAL FREQUENCY
RESPONSE
HARMONIC DISTORTION
vs
FREQUENCY
HARMONIC DISTORTION
vs
FREQUENCY
0
0
Single-Ended Input to
Differential Output
Gain = 1
RL = 800 Ω
Rf = 392 Ω
VO = 1 VPP
VS = 5 V
−10
Harmonic Distortion − dBc
0
−1
−2
Gain = 1
RL = 800 Ω
Rf = 392 Ω
PIN = 10 dBm
VS = 5 V
−3
−20
−30
−40
−50
−60
HD2
−70
−80
HD3
−90
−4
1
10
100
1000
−30
−40
−60
HD2
−70
−80
HD3
−100
0.1
Figure 55.
Figure 56.
HARMONIC DISTORTION
vs
FREQUENCY
HARMONIC DISTORTION
vs
FREQUENCY
HARMONIC DISTORTION
vs
OUTPUT VOLTAGE SWING
0
Single-Ended Input to
Differential Output
Gain = 1
RL = 800 Ω
Rf = 392 Ω
VO = 2 VPP
VS = 5 V
−10
−50
−60
−70
HD3
HD2
−80
−90
−100
0.1
−50
Figure 54.
Harmonic Distortion − dBc
−20
−40
f − Frequency − MHz
0
−10
−30
1
10
f − Frequency − MHz
−20
−30
−40
100
−60
HD3
HD2
−80
10
100
−100
0.1
−30
−40
−50
−60
HD3
−70
−80
−90
−100
f − Frequency − MHz
1
10
f − Frequency − MHz
Figure 57.
Figure 58.
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−20
HD2
−90
1
100
Single-Ended Input to
Differential Output
Gain = 1
RL = 800 Ω
Rf = 392 Ω
f= 8 MHz
VS = 5 V
−10
−50
−70
1
10
f − Frequency − MHz
0
Differential Input to
Differential Output
Gain = 1
RL = 800 Ω
Rf = 499 Ω
VO = 2 VPP
VS = 5 V
Harmonic Distortion − dBc
0.1
−20
−90
−100
0.1
Differential Input to
Differential Output
Gain = 1
RL = 800 Ω
Rf = 392 Ω
VO = 1 VPP
VS = 5 V
−10
Harmonic Distortion − dBc
1
Large Signal Gain − dB
0.1 dB Gain Flatness − dB
−1
−4
Harmonic Distortion − dBc
Rf = 499 Ω
0.1
18
0
Small Signal Gain − dB
Small Signal Unity Gain − dB
20
16
0.1-dB GAIN FLATNESS
FREQUENCY RESPONSE
100
0
0.5
1
1.5
2
2.5
3
VO − Output Voltage Swing − V
Figure 59.
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THS4501
SLOS350F – APRIL 2002 – REVISED OCTOBER 2011
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TYPICAL CHARACTERISTICS: 5 V (continued)
HARMONIC DISTORTION
vs
OUTPUT VOLTAGE SWING
HARMONIC DISTORTION
vs
OUTPUT VOLTAGE SWING
0
Differentia Input to
Differential Output
Gain = 1
RL = 800 Ω
Rf = 392 Ω
f= 8 MHz
VS = 5 V
−20
−30
−40
−50
−60
HD3
−70
−80
HD2
−90
−100
0
0.5
1
1.5
2
2.5
−20
−30
−40
HD2
−60
−70
−80
−20
−30
−40
−70
−80
−90
−100
1
1.5
2
2.5
3
0
HARMONIC DISTORTION
vs
FREQUENCY
HARMONIC DISTORTION
vs
FREQUENCY
HARMONIC DISTORTION
vs
FREQUENCY
HD2
−80
−20
−30
−40
−50
HD2
−60
−70
−80
−20
−30
−40
−50
−60
1
10
−80
HD3
1
10
f − Frequency − MHz
f − Frequency − MHz
100
HARMONIC DISTORTION
vs
FREQUENCY
HARMONIC DISTORTION
vs
OUTPUT VOLTAGE SWING
HARMONIC DISTORTION
vs
OUTPUT VOLTAGE SWING
0
0
Differential Input to
Differential Output
Gain = 2
RL = 800 Ω
Rf = 1 kΩ
VO = 2 VPP
VS = 5 V
HD3
−70
HD2
−80
−90
1
Single-Ended Input to
Differential Output
Gain = 2
RL = 800 Ω
Rf = 1 kΩ
f = 8 MHz
VS = 5 V
−10
−60
−100
0.1
1
10
f − Frequency − MHz
Figure 65.
10
−20
−30
−40
−50
−60
HD3
−70
−80
HD2
20
30
40
50
60
80
90
−100
100
0.5
1
1.5
2
2.5
3
HD3
70
−90
0
100
Differential Input to
Differential Output
Gain = 2
RL = 800 W
RF = 1 kW
f = 8 MHz
VS = 5 V
10
Harmonic Distortion - dBc
−50
−100
0.1
Figure 64.
Harmonic Distortion − dBc
−40
100
Figure 63.
0
−30
HD2
−90
−100
0.1
100
HD3
−70
−90
−100
Single-Ended Input to
Differential Output
Gain = 2
RL = 800 Ω
Rf = 1 kΩ
VO = 2 VPP
VS = 5 V
−10
Harmonic Distortion − dBc
−70
3
0
Differential Input to
Differential Output
Gain = 2
RL = 800 Ω
Rf = 1 kΩ
VO = 1 VPP
VS = 5 V
−10
HD3
−20
2.5
Figure 62.
−60
−10
2
Figure 61.
−50
0.1
1.5
Figure 60.
Harmonic Distortion − dBc
−40
1
VO − Output Voltage Swing − V
0
−30
0.5
VO − Output Voltage Swing − V
Single-Ended Input to
Differential Output
Gain = 2
RL = 800 Ω
Rf = 1 kΩ
VO = 1 VPP
VS = 5 V
−20
HD2
−60
−100
0.5
HD3
−50
−90
0
−90
Harmonic Distortion − dBc
HD3
−50
3
Differentia Input to
Differential Output
Gain = 1
RL = 800 Ω
Rf = 392 Ω
f= 30 MHz
VS = 5 V
−10
VO − Output Voltage Swing − V
0
−10
Harmonic Distortion − dBc
0
Single-Ended Input to
Differential Output
Gain = 1
RL = 800 Ω
Rf = 392 Ω
f = 30 MHz
VS = 5 V
−10
Harmonic Distortion − dBc
Harmonic Distortion − dBc
−10
Harmonic Distortion − dBc
0
HARMONIC DISTORTION
vs
OUTPUT VOLTAGE SWING
HD2
0
0.5
1
1.5
2
2.5
f − Frequency − MHz
VO − Output Voltage Swing − V
VO - Output Voltage Swing - V
Figure 66.
Figure 67.
Figure 68.
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TYPICAL CHARACTERISTICS: 5 V (continued)
HARMONIC DISTORTION
vs
OUTPUT VOLTAGE SWING
HARMONIC DISTORTION
vs
OUTPUT VOLTAGE SWING
−40
−50
HD2
−60
HD3
−70
−80
30
40
HD2
50
−20
60
HD3
70
80
−30
−40
−60
−80
−90
−100
100
−100
1
1.5
2
2.5
3
2.5
0
3
400
800
1200
1600
RL − Load Resistance − Ω
Figure 70.
Figure 71.
HARMONIC DISTORTION
vs
LOAD RESISTANCE
THIRD-ORDER INTERMODULATION
DISTORTION
vs
FREQUENCY
THIRD-ORDER OUTPUT INTERCEPT
POINT
vs
FREQUENCY
HD2
-60
HD3
-70
-80
-90
0
2
Figure 69.
-50
400
800
1200
1600
−50
Single-Ended Input to
Differential Output
Gain = 1
VO = 2 VPP
Rf = 392 Ω
RL = 800 Ω
VS = 5 V
−60
−70
−80
−90
−100
10
55
Gain = 1
VO = 2 VPP
RF = 392 W
RL = 800 W
VS = 5 V
50
45
40
35
30
0
100
20
40
60
80
100
120
RL - Load Resistance - W
f − Frequency − MHz
Figure 72.
Figure 73.
Figure 74.
SLEW RATE
vs
DIFFERENTIAL OUTPUT VOLTAGE
STEP
LARGE-SIGNAL TRANSIENT
RESPONSE
SMALL-SIGNAL TRANSIENT
RESPONSE
1400
1200
1000
VO − Output Voltage − V
Gain = 1
RL = 800 Ω
Rf = 392 Ω
VS = 5 V
800
600
400
200
f - Frequency - MHz
2
0.4
1.5
0.3
VO − Output Voltage − V
-100
1.5
Third-Order Output Intercept Point - dBm
-40
1
VO - Output Voltage Swing - V
Differential Input to
Differential Output
Gain = 1
VO = 2 VPP
RF = 392 W
f = 30 MHz
VS = 5 V
-30
0.5
VO − Output Voltage Swing − V
0
-20
0
HD3
−70
90
0.5
HD2
−50
−90
-10
Harmonic Distortion - dBc
20
Single-Ended Input to
Differential Output
Gain = 1
VO = 2 VPP
Rf = 392 Ω
f= 30 MHz
VS = 5 V
−10
Harmonic Distortion − dBc
−30
Third-Order Intermodulation Distortion − dBc
Harmonic Distortion − dBc
−20
Differential Input to
Differential Output
Gain = 2
RL = 800 W
RF = 1 kW
f = 30 MHz
VS = 5 V
10
Harmonic Distortion - dBc
Single-Ended Input to
Differential Output
Gain = 2
RL = 800 Ω
Rf = 1 kΩ
f = 30 MHz
VS = 5 V
0
SR − Slew Rate − V/ µ s
0
0
0
−10
1
Gain = 1
RL = 800 Ω
Rf = 392 Ω
tr/tf = 300 ps
VS = 5 V
0.5
0
−0.5
−1
0
0.5
1
1.5
2
2.5
VO − Differential Output Voltage Step − V
Figure 75.
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3
0.2
Gain = 1
RL = 800 Ω
Rf = 392 Ω
tr/tf = 300 ps
VS = 5 V
0.1
0
−0.1
−0.2
−0.3
−1.5
0
18
HARMONIC DISTORTION
vs
LOAD RESISTANCE
−2
−100
0
100
200
300
t − Time − ns
400
500
−0.4
−100
0
100
200
300
400
500
t − Time − ns
Figure 76.
Figure 77.
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TYPICAL CHARACTERISTICS: 5 V (continued)
REJECTION RATIOS
vs
FREQUENCY
VOLTAGE AND CURRENT NOISE
vs
FREQUENCY
90
60
50
CMMR
PSRR−
40
30
20
In
10
0.1
1
10
100
1000
0.1
1
10
f − Frequency − MHz
f − Frequency − kHz
OPEN-LOOP GAIN AND PHASE
vs
FREQUENCY
OPEN-LOOP GAIN
vs
CASE TEMPERATURE
60
−60
Phase
−90
Open-Loop Gain − dB
30
Phase − °
Open-Loop Gain − dB
54
53
52
51
50
49
48
−120
47
1
10
f − Frequency − MHz
0
0.01
100
0.1
1
10
46
−150
1000
100
−40−30−20−100 10 20 30 40 50 60 70 80 90
Case Temperature − °C
f − Frequency − MHz
Figure 81.
Figure 82.
Figure 83.
INPUT BIAS AND OFFSET
CURRENT
vs
CASE TEMPERATURE
QUIESCENT CURRENT
vs
SUPPLY VOLTAGE
INPUT OFFSET VOLTAGE
vs
CASE TEMPERATURE
3.75
0
VS = 5 V
35
4
TA = 85°C
IIB+
I OS − Input Offset Current − µ A
−0.01
30
IIB−
−0.03
2.75
−0.04
2.5
−0.05
IOS
Quiescent Current − mA
−0.02
−0.06
2
−0.07
1.75
−0.08
1.5
−0.09
TA = 25°C
VOS − Input Offset Voltage − mV
Output Balance Error − dB
55
−30
20
RL = 800 Ω
VS = 5 V
56
0
40
10
−70
0.1
I IB − Input Bias Current − µ A
PIN = −30 dBm
RL = 800 Ω
VS = 5 V
50
−60
57
30
Gain
−50
2.25
Case Temperature − °C
OUTPUT BALANCE ERROR
vs
FREQUENCY
−40
3
100
Figure 80.
−30
3.25
RL = 800 Ω
VS = 5 V
Figure 79.
PIN = −20 dBm
RL = 800 Ω
Rf = 499 Ω
VS = 5 V
−20
40
Figure 78.
0
−10
PSRR+
60
0
−40−30−20−10 0 10 20 30 40 50 60 70 80 90
−10
10 k
PSRR−
80
20
RL = 800 Ω
VS = 5 V
0
1
0.01
CMMR
100
70
Rejection Ratios − dB
Hz
I n − Current Noise − pA/
Vn
10
120
PSRR+
80
Rejection Ratios − dB
Hz
Vn − Voltage Noise − nV/
100
3.5
REJECTION RATIOS
vs
CASE TEMPERATURE
25
TA = −40°C
20
15
10
5
−0.1
1.25
−40−30−20−10 0 10 20 30 40 50 60 70 80 90
0
Case Temperature − °C
Figure 84.
0 0.5
1
1.5
2
2.5
3
3.5
4 4.5
5
3.5
VS = 5 V
3
2.5
2
1.5
1
0.5
0
−40 −30−20−10 0 10 20 30 40 50 60 70 80 90
VS − Supply Voltage − ±V
Case Temperature − °C
Figure 85.
Figure 86.
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TYPICAL CHARACTERISTICS: 5 V (continued)
0
150
VS = 5 V
VS = 5 V
90
Source
100
Output Drive − mA
80
70
60
50
40
30
50
0
−50
20
−100
10
0
−10
0
1
2
3
4
Sink
5
−20
−30
−40
−50
HD3-SE
and Diff
−60
−70
−80
−90
HD2-SE
1 1.25 1.5 1.75 2 2.25 2.5 2.75 3 3.25 3.5
Input Common-Mode Range − V
Case Temperature − °C
VOCM − Output Common-Mode Voltage − V
Figure 87.
Figure 88.
Figure 89.
SMALL-SIGNAL FREQUENCY
RESPONSE AT VOCM
OUTPUT OFFSET VOLTAGE
vs
OUTPUT COMMON-MODE
VOLTAGE
QUIESCENT CURRENT
vs
POWER-DOWN VOLTAGE
25
800
4
2
1
VS = 5 V
600
Quiescent Current − mA
Gain = 1
RL = 800 Ω
Rf = 392 Ω
PIN= −20 dBm
VS = 5 V
3
HD2-Diff
−100
−150
−40−30−20−10 0 10 20 30 40 50 60 70 80 90
VOS − Output Offset Voltage − mV
Small Signal Frequency Response at VOCM − dB
−1
Single-Ended and
Differential Input
Gain = 1
VO = 2 VPP
Rf = 392 Ω
f= 8 MHz, VS = 5 V
−10
Harmonic Distortion − dBc
CMRR − Common-Mode Rejection Ratio − dB
110
100
HARMONIC DISTORTION
vs
OUTPUT COMMON-MODE
VOLTAGE
OUTPUT DRIVE
vs
CASE TEMPERATURE
COMMON-MODE REJECTION RATIO
vs
INPUT COMMON-MODE RANGE
400
200
0
−200
0
−400
−1
20
15
10
5
−600
−2
−800
0
0 0.5
−3
0.1
1
10
100
1000
1
1.5
2
2.5
3
3.5
4
4.5
5
0 0.25 0.5 0.75 1 1.25 1.5 1.75 2 2.25 2.5
Power-down Voltage − V
VOC − Output Common-Mode Voltage − V
f − Frequency − MHz
Figure 90.
Figure 91.
Figure 92.
TURN-ON AND TURN-OFF DELAY
TIME
SINGLE-ENDED OUTPUT
IMPEDANCE IN POWER-DOWN
vs
FREQUENCY
POWER-DOWN QUIESCENT
CURRENT
vs
CASE TEMPERATURE
−2
−3
−4
−6
0 0.5 1 1.5 2 2.5 3 100.5 101
t − Time − ms
102
103
Figure 93.
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Power-Down Quiescent Current − µ A
0
−1
ZO− Single-Ended Output Impedance
in Power Down − Ω
0
Quiescent Current − mA
Power-Down Voltage Signal − V
0.01
−5
20
1000
0.02
Current
800
1100
0.03
900
800
700
600
500
400
300
200
100
0
0.1
Gain = 1
RL = 400 Ω
Rf = 499 Ω
PIN = −1 dBm
VS = 5 V
1
10
100
f − Frequency − MHz
1000
700
RL = 800 Ω
VS = 5 V
600
500
400
300
200
100
0
−40 −30−20−10 0 10 20 30 40 50 60 70 80 90
Case Temperature − °C
Figure 94.
Figure 95.
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Product Folder Link(s): THS4500 THS4501
THS4500
THS4501
SLOS350F – APRIL 2002 – REVISED OCTOBER 2011
www.ti.com
TYPICAL CHARACTERISTICS: 5 V (continued)
POWER-DOWN QUIESCENT CURRENT
vs
SUPPLY VOLTAGE
Power-Down Quiescent Current − µ A
1000
900
800
700
600
500
400
300
200
100
0
0
0.5 1
1.5 2
2.5 3
3.5 4
4.5 5
VS − Supply Voltage − V
Figure 96.
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APPLICATION INFORMATION
FULLY DIFFERENTIAL AMPLIFIERS
FULLY DIFFERENTIAL AMPLIFIER
TERMINAL FUNCTIONS
Differential signaling offers a number of performance
advantages in high-speed analog signal processing
systems,
including
immunity
to
external
common-mode noise, suppression of even-order
nonlinearities, and increased dynamic range. Fully
differential amplifiers not only serve as the primary
means of providing gain to a differential signal chain,
but also provide a monolithic solution for converting
single-ended signals into differential signals for
easier, higher performance processing. The THS4500
family of amplifiers contains products in Texas
Instruments' expanding line of high-performance, fully
differential amplifiers. Information on fully differential
amplifier fundamentals, as well as implementation
specific information, is presented in the Applications
Section of this data sheet to provide a better
understanding of the operation of the THS4500 family
of devices, and to simplify the design process for
designs using these amplifiers.
Fully differential amplifiers are typically packaged in
eight-pin packages, as shown in Figure 97. The
device pins include two inputs (VIN+, VIN–), two
outputs (VOUT–, VOUT+), two power supplies (VS+, VS–),
an output common-mode control pin (VOCM), and an
optional power-down pin (PD).
APPLICATIONS SECTION
A standard configuration for the device is shown in
Figure 97. The functionality of a fully differential
amplifier can be imagined as two inverting amplifiers
that share a common noninverting terminal (though
the voltage is not necessarily fixed). For more
information on the basic theory of operation for fully
differential amplifiers, refer to the Texas Instruments
application note Fully Differential Amplifiers, literature
number SLOA054
, available for download
at www.ti.com.
•
•
•
•
•
•
•
•
•
•
•
•
•
•
22
Fully Differential Amplifier Terminal Functions
Input Common-Mode Voltage Range and the
THS4500 Family
Choosing the Proper Value for the Feedback and
Gain Resistors
Application Circuits Using Fully Differential
Amplifiers
Key Design Considerations for Interfacing to an
Analog-to-Digital Converter
Setting the Output Common-Mode Voltage With
the VOCM Input
Saving Power with Power-Down Functionality
Linearity:
Definitions,
Terminology,
Circuit
Techniques, and Design Tradeoffs
An Abbreviated Analysis of Noise in Fully
Differential Amplifiers
Printed-Circuit Board Layout Techniques for
Optimal Performance
Power Dissipation and Thermal Considerations
Power Supply Decoupling Techniques and
Recommendations
Evaluation
Fixtures,
Spice
Models,
and
Applications Support
Additional Reference Material
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VIN- 1
8 VIN+
VOCM 2
7 PD
VS+ 3
6 VS-
VOUT+ 4
5 VOUT-
Figure 97. Fully Differential Amplifier Pin Diagram
INPUT COMMON-MODE VOLTAGE RANGE
AND THE THS4500 FAMILY
The key difference between the THS4500/1 and the
THS4502/3 is the input common-mode range for the
four devices. The THS4502 and THS4503 have an
input common-mode range that is centered around
midrail, and the THS4500 and THS4501 have an
input common-mode range that is shifted to include
the negative power-supply rail. Selection of one or
the other amplifier is determined by the nature of the
application. Specifically, the THS4500 and THS4501
are designed for use in single-supply applications
where the input signal is ground-referenced, as
depicted in Figure 98. The THS4502 and THS4503
are designed for use in single-supply or split-supply
applications where the input signal is centered
between the power-supply voltages, as depicted in
Figure 99.
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RF1
RG1
RS
+VS
RT
VS
V OUT) +
V OUT–
VOCM
+ - +
RG2
RF2
Figure 98. Application Circuit for the THS4500
and THS4501, Featuring Single-Supply Operation
With a Ground-Reference Input Signal
VS
VOCM
Where:
RG
β+
RF ) RG
(4)
V P + V IN)(1–β) ) V OUT–β
NOTE: The equations denote the device inputs as VN and VP, and
the circuit inputs as VIN+ and VIN–.
(5)
RF
RG
VIN+
VP
+ - +
VOCM
VN
-VS
RG2
(2)
(3)
+VS
RT
(1)
–V IN)(1–β) ) V IN–(1–β) ) 2V OCMβ
+
2β
V N + V IN–(1–β) ) V OUT)β
RF1
RG1
RS
V IN)(1–β)–V IN–(1–β) ) 2V OCMβ
2β
+ - +
VOUTVOUT+
VINRG
RF2
Figure 99. Application Circuit for the THS4500
and THS4501, Featuring Split-Supply Operation
With an Input Signal Referenced at the Midrail
Equation 1 through Equation 5 are used to calculate
the required input common-mode range for a given
set of input conditions.
The equations allow calculation of the input
common-mode range requirements, given information
about the input signal, the output voltage swing, the
gain, and the output common-mode voltage.
Calculating the maximum and minimum voltage
required for VN and VP (the amplifier input nodes)
determines whether or not the input common-mode
range is violated or not. Four equations are required:
two calculate the output voltages and two calculate
the node voltages at VN and VP (note that only one of
these nodes needs calculation, because the amplifier
forces a virtual short between the two nodes).
RF
Figure 100. Diagram For Input Common-Mode
Range Equations
Table 1 and Table 2 depict the input common-mode
range requirements for two different input scenarios,
an input referenced around the negative rail and an
input referenced around midrail. The tables highlight
the differing requirements on input common-mode
range, and illustrate the reasoning to choose either
the THS4500/1 or the THS4502/3. For signals
referenced around the negative power supply, the
THS4500/1 should be chosen because its input
common-mode range includes the negative supply
rail. For all other situations, the THS4502/3 offers
slightly improved distortion and noise performance for
applications with input signals centered between the
power-supply rails.
Table 1. Negative-Rail Referenced
Gain
(V/V)
VIN+
(V)
VIN–
(V)
VIN
(VPP)
VOCM
(V)
VOD
(VPP)
VNMIN
(V)
VNMAX
(V)
1
–2.0 to 2.0
0
4
2.5
4
0.75
1.75
2
–1.0 to 1.0
0
2
2.5
4
0.5
1.167
4
–0.5 to 0.5
0
1
2.5
4
0.3
0.7
8
–0.25 to 0.25
0
0.5
2.5
4
0.167
0.389
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Table 2. Midrail Referenced
Gain
(V/V)
VIN+
(V)
VIN–
(V)
VIN
(VPP)
VOCM
(V)
VOD
(VPP)
VNMIN
(V)
VNMAX
(V)
1
0.5 to 4.5
2.5
4
2.5
4
2
3
2
1.5 to 3.5
2.5
2
2.5
4
2.16
2.83
4
2.0 to 3.0
2.5
1
2.5
4
2.3
2.7
8
2.25 to 2.75
2.5
0.5
2.5
4
2.389
2.61
CHOOSING THE PROPER VALUE FOR THE
FEEDBACK AND GAIN RESISTORS
The selection of feedback and gain resistors impacts
circuit performance in a number of ways. The values
presented in this section provide the optimum
high-frequency performance (lowest distortion, flat
frequency response). Since the THS4500 family of
amplifiers is developed with a voltage feedback
architecture, the choice of resistor values does not
have a dominant effect on bandwidth, unlike a
current-feedback amplifier. However, resistor choices
do have second-order effects. For optimal
performance, the following feedback resistor values
are recommended. In higher gain configurations (gain
greater than two), the feedback resistor values have
much less effect on the high-frequency performance.
Example feedback and gain resistor values are given
in the section on basic design considerations
(Table 3).
Amplifier loading, noise, and the flatness of the
frequency response are three design parameters that
should be considered when selecting feedback
resistors. Larger resistor values contribute more noise
and can induce peaking in the ac response in low
gain configurations; smaller resistor values can load
the amplifier more heavily, resulting in a reduction in
distortion performance. In addition, feedback resistor
values, coupled with gain requirements, determine
the value of the gain resistors and directly impact the
input impedance of the entire circuit. While there are
no strict rules about resistor selection, these trends
can provide qualitative design guidance.
APPLICATION CIRCUITS USING FULLY
DIFFERENTIAL AMPLIFIERS
Fully differential amplifiers provide designers with a
great deal of flexibility in a wide variety of
applications. This section provides an overview of
some common circuit configurations and gives some
design guidelines. Designing the interface to an
analog-to-digital converter (ADC), driving lines
differentially, and filtering with fully differential
amplifiers are a few of the circuits that are covered.
BASIC DESIGN CONSIDERATIONS
Table 3. Resistor Values for Balanced Operation
in Various Gain Configurations
Gain
R2 and R4
(Ω)
R1 (Ω)
R3 (Ω)
RT (Ω)
1
392
412
383
54.9
1
499
523
487
53.6
2
392
215
187
60.4
2
1.3 k
665
634
52.3
5
1.3 k
274
249
56.2
5
3.32 k
681
649
52.3
10
1.3 k
147
118
64.9
10
6.81 k
698
681
52.3
ǒ Ǔ
VOD
VIN
R2
R1
Vn
Vout+
R3
RS
+
+
-
VP
VS
VoutVOCM
RT
R4
Figure 101. Diagram for Design Calculations
Equations for calculating fully differential amplifier
resistor values in order to obtain balanced operation
in the presence of a 50-Ω source impedance are
given in Equation 6 through Equation 9.
1
K + R2 R2 + R4
RT +
R1
K
1–
2(1)K)
1 –
RS
R3
R3 + R1 * ǒRs || R TǓ
(6)
β1 +
R3 ) RT || R S
R1
β +
R1 ) R2 2
R3 ) RT || R S ) R4
ǒ Ǔǒ
1–β
Ǔ
+ 2ǒ
β )β
V OD
1–β 2
+2
β1 ) β 2
VS
V OD
V IN
RT
RT ) RS
Ǔ
(7)
(8)
2
1
2
(9)
The circuits in Figure 98 through Figure 101 are used
to highlight basic design considerations for fully
differential amplifier circuit designs.
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For more detailed information about balance in fully
differential amplifiers, see the application report, Fully
Differential Amplifiers (SLOA054), referenced at the
end of this data sheet.
INTERFACING TO AN ANALOG-TO-DIGITAL
CONVERTER
The THS4500 family of amplifiers are designed
specifically
to
interface
to
today's
highest-performance ADCs. This section highlights
the key concerns when interfacing to an ADC and
provides example interface circuits.
There are several key design concerns when
interfacing to an analog-to-digital converter:
• Terminate the input source properly. In
high-frequency receiver chains, the source that
feeds the fully differential amplifier requires a
specific load impedance (that is, 50 Ω).
• Design a symmetric printed circuit board (PCB)
layout. Even-order distortion products are heavily
influenced by layout, and careful attention to a
symmetric layout minimizes these distortion
products.
• Minimize inductance in power-supply decoupling
traces and components. Poor power-supply
decoupling can have a dramatic effect on circuit
performance. Since the outputs are differential,
differential currents exist in the power-supply pins.
Thus, decoupling capacitors should be placed in a
manner that minimizes the impedance of the
current loop.
• Use separate analog and digital power supplies
and grounds. Noise (bounce) in the power
supplies (created by digital switching currents) can
couple directly into the signal path, and
power-supply noise can create higher distortion
products as well.
• Use care when filtering. While an RC low-pass
filter may be desirable on the output of the
amplifier to filter broadband noise, the excess
loading can negatively impact the amplifier
linearity. Filtering in the feedback path does not
have this effect.
• AC-coupling allows easier circuit design. If
dc-coupling is required, be aware of the excess
power dissipation that can occur due to
level-shifting the output through the output
common-mode voltage control.
• Do not terminate the output unless required. Many
open-loop, class-A amplifiers require 50-Ω
termination for proper operation, but closed-loop
fully differential amplifiers drive a specific output
•
•
•
•
•
voltage regardless of the load impedance present.
Terminating the output of a fully differential
amplifier with a heavy load adversely affects the
amplifier linearity.
Comprehend the VOCM input drive requirements.
Determine if the ADC voltage reference can
provide the required amount of current to move
VOCM to the desired value. A buffer may be
needed.
Decouple the VOCM pin to eliminate the antenna
effect. VOCM is a high-impedance node that can
act as an antenna. A large decoupling capacitor
on this node eliminates this problem.
Know the input common-mode range. If the input
signal is referenced around the negative
power-supply rail (for example, around ground on
a single 5 V supply), then the THS4500/1
accommodates the input signal. If the input signal
is referenced around midrail, choose the
THS4502/3 for the best operation.
Packaging makes a difference at higher
frequencies. If possible, choose the smaller,
thermally-enhanced MSOP package for the best
performance. As a rule, lower junction
temperatures provide better performance. If
possible, use a thermally-enhanced package,
even if the power dissipation is relatively small
compared to the maximum power dissipation
rating to achieve the best results.
Understand the effect of the load impedance seen
by the fully differential amplifier when performing
system-level intercept point calculations. Lighter
loads (such as those presented by an ADC) allow
smaller intercept points to support the same level
of intermodulation distortion performance.
EXAMPLE ANALOG-TO-DIGITAL
CONVERTER DRIVER CIRCUITS
The THS4500 family of devices is designed to drive
high-performance ADCs with extremely high linearity,
allowing for the maximum effective number of bits at
the output of the data converter. Two representative
circuits shown below highlight single-supply operation
and split supply operation, respectively. Specific
feedback resistor, gain resistor, and feedback
capacitor values are not shown, as these values
depend on the frequency of interest. Information on
calculating these values can be found in the
applications material above.
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CF
RS
VS
RG
CG
RS
RF
RF
15 V
RT
VS
5V
RT
RG
RISO
+
VOCM
0.1 mF
+
VOCM
+
1 mF
THS4503
RL
+
-
0.1 mF
RISO
RISO
CS
VDD
THS4500/2
5V
10 mF
-
RISO
IN ADS5410
12-Bit/80 MSPS
IN
CM
RF
RG
CS
VOD = 26 VPP
CG
5 V
RG
10 mF 0.1 mF
0.1 mF
RF
Figure 104. Fully Differential Line Driver With
High Output Swing
CF
Figure 102. Using the THS4503 With the ADS5410
CF
RS
VS
RG
RF
5V
RT
5V
10 mF
0.1 mF
+
VOCM
+
1 mF
RISO
IN ADS5421
14-Bit/40 MSPS
IN
CM
THS4501
RISO
RG
RF
CF
FILTERING WITH FULLY DIFFERENTIAL
AMPLIFIERS
Similar to single-ended counterparts, fully differential
amplifiers have the ability to couple filtering
functionality with voltage gain. Numerous filter
topologies can be based on fully differential
amplifiers. Several of these are outlined in the
application report A Differential Circuit Collection
(literature number SLOA064), referenced at the end
of this data sheet. The circuit below depicts a simple,
two-pole, low-pass filter applicable to many different
types of systems. The first pole is set by the resistors
and capacitors in the feedback paths, and the second
pole is set by the isolation resistors and the capacitor
across the outputs of the isolation resistors.
0.1 mF
CF1
Figure 103. Using the THS4501 With the ADS5421
FULLY DIFFERENTIAL LINE DRIVERS
RG1
RS
RF1
RISO
RT
VS
+
-
The THS4500 family of amplifiers can be used as
high-frequency, high-swing differential line drivers.
The high power-supply voltage rating (16.5 V
absolute maximum) allows operation on a single 12-V
or a single 15-V supply. The high supply voltage,
coupled with the ability to provide differential outputs,
enables the ability to drive 26 VPP into reasonably
heavy loads (250 Ω or greater). The circuit in
Figure 104 illustrates the THS4500 family of devices
used as high-speed line drivers. For line driver
applications, close attention must be paid to thermal
design constraints because of the typically high level
of power dissipation.
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C
VO
+
RG2
RISO
RF2
CF2
Figure 105. A Two-Pole, Low-Pass Filter Design
Using a Fully Differential Amplifier With Poles
Located at: P1 = (2πRFCF)–1 in Hz and
P2 = (4πRISOC)–1 in Hz
Often, filters like these are used to eliminate
broadband noise and out-of-band distortion products
in signal acquisition systems. It should be noted that
the increased load placed on the output of the
amplifier by the second low-pass filter has a
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detrimental effect on the distortion performance. The
preferred method of filtering is to use the feedback
network, as the typically smaller capacitances
required at these points in the circuit do not load the
amplifier nearly as heavily in the passband.
SETTING THE OUTPUT COMMON-MODE
VOLTAGE WITH THE VOCM INPUT
The output common-mode voltage pin provides a
critical function to the fully differential amplifier; it
accepts an input voltage and reproduces that input
voltage as the output common-mode voltage. In other
words, the VOCM input provides the ability to level-shift
the outputs to any voltage inside the output voltage
swing of the amplifier.
A description of the input circuitry of the VOCM pin is
shown in Figure 106 to facilitate an easier
understanding of the VOCM interface requirements.
The VOCM pin has two 50-kΩ resistors between the
power supply rails to set the default output
common-mode voltage to midrail. A voltage applied to
the VOCM pin alters the output common-mode voltage
as long as the source has the ability to provide
enough current to overdrive the two 50-kΩ resistors.
This phenomenon is depicted in the VOCM equivalent
circuit diagram. Current drive is especially important
when using the reference voltage of an
analog-to-digital converter to drive VOCM. Output
current drive capabilities differ from part to part, so a
voltage buffer may be necessary in some
applications.
capacitance is a reasonable value for eliminating a
great deal of broadband interference, but additional,
tuned decoupling capacitors should be considered if a
specific source of electromagnetic or radio frequency
interference is present elsewhere in the system.
Information on the ac performance (bandwidth, slew
rate) of the VOCM circuitry is included in the Electrical
Characteristics and Typical Characterisitcs sections.
Since the VOCM pin provides the ability to set an
output common-mode voltage, the ability for
increased power dissipation exists. While this
possibility does not pose a performance problem for
the amplifier, it can cause additional power
dissipation of which the system designer should be
aware. The circuit shown in Figure 107 demonstrates
an example of this phenomenon. For a device
operating on a single 5-V supply with an input signal
referenced around ground and an output
common-mode voltage of 2.5 V, a dc potential exists
between the outputs and the inputs of the device. The
amplifier sources current into the feedback network in
order to provide the circuit with the proper operating
point. While there are no serious effects on the circuit
performance, the extra power dissipation may need to
be included in the system power budget.
I1 =
DC Current Path to Ground
5V
RT
VOCM = 2.5 V
R = 50 kW
+ -
RL
+
2 VOCM - V S+ - VSR
RG2
IIN
2.5-V DC
-
IIN =
VOCM
RF1
RG1
RS
VS
VS+
VOCM
RF1+ RG1 + RS || RT
RF2
2.5-V DC
R = 50 kW
DC Current Path to Ground
I2 =
VS-
Figure 106. Equivalent Input Circuit for VOCM
By design, the input signal applied to the VOCM pin
propagates to the outputs as a common-mode signal.
As shown in Figure 106, the VOCM input has a high
impedance associated with it, dictated by the two
50-kΩ resistors. While the high impedance allows for
relaxed drive requirements, it also allows the pin and
any associated PCB traces to act as an antenna. For
this reason, a decoupling capacitor is recommended
on this node for the sole purpose of filtering any
high-frequency noise that could couple into the signal
path through the VOCM circuitry. A 0.1-μF or 1-μF
VOCM
RF2 + RG2
Figure 107. Depiction of DC Power Dissipation
Caused By Output Level-Shifting in a DC-Coupled
Circuit
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The THS4500 family of fully differential amplifiers
contains devices that come with and without the
power-down option. Even-numbered devices have
power-down capability, which is described in detail
here.
The power-down pin of the amplifiers defaults to the
positive supply voltage in the absence of an applied
voltage (that is, an internal pull-up resistor is present),
putting the amplifier in the power-on mode of
operation. To turn off the amplifier in an effort to
conserve power, the power-down pin can be driven
towards the negative rail. The threshold voltages for
power-on and power-down are relative to the supply
rails and given in the specification tables. Above the
enable threshold voltage, the device is on. Below the
disable threshold voltage, the device is off. Behavior
between these threshold voltages is not specified.
Note that this power-down functionality is just that;
the amplifier consumes less power in power-down
mode. The power-down mode is not intended to
provide a high-impedance output. In other words, the
power-down functionality is not intended to allow use
as a 3-state bus driver. When in power-down mode,
the impedance looking back into the output of the
amplifier is dominated by the feedback and gain
setting resistors.
The time delays associated with turning the device on
and off are specified as the time it takes for the
amplifier to reach 50% of the nominal quiescent
current. The time delays are on the order of
microseconds because the amplifier moves in and out
of the linear mode of operation in these transitions.
LINEARITY: DEFINITIONS, TERMINOLOGY,
CIRCUIT TECHNIQUES, AND DESIGN
TRADEOFFS
The
THS4500
family
of
devices
features
unprecedented distortion performance for monolithic
fully differential amplifiers. This section focuses on
the fundamentals of distortion, circuit techniques for
reducing nonlinearity, and methods for equating
distortion of fully differential amplifiers to desired
linearity specifications in RF receiver chains.
Amplifiers are generally thought of as linear devices.
In other words, the output of an amplifier is a linearly
scaled version of the input signal applied to it. In
reality, however, amplifier transfer functions are
nonlinear. Minimizing amplifier nonlinearity is a
primary design goal in many applications.
Intercept points are specifications that have long
been used as key design criteria in the RF
communications world as a metric for the
intermodulation distortion performance of a device in
the signal chain (for example, amplifiers, mixers,
28
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etc.). Use of the intercept point, rather than strictly the
intermodulation distortion, allows for simpler
system-level calculations. Intercept points, like noise
figures, can be easily cascaded back and forth
through a signal chain to determine the overall
receiver chain intermodulation distortion performance.
The relationship between intermodulation distortion
and intercept point is depicted in Figure 108 and
Figure 109.
PO
PO
∆fc = fc − f1
Power
SAVING POWER WITH POWER-DOWN
FUNCTIONALITY
∆fc = f2 − fc
IMD3 = PS − PO
PS
fc − 3∆f
PS
f1 fc
f2
fc + 3∆f
f − Frequency − MHz
Figure 108. 2-Tone and 3rd-Order
Intermodulation Products
POUT
(dBm)
1X
OIP3
PO
IMD3
IIP3
3X
PIN
(dBm)
PS
Figure 109. Graphical Representation of 2-Tone
and 3rd-Order Intercept Point
Due to the intercept point ease-of-use in system level
calculations for receiver chains, it has become the
specification of choice for guiding distortion-related
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design decisions. Traditionally, these systems use
primarily class-A, single-ended RF amplifiers as gain
blocks. These RF amplifiers are typically designed to
operate in a 50-Ω environment, just like the rest of
the receiver chain. Since intercept points are given in
dBm, this implies an associated impedance (50 Ω).
However, with a fully differential amplifier, the output
does not require termination as an RF amplifier
would. Because closed-loop amplifiers deliver signals
to the outputs regardless of the impedance present, it
is important to comprehend this feature when
evaluating the intercept point of a fully differential
amplifier. The THS4500 series of devices yields
optimum distortion performance when loaded with
200 Ω to 1 kΩ, very similar to the input impedance of
an analog-to-digital converter over its input frequency
band. As a result, terminating the input of the ADC to
50 Ω can actually be detrimental to system
performance.
This discontinuity between open-loop, class-A
amplifiers and closed-loop, class-AB amplifiers
becomes apparent when comparing the intercept
points of the two types of devices. Equation 10 gives
the definition of an intercept point, relative to the
intermodulation distortion.
ŤIMD 3Ť
OIP 3 + P O )
where
2
(10)
ǒ
ǒ
P O + 10 log
Ǔ
Ǔ
V 2Pdiff
2RL 0.001
NOTE: Po is the output power of a single tone, RL is the
differential load resistance, and VP(diff) is the differential
peak voltage for a single tone.
(11)
As can be seen in the equations, when a higher
impedance is used, the same level of intermodulation
distortion performance results in a lower intercept
point. Therefore, it is important to understand the
impedance seen by the output of the fully differential
amplifier when selecting a minimum intercept point.
Figure 110 shows the relationship between the strict
definition of an intercept point with a normalized, or
equivalent, intercept point for the THS4500.
OIP 3 − Third-Order Output Intercept Point − dBm
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60
Normalized to 200 Ω
55
Normalized to 50 Ω
50
45
40
35
OIP3 RL= 800 Ω
30
Gain = 1
Rf = 392 Ω
VS = ± 5 V
Tone Spacing = 200 kHz
25
20
15
0
10 20 30 40 50 60 70 80 90 100
f − Frequency − MHz
Figure 110. Equivalent 3rd-Order Intercept Point
for the THS4500
Comparing specifications between different device
types becomes easier when a common impedance
level is assumed. For this reason, the intercept points
on the THS4500 family of devices are reported
normalized to a 50-Ω load impedance.
AN ANALYSIS OF NOISE IN FULLY
DIFFERENTIAL AMPLIFIERS
Noise analysis in fully differential amplifiers is
analogous to noise analysis in single-ended
amplifiers; the same concepts apply. Figure 111
shows a generic circuit diagram consisting of a
voltage source, a termination resistor, two gain
setting resistors, two feedback resistors, and a fully
differential amplifier is shown, including all the
relevant noise sources. From this circuit, the noise
factor (F) and noise figure (NF) are calculated. The
figures indicate the appropriate scaling factor for each
of the noise sources in two different cases. The first
case includes the termination resistor, and the
second, simplified case assumes that the voltage
source is properly terminated by the gain-setting
resistors. With these scaling factors, the amplifier
input noise power (NA) can be calculated by summing
each individual noise source with its scaling factor.
The noise delivered to the amplifier by the source (NI)
and input noise power are used to calculate the noise
factor and noise figure as shown in Equation 23
through Equation 27.
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Ni
eg
NA
Rg
www.ti.com
ef
Rf
Si
Scaling Factors for Individual Noise Sources
Asseming No Termination Resistance is Used
(that is, RT is Open)
en
Ni
No
Rs
+
Rt
So
fully-diff
amp
−
ini
es
NA: Fully Differential Amplifier; termination = 2Rg
Noise
Source
No
et
eg
(ini)2
Rg
Rf
ef
ȡR )
ȧR R
Ȣ
(iii)2
f
s
2
(18)
(19)
Rg2
(20)
ǒ Ǔ
Rg
Rf
2
Figure 111. Noise Sources in a Fully Differential
Amplifier Circuit
2
(21)
2
ȡ R ȣ
ȧR ) R ȧ
Ȣ 2Ȥ
g
2
4kTRg
NA: Fully Differential Amplifier
Noise
Source Scale Factor
g
Rg2
4kTRf
Scaling Factors for Individual Noise Sources
Assuming a Finite Value Termination Resistor
2
ȣ
Rȧ
) Ȥ
Rg
g
(eni)2
iii
Scale Factor
s
g
(22)
Input Noise With a Termination Resistor
2
(12)
ȡ 2R R ȣ
ȧ R )2R ȧ
N + 4kTR ȧ
R ȧ
ȧ
ȧR )R2R)2R
Ȥ
Ȣ
(ini)2
Rg2
(13)
Input Noise Assuming No Termination Resistor
(iii)2
Rg2
(14)
Ni = 4kTRS
4kTRt
R
ȣ
ȡ R2R)2R
ȧR ) 2R R ȧ
Ȣ R )2R Ȥ
ȡR
ȧR ) R
Ȣ
g
(eni)2
f
t
2
ȣ
ȧ
Ȥ
Rg
R sR t
g) ǒ
2 Rs)R tǓ
i
2
4kTRf
4kTRg
30
g
s
t
2
s
ǒ Ǔ
Rg
Rf
ȡ
ȧR
Ȣ
t
t
G
s
g
t
s
s
2
s
g
g
g
(15)
2
(16)
2
ȣ
ȧ
Ȥ
g
g
2RG
(23)
2
RS + 2RG
Noise Factor and Noise Figure Calculations
NA = S(Noise Source ´ Scale Factor)
NA
F=1+
NI
NF + 10 log (F)
(24)
(25)
(26)
(27)
Rg
R sR t
g)
2ǒR s)RtǓ
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PRINTED CIRCUIT BOARD LAYOUT
TECHNIQUES FOR OPTIMAL
PERFORMANCE
Achieving optimum performance with high frequency
amplifier-like devices in the THS4500 family requires
careful attention to PCB layout parasitic and external
component types.
Recommendations that optimize performance include:
• Minimize parasitic capacitance to any ac ground
for all of the signal I/O pins. Parasitic capacitance
on the output and input pins can cause instability.
To reduce unwanted capacitance, a window
around the signal I/O pins should be opened in all
of the ground and power planes around those
pins. Otherwise, ground and power planes should
be unbroken elsewhere on the board.
• Minimize the distance (< 0.25”, 6.35 mm) from the
power-supply pins to high frequency 0.1-μF
decoupling capacitors. At the device pins, the
ground and power-plane layout should not be in
close proximity to the signal I/O pins. Avoid
narrow power and ground traces to minimize
inductance between the pins and the decoupling
capacitors. The power supply connections should
always be decoupled with these capacitors.
Larger (6.8 μF or more) tantalum decoupling
capacitors, effective at lower frequency, should
also be used on the main supply pins. These may
be placed somewhat farther from the device and
may be shared among several devices in the
same area of the PCB. The primary goal is to
minimize
the
impedance
seen
in
the
differential-current return paths.
• Careful selection and placement of external
components
preserve
the
high-frequency
performance of the THS4500 family. Resistors
should be a very low reactance type.
Surface-mount resistors work best and allow a
tighter overall layout. Metal-film and carbon
composition, axially-leaded resistors can also
provide good high frequency performance. Again,
keep the leads and PCB trace length as short as
possible. Never use wirewound type resistors in a
high-frequency application. Since the output pin
and inverting input pins are the most sensitive to
parasitic capacitance, always position the
feedback and series output resistors, if any, as
close as possible to the inverting input pins and
output pins. Other network components, such as
input termination resistors, should be placed close
to the gain-setting resistors. Even with a low
parasitic capacitance shunting the external
resistors, excessively high resistor values can
create significant time constants that can degrade
performance.
Good
axial
metal-film
or
surface-mount resistors have approximately
0.2 pF in shunt with the resistor. For resistor
values greater than 2.0 kΩ, this parasitic
•
•
•
•
capacitance can add a pole and/or a zero below
400 MHz that can affect circuit operation. Keep
resistor values as low as possible, consistent with
load driving considerations.
Connections to other wideband devices on the
board may be made with short direct traces or
through onboard transmission lines. For short
connections, consider the trace and the input to
the next device as a lumped capacitive load.
Relatively wide traces (50 mils to 100 mils, or 1.27
mm to 2.54 mm) should be used, preferably with
ground and power planes opened up around
them. Estimate the total capacitive load and
determine if isolation resistors on the outputs are
necessary. Low parasitic capacitive loads (less
than 4 pF) may not need an RS since the
THS4500 family is nominally compensated to
operate with a 2-pF parasitic load. Higher parasitic
capacitive loads without an RS are allowed as the
signal gain increases (increasing the unloaded
phase margin). If a long trace is required, and the
6-dB signal loss intrinsic to a doubly-terminated
transmission line is acceptable, implement a
matched impedance transmission line using
microstrip or stripline techniques (consult an ECL
design handbook for microstrip and stripline layout
techniques).
A 50-Ω environment is normally not necessary
onboard, and in fact, a higher impedance
environment improves distortion as shown in the
distortion versus load plots. With a characteristic
board trace impedance defined based onboard
material and trace dimensions, a matching series
resistor into the trace from the output of the
THS4500 family is used as well as a terminating
shunt resistor at the input of the destination
device.
Remember also that the terminating impedance is
the parallel combination of the shunt resistor and
the input impedance of the destination device: this
total effective impedance should be set to match
the trace impedance. If the 6-dB attenuation of a
doubly-terminated
transmission
line
is
unacceptable,
a
long
trace
can
be
series-terminated at the source end only. Treat
the trace as a capacitive load in this case. This
configuration does not preserve signal integrity as
well as a doubly-terminated line. If the input
impedance of the destination device is low, there
is some signal attenuation due to the voltage
divider formed by the series output into the
terminating impedance.
Socketing a high-speed part such as the THS4500
family is not recommended. The additional lead
length and pin-to-pin capacitance introduced by
the socket can create an extremely troublesome
parasitic network that can make it almost
impossible to achieve a smooth, stable frequency
response. Best results are obtained by soldering
Copyright © 2002–2011, Texas Instruments Incorporated
Product Folder Link(s): THS4500 THS4501
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31
THS4500
THS4501
SLOS350F – APRIL 2002 – REVISED OCTOBER 2011
www.ti.com
the THS4500 family parts directly onto the board.
0.205
PowerPAD DESIGN CONSIDERATIONS
The
THS4500
family
is
available
in
a
thermally-enhanced PowerPAD set of packages.
These packages are constructed using a downset
leadframe upon which the die is mounted [see
Figure 112(a) and Figure 112(b)]. This arrangement
results in the lead frame being exposed as a thermal
pad on the underside of the package [see
Figure 112(c)]. Because this thermal pad has direct
thermal contact with the die, excellent thermal
performance can be achieved by providing a good
thermal path away from the thermal pad.
The PowerPAD package allows for both assembly
and thermal management in one manufacturing
operation. During the surface-mount solder operation
(when the leads are being soldered), the thermal pad
can also be soldered to a copper area underneath the
package. Through the use of thermal paths within this
copper area, heat can be conducted away from the
package into either a ground plane or other heat
dissipating device.
The PowerPAD package represents a breakthrough
in combining the small area and ease of assembly of
surface mount with the, heretofore, awkward
mechanical methods of heatsinking.
DIE
Side View (a)
Thermal
Pad
DIE
End View (b)
Bottom View (c)
Figure 112. Views of PowerPAD,
Thermally-Enhanced Package
Although there are many ways to properly heatsink
the PowerPAD package, the following steps illustrate
the recommended approach.
32
Submit Documentation Feedback
0.060
0.017
Pin 1
0.013
0.030
0.075
0.025 0.094
0.010
vias
0.035
0.040
Top View
Figure 113. PowerPAD PCB Etch and Via Pattern
PowerPAD PCB LAYOUT CONSIDERATIONS
1. Prepare the PCB with a top side etch pattern as
shown in Figure 113. There should be etch for
the leads as well as etch for the thermal pad.
2. Place five holes in the area of the thermal pad.
These holes should be 13 mils (0.33 mm) in
diameter. Keep them small so that solder wicking
through the holes is not a problem during reflow.
3. Additional vias may be placed anywhere along
the thermal plane outside of the thermal pad
area. These holes help dissipate the heat
generated by the THS4500 family IC. These
additional vias may be larger than the 13-mil
diameter vias directly under the thermal pad.
They can be larger because they are not in the
thermal pad area to be soldered so that wicking
is not a problem.
4. Connect all holes to the internal ground plane.
5. When connecting these holes to the ground
plane, do not use the typical web or spoke via
connection methodology. Web connections have
a high thermal resistance connection that is
useful for slowing the heat transfer during
soldering operations. This transfer slowing makes
the soldering of vias that have plane connections
easier. In this application, however, low thermal
resistance is desired for the most efficient heat
transfer. Therefore, the holes under the THS4500
family PowerPAD package should make their
connection to the internal ground plane with a
complete
connection
around
the
entire
circumference of the plated-through hole.
Copyright © 2002–2011, Texas Instruments Incorporated
Product Folder Link(s): THS4500 THS4501
THS4500
THS4501
SLOS350F – APRIL 2002 – REVISED OCTOBER 2011
www.ti.com
POWER DISSIPATION AND THERMAL
CONSIDERATIONS
The THS4500 family of devices does not incorporate
automatic thermal shutoff protection, so the designer
must take care to ensure that the design does not
violate the absolute maximum junction temperature of
the device. Failure may result if the absolute
maximum junction temperature of +150°C is
exceeded. For best performance, design for a
maximum junction temperature of +125°C. Between
+125°C and +150°C, damage does not occur, but the
performance of the amplifier begins to degrade.
The thermal characteristics of the device are dictated
by the package and the PCB. Maximum power
dissipation for a given package can be calculated
using the following formula.
TMAX - TA
PDmax =
qJA
Where:
PDmax is the maximum power dissipation in the
amplifier (W).
TMAX is the absolute maximum junction
temperature (°C).
TA is the ambient temperature (°C).
θJA = θJC + θCA
θJC is the thermal coefficient from the silicon
junctions to the case (°C/W).
θCA is the thermal coefficient from the case to
ambient air (°C/W).
(28)
For systems where heat dissipation is more critical,
the THS4500 family of devices is offered in an
MSOP-8 package with PowerPAD. The thermal
coefficient for the MSOP PowerPAD package is
substantially improved over the traditional SOIC.
Maximum power dissipation levels are depicted in
Figure 114 for the two packages. The data for the
DGN package assumes a board layout that follows
the PowerPAD layout guidelines referenced above
and detailed in the PowerPAD application notes in
the Additional Reference Material section at the end
of the data sheet.
3.5
PD − Maximum Power Dissipation − W
6. The top-side solder mask should leave the
terminals of the package and the thermal pad
area with its five holes exposed. The bottom-side
solder mask should cover the five holes of the
thermal pad area. This configuration prevents
solder from being pulled away from the thermal
pad area during the reflow process.
7. Apply solder paste to the exposed thermal pad
area and all of the IC terminals.
8. With these preparatory steps in place, the IC is
simply placed in position and run through the
solder reflow operation as any standard
surface-mount component. This process results
in a part that is properly installed.
8-Pin DGN Package
3
2.5
2
8-Pin D Package
1.5
1
0.5
0
−40
−20
0
20
40
60
TA − Ambient Temperature − °C
80
θJA = 170°C/W for 8-Pin SOIC (D)
θJA = 58.4°C/W for 8-Pin MSOP (DGN)
ΤJ = 150°C, No Airflow
Figure 114. Maximum Power Dissipation vs
Ambient Temperature
When determining whether or not the device satisfies
the maximum power dissipation requirement, it is
important to not only consider quiescent power
dissipation, but also dynamic power dissipation. Often
times, this consideration is difficult to quantify
because the signal pattern is inconsistent; an
estimate of the RMS power dissipation can provide
visibility into a possible problem.
DRIVING CAPACITIVE LOADS
High-speed amplifiers are typically not well-suited for
driving large capacitive loads. If necessary, however,
the load capacitance should be isolated by two
isolation resistors in series with the output. The
requisite isolation resistor size depends on the value
of the capacitance, but 10 Ω to 25 Ω is a good place
to begin the optimization process. Larger isolation
resistors decrease the amount of peaking in the
frequency response induced by the capacitive load,
but this decreased peaking comes at the expense ofa
larger voltage drop across the resistors, increasing
the output swing requirements of the system.
Copyright © 2002–2011, Texas Instruments Incorporated
Product Folder Link(s): THS4500 THS4501
Submit Documentation Feedback
33
THS4500
THS4501
SLOS350F – APRIL 2002 – REVISED OCTOBER 2011
www.ti.com
RF
VS
RG
RS
RISO
+
VS
RT
-
-
CL
+
THS4501 product folder on the Texas Instruments
web site, www.ti.com, or through your local Texas
Instruments sales representative. A schematic for the
evaluation board is shown in Figure 116 with the
default component values. Unpopulated footprints are
shown to provide insight into design flexibility.
RISO
-VS
C4
Riso = 10 - 25 W
RF
R0805
VS
RG
J1
Figure 115. Use of Isolation Resistors With a
Capacitive Load
C1
R1
C0805
C2
R1206
C0805
R2
1
R0805
R0805
R3
POWER-SUPPLY DECOUPLING
TECHNIQUES AND RECOMMENDATIONS
Power-supply decoupling is a critical aspect of any
high-performance amplifier design process. Careful
decoupling provides higher quality ac performance
(most notably improved distortion performance). The
following guidelines ensure the highest level of
performance.
1. Place decoupling capacitors as close to the
power-supply inputs as possible, with the goal of
minimizing the inductance of the path from
ground to the power supply.
2. Placement priority should be as follows: smaller
capacitors should be closer to the device.
3. Use of solid power and ground planes is
recommended to reduce the inductance along
power-supply return current paths.
4. Recommended
values
for
power-supply
decoupling include 10-μF and 0.1-μF capacitors
for each supply. A 1000-pF capacitor can be
used across the supplies as well for extremely
high frequency return currents, but often is not
required.
EVALUATION FIXTURES, SPICE MODELS,
AND APPLICATIONS SUPPORT
Texas Instruments is committed to providing its
customers with the highest quality of applications
support. To support this goal, an evaluation board
has been developed for the THS4500 family of fully
differential amplifiers. The evaluation board can be
obtained by ordering through the THS4500 or
Submit Documentation Feedback
PD
U1
THS450X
R6
4
7
R0805
3
_
8
+
2
5
6
VOCM
34
C0805
R4
PwrPad
C5
C0805
C7
C0805
R0805
R7
J2
J3
J2
J3
C6
C0805
-V S
R5 R0805
C3
C0805
J2
R8
R0805
J3
R9
R0805
R0805
R9
4
5
6
J4
3
R11
R1206
T1
1
Figure 116. Simplified Schematic of the
Evaluation Board
Computer simulation of circuit performance using
SPICE is often useful when analyzing the
performance of analog circuits and systems. This
practice is particularly true for video and RF amplifier
circuits where parasitic capacitance and inductance
can have a major effect on circuit performance. A
SPICE model for the THS4500 family of devices is
available through either the Texas Instruments web
site (www.ti.com) or as one model on a disk from the
Texas Instruments Product Information Center
(1-800-548-6132). The PIC is also available for
design assistance and detailed product information at
this number. These models do a good job of
predicting small-signal ac and transient performance
under a wide variety of operating conditions. They are
not intended to model the distortion characteristics of
the amplifier, nor do they attempt to distinguish
between the package types in their small-signal ac
performance. Detailed information about what is and
is not modeled is contained in the model file itself.
Copyright © 2002–2011, Texas Instruments Incorporated
Product Folder Link(s): THS4500 THS4501
THS4500
THS4501
SLOS350F – APRIL 2002 – REVISED OCTOBER 2011
www.ti.com
ADDITIONAL REFERENCE MATERIAL
•
•
•
•
•
•
•
PowerPAD Made Easy, application brief, Texas Instruments Literature Number SLMA004.
PowerPAD Thermally-Enhanced Package, technical brief, Texas Instruments Literature Number SLMA002.
Karki, James. Fully Differential Amplifiers. application report, Texas Instruments Literature Number
SLOA054D.
Karki, James. Fully Differential Amplifiers Applications: Line Termination, Driving High-Speed ADCs, and
Differential Transmission Lines. Texas Instruments Analog Applications Journal, February 2001.
Carter, Bruce. A Differential Op-Amp Circuit Collection. application report, Texas Instruments Literature
Number SLOA064.
Carter, Bruce. Differential Op-Amp Single-Supply Design Technique, application report, Texas Instruments
Literature Number SLOA072.
Karki, James. Designing for Low Distortion with High-Speed Op Amps. Texas Instruments Analog
Applications Journal, July 2001.
REVISION HISTORY
Changes from Revision D (January 2004) to Revision E
Page
•
Updated document format .................................................................................................................................................... 1
•
Added footnote 1 to Package/Ordering Information table .................................................................................................... 3
•
Changed x-axis of Figure 27 ............................................................................................................................................... 12
•
Updated crossreferences for Figure 97 in first two paragraphs of the Fully Differential Amplifier Terminal Functions
section ................................................................................................................................................................................. 22
•
Added available for download at www.ti.com and end of second paragraph of the Fully Differential Amplifier
Terminal Functions section ................................................................................................................................................. 22
•
Changed allow for calculation of to are used to calculate in second paragraph of Input Common-Mode Voltage
Range and the THS4500 Family section ............................................................................................................................ 22
•
Clarified last sentence of third paragraph of Input Common-Mode Voltage Range and the THS4500 Family section ..... 22
•
Changed two to four in first sentence of Input Common-Mode Voltage Range and the THS4500 Family section ........... 22
•
Corrected title of Basic Design Considerations section ...................................................................................................... 24
•
Clarified cross-references of the circuits mentioned in the first sentence of the Basic Design Considerations section .... 24
•
Deleted figure from Basic Design Considerations section .................................................................................................. 24
•
Corrected cross-references in first sentence of Basic Design Considerations section ...................................................... 24
•
Clarified the Interfacing to an Analog-to-Digital Converter section ..................................................................................... 25
•
Removed cross-reference to nonexistant tble in second paragraph of Setting the Output Common-Mode Voltage
with the V OCM Input section ................................................................................................................................................ 27
•
Added caption titles to figures in the Linearity: Definitions, Terminology, Circuit Techniques, and Design Treadeoffs
section ................................................................................................................................................................................. 28
•
Changed THS4502 to THS4500 in seventh paragraph of Linearity: Definitions, Terminology, Circuit Techniques, and
Design Treadeoffs section .................................................................................................................................................. 28
•
Corrected spelling in title of An Analysis of Noise in Fully Differential Amplifiers section .................................................. 29
•
Added 6.35 mm to second bullet of Printed Circuit Board Layout Techniques for Optimal Performance list .................... 31
•
Added 1.27 mm to 2.54 mm to fourth bullet of Printed Circuit Board Layout Techniques for Optimal Performance list ... 31
•
Added 0.33 mm to second list item in the PowerPAD PCB Layout Considerations section .............................................. 32
•
Changed title of Figure 116 ................................................................................................................................................ 34
Changes from Revision E (MAy 2008) to Revision F
•
Page
Added Figure 101 to the Basic Design Considerations section ......................................................................................... 24
Copyright © 2002–2011, Texas Instruments Incorporated
Product Folder Link(s): THS4500 THS4501
Submit Documentation Feedback
35
PACKAGE OPTION ADDENDUM
www.ti.com
14-Oct-2022
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
Samples
(4/5)
(6)
THS4500CD
ACTIVE
SOIC
D
8
75
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
0 to 70
4500C
Samples
THS4500CDGNR
ACTIVE
HVSSOP
DGN
8
2500
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
0 to 70
BFB
Samples
THS4500ID
ACTIVE
SOIC
D
8
75
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 85
4500I
Samples
THS4500IDGK
ACTIVE
VSSOP
DGK
8
80
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 85
ASV
Samples
THS4500IDGN
ACTIVE
HVSSOP
DGN
8
80
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 85
BFC
Samples
THS4500IDGNG4
ACTIVE
HVSSOP
DGN
8
80
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 85
BFC
Samples
THS4500IDGNR
ACTIVE
HVSSOP
DGN
8
2500
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 85
BFC
Samples
THS4501CD
ACTIVE
SOIC
D
8
75
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
0 to 70
4501C
Samples
THS4501CDGN
ACTIVE
HVSSOP
DGN
8
80
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
0 to 70
BFD
Samples
THS4501ID
ACTIVE
SOIC
D
8
75
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 85
4501I
Samples
THS4501IDGK
ACTIVE
VSSOP
DGK
8
80
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 85
ASW
Samples
THS4501IDGN
ACTIVE
HVSSOP
DGN
8
80
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 85
BFE
Samples
THS4501IDR
ACTIVE
SOIC
D
8
2500
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 85
4501I
Samples
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of
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