THS4601CDDAG3

 SLOS388B − OCTOBER 2001 − REVISED JUNE 2002     
 

     FEATURES D Gain Bandwidth Product: 180 MHz D Slew Rate: 100 V/µs D Maximum Input Bias Current: 100 pA D Input Voltage Noise: 5.4 nV/√Hz D Maximum Input Offset Voltage: 4 mV D Input Impedance: 109 Ω || 10 pF D Power Supply Voltage Range: ±5 to ±15 V D Unity Gain Stable DESCRIPTION APPLICATIONS D Wideband Photodiode Amplifier D High-Speed Transimpedance Gain Stage D Test and Measurement Systems D Current-DAC Output Buffer D Active Filtering D High-Speed Signal Integrator D High-Impedance Buffer The characteristics of the THS4601 ideally suit it for use as a wideband photodiode amplifier. Photodiode output current is a prime candidate for transimpedance amplification, an application of which is illustrated in Figure 1. Other potential applications include test and measurement systems requiring high-input impedance, digital-to-analog converter output buffering, high-speed integration, and active filtering. The THS4601 is a high-speed, FET-input operational amplifier designed for applications requiring wideband operation, high-input impedance, and high-power supply voltages. By providing a 180-MHz gainbandwidth product, ±15-V supply operation, and 100-pA input bias current, the THS4601 is capable of wideband transimpedance gain and large output signal swing simultaneously. Low current and voltage noise allow amplification of extremely low-level input signals while still maintaining a large signal-to-noise ratio. A SELECTION OF RELATED OPERATIONAL AMPLIFIER PRODUCTS DEVICE VS (V) BW (MHz) SLEW RATE (V/µs) VOLTAGE NOISE (nV√Hz) OPA627 ±15 OPA637 ±15 16 55 4.5 Unity-gain stable FET-input amplifier 80 135 4.5 Gain of +5 stable FET-input amplifier OPA655 ±5 400 290 6 Unity-gain stable FET-input amplifier DESCRIPTION 100 kΩ TRANSIMPEDANCE BANDWIDTH 105 CF = 0.7 pF Transimpedance Gain − dB 100 RF = 100 kΩ _ λ 18 pF + THS4601 RL = 1 kΩ −VBias 95 90 85 80 75 70 Diode Capacitance: 18 pF −3 dB Bandwidth: 4 MHz 65 60 Figure 1. Wideband Photodiode Transimpedance Amplifier 0.1 1 10 Frequency − MHz 100 Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.  

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 SLOS388B − OCTOBER 2001 − REVISED JUNE 2002 Terminal Functions THS4601 D AND DDA PACKAGE (TOP VIEW) NC IN − IN + VS− 1 8 2 7 3 6 4 5 TERMINAL NAME NC VS+ OUT NC NC − No internal connection DESCRIPTION NO. NC 1, 5, 8 These pins have no internal connection. IN− 2 Inverting input of the amplifier IN+ 3 Noninverting input of the amplifier VS− OUT 4 Negative power supply 6 Output of the amplifier VS+ 7 Positive power supply absolute maximum ratings over operating free-air temperature (unless otherwise noted)† Supply voltage, VS+ . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 16.5 V Supply voltage, VS− . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . −16.5 V Input voltage, VI . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . ±VS Output current, IO . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 100 mA Differential input voltage, VID . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . ±4 V Maximum junction temperature, TJ . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 150°C Operating free-air temperature, TA: C-suffix . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 0°C to 70°C I-suffix . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . −40°C to 85°C Storage temperature, Tstg . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . −65°C to 125°C Lead temperature 1,6 mm (1/16 inch) from cases for 10 seconds . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 300°C † Stresses beyond those listed under “absolute maximum ratings” may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated under “recommended operating conditions” is not implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability. PACKAGE AND ORDERING INFORMATION PRODUCT PACKAGE PACKAGE DESIGNATOR SPECIFIED TEMPERATURE RANGE PACKAGE MARKING THS4601CD SOIC surface mount 8D 0°C to 70°C 4601C THS4601ID SOIC surface mount 8D −40°C to 85°C 4601I THS4601CDDA SOIC surface mount with PowerPAD 8DDA 0°C to 70°C 4601C THS4601IDDA SOIC surface mount with PowerPAD 8DDA −40°C to 85°C 4601I NOTE: The THS4601 is available taped and reeled. Add an R suffix to the device type when ordering (e.g., THS4601IDR). PowerPAD is a trademark of Texas Instruments. 2 www.ti.com 
 SLOS388B − OCTOBER 2001 − REVISED JUNE 2002 electrical specifications: VS = ±15 V: RF = 250 Ω, RL = 1 kΩ and G = +2 (unless otherwise noted) THS4601 PARAMETER TEST CONDITIONS TYP 25°C OVER TEMPERATURE 25°C 0°C to 70°C −40°C to 85°C MIN/ MAX UNIT AC PERFORMANCE G = +1, VO = 20 mVpp, RF = 0 Ω 440 Typ MHz G = +2, VO = 40 mVpp, RF = 62 Ω 95 Typ MHz G = +5, VO = 100 mVpp, RF = 500 Ω 36 Typ MHz G = +10, VO = 200 mVpp, RF = 1 kΩ 18 Typ MHz Gain-bandwidth product G > +10 180 Typ MHz Bandwidth for 0.1 dB flatness G = +2, VO = 200 mVpp 5 Typ MHz Large-signal bandwidth G = +5, VO = 10 Vpp Slew rate, SR G = +5, 10 V Step Rise/fall time, tr/tf 1.0 V Step Small-signal bandwidth 3 Typ MHz 100 Typ V/µs 7 Typ ns 0.01% G = +5, VO = 5 V Step 170 Typ ns 0.1% G = +5, VO = 5 V Step 135 Typ ns G = +2, f = 1 MHz, VO = 2Vpp RL = 100 Ω −65 Typ RL = 1 kΩ −77 Typ RL = 100 Ω −73 Typ RL = 1 kΩ −96 Typ Input voltage noise, Vn f > 10 kHz 5.4 Typ nV/√Hz Input current noise, In f > 10 kHz 5.5 Typ fA/√Hz Differential gain (NTSC, PAL) G = +2, RL = 150 Ω 0.02% Typ Differential phase (NTSC, PAL) G = +2, RL = 150 Ω 0.08 Typ _ Open-loop voltage gain G = −10, RL = 1 kΩ 105 94 90 Min dB Input offset voltage, VIO VCM = 0 V VCM = 0 V 1.0 4.0 VCM = 0 V VCM = 0 V 30 VCM = 0 V VCM = 0 V 2 Settling time, ts Harmonic distortion 2nd Harmonic 3rd Harmonic dBc dBc DC PERFORMANCE Average offset voltage drift Input bias current, IIB Average bias current drift Input offset current, IIO Average offset current drift 100 100 92 4.5 5.0 Max mV ±10 ±10 Typ µV/_C 550 1100 Max pA 50 50 Typ pA/°C 200 300 Max pA 5 5 Typ pA/°C 12.4 to −11.8 Min V INPUT ±13.0 Common-mode input range, VIC Common-mode rejection ratio, CMRR Input impedance, Zid Differential Input impedance, Zic Common-mode 110 109 || 3.5 109 || 6.5 www.ti.com 12.6 12.5 to to −11.9 −12.0 100 95 90 Min dB Typ Ω || pF Typ Ω || pF 3 
 SLOS388B − OCTOBER 2001 − REVISED JUNE 2002 electrical specifications: VS = ±15 V: RF = 250 Ω, RL = 1kΩ and G = +2 (unless otherwise noted) (continued) THS4601 PARAMETER TEST CONDITIONS TYP OVER TEMPERATURE 25°C 25°C 0°C to 70°C −40°C to 85°C MIN/ MAX UNIT 12.8 to –13.4 12.4 to −13.1 12.3 to −13.0 12.1 to −12.8 Min V −80 −60 −60 −59 Min 50 35 35 34 Min OUTPUT Voltage output swing RL = 1 kΩ Sourcing Current output, IO Sinking Closed-loop output impedance, Zo RL = 20 Ω G = +1, f = 1 MHz 0.1 Typ mA Ω POWER SUPPLY Specified operating voltage ±15 ±16.5 ±16.5 ±16.5 Max V Maximum quiescent current 10.0 11.5 11.7 12.0 Max mA Minimum quiescent current 10.0 8.5 8.3 8.0 Min mA +PSRR 115 90 88 86 Min −PSRR 115 90 88 86 Min Power supply rejection dB TEMPERATURE −40 to 85 Typ °C 8D: SO−8 170 Typ °C/W 8DDA: SO−8 with PowerPAD 66.6 Typ °C/W Specified operating range, TA Thermal resistance, θJA 4 Junction-to-ambient www.ti.com 
 SLOS388B − OCTOBER 2001 − REVISED JUNE 2002 electrical specifications: VS = ±5 V: RF = 250 Ω, RL = 1 kΩ and G = +2 (unless otherwise noted) THS4601 PARAMETER TEST CONDITIONS TYP 25°C OVER TEMPERATURE 25°C 0°C to 70°C −40°C to 85°C MIN/ MAX UNIT AC PERFORMANCE G = +1, VO = 20 mVpp 400 Typ MHz G = +2, VO = 40 mVpp 100 Typ MHz G = +5, VO = 100 mVpp 50 Typ MHz G = +10, VO = 200 mVpp 18 Typ MHz Gain-bandwidth product G > +10 180 Typ MHz Bandwidth for 0.1 dB flatness G = +2, VO = 200 mVpp 5 Typ MHz Large-signal bandwidth G = +5, VO = 5 Vpp 6 Typ MHz Slew rate, SR G = +5, 5 V Step 100 Typ V/µs Small-signal bandwidth Rise/fall time, tr/tf 8 Typ ns 0.01% G = +5, VO = 2 V Step 140 Typ ns 0.1% G = +5, VO = 2 V Step 170 Typ ns G = +2, f = 1 MHz, VO = 2Vpp RL = 100 Ω −74 Typ RL = 1 kΩ −84 Typ RL = 100 Ω −79 Typ RL = 1 kΩ −94 Typ Input voltage noise, Vn f > 10 kHz 5.4 Typ nV/√Hz Input current noise, In f > 10 kHz 5.5 Typ fA/√Hz Differential gain (NTSC and PAL) G = +2, RL = 150 Ω 0.02% Typ Differential phase (NTSC and PAL) G = +2, RL = 150 Ω 0.08 Typ Open-loop voltage gain G = −10, RL = 1 kΩ 105 94 92 90 Min dB Input offset voltage, VIO VCM = 0 V VCM = 0 V 1.0 4.0 4.5 5.0 Max mV ±10 ±10 Typ µV/_C VCM = 0 V VCM = 0 V 20 550 1100 Max pA 50 50 Typ pA/°C VCM = 0 V VCM = 0 V 1 200 300 Max pA 5 5 Typ pA/°C Min V Settling time, ts 1.0 V Step Harmonic distortion 2nd Harmonic 3rd Harmonic dBc dBc _ DC PERFORMANCE Average offset voltage drift Input bias current, IIB Average bias current drift Input offset current, IIO Average offset current drift 100 100 INPUT Common-mode input range, VIC Common-mode rejection ratio, CMRR Input impedance, Zid Differential Input impedance, Zic Common-mode ±2.2 2.7 to −2.0 2.6 to −1.9 2.5 to −1.8 110 109 || 3.5 109 || 6.5 100 95 90 2.9 to –3.5 2.6 to −3.3 2.5 to −3.2 −65 −48 45 30 Min dB Typ Ω || pF Typ Ω || pF 2.3 to −3.1 Min V −48 −47 Min 30 29 Min OUTPUT Voltage output swing RL = 1 kΩ Sourcing Current output, IO Sinking Closed-loop output impedance, Zo RL = 20 Ω G = +1, f = 1 MHz 0.1 www.ti.com Typ mA Ω 5 
 SLOS388B − OCTOBER 2001 − REVISED JUNE 2002 electrical specifications: VS = ±5 V; RF = 250 Ω, RL = 1 kΩ and G = +2 (unless otherwise noted) (continued) THS4601 PARAMETER TEST CONDITIONS TYP OVER TEMPERATURE UNIT 25°C 25°C 0°C to 70°C −40°C to 85°C MIN/ MAX Specified operating voltage ±5 ±16.5 ±16.5 ±16.5 Max V Maximum quiescent current 9.6 11.2 11.4 11.7 Max mA mA POWER SUPPLY Minimum quiescent current Power supply rejection 9.6 8.2 8.0 7.7 Min +PSRR 110 90 88 86 Min −PSRR 110 90 88 86 Min dB TEMPERATURE −40 to 85 Typ °C 8D: SO−8 170 Typ °C/W 8DDA: SO−8 with PowerPAD 67 Typ °C/W Specified operating range, TA Thermal resistance, θJA 6 Junction-to-ambient www.ti.com 
 SLOS388B − OCTOBER 2001 − REVISED JUNE 2002 TYPICAL CHARACTERISTICS Table of Graphs FIGURE Small-Signal Unity Gain Frequency Response 2 Large-Signal Unity Gain Frequency Response 3 Small-Signal Frequency Response, Gain = +2 4 Small-Signal Frequency Response, Gain = +5 5 Small-Signal Frequency Response, Gain = +10 6 Small-Signal Frequency Response, Gain = +100 7 Open-Loop Gain and Phase vs Frequency 8 Voltage Noise vs Frequency 9 Rejection Ratios vs Frequency 10 Closed-Loop Output Impedance vs Frequency 11 Large-Signal Pulse Response 12 Harmonic Distortion vs Frequency 13 Harmonic Distortion vs Output Voltage Swing 14 Slew Rate vs Output Voltage Step 15 Input Bias Current vs Input Common-Mode Range 16 Common-Mode Rejection Ratio vs Input Common-Mode Range 17 Open-Loop Gain vs Temperature 18 Input Bias Current vs Temperature 19 Input Offset Current vs Temperature 20 Offset Voltage vs Temperature 21 Quiescent Current vs Temperature 22 Output Current vs Temperature 23 Output Voltage Swing vs Temperature 24 Rejection Ratios vs Temperature 25 www.ti.com 7 
 SLOS388B − OCTOBER 2001 − REVISED JUNE 2002 TYPICAL CHARACTERISTICS measurement conditions: TA = 25°C, RL = 1 kΩ, VS = ±15 V (unless otherwise noted) SMALL-SIGNAL UNITY GAIN FREQUENCY RESPONSE LARGE-SIGNAL UNITY GAIN FREQUENCY RESPONSE 5 8 Gain = 1, RF = 0Ω, RL = 1 kΩ, PIN = −30 dBm 6 4 0 −5 Gain − dB Gain − dB 2 0 −2 −10 −15 −4 −20 −6 Gain = 1, RF = 0 Ω, RL = 1 kΩ, PIN = 0 dBm −25 −8 100 k 1M 10 M 100 M Frequency − Hz 100 k 1G 1M 10 M 100 M Frequency − Hz Figure 2 Figure 3 SMALL-SIGNAL FREQUENCY RESPONSE, GAIN = +2 SMALL-SIGNAL FREQUENCY RESPONSE, GAIN = +5 10 20 8 15 6 10 Gain − dB Gain − dB 4 2 0 −2 −4 −8 −10 0 −5 Gain = 2, RF = 62Ω, RL = 1 kΩ, PIN = −30 dBm −6 5 Gain = 5, RF = 500Ω, RL = 1 kΩ, PIN = −30 dBm −10 −15 100 k 1M 10 M 100 M Frequency − Hz 1G 100 k Figure 4 1M 10 M 100 M Frequency − Hz Figure 5 SMALL-SIGNAL FREQUENCY RESPONSE, GAIN = +100 SMALL-SIGNAL FREQUENCY RESPONSE, GAIN = +10 50 25 Gain = 100, RF = 5 kΩ, RL = 1 kΩ, PIN = −30 dBm 20 40 30 10 Gain − dB Gain − dB 15 5 0 −5 −10 −15 20 10 Gain = 10, RF = 1 kΩ, RL = 1 kΩ, PIN = −30 dBm 100 k 1M 10 M 100 M Frequency − Hz 0 −10 1G 100 k Figure 6 8 1G 1M 10 M 100 M Frequency − Hz Figure 7 www.ti.com 1G 
 SLOS388B − OCTOBER 2001 − REVISED JUNE 2002 TYPICAL CHARACTERISTICS measurement conditions: TA = 25°C, RL = 1 kΩ, VS = ±15 V (unless otherwise noted) VOLTAGE NOISE vs FREQUENCY 60 90 30 80 0 70 −30 60 −60 50 −90 40 −120 30 −150 20 −180 10 −210 0 −240 −270 10 k 100 k 1 M 10 M 100 M 1 G −10 10 100 1k 60 50 Hz 90 Voltage Noise − nV/ 110 100 Phase − ° Gain − dB OPEN-LOOP GAIN AND PHASE vs FREQUENCY 40 30 20 10 0 10 100 Frequency − Hz Figure 8 100 k CLOSED-LOOP OUTPUT IMPEDANCE vs FREQUENCY 100 CMRR Output Impedance − Ω PSRR+ 100 Rejection Ratio − dB 10 k Figure 9 REJECTION RATIOS vs FREQUENCY 120 1k Frequency − Hz PSRR− 80 60 40 10 1 0.1 20 0 100 1k 10 k 100 k 1M 10 M 0.01 100 k 100 M 1M 10 M Frequency − Hz Frequency − Hz Figure 10 Figure 11 HARMONIC DISTORTION vs FREQUENCY LARGE-SIGNAL PULSE RESPONSE 3 −20 −30 2 −40 1 Distortion − dBc Output Voltage − V 100 M 0 −1 Gain = 2, RF = 250 Ω, RL = 1 kΩ, VO = 2 VPP 3rd Harmonic −50 −60 −70 2nd Harmonic −80 −2 −90 −3 0 0.2 0.4 0.6 0.8 −100 100 k 1 t − Time − µs 1M 10 M Frequency − Hz Figure 12 Figure 13 www.ti.com 9 
 SLOS388B − OCTOBER 2001 − REVISED JUNE 2002 TYPICAL CHARACTERISTICS measurement conditions: TA = 25°C, RL = 1 kΩ, VS = ±15 V (unless otherwise noted) SLEW RATE vs OUTPUT VOLTAGE STEP HARMONIC DISTORTION vs OUTPUT VOLTAGE SWING 130 −50 −55 120 Slew Rate − V/ µ s Distortion − dBc SR− 3rd Harmonic −60 2nd Harmonic −65 −70 −75 −80 −85 −95 SR+ 100 90 80 Gain = 2, RF = 250 Ω, RL = 1 kΩ, f = 1 MHz −90 110 Gain = 5, RL = 1 kΩ 70 60 −100 0 2 4 6 8 10 0 12 5 10 15 20 Output Voltage Step − V Peak-to-Peak Output Swing − V Figure 14 Figure 15 COMMON-MODE REJECTION RATIO vs INPUT COMMON-MODE RANGE INPUT BIAS CURRENT vs INPUT COMMON-MODE RANGE 120 10 k 60 CMRR − dB Input Bias Current − pA 100 1k 100 80 40 10 20 1 −15 −10 −5 0 5 10 0 −15 15 −10 −5 0 5 10 15 Input Common-Mode Range − V Input Common-Mode Range − V Figure 16 Figure 17 OPEN-LOOP GAIN vs TEMPERATURE INPUT BIAS CURRENT vs TEMPERATURE 110 100 k 108 10 k Input Bias Current − pA Open-Loop Gain − dB 106 104 102 100 98 96 1k 100 10 94 92 −40 −20 IIB− 1 0 20 40 60 80 100 120 −40 −20 0 Case Temperature − °C 20 40 60 80 Case Temperature − °C Figure 19 Figure 18 10 IIB+ www.ti.com 100 120 
 SLOS388B − OCTOBER 2001 − REVISED JUNE 2002 TYPICAL CHARACTERISTICS measurement conditions: TA = 25°C, RL = 1 kΩ, VS = ±15 V (unless otherwise noted) OFFSET VOLTAGE vs TEMPERATURE INPUT OFFSET CURRENT vs TEMPERATURE 2.50 15 10 2 0 Offset Voltage − mV Input Offset Current − pA 5 −5 −10 −15 −20 −25 1.50 1 0.50 −30 −35 −40 0 −40 −20 0 20 40 60 80 100 120 −40 −20 Case Temperature − °C 0 20 40 60 80 Case Temperature − °C Figure 20 Figure 21 QUIESCENT CURRENT vs TEMPERATURE OUTPUT CURRENT vs TEMPERATURE 10.2 90 Sourcing Current 85 10 80 Output Current − mA Quiescent Current − mA 100 120 9.8 9.6 9.4 75 70 65 60 55 Sinking Current 50 9.2 45 9 −40 −20 40 0 20 40 60 80 100 120 −40 −20 0 20 40 60 80 Case Temperature − °C Case Temperature − °C Figure 22 Figure 23 OUTPUT VOLTAGE SWING vs TEMPERATURE VO− −13.6 −13.4 13.4 −13.2 13.2 VO+ 13 −13 12.8 −12.8 12.6 −12.6 130 125 PSRR+ 120 Rejection Ratios − dB −13.8 13.8 Negative Output Voltage Swing − V Positive Output Voltage Sling − V REJECTION RATIOS vs TEMPERATURE −14 14 13.6 100 120 115 CMRR 110 105 100 95 PSRR− 90 85 80 75 12.4 −40 −20 0 20 40 60 80 −12.4 100 120 70 −40 −20 0 Case Temperature − °C 20 40 60 80 100 120 Case Temperature − °C Figure 24 Figure 25 www.ti.com 11 
 SLOS388B − OCTOBER 2001 − REVISED JUNE 2002 APPLICATION INFORMATION introduction The THS4601 is a high-speed, FET-input operational amplifier. The combination of its high frequency capabilities and its DC precision make it a design option for a wide variety of applications, including test and measurement, optical monitoring, transimpedance gain circuits, and high-impedance buffers. The applications section of the data sheet discusses these particular applications in addition to general information about the device and its features. transimpedance fundamentals FET-input amplifiers are often used in transimpedance applications because of their extremely high input impedance. A transimpedance block accepts a current as an input and converts this current to a voltage at the output. The high-input impedance associated with FET-input amplifiers minimizes errors in this process caused by the input bias currents, IIB, of the amplifier. designing the transimpedance circuit Typically, design of a transimpedance circuit is driven by the characteristics of the current source that provides the input to the gain block. A photodiode is the most common example of a capacitive current source that would interface with a transimpedance gain block. Continuing with the photodiode example, the system designer traditionally chooses a photodiode based on two opposing criteria: speed and sensitivity. Faster photodiodes cause a need for faster gain stages, and more sensitive photodiodes require higher gains in order to develop appreciable signal levels at the output of the gain stage. These parameters affect the design of the transimpedance circuit in a few ways. First, the speed of the photodiode signal determines the required bandwidth of the gain circuit. However, the required gain, based on the sensitivity of the photodiode, limits the bandwidth of the circuit. Additionally, the larger capacitance associated with a more sensitive signal source also detracts from the achievable speed of the gain block. The dynamic range of the input signal also places requirements on the amplifier’s dynamic range. Knowledge of the source’s output current levels, coupled with a desired voltage swing on the output, dictates the value of the feedback resistor, RF. The transfer function from input to output is VOUT = IINRF. The large gain-bandwidth product of the THS4601 provides the capability for achieving both high transimpedance gain and wide bandwidth simultaneously. In addition, the high power supply rails provide the potential for a very wide dynamic range at the output, allowing for the use of input sources which possess wide dynamic range. The combination of these characteristics makes the THS4601 a design option for systems that require transimpedance amplification of wideband, low-level input signals. A standard transimpedance circuit is shown in Figure 26. CF RF _ λ + RL −VBias Figure 26. Wideband Photodiode Transimpedance Amplifier 12 www.ti.com 
 SLOS388B − OCTOBER 2001 − REVISED JUNE 2002 APPLICATION INFORMATION designing the transimpedance circuit (continued) As indicated, the current source typically sets the requirements for gain, speed, and dynamic range of the amplifier. For a given amplifier and source combination, achievable performance is dictated by the following parameters: the amplifier’s gain-bandwidth product, the amplifier’s input capacitance, the source capacitance, the transimpedance gain, the amplifier’s slew rate, and the amplifier’s output swing. From this information, the optimal performance of a transimpedance circuit using a given amplifier can be determined. Optimal is defined here as providing the required transimpedance gain with a maximally flat frequency response. For the circuit shown in Figure 26, all but one of the design parameters is known; the feedback capacitor must be determined. Proper selection of the feedback capacitor prevents an unstable design, controls pulse response characteristics, provides maximally flat transimpedance bandwidth, and limits broadband integrated noise. The maximally flat frequency response results with CF calculated as shown in equation 1, where CF is the feedback capacitor, RF is the feedback resistor, CS is the total source capacitance (including amplifier input capacitance and parasitic capacitance at the inverting node), and GBP is the gain-bandwidth product of the amplifier in hertz. 1 ) pR GBP F Ǹǒ Ǔ 2 1 pR GBP F C + F ) 4C S pR GBP F (1) 2 Once the optimal feedback capacitor has been selected, the transimpedance bandwidth can be calculated with equation 2. Ǹ GBP CIDIFF + _ F –3 dB + CICM IDIODE 2pR F ǒC S ) C FǓ (2) Cs = CICM + CIDIFF + CP + CD CP RF CD CF Where: CICM is the common-mode input capacitance. CIDIFF is the differential input capacitance. CD is the diode capacitance. CP is parasitic capacitance at the inverting node. NOTE: The total source capacitance is the sum of several distinct capacitances. Figure 27. Transimpedance Analysis Circuit www.ti.com 13 
 SLOS388B − OCTOBER 2001 − REVISED JUNE 2002 APPLICATION INFORMATION designing the transimpedance circuit (continued) The feedback capacitor provides a pole in the noise gain of the circuit, counteracting the zero in the noise gain caused by the source capacitance. The pole is set such that the noise gain achieves a 20 dB per decade rate-of-closure with the open-loop gain response of the amplifier, resulting in a stable circuit. As indicated, the formula given provides the feedback capacitance for maximally flat bandwidth. Reduction in the value of the feedback capacitor can increase the signal bandwidth, but this occurs at the expense of peaking in the AC response. Gain AOL −20 dB/ Decade 20 dB/Decade Rate-of-Closure Noise Gain GBP 20 dB/ Decade 0 f Zero Pole Figure 28. Transimpedance Circuit Bode Plot The performance of the THS4601 has been measured for a variety of transimpedance gains with a variety of source capacitances. The achievable bandwidths of the various circuit configurations are summarized numerically in the table. The frequency responses are presented in the Figures 27, 28, and 29. Note that the feedback capacitances do not correspond exactly with the values predicted by the equation. They have been tuned to account for the parasitic capacitance of the feedback resistor (typically 0.2 pF for 0805 surface mount devices) as well as the additional capacitance associated with the PC board. The equation should be used as a starting point for the design, with final values for CF optimized in the laboratory. 14 www.ti.com 
 SLOS388B − OCTOBER 2001 − REVISED JUNE 2002 APPLICATION INFORMATION designing the transimpedance circuit (continued) Table 1. Transimpedance Performance Summary for Various Configurations SOURCE CAPACITANCE (pF) TRANSIMPEDANCE GAIN (Ω) FEEDBACK CAPACITANCE (pF) −3 dB FREQUENCY (MHz) 18 10k 2.2 10.4 18 100k 0.6 3.3 18 1M 0 1.1 47 10k 3.3 7.6 47 100k 0.6 2.8 47 1M 0 0.88 100 10k 3.9 5.9 100 100k 1.5 1.3 100 1M 0 0.62 220 10k 5.6 3.8 220 100k 1.8 1.1 220 1M 0.4 0.36 10 kΩ TRANSIMPEDANCE BANDWIDTH FOR VARIOUS SOURCE CAPACITANCES 100 kΩ TRANSIMPEDANCE BANDWIDTH FOR VARIOUS SOURCE CAPACITANCES 1 MΩ TRANSIMPEDANCE BANDWIDTH FOR VARIOUS SOURCE CAPACITANCES 105 90 130 CS = 18 pF, CF = 2.2 pF 80 CS = 47 pF, CF = 3.3 pF 75 CS = 100 pF, CF = 3.9 pF 70 CS = 220 pF, CF = 5.6 pF 65 10 k 100 k 1M 95 90 85 100 M CS = 47 pF, CF = 0.6 pF 80 75 CS = 100 pF, CF = 1.5 pF 70 65 10 M 125 CS = 18 pF, CF = 0.6 pF Transimpedance Gain − dB 85 Transimpedance Gain − dB Transimpedance Gain − dB 100 60 10 k Frequency − Hz Figure 29 CS = 220 pF, CF = 1.8 pF CS = 18 pF, CF = 0 120 CS = 47 pF, CF = 0 115 110 CS = 100 pF, CF = 0 105 100 CS = 220 pF, CF = 0.4 pF 95 90 100 k 1M Frequency − Hz Figure 30 10 M 10 k 100 k 1M 10 M Frequency − Hz Figure 31 measuring transimpedance bandwidth While there is no substitute for measuring the performance of a particular circuit under the exact conditions that are used in the application, the complete system environment often makes measurements harder. For transimpedance circuits, it is difficult to measure the frequency response with traditional laboratory equipment because the circuit requires a current as an input rather than a voltage. Also, the capacitance of the current source has a direct effect on the frequency response. A simple interface circuit can be used to emulate a capacitive current source with a network analyzer. With this circuit, transimpedance bandwidth measurements are simplified, making amplifier evaluation easier and faster. www.ti.com 15 
 SLOS388B − OCTOBER 2001 − REVISED JUNE 2002 APPLICATION INFORMATION measuring transimpedance bandwidth (continued) IO Network Analyzer 50 Ω C2 50 Ω RS I O + V S C1 VS ǒ 1 Ǔ C 2R S 1 ) 1 C2 (above the pole frequency) NOTE: This interface network creates a capacitive, constant current source from a network analyzer and properly terminates the network analyzer at high frequencies. Figure 32. Emulating a Capacitive Current Source With a Network Analyzer The transconductance transfer function of the interface circuit is ǒ s Ǔ C 2R S 1) 1 C2 I O (s) + Vs s) 1 2R SǒC 1 ) . C Ǔ 2 This transfer function contains a zero at DC and a pole at s + constant at ǒ 1 1 . The transconductance is 2R SǒC 1 ) C 2Ǔ Ǔ , above the pole frequency, providing a controllable AC current source. This circuit C 2R S 1 ) 1 C2 also properly terminates the network analyzer with 50 Ω at high frequencies. The second requirement for this current source is to provide the desired output impedance, emulating the output impedance of a photodiode or other current source. The output impedance of this circuit is given by 1 ȱs ) ȳ ǒ Ǔ 2R C C s 1 C ) C2 ȧ 2 ȧ Z O(s) + 1 ȧ ȧ. C 1C 2 ȧ ȧ 1 Ǔ sǒs ) 2R sC 1 ȴ Ȳ ) Assuming C1 >> C2, the equation reduces to Z O [ 1 , giving the appearance of a capacitive source at higher sC 2 frequency. Capacitor values should be chosen to satisfy two requirements. First, C2 should represent the anticipated capacitance of the true source. C1 should then be chosen such that the corner frequency of the transconductance network is much less than the transimpedance bandwidth of the circuit. Choosing this corner frequency properly leads to more accurate measurements of the transimpedance bandwidth. If the interface circuit’s corner frequency is too close to the bandwidth of the circuit, determining the power level in the flatband is difficult. A decade or more of flat bandwidth provides a good basis for determining the proper transimpedance bandwidth. 16 www.ti.com 
 SLOS388B − OCTOBER 2001 − REVISED JUNE 2002 APPLICATION INFORMATION alternative transimpedance configurations Other transimpedance configurations are possible. Three possibilities are shown below. The first configuration is a slight modification of the basic transimpedance circuit. By splitting the feedback resistor, the feedback capacitor value becomes more manageable and easier to control. This type of compensation scheme is useful when the feedback capacitor required in the basic configuration becomes so small that the parasitic effects of the board and components begin to dominate the total feedback capacitance. By reducing the resistance across the capacitor, the capacitor value can be increased. This mitigates the dominance of the parasitic effects. CF RF2 RF1 _ λ + RL −VBias NOTE: Splitting the feedback resistor enables use of a larger, more manageable feedback capacitor. Figure 33. Alternative Transimpedance Configuration #1 The second configuration uses a resistive T-network to achieve very high transimpedance gains using relatively small resistor values. This topology can be very useful when the desired transimpedance gain exceeds the value of available resistors. The transimpedance gain is given by equation 3. ǒ Ǔ R R EQ + R F1 1 ) F2 R F3 (3) RF3 CF RF1 RF2 _ λ + RL −VBias NOTE: A resistive T-network enables high transimpedance gain with reasonable resistor values. Figure 34. Alternative Transimpedance Configuration #2 www.ti.com 17 
 SLOS388B − OCTOBER 2001 − REVISED JUNE 2002 APPLICATION INFORMATION alternative transimpedance configurations (continued) The third configuration uses a capacitive T-network to achieve fine control of the compensation capacitance. The capacitor CF3 can be used to tune the total effective feedback capacitance to a very fine degree. This circuit behaves the same as the basic transimpedance configuration, with the effective CF given by equation 4. ǒ Ǔ C + 1 1 ) F3 C F2 C F1 C FEQ 1 (4) CF3 CF1 CF2 RF _ λ + RL −VBias NOTE: A capacitive T-network enables fine control of the effective feedback capacitance using relatively large capacitor values. Figure 35. Alternative Transimpedance Configuration #3 summary of key decisions in transimpedance design The following is a quick, simplified process for basic transimpedance circuit design. This process gives a quick start to the design process, though it does ignore some aspects that may be critical to the circuit. 18 Step 1: Determine the capacitance of the source. Step 2: Calculate the total source capacitance, including the amplifier input capacitance, CICM and CIDIFF . Step 3: Determine the magnitude of the possible current output from the source, including the minimum signal current anticipated and maximum signal current anticipated. Step 4: Choose a feedback resistor value such that the input current levels create the desired output signal voltages, and ensure that the output voltages can accommodate the dynamic range of the input signal. Step 5: Calculate the optimum feedback capacitance using equation 1. Step 6: Calculate the bandwidth given the resulting component values. Step 7: Evaluate the circuit to see if all design goals are satisfied. www.ti.com 
 SLOS388B − OCTOBER 2001 − REVISED JUNE 2002 APPLICATION INFORMATION selection of feedback resistors Feedback resistor selection can have a significant effect on the performance of the THS4601 in a given application, especially in configurations with low closed-loop gain. If the amplifier is configured for unity gain, the output should be directly connected to the inverting input. Any resistance between these two points interacts with the input capacitance of the amplifier and causes an additional pole in the frequency response. For non-unity gain configurations, low resistances are desirable for flat frequency response. However, care must be taken not to load the amplifier too heavily with the feedback network if large output signals are expected. In most cases, a tradeoff will be made between the frequency response characteristics and the loading of the amplifier. For a gain of 2, a 250 Ω feedback resistor is a suitable operating point from both perspectives. If resistor values are chosen too large, the THS4601 is subject to oscillation problems. For example, an inverting amplifier configuration with a 1-kΩ gain resistor and a 1-kΩ feedback resistor develops an oscillation due to the interaction of the large resistors with the input capacitance. In low gain configurations, avoid feedback resistors this large or anticipate using an external compensation scheme to stabilize the circuit. overdrive recovery The THS4601 has an overdrive recovery period when the output is driven close to one power supply rail or the other. The overdrive recovery time period is dependent upon the magnitude of the overdrive and whether the output is driven towards the positive or the negative power supply. The four graphs shown here depict the overdrive recovery time in two cases, an attempted 28 VPP signal on the output and an attempted 30 VPP signal on the output. Note that in both of these cases, the output does not achieve these levels as the output voltage swing is limited to less than these values, but these values are representative of the desired signal swing on the output for the given inputs. As shown in the figures, the recovery period increases as the magnitude of the overdrive increases, with the worst case recovery occurring with the negative rail. The recovery times are summarized in Table 2. Table 2. Overdrive Recovery Characteristics IDEAL OUTPUT SWING (VPP) OVERDRIVE RECOVERY TIME (ns) +VS −VS 28 320 28 340 +VS −VS 30 540 30 680 VOLTAGE RAIL www.ti.com 19 
 SLOS388B − OCTOBER 2001 − REVISED JUNE 2002 APPLICATION INFORMATION overdrive recovery (continued) FALLING EDGE OVERDRIVE RECOVERY RISING EDGE OVERDRIVE RECOVERY 3 Input −5 −1 −10 −2 −15 −3 −20 Output Voltage − V 0 0 2 3 0 0 −5 −1 −2 Gain = 5, VIN = 5.57 VPP, Recovery Time = 320 ns −15 −20 1 2 FALLING EDGE OVERDRIVE RECOVERY Gain = 5, VIN = 6 VPP, Recovery Time = 680 ns 0 0 −1 −5 Input Output Voltage − V 1 Output −20 1 0 0 −5 −1 −2 Gain = 5, VIN = 6 VPP, Recovery Time = 540 ns −20 −4 3 5 −15 −3 −15 Input −10 −2 −10 2 10 2 5 3 Output Input Voltage − V Output Voltage − V 15 3 10 4 20 4 20 2 4 Figure 37 RISING EDGE OVERDRIVE RECOVERY 1 3 Time − µs Figure 36 0 −3 −4 0 4 Time − µs 15 1 −3 −4 0 4 Input Voltage − V 1 Input 5 −10 −4 0 2 10 Input Voltage − V 1 5 3 Output 2 Output 10 4 15 Input Voltage − V Gain = 5, VIN = 5.57 VPP, Recovery Time = 340 ns 15 Output Voltage − V 20 4 20 1 2 3 4 Time − µs Time − µs Figure 38 Figure 39 high frequency continuous wave amplification When presented with high frequency sinusoids in low-gain configurations (G < 5), the THS4601 experiences a relatively large differential input voltage between the two input terminals of the amplifier. As this differential input voltage increases, the internal slew-boosting circuitry can cause some transistors in the signal path to enter the cutoff region of operation. As the derivative of the signal changes signs, these transistors suffer from a short recovery time period, generating appreciable levels of distortion. This behavior is depicted in the graph Harmonic Distortion vs Frequency. At 2 MHz with a 2 VPP output signal, the distortion rises significantly. For most high-gain configurations including transimpedance applications, this phenomena is not problematic. slew rate performance with varying input step amplitude and rise/fall time Some FET input amplifiers exhibit the peculiar behavior of having a larger slew rate when presented with smaller input voltage steps and slower edge rates due to a change in bias conditions in the input stage of the amplifier under these circumstances. This phenomena is most commonly seen when FET input amplifiers are used as voltage followers. As this behavior is typically undesirable, the THS4601 has been designed to avoid these issues. Larger amplitudes lead to higher slew rates, as would be anticipated, and fast edges do not degrade the slew rate of the device. 20 www.ti.com 
 SLOS388B − OCTOBER 2001 − REVISED JUNE 2002 APPLICATION INFORMATION power dissipation and thermal characteristics The THS4601 does not incorporate automatic thermal shutoff protection, so the designer must take care to ensure that the design does not violate the absolute maximum junction temperature of the device. Failure may result if the absolute maximum junction temperature of 150°C is exceeded. The thermal characteristics of the device are dictated by the package and the PC board. Maximum power dissipation for a given package can be calculated using the following formula. P Dmax + T max–T q JA A Where: PDmax is the maximum power dissipation (W) Tmax is the absolute maximum junction temperature (°C) TA is the ambient temperature (°C) θJA is the thermal coefficient from the silicon junctions to the ambient air (°C/W) For systems where heat dissipation is more critical, the THS4601 is offered in an 8-pin SOIC with PowerPAD. The thermal coefficient for the SOIC PowerPAD is substantially improved over the traditional SOIC. Maximum power dissipation levels are depicted in the graph for the two packages. The data for the 8DDA package assumes a board layout that follows the PowerPAD layout guidelines. MAXIMUM POWER DISSIPATION vs TEMPERATURE Maximum Power Dissipation − W 3 θJA = 170°C/W for 8D, θJA = 66.6°C/W for 8DDA 2.5 2 8DDA Package 1.5 1 8D Package 0.5 0 −40 −20 0 20 40 60 80 Ambient Temperature − °C Figure 40 When determining whether or not the device satisfies the maximum power dissipation requirement, it is important to not only consider quiescent power dissipation, but also dynamic power dissipation. Often times, this is difficult to quantify because the signal pattern is inconsistent, but an estimate of the RMS power dissipation can provide visibility into a possible problem. www.ti.com 21 
 SLOS388B − OCTOBER 2001 − REVISED JUNE 2002 APPLICATION INFORMATION PC board layout guidelines Achieving optimum performance with a high frequency amplifier requires careful attention to board layout parasitics and external component selection. Recommendations that optimize performance include the following. D Use of a ground plane—It is highly recommended that a ground plane be used on the board to provide all components with a low impedance connection to ground. However, the ground plane should be cleared around the amplifier inputs and outputs to minimize parasitic capacitance. A solid ground plane is recommended wherever possible. D Proper power supply decoupling—A 6.8 µF tantalum capacitor and a 0.1 µF ceramic capacitor should be used on each power supply node. Good performance is possible if the 6.8 µF capacitor is shared among several amplifiers, but each amplifier should have a dedicated 0.1 µF capacitor for each supply. The 0.1 µF capacitor should be placed as close to the power supply pins as possible. As the distance from the device increases, the trace inductance rises and decreases the effectiveness of the capacitor. A good design has less than 2.5 mm separating the ceramic capacitor and the power supply pin. The tantalum capacitors can be placed significantly further away from the device. D Avoid sockets—Sockets are not recommended for high-speed amplifiers. The lead inductance associated with the socket pins often leads to stability problems. Direct soldering to a printed-circuit board yields the best performance. D Minimize trace length and place parts compactly—Shorter traces minimize stray parasitic elements of the design and lead to better high-frequency performance. D Use of surface mount passive components—Surface mount passive components are recommended due to the extremely low lead inductance and the small component footprint. These characteristics minimize problems with stray series inductance and allow for a more compact circuit layout. Compact layout reduces both parasitic inductance and capacitance in the design. D Minimize parasitic capacitance on the signal input and output pins—Parasitic capacitance on the input and output pins can degrade high frequency behavior or cause instability in the circuit. Capacitance on the inverting input or the output is a common cause of instability in high performance amplifiers, and capacitance on the noninverting input can react with the source impedance to cause unintentional band-limiting. To reduce unwanted capacitance around these pins, a window should be opened up in the signal/power layers that are underneath those pins. Power and ground planes should otherwise be unbroken. PowerPAD design considerations The THS4601 is available in a thermally-enhanced PowerPAD package. This package is constructed using a downset leadframe upon which the die is mounted (see Figure 39). This arrangement results in the lead frame exposed as a thermal pad on the underside of the package. Because this thermal pad has direct thermal contact with the die, excellent thermal performance can be achieved by providing a good thermal path away from the thermal pad. The PowerPAD package allows for both assembly and thermal management in one manufacturing operation. During the surface-mount solder operation (when the leads are being soldered), the thermal pad can also be soldered to a copper area underneath the package. Through the use of thermal paths within this copper area, heat can be conducted away from the package into either a ground plane or other heat dissipating device. The PowerPAD is electrically insulated from the amplifier circuitry, but connection to the ground plane is recommended due to the high thermal mass typically associated with a ground plane. The PowerPAD package represents a breakthrough in combining the small area and ease of assembly of surface mount with the, heretofore, awkward mechanical methods of heatsinking. 22 www.ti.com 
 SLOS388B − OCTOBER 2001 − REVISED JUNE 2002 APPLICATION INFORMATION PowerPAD design considerations (continued) DIE Side View (a) Thermal Pad DIE End View (b) Bottom View (c) NOTE A: The thermal pad is electrically isolated from all terminals in the package. Figure 41. Views of Thermally Enhanced Package Although there are many ways to properly heatsink the PowerPAD package, the following steps illustrate the recommended approach. Thermal pad area (68 mils x 70 mils) with 5 vias (Via diameter = 13 mils) Figure 42. PowerPAD PCB Etch and Via Pattern PowerPAD PCB LAYOUT CONSIDERATIONS 1. Prepare the PCB with a top side etch pattern as shown in Figure 42. There should be etch for the leads as well as etch for the thermal pad. 2. Place five vias in the area of the thermal pad. These holes should be 13 mils in diameter. Keep them small so that solder wicking through the holes does not occur during reflow. 3. Additional vias may be placed anywhere along the thermal plane outside of the thermal pad area. This helps dissipate the heat generated by the IC. These additional vias may be larger than the 13-mil diameter vias directly under the thermal pad. Larger vias are permissible here because they are not susceptible to solder wicking as the vias underneath the device. 4. Connect all vias to the internal ground plane for best thermal characteristics 5. When connecting these holes to the ground plane, do not use the typical web or spoke via connection methodology. Web connections have a high thermal resistance connection that is useful for slowing the heat transfer during soldering operations. This makes the soldering of vias that have plane connections easier. In this application, however, low thermal resistance is desired for the most efficient heat transfer. Therefore, the holes under the PowerPAD package should make their connection to the internal ground plane with a complete connection around the entire circumference of the plated-through hole. 6. The top-side solder mask should leave the terminals of the package and the thermal pad area with its five holes exposed. The bottom-side solder mask should cover the five holes of the thermal pad area. This prevents solder from being pulled away from the thermal pad area during the reflow process. 7. Apply solder paste to the exposed thermal pad area and all of the IC terminals. 8. With these preparatory steps in place, the IC is simply placed in position and run through the solder reflow operation as any standard surface-mount component. This results in a part that is properly installed. www.ti.com 23 
 SLOS388B − OCTOBER 2001 − REVISED JUNE 2002 APPLICATION INFORMATION evaluation module and applications support An evaluation board is available for quick laboratory verification of performance. An evaluation module can be ordered from Texas Instruments’ web site (www.ti.com) or from your local TI sales representative. Applications support is also available for designers. The Product Information Center (PIC) can put designers in touch with applications engineers at Texas Instruments. The PIC be contacted via the web site as well. additional reference material D PowerPAD Made Easy, application brief, Texas Instruments Literature Number SLMA004. D PowerPAD Thermally Enhanced Package, technical brief, Texas Instruments Literature Number SLMA002. D Noise Analysis of FET Transimpedance Amplifiers, application bulletin, Texas Instruments Literature Number SBOA060. D Tame Photodiodes With Op Amp Bootstrap, application bulletin, Texas Instruments Literature Number SBBA002. D Designing Photodiode Amplifier Circuits With OPA128, application bulletin, Texas Instruments Literature Number SBOA061. D Photodiode Monitoring With Op Amps, application bulletin, Texas Instruments Literature Number SBOA035. D Comparison of Noise Performance Between a FET Transimpedance Amplifier and a Switched Integrator, Application Bulletin, Texas Instruments Literature Number SBOA034. 24 www.ti.com PACKAGE OPTION ADDENDUM www.ti.com 13-Aug-2021 PACKAGING INFORMATION Orderable Device Status (1) Package Type Package Pins Package Drawing Qty Eco Plan (2) Lead finish/ Ball material MSL Peak Temp Op Temp (°C) Device Marking (3) (4/5) (6) THS4601CD ACTIVE SOIC D 8 75 RoHS & Green NIPDAU Level-1-260C-UNLIM 0 to 70 4601C THS4601ID ACTIVE SOIC D 8 75 RoHS & Green NIPDAU Level-1-260C-UNLIM -40 to 85 4601I THS4601IDDA ACTIVE SO PowerPAD DDA 8 75 RoHS & Green SN Level-1-260C-UNLIM -40 to 85 4601I THS4601IDDAG3 ACTIVE SO PowerPAD DDA 8 75 RoHS & Green SN Level-1-260C-UNLIM -40 to 85 4601I (1) The marketing status values are defined as follows: ACTIVE: Product device recommended for new designs. LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect. NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design. PREVIEW: Device has been announced but is not in production. Samples may or may not be available. OBSOLETE: TI has discontinued the production of the device. (2) RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may reference these types of products as "Pb-Free". RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption. Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of