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TLV1805-Q1
SNOSD52B – AUGUST 2018 – REVISED JANUARY 2020
TLV1805-Q1 40V, Rail-to-Rail Input, Push-Pull Output, High Voltage Automotive
Comparator with Shutdown
1 Features
3 Description
•
The TLV1805-Q1 high voltage comparator offers the
unique combination of wide supply range, push-pull
output, rail-to-rail inputs, low quiescent current,
shutdown capability and fast output response. All
these features make this comparator well-suited for
applications that require sensing at the positive or
negative voltage rails such as reverse current
protection for a smart diode controller, overcurrent
sensing, and overvoltage protection circuits where the
push-pull output stage is used to drive the gate of a
p-channel or n-channel MOSFET switch.
1
•
•
•
•
•
•
•
•
•
•
AEC-Q100 qualified with the following results:
– Device temperature grade 1: –40°C to +125°C
ambient operating temperature
– Device HBM ESD Classification Level 2
– Device CDM ESD Classification Level C6
3.3 V to 40 V supply range
Low quiescent current: 135 µA
High peak current push-pull output
Rail-to-rail inputs with phase reversal protection
Built-In hysteresis: 14mV
250ns propagation delay
Low input offset voltage: 500 µV
Shutdown with high-z output
Power-On Reset (POR)
SOT-23-6 package
2 Applications
•
•
•
•
•
•
Reverse current protection smart diode controller
Overvoltage, undervoltage, and overcurrent
detection
OR-ing MOSFET controller
MOSFET gate driver
High voltage oscillators
System monitoring for:
– Automotive infotainment & cluster
– HEV/EV & powertrain
The high peak current push-pull output stage, which
is unique for high-voltage comparators, offers the
advantage of allowing the output to actively drive the
load to either supply rail with a fast edge rate. This is
especially valuable in applications where a MOSFET
gate needs to be driven high or low quickly in order to
connect or disconnect a host from an unexpected
high voltage supply. Additional features such as low
input offset voltage, low input bias currents and HighZ shutdown make the TLV1805-Q1 flexible enough to
handle a broad range of applications. Power-On reset
prevents false outputs at power-up.
The TLV1805-Q1 is AEC-Q100 qualified in a 6-pin
SOT-23 package and is specified for operation across
the automotive Grade 1 temperature range of –40°C
to +125°C.
Device Information(1)
PART NUMBER
PACKAGE
TLV1805-Q1
SOT-23 (6)
BODY SIZE (NOM)
1.60 mm × 2.90 mm
(1) For all available packages, see the package option addendum
at the end of the datasheet.
Reverse Current Protection Using an N-Channel
MOSFET
Clock
Source
Reverse Current & Overvoltage Protection Using
P-Channel MOSFETs
P-Channel MOSFETs
Charge
Pump
iBAT
+
SD
-
S
S
D
+
TLV1805-Q1
N-Channel MOSFET
iBAT
VBAT
D
+
D2
System
Power
Q1
System
Power
Q2
VBAT
D1
SD
+
TLV1805-Q1
1
An IMPORTANT NOTICE at the end of this data sheet addresses availability, warranty, changes, use in safety-critical applications,
intellectual property matters and other important disclaimers. PRODUCTION DATA.
TLV1805-Q1
SNOSD52B – AUGUST 2018 – REVISED JANUARY 2020
www.ti.com
Table of Contents
1
2
3
4
5
6
7
Features ..................................................................
Applications ...........................................................
Description .............................................................
Revision History.....................................................
Pin Configuration and Functions .........................
Specifications.........................................................
1
1
1
2
3
4
6.1
6.2
6.3
6.4
6.5
6.6
6.7
4
4
4
4
5
5
7
Absolute Maximum Ratings ......................................
ESD Ratings..............................................................
Recommended Operating Conditions.......................
Thermal Information ..................................................
Electrical Characteristics...........................................
Switching Characteristics ..........................................
Typical Characteristics ..............................................
Detailed Description ............................................ 17
7.1 Overview ................................................................. 17
7.2 Functional Block Diagram ....................................... 17
7.3 Feature Description................................................. 17
7.4 Device Functional Modes........................................ 18
8
Application and Implementation ........................ 21
8.1 Application Information............................................ 21
8.2 Typical Applications ............................................... 21
9 Power Supply Recommendations...................... 28
10 Layout................................................................... 28
10.1 Layout Guidelines ................................................. 28
10.2 Layout Example .................................................... 28
11 Device and Documentation Support ................. 29
11.1
11.2
11.3
11.4
11.5
11.6
Documentation Support ........................................
Receiving Notification of Documentation Updates
Support Resources ...............................................
Trademarks ...........................................................
Electrostatic Discharge Caution ............................
Glossary ................................................................
29
29
29
29
29
29
12 Mechanical, Packaging, and Orderable
Information ........................................................... 29
4 Revision History
NOTE: Page numbers for previous revisions may differ from page numbers in the current version.
Changes from Revision A (May 2019) to Revision B
Page
•
Added links to Applications list ............................................................................................................................................... 1
•
Changed Output High and Low vs Supply Graphs .............................................................................................................. 10
Changes from Original (August 2018) to Revision A
•
Page
Changed Advance Information to Production Data ............................................................................................................... 1
Changes from Revision A (May 2019) to Revision B
Page
•
Added links to Applications list ............................................................................................................................................... 1
•
Changed Output High and Low vs Supply Graphs .............................................................................................................. 10
2
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SNOSD52B – AUGUST 2018 – REVISED JANUARY 2020
5 Pin Configuration and Functions
TLV1805-Q1 DBV Package
6-Pin SOT-23
Top View
1
6
V+
V-
2
5 SHDN
-IN
3
+
OUT
4
+IN
Note the reversed positions of the input pins. This differs from a similar popular pinout.
Pin Functions
PIN
NAME
NO.
IN+
4
IN–
OUT
TYPE
DESCRIPTION
I
Noninverting input
3
I
Inverting input
1
O
Output
SHDN
5
I
Shutdown (active high)
V+
6
P
Positive (highest) power supply
V–
2
P
Negative (lowest) power supply
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3
TLV1805-Q1
SNOSD52B – AUGUST 2018 – REVISED JANUARY 2020
www.ti.com
6 Specifications
6.1 Absolute Maximum Ratings
over operating free-air temperature range (unless otherwise noted) (1)
MIN
MAX
-0.3
42
V
Input pins (IN+, IN–) (2)
(V–) – 0.3
(V+) + 0.3
V
Shutdown pin (SHDN) (3)
(V–) – 0.3
(V–) + 5.5
Supply voltage: VS = (V+) – (V–)
Current into Input pins (IN+, IN–, SHDN) (2)
(V–) – 0.3
Storage temperature, Tstg
(3)
mA
(V+) + 0.3
V
150
°C
150
°C
Junction temperature, TJ
(2)
V
±10
Output (OUT)
(1)
UNIT
–65
Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. These are stress ratings
only, which do not imply functional operation of the device at these or any other conditions beyond those indicated under Recommended
Operating Conditions. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
Input terminals are diode-clamped to the power-supply rails. Input signals that can swing more than 0.3 V beyond the supply rails must
be current-limited to 10 mA or less.
Shutdown pin is diode-clamped to (V–). Input to SHDN that can swing more than 0.3 V below (V–) must be current-limited to 10 mA or
less.
6.2 ESD Ratings
VALUE
V(ESD)
(1)
Electrostatic
discharge
Human-body model (HBM), per AEC Q100-002 (1)
±2000
Charged-device model (CDM), per AEC Q100-011
±1500
UNIT
V
AEC Q100-002 indicates that HBM stressing shall be in accordance with the ANSI/ESDA/JEDEC JS-001 specification.
6.3 Recommended Operating Conditions
over operating free-air temperature range (unless otherwise noted)
MIN
MAX
Supply voltage: VS = (V+) – (V–)
3.3
40
UNIT
V
Ambient temperature, TA
–40
125
°C
6.4 Thermal Information
TLV1805-Q1
THERMAL METRIC (1)
DBV (SOT23)
UNIT
6 PINS
RθJA
Junction-to-ambient thermal resistance
166.9
°C/W
RθJC(top)
Junction-to-case (top) thermal resistance
104.2
°C/W
RθJB
Junction-to-board thermal resistance
46.8
°C/W
ψJT
Junction-to-top characterization parameter
31.3
°C/W
ψJB
Junction-to-board characterization parameter
46.6
°C/W
(1)
4
For more information about traditional and new thermal metrics, see the Semiconductor and IC Package Thermal Metrics application
report.
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SNOSD52B – AUGUST 2018 – REVISED JANUARY 2020
6.5 Electrical Characteristics
VS = 3.3 V to 40 V, VCM = VS / 2; TA = 25°C (unless otherwise noted). Typical values are at VS = 12 V and TA = 25°C,VCM = VS
/2
PARAMETER
MIN
TYP
MAX
VS = 3.3V, 12V and 40V
TEST CONDITIONS
-4.5
±0.5
4.5
VS = 3.3V, 12V and 40V, TA = –40°C to +125°C
-6.5
UNIT
VIO
Input offset voltage
dVIO/dT
Input offset voltage drift
VHYS
Input hysteresis voltage
VCM
Common-mode voltage range
IB
Input bias current
IOS
Input offset current
PSRR
Power-supply rejection ratio
VCM = V-
CMRR
Common-mode rejection ratio
(V–) < VCM < (V+)
Voltage output swing from (V–)
ISINK ≤ 5mA, input overdrive = –100 mV,
VS = 5V, TA = –40°C to +125°C
300
mV
ISOURCE ≤ 5mA, input overdrive = +100 mV,
VS = 5V, TA = –40°C to +125°C
300
mV
VOL
TA = –40°C to +125°C
±2.5
TA = -40℃ to +125℃
(V–) – 0.2
mV
μV/°C
14
Voltage output swing from (V+)
VOH
6.5
mV
(V+) + 0.2
V
0.05
pA
0.05
pA
95
dB
80
dB
Isc_source
Peak charging current (sourcing)
with output shorted to V- (1)
Vs = 5 V to 40 V
100
mA
Isc_sink
Peak dis-charging current (sinking)
with output shorted to V+ (1)
Vs = 5 V to 40 V
100
mA
Quiescent current
VS = 12 V, no load, VID = –0.1 V (output low), TA
= 25°C
135
IQ
VS =12V to 40V no load, VID = –0.1 V (output
low), TA = –40°C to +125°C
tOFF
Time to enter shutdown
CL = 15 pF
1.0
tON
Time to exit shutdown
CL = 15 pF
2.3
VSD
Shutdown input: voltage range
VSD_VIH
VSD_VIL
V s= 3.3 to 40V, TA = -40 to 125 °C
0
SHDN pin input high level
VS = 3.3 V and 40V, TA = -40 to 125 °C
2
SHDN pin input low level
VS = 3.3 V and 40V, TA = -40 to 125 °C
IB-SDH
SHDN bias current
IQ-SD
Quiescent current (Shutdown)
(1)
(2)
(2)
µA
400
µA
µs
µs
5.5
V
1.35
0.65
VS = VSD = 5.5 V
0.015
VS = 5 V, VSD = 0 V
0.001
VS = 12V; TS = 25°C; VSD > VSD_VIH Min
200
9.5
V
0.4
V
nA
nA
13
µA
Continuous short circuit can result in excessive heating and exceeding the maximum allowed junction temperature of 150°C. Please
refer to the Maximum Output Current Derating curve in the Typical Operation Plots.
The recommended voltage range if VSD is independent of VS.
6.6 Switching Characteristics
Typical values are at TA = 25°C, VS = 12 V, VCM = VS / 2; Input overdrive = 100 mV (unless otherwise noted).
PARAMETER
Propagation delay time, high-to-low
tPHL
(1)
Propagation delay time, low-to-high
tPLH
(1)
tR
Rise time
tF
Fall time
tSTART
Power-up time
(1)
(2)
TEST CONDITIONS
MIN
TYP
MAX
UNIT
CL = 15 pF
250
ns
CL = 4 nF
450
ns
CL = 15 pF
250
ns
CL = 4 nF
500
ns
20% to 80%, CL = 15 pF
18
ns
20% to 80%, CL = 4 nF
0.3
µs
20% to 80%, CL = 15 pF
20% to 80%, CL = 4 nF
(2)
10
ns
0.26
µs
45
µs
High-to-low and low-to-high refers to the transition at the input.
During power on, VS must exceed 3.3 V for tON before the output is in a correct state.
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SNOSD52B – AUGUST 2018 – REVISED JANUARY 2020
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Figure 1. Propagation Delay
Figure 2. Shutdown Timing
6
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SNOSD52B – AUGUST 2018 – REVISED JANUARY 2020
6.7 Typical Characteristics
at TA = 25°C, VS = 12 V, VCM = VS/2, and input overdrive = 100 mV (unless otherwise noted)
500
3
30 Typical Units Shown
2
400
350
1
Device Count
Offset Voltage (mV)
Distribution of 2917 Units
450
0
-1
300
250
200
150
100
-2
50
0
0
4
8
12
16
20
24
28
Supply Voltage (V)
32
36
-3
-2.5
-1.8
-1.6
-1.4
-1.2
-1
-0.8
-0.6
-0.4
-0.2
0
0.2
0.4
0.6
0.8
1
1.2
1.4
1.6
1.8
2.5
3
-3
40
Offset Voltage (mV)
Figure 4. Offset Voltage Histogram
Figure 3. Input Offset Voltage vs. Supply Voltage
3
3
30 Typical Units Shown
30 Typical Units Shown
2
Offset Voltage (mV)
Offset Voltage (mV)
2
1
0
-1
1
0
-1
-2
-2
-3
-1
0
1
2
-3
-40
3 4 5 6 7 8 9 10 11 12 13
Input Common Mode Voltage (V)
250
225
225
200
175
150
125
100
75
125°C
85°C
25°C
-40°C
50
25
-10
5
20 35 50 65
Temperature (°C)
80
95
110 125
Figure 6. Offset Voltage vs Temperature
250
0
Quiecent Supply Current (PA)
Quiecent Supply Current (PA)
Figure 5. Offset Voltage vs. Common Mode
-25
200
175
150
125
100
75
125°C
85°C
25°C
-40°C
50
25
0
0
4
VCM = V-
8
12
16
20
24
28
Supply Voltage (V)
32
36
40
Output Low (VID = -0.1 V)
Figure 7. Supply Current vs. Supply Voltage, Output Low
0
4
8
VCM = V-
12
16
20
24
28
Supply Voltage (V)
32
36
40
Output High (VID = 0.1 V)
Figure 8. Supply Current vs. Temperature, Output High
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Typical Characteristics (continued)
at TA = 25°C, VS = 12 V, VCM = VS/2, and input overdrive = 100 mV (unless otherwise noted)
200
40
40V
12V
3.3V
30
160
120
Input Bias Current (pA)
Input Bias Current (pA)
35
25
20
15
10
5
80
40
0
-40
-80
-120
0
-160
-5
-40
-200
-25
-10
5
20 35 50 65
Temperature (°C)
80
95
0
110 125
1
2
3
4
5
6
7
VCM (V)
8
9
10
11
12
TA = 125 °C
Figure 10. Input Bias Current vs. Common Mode, 125°C
0.1
0.4
0.08
0.3
0.06
Input Bias Current (pA)
Input Bias Current (pA)
Figure 9. Input Bias Current vs. Temperature
0.5
0.2
0.1
0
-0.1
-0.2
0.04
0.02
0
-0.02
-0.04
-0.3
-0.06
-0.4
-0.08
-0.5
-0.1
0
1
2
3
4
5
6
7
8
9
10
11
12
0
1
2
3
VCM (V)
6
7
VCM (V)
8
9
10
11
12
Figure 11. Input Bias Current vs. Common Mode, 25°C
Figure 12. Input Bias Current vs. Common Mode, -40°C
20
20
18
16
14
12
10
8
6
125 °C
85 °C
25 °C
-40 °C
4
2
0
0
4
8
12
16
20
24
28
Supply Voltage (V)
32
36
40
VCM = V-
18
16
14
12
10
8
6
125 °C
85 °C
25 °C
-40 °C
4
2
0
0
4
8
12
16
20
24
28
Supply Voltage (V)
32
36
40
VCM = V-
Figure 13. Shutdown Supply Current vs. Supply Votlage
8
5
TA = -40 °C
Shutdown Supply Current per Channel ( PA)
Shutdown Supply Current per Channel ( PA)
TA = 25 °C
4
Figure 14. Shutdown Supply Current vs. Supply Votlage
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Typical Characteristics (continued)
at TA = 25°C, VS = 12 V, VCM = VS/2, and input overdrive = 100 mV (unless otherwise noted)
1.85
0.8
Referenced to V-
24V
12V
5V
3.3V
1.55
1.4
1.25
1.1
5V
3.3V
0.7
0.65
0.6
0.55
0.8
-40
-25
-10
5
20 35 50 65
Temperature (°C)
80
95
0.5
-40
110 125
Figure 15. Shutdown Voltage High Threshold vs.
Temperature
-25
-10
5
20 35 50 65
Temperature (°C)
80
95
110 125
Figure 16. Shutdown Voltage Low Threshold vs.
Temperature
2
0.8
Referenced to V-
125°C
85°C
50°C
25°C
0°C
-40°C
Referenced to V-
-40°C
0°C
25°C
50°C
85°C
125°C
0.75
Shutdown VSD_VIL (V)
1.8
Shutdown VSD_VIH (V)
24V
12V
0.75
0.95
1.6
1.4
1.2
0.7
0.65
0.6
0.55
1
0.5
0
5
10
15
20
25
Supply Voltage (V)
30
35
40
0
Figure 17. Shutdown Voltage High Threshold vs. Supply
Voltage
5
10
15
20
25
Supply Voltage (V)
30
35
40
Figure 18. Shutdown Voltage Low Threshold vs. Supply
Voltage
12
18
125°C
85°C
10
Shutdown Input Bias Current (pA)
Shutdown Input Bias Current (nA)
40V
36V
Referenced to V-
Shutdown VSD_VIL (V)
Shutdown VSD_VIH (V)
1.7
40V
36V
8
6
4
2
0
-2
25°C
-40°C
16
14
12
10
8
6
4
2
0
-2
0
1
2
3
4
Shutdown Voltage (V)
5
6
Figure 19. Shutdown Input Bias Current vs. Shutdown Input
Voltage, High Temperatures
0
1
2
3
4
Shutdown Voltage (V)
5
6
Figure 20. Shutdown Input Bias Current vs. Shutdown Input
Voltage, Low Temperatures
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Typical Characteristics (continued)
20
10
5
100
5V
12V
21V
30V
40V
90
80
2
1
0.5
Output Current (mA)
Shutdown Input Bias Current (nA)
at TA = 25°C, VS = 12 V, VCM = VS/2, and input overdrive = 100 mV (unless otherwise noted)
0.2
0.1
0.05
0.02
0.01
0.005
70
60
50
40
30
20
0.002
0.001
-40
10
-25
-10
5
20 35 50 65
Temperature (°C)
80
95
0
-40
110 125
-25
-10
VSD = 5 V
110 125
0.10
0.08
0.07
0.06
0.05
0.04
0.03
0.02
0.08
0.07
0.06
0.05
0.04
0.03
0.02
0.01
0.01
0.00
0.00
0
4
8
12
16
20
24
28
Supply Voltage (V)
32
36
40
ISINK = 1 mA
0
4
8
12
16
20
24
28
Supply Voltage (V)
32
36
40
ISOURCE = 1 mA
Figure 23. Output Low Voltage vs. Supply Voltage
Figure 24. Output High Voltage vs. Supply Voltage
10
10
125 °C
85 °C
50 °C
25 °C
0 °C
-40 °C
Output Voltage from V+ (V)
1
125 °C
85°C
25 °C
0 °C
-40 °C
0.09
Output Voltage from V+ (V)
125 °C
85 °C
25 °C
0 °C
-40 °C
0.09
Output Voltage from V- (V)
95
Figure 22. Maximum Continuous Output Current vs.
Ambient Temperature
0.10
Output Voltage from V- (V)
20 35 50 65 80
Ambient Temperature (°C)
DBV SOT-23-6 Package
Figure 21. Shutdown Input Bias Current vs. Temperature
100m
10m
1m
100P
100m
1
10
Output Sinking Current (mA)
100
VS = 3.3 V
1
125 °C
85 °C
50 °C
25 °C
0 °C
-40 °C
100m
10m
1m
100m
1
10
Output Sourcing Current (mA)
100
VS = 3.3 V
Figure 25. Output Voltage vs. Output Sinking Current
at 3.3V
10
5
Figure 26. Output Voltage vs. Output Sourcing Current
at 3.3V
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Typical Characteristics (continued)
at TA = 25°C, VS = 12 V, VCM = VS/2, and input overdrive = 100 mV (unless otherwise noted)
10
125 °C
85 °C
50 °C
25 °C
0 °C
-40 °C
1
Output Voltage from V+ (V)
Output Voltage from V- (V)
10
100m
10m
1m
100P
100m
1
10
Output Sinking Current (mA)
1
100m
10m
1m
100m
100
VS = 12 V
10
125 °C
85 °C
50 °C
25 °C
0 °C
-40 °C
1
Output Voltage from V+ (V)
Output Voltage from V- (V)
100
Figure 28. Output Voltage vs. Output Sourcing Current
at 12V
10
100m
10m
1m
100P
100m
1
10
Output Sinking Current (mA)
125 °C
85 °C
50 °C
25 °C
0 °C
-40 °C
1
100m
10m
1m
100m
100
VS = 40 V
1
10
Output Sourcing Current (mA)
100
VS = 40 V
Figure 29. Output Voltage vs. Output Sinking Current
at 40V
Figure 30. Output Voltage vs. Output Sourcing Current
at 40V
25
25
20
20
15
10
3V
5V
12V
24V
40V
5
Hysteresis Voltage (mV)
Hystersis Voltage (mV)
1
10
Output Sourcing Current (mA)
VS = 12 V
Figure 27. Output Voltage vs. Output Sinking Current
at 12V
0
-40
125 °C
85 °C
50 °C
25 °C
0 °C
-40 °C
15
10
125°C
85°C
25°C
-40°C
5
0
-25
-10
5
20 35 50 65
Temperature (°C)
80
95
110 125
Figure 31. Hysteresis vs. Temperature
2
6
10
14
18
22
26
Supply Voltage (V)
30
34
38
Figure 32. Hysteresis vs. Supply Voltage
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Typical Characteristics (continued)
at TA = 25°C, VS = 12 V, VCM = VS/2, and input overdrive = 100 mV (unless otherwise noted)
25
1000
Distribution of 2917 Units
900
800
700
Device Count
15
10
600
500
400
300
125°C
85°C
25°C
-40°C
5
200
100
15
14.8
12
14.6
11
14.4
10
14.2
4
5
6
7
8
9
Common Mode Voltage (V)
14
3
13.8
2
13.6
1
13
0
13.4
0
0
13.2
Hystersis Voltage (mV)
20
Hysteresis (mV)
Figure 33. Hysteresis vs Common-Mode Voltage
Figure 34. Hysteresis Histogram
1000
-40 °C
0 °C
25 °C
50 °C
85 °C
125 °C
800
700
600
500
400
300
200
100
20
30 40 50
VS = 3.3 V
70 100
200 300
Input Overdrive Voltage (mV)
Propagation Delay, High to Low (ns)
Propagation Delay, Low to High (ns)
1000
125 °C
85 °C
50 °C
25 °C
0 °C
-40 °C
800
700
600
500
400
300
200
100
20
500 700 1000
CL = 15pF
VS = 3.3 V
Figure 35. TPLH Response Time vs. Overdrive
at 3.3V
500
400
300
200
30 40 50
70 100
200 300
Input Overdrive Voltage (mV)
500 700 1000
CL = 15pF
Propagation Delay, High to Low (ns)
Propagation Delay, Low to High (ns)
500 700 1000
CL = 15pF
1000
-40 °C
0 °C
25 °C
50 °C
85 °C
125 °C
800
700
600
VS = 5 V
-40 °C
0 °C
25 °C
50 °C
85 °C
125 °C
800
700
600
500
400
300
200
100
20
30 40 50
VS = 5 V
Figure 37. TPLH Response Time vs. Overdrive
at 5V
12
70 100
200 300
Input Overdrive Voltage (mV)
Figure 36. TPHL Response Time vs. Overdrive
at 3.3V
1000
100
20
30 40 50
70 100
200 300
Input Overdrive Voltage (mV)
500 700 1000
CL = 15pF
Figure 38. TPHL Response Time vs. Overdrive
at 5V
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Typical Characteristics (continued)
at TA = 25°C, VS = 12 V, VCM = VS/2, and input overdrive = 100 mV (unless otherwise noted)
1000
-40 °C
0 °C
25 °C
50 °C
85 °C
125 °C
800
700
600
500
400
300
200
100
20
30 40 50
VS = 12 V
70 100
200 300
Input Overdrive Voltage (mV)
Propagation Delay, High to Low (ns)
Propagation Delay, Low to High (ns)
1000
-40 °C
0 °C
25 °C
50 °C
85 °C
125 °C
800
700
600
500
400
300
200
100
20
500 700 1000
CL = 15pF
VS = 12 V
Figure 39. TPLH Response Time vs.
Overdrive at 12V
500 700 1000
CL = 15pF
2000
-40 °C
0 °C
25 °C
50 °C
85 °C
125 °C
1000
800
700
600
500
400
300
200
20
30 40 50
VS = 40 V
70 100
200 300
Input Overdrive Voltage (mV)
Propagation Delay, High to Low (ns)
Propagation Delay, Low to High (ns)
70 100
200 300
Input Overdrive Voltage (mV)
Figure 40. TPHL Response Time vs. Overdrive
at 12V
2000
-40 °C
0 °C
25 °C
50 °C
85 °C
125 °C
1000
800
700
600
500
400
300
200
20
500 700 1000
CL = 15pF
900
700
600
500
-40 °C
0 °C
25 °C
50 °C
85 °C
125 °C
300
20
30 40 50
VS = 3.3 V
70 100
200 300
Input Overdrive Voltage (mV)
500 700 1000
CL = 4nF
Propagation Delay, High to Low (ns)
1000
900
800
500 700 1000
CL = 15pF
-40 °C
0 °C
25 °C
50 °C
85 °C
125 °C
800
700
600
500
400
300
20
30 40 50
VS = 3.3 V
Figure 43. TPLH Response Time vs. Overdrive
at 3.3V
70 100
200 300
Input Overdrive Voltage (mV)
Figure 42. TPHL Response Time vs. Overdrive
at 40V
1000
400
30 40 50
VS = 40 V
Figure 41. TPLH Response Time vs. Overdrive
at 40V
Propagation Delay,Low to High (ns)
30 40 50
70 100
200 300
Input Overdrive Voltage (mV)
500 700 1000
CL = 4nF
Figure 44. TPHL Response Time vs. Overdrive
at 3.3V
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Typical Characteristics (continued)
at TA = 25°C, VS = 12 V, VCM = VS/2, and input overdrive = 100 mV (unless otherwise noted)
1000
125 °C
85 °C
50 °C
25 °C
0 °C
-40 °C
900
800
700
600
500
400
300
20
30 40 50
VS = 5 V
70 100
200 300
Input Overdrive Voltage (mV)
Propagation Delay, High to Low (ns)
Propagation Delay,Low to High (ns)
1000
125 °C
85 °C
50 °C
25 °C
0 °C
-40 °C
900
800
700
600
500
400
300
20
500 700 1000
CL = 4nF
VS = 5 V
Figure 45. TPLH Response Time vs. Overdrive
at 5V
800
700
600
500
400
30 40 50
VS = 12 V
70 100
200 300
Input Overdrive Voltage (mV)
Propagation Delay, High to Low (ns)
Propagation Delay,Low to High (ns)
500 700 1000
CL = 4nF
1000
125 °C
85 °C
50 °C
25 °C
0 °C
-40 °C
900
300
20
125 °C
85 °C
50 °C
25 °C
0 °C
-40 °C
900
800
700
600
500
400
300
20
500 700 1000
CL = 4nF
30 40 50
VS = 12 V
Figure 47. TPLH Response Time vs.
Overdrive at 12V
70 100
200 300
Input Overdrive Voltage (mV)
500 700 1000
CL = 4nF
Figure 48. TPHL Response Time vs. Overdrive
at 12V
3000
125 °C
85 °C
50 °C
25 °C
0 °C
-40 °C
2000
1000
900
800
700
600
500
20
30 40 50
VS = 40 V
70 100
200 300
Input Overdrive Voltage (mV)
500 700 1000
CL = 4nF
Propagation Delay, High to Low (ns)
3000
Propagation Delay,Low to High (ns)
70 100
200 300
Input Overdrive Voltage (mV)
Figure 46. TPHL Response Time vs. Overdrive
at 5V
1000
125 °C
85 °C
50 °C
25 °C
0 °C
-40 °C
2000
1000
900
800
700
600
500
20
30 40 50
VS = 40 V
Figure 49. TPLH Response Time vs. Overdrive
at 40V
14
30 40 50
70 100
200 300
Input Overdrive Voltage (mV)
500 700 1000
CL = 4nF
Figure 50. TPHL Response Time vs. Overdrive
at 40V
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Typical Characteristics (continued)
1000
1000
500
300
200
500
300
200
100
100
Falltime (ns)
Risetime (ns)
at TA = 25°C, VS = 12 V, VCM = VS/2, and input overdrive = 100 mV (unless otherwise noted)
50
30
20
10
50
30
20
10
125 °C
85 °C
25 °C
- 40°C
5
3
2
1
0.01 0.02
0.05
0.1 0.2 0.3 0.5
1
Capacitive Load (nF)
2
125 °C
85 °C
25 °C
-40 °C
5
3
2
1
0.01 0.02
3 4 5 67 10
VS = 3.3 V
2
3 4 5 67 10
Figure 52. tFALL vs. Capacitive Load
at 3.3V
1000
1000
500
300
200
500
300
200
100
100
Falltime (ns)
Risetime (ns)
0.1 0.2 0.3 0.5
1
Capacitive Load (nF)
VS = 3.3 V
Figure 51. tRISE vs. Capacitive Load
at 3.3V
50
30
20
10
50
30
20
10
125 °C
85 °C
25 °C
- 40°C
5
3
2
1
0.01 0.02
0.05
0.1 0.2 0.3 0.5
1
Capacitive Load (nF)
2
125 °C
85 °C
25 °C
-40 °C
5
3
2
1
0.01 0.02
3 4 5 67 10
VS = 5 V
0.05
0.1 0.2 0.3 0.5
1
Capacitive Load (nF)
2
3 4 5 67 10
VS = 5 V
Figure 53. tRISE vs. Capacitive Load
at 5V
Figure 54. tFALL vs. Capacitive Load
at 5V
1000
1000
500
300
200
500
300
200
100
100
Falltime (ns)
Risetime (ns)
0.05
50
30
20
10
50
30
20
10
125 °C
85 °C
25 °C
- 40°C
5
3
2
1
0.01 0.02
0.05
0.1 0.2 0.3 0.5
1
Capacitive Load (nF)
2
3 4 5 67 10
VS = 12 V
125 °C
85 °C
25 °C
-40 °C
5
3
2
1
0.01 0.02
0.05
0.1 0.2 0.3 0.5
1
Capacitive Load (nF)
2
3 4 5 67 10
VS = 12 V
Figure 55. tRISE vs. Capacitive Load
at 12V
Figure 56. tFALL vs. Capacitive Load
at 12V
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Typical Characteristics (continued)
3000
2000
3000
2000
1000
1000
500
500
200
200
Falltime (ns)
Risetime (ns)
at TA = 25°C, VS = 12 V, VCM = VS/2, and input overdrive = 100 mV (unless otherwise noted)
100
50
20
10
100
50
20
10
125 °C
85 °C
25 °C
- 40°C
5
2
1
0.01 0.02
0.05
0.1 0.2 0.3 0.5
1
Capacitive Load (nF)
2
125 °C
85 °C
25 °C
-40 °C
5
2
1
0.01 0.02
3 4 5 67 10
VS = 40 V
3 4 5 67 10
10
-40 °C
25 °C
85 °C
125 °C
9
8
-40 °C
25 °C
85 °C
125 °C
9
8
7
Turn-Off Time (Ps)
Turn-On Time (Ps)
2
Figure 58. tFALL vs. Capacitive Load
at 40V
10
6
5
4
3
7
6
5
4
3
2
2
1
1
0
0
0
4
8
12
16
20
24
28
Supply Voltage (V)
32
36
40
0
Figure 59. Turn-On Time vs Supply Voltage
4
8
12
16
20
24
28
Supply Voltage (V)
32
36
40
Figure 60. Turn-Off Time vs Supply Voltage
110
-40 °C
0 °C
25 °C
85 °C
125 °C
90
80
70
60
50
40
30
20
0
5
10
15
20
25
Supply Voltage (V)
30
35
Figure 61. Start-Up Time vs Supply Voltage
40
Power On Reset Threshold Voltage (V)
3
100
Start-Up Time (Ps)
0.1 0.2 0.3 0.5
1
Capacitive Load (nF)
VS = 40 V
Figure 57. tRISE vs. Capacitive Load
at 40V
16
0.05
2.9
2.8
2.7
2.6
2.5
2.4
2.3
2.2
2.1
2
-40
-25
-10
5
20 35 50 65
Temperature (°C)
80
95
110 125
Figure 62. Power On Reset Voltage vs. Temperature
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7 Detailed Description
7.1 Overview
The TLV1805-Q1 comparator features a rail-to-rail inputs with a push-pull output stage that operates at supply
voltages as high as 40 V or ±20 V. The rail-to-rail input stage enables detection of signals close to the supply
and ground while the push-pull output stage creates fast transition edges to either supply rail. A low supply
current of 135 μA per channel with small, space-saving packages, makes these comparators versatile for use in
a wide range of applications, from portable to industrial.
7.2 Functional Block Diagram
VCC
IN+
+
IN-
±
OUT
SHDN
Bias
Power-on-reset
GND
7.3 Feature Description
7.3.1 Rail to Rail Inputs
The TLV1805-Q1 comparator features a CMOS input with a common-mode range that includes both supply rails.
The TLV1805-Q1 is designed to prevent phase inversion when the input pins exceed the supply voltage.
7.3.2 Power On Reset
The TLV1805-Q1 incorporates a power-on reset that holds the output in a High-Z state until the minimum
operating supply voltage has been reached for at least 20µs. After this time the output will start responding to the
inputs. This feature prevents false outputs during power-up and power-down.
7.3.3 High Power Push-Pull Output
The push-pull output stage, which is unique for high-voltage comparators, offers the advantage of allowing the
output to actively drive the load to either supply rail with a fast edge rate. A high output sink and source peak
current of over 100mA allows quickly charging and dis-cahrging capacitive loads such as cables and power
MOSFET gates. Caution must be taken to ensure that the package power dissipation is not exceeded when
switching at these high supply voltages. See Figure 22 for the output current derating curve.
7.3.4 Shutdown Function
The TLV1805-Q1 has a logic level SHDN input. When the shutdown SHDN input is 1.4V above V-, the TLV1805Q1 is disabled. When disabled, the output becomes high impedance (Hi-Z), and the supply current drops to
below 10µA. The input bias current remains unchanged. Voltages may still be applied to the comparator inputs
as long as V+ power is still applied and the applied input voltages are still within the specified input voltage
range.
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Feature Description (continued)
CAUTION
The maximum voltage on the shutdown pin is +5.5V referred to V-, regardless of
supply voltage. Connect the SHDN pin to V- if shutdown is not used. Do not float the
SHDN pin.
A high value pull-up or pull-down resistor on the output may be required if a specific logic level is required during
shutdown (when the output is High-Z). This prevents logic inputs from floating to illegal states when the
comparator output is in High-Z mode.
Since the Shutdown threshold voltage is a tested parameter, the shutdown pin can also be used as a second
comparison input to provide a secondary measurment, such as overvoltage monitoring, as shown in the PChannel Reverse Current Protection With Overvotlage Protection circuit.
7.3.5 Internal Hysteresis
The TLV1805-Q1 contains 14mV of internal hysteresis.
The hysteresis transfer curve is shown in Figure 63. This curve is a function of three components: VTH, VOS, and
VHYST:
• VTH is the actual set voltage or threshold trip voltage.
• VOS is the internal offset voltage between VIN+ and VIN–. This voltage is added to VTH to form the actual trip
point at which the comparator must respond to change output states.
• VHYST is the internal hysteresis (or trip window) that is designed to reduce comparator sensitivity to noise
(14 mV for the TLV1805-Q1).
VTH + VOS - (VHYST / 2)
VTH + VOS
VTH + VOS + (VHYST / 2)
Figure 63. Hysteresis Transfer Curve
7.4 Device Functional Modes
7.4.1 External Hysteresis
External Hysteresis may be added to further improve response to noisy or slow-moving input signals.
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Device Functional Modes (continued)
7.4.1.1 Inverting Comparator With Hysteresis
+VCC
+5 V
R1
1 MW
VIN
5V
RLOAD
100 kW
VA
VO
VA2
VA1
0V
1.67 V
R3
1 MW
R2
1 MW
VO High
+VCC
R1
VIN
3.33 V
VO Low
+VCC
R3
R1
VA1
VA2
R2
R2
R3
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Figure 64. TLV1805-Q1 in an Inverting Configuration With Hysteresis
The inverting comparator with hysteresis requires a three-resistor network that is referenced to the comparator
supply voltage (VCC), as shown in Figure 64. When VIN at the inverting input is less than VA, the output voltage is
high (for simplicity, assume VO switches as high as VCC). The three network resistors can be represented as R1
|| R3 in series with R2. Equation 1 defines the high-to-low trip voltage (VA1).
R2
VA1 = VCC ´
(R1 || R3) + R2
(1)
When VIN is greater than VA, the output voltage is low, very close to ground. In this case, the three network
resistors can be presented as R2 || R3 in series with R1. Use Equation 2 to define the low to high trip voltage
(VA2).
R2 || R3
VA2 = VCC ´
R1 + (R2 || R3)
(2)
Equation 3 defines the total hysteresis provided by the network.
DVA = VA1 - VA2
(3)
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Device Functional Modes (continued)
7.4.1.2 Noninverting Comparator With Hysteresis
+VCC
+5 V
VREF
+2.5 V
VO
VA
VIN
RLOAD
R1
330 kW
R2
1 MW
VO High
+VCC
VO Low
VIN1
5V
R2
R1
VA = VREF
VA = VREF
R1
R2
VO
VIN2
VIN1
0V
1.675 V
3.325 V
VIN
VIN2
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Figure 65. TLV1805-Q1 in a Noninverting Configuration With Hysteresis
A noninverting comparator with hysteresis requires a two-resistor network, as shown in Figure 65, and a voltage
reference (VREF) at the inverting input. When VIN is low, the output is also low. For the output to switch from low
to high, VIN must rise to VIN1. Use Equation 4 to calculate VIN1.
VREF
VIN1 = R1 ´
+ VREF
(4)
R2
When VIN is high, the output is also high. For the comparator to switch back to a low state, VIN must drop to VIN2
such that VA is equal to VREF. Use Equation 5 to calculate VIN2.
VREF (R1 + R2) - VCC ´ R1
VIN2 =
(5)
R2
The hysteresis of this circuit is the difference between VIN1 and VIN2, as shown in Equation 6.
R1
DVIN = VCC ´
R2
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8 Application and Implementation
NOTE
Information in the following applications sections is not part of the TI component
specification, and TI does not warrant its accuracy or completeness. TI’s customers are
responsible for determining suitability of components for their purposes. Customers should
validate and test their design implementation to confirm system functionality.
8.1 Application Information
The TLV1805-Q1 family of devices can be used in a wide variety of applications, such as MOSFET gate drivers,
zero crossing detectors, window comparators, over and undervoltage detectors, and high-side voltage sense
circuits.
8.2 Typical Applications
Comparators are used to differentiate between two different signal levels. For example, a comparator
differentiates between an over-temperature and normal-temperature condition. However, noise or signal variation
at the comparison threshold causes multiple transitions. This application example sets upper and lower
hysteresis thresholds to eliminate the multiple transitions caused by noise.
RH
576 NŸ
5V
5V
RX
100 NŸ
+
VOUT
±
RY
100 NŸ
+
±
VIN
Figure 66. Comparator with Hysteresis
8.2.1 Design Requirements
The design requirements are as follows:
• Supply voltage: 5 V
• Input: 0 V to 5 V
• Lower threshold (VL) = 2.3 V ±0.1 V
• Upper threshold (VH) = 2.7 V ±0.1 V
• VH – VL = 2.4 V ±0.1 V
• Low-power consumption
8.2.2 Detailed Design Procedure
A small change to the comparator circuit can be made to add hysteresis. Hysteresis uses two different threshold
voltages to avoid the multiple transitions introduced in the previous circuit. The input signal must exceed the
upper threshold (VH) to transition low, or below the lower threshold (VL) to transition high.
Figure 66 illustrates hysteresis on a comparator. Resistor RH sets the hysteresis level.
When the output is at a logic high (5 V), RH is in parallel with RX. This configuration drives more current into Ry,
and raises the threshold voltage (VH) to 2.7 V. The input signal must drive above VH = 2.7 V to cause the output
to transition to logic low (0 V).
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Typical Applications (continued)
When the output is at logic low (0 V), Rh is in parallel with Ry. This configuration reduces the current into Ry, and
reduces the threshold voltage to 2.3 V. The input signal must drive below VL = 2.3 V to cause the output to
transition to logic high (5 V).
For more details on this design, refer to Precision Design TIPD144, Comparator with Hysteresis Reference
Design.
8.2.3 Application Curve
Figure 67 shows the upper and lower thresholds for hysteresis. The upper threshold is 2.76 V and the lower
threshold is 2.34 V, both of which are close to the design target.
Figure 67. TLV1805-Q1 Upper and Lower Threshold with Hysteresis
22
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Typical Applications (continued)
8.2.4 Reverse Current Protection Using MOSFET and TLV1805-Q1
An N-Channel or P-Channel MOSFET may be used to protect against reverse current. Reverse current is
defined as current flowing from the load (VLOAD) to the source (VBATT). Both the P-Channel and N-Channel
circuits work on the same basic principle, where a comparator monitors the voltage across the MOSFET's
Source and Drain terminals (monitoring VDS). The described circuits also protect against reverse voltage.
MOSFET
³2II´
VBATT
Intrinsic
Body Diode
Source
Drain
³2Q´
VLOAD
RDS(ON)
Gate
+
Load
Battery
Figure 68. Simplified Operational Theory
When the current is flowing from the battery (VBATT) to the load (VLOAD), the battery voltage will be higher than
the load voltage due to voltage drop across the MOSFET caused by the RDS(ON) or the intrinsic body diode
forward voltage drop. The comparator will detect this and turn "on" the MOSFET so that the load current is now
flowing through the low loss RDS(ON) path.
In a reverse current condition, VLOAD will be higher than VBATT. The comparator will detect this and drive the gate
to set VGS = 0 to turn "off" the MOSFET (non-conducting). The body diode is reverse biased and will block
current flow.
For a P-Channel MOSFET, the gate must be driven at least 4V or more below the battery voltage to turn "on" the
MOSFET.
For a N-Channel MOSFET, the gate must be driven 4V or more above the battery voltage to turn "on" the
MOSFET. If a higher voltage is not available in the system, a charge pump is usually required to generate a
voltage higher than the battery voltage to provide the necessary positive gate drive voltage.
8.2.4.1 Minimum Reverse Current
There is a minimum amount of reverse current that is needed to trip the comparator. To detect this reverse
current, a voltage must be dropped across the MOSFET (VMEAS).
When the MOSFET is off, VGS will be in the -600mV to -1V range due to the forward voltage drop (VF) of the
MOSFET body diode. Response to this large voltage will be immediate.
However, with the MOSFET "on" (conducting), the current required to create the trip voltage will be much
greater. The trip voltage drop required across the MOSFET RDS(ON) will be the comparator offset voltage plus half
of the hysteresis.
The maximum offset voltage of the TLV1805-Q1 is 5mV with a typical hysteresis of 14mV. The trip voltage can
be calculated from:
VTRIP = VOS(max) + ( VHYST / 2) = 5 mV + 7 mV = 12 mV
(7)
The actual current trip point will depend on the MOSFET RDS(ON) and VGS drive level. Assuming the MOSFET
has a 22 mΩ on resistance, the trip current is found from:
ITRIP = VTRIP / RDS(ON) = 12 mV / 22 mΩ = 546mA
(8)
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Typical Applications (continued)
8.2.4.2 N-Channel Reverse Current Protection Circuit
In order to turn "on" the N-Channel MOSFET, the MOSFET gate must be brought "High" above VBATT. If a higher
voltage is not available, a charge pump circuit is required to provide the comparator with a supply voltage above
VBATT.
J1
External Clock
Input
EXT
INT
Clock
Select
C1
100V
1µF
Charge
Pump
R2
D2
BAT46
D1
BAT64
VBATT_NCH
Comparator
Supply Clamp
R1
47 Ÿ
D3
15 V
C3
35V
10µF
C2
100V
1µF
GND
3
4
D4
BAT64
6
±
1
+
5
2
R3
Q1
SQ4850EY
N-Channel
47 Ÿ
G
R4
100 NŸ
VLOAD_OUT_NCH
S
Input
Protection
D5
SMAJ28CA
10 NŸ
U1
TLV1805-Q1
D
Internal Oscillator Output ± 10kHz
C6
50V
0.22µF
C7
50V
0.22µF
Oscillator Supply
R6
56 NŸ
R7
56 NŸ
R5
56 NŸ
3
4
R8
56 NŸ
C4
50V
0.22µF
C5
1nF
Oscillator
Circuit
D6
DB2430100L
GND
6
±
+
JP1
Short
1
5
2 U2
TLV1805-Q1
Figure 69. N-Channel Reverse Current Schematic with Oscillator
C1, D1, D2 & C2 form the charge pump. The AC drive signal is applied through C1 into the charge pump. The
result is a voltage across C2 that is approximately equal to the peak-to-peak amplitude of the AC waveform,
minus 700mV. If a 12Vpp waveform is applied to the C1 input, 11.3V will be generated across C2. This voltage is
on top of the VBATT voltage, so the voltage seen from the D2-C2 junction ground is 23.3V. This provides the
needed higher voltage to drive the MOSFET and power the comparator.
An external oscillator source may be used, such as the gate drive output of a switcher, system clock or any
avaialbe clock source in the 1kHz to 10MHz range. The charge pump should be fed by a 50 percent duty cycle
square wave source of 5Vpp or more. Since the input capacitor of the charge-pump effectively AC-couples the
input, the oscillator may be ground referenced.
R1 and D3 form the comparator supply clamp to limit the gate drive to prevent exceeding the VGS(MAX) of the
MOSFET during an overvotlage event. R1 must be sized to dissapate any expected overvoltage.
D4 and R2 clamp the input should VBATT drop below VLOAD (as in a supply reversal).
The output diode D6 is used to anchor the output during light or floating loads. At light or no loads, there is a
possibility the MOSFET could turn on due to the comparator offset voltage. The diode provides enough of a
negative leakage to turn the MOSFET off.
8.2.4.2.1 N-Channel Oscillator Circuit
The oscillation frequency is determined by R5 and C5. The default configuration oscillates around 10kHz
(depending on RC component tolerances). For further information on selecting these RC values, please see the
Engineers Cookbook Circuit entitled Oscillator Circuit (SNOA990). Do note that R5 does present an AC load to
the oscillator output, and should be sized appropriately to minimize the peak charging currents of C5 (use large
resistors and small capacitors).
24
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Typical Applications (continued)
The output amplitude is roughly equivalent to the VLOAD voltage minus the TLV1805-Q1 output saturation
(approximately 300mV). With a maximum supply voltage of 40V for the TLV1805-Q1, the oscillator circuit is
capable of generating up to 39Vpp!
The TLV1805-Q1 oscillator typically starts oscillating when VLOAD reaches 2.8V, though full specified operation
does not occur until 3.3V.
For more information, please see the TLV1805-Q1 Evaluation Module Users Guide TLV1805-Q1 Evaluation
Module Users Guide (SNOU158).
8.2.5 P-Channel Reverse Current Protection Circuit
Figure 70 shows the P-Channel circuit. In order to turn "on" the P-Channel MOSFET, the gate must be brought
"Low" below VBATT . To accomplish this, the comparators Inverting input is tied to the battery side of the MOSFET
to set the output low during forward current.
R2
Q1
SQJ459EP
P-Channel
VBATT_PCH
D1
SMAJ28CA
GND
4
C2
50V
0.22µF
C3
50V
0.22µF
VLOAD_OUT_PCH
S
D
Input
Protection
47 Ÿ
R3
100 NŸ
G
R1
10 NŸ
3
D2
BAT64
U1
TLV1805-Q1
6
1
+
±
5
C1
25V
10µF
2
D3
15 V
D4
BAT64
D5
DB2430100L
GND
R4
560 Ÿ
Figure 70. P-Channel Reverse Current Schematic
This design implements a "floating ground" topology, using D3, D4 and R12, to allow for clamping the
comparator supply voltage as to not exceed the VGS(MAX) of the MOSFET. During a reverse voltage or supply
drop, D4 also prevents C1 from discharging to allow some standby time to keep the comparator powered during
the event.
During "normal" forward current operation, the quiescent current of the comparator circuit flows through D4 and
R4. D3 provides the clamping during an overvoltage event.
R4 is sized to allow for minimum voltage drop during "normal" operation, but also to allow for dissipation during
overvoltage events. R4 will see the battery voltage minus the D3 Zener voltage during an overvoltage event.
Since the comparator supply voltage is clamped by D3, the maximum battery voltage is determined by the power
dissipated by R4 and the VDS(MAX) of the MOSFET.
R2 limits the gate current should there be any transients and should be a low value to allow the peak currents
needed to drive the MOSFET gate capacitance. R3 provides the pull-down needed when the comparator output
goes high-Z during power-off to ensure the gate is pulled to zero volts to turn off the MOSFET.
R1 and D2 clamp the input voltage should the VBATT input go below the floating ground Voltage (such as in a
battery reversal). A bonus feature is that during a reverse battery voltage condition, D2 and R1 pull the floating
ground down towards the negative potential, providing power to the comparator during reverse voltage.
The output clamp diode D5 is used to anchor the output during light or floating loads. At light or no loads, there is
a possibility the MOSFET could turn on due to the comparator offset voltage. The diode provides enough of a
negative leakage to turn the MOSFET off.
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Typical Applications (continued)
If shutdown of the comparator circuit is desired, a transistor or MOSFET switch can be placed between the
ground end of R4 and ground. The MOSFET will be in body diode mode when the comparator is disabled.
8.2.6 P-Channel Reverse Current Protection With Overvotlage Protection
The SHDN pin can be utilized to add Overvotlage Protection (OVP) by adding a second MOSFET, zener diode
and resistor, as shown in Figure 71.
G
VBATT
R2
100 NŸ
Q1
P-Channel
R3
22 Ÿ
S
D
4
R1
3
10 NŸ
U1
TLV1805-Q1
6
1
+
±
5
G
Q2
P-Channel
S
VLOAD_OUT
D
C1
50V
1µF
2
ZD1
13 V
GND
GND
Shutdown when > 1.35V
RPD
13.7 NŸ
Figure 71. Adding Overvoltage Protection Using SHDN Pin
When the SHDN pin is pulled 1.35 V above V-, the comparator is placed in shutdown. During shutdown, the
comparator output goes Hi-Z and R2 pulls the gate and source together to turn off the MOSFET (VGS = 0 V).
RPD pulls the SHDN pin low while the Zener diode is not conducting (< VZ). When ZD1 reaches its breakdown
voltage and starts conducting, it will pull RPD up to a voltage calculated to place >1.35 V on the shutdown pin.
The Zener diode ZD1 should be chosen so that the breakdown voltage (VB) is 1.35 V below the desired
overvoltage point. The Zener should have low sub-threshold leakage and a sharp knee, such as the low power
1N47xx or BZD series.
The pull-down resistor RPD should be chosen to create 1.35 V at the desired Zener diode current (usually 100uA
to 1mA) at the Zener breakdown voltage. Actual resistor value should be verified on the bench due to differences
in actual Zener diode threshold voltages.
If a 14.3 V overvotlage trip point (OVP) is desired, the Zener Diode voltage should be 12.95 V. We will choose a
100uA Zener current. The required Zener diode breakdown voltage is determined from:
VB = VOV - 1.35 V = 14 .3V - 1.35 V = 12.95 V
RPD = 1.35 V / 100 µA = 13.5 kΩ (13.7kΩ nearest value)
(9)
(10)
Resistor RPD may be split into two resistors to create a voltage divider if more precise trip points are needed, or
a more convenient zener voltage is desired. Series voltage references can also be used if more accuracy is
desired. A second resistor in series with the Zener or reference can extend the breakdown voltage.
The maximum voltage allowed on the Shutdown pin is 5.5V, so make sure the highest VBATT voltage does not
exceed 5.5 V.
Note that the above circuit, as shown for simplicity, does not protect against reverse voltage. Reverse clamping
diodes would be needed on the -IN, SHDN and Load Output. Also make sure VBATT does not exceed the
VGS(MAX) of the MOSFET.
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Typical Applications (continued)
8.2.7 ORing MOSFET Controller
The previous reverse current circuits may be combined to create an OR'ing supply controller, utilizing either the
P-Channel or N-Channel topologies.
For the previous P-Channel circuit, if no negative input voltages are possible, and the input voltage is below the
MOSFET's VGS(MAX) , then D3, D4 and R4 may be eliminated (the D2 anode, U1 pins 2 and 5, and C1 can be
directly grounded).
For the N-Channel circuit, the oscillator drive can be shared between the channels, or eliminated if a higher
system voltage is available to provide the higher votlage.
Charge Pump
Gate
Drive
+
+
TLV1805-Q1
SD
Q1
Power
Supply
#1
Charge Pump
System
Power
Gate
Drive
+
+
TLV1805-Q1
SD
Q2
Power
Supply
#2
Figure 72. N-Channel OR'ing MOSFET Controller
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9 Power Supply Recommendations
The TLV1805-Q1 family of devices is specified for operation from 3.3 V to 40 V (±1.65 to ±20 V); many
specifications apply from –40°C to +125°C. Parameters that can exhibit significant variance with regard to
operating voltage or temperature are presented in the Typical Characteristics section.
CAUTION
Supply voltages larger than 40 V can permanently damage the device; see the
Recommended Operating Conditions section.
Place 0.1-μF bypass capacitors close to the power-supply pins to reduce errors coupling in from noisy or highimpedance power supplies. For more detailed information on bypass capacitor placement; see the Layout
Guidelines section.
The TLV1805-Q1 does not contain reverse battery protection, so applying negative voltage to the supply pins
must be avoided. The TLV1805-Q1 cannot withstand ISO 16750 type waveforms alone and requires external
protection circuitry.
10 Layout
10.1 Layout Guidelines
Comparators are very sensitive to input noise. For best results, maintain the following layout guidelines:
• Use a printed circuit board (PCB) with a good, unbroken low-inductance ground plane. Proper grounding (use
of ground plane) helps maintain specified performance of the TLV1805-Q1 family of devices.
• To minimize supply noise, place a decoupling capacitor (0.1-μF ceramic, surface-mount capacitor) as close
as possible to VS as shown in Figure 73.
• On the inputs and the output, keep lead lengths as short as possible to avoid unwanted parasitic feedback
around the comparator. Keep inputs away from the output.
• Solder the device directly to the PCB rather than using a socket.
• Run the ground pin ground trace under the device up to the bypass capacitor, shielding the inputs from the
outputs.
10.2 Layout Example
Figure 73. Oscillator Circuit Layout Example
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11 Device and Documentation Support
11.1 Documentation Support
11.1.1 Related Documentation
Precision Design, Comparator with Hysteresis Reference Design— TIDU020
Reference Design, Window Comparator Reference Design— TIPD178
Application Report, Using Comparators in Reverse Current Applications— SNOAA23
Application Report, TLV1805-Q1 EVM ISO Testing Results— SNOAA13
EVM Users Guide, TLV1805-Q1 Reverse Current Evaluation Module Users Guide— SNOU158
11.2 Receiving Notification of Documentation Updates
To receive notification of documentation updates, navigate to the device product folder on ti.com. In the upper
right corner, click on Alert me to register and receive a weekly digest of any product information that has
changed. For change details, review the revision history included in any revised document.
11.3 Support Resources
TI E2E™ support forums are an engineer's go-to source for fast, verified answers and design help — straight
from the experts. Search existing answers or ask your own question to get the quick design help you need.
Linked content is provided "AS IS" by the respective contributors. They do not constitute TI specifications and do
not necessarily reflect TI's views; see TI's Terms of Use.
11.4 Trademarks
E2E is a trademark of Texas Instruments.
11.5 Electrostatic Discharge Caution
This integrated circuit can be damaged by ESD. Texas Instruments recommends that all integrated circuits be handled with
appropriate precautions. Failure to observe proper handling and installation procedures can cause damage.
ESD damage can range from subtle performance degradation to complete device failure. Precision integrated circuits may be more
susceptible to damage because very small parametric changes could cause the device not to meet its published specifications.
11.6 Glossary
SLYZ022 — TI Glossary.
This glossary lists and explains terms, acronyms, and definitions.
12 Mechanical, Packaging, and Orderable Information
The following pages include mechanical packaging and orderable information. This information is the most
current data available for the designated devices. This data is subject to change without notice and revision of
this document. For browser-based versions of this data sheet, refer to the left-hand navigation.
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PACKAGE OPTION ADDENDUM
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10-Dec-2020
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
(4/5)
(6)
TLV1805QDBVRQ1
ACTIVE
SOT-23
DBV
6
3000
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 125
1ULF
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of
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