TPS40060
TPS40061
8
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SLUS543F – DECEMBER 2002 – REVISED JUNE 2013
WIDE-INPUT SYNCHRONOUS BUCK CONTROLLER
Check for Samples: TPS40060, TPS40061
FEATURES
APPLICATIONS
•
•
•
•
•
•
•
•
1
2
•
•
•
•
•
•
•
Operating Input Voltage 10 V to 55 V
Input Voltage Feed-Forward Compensation
< 1% Internal 0.7-V Reference
Programmable Fixed-Frequency, Up to 1-MHz
Voltage Mode Controller
Internal Gate Drive Outputs for High-Side PChannel and Synchronous N-Channel
MOSFETs
16-Pin PowerPAD™ Package (θJC = 2°C/W)
Thermal Shutdown
Externally Synchronizable
Programmable High-Side Sense Short Circuit
Protection
Programmable Closed-Loop Soft-Start
TPS40060 Source Only/TPS40061 Source/Sink
Networking Equipment
Telecom Equipment
Base Stations
Servers
DESCRIPTION
The TPS40060 and TPS40061 are high-voltage, wide
input (10 V to 55 V) synchronous, step-down
converters.
This family of devices offers design flexibility with a
variety of user programmable functions, including;
soft-start, UVLO, operating frequency, voltage feedforward,
high-side
current
limit,
and
loop
compensation.
These
devices
are
also
synchronizable to an external supply.
The TPS40060 and TPS40061 incorporate MOSFET
gate drivers for external P-channel high-side and Nchannel synchronous rectifier (SR) MOSFETs. Gate
drive logic incorporates anti-cross conduction circuitry
to prevent simultaneous high-side and synchronous
rectifier conduction.
SIMPLIFIED APPLICATION DIAGRAM
1
TPS40060PWP
KFF
ILIM
16
2
RT
VIN
15
3
BP5
HDRV
14
4
SYNC
BPN10
13
5
SGND
SW
12
6
SS/SD
BP10
11
7
VFB
LDRV
10
8
COMP
PGND
9
VIN
+
VOUT
UDG-02157
1
2
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
PowerPAD is a trademark of Texas Instruments.
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of the Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
Copyright © 2002–2013, Texas Instruments Incorporated
TPS40060
TPS40061
SLUS543F – DECEMBER 2002 – REVISED JUNE 2013
www.ti.com
This integrated circuit can be damaged by ESD. Texas Instruments recommends that all integrated circuits be handled with
appropriate precautions. Failure to observe proper handling and installation procedures can cause damage.
ESD damage can range from subtle performance degradation to complete device failure. Precision integrated circuits may be more
susceptible to damage because very small parametric changes could cause the device not to meet its published specifications.
ORDERING INFORMATION
TA
SOURCE
–40°C to 85°C
(1)
(2)
PACKAGE (1)
PART NUMBER
Plastic HTSSOP (PWP)
TPS40060PWP
Plastic HTSSOP (PWP)
TPS40061PWP
LOAD CURRENT
(2)
SOURCE/SIN (2)
The PWP package is also available taped and reeled. Add an R suffix to the device type (i.e., TPS40060PWPR). See the Application
Information of the data sheet for PowerPAD drawing and layout information.
See Application Information section.
ABSOLUTE MAXIMUM RATINGS
over operating free-air temperature range unless otherwise noted (1)
TPS40060
TPS40061
VIN
60 V
VFB, SS/SD, SYNC
VIN
Input voltage range
–0.3 V to 6 V
–0.3 V to 60 V or VIN+5 V
(whichever is less)
SW
SW. transient < 50 ns
–2.5 V
–0.3 V to 6 V
VOUT
Output voltage range
COMP, RT, KFF, SS
IIN
Input current
KFF
5 mA
IOUT
Output current
RT
200 µA
TJ
Operating junction temperature range
–40°C to 125°C
Tstg
Storage temperature
–55°C to 150°C
Lead temperature 1,6 mm (1/16 inch) from case for 10 seconds
(1)
260°C
Stresses beyond those listed under absolute maximum ratings may cause permanent damage to the device. These are stress ratings
only, and functional operation of the device at these or any other conditions beyond those indicated under recommended operating
conditions is not implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
RECOMMENDED OPERATING CONDITIONS
MIN NOM MAX
VIN
Input voltage
TA
Operating free-air temperature
UNIT
10
55
V
–40
85
°C
PWP PACKAGE (1)(2)
(TOP VIEW)
KFF
RT
BP5
SYNC
SGND
SS/SD
VFB
COMP
2
1
2
3
4
5
6
7
8
THERMAL
PAD
16
15
14
13
12
11
10
9
ILIM
VIN
HDRV
BPN10
SW
BP10
LDRV
PGND
(1)
For more information on the PWP package, refer to TI Technical Brief (SLMA002).
(2)
PowerPAD™ heat slug must be connected to SGND (Pin 5), or electrically isolated from all other pins.
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SLUS543F – DECEMBER 2002 – REVISED JUNE 2013
ELECTRICAL CHARACTERISTICS
TA = –40°C to 85°C, VIN = 24 Vdc, RT = 165 kΩ, IKFF = 113 µA, fSW = 300 kHz, all parameters at zero power dissipation (unless
otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
INPUT SUPPLY
VIN
Input voltage range, VIN
10
55
V
1.5
2.5
mA
4.5
5.0
5.5
V
270
300
330
kHz
OPERATING CURRENT
IDD
Quiescent current
Output drivers not switching
5-V REFERENCE
VBP5
Input voltage
OSCILLATOR/RAMP GENERATOR (1)
fOSC
Frequency
VRAMP PWM ramp voltage
(2)
VIH
High-level input voltage, SYNC
VIL
Low-level input voltage, SYNC
ISYNC
Input current, SYNC
2
2
Pulse width, SYNC
VRT
V
0.8
Pulse amplitude = 5 V
RT voltage
2.32
Maximum duty cycle
VFB = 0 V, 100 kHz ≤ fSW≤ 1 MHz
Minimum duty cycle
VFB ≥ 0.75 V
VKFF
Feed-forward voltage
IKFF
Feed-forward current operating range (2)
5
10
2.50
2.68
50
µA
ns
85%
V
98%
0%
3.35
3.50
20
3.65
V
1100
µA
SS/SD (SOFT START)
ISS
Soft-start source current
1.5
2.3
2.9
µA
VSS
Soft-start clamp voltage
3.1
3.7
4.0
V
tDSCH
Discharge time
CSS = 220 pF
1.6
2.2
2.9
tSS
Soft-start time
CSS = 220 pF, 0 V ≤ VSS ≤ 1.6 V
120
155
235
90
130
160
170
210
260
mV
V
µs
SS/SD (SHUTDOWN)
VSD
Shutdown threshold voltage
VEN
Device action threshold voltage
Hysteresis
80
10-V REFERENCE
VBP10
Input voltage
9.0
9.7
10.7
TA = 25°C
0.698
0.700
0.704
0°C ≤ TA ≤ 85°C
0.690
0.700
0.707
0.690
0.700
0.715
ERROR AMPLIFIER
VFB
Feedback regulation voltage
V
GBW
Gain bandwidth
3
5
MHz
AVOL
Open loop gain
60
80
dB
IOH
High-level output source current
VCOMP = 2.0 V, VFB = 0 V
1.5
4.0
IOL
Low-level output sink current
VCOMP = 2.0 V, VFB = 1 V
2.5
4.0
IBIAS
Input bias current
VFB = 0.7 V
VOH
High-level output voltage
IOH = 0.5 mA, VFB = 0 V
3.25
3.45
3.60
VOL
Low-level output voltage
IOL = 0.5 mA, VFB = 1 V
0.050
0.215
0.350
(1)
(2)
100
mA
300
nA
V
KFF current (IKFF) increases with SYNC frequency (fSYNC) and decreases with maximum duty cycle (DMAX).
Ensured by design. Not production tested.
Copyright © 2002–2013, Texas Instruments Incorporated
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TPS40060
TPS40061
SLUS543F – DECEMBER 2002 – REVISED JUNE 2013
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ELECTRICAL CHARACTERISTICS (continued)
TA = –40°C to 85°C, VIN = 24 Vdc, RT = 165 kΩ, IKFF = 113 µA, fSW = 300 kHz, all parameters at zero power dissipation (unless
otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
TA = 25°C
8.8
10.0
11.4
0°C ≤ TA ≤ 85°C
8.3
11.9
-40°C ≤ TA ≤ 0°C
7.5
11.5
UNIT
CURRENT LIMIT
ISINK
Current limit sink current
VILIM = 23.7 V, VSW = (VILIM – 0.5 V)
tDELAY
Propagation delay to output
tON
Switch leading-edge blanking pulse time (3)
tOFF
Off time during a fault
VOS
Overcurrent comparator offset voltage
VILIM = 23.7 V, VSW = (VILIM – 2 V)
330
500
275
375
µA
ns
100
7
-200
cycles
-60
50
CHDRV = 2200 pF, (VIN – VBPN10)
48
96
mV
OUTPUT DRIVER
tHFALL
High-side driver fall time (3)
(3)
tHRISE
High-side driver rise time
CHDRV = 2200 pF, (VIN – VBPN10)
36
72
tLFALL
Low-side driver fall time (3)
CLDRV = 2200 pF, BP10
24
48
tLRISE
Low-side driver rise time (3)
CLDRV = 2200 pF, BP10
48
96
VOH
High-level ouput voltage, HDRV
IHDRV = 0.1 A , (VIN – VHDRV)
1.0
1.4
VOL
Low-level ouput voltage, HDRV
IHDRV = 0.1 A , (VHDRV – VBPN10)
VOH
High-level ouput voltage, LDRV
ILDRV = 0.1 A, (VBP10 – VLDRV)
VOL
Low-level ouput voltage, LDRV
ILDRV = 0.1 A
0.75
1.0
1.5
ns
V
0.5
Minimum controllable pulse width
100
150
ns
–7.5
–8.5
–9.5
V
–6
0
6
mV
1
µA
BPN10 REGULATOR
VBPN1
Output voltage
Outputs off
0
RECTIFIER ZERO CURRENT COMPARATOR (TPS40060 ONLY)
VSW
Switch voltage
LDRV output OFF
SW NODE
ILEAK
Leakage current (3)
THERMAL SHUTDOWN
Shutdown temperature (3)
TSD
Hysteresis
165
(3)
°C
25
UNDERVOLTAGE LOCKOUT
VUVLO
Undervoltage lockout threshold voltage, BP10
RKFF = 10 kΩ
6.25
Undervoltage lockout hysteresis
VKFF
(3)
4
KFF programmable threshold voltage
6.5
7.5
0.4
RKFF = 82.5 kΩ
9
10
V
11
Ensured by design. Not production tested.
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SLUS543F – DECEMBER 2002 – REVISED JUNE 2013
Terminal Functions
TERMINAL
NAME
NO.
I/O
DESCRIPTION
BP5
3
O
5-V reference. This pin should be bypassed to ground with a 0.1-µF ceramic capacitor. This pin may be used with
an external DC load of 1 mA or less.
BP10
11
O
10-V reference used for gate drive of the N-channel synchronous rectifier. This pin should be bypassed by a 1-µF
ceramic capacitor. This pin may be used with an external DC load of 1 mA or less.
BPN10
13
O
Negative 8-V reference with respect to VIN. This voltage is used to provide gate drive for the high side P-channel
MOSFET. This pin should be bypassed to VIN with a 0.1-µF capacitor
COMP
8
I
Output of the error amplifier, input to the PWM comparator. A feedback network is connected from this pin to the
VFB pin to compensate the overall loop. The comp pin is internally clamped above the peak of the ramp to
improve large signal transient response.
HDRV
14
O
Floating gate drive for the high-side P-channel MOSFET. This pin switches from VIN (MOSFET off) to BPN10
(MOSFET on).
ILIM
16
I
Current limit pin, used to set the overcurrent threshold. An internal current sink from this pin to ground sets a
voltage drop across an external resistor connected from this pin to VIN. The voltage on this pin is compared to the
voltage drop (VIN -SW) across the high side MOSFET during conduction.
KFF
1
I
A resistor is connected from this pin to VIN to program the amount of voltage feed-forward. The current fed into
this pin is internally divided and used to control the slope of the PWM ramp.
LDRV
10
I
Gate drive for the N-channel synchronous rectifier. This pin switches from BP10 (MOSFET on) to ground
(MOSFET off).
PGND
9
RT
2
SGND
5
Signal ground reference for the device.
SS/SD
6
I
Soft-start programming pin. A capacitor connected from this pin to ground programs the soft-start time. The
capacitor is charged with an internal current source of 2.3 µA. The resulting voltage ramp on the SS pin is used as
a second non-inverting input to the error amplifier. The output voltage begins to rise when VSS/SD is approximately
0.85 V. The output continues to rise and reaches regulation when VSS/SD is approximately 1.55 V. The controller is
considered shut down when VSS/SD is 125 mV or less. All internal circuitry is inactive. The internal circuitry is
enabled when VSS/SD is 210 mV or greater. When VSS/SD is less than approximately 0.85 V, the outputs cease
switching and the output voltage (VOUT) decays while the internal circuitry remains active.
SW
12
I
This pin is connected to the switched node of the converter and used for overcurrent sensing. This pin is used for
zero current sensing in the TPS40060.
SYNC
4
I
Synchronization input for the device. This pin can be used to synchronize the oscillator to an external master
frequency.
VFB
7
I
Inverting input to the error amplifier. In normal operation the voltage on this pin is equal to the internal reference
voltage, 0.7 V.
VIN
15
I
Supply voltage for the device.
Power ground reference for the device. There should be a low-impedance connection from this point to the source
of the power MOSFET.
I
A resistor is connected from this pin to ground to set the internal oscillator ramp charging current and switching
frequency.
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TPS40060
TPS40061
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SIMPLIFIED BLOCK DIAGRAM
ILIM
16
VIN 15
11
BP10
13
BPN10
14
HDRV
12
SW
BP10
+
CLK
RT
2
SYNC
4
CLK
7
Clock
Oscillator
7
10−V Regulator
VIN
7
1V5REF
7 HDRV
7
Ramp Generator
KFF
Reference
Voltages
1
07VREF 7
1V5REF 7
3V5REF 7
BP5
HDRV
7
7 CL
3−bit up/down
Fault Counter
P-Channel
Driver
7
7
7
7
BPN10
Restart Fault
BP5
COMP
3 BP5 7
7 Fault
8
07VREF
7
VFB 7
7
+
0.85 V
+
7
7
Q
R
Q
N-Channel
Driver
7
7 07VREF
6
S
CL
+
+
SS/SD
7 BP10
SW
CLK
Zero Current Detector
(TPS40060 Only)
Restart
10 LDRV
5
S
Q
R
Q
9 PGND
UDG−02160
SGND
6
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TPS40061
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SLUS543F – DECEMBER 2002 – REVISED JUNE 2013
APPLICATION INFORMATION
The TPS40060/61 family of parts allows the user to optimize the PWM controller to the specific application.
The TPS40061 is the controller of choice for synchronous buck designs which will include most applications. It
has two quadrant operation and will source or sink output current. This provides the best transient response.
The TPS40060 operates in one quadrant and sources output current only, allowing for paralleling of converters
and ensures that one converter does not sink current from another converter. This controller also emulates a
standard buck converter at light loads where the inductor current goes discontinuous. At continuous output
inductor currents the controller operates as a synchronous buck converter to optimize efficiency.
SW NODE RESISTOR
The SW node of the converter will be negative during the dead time when both the upper and lower MOSFETs
are off. The magnitude of this negative voltage is dependent on the lower MOSFET body diode and the output
current which flows during this dead time. This negative voltage could affect the operation of the controller,
especially at low input voltages.
Therefore, a 10-Ω resistor must be placed between the lower MOSFET drain and pin 12 (SW) of the controller as
shown in Figure 14 as RSW.
SETTING THE SWITCHING FREQUENCY (PROGRAMMING THE CLOCK OSCILLATOR)
The TPS40060 and TPS40061 have independent clock oscillator and ramp generator circuits. The clock
oscillator serves as the master clock to the ramp generator circuit. The switching frequency, fSW in kHz, of the
clock oscillator is set by a single resistor (RT) to ground. The clock frequency is related to RT, in kΩ by
Equation 1 and the relationship is charted in Figure 2.
RT +
ǒ
f SW
1
17.82
10 *6
Ǔ
* 23
kW
(1)
PROGRAMMING THE RAMP GENERATOR CIRCUIT
The ramp generator circuit provides the actual ramp used by the PWM comparator. The ramp generator provides
voltage feed-forward control by varying the PWM ramp slope with line voltage, while maintaining a constant ramp
magnitude. Varying the PWM ramp directly with line voltage provides excellent response to line variations since
the PWM does not have to wait for loop delays before changing the duty cycle. (See Figure 1).
VIN
VIN
SW
SW
RAMP
VPEAK
COMP
COMP
RAMP
VVALLEY
tON1
t
d + ON
T
T1
tON2
T2
tON1 > tON2 and d1 > d2
UDG-02131
Figure 1. Voltage Feed-Forward Effect on PWM Duty Cycle
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TPS40060
TPS40061
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The PWM ramp must be faster than the master clock frequency or the PWM is prevented from starting. The
PWM ramp time is programmed via a single resistor (RKFF) pulled up to VIN. RKFF is related to RT, and the
minimum input voltage, VIN(min) through the following:
ǒ
Ǔ ǒ65.27
R KFF + VIN (min) * 3.5
RT ) 1502Ǔ (W)
where:
•
•
VIN is the desired start-up (UVLO) input voltage
RT is the timing resistor in kΩ
(2)
See the section on UVLO operation for further description.
The curve showing the feedforward impedance required for a given switching frequency, fSW, at various input
voltages is shown in Figure 3.
For low input voltage and high duty cycle applications, the voltage feed-forward may limit the duty cycle
prematurely. This does not occur for most applications. The voltage control loop controls the duty cycle and
regulates the output voltages. For more information on large duty cycle operation, refer to Application Note
(SLUA310).
FEED-FORWARD IMPEDANCE
vs
SWITCHING FREQUENCY
TIMING RESISTANCE
vs
SWITCHING FREQUENCY
600
800
700
RKFF - Feed-Forward Impedance - kΩ
RT - Timing Resistance - kΩ
500
400
300
200
100
0
VIN = 25 V
600
500
400
VIN = 15 V
300
200
VIN = 9 V
100
0
200
0
400
600
800
1000
200
400
fSW - Switching Frequency - kHz
600
800
1000
fSW - Switching Frequency - kHz
Figure 2.
Figure 3.
UVLO OPERATION
The TPS40060 and TPS40061 use both fixed and variable (user programmable) UVLO protection. The fixed
UVLO monitors the BP10 and BP5 bypass voltages. The UVLO circuit holds the soft-start low until the BP5 and
BP10 voltage rails have exceeded their thresholds and the input voltage has exceed the user programmable
undervoltage threshold.
The TPS40060 and TPS40061 use the feed-forward pin, KFF, as a user programmable low-line UVLO detection.
This variable low-line UVLO threshold compares the PWM ramp duration to the oscillator clock period. An
undervoltage condition exists if the device receives a clock pulse before the ramp has reached 90% of its full
amplitude. The ramp duration is a function of the ramp slope, which is directly related to the current into the KFF
pin. The KFF current is a function of the input voltage and the resistance from KFF to the input voltage. The KFF
resistor can be referenced to the oscillator frequency as described in Equation 3:
ǒ
Ǔ ǒ65.27
R KFF + VIN (min) * 3.5
8
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RT ) 1502Ǔ (W)
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TPS40061
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SLUS543F – DECEMBER 2002 – REVISED JUNE 2013
where:
•
•
VIN is the desired start-up (UVLO) input voltage
RT is the timing resistor in kΩ
(3)
The variable UVLO function utilizes a 3-bit full adder to prevent spurious shut-downs or turn-ons due to spikes or
fast line transients. When the adder reaches a total of seven counts in which the ramp duration is shorter the
clock cycle a powergood signal is asserted, a soft-start initiated, and the upper and lower MOSFETs are turned
off.
Once the soft-start is initiated, the UVLO circuit must see a total count of seven cycles in which the ramp
duration is longer than the clock cycle before an undervoltage condition is declared (See Figure 4).
UVLO Threshold
VIN
Clock
PWM RAMP
1 2 3 4 5 6 7
1 2
1 2 3 4 5 6 7
PowerGood
UDG-02132
Figure 4. Undervoltage Lockout Operation
UNDERVOLTAGE LOCKOUT
vs
HYSTERESIS
3.0
VUVLO - Output Voltage - V
2.5
2.0
1.5
1.0
0.5
0
10
15
20
25
30
35
40
45
50
45
VUVLO - Undervoltage Lockout Threshold - V
Figure 5.
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TPS40060
TPS40061
SLUS543F – DECEMBER 2002 – REVISED JUNE 2013
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The impedance of the input voltage can cause the input voltage, at the TPS4006x, to sag when the converter
starts to operate and draw current from the input source. Therefore, there is voltage hysteresis that prevents
nuisance shutdowns at the UVLO point.
With RT chosen to select the operating frequency and RKFF chosen to select the start-up voltage, the amount of
hysteresis voltage is shown in Figure 5.
PROGRAMMING SOFT START
TPS4006x uses a closed-loop approach to ensure a controlled ramp on the output during start-up. Soft-start is
programmed by charging an external capacitor (CSS) via an internally generated current source. The voltage on
CSS minus 0.85 V, is fed into a separate non-inverting input to the error amplifier (in addition to FB and 0.7-V
VREF). The loop is closed on the lower of the (VCSS – 0.85 V) voltage or the internal reference voltage (0.7-V
VREF). Once the (VCSS – 0.85 V) voltage rises above the internal reference voltage, regulation is based on the
internal reference. To ensure a controlled ramp-up of the output voltage the soft-start time should be greater than
the L-CO time constant as described in Equation 4.
t START w 2p
ǸL
CO
(seconds)
(4)
There is a direct correlation between tSTART and the input current required during start-up. The faster tSTART, the
higher the input current required during start-up. This relationship is describe in more detail in the section titled,
Programming the Current Limit, which follows. The soft-start capacitance, CSS, is described in Equation 5.
For applications in which the VIN supply ramps up slowly, (typically between 50 ms and 100 ms) it may be
necessary to increase the soft-start time to between approximately 2 ms and 5 ms to prevent nuisance UVLO
tripping. The soft-start time should be longer than the time that the VINsupply transitions between 6 V and 7 V.
2.3 mA
C SS +
t START (Farads)
0.7 V
(5)
10
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SLUS543F – DECEMBER 2002 – REVISED JUNE 2013
PROGRAMMING CURRENT LIMIT
This device uses a two-tier approach for overcurrent protection. The first tier is a pulse-by-pulse protection
scheme. Current limit is implemented on the high-side MOSFET by sensing the voltage drop across the
MOSFET when the gate is driven low. The MOSFET voltage is compared to the voltage dropped across a
resistor connected from VIN pin to the ILIM pin when driven by a constant current sink. If the voltage drop across
the MOSFET exceeds the voltage drop across the ILIM resistor, the switching pulse is immediately terminated.
The MOSFET remains off until the next switching cycle is initiated.
The second tier consists of a fault counter. The fault counter is incremented on an overcurrent pulse and
decremented on a clock cycle without an overcurrent pulse. When the counter reaches seven (7) a restart is
issued and seven soft-start cycles are initiated. Both the upper and lower MOSFETs are turned off during this
period. The counter is decremented on each soft-start cycle. When the counter is decremented to zero, the PWM
is re-enabled. If the fault has been removed the output starts up normally. If the output is still present the counter
counts seven overcurrent pulses and re-enters the second-tier fault mode. See Figure 7 for typical overcurrent
protection waveforms.
The minimum current limit setpoint (ILIM) depends on tSTART, CO, VO, and the load current at start-up (ILOAD).
é (C ´ VO )ù
I LIM = ê O
ú + I LOAD (A )
ë t START û
(6)
The current limit programming resistor (RILIM) is calculated using Equation 7. Care must be taken in choosing the
values used for VOS and ISINK in the equation. In order to ensure the output current at the overcurrent level, the
minimum value of ISINK and the maximum value of VOS must be used.
I OC RDS(on)[max]
V
R ILIM +
) OS (W)
I SINK
I SINK
where:
•
•
•
ISINK is the current into the ILIM pin and is nominally 8.3 µA, minimum
IOC is the overcurrent setpoint which is the DC output current plus one-half of the peak inductor current
VOS is the overcurrent comparator offset and is 50 mV maximum
(7)
BP5, BP10 AND BPN10 INTERNAL VOLTAGE REGULATOR
Start-up characteristics of the BP5, BP10 and BPN10 regulators are shown in Figure 7. Slight variations in the
BP5 occurs dependent upon the switching frequency. Variation in the BPN10 and BP10 regulation characteristics
is also based on the load presented by switching the external MOSFETs.
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HDRV
CLOCK
tBLANKING
VILIM
VVIN-VSW
SS
7 CURRENT LIMIT TRIPS
(HDRV CYCLE TERMINATED BY CURRENT LIMIT TRIP)
7 SOFT-START CYCLES
UDG-02136
Figure 6. Typical Current Limit Protection Waveforms
INTERNAL REGULATOR OUTPUT VOLTAGE
vs
INPUT VOLTAGE
12
BP10
VBPx - Output Voltage - V
10
8
BP5
6
BPN10
4
2
0
2
4
6
8
10
12
VIN - Input Voltage - V
Figure 7.
CALCULATING THE BPN10 AND BP10V BYPASS CAPACITOR
The BPN10 capacitance provides energy for the high-side driver. The BPN10 capacitor should be a good quality,
high-frequency capacitor. The size of the bypass capacitor depends on the total gate charge of the high-side
MOSFET and the amount of droop allowed on the bypass capacitor. The BPN10 capacitance is described in
Equation 8.
12
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C BPN10 +
Qg
DV
(F)
(8)
The 10-V reference pin, BP10V needs to provide energy for the synchronous MOSFET gate drive via the BP10V
capacitor. Neglecting any efficiency penalty, the BP10V capacitance is described in Equation 9.
Q gSR
C BP10V +
(F)
(9)
DV
SYNCHRONIZING TO AN EXTERNAL SUPPLY
The TPS4006x can be synchronized to an external clock through the SYNC pin. The SW node rises on the
falling edge of the SYNC signal. The synchronization frequency should be in the range of 20% to 30% higher
than its programmed free-run frequency. The clock frequency at the SYNC pin replaces the master clock
generated by the oscillator circuit. Pulling the SYNC pin low programs the TPS4006x to freely run at the
frequency programmed by RT.
Internally, the SYNC pin has a pull-down current between 5 µA and 10 µA. In order to synchronize the device to
an external clock signal, the SYNC pin has to be overdriven from the external clock circuit. Normal logic gates or
an external MOSFET with a pull-up resistor of 10 kΩ is adequate.
Internally there is a delay of between approximately 50 ns and 100 ns from the time the SYNC pin is pulled low
and the HDRV signal goes low to turn on the upper MOSFET. Additionally, there is some delay as the MOSFET
gate charges to turn on the upper MOSFET, typically between 20 ns and 50 ns.
The higher synchronization must be factored in when programming the PWM ramp generator circuit. If the PWM
ramp is interrupted by the SYNC pulse, a UVLO condition is declared and the PWM becomes disabled. Typically
this is of concern under low-line conditions only. In any case, RKFF needs to be adjusted for the higher switching
frequency. In order to specify the correct value for RKFF at the synchronizing frequency, calculate a 'dummy'
value for RT that would cause the oscillator to run at the synchronizing frequency. Do not use this value of RT in
the design.
R T(dummy) +
ǒ
1
17.82
f SYNC
10 *6
Ǔ
* 23
kW
where:
•
fSYNC is the synchronous frequency in kHz
(10)
Use the value of RT(dummy) to calculate the value for RKFF.
(
)(
)
RKFF = VIN(min ) - 3.5 V ´ 65.27 ´ RT(dummy ) + 1502 W
where:
•
RT(dummy) is in kΩ
(11)
This value of RKFF ensures that UVLO is not engaged when operating at the synchronization frequency.
SELECTING THE INDUCTOR VALUE
The inductor value determines the magnitude of ripple current in the output capacitors as well as the load current
at which the converter enters discontinuous mode. Too large an inductance results in lower ripple current but is
physically larger for the same load current. Too small an inductance results in larger ripple currents and a greater
number of (or more expensive output capacitors for) the same output ripple voltage requirement. A good
compromise is to select the inductance value such that the converter doesn't enter discontinuous mode until the
load approximated somewhere between 10% and 30% of the rated output. The inductance value is described in
Equation 12.
L+
ǒV IN * V OǓ
VIN
DI
VO
f SW
(H)
where:
•
•
VO is the output voltage
ΔI is the peak-to-peak inductor current
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CALCULATING THE OUTPUT CAPACITANCE
The output capacitance depends on the output ripple voltage requirement, output ripple current, as well as any
output voltage deviation requirement during a load transient.
The output ripple voltage is a function of both the output capacitance and capacitor ESR. The worst case output
ripple is described in Equation 13.
DV + DI
ƪ
ESR )
ǒ
1
CO
8
Ǔƫ
f SW
ǒVP*PǓ
(13)
The output ripple voltage is typically between 90% and 95% due to the ESR component.
The output capacitance requirement typically increases in the presence of a load transient requirement. During a
step load, the output capacitance must provide energy to the load (light to heavy load step) or absorb excess
inductor energy (heavy-to-light load step) while maintaining the output voltage within acceptable limits. The
amount of capacitance depends on the magnitude of the load step, the speed of the loop and the size of the
inductor.
Stepping the load from a heavy load to a light load results in an output overshoot. Excess energy stored in the
inductor must be absorbed by the output capacitance. The energy stored in the inductor is described in
Equation 14 and Equation 15.
E L + 1 L I 2 (J)
2
(14)
where:
I2 +
ƪǒI
Ǔ
OH
2
* ǒI OLǓ
ƫ
2
ǒ(Amperes)2Ǔ
where:
•
•
IOH is the output current under heavy load conditions
IOL is the output current under light load conditions
(15)
Energy in the capacitor is given by the following equation:
E C + 1 C V2 (J)
2
(16)
where:
2
V 2 + ǒVfǓ * ǒV iǓ
2
ǒVolts2Ǔ
where:
•
•
Vf is the final peak capacitor voltage
Vi is the initial capacitor voltage
(17)
By substituting Equation 15 into Equation 14, substituting Equation 17 into Equation 16, setting Equation 14
equal to Equation 16 and solving for CO yields the following equation.
L
CO +
ƪǒI Ǔ * ǒI Ǔ ƫ
ƪǒV Ǔ * ǒV Ǔ ƫ
2
2
OH
OL
2
f
(F)
2
i
(18)
Loop Compensation
Voltage-mode buck-type converters are typically compensated using Type III networks. Since the TPS40060 and
TPS40061 use voltage feedforward control, the gain of the PWM modulator with voltage feedforward circuit must
be included. The generic modulator gain is described in Figure 8.
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Duty cycle, D, varies from 0 to 1 as the control voltage, VC, varies from the minimum ramp voltage to the
maximum ramp voltage, VS. Also, for a synchronous buck converter, D = VO / VIN. To get the control voltage to
output voltage modulator gain in terms of the input voltage and ramp voltage,
V
V
VO
V
D + O + C or
+ IN
VS
V IN
VS
VC
(19)
With the voltage feedforward function, the ramp slope is proportional to the input voltage. Therefore, the
moderator DC gain is independent of the change of input voltage. For the TPS40060 and TPS40061 the
modulator dc gain is shown in Equation 20, with VIN(min) as the minimum input voltage required to cause the ramp
excursion to reach the maximum ramp amplitude of VRAMP.
æ VIN(min ) ö
æ VIN(min ) ö
÷ or AMOD(dB ) = 20 ´ log ç
÷
AMOD = ç
ç VRAMP ÷
ç VRAMP ÷
è
ø
è
ø
(20)
Calculate the Poles and Zeros
For a buck converter using voltage mode control there is a double pole due to the output L-CO. The double pole
is located at the frequency calculated in Equation 21.
1
f LC +
(Hz)
Ǹ
2p
L CO
(21)
There is also a zero created by the output capacitance, CO, and its associated ESR. The ESR zero is located at
the frequency calculated in Equation 22.
1
fZ +
(Hz)
2p ESR CO
(22)
Calculate the value of RBIAS to set the output voltage, VO.
0.7 ´ R1
RBIAS =
W
VO - 0.7
(23)
The maximum crossover frequency (0 dB loop gain) is set by Equation 24.
f
f C + SW (Hertz)
4
(24)
Typically, fC is selected to be close to the midpoint between the L-CO double pole and the ESR zero. At this
frequency, the control to output gain has a –2 slope (-40 dB/decade), while the Type III topology has a +1 slope
(20 dB/decade), resulting in an overall closed loop –1 slope (–20 dB/decade). Figure 9 shows the modulator
gain, L-C filter, output capacitor ESR zero, and the resulting response to be compensated.
A Type III topology, shown in Figure 10, has two zero-pole pairs in addition to a pole at the origin. The gain and
phase boost of a Type III topology is shown in Figure 11. The two zeros are used to compensate the L-CO
double pole and provide phase boost. The double pole is used to compensate for the ESR zero and provide
controlled gain roll-off. In many cases the second pole can be eliminated and the amplifier's gain roll-off used to
roll-off the overall gain at higher frequencies.
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MODULATOR GAIN
vs
SWITCHING FREQUENCY
PWM MODULATOR RELATIONSHIPS
ESR Zero, + 1
Modulator Gain - dB
AMOD = VIN(min) / VRAMP
VS
VC
Resultant, - 1
D = VC / VS
LC Filter, - 2
100
1k
10 k
fSW - Switching Frequency - Hz
Figure 8.
Figure 9.
C2
(optional)
C1
R3
−1
+1
0 dB
R2
−1
GAIN
−90°
C3
100 k
VFB
R1
180°
7
8
VOUT
COMP
+
RBIAS
VREF
PHASE
−270°
UDG−02189
Figure 10. Type III Compensation of Configuration
Figure 11. Type III Compensation Gain and Phase
The poles and zeros for a type III network are described in Equation 25.
1
1
f Z1 +
(Hz)
f Z2 +
(Hz)
2p R2 C1
2p R1 C3
f P1 +
2p
1
R2
C2
(Hz)
f P2 +
2p
(25)
1
R3
C3
(Hz)
The value of R1 is somewhat arbitrary, but influences other component values. A value between 50kΩ and
100kΩ usually yields reasonable values.
The unity gain frequency is described in Equation 26.
1
fC +
(Hertz)
2p R1 C2 G
where
•
16
G is the reciprocal of the modulator gain at fC
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The modulator gain as a function of frequency at fC, is described in Equation 27.
AMOD(f) + AMOD
ǒ Ǔ
f LC
fC
2
and
G+
1
AMOD(f)
(27)
Care must be taken not to load down the output of the error amplifier with the feedback resistor, R2, that is too
small. The error amplifier has a finite output source and sink current which must be considered when sizing R2.
Too small a value does not allow the output to swing over its full range.
VC (max)
R2 (MIN) +
(W) + 3.45 V + 1.725 kW
I SOURCE (min)
2.0 mA
(28)
dv/dt INDUCED TURN-ON
MOSFETs are susceptible to dv/dt turn-on particularly in high-voltage (VDS) applications. The turn-on is caused
by the capacitor divider that is formed by CGD and CGS. High dv/dt conditions and drain-to-source voltage, on the
MOSFET causes current flow through CGD and causes the gate-to-source voltage to rise. If the gate-to-source
voltage rises above the MOSFET threshold voltage, the MOSFET turns on, resulting in large shoot-through
currents. Therefore the SR MOSFET should be chosen so that the CGD capacitance is smaller than the CGS
capacitance. A 2-Ω to 5-Ω resistor in the upper MOSFET gate lead shapes the turn-on and dv/dt of the SW node
and helps reduce the induced turn-on.
HIGH-SIDE MOSFET POWER DISSIPATION
The power dissipated in the external high-side MOSFET is comprised of conduction and switching losses. The
conduction losses are a function of the IRMS current through the MOSFET and the RDS(on) of the MOSFET. The
high-side MOSFET conduction losses are defined by Equation 29.
P COND + ǒI RMSǓ
2
R DS(on)
ǒ1 ) TCR ƪT J * 25 CƫǓ
O
(W)
where:
•
TCR is the temperature coefficient of the MOSFET RDS(on)
(29)
The TCR varies depending on MOSFET technology and manufacturer but is typically ranges between 3500
ppm/°C and 1000 ppm/°C.
The IRMS current for the high side MOSFET is described in Equation 30.
Ǹd ǒAmperes
Ǔ
I
+I
RMS
RMS
O
(30)
The switching losses for the high-side MOSFET are described in Equation 31.
P SW(fsw) + ǒVIN
I OUT
t SWǓ
f SW (Watts)
where:
•
•
•
IO is the DC output current
tSW is the switching rise time, typically < 20 ns
fSW is the switching frequency
(31)
Typical switching waveforms are shown in Figure 12.
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ID2
IO
ID1
d
}
∆I
1-d
BODY DIODE
CONDUCTION
BODY DIODE
CONDUCTION
SW
0
SYNCHRONOUS
RECTIFIER ON
ANTI-CROSS
CONDUCTION
HIGH SIDE ON
UDG-02179
Figure 12. Inductor Current and SW Node Waveforms
The maximum allowable power dissipation in the MOSFET is determined by the following equation.
PT +
ǒTJ * TAǓ
q JA
(W)
(32)
where:
P T + PCOND ) PSW(fsw) (W)
(33)
and ΘJA is the package thermal impedance.
SYNCHRONOUS RECTIFIER MOSFET POWER DISSIPATION
The power dissipated in the synchronous rectifier MOSFET is comprised of three components: RDS(on) conduction
losses, body diode conduction losses, and reverse recovery losses. RDS(on) conduction losses can be found
using Equation 29 and the RMS current through the synchronous rectifier MOSFET is described in Equation 34.
Ǹ1 * d ǒA
Ǔ
I
+I
RMS
O
RMS
(34)
The body-diode conduction losses are due to forward conduction of the body diode during the anti-cross
conduction delay time. The body diode conduction losses are described by Equation 35.
P DC + 2 I O V F t DELAY f SW (W)
where:
•
•
VF is the body diode forward voltage
tDELAY is the delay time just before the SW node rises
(35)
The 2-multiplier is used because the body-diode conducts twice during each cycle (once on the rising edge and
once on the falling edge)
The reverse recovery losses are due to the time it takes for the body diode to recovery from a forward bias to a
reverse blocking state. The reverse recovery losses are described in Equation 36.
P RR + 0.5 Q RR V IN f SW (W)
where:
•
QRR is the reverse recovery charge of the body diode
(36)
The total synchronous rectifier MOSFET power dissipation is described in Equation 37.
P SR + PDC ) PRR ) PCOND (W)
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TPS40060/TPS40061 POWER DISSIPATION
The power dissipation in the TPS40060 and TPS40061 is largely dependent on the MOSFET driver currents and
the input voltage. The driver current is proportional to the total gate charge, Qg, of the external MOSFETs. Driver
power (neglecting external gate resistance, (refer to the second reference in the REFERENCES section) can be
calculated from Equation 38.
PD = Qg ´ VDR ´ fSW (W / driver)
(38)
And the total power dissipation in the device, assuming MOSFETs with similar gate charges for both the highside and synchronous rectifier is described in Equation 39.
2 PD
PT +
) IQ
V IN (W)
V DR
ǒ
Ǔ
(39)
or
P T + ƪǒ2
Qg
f SWǓ ) I Qƫ
V IN (W)
where:
•
IQ is the quiescent operating current (neglecting drivers)
(40)
The maximum power capability of the device's PowerPad package is dependent on the layout as well as air flow.
The thermal impedance from junction to air, assuming 2 oz. copper trace and thermal pad with solder and no air
flow.
ΘJA = 36.51°C/W
The maximum allowable package power dissipation is related to ambient temperature by Equation 36.
Substituting Equation 32 into Equation 40 and solving for fSW yields the maximum operating frequency for the
TPS40060 and TPS40061. The result is:
æ é (TJ - TA ) ù
ö
-I ÷
çê
ç ê (qJA ´ VIN )úú Q ÷
ë
û
ø Hz
fSW = è
( )
2 ´ Qg
(
)
(41)
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LAYOUT CONSIDERATIONS
THE PowerPAD™ PACKAGE
The PowerPAD package provides low thermal impedance for heat removal from the device. The PowerPAD
derives its name and low thermal impedance from the large bonding pad on the bottom of the device. For
maximum thermal performance, the circuit board must have an area of solder-tinned-copper underneath the
package. The dimensions of this area depends on the size of the PowerPAD package. For a 16-pin TSSOP
(PWP) package the dimensions of the circuit board pad are 5 mm x 3.4 mm. The dimensions of the package pad
are shown in Figure 13.
Thermal vias connect this area to internal or external copper planes and should have a drill diameter sufficiently
small so that the via hole is effectively plugged when the barrel of the via is plated with copper. This plug is
needed to prevent wicking the solder away from the interface between the package body and the solder-tinned
area under the device during solder reflow. Drill diameters of 0.33 mm (13 mils) works well when 1-oz copper is
plated at the surface of the board while simultaneously plating the barrel of the via. If the thermal vias are not
plugged when the copper plating is performed, then a solder mask material should be used to cap the vias with a
diameter equal to the via diameter of 0.1 mm minimum. This capping prevents the solder from being wicked
through the thermal vias and potentially creating a solder void under the package. Refer to PowerPAD Thermally
Enhanced Package (see REFERENCES section) for more information on the PowerPAD package.
Figure 13. PowerPAD Dimensions
MOSFET PACKAGING
MOSFET package selection depends on MOSFET power dissipation and the projected operating conditions. In
general, for a surface-mount applications, the DPAK style package provides the lowest thermal impedance (θJA)
and, therefore, the highest power dissipation capability. However, the effectiveness of the DPAK depends on
proper layout and thermal management. The θJAspecified in the MOSFET data sheet refers to a given copper
area and thickness. In most cases, a thermal impedance of 40°C/W requires one square inch of 2-ounce copper
on a G-10/FR-4 board. Lower thermal impedances can be achieved at the expense of board area. Please refer
to the selected MOSFET's data sheet for more information regarding proper mounting.
GROUNDING AND CIRCUIT LAYOUT CONSIDERATIONS
The device provides separate signal ground (SGND) and power ground (PGND) pins. It is important that circuit
grounds are properly separated. Each ground should consist of a plane to minimize its impedance if possible.
The high power noisy circuits such as the output, synchronous rectifier, MOSFET driver decoupling capacitor
(BP10), and the input capacitor should be connected to PGND plane at the input capacitor.
Sensitive nodes such as the FB resistor divider, RT, and ILIM should be connected to the SGND plane. The
SGND plane should only make a single point connection to the PGND plane.
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Component placement should ensure that bypass capacitors (BP10, BP5, and BPN10) are located as close as
possible to their respective power and ground pins. Also, sensitive circuits such as FB, RT and ILIM should not
be located near high dv/dt nodes such as HDRV, LDRV, BPN10, and the switch node (SW).
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DESIGN EXAMPLE
•
•
•
•
•
•
•
Input voltage: 18 VDC to 55 VDC
Output voltage: 3.3 V ±2%
Output current: 5 A (maximum, steady-state), 7 A (surge, 10-ms duration, 10% duty cycle maximum)
Output ripple: 33 mVP-P at 5 A
Output load response: 0.3 V => 10% to 90% step load change
Operating temperature: –40°C to 85°C
fSW = 130 kHz
1. Calculate maximum and minimum duty cycles
d MIN +
V O(min)
VIN(max)
+ 0.0588
d MAX +
V O(max)
V IN(min)
++ 0.187
(42)
2. Select switching frequency
The switching frequency is based on the minimum duty cycle ratio and the propagation delay of the current limit
comparator. In order to maintain current limit capability, the on time of the upper MOSFET, tON, must be greater
than 330 ns (see Electrical Characteristics table). Therefore
V O(min)
t
+ ON
or
V IN(max)
T SW
(43)
ȡǒ VO(min) Ǔȣ
ȧ VIN(max) ȧ
1 +f
+
ȧ TON ȧ
SW
T SW
ȧ
ȧ
Ȥ
Ȣ
(44)
Using 400 ns to provide margin,
f SW + 0.0588 + 147 kHz
400 ns
(45)
Since the oscillator can vary by 10%, decrease fSW, by 10%
fSW = 0.9 × 147 kHz = 130 kHz
and therefore choose a frequency of 130 kHz.
3. Select ΔI
In this case ΔI is chosen so that the converter enters discontinuous mode at 20% of nominal load.
DI + I O 2 0.2 + 5 2 0.2 + 2.0 A
(46)
4. Calculate the high-side MOSFET power losses
Power losses in the high-side MOSFET (Si9407AGY) at 55-VIN where switching losses dominate can be
calculated from Equation 46 through Equation 49.
Ǹd + 5 Ǹ0.0588 + 1.2 A
I
+I
RMS
(47)
O
substituting Equation 47 into Equation 29 yields
P COND + 1.22 0.12 (1 ) 0.007 (150 * 25)) + 0.324 W
(48)
and from Equation 31, the switching losses can be determined.
P SW(fsw) + ǒVIN
IO
t SWǓ
f SW + 55 V
5A
20 ns
130 kHz + 0.715 W
(49)
The MOSFET junction temperature can be found by substituting Equation 33 into Equation 32
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T J + ǒPCOND ) PSWǓ
q JA ) T A + (0.324 ) 0.715)
40 ) 85 + 127 C
O
(50)
5. Calculate synchronous rectifier losses
The synchronous rectifier MOSFET has two loss components, conduction, and diode reverse recovery losses.
The conduction losses are due to IRMS losses as well as body diode conduction losses during the dead time
associated with the anti-cross conduction delay.
The IRMS current through the synchronous rectifier from Equation 51
Ǹ1 * d + 5 Ǹ1 * 0.0588 + 4.85 A
I
+I
RMS
O
(51)
RMS
The synchronous MOSFET conduction loss from Equation 29 is:
PCOND = 4.852 ´ 0.011´ (1 + 0.007 (150 - 25 )) = 0.485 W
(52)
The body diode conduction loss from Equation 35 is:
PDC = 2 ´ IO ´ VFD ´ tDELAY ´ fSW = 2 ´ 5 A ´ 0.8 V ´ 50 ns ´ 130 kHZ = 0.052 W
(53)
The body diode reverse recovery loss from Equation 36 is:
P RR + 0.5 Q RR V IN f SW + 0.5 30 nC 55 V
130 kHz + 0.107 W
(54)
The total power dissipated in the synchronous rectifier MOSFET from Equation 37 is:
PSR = PRR ´ PCOND ´ PDC = 0.107 + 0.485 + 0.052 = 0.644 W
(55)
The junction temperature of the synchronous rectifier at 85°C is:
TJ = PSR ´ qJA + TA = (0.644 )´ 40 + 85 = 111°C
(56)
In typical applications, paralleling the synchronous rectifier MOSFET with a Schottky rectifier increases the
overall converter efficiency by approximately 2% due to the lower power dissipation during the body diode
conduction and reverse recovery periods.
6. Calculate the Inductor Value
The inductor value is calculated from Equation 12.
(55 - 3.3 )´ 3.3
L=
= 11.9 mH
55 ´ 2 ´ 130 kHZ
(57)
A standard inductor value of 10-µH is chosen. A Coev DXM1306-10RO or Panasonic ETQPF102HFA could be
used.
7. Setting the switching frequency
The clock frequency is set with a resistor (RT) from the RT pin to ground. The value of RT can be derived from
following Equation 58, with fSW in kHz.
RT +
ǒ
f SW
Ǔ
1
* 23 kW + 408 kW,
17.82 E * 06
use 412 kW
(58)
8. Programming the Ramp Generator Circuit
The PWM ramp is programmed through a resistor (RKFF) from the KFF pin to VIN. The ramp generator also
controls the input UVLO voltage. For an undervoltage level of 14.4V (20% below the 18 VIN(min)), RKFF is
calculated in Equation 59.
RKFF = (80%xVIN(min) – 3.5)(65.27 ×RT + 1502) Ω = 309 kΩ, use 301 kΩ
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(59)
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9. Calculating the Output Capacitance (CO)
In this example. the output capacitance is determined by the load response requirement of ΔV = 0.3 V for a 1 A
to 5 A step load. CO can be calculated using Equation 18.
10 mH ǒ5 2 * 1 2Ǔ
CO +
+ 127 mF
ǒ3.3 2 * 3.0 2Ǔ
(60)
Using Equation 13 calculate the ESR required to meet the output ripple requirements.
ǒ
33 mV + 2.0 ESR )
8
1
127 mF
130 kHz
Ǔ
(61)
ESR = 8.9 mΩ
In order to get the required ESR, the capacitance needs to be greater than the 127-µF calculated. For example,
a single Panasonic SP capacitor, 180-µF with ESR of 12 mΩ can be used. Re-calculating the ESR required with
the new value of 180-µF is shown in Equation 62.
ǒ
33 mV + 2.0 ESR )
8
1
180 mF
130 kHz
Ǔ
(62)
ESR = 11.1 mΩ
10. Calculate the Soft-Start Capacitor (CSS)
This design requires a soft-start time (tSTART) of 1 ms. CSS is calculated in Equation 63.
2.3 mA
C SS +
1 ms + 3.28 nF + 3300 pF
0.7 V
(63)
11. Calculate the Current Limit Resistor (RILIM)
The current limit set point depends on tSTART, VO, CO and ILOAD at start up as shown in Equation 7.
180 mF 3.3
I LIM u
) 7.0 + 7.6 A
1m
(64)
Set ILIM for 10.0 A minimum, then from Equation 7
V
(50 mV)
R ILIM + 10 0.14 ) OS W + 10 0.14 )
W + 175 kW + 174 kW
I SINK
I SINK
8.3 mA
8.3 mA
(65)
12. Calculate Loop Compensation Values
Calculate the DC modulator gain (AMOD) from Equation 20.
18
AMOD =
=9
2
AMOD(dB) = 20 ´ log(9) = 19 dB
(66)
(67)
Calculate the output poles and zeros from Equation 21 and Equation 22 of the L-C filter.
1
f LC +
+ 3.7 kHz
2p Ǹ10 mH 180 mF
(68)
and
fZ +
2p
1
0.012
180 mF
+ 74 kHz
(69)
Select the close-loop 0 dB crossover frequency, fC. For this example fC = 10 kHz.
Select the double zero location for the Type III compensation network at the output filter double pole at 3.7 kHz.
Select the double pole location for the Type III compensation network at the output capacitor ESR zero at 73.7
kHz.
24
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SLUS543F – DECEMBER 2002 – REVISED JUNE 2013
The amplifier gain at the crossover frequency of 10 kHz is determined by the reciprocal of the modulator gain
AMOD at the crossover frequency from Equation 27.
2
2
æf ö
æ 3.7 kHz ö
AMOD(f ) = AMOD ´ ç LC ÷ = 9 ´ ç
÷ = 1.23
è 10 kHz ø
è fC ø
(70)
And also from Equation 27.
1
1
G=
=
= 0.81
AMOD(f ) 1.23
(71)
Choose R1 = 100 kΩ
The poles and zeros for a Type III network are described in Equation 25 and Equation 26.
1
1
f Z2 +
N C3 +
+ 430 pF, choose 470 pF
2p R1 C3
2p 100 kW 3.7 kHz
1
1
fP2 =
\ R3 =
= 4.59 kW, choose 4.64 kW
2p ´ R3 ´ C3
2p ´ 470 pF ´ 73.7 kHz
1
1
\ C2 =
= 196 pF, choose 220 pF
2p ´ R1´ C2 ´ G
2p ´ 100 kW ´ 0.81´ 10 kHz
1
1
fP1 =
\ R2 =
= 9.82 kW, choose 10 kW
2p ´ R2 ´ C2
2p ´ 220 pF ´ 73.7 kHz
fC =
1
1
fZ1 =
\ C1 =
= 4301 pF, choose 3900 pF
2p ´ R2 ´ C1
2p ´ 10 kW ´ 3.7 kHz
(72)
(73)
(74)
(75)
(76)
Calculate the value of RBIAS from Equation 23 with R1 = 100 kΩ.
R BIAS + 0.7 V R1 + 0.7 V 100kW + 26.9 kW, choose 26.7 kW
VO * 0.7 V
3.3 V * 0.7 V
(77)
CALCULATING THE BPN10 AND BP10V BYPASS CAPACITANCE
The size of the bypass capacitor depends on the total gate charge of the MOSFET being used and the amount
of droop allowed on the bypass capacitor. The BPN10 capacitance, allowing for a 0.5-V droop on the BPN10 pin
from Equation 8 is shown in Equation 78.
Qg
C BPN10 +
+ 30 nC + 60 nF
0.5
DV
(78)
and the BP10V capacitance from Equation 9 is shown in Equation 79.
Q gSR
C BP10V +
+ 57 nC + 114 nF
0.5
DV
(79)
For this application, a 0.1-µF capacitor was used for the BPN10V and a 1.0-µF was used for the BP10V bypass
capacitor. Figure 14 shows component selection for the 18-V through 55-V to 3.3-V at 5-A dc-to-dc converter
specified in the design example.
GATE DRIVE CONFIGURATION
Due to the possibility of dv/dt induced turn-on from the fast MOSFET switching times, high VDS voltage and low
gate threshold voltage of the Si4470, the design includes a 2-Ω in the gate lead of the upper MOSFET. The
resistor can be used to shape the low-to-high transition of the Switch node and reduce the tendency of dv/dtinduced turn on.
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SLUS543F – DECEMBER 2002 – REVISED JUNE 2013
RKFF
301 kΩ
+
www.ti.com
RILIM
174 kΩ
1
TPS40060PWP
KFF
ILIM
16
2
RT
VIN
15
3
BP5
HDRV
14
RT 412 kΩ
VIN
2Ω
Si9407
0.1 µF
−
4
SYNC
BPN10
13
0.1 µF
RSW
10 Ω
CSS
5
SGND
SW
12
6
SS/SD
BP10
11
7
VFB
LDRV
10
8
COMP
PGND
9
30BQ060
10 µH
1.0 µF
Si4470
3300 pF
R3
4.64 kΩ
R1
100kΩ
+
CO
180 µF
C1 3900 pF
R2
10 kΩ
VOUT
C3
470 pF
−
PGND
RBIAS
26.7 kΩ
C2
220 pF
UDG−02161
Figure 14. Design Example, 48 V to 3.3 V at 5 A dc-to-dc Converter
REFERENCES
1. Balogh, Laszlo, Design and Application Guide for High Speed MOSFET Gate Drive Circuits, Texas
Instruments/Unitrode Corporation, Power Supply Design Seminar, SEM-1400 Topic 2.
2. PowerPAD Thermally Enhanced Package Texas Instruments, Semiconductor Group, Technical Brief: TI
Literature No. SLMA002
26
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SLUS543F – DECEMBER 2002 – REVISED JUNE 2013
REVISION HISTORY
Changes from Revision E (June 2006) to Revision F
Page
•
Changed reference to Figure 13, PowerPad Dimensions, to Figure 14, Design Example, 48 V to 3.3 V at 5 A dc-todc Converter ......................................................................................................................................................................... 7
•
Changed both (CSS – 0.85 V) voltages to (VCSS – 0.85 V) in Programming Soft Start ....................................................... 10
•
Changed turn-on (IL) to start-up (ILOAD) in the third paragraph of Programming Current Limit section. ............................. 11
•
Changed first instance of BPN10 to BP10 in respective section title. ................................................................................ 11
•
Added high-side before MOSFET in the Calculating the BP10 and BP10V Bypass Capacitor section ............................. 12
•
Changed HDRV signal goes high to ...goes low in the Synchronizing to an External Supply section ............................... 13
•
Added equation definition for fSYNC to Equation 10 ............................................................................................................. 13
•
Deleted k from KΩ at the end of equation Equation 11 ...................................................................................................... 13
•
Added (dummy) to RT in Equation 11 definition ................................................................................................................. 13
•
Changed sequence of equation substitutions from: Equation 14 into Equation 13, Equation 16 into Equation 15,
Equation 13 equal to Equation 15, to: Equation 15 into Equation 14, Equation 17 into Equation 16, Equation 14
equal to Equation 16 ........................................................................................................................................................... 14
•
Added generic before modulator gain in first paragraph of the Loop Compensation section ............................................ 14
•
Deleted with VIN being the minimum input voltage required to cause the ramp excursion to cover the entire switching
period. from first paragraph of the Loop Compensation section ........................................................................................ 14
•
Deleted previous Equation 19, which was AMOD = VIN / VS or AMOD(db) = 20 × log (VIN / VS ) ............................................. 14
•
Changed figure reference for modulator gain in the Loop Compensation from Figure 6 (Typical Current Limit
Protection Waveforms) to Figure 8 (PWM MODULATOR RELATIONSHIPS) ................................................................... 14
•
Added moderator DC gain and new Equation 20 to Loop Compensation section ............................................................. 15
•
Changed VOUT to VOin sentence before and in Equation 23 .............................................................................................. 15
•
Changed calculated in to set by in sentence before Equation 24 ...................................................................................... 15
•
Changed VIN / VS to VIN(min) / VRAMP in the Modulator Gain vs Switching Frequency graph ............................................... 15
•
Changed the TCR minimum value from 0.0035 to 3500 and the maximum from 0.010 to 10000 in the second
paragraph of the High-Side MOSFET Power Dissipation section ...................................................................................... 17
•
Changed VDD to VIN in Equation 41 .................................................................................................................................... 19
•
Changed PowerPAD Dimensions to include x and y axis values ....................................................................................... 20
•
Added high-side MOSFET to step four title ........................................................................................................................ 22
•
Changed reference to substituting Equation 30 to Equation 47 ......................................................................................... 22
•
Deleted IRMS2 × RDS(ON) from synchronous MOSFET conduction equation ........................................................................ 23
•
Changed synchronous MOSFET conduction equation equals value from 0.10 to 0.485 ................................................... 23
•
Changed body diode conduction equation values: 100 ns to 50 ns and 0.104 W to 0.052 W ........................................... 23
•
Changed power dissipation equation values: 0.1 to 0.485, 0.104 to 0.052, 0.311 W to 0.644 W ..................................... 23
•
Changed junction temperature equation values: (0.311) to 0.644, 97°C to 111°C ............................................................ 23
•
Changed Step 6 reference to Equation 11 to Equation 12 ................................................................................................. 23
•
Changed inductor value equation in Step 6: replaced value of 48 with 55 and 11.8 with 11.9 .......................................... 23
•
Changed RKFF equation values in Step 8:133.7 to 309 kΩ, 133 to 301 kΩ ........................................................................ 23
•
Added 80%x before VIN(min) in RKFF equation in Step 8 ....................................................................................................... 23
•
Changed first ESR value in Step 9 from 12.7 to 8.9 mΩ .................................................................................................... 24
•
Changed second ESR value in Step 9 from 13.8 to 11.1 mΩ ............................................................................................ 24
•
Changed DC modulator gain values in both equations: 10 to 18, 5 to 9; (5.0) to 9, 14 to 19 dB ...................................... 24
•
Changed AMOD crossover frequency equation values: 5 to 9, 0.68 to 1.23 ..................................................................... 25
•
Changed gain (G) equation values: 0.68 to 1.23, 1.46 to 0.81 .......................................................................................... 25
•
Changed poles and zeros equation values: Equation 73, 73.3 to 73.7 kHZ, 4.62 to 4.59 kΩ; Equation 74, 3.29 to
0.81, 1.46 to 10 kHZ, 109 to 196 pF, 100 to 220 pF; Equation 75, 100 to 200 pF, 73.3 to 73.7 kHz, 21.7 to 9.82 kΩ,
Copyright © 2002–2013, Texas Instruments Incorporated
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21.5 to 10 kΩ; Equation 76, 21.5 to 10 kΩ, 2000 to 4301 pF, 1800 to 3900 pF ................................................................ 25
•
Changed Design Example graphic to include new values from equation: 133 to 301 kΩ, 1800 to 3900 pF, 21.5 to 10
kΩ, 100 to 220 pF. Si9470 to Si9407 ................................................................................................................................. 25
•
Added link references to hard-coded references throughout document ............................................................................. 26
28
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PACKAGE OPTION ADDENDUM
www.ti.com
19-Oct-2022
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
Samples
(4/5)
(6)
TPS40060PWP
ACTIVE
HTSSOP
PWP
16
90
RoHS & Green
NIPDAU
Level-2-260C-1 YEAR
-40 to 85
40060
Samples
TPS40060PWPR
ACTIVE
HTSSOP
PWP
16
2000
RoHS & Green
NIPDAU
Level-2-260C-1 YEAR
-40 to 85
40060
Samples
TPS40060PWPRG4
ACTIVE
HTSSOP
PWP
16
2000
RoHS & Green
NIPDAU
Level-2-260C-1 YEAR
-40 to 85
40060
Samples
TPS40061PWP
ACTIVE
HTSSOP
PWP
16
90
RoHS & Green
NIPDAU
Level-2-260C-1 YEAR
-40 to 85
40061
Samples
TPS40061PWPG4
ACTIVE
HTSSOP
PWP
16
90
RoHS & Green
NIPDAU
Level-2-260C-1 YEAR
-40 to 85
40061
Samples
TPS40061PWPR
ACTIVE
HTSSOP
PWP
16
2000
RoHS & Green
NIPDAU
Level-2-260C-1 YEAR
-40 to 85
40061
Samples
TPS40061PWPRG4
ACTIVE
HTSSOP
PWP
16
2000
RoHS & Green
NIPDAU
Level-2-260C-1 YEAR
-40 to 85
40061
Samples
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of
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