TPS40130
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SLUS602B – JUNE 2004 – REVISED SEPTEMBER 2005
TWO-PHASE, SYNCHRONOUS BUCK CONTROLLER WITH INTEGRATED MOSFET
DRIVERS
FEATURES
APPLICATIONS
•
•
•
•
•
•
•
•
•
Each phase can be operated at a switching
frequency up to 1 MHz, resulting in an effective
ripple frequency of up to 2 MHz at the input and the
output. The two phases operates 180 degrees
out-of-phase.
HDRV2
SW2
32 31 30 29 28 27 26 25
24
OVSET
2
23
BOOT2
3
22
SS
4
21
UVLO
5
20
BP5
6
19
AGND
CSRT1
7
18
CS2
NC
8
17
9 10 11 12 13 14 15 16
GSNS
DIFFO
CS1
NC
CSRT2
RT
VOUT
PGOOD
1
ILIM
BOOT1
FB
PGND
LDRV2
SW2
HDRV2
BOOT2
SS
UVLO
BP5
AGND
CS2
CSRT2
RT
PGOOD
ILIM
EN/SYNC
EN/SYNC
30
29
28
27
26
25
24
23
22
21
20
19
18
17
16
HDRV1
1
2
3
4
5
6
7
8
9
10
11
12
13
14
15
COMP
LDRV1
VIN5
SW1
HDRV1
BOOT1
OVSET
VOUT
GSNS
DIFFO
CS1
CSRT1
COMP
VREF
DROOP
FB
PGND
LDRV2
RHB PACKAGE
(TOP VIEW)
DBT PACKAGE
(TOP VIEW)
VIN5
LDRV1
•
•
•
•
The TPS40130 is a two-phase synchronous buck
controller that is optimized for low-output voltage,
high-output current applications powered from a
supply between 3 V and 40 V. A multi-phase
converter offers several advantages over a single
power stage including lower current ripple on the
input and output capacitors, faster transient response
to load steps, improved power handling capabilities,
and higher system efficiency.
VREF
•
•
DESCRIPTION
DROOP
•
Graphic Cards
Internet Servers
Networking Equipment
Telecommunications Equipment
DC Power Distributed Systems
SW1
•
•
Two-Phase Interleaved Operation
3-V to 40-V Power Stage Operation Range
Supports Up to 6-V VOUT With External Divider
Requires VIN5 @ 50 mA, Typical, Depending
on External MOSFETs and Switching
Frequency
1-µA Shutdown Current
Programmable Switching Frequency up to 1
MHz/Phase
Current Mode Control with Forced Current
Sharing
Better than 1% Internal 0.7-V Reference
Resistive Divider Sets Direct Output Over
Voltage Threshold and Sets Input
Undervoltage Lockout
True Remote Sensing Differential Amplifier
Resistive or Inductor’s DCR Current Sensing
30-pin TSSOP or 32-Pin QFN Packages
Can Be Used with TPS40120 to Provide a 6-Bit
Digitally Controlled Output
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas
Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of the Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
Copyright © 2004–2005, Texas Instruments Incorporated
TPS40130
www.ti.com
SLUS602B – JUNE 2004 – REVISED SEPTEMBER 2005
These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
SIMPLIFIED APPLICATION DIAGRAM
VOUT
VIN
VIN
5V
TPS40130DBT
1 LDRV1
DIFFO
PGND 30
2 VIN5
LDRV2 29
3 SW1
SW2 28
4 HDRV1
HDRV2 27
VIN
5 BOOT1 BOOT2 26
6 OVSET
VOUT
SS 25
7 VOUT
UVLO 24
LOAD
BP5 23
5V
8 GSNS
9 DIFFO
AGND 22
10 CS1
CS2 21
11 CSRT1
CSRT2 20
12 COMP
RT 19
VREF
13 VREF
PGOOD 18
14 DROOP
15 FB
ILIM 17
EN/SYNC 16
5V
UDG−04017
2
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TPS40130
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SLUS602B – JUNE 2004 – REVISED SEPTEMBER 2005
ORDERING INFORMATION
TA
-40°C to 85°C
(1)
(2)
(3)
PACKAGE
PART NUMBER
Plastic TSSOP(DBT) (1)
TPS40130DBT (2) (3)
Plastic QFN (RHB)
TPS40130RHB
The DBTpackage is also available taped and reeled. Add an R suffix to the device type (i.e.,
TPS40130DBTR).
The TPS40130DBTRG4 is a lead (Pb) free product, which means that it is compatible with current
RoHS requirements for all 6 substances, including the requirement that lead not exceed 0.1% by
weight in homogeneous material. In addition, this part has NiPdAu plated copper lead frame and is
rated at MSL level 2 at 260°C according to JEDEC 020C Standards.
Release date for the TPS40130DBTRG4 TBD.
ABSOLUTE MAXIMUM RATING
over operating free-air temperature range unless otherwise noted (1)
TPS40130
SW1, SW2
Input voltage range
BOOT1, BOOT2
-0.3 to VSW + 6.0
All other pins
Sourcing current
UNITS
-1 to 44
V
-0.3 to 6.0
RT
200
µA
TJ
Operating junction temperature range
-40 to 125
°C
Tstg
Storage temperature
-55 to 150
°C
260
°C
Lead temperature 1,6 mm (1/16 inch) from case for 10 seconds
(1)
Stresses beyond those listed under "absolute maximum ratings" may cause permanent damage to the device. These are stress ratings
only, and functional operation of the device at these or any other conditions beyond those indicated under "recommended operating
conditions" is not implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
RECOMMENDED OPERATING CONDITIONS
MIN
NOM
MAX
UNIT
VIN
Input voltage
3.0
40
V
TA
Operating free-air temperature
-40
85
°C
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TPS40130
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SLUS602B – JUNE 2004 – REVISED SEPTEMBER 2005
ELECTRICAL CHARACTERISTICS
TA = -40°C to 85°C, VIN = 12 V, RRT = 64.9 kΩ, TJ = TA (unless otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
4.5
5.0
5.5
1
5
µA
0.5
1.0
1.5
mA
4.3
5.0
5.5
2
3
5
4.00
4.25
4.45
VIN5 INPUT SUPPLY
VIN
Operating voltage range, VIN5
IIN
Shutdown current, VIN5
EN/SYNC = GND
Operating current
Outputs switching, No load
V
BP5 INPUT SUPPLY
Operating voltage range
IBP5
Operating current
VFB < VREF, Outputs switching, no external FETs
Turn-on BP5 rising
Turn-off
hysteresis (1)
150
V
mA
V
mV
OSCILLATOR/SYNCHRONIZATION
Phase frequency accuracy
RT = 64.9 kΩ
Phase frequency set range (1)
Synchronization frequency
360
415
100
range (1)
800
Synchronization input threshold (1)
455
1200
kHz
9600
VBP5/2
V
EN/SYNC
Enable threshold
Pulse width > 50 ns
0.8
Voltage capability (1)
1.0
1.5
VBP5
V
PWM
Maximum duty cycle per channel (1)
87.5%
Minimum duty cycle per channel (1)
0
VREF
Voltage reference
ILOAD = 100 µA
0.687
0.700
0.709
0.691
0.700
0.705
0.0
0.7
2.0
55
150
V
ERROR AMPLIFIER
VFB
Voltage feedback, trimmed (including
differential amplifier)
CMRR
Input common mode range (1)
Input bias current
VFB = 0.7 V
Input offset voltage
Value trimmed to zero
ISRC
Output source current (1)
VCOMP = 1.1 V, VFB = 0.6 V
1
2
ISINK
Output sink current (1)
VCOMP = 1.1 V, VFB =VBP5
1
2
VOH
High-level output voltage
ICOMP = -1 mA
2.5
2.9
VOL
low-level output voltage
ICOMP = 1 mA
GBW
Gain bandwidth (1)
3
5
MHz
AVOL
Open loop gain (1)
60
90
dB
(1)
4
V
Ensured by design. Not production tested.
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0
0.5
nA
V
mA
0.8
V
TPS40130
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SLUS602B – JUNE 2004 – REVISED SEPTEMBER 2005
ELECTRICAL CHARACTERISTICS (continued)
TA = -40°C to 85°C, VIN = 12 V, RRT = 64.9 kΩ, TJ = TA (unless otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
3.5
5.0
6.5
µA
0.95
1.00
1.05
V
-5
4
10
mV
5.1
5.6
6.1
V/V
SOFT START
ISS
Soft-start source current
VSS
Fault enable threshold voltage
32 clocks after EN/SYNC before SS current
begins
CURRENT SENSE AMPLIFIER
Input offset voltage
CS1, CS2
Gain transfer to PWM comparator
-100 mV ≤ VCS≤ 100 mV, VCSRT = 1.5 V
Transconductance to DROOP
VCS - VCSRTn = 100 mV
Gain variance between phases
VCS - VCSRTn = 100 mV
-4%
0
4%
Input offset variance
VCS = 0 V
-3.5
0
3.5
mV
6
µA
Offset current at DROOP
Input common mode
40
VCS - VCSRTn = 0 V
(2)
0
Bandwidth (2)
µA
VBP5-0.7
18
V
MHz
DIFFERENTIAL AMPLIFIER
Gain
CMRR
1
Gain tolerance
VOUT = 4 V vs VOUT = 0.7 V, VGSNS = 0 V
Common mode rejection ratio (2)
0.7 V≤ VOUT ≤ 4.0 V
Output source current
VOUT - VGSNS = 2.0 V, VDIFFO≥ 1.98 V
2
4
Output sink current
VOUT - VGSNS = 2.0 V, VDIFFO≥ 2.02 V
2
4
Input offset voltage (2)
0.7 V≤ VOUT ≤ 4.0 V
Bandwidth (2)
-0.5%
V/V
0.5%
60
dB
mA
5
5
mV
MHz
Input impedance, non-inverting (2)
VOUT to GND
40
Input impedance, inverting (2)
VGSNS to VDIFFO
40
kΩ
GATE DRIVERS
Source on-resistance, HDRV1,
HDRV2
VBOOT1 = 5 V, VBOOT2 = 5 V, VSW1 = 0 V,
VSW2 = 0 V, Sourcing 100 mA
1.0
2.0
3.5
Sink on-resistance, HDRV1, HDRV2
VBOOT1 = 5 V, VBOOT2 = 5 V, VVIN5 = 5 V,
VSW1 = 0 V, VSW2 = 0 V, Sinking 100 mA
0.5
1.0
2.0
Source on-resistance, LDRV1,
LDRV2
VVIN5 = 5 V, VSW1 = 0 V, VSW2 = 0 V,
Sourcing 100 mA
1
2
3.5
Sink on-resistance, LDRV1, LDRV2
VVIN5 = 5 V, VSW1 = 0 V, VSW2 = 0 V,
Sinking 100 mA
0.30
0.75
1.50
tRISE
Rise time, HDRV (2)
CLOAD = 3.3 nF
25
75
tFALL
Fall time, HDRV (2)
CLOAD = 3.3 nF
25
75
tRISE
Rise time,
LDRV (2)
CLOAD = 3.3 nF
25
75
tFALL
Fall time, LDRV (2)
CLOAD = 3.3 nF
25
60
ns
SW falling to LDRV rising
50
mV
tDEAD
Dead time (2)
tON
Minimum controllable on-time (2)
Ω
Ω
LDRV falling to SW rising
30
CLOAD = 3.3 nF
150
OUTPUT UNDERVOLTAGE FAULT
Undervoltage fault threshold
VFB relative to GND
560
588
610
-20%
-16%
-13%
VOVSET relative to GND
796
817
832
VOVSET relative to VVREF
14%
16%
19%
VFB relative to VVREF
OUTPUT OVERVOLTAGE SET
Overvoltage threshold
(2)
mV
Ensured by design. Not production tested.
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TPS40130
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SLUS602B – JUNE 2004 – REVISED SEPTEMBER 2005
ELECTRICAL CHARACTERISTICS (continued)
TA = -40°C to 85°C, VIN = 12 V, RRT = 64.9 kΩ, TJ = TA (unless otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
0.4
0.5
0.6
UNIT
RAMP
Ramp amplitude
(3)
Ramp valley (3)
1.4
V
POWER GOOD
PGOOD high threshold
VFB relative to VREF
10%
14%
PGOOD low threshold
VFB relative to VREF
-14%
-10%
VOL
Low-level output voltage
IPGOOD = 4 mA
ILEAK
PGOOD bias current
VPGOOD = 5.0 V
Current sense fault
(3)
Current from CS1, CS2
0.35
0.60
50
80
5
V
µA
INPUT UVLO PROGRAMMABLE
Input threshold voltage, turn-on
0.9
Input threshold voltage, turn-off
1.0
1.1
0.810
V
LOAD LINE PROGRAMMING
IDROOP
(3)
6
Pull-down current
VCS = 100 mV
Ensured by design. Not production tested.
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30
40
50
µA
TPS40130
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SLUS602B – JUNE 2004 – REVISED SEPTEMBER 2005
HDRV2
SW2
PGND
LDRV2
VIN5
LDRV1
32 31 30 29 28 27 26 25
24
BOOT1
1
NC
SS
4
21
UVLO
5
20
BP5
6
19
AGND
CSRT1
7
18
CS2
NC
8
17
9 10 11 12 13 14 15 16
CS1
CSRT2
RT
DIFFO
PGOOD
3
VOUT
GSNS
ILIM
BOOT2
22
FB
2
23
EN/SYNC
OVSET
VREF
PGND
LDRV2
SW2
HDRV2
BOOT2
SS
UVLO
BP5
AGND
CS2
CSRT2
RT
PGOOD
ILIM
EN/SYNC
DROOP
30
29
28
27
26
25
24
23
22
21
20
19
18
17
16
HDRV1
1
2
3
4
5
6
7
8
9
10
11
12
13
14
15
COMP
LDRV1
VIN5
SW1
HDRV1
BOOT1
OVSET
VOUT
GSNS
DIFFO
CS1
CSRT1
COMP
VREF
DROOP
FB
SW1
RHB PACKAGE
(TOP VIEW)
DBT PACKAGE
(TOP VIEW)
Terminal Functions
TERMINAL
NAME
I/O
NO.
DESCRIPTION
RHB
DBT
AGND
19
22
-
Low noise ground connection to the device.
BOOT1
1
5
I
Provides a bootstrapped supply for the high-side FET driver for PWM1, enabling the gate of the
high-side FET to be driven above the input supply rail. Connect a capacitor from this pin to SW1 pin
and a Schottky diode from this pin to VIN5.
BOOT2
23
26
I
Provides a bootstrapped supply for the high-side FET driver for PWM2, enabling the gate of the
high-side FET to be driven above the input supply rail. Connect a capacitor from this pin to SW2 pin
and a Schottky diode from this pin to VIN5.
BP5
20
23
O
Filtered input from the VIN5 pin. A 10-Ω resistor should be connected between VIN5 and BP5 and a
1.0-µF ceramic capacitor should be connected from this pin to ground.
COMP
9
12
O
Output of the error amplifier. The voltage at this pin determines the duty cycle for the PWM.
CS1
6
10
I
CS2
18
21
I
These pins are used to sense the inductor phase current. Inductor current can be sensed with an
external current sense resistor or by using an external R-C circuit and the inductor's DC resistance.
The traces for these signals must be connected directly at the current sense element. See Layout
Guidelines for more information. After the device is enabled and prior to the device starting (during the
first 32 clock cycles), a 5-µA current flows out of these pins. The current flows through the external
components: current sense resistor, RCS, the output inductor and the output capacitor(s) to ground. If
the voltage on the CS1, and CS2 pins exceed 0.2 V (resistance greater than 40 kΩ), a fault is
declared and the device does not start. This is a fault detection feature that insures the output
inductor, current sense resistor and output capacitors are installed properly on the board.
CSRT1
7
11
O
CSRT2
17
20
O
DIFFO
5
9
O
Output of the differential amplifier. The voltage at this pin represents the true output voltage without IR
drops that result from high-current in the PCB traces. The VOUT and GSNS pins must be connected
directly at the point of load where regulation is required. See Layout Guidelines for more information.
Return point of current sense voltage. The traces for these signals must be connected directly at the
current sense element. See Layout Guidelines for more information.
DROOP
11
14
I
This is the input to the non-inverting input of the Error Amplifier. This pin is normally connected to the
VREF pin and is the voltage that the feedback loop regulates to. This pin is also used to program
droop function. A resistor between this pin and the VREF pin sets the desired droop value. The value
of the DROOP resistor is described in Equation 22.
EN/SYNC
13
16
I
A logic high signal on this input enables the controller operation. A pulsing signal to this pin
synchronizes the rising edge of SW to the falling edge of an external clock source. These pulses must
be greater than 8.2 times the free running frequency of the main oscillator set by the RT resistor.
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Terminal Functions (continued)
TERMINAL
NAME
8
I/O
NO.
DESCRIPTION
RHB
DBT
FB
12
15
I
Inverting input of the error amplifier. In closed loop operation, the voltage at this pin is the internal
reference level of 700 mV. This pin is also used for the PGOOD and undervoltage comparators.
GSNS
4
8
I
Inverting input of the differential amplifier. This pin should be connected to ground at the point of load.
HDRV1
32
4
O
Gate drive output for the high-side N-channel MOSFET switch for PWM1. Output is referenced to
SW1 and is bootstrapped for enhancement of the high-side switch.
HDRV2
25
27
O
Gate drive output for the high-side N-channel MOSFET switch for PWM2. Output is referenced to
SW2 and is bootstrapped for enhancement of the high-side switch
ILIM
14
17
I
Used to set the cycle-by-cycle current limit threshold. If ILIM threshold is reached, the PWM cycle is
terminated and the converter delivers limited current to the output. Under these conditions the
undervoltage threshold eventually is reached and the controller enters the hiccup mode. The
controller stays in the hiccup mode for seven (7) consecutive cycles of SS voltage rising from zero to
1.0 V. At the eighth cycle the controller attempts a full start-up sequence. The relationship between
ILIM and the maximum phase current is described in Equation 4 and Equation 5. See the Overcurrent
Protection section for more details.
LDRV1
29
1
O
Gate drive output for the low-side synchronous rectifier (SR) N-channel MOSFET for PWM1. See
Layout Considerations section.
LDRV2
27
29
O
Gate drive output for the low-side synchronous rectifier (SR) N-channel MOSFET for PWM2. See
Layout Considerations section.
NC
8
NC
24
-
-
No connect. This pin is mechanical only.
OVSET
2
6
I
A resistor divider, on this pin connected to the output voltage sets the overvoltage sense point.
PGOOD
15
18
O
Power good indicator of the output voltage. This open-drain output connects to a voltage via an
external resistor. When the FB pin voltage is between 0.616 V to 0.784 V (88% to 112% of VREF),
the PGOOD output is in a high impedance state. If the DROOP function is implemented, the
programmed droop voltage must be within this window.
PGND
28
30
-
Power ground reference for the controller lower gate drivers. There should be a high-current return
path from the sources of the lower MOSFETs to this pin.
RT
16
19
I
Connecting a resistor from this pin to ground sets the oscillator frequency.
SS
22
25
I
Provides user programmable soft-start by means of a capacitor connected to the pin. If an
undervoltage fault is detected the soft-start capacitor cycles 7 times with no switching before a normal
soft-start sequence allowed.
SW1
31
3
I
Connect to the switched node on converter 1. Power return for the channel 1 upper gate driver. There
should be a high-current return path from the source of the upper MOSFET to this pin. It is also used
by the adaptive gate drive circuits to minimize the dead time between upper and lower MOSFET
conduction.
SW2
26
28
I
Connect to the switched node on converter 2. Power return for the channel 2 upper gate driver. There
should be a high-current return path from the source of the upper MOSFET to this pin. It is also used
by the adaptive gate drive circuits to minimize the dead time between upper and lower MOSFET
conduction.
UVLO
21
24
O
A voltage divider from VIN to this pin, set to 1V, determines the input voltage that starts the controller.
VOUT
3
7
O
Non-inverting input of the differential amplifier. This pin should be connected to VOUT at the point of
load.
VREF
10
13
O
Output of an internal reference voltage. The load may be up to 100 µA DC.
VIN5
30
2
I
Power input for the device. A 1.0-µF ceramic capacitor should be connected from this pin to ground.
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SLUS602B – JUNE 2004 – REVISED SEPTEMBER 2005
FUNCTIONAL BLOCK DIAGRAM
BP5
23
AGND
22
CS1
10
CSRT1
11
VOUT
7
TPS40130DBT
U1
+
20 kΩ
U7
20 kΩ
+
5
BOOT1
4
HDRV1
3
SW1
2
VIN5
1
LDRV1
30
PGND
26
BOOT2
27
HDRV2
28
SW2
29
LDRV2
18
PGOOD
20 kΩ
GSNS
8
U6
20 kΩ
DIFFO
13
Ramp1
U2
0.7 V
U8
+
VREF
9
U11
DROOP
14
FB
15
SS
25
U3
U12
PWM1
+
U9
ICTLR
U10
+
U4
PWM
LOGIC
U13
5 µA
COMP
12
CS2
21
CSRT2
20
ILIM
17
UVLO
24
EN/SYNC
16
RT
19
U15
U18
+
6
+
U17
Ramp2
U14
UV
Power-On Reset FB
OVSET
U5
Anti
Cross
Conduction
VIN5
BP5
U22
Power−On Reset
U23
Clock
U19
OC/UV
Detect
U24
Ramp Gen
U20
SS
PWM2
U16
Anti
Cross
Conduction
VIN5
OV
OC
U21
Ramp1
Ramp2
U25
OV Detect
UDG−04030
FUNCTIONAL DESCRIPTION
The TPS40130 uses programmable fixed-frequency, peak current mode control with forced phase current
balancing. When compared to voltage-mode control, current mode results in a simplified feedback network and
reduced input line sensitivity. Phase current is sensed by using either the DCR (direct current resistance) of the
filter inductors or current sense resistors installed in series with output. The first method involves generation of a
current signal with an R-C circuit (shown in the applications diagram). The R-C values are selected by matching
time constants of the RC circuit and the inductor time constant, R×C = L/DCR. With either current sense
method, the current signal is amplified and superimposed on the amplified voltage error signal to provide current
mode PWM control.
Output voltage droop can be programmed to improve the transient window and reduce size of the output filter.
Other features include: a true differential output sense amplifier, programmable current limit, programmable
output over-voltage set-point, capacitor set soft-start, power good indicator, programmable input undervoltage
lockout (UVLO), user programmable operation frequency for design flexibility, external synchronization
capability, programmable pulse-by-pulse overcurrent protection, output undervoltage shutdown and restart.
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FUNCTIONAL DESCRIPTION (continued)
Startup Sequence
Figure 1 shows a typical start up with the VIN5 and BP5 applied to the controller and then the EN/SYNC being
enabled. Shut down occurs when the VIN5 is removed
VIN5
BP5
EN/SYNC
1.0V
0.7V
SS
SSWAIT
VOUT
PGOOD
POR
UDG−04031
Figure 1. Startup and Shutdown Sequence
Differential Amplifier (U7)
The unity gain differential amplifier with high bandwidth allows improved regulation at a user-defined point and
eases layout constraints. The output voltage is sensed between the VOUT and GSNS pins. The output voltage
programming divider is connected to the output of the amplifier (DIFFO). The differential amplifier input voltage
must be lower than (VBP5 - 0.7 V).
If there is no need for a differential amplifer, the differential amplifier can be disabled by connecting the GSNS
pin to the BP5 pin and leaving VOUT and DIFFO open. The voltage programming divider in this case should be
connected directly to the output of the converter.
TPS40130DBT
VOUT
20 kΩ
7
20 kΩ
GSNS
20 kΩ
Differential
Amplifier
+
DIFFO
9
20 kΩ
8
UDG−04081
Figure 2. Differential Amplifier Configuration
Because of the resistor configuration of the differential amplifier, the input impedance must be kept very low or
there will be error in setting the output voltage.
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FUNCTIONAL DESCRIPTION (continued)
Current Sensing and Balancing (U1, U9 and U18)
The controller employs peak current mode control scheme, thus naturally provides certain degree of current
balancing. With current mode, the level of current feedback should comply with certain guidelines depending on
duty factor known as “slope compensation” to avoid the sub-harmonic instability. This requirement can prohibit
achieving a higher degree of phase current balance. To avoid the controversy, a separate current loop that
forces phase currents to match is added to the proprietary control scheme. This effectively provides high degree
of current sharing independent of the controller’s small signal response and is implemented in U9, ICTLR.
High bandwidth current amplifiers, U1 and U18 can accept as an input voltage either the voltage drop across
dedicated precise current sense resistors, or inductor’s DCR voltage derived by an RC network, or thermally
compensated voltage derived from the inductor’s DCR. The wide range of current sense arrangements ease the
cost/complexity constrains and provides superior performance compared to controllers utilizing the low-side
MOSFET current sensing. The current sense amplifier inputs must not exceed 4 V. See the Inductor DCR
Current Sense section for more information on selecting component values for the R-C network.
PowerGood
The PGOOD pin indicates when the inputs and output are within their specified ranges of operation. Also
monitored are the EN/SYNC and SS pins. PGOOD has high impedance when indicating inputs and outputs are
within specified limits and is pulled low to indicate an out-of-limits condition. Some applications may require
hysteresis on the PGOOD signal to avoid a PGOOD signal bounce. A simple method to achieve this (and
thereby eliminate any PGOOD signal bounce) is to add a small resistor (RA) and capacitor (CA) between the FB
pin and the PGOOD pin. See Figure 3 for implementation.
VOUT
VPU
Optional components for
hysteresis
10 kΩ
10 kΩ
RA
CA
TPS40130DBT
PGOOD
18
0.616 V
+
FB
15
RBIAS
+
0.784 V
PGOOD window
comparator
UDG−05078
Figure 3. Adding Hysteresis to the PGOOD Signal
To select RA and CA, the following criteria can be used.
R A + RBIAS
ƪǒ
Ǔ ƫ
VPU * VFB
*1
1.3 0.16 VFB
(1)
VPU and the PGOOD pull-up voltage and VFB is the TPS40130 internal reference voltage. The factor 1.3 in
Equation 1 provides a margin for robustness.
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FUNCTIONAL DESCRIPTION (continued)
C A + 100 ns
RA
(2)
Even though CA may calculate to less than 1 pF, a capacitance of no more than 4.7 pF is recommended.
Soft-Start
A capacitor connected to the soft start pin (SS) sets the power-up time. When EN is high and POR is cleared,
the calibrated current source, U13, starts charging the external soft start capacitor. The PGOOD pin is held low
during the start up. The rising voltage across the capacitor serves as a reference for the error amplifier, U12.
When the soft-start voltage reaches the level of the reference voltage, U8 (VVREF=0.7V), the converter’s output
reaches the regulation point and further voltage rise of the soft start voltage has no effect on the output. When
the soft start voltage reaches 1.0 V, the power good (PGOOD) function is cleared to be reported on the PGOOD
pin. Normally the PGOOD pin goes high at this time. Equation 3 is used to calculate the value of the soft-start
capacitor.
0.7 C SS
t SS +
5 10 *6
(3)
Overcurrent Protection
The overcurrent function, U19, monitors the output of current sense amplifiers U1 and U18. These currents are
converted to voltages and compared to the voltage on the ILIM pin. The relationship between the maximum
phase current and the current sense resistance is given in the following equation. In case a threshold of VILIM/2.7
is exceeded the PWM cycle on the associated phase is terminated. The overcurrent threshold, IPH(max), and the
voltage to set on the ILIM pin is determined by Equation 4 and Equation 5.
V ILIM + 2.7 I PH(max) R CS
(4)
I PH(max) +
ǒV IN * VOUTǓ VOUT
I OUT
)
2
2 L OUT f SW V IN
(5)
where
•
•
•
IPH(max) is a maximum value of the phase current allowed
IOUT is the total maximum DC output current
RCS is a value of the current sense resistor used or the DCR value of the output inductor, LOUT
If the overcurrent condition persists, both phases have PWM cycles terminated by the overcurrent signals. This
puts a converter in a constant current mode with the output current programmed by the ILIM voltage. Eventually
the supply-and-demand equilibrium on the converter output is not satisfied and the output voltage starts to
decline. When the undervoltage threshold is reached, the converter enters a hiccup mode. The controller is
stopped and the output is not regulated any more, the soft-start pin function changes.
It now serves as a hiccup timing capacitor controlled by U20, the fault control circuit. The soft-start pin is
periodically charged and discharged by U20. After seven hiccup cycles, the controller attempts another soft-start
cycle to restore normal operation. If the overload condition persists, the controller returns to the hiccup mode.
This condition may continue indefinitely. In such conditions the average current delivered to the load is
approximately 1/8 of the set overcurrent value.
Current Sense Fault Protection
Multiphase controllers with forced current sharing are inherently sensitive to a failure of the current sense
component or a defect in the assembly process. In case of such failure the entire load current can be steered
with catastrophic consequences into a single channel where the fault has occurred. A dedicated circuit in the
TPS40130 controller detects this defect and prevents the controller from starting up. This fault detection circuit is
active only during chip initialization and does not protect should current sense failure happen during normal
operation.
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FUNCTIONAL DESCRIPTION (continued)
After the device is enabled and prior to the IC starting (during the first 32 clock cycles), a 5-µA current flows out
of the CS1 and CS2 pins. The current flows through the external components: current sense resistor, RCS, the
output inductor and the output capacitor(s) to ground. If the voltage on the CS1 and CS2 pins exceed 0.2 V
(resistance greater than 40 kΩ), a fault is declared and the device does not start. This is a fault detection feature
that insures the output inductor, current sense resistor and output capacitors are installed properly on the board.
Overvoltage Protection
The voltage on OVSET is compared with 0.817 V, 16% higher than VREF, in U25 to determine the output
overvoltage point. When an overvoltage is detected, the output drivers command the upper MOSFETs off and
the lower MOSFETs on. If the overvoltage is caused by a shorted upper MOSFET, latching on the lower
MOSFET should blow the input fuse and protect the output. Hiccup mode consisting of seven (7) soft-start
timing cycles is initiated and then attempts to restart. If the overvoltage condition has been cleared and the input
fuse has not opened, the output comes up and normal operation continues. If the overvoltage condition persists,
the controller restarts to allow the output to rise to the overvoltage level and return to the hiccup mode. Using a
voltage divider with the same ratio, that sets the output voltage, an output overvoltage is declared when the
output rises 16% above nominal.
Output Undervoltage Protection
If the output voltage, as sensed by U19 on the FB pin becomes less than 0.588 V, the undervoltage protection
threshold (84% of VREF), the controller enters the hiccup mode as it is described in the Overcurrent Protection
section.
Programmable Input Undervoltage Lockout Protection
A voltage divider that sets 1V on the UVLO pin determines when the controller starts operating. Operation
commences when the voltage on the UVLO pin exceeds 1.0 V.
Power-On Reset (POR)
The power-on reset (POR) function, U22, insures the VIN5 and BP5 voltages are within their regulation windows
before the controller is allowed to start.
Fault Masking Operation
If the SS pin voltage is externally limited below the 1-V threshold, the controller does not respond to most faults
and the PGOOD output is always low. Only the overcurrent function and current sense fault remain active. The
overcurrent protection still continues to terminate PWM cycle every time when the threshold is exceeded but the
hiccup mode is not entered.
Fault Conditions and MOSFET Control
Table 1 shows a summary of the fault conditions and the state of the MOSFETs.
Table 1. Fault Condifions
FAULT MODE
UPPER MOSFET
LOWER MOSFET
EN/SYNC = LOW
OFF
OFF
FIXED UVLO, VBP5 < 4.25 V
OFF
OFF
Programmable UVLO, < 1.0 V
OFF
ON
Output undervoltage
OFF, Hiccup mode
ON, Hiccup mode
Output overvoltage
OFF, Hiccup mode
ON, Hiccup mode
ISF, current sense fault
OFF
ON
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Setting the Switching Frequency
The clock frequency is programmed by the value of the timing resistor connected from the RT pin to ground.
See Equation 6.
R T + 0.8
ƪǒ
36
Ǔ ƫ
103 * 9
f PH
(6)
fPH is a single phase frequency, kHz. The RT resistor value is expressed in kΩ. See Figure 4.
500
RT − Timing Resistance − kΩ
450
400
350
300
250
200
150
100
50
0
0
200
400
600
800
1000
fSW − Phase Switching Frequency − kHz
Figure 4. Phase Switching Frequency vs. Timing Resistance
EN/SYNC Function
The output ripple frequency is twice that of the single phase frequency. The switching frequency of the controller
can be synchronized to an external clock applied to the EN/SYNC pin. The external clock synchronizes the
rising edge of HDRV and the falling edge of an external clock source. The external clock pulses must be at a
frequency at least 8.2 times higher than the switching frequency set by the RT resistor.
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Setting Overcurrent Protection
Setting the overcurrent protection is given in the following equations. Care must be taken when calculating VILIM
to include the increase in RCS caused by the output current as it approaches the overcurrent trip point. The DCR
(RCS in the equation) of the inductor increases approximately 0.39% per degree Centigrade.
V ILIM + 2.7 I PH(max) R CS
(7)
I PH(max) +
ǒV IN * VOUTǓ VOUT
I OUT
)
2
2 L OUT f SW V IN
(8)
where
•
•
•
•
•
•
•
IPH(max) is a maximum value of the phase current allowed
IOUT is the total maximum DC output current
LOUT is the output inductor value
fSW is the switching frequency
VOUT is the output voltage
VIN is the input voltage
RCS is a value of the current sense resistor used or the DCR value of the output inductor, LOUT
Resistor Divider Calculation for VOUT, ILIM, OVSET and UVLO
Use Figure 9 for setting the output voltage, current limit voltage and overvoltage setting voltage. Select RBIAS
using Equation 9. With a voltage divider from VREF, select R6 using Equation 10. WIth a voltage from DIFFO
select R4 using Equation 11. With a voltage divider from VIN, select R8 using Equation 12.
R1
R BIAS + 0.7
ǒV OUT * 0.7Ǔ
(9)
R6 + R5
R4 + 0.812
R8 + 1.0
VILIM
ǒ0.7 * V ILIMǓ
(10)
R3
ǒVOUT(ov) * 0.812Ǔ
(11)
R7
ǒVIN * 1.0Ǔ
(12)
Feedback Loop Compensation
The TPS40130 operates in a peak-current mode and the converter exhibits a single pole response with ESR
zero for which Type II compensation network is usually adequate as shown in Figure 5.
The load pole is situated at a value calculated using Equation 13.
1
f OP +
2p R OUT C OUT
(13)
and the ESR zero is situated at a value calculated using Equation 14.
1
f ESRZ +
2p R ESR C OUT
(14)
To achieve the desired bandwidth the error amplifier has to compensate for modulator gain loss at the crossover
frequency. A zero placed at the load pole frequency facilitates that. The ESR zero alters the modulator -1 slope
at higher frequencies. To compensate for the ESR zero, a pole in the error amplifier transfer function should be
placed at the ESR zero frequency.
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VOUT
TPS40130DBT
DIFFO
7
+
9
GSNS
8
COMP
12
C1
R1
C2
R2
FB
15
Modulator
DROOP
LOUT
+
14
VOUT
COUT
RBIAS
ROUT
13
VREF
+
VREF
RESR
UDG−04029
Figure 5. Compensation Components
The following expressions help in choosing components of the EA compensation network. It is recommended to
fix value of the resistor R1 first as it further simplifies adjustments of the output voltage without altering the
compensation network.
R1
R2 +
AMOD(f)
(15)
AMOD +
V VIN
0.4
(16)
where AMOD is the modulator gain at DC
f OP
AMOD(f) + AMOD
fC
(17)
where AMOD(f) is the modulator gain at the crossover frequency
1
C1 +
ǒ2p fOP R2Ǔ
(18)
C2 +
1
ǒ2p
f ESRZ
R2Ǔ
(19)
Introduction of output voltage droop as a measure to reduce amount of filter capacitors changes the transfer
function of the modulator as it is shown in Figure 6 and Figure 7. The droop function introduces another zero in
the modulator gain function.
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GAIN AND PHASE
vs
FREQUENCY WITHOUT DROOP
GAIN AND PHASE
vs
FREQUENCY WITH DROOP
80
80
Converter Overall
Converter Overall
60
EA
60
EA
20
Type II
G − Gain − dB
G − Gain − dB
40
Modulator
40
Droop Zero
20
Modulator
0
0
Load Pole
Load Pole
ESR Zero
−20
−20
ESR Zero
−40
−40
200
200
EA
150
150
Phase
EA
100
Phase − °
Phase − °
100
Converter Overall
50
Converter Overall
50
0
0
Modulator
Modulator
−50
−50
−100
10
100
1k
10 k
f − Frequency − Hz
100 k
1M
−100
10
Figure 6.
100
1k
10 k
f − Frequency − Hz
100 k
1M
Figure 7.
The droop function, as well as the the output capacitor ESR, introduce a zero on some frequency left from the
crossover point. See Equation 20
1
f DROOPZ +
VDROOP
2p
COUT
I OUT(max)
ǒ
Ǔ
(20)
To compensate for this zero, pole on the same frequency should be added to the error amplifier transfer
function. With Type II compensation network a new value for the capacitor C2 is required compared to the case
without droop.
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C2 +
C1
2p
R2
C1
ǒf DROOPZ * 1 Ǔ
(21)
When attempting closing the feedback loop at frequency that is close to the theoretical limit, use the above
considerations as a first approximation and perform on bench measurements of closed loop parameters as
effects of switching frequency proximity and finite bandwidth of voltage and current amplifiers may substantially
alter them as it is shown in Figure 8.
GAIN AND PHASE
vs
FREQUENCY
60
Phase
80
50
60
30
40
20
10
Phase − °
G − Gain − dB
40
20
Gain
0
0
−10
−20
100
VIN = 12 V
VOUT = 1.5 V
1k
10 k
100 k
f − Frequency − Hz
−20
1M
Figure 8.
Setting the Output Voltage Droop
In many applications the output voltage of the converter intentionally allowed to droop as load current increases.
This approach also called active load line programming and allows for better use of regulation window and
reduces the amount of the output capacitors required to handle a load current step. A resistor from the VREF pin
to the DROOP pin sets the desired value of the output voltage droop. See Equation 22.
5000 V DROOP
R BIAS
R DROOP +
I OUT R CS
R1 ) RBIAS
(22)
where
•
•
18
VDROOP is the value of droop at maximum load current (ILOAD)
RCS is a value of the current sense resistor used or the DCR value of the output inductor
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TPS40130DBT
OVSET
6
GSNS
Differential
Amplifier
8
VOUT
7
+
DIFFO
9
R2
R3
C1
COMP
12
R1
Error
Amplifier
FB
15
+
DROOP
R4
RBIAS
VIN
14
VREF
RDROOP
IDROOP
13
R5
ILIM
17
R6
R7
UVLO
24
+
R8
700 mV
UDG−04032
Figure 9. Implementing the Droop Function, Resistor Between DROOP and VREF.
VOUT − Output Voltage − V
VOUT
VDROOP
IOUT(max)
0
IOUT − Output Current − A
UDG−03116
Figure 10. Output Voltage Droop Characteristic as Output Current Varies.
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Inductor DCR Current Sense
Inductor DCR current sensing is a known lossless technique to retrieve current proportional signal. Referring to
Figure 11.
At any given frequency the DCR voltage can be calculated using Equation 23 and Equation 24.
DCR
V DCR + ǒVIN * VOUTǓ
DCR ) w L
1
V C + ǒVIN * VOUTǓ
w C
RCS ) 1
w C
ǒ
(23)
Ǔ
(24)
Voltage across the capacitor is equal to voltage drop across the inductor DCR, VC=VDCR when time constant of
the inductor and the time constant of the RC network are equal, see Equation 25. Setting the value of the
capacitor to 0.1 µF or 0.01 µF provides for reasonable resistor values.
DCR
1
L +R
VC +
+
;
C; tDCRL + t RC
CS
DCR ) w L DCR
1
w C
RCS )
w C
(25)
ǒ
Ǔ
The output signal generated by the network shown in Figure 11 is temperature dependent due to positive
thermal coefficient of copper specific resistance KT=1+0.0039 ×(T-25). The temperature variation of the inductor
coil can easily exceed 100°C in a practical application leading to approximately 40% variation in the output
signal and, in turn, respectively moving the overcurrent threshold and the load line.
VDCR
VDCR
L
L
DCR
DCR
Switch
Node
VOUT
Switch
Node
VOUT
R
0.1%
RCS
C
RCS
R
0.1%
C
VC
CS
R
0.1%
CSRT
UDG−03142
VC
CS
CSRT
R
0.1%
UDG−05079
Figure 11. Inductor Current Sense Configuration
for 1.2-V Output
Figure 12. Inductor Current Sense Configuration
for 5-V Output
Inductor DCR Current Sensing with 5-V Outputs
Due to the current sense operational amplifier input common mode voltage range, it is necessary to divide the
current sense information before connecting to the TPS40130 current sense pins (CS1, CSRT1, CS2 and
CSRT2). Figure 12 shows how this is achieved. RCS and C are selected as normal using the method described
in Equation 24 and Equation 25. The divider resistors (R) are chosen to be much smaller than RCS. They are
typically 100 times smaller. However, if these resitors are too small, they may dissipate too much power. In that
case, choose a smaller capacitance value, and select a new RCS. The resistors used in the divider should be
precision resistors each having the same value.
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APPLICATION INFORMATION
Applications Circuit
Figure 13 shows a typical applications circuit providing 1.5 VOUT at 40 A.
VOUT
R2
C5
C4
R1
C19
L1
L2
12 V
12 V
5V
C1
0.1 µF
Q1
D1
BAT54A
C13
C21
C12
C20
C2
0.1 µF
D2
Q3,
Q4
C17
C15
Q2
TPS40130DBT
1.0 Ω
1
LDRV1
PGND 30
2
VIN5
LDRV2 29
3
SW1
SW2 28
1.0 Ω
Q5,
Q6
PGND
PGND
DIFFO
5V
4
HDRV1
HDRV2 27
5
BOOT1
BOOT2 26
12 V
R13
10 kΩ
C7 2200 pF
6
OVSET
SS 25
7
VOUT
UVLO 24
8
GSNS
BP5 23
R22
10 Ω
R21 51 Ω
R14
10 kΩ
VOUT
R20 51 Ω
9
VOUT
DIFFO
R6
R7
C6
0.1 µF
AGND 22
10 CS1
CS2 21
R10
10 kΩ
LOAD
11 CSRT1
C8
CSRT2 20
R11
VREF
R5 90.9 kΩ
12 COMP
RT 19
C10
R8 10 kΩ
R17
PGOOD 18
13 VREF
R12 10 kΩ
C9
0.1 µF
ILIM 17
14 DROOP
R16 10 kΩ
EN/SYNC 16
FB
5V
R18
15 FB
EN/SYNC
R19
UDG−04018
Figure 13. Typical Application Circuit
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APPLICATION INFORMATION (continued)
Additional Application Circuits
Figure 14 shows a VRM10.x compliant solution where the output voltage is controlled by the VID code of the
TPS40120. The six-bit controller provides outputs from 0.8375 V to 1.600 V in 12.5 mV steps for VRM 10.x or
provides five-bit control for other Intel processors. When the TPS40120 receives a VID of x11111, indicating the
no CPU state, output NCPU1# pulls the soft-start (SS) pin low insuring the output voltage soft-starts with a valid
VID code.
VOUT
R2
C5
C4
R1
C19
L1
L2
12 V
12 V
5V
Q1
C1
0.1 µF
D1
BAT54A
C13
C21
C12
C20
C2
0.1 µF
D2
Q3,
Q4
Q2
1 LDRV1
PGND 30
1.0 Ω
PGND
DIFFO
2 VIN5
LDRV2 29
3 SW1
SW2 28
4 HDRV1
6 OVSET
1 VID5
VCC 14
2 VID0
VOUT 8
VOUT
R20 51 Ω
N/C 11
6 VID4
FB 9
7 GND
BIAS 10
C7 2200 pF
R22
10 Ω
12 V
SS 25
7 VOUT
UVLO 24
R6
8 GSNS
BP5 23
R7
BP5
9 DIFFO
10 CS1
SS
11 CSRT1
C8
5 VID3
PGND
R21 51 Ω
3 VID1 NCPU2 13
4 VID2 NCPU1 12
Q5,
Q6
5V
HDRV2 27
5 BOOT1 BOOT2 26
R13
10 kΩ
TPS40120
C15
TPS40130DBT
1.0 Ω
R14
10 kΩ
C17
R11
12 COMP
C10
AGND 22
C6
0.1 µF
CS2 21
CSRT2 20
VREF
R5 90.9 kΩ
RT 19
PGOOD 18
13 VREF
R12 10 Ω
C9
0.1 µF
R17
R8 10 kΩ
ILIM 17
14 DROOP
EN/SYNC 16
R16 10 kΩ
15 FB
FB
5V
R18
EN/SYNC
UDG−04088
Figure 14. Application Circuit with VID Control
Figure 15 shows the configuration with the TPS40130 processing power from two different input power sources,
12 V and 5 V is shown. This is useful when there is not sufficient power from a single input source to provide the
required output power. The inductor currents are not equal and the difference in the peak currents are
approximately:
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APPLICATION INFORMATION (continued)
(D1 * D2)
DI PEAK ^ 0.067
DCR h
(26)
where
• D1 is the duty cycle for VIN1
• D2 is the duty cycle for VIN2
• DCR is the resistance of the output inductor
• η is the efficiency of the converter
VOUT
R2
C5
C4
R1
C19
L1
L2
12 V
5V
5V
Q1
C1
0.1 µF
D1
BAT54A
C13
C21
C12
C20
C2
0.1 µF
D2
Q3,
Q4
Q2
C15
TPS40130DBT
1.0 Ω
1 LDRV1
PGND 30
1.0 Ω
PGND
DIFFO
2 VIN5
LDRV2 29
3 SW1
SW2 28
Q5,
Q6
PGND
5V
R13
10 kΩ
R14
10 kΩ
4 HDRV1
HDRV2 27
5 BOOT1
BOOT2 26
6 OVSET
SS 25
VOUT
C7 2200 pF
R22
10 Ω
7 VOUT
UVLO 24
R6
8 GSNS
BP5 23
R20 51 Ω
VOUT
AGND 22
10 CS1
R10
10 kΩ
12 V
R21 51 Ω
9 DIFFO
LOAD
C17
C6
0.1 µF
CS2 21
11 CSRT1
C8
R7
CSRT2 20
R11
VREF
R5 90.9 kΩ
12 COMP
RT 19
C10
R8 10 kΩ
PGOOD 18
13 VREF
R12 10 Ω
C9
0.1 µF
14 DROOP
FB
15 FB
R17
ILIM 17
R16 10 kΩ
EN/SYNC 16
5V
R18
EN/SYNC
R19
UDG−04089
Figure 15. Application Circuit with Input Voltage Power Sharing from Two Separate Voltage Sources
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APPLICATION INFORMATION (continued)
Figure 16 shows the required 5-V input being generated with an external linear regulator. The regulator shown is
the TL431 shunt regulator which is a very cost effective solution. Depending on the required current to the
MOSFET gates, the 115 Ω resistor may need to be a ¼ W or ½ W resistor.
L1
VOUT
12 V
Q1
Q3
C21
Q4
C15
C12 C13
115 Ω
L2
C17
Q5
Q2
Q6
C19 C20
10 kΩ
1
BOOT1
VOUT
R13
10 kΩ
C2
0.1 µF
R1
NC 24
2 OVSET
R2
D2
BAT54A
HDRV2 25
SW2 26
LDRV2 27
PGND 28
LDRV1 29
VIN5 30
D1
BAT54A
R14
10 kΩ
1Ω
1Ω
C1
0.1 µF
SW1 31
TL431
HDRV1 32
10 kΩ
R21 51 Ω
12 V
5V
BOOT2 23
C7 2200 pF
3 VOUT
SS 22
R22
10 Ω
R22 51 Ω
4 GSNS
R6
UVLO 21
TPS40130RHB
5 DIFFO
R7
BP5 20
6 CS1
C6
0.1 µF
AGND 19
C5
7 CSRT1
R10
10 kΩ
VREF
R12
C19
0.1 µF
R11
C10
R17
R5
90.9 kΩ
R16
10 kΩ
C8
C4
CSRT2 17
16 RT
15 PGOOD
14 ILIM
13 EN/SYNC
12 FB
10 VREF
9
COMP
8 NC
11 DROOP
CS2 18
R18
R8
10 kΩ
5V
R19
EN/SYNC
UDG−05080
Figure 16. Application Circuit with an External Linear Regulator Providing VIN5
24
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SLUS602B – JUNE 2004 – REVISED SEPTEMBER 2005
APPLICATION INFORMATION (continued)
Figure 17 shows the configuration for efficiently operating at high frequencies. With the power stages input at 5
V, the switching losses in the upper MOSFET are significantly reduced. The upper MOSFET should be selected
for lower RDS(on) because the conduction losses are somewhat higher at the higher duty cycle.
VOUT
R2
C5
C4
R1
C19
L1
L2
5V
5V
5V
Q1
C1
0.1 µF
D1
BAT54A
C13
C21
C12
C20
C2
0.1 µF
C17
D2
Q3,
Q4
C15
Q2
TPS40130DBT
1.0 Ω
1
LDRV1
PGND 30
2
VIN5
LDRV2 29
3
SW1
SW2 28
1.0 Ω
PGND
Q5,
Q6
PGND
DIFFO
5V
4
HDRV1
HDRV2 27
5
BOOT1
BOOT2 26
5V
R13
10 kΩ
C7 2200 pF
6
OVSET
R22
10 Ω
SS 25
R6
R21 51 Ω
R14
10 kΩ
VOUT
7
VOUT
UVLO 24
8
GSNS
BP5 23
9
DIFFO
R20 51 Ω
AGND 22
VOUT
R7
C6
0.1 µF
10 CS1
CS2 21
R10
10 kΩ
LOAD
11 CSRT1
C8
CSRT2 20
R11
VREF
R5 33.2 kΩ
12 COMP
RT 19
C10
R8 10 kΩ
PGOOD 18
R17
13 VREF
R12 10 Ω
C9
0.1 µF
ILIM 17
14 DROOP
R16 10 kΩ
EN/SYNC
FB
16
5V
R18
15 FB
EN/SYNC
R19
UDG−04091
Figure 17. Application Circuit For High-Frequency Operation With Input Voltage of 5 V
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APPLICATION INFORMATION (continued)
Figure 18 shows how to configure for a 5-V output. The resistor dividers on the CSx and CSRTx inputs are
necessary to reduce the common mode voltage into the current sense amplifiers. The differential amplifier is not
used because with a 5-V output, remote sensing is not generally necessary. If the differential Amplifier is
necessary, a voltage divider of 2/3 should be used and the magnitude of the resistors should be about 500 Ω
and 1 kΩ.
VOUT
L1
12 V
L2
12 V
Q1
C13
5V
C15
C1
0.1 µF
C21
C19
C12
C20
C2
0.1 µF
D1 BAT54A
Q2
Q3, Q4
R32
0.1 %
D2
BAT54A
1.0 Ω
1
LDRV1
PGND 30
Q5, Q6
1.0 Ω
2
VIN5
LDRV2 29
3
SW1
SW2 28
4
HDRV1
HDRV2 27
5
BOOT1
BOOT2 26
6
OVSET
SS 25
R14
10 kΩ
7
VOUT
UVLO 24
BP5
8
GSNS
BP5 23
9
DIFFO
VOUT
R13
10 kΩ
R33
0.1 %
R2
R40
0.1 %
5V
C7 2200 pF
PGND
R22
10 kΩ
R34
0.1%
C5
R42
0.1 %
C6
0.1 µF
AGND 22
10 CS1
R1
CS2 21
VOUT
11 CSRT1
R10
10 kΩ
C8
C17
TPS40130DBT
C4
R35
0.1 %
R43
0.1%
CSRT2 20
R11
R5 90.9 kΩ
12 COMP
R44
0.1%
RT 19
C10
LOAD
R8 10 kΩ
PGOOD 18
13 VREF
C9
0.1 µF
R17
R12 10 Ω
14 DROOP
FB
15 FB
ILIM 17
EN/SYNC 16
R19
1.62 kΩ
VREF
R16 10 kΩ
5V
R18
EN/SYNC
UDG−04092
Figure 18. Application Circuit for Providing 5-V Output
26
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TYPICAL CHARACTERISTRICS
VDIFFOUT (50 mV/div)
VDIFFOUT
(50 mV/div)
IL1 (5 A / div)
IL1, IL2
(5 A/div)
IL2 (5 A / div)
VSW2 (10 V / div)
VSW2 (10 V / div)
t − Time − 20 µs/div
t − Time − 4 µs/div
Figure 19. Load Transient
Figure 20. Load Transient Rising Edge
VSW1 (10 V/div)
VDIFFOUT (50 mV/div)
IL1 (5 A / div)
VEN/SYNC
(5 V / div)
IL2 (5 A / div)
VSS
(100 mV / div)
VSW2
(10 V / div)
VDIFFOUT
(500 mV/div)
t − Time − 40 µs/div
t − Time − 4 µs/div
Figure 21. Load Transient Falling Edge
Figure 22. Start-Up with EN/SYNC and Showing
Soft-Wait Time
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TYPICAL CHARACTERISTRICS (continued)
VDIFFOUT
(500 mV/div)
VEN/SYNC
(5 V/div)
VSS
(500 mV / div)
VEN/SYNC
(5 V / div)
VHDRV1
(10 V / div)
VHDRV2
(10 V / div)
t − Time − 400 µs/div
t − Time − 400 ns/div
Figure 23. Start-Up with EN/SYNC
VEN/SYNC
(5 V/div)
Figure 24. External Clock on EN/SYNC
VSS
(500 mV / div)
VLDRV1
(5 V / div)
VHDRV1
(10 V / div)
VHDRV2
(10 V / div)
VOVSET
(1 V / div)
t − Time − 40 ns/div
t − Time − 4 ms/div
Figure 25. External Clock on EN/SYN and Delay to HDRV
28
Figure 26. Overvoltage, Latch and Re-Start
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SLUS602B – JUNE 2004 – REVISED SEPTEMBER 2005
TYPICAL CHARACTERISTRICS (continued)
VFB
(200 mV / div)
VFB
(200 mV/ div)
VSS (1 V/ div)
VSS (1 V/div)
IL2, IL2
(10 A / div)
IL2, IL2
(10 A / div)
VFB = 0.588 V
t − Time − 40 µs/div
t − Time − 4 ms/div
Figure 27. Overcurrent, Hiccup Mode
Figure 28. Overcurrent
LAYOUT CONSIDERATIONS
Introduction
Any pin number references are for the DBT, TSSOP package. There are two general classes of signals to
consider for proper layout, high-current switching and low-level analog. Refer to Figure 13 for references to
components. A printed wiring board (PWB) with a minimum of four layers should be used.
Two Ground Planes
A basic requirement is two separate ground planes that ultimately get connected together at a point where no
switching currents are present, the power ground (PGND) and the analog ground (AGND). They should be
implemented as split planes on the top and bottom layers. The PGND is used for all high-current signals
including LDRV1, LDRV2, lower MOSFETs and input and output decoupling capacitors. PGND should be used
on the top layer around the high current components and on the bottom layer as a minimum. The AGND is used
for low level signals such as: soft-start, RT, VREF, FB, BP5 decoupling to AGND. AGND should be used on the
top layer around the device and low level components and on the bottom layer as a minimum. The signals which
connect to the two different ground planes are shown in Figure 13 using different symbols for each ground.
Low-Level Signal Connections and Routing
Current Sense Signals
Using inductor current sense has advantages over using a low-value, high-power current-sense resistor, but
attention must be paid to how the current sense signals are generated and routed.
Connection
Resistor R2 and capacitor C5 generate the current sense signal for phase 1 and resister R1 and capacitor C4
generate the current sense signal for phase 2. The R2-C5 and R1-C4 components must be connected directly to
the pads for L1 and L2, respectively.
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LAYOUT CONSIDERATIONS (continued)
Routing
The traces that connect to C5 and C4 should be made directly at the capacitor(s) and routed on an internal
signal plane to CS1, CSRT1 and CS2, CSRT2, respectively. In addition, a small value of R-C filter may be used
on the CSx and CSRTx lines, with these components placed close to the device. A 5.1-Ω resistor in series with
the CSx and CSRTx lines and a 100-pF capacitor between the CSx and CSRTx lines, provides additional
filtering, a prudent measure since the level of switching noise in a given layout is not fully known until the board
is being tested for the first time.
Differential Amplifier Signals
The differential amplifier provides optimum regulation at the load point.
Connection
The signal connections for VOUT and GSNS should be made across the closest capacitor to the load point. This
ensures the most accurate DC sensing and most noise free connection also.
Routing
Since the load point may be physically several inches, or more, from the device, it is very likely that the VOUT
and GSNS inputs to the differential amplifier are corrupted by switching noise. The signals should be routed on
an internal layer, and the R-C filter approach recommended for the CSx and CSRTx lines is applicable for these
lines as well.
High-Current Connections and Routing
Device Decoupling for VIN5 and BP5
The 1.0-µF decoupling capacitor for VIN5 should be placed close to pins 1 and 30 of the device. The decoupling
capacitor for BP5 should be placed close to pins 22 and 23 of the device.
Symmetry
Symmetry is especially important in the power processing components when considering the device placement
between the two phases. Input ceramic decoupling capacitors should be placed close to the upper MOSFETs
and the current path from the upper MOSFET drain to the lower MOSFET source should be on the PGND with
maximum copper area. Output capacitors should be placed symmetrically between the output inductor and lower
MOSFET for each phase.
SW Node
The SW node consists of the source of the upper MOSFET, the drain of the lower MOSFET,and the output
inductor. These components should be placed to minimize the area of the SW node. The area of the SW node
determines the amount of stray capacitance and inductance that causes ringing during switching transitions.
Lower MOSFET Gate Drive, LDRV1 and LDRV2
A resistor, with a value of between approximately 1.0 Ω and 2.2 Ω should be placed between LDRVx and the
gate of the respective MOSFET. The resistors are necessary if the falling SW node pulls the gate voltage below
GND. This can occur if the MOSFET QGD is larger than QGS. The traces for LDRVx should be wide, (0.05 to 0.1
inches) and routed on the top layer if possible. If routing must go to another layer, use multiple vias for
interconnect. The return signal from the MOSFET drain to PGND on the device should be as wide as the return
for LDRVx.
Upper MOSFET Gate Drive, HDRV1 and HDRV2
The traces for HDRVx and SWx should be wide, (0.05 to 0.1inches), and routed on the top layer if possible. If
routing must go to another layer, use multiple vias for interconnect.
30
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PACKAGE OPTION ADDENDUM
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10-Dec-2020
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
(4/5)
(6)
TPS40130DBT
NRND
TSSOP
DBT
30
60
RoHS & Green
NIPDAU
Level-2-260C-1 YEAR
-40 to 85
TPS40130
TPS40130DBTR
NRND
TSSOP
DBT
30
2000
RoHS & Green
NIPDAU
Level-2-260C-1 YEAR
-40 to 85
TPS40130
TPS40130RHBR
NRND
VQFN
RHB
32
3000
RoHS & Green
NIPDAU
Level-2-260C-1 YEAR
-40 to 85
TPS
40130
TPS40130RHBT
NRND
VQFN
RHB
32
250
RoHS & Green
NIPDAU
Level-2-260C-1 YEAR
-40 to 85
TPS
40130
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of
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