TPS54331
SLVS839G – JULY 2008 – REVISED JULY 2022
TPS54331 3-A, 28-V Input, Step Down DC-DC Converter With Eco-mode
1 Features
3 Description
•
•
•
The TPS54331 device is a 28-V, 3-A nonsynchronous buck converter that integrates a low
RDS(on) high-side MOSFET. To increase efficiency
at light loads, a pulse skipping Eco-mode feature
is automatically activated. Furthermore, the 1-μA
shutdown supply-current allows the device to be
used in battery-powered applications. Current mode
control with internal slope compensation simplifies
the external compensation calculations and reduces
component count while allowing the use of ceramic
output capacitors. A resistor divider programs the
hysteresis of the input undervoltage lockout. An
overvoltage transient protection circuit limits voltage
overshoots during startup and transient conditions.
A cycle-by-cycle current-limit scheme, frequency
foldback and thermal shutdown protect the device
and the load in the event of an overload condition.
The TPS54331 device is available in an 8-pin SOIC
package and 8-pin SO PowerPAD package that
have been internally optimized to improve thermal
performance.
•
•
•
•
•
•
•
•
2 Applications
•
•
•
Consumer applications such as set-top boxes,
CPE equipment, LCD displays, peripherals, and
battery chargers
Industrial and car-audio power supplies
5-V, 12-V, and 24-V distributed power systems
Device Information
PART NUMBER
(1)
PACKAGE(1)
BODY SIZE (NOM)
SOIC (8)
TPS54331
4.90 mm × 3.90 mm
SO PowerPAD (8)
For all available packages, see the orderable addendum at
the end of the datasheet.
100
Ren1
90
EN
VIN
Ren2
VIN
CI
80
70
TPS54331D
CBOOT
BOOT
LO
VOUT
PH
SS
COMP
D1
CO
RO1
Efficiency - %
•
3.5 to 28-V input voltage range
Adjustable output voltage down to 0.8 V
Integrated 80-mΩ high-side MOSFET supports up
to 3-A continuous output current
High efficiency at light loads with a pulse skipping
Eco-mode
Fixed 570-kHz switching frequency
Typical 1-μA shutdown quiescent current
Adjustable slow-start limits inrush currents
Programmable UVLO threshold
Overvoltage transient protection
Cycle-by-cycle current limit, frequency foldback,
and thermal shutdown protection
Available in easy-to-use SOIC8 package or
thermally-enhanced SOIC8 PowerPAD™ package
Create a custom design using the TPS54331 with
the WEBENCH® Power Designer
60
50
40
VI = 12 V
VI = 18 V
VI = 24 V
VI = 28 V
VI = 5 V
30
20
C1
CSS
C2
R3
10
VSENSE
GND
0
0.01
RO2
VO = 3.3 V
0.1
1
10
IL - Load Current - A
TPS54331 (D Package) Efficiency
Simplified Schematic
An IMPORTANT NOTICE at the end of this data sheet addresses availability, warranty, changes, use in safety-critical applications,
intellectual property matters and other important disclaimers. PRODUCTION DATA.
TPS54331
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Table of Contents
1 Features............................................................................1
2 Applications..................................................................... 1
3 Description.......................................................................1
4 Revision History.............................................................. 2
5 Pin Configuration and Functions...................................3
6 Specifications.................................................................. 4
6.1 Absolute Maximum Ratings........................................ 4
6.2 Handling Ratings.........................................................4
6.3 Recommended Operating Conditions.........................4
6.4 Thermal Information....................................................5
6.5 Electrical Characteristics.............................................5
6.6 Switching Characteristics............................................6
6.7 Typical Characteristics................................................ 7
7 Detailed Description........................................................9
7.1 Overview..................................................................... 9
7.2 Functional Block Diagram......................................... 10
7.3 Feature Description...................................................10
7.4 Device Functional Modes..........................................13
8 Application and Implementation.................................. 14
8.1 Application Information............................................. 14
8.2 Typical Application.................................................... 14
9 Power Supply Recommendations................................25
10 Layout...........................................................................26
10.1 Layout Guidelines................................................... 26
10.2 Layout Example...................................................... 26
10.3 Electromagnetic Interference (EMI)
Considerations............................................................ 27
11 Device and Documentation Support..........................28
11.1 Device Support........................................................28
11.2 Support Resources................................................. 28
11.3 Receiving Notification of Documentation Updates.. 28
11.4 Trademarks............................................................. 28
11.5 Electrostatic Discharge Caution.............................. 28
11.6 Glossary.................................................................. 28
12 Mechanical, Packaging, and Orderable
Information.................................................................... 28
4 Revision History
NOTE: Page numbers for previous revisions may differ from page numbers in the current version.
Changes from Revision F (October 2014) to Revision G (July 2022)
Page
• Updated the numbering format for tables, figures and cross-references throughout the document. .................1
• Updated Equation 2 ......................................................................................................................................... 10
Changes from Revision E (February 2012) to Revision F (October 2014)
Page
• Added Handling Ratings table, Feature Description section, Device Functional Modes, Application and
Implementation section, Power Supply Recommendations section, Layout section, Device and
Documentation Support section, and Mechanical, Packaging, and Orderable Information section. ................. 1
• Updated the inductor current equations for IL(RMS) and IL(PK) .......................................................................... 16
2
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5 Pin Configuration and Functions
BOOT
1
8
PH
VIN
2
7
EN
3
SS
4
8
PH
7
GND
3
6
COMP
4
5
VSENSE
BOOT
1
GND
VIN
2
6
COMP
EN
5
VSENSE
SS
Figure 5-1. 8-Pin SOIC D Package (Top View)
PowerPAD
(Pin 9)
TM
Figure 5-2. 8-Pin SO With PowerPAD™ DDA
Package (Top View)
Table 5-1. Pin Functions
PIN
I/O
DESCRIPTION
BOOT
O
A 0.1-μF bootstrap capacitor is required between the BOOT and PH pins. If the voltage on this
capacitor falls below the minimum requirement, the high-side MOSFET is forced to switch off
until the capacitor is refreshed.
2
VIN
I
This pin is the 3.5-V to 28-V input supply voltage.
3
EN
I
This pin is the enable pin. To disable, pull below 1.25 V. Float this pin to enable. Programming
the input undervoltage lockout with two resistors is recommended.
4
SS
I
This pin is slow-start pin. An external capacitor connected to this pin sets the output rise time.
5
VSENSE
I
This pin is the inverting node of the transconductance (gm) error amplifier.
6
COMP
O
This pin is the error-amplifier output and the input to the PWM comparator. Connect frequency
compensation components to this pin.
7
GND
—
Ground pin
8
PH
O
The PH pin is the source of the internal high-side power MOSFET.
9
PowerPAD
—
The PowerPAD is only available on the DDA package. For proper operation, the GND pin must
be connected to the exposed pad.
NO.
NAME
1
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6 Specifications
6.1 Absolute Maximum Ratings
over operating free-air temperature range (unless otherwise noted) (1)
Input voltage
MIN
MAX
VIN
–0.3
30
EN
–0.3
6
BOOT
38
VSENSE
–0.3
3
COMP
–0.3
3
SS
–0.3
3
BOOT-PH
Output voltage
Source current
Sink current
V
8
PH
–0.6
30
V
PH (10-ns transient from ground to negative peak)
–5
EN
100
μA
BOOT
100
mA
VSENSE
10
μA
PH
9
A
VIN
9
A
COMP
100
SS
200
Operating junction temperature, TJ
(1)
UNIT
–40
μA
150
°C
Stresses beyond those listed under absolute maximum ratings may cause permanent damage to the device. These are stress ratings
only, and functional operation of the device at these or any other conditions beyond those indicated under recommended operating
conditions is not implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
6.2 Handling Ratings
Tstg
Storage temperature range
V(ESD)
(1)
(2)
Electrostatic
discharge
Human body model (HBM), per ANSI/ESDA/JEDEC JS-001, all pins(1)
Charged device model (CDM), per JEDEC specification JESD22C101, all pins(2)
MIN
MAX
UNIT
–65
150
°C
–2
2
kV
–500
500
V
JEDEC document JEP155 states that 500-V HBM allows safe manufacturing with a standard ESD control process.
JEDEC document JEP157 states that 250-V CDM allows safe manufacturing with a standard ESD control process.
6.3 Recommended Operating Conditions
over operating free-air temperature range (unless otherwise noted)
TJ
4
MIN
MAX
Operating input voltage on (VIN pin)
3.5
28
V
Operating junction temperature
–40
150
°C
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6.4 Thermal Information
THERMAL METRIC(1)
D
DDA
8 PINS
8 PINS
RθJA
Junction-to-ambient thermal resistance
116.3
48.7
RθJC(top)
Junction-to-case (top) thermal resistance
53.7
52.4
RθJB
Junction-to-board thermal resistance
57.1
25.5
ψJT
Junction-to-top characterization parameter
12.9
8.4
ψJB
Junction-to-board characterization parameter
56.5
25.2
RθJC(bot)
Junction-to-case (bottom) thermal resistance
—
2.3
(1)
UNIT
°C/W
For more information about traditional and new thermal metrics, see the IC Package Thermal Metrics application report, SPRA953.
6.5 Electrical Characteristics
TJ = –40°C to 150°C, VIN = 3.5 to 28 V (unless otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
SUPPLY VOLTAGE (VIN PIN)
Internal undervoltage lockout threshold
Rising and falling
Shutdown supply current
EN = 0 V, VIN = 12 V, –40°C to 85°C
3.5
V
1
4
μA
Operating – non-switching supply current
VSENSE = 0.85 V
110
190
μA
Enable threshold
Rising and falling
1.25
1.35
Input current
Enable threshold – 50 mV
–1
μA
Input current
Enable threshold + 50 mV
–4
μA
ENABLE AND UVLO (EN PIN)
V
VOLTAGE REFERENCE
Voltage reference
0.772
0.8
0.828
BOOT-PH = 3 V, VIN = 3.5 V
115
200
BOOT-PH = 6 V, VIN = 12 V
80
150
–2 μA < I(COMP) < 2 μA, V(COMP) = 1 V
92
V
HIGH-SIDE MOSFET
On resistance
mΩ
ERROR AMPLIFIER
Error amplifier transconductance (gm)
Error amplifier DC
gain(1)
μmhos
VSENSE = 0.8 V
800
V/V
Error amplifier unity gain bandwidth(1)
5-pF capacitance from COMP to GND pins
2.7
MHz
Error amplifier source and sink current
V(COMP) = 1 V, 100-mV overdrive
±7
μA
Switch current to COMP
transconductance
VIN = 12 V
12
A/V
160
mA
5.8
A
165
°C
PULSE SKIPPING ECO-MODE™
Pulse skipping Eco-mode switch current
threshold
CURRENT LIMIT
Current-limit threshold
VIN = 12 V
3.5
THERMAL SHUTDOWN
Thermal shutdown
SLOW START (SS PIN)
(1)
Charge current
V(SS) = 0.4 V
2
μA
SS to VSENSE matching
V(SS) = 0.4 V
10
mV
Specified by design
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6.6 Switching Characteristics
TJ = –40°C to 150°C, VIN = 3.5 to 28 V (unless otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
456
570
684
kHz
105
130
ns
90%
93%
SWITCHING FREQUENCY
6
Switching frequency
VIN = 12 V, 25°C
Minimum controllable on time
VIN = 12 V, 25°C
Maximum controllable duty ratio(1)
BOOT-PH = 6 V
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6.7 Typical Characteristics
4
110
105
VIN = 12 V
EN = 0 V
Isd - Shutdown Current - mA
Rdson - On Resistance - mW
100
95
90
85
80
75
TJ = 150°C
3
2
TJ = 25°C
1
TJ = -40°C
70
65
60
-50
0
-25
0
25
50
75
100
TJ - Junction Temperature - °C
125
3
150
8
13
18
VI - Input Voltage - V
23
28
Figure 6-2. Shutdown Quiescent Current vs Input
Voltage
Figure 6-1. ON Resistance vs Junction
Temperature
0.8240
590
VIN = 12 V
0.8180
Vref - Voltage Reference - V
fsw - Oscillator Frequency - kHz
585
580
575
570
565
560
-25
0
25
50
75
100
125
0.8000
0.7940
0.7880
0.7760
-50
150
TJ - Junction Temperature - °C
-25
0
25
50
75
100
TJ - Junction Temperature - °C
125
150
Figure 6-4. Voltage Reference vs Junction
Temperature
Figure 6-3. Switching Frequency vs Junction
Temperature
7.50
140
VIN = 12 V
7.25
VIN = 12 V
Minimum Controllable Duty Ratio - %
Tonmin - Minimum Controllable On Time - ns
0.8060
0.7820
555
550
-50
0.8120
130
120
110
100
-50
-25
0
25
50
75
100
TJ - Junction Temperature - °C
125
150
Figure 6-5. Minimum Controllable On Time vs
Junction Temperature
7
6.75
6.50
6.25
6
5.75
5.50
-50
-25
0
25
50
75
100
TJ - Junction Temperature - °C
125
150
Figure 6-6. Minimum Controllable Duty Ratio vs
Junction Temperature
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6
2.10
Current Limit Threshold - A
ISS - Slow Start Charge Current - mA
TJ = 150°C
2
1.90
-50
TJ = 25°C
5
TJ = -40°C
4
3
-25
0
25
50
75
100
125
150
3
TJ - Junction Temperature - °C
Figure 6-7. SS Charge Current vs Junction
Temperature
8
8
13
18
VI - Input Voltage - V
23
28
Figure 6-8. Current-Limit Threshold vs Input
Voltage
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7 Detailed Description
7.1 Overview
The TPS54331 device is a 28-V, 3-A, step-down (buck) converter with an integrated high-side n-channel
MOSFET. To improve performance during line and load transients, the device implements a constant-frequency
current mode control, which reduces output capacitance and simplifies external frequency compensation design.
The TPS54331 device has a preset switching frequency of 570 kHz.
The TPS54331 device requires a minimum input voltage of 3.5 V for normal operation. The EN pin has an
internal pullup current source that can adjust the input-voltage undervoltage lockout (UVLO) with two external
resistors. In addition, the pullup current provides a default condition when the EN pin is floating for the device
to operate. The operating current is 110 μA (typical) when not switching and under no load. When the device is
disabled, the supply current is 1 μA (typical).
The integrated 80-mΩ high-side MOSFET allows for high-efficiency power-supply designs with continuous
output currents up to 3 A.
The TPS54331 device reduces the external component count by integrating the boot recharge diode. The bias
voltage for the integrated high-side MOSFET is supplied by an external capacitor on the BOOT to PH pin. The
boot capacitor voltage is monitored by an UVLO circuit and turns the high-side MOSFET off when the voltage
falls below a preset threshold of 2.1 V (typical). The output voltage can be stepped down to as low as the
reference voltage.
By adding an external capacitor, the slow-start time of the TPS54331 device can be adjustable, which enables
flexible output filter selection.
To improve the efficiency at light load conditions, the TPS54331 device enters a special pulse skipping Ecomode when the peak inductor current drops below 160 mA (typical).
The frequency foldback reduces the switching frequency during start-up and overcurrent conditions to help
control the inductor current. The thermal shutdown provides additional protection under fault conditions.
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7.2 Functional Block Diagram
EN
VIN
165 C
Thermal
Shutdown
1 mA
3 mA
Shutdown
Shutdown
Logic
1.25 V
Enable
Threshold
Enable
Comparator
Boot
Charge
™
ECO-MODE
Minimum Clamp
Boot
UVLO
BOOT
2.1V
Error
Amplifier
VSENSE
2 mA
PWM
Comparator
Gate
Drive
Logic
gm = 92 mA/V
DC gain = 800 V/V
BW = 2.7 MHz
Voltage
Reference
SS
2 kW
0.8 V
S
Shutdown
PWM
Latch
12 A/V
Current
Sense
R
80 mW
Q
S
Slope
Compensation
PH
Discharge
Logic
VSENSE
Frequency
Shift
Oscillator
GND
COMP
Maximum
Clamp
TPS54331D
7.3 Feature Description
7.3.1 Fixed-Frequency PWM Control
The TPS54331 device uses a fixed-frequency, peak-current mode control. The internal switching frequency of
the TPS54331 device is fixed at 570 kHz.
7.3.2 Voltage Reference (VREF)
The voltage reference system produces a ±2% initial accuracy voltage reference (±3.5% over temperature) by
scaling the output of a temperature-stable band-gap circuit. The typical voltage reference is designed at 0.8 V.
7.3.3 Bootstrap Voltage (BOOT)
The TPS54331 device has an integrated boot regulator and requires a 0.1-μF ceramic capacitor between the
BOOT and PH pins to provide the gate-drive voltage for the high-side MOSFET. A ceramic capacitor with
an X7R- or X5R-grade dielectric is recommended because of the stable characteristics over temperature and
voltage. To improve dropout, the TPS54331 device is designed to operate at 100% duty cycle as long as the
BOOT-to-PH pin voltage is greater than 2.1 V (typical).
7.3.4 Enable and Adjustable Input Undervoltage Lockout (VIN UVLO)
The EN pin has an internal pullup current-source that provides the default condition of the device while operating
when the EN pin floats.
The TPS54331 device is disabled when the VIN pin voltage falls below the internal VIN UVLO threshold. Using
an external VIN UVLO to add hysteresis is recommended unless the VIN voltage is greater than (VOUT + 2 V).
To adjust the VIN UVLO with hysteresis, use the external circuitry connected to the EN pin as shown in Figure
7-1. When the EN pin voltage exceeds 1.25 V, an additional 3 μA of hysteresis is added. Use Equation 1 and
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Equation 2 to calculate the resistor values required for the desired VIN UVLO threshold voltages. The VSTOP
threshold must always be greater than 3.5 V.
TPS54331
VIN
Ren1
1 mA
3 mA
+
EN
Ren2
1.25 V
-
Figure 7-1. Adjustable Input Undervoltage Lockout
Ren1 =
VSTART - VSTOP
3 mA
(1)
where
•
•
VSTART is the input start threshold voltage.
VSTOP is the input stop threshold voltage.
(2)
where
•
VEN is the enable threshold voltage of 1.25 V.
7.3.5 Programmable Slow Start Using SS Pin
Programming the slow-start time externally is highly recommended because no slow-start time is implemented
internally. The TPS54331 device effectively uses the lower voltage of the internal voltage reference or the SS pin
voltage as the reference voltage of the power supply that is fed into the error amplifier and regulates the output
accordingly. A capacitor (CSS) on the SS pin to ground implements a slow-start time. The TPS54331 device
has an internal pullup current-source of 2 μA that charges the external slow-start capacitor. Use Equation 3 to
calculate the slow-start time (10% to 90%).
TSS (ms ) =
CSS (nF ) ´ Vref (V )
ISS (mA )
(3)
where
•
•
Vref is 0.8 V.
ISS is 2 μA.
The slow-start time must be set between 1 ms to 10 ms to ensure good start-up behavior. The value of the
slow-start capacitor must not exceed 27 nF.
During normal operation, the TPS54331 device stops switching if the input voltage drops below the VIN UVLO
threshold, the EN pin is pulled below 1.25 V, or a thermal shutdown event occurs.
7.3.6 Error Amplifier
The TPS54331 device has a transconductance amplifier for the error amplifier. The error amplifier compares
the VSENSE voltage to the internal effective voltage reference presented at the input of the error amplifier.
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The transconductance of the error amplifier is 92 μA/V during normal operation. Frequency compensation
components are connected between the COMP pin and ground.
7.3.7 Slope Compensation
To prevent the subharmonic oscillations when operating the device at duty cycles greater than 50%, the
TPS54331 device adds a built-in slope compensation, which is a compensating ramp to the switch-current
signal.
7.3.8 Current-Mode Compensation Design
To simplify design efforts using the TPS54331 device, the typical designs for common applications are listed
in Table 7-1. For designs using ceramic output capacitors, proper derating of ceramic output capacitance
is recommended when performing the stability analysis because the actual ceramic capacitance drops
considerably from the nominal value when the applied voltage increases. See Section 8.2.2 for the detailed
guidelines or use the WEBENCH software tool (www.TI.com/WEBENCH).
Table 7-1. Typical Designs (Refer to the Simplified Schematic)
VIN
(V)
VOUT
(V)
ƒSW
(kHz)
Lo
(μH)
Co
RO1
(kΩ)
RO2
(kΩ)
C2
(pF)
C1
(pF)
R3
(kΩ)
12
5
570
6.8
Ceramic 33 μF, ×2
10
1.91
39
4700
49.9
12
3.3
570
6.8
Ceramic 47μF, ×2
10
3.24
47
1000
29.4
12
1.8
570
4.7
Ceramic 100 μF
10
8.06
68
5600
29.4
12
0.9
570
3.3
Ceramic 100 μF, ×2
10
80.6
56
5600
29.4
12
5
570
6.8
Aluminum 330 μF, 160 mΩ
10
1.91
68
120
29.4
12
3.3
570
6.8
Aluminum 470 μF, 160 mΩ
10
3.24
82
220
10
12
1.8
570
4.7
SP 100 μF, 15 mΩ
10
8.06
68
5600
29.4
12
0.9
570
3.3
SP 330 μF, 12 mΩ
10
80.6
100
1200
49.9
7.3.9 Overcurrent Protection and Frequency Shift
The TPS54331 device implements current mode control that uses the COMP pin voltage to turn off the high-side
MOSFET on a cycle-by-cycle basis. During each cycle, the switch current and the COMP pin voltage are
compared. When the peak inductor current intersects the COMP pin voltage, the high-side switch is turned off.
During overcurrent conditions that pull the output voltage low, the error amplifier responds by driving the COMP
pin high, causing the switch current to increase. The COMP pin has a maximum clamp internally, which limits the
output current.
The TPS54331 device provides robust protection during short circuits. Overcurrent runaway is possible in the
output inductor during a short circuit at the output. The TPS54331 device solves this issue by increasing the off
time during short-circuit conditions by lowering the switching frequency. The switching frequency is divided by
1, 2, 4, and 8 as the voltage ramps from 0 to 0.8 V on VSENSE pin. The relationship between the switching
frequency and the VSENSE pin voltage is listed in Table 7-2.
Table 7-2. Switching Frequency Conditions
Switching Frequency
VSENSE Pin Voltage
570 kHz
VSENSE ≥ 0.6 V
570 kHz / 2
0.6 V > VSENSE ≥ 0.4 V
570 kHz / 4
0.4 V > VSENSE ≥ 0.2 V
570 kHz / 8
0.2 V > VSENSE
7.3.10 Overvoltage Transient Protection
The TPS54331 device incorporates an overvoltage transient-protection (OVTP) circuit to minimize output voltage
overshoot when recovering from output fault conditions or strong unload transients. The OVTP circuit includes
an overvoltage comparator to compare the VSENSE pin voltage and internal thresholds. When the VSENSE
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pin voltage goes above 109% × VREF, the high-side MOSFET is forced off. When the VSENSE pin voltage falls
below 107% × VREF, the high-side MOSFET is enabled again.
7.3.11 Thermal Shutdown
The device implements an internal thermal shutdown to protect the device if the junction temperature exceeds
165°C. The thermal shutdown forces the device to stop switching when the junction temperature exceeds the
thermal trip threshold. When the die temperature decreases below 165°C, the device reinitiates the power-up
sequence.
7.4 Device Functional Modes
7.4.1 Eco-mode
The TPS54331 device is designed to operate in pulse skipping Eco-mode at light load currents to boost light
load efficiency. When the peak inductor current is lower than 160 mA (typical), the COMP pin voltage falls to
0.5 V (typical) and the device enters Eco-mode. When the device is in Eco-mode, the COMP pin voltage is
clamped at 0.5 V internally, which prevents the high-side integrated MOSFET from switching. The peak inductor
current must rise above 160 mA for the COMP pin voltage to rise above 0.5 V and exit Eco-mode. Because
the integrated current comparator catches the peak inductor current only, the average load current entering
Eco-mode varies with the applications and external output filters.
7.4.2 Operation With VIN < 3.5 V
The device is recommended to operate with input voltages above 3.5 V. The typical VIN UVLO threshold is not
specified and the device can operate at input voltages down to the UVLO voltage. At input voltages below the
actual UVLO voltage, the device does not switch. If the EN pin is externally pulled up or left floating, the device
becomes active when the VIN pin passes the UVLO threshold. Switching begins when the slow-start sequence
is initiated.
7.4.3 Operation With EN Control
The enable threshold voltage is 1.25 V (typical). When the EN pin is held below that voltage, the device is
disabled and switching is inhibited even if the VIN pin is above the UVLO threshold. The IC quiescent current
is reduced in this state. If the EN voltage increases above the threshold while the VIN pin is above the UVLO
threshold, the device becomes active. Switching is enabled, and the slow-start sequence is initiated.
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8 Application and Implementation
Note
Information in the following applications sections is not part of the TI component specification,
and TI does not warrant its accuracy or completeness. TI’s customers are responsible for
determining suitability of components for their purposes, as well as validating and testing their design
implementation to confirm system functionality.
8.1 Application Information
The TPS54331 device is typically used as a step-down converter, which converts a voltage from 3.5 V to 28 V to
a lower voltage. WEBENCH software is available to aid in the design and analysis of circuits.
For additional design needs, see the following devices:
Parameter
TPS54231
TPS54232
TPS54233
TPS54331
TPS54332
IO (maximum)
2A
2A
2A
3A
3.5 A
Input voltage range
3.5 to 28 V
3.5 to 28 V
3.5 to 28 V
3.5 to 28 V
3.5 to 28 V
Switching frequency (typical)
570 kHz
1000 kHz
285 kHz
570 kHz
1000 kHz
Switch current limit (minimum)
2.3 A
2.3 A
2.3 A
3.5 A
4.2 A
8SOIC
8SOIC
8SO PowerPAD
8SO PowerPAD
Pin and package
8SOIC
8SOIC
8.2 Typical Application
Vout 3.3 V
Iout Max 3 A
6.8 µH
0.1 μF
47 µF
Vin 7 V – 28 V
4.7µF
4.7µF
0.01 μF
0Ω
10.2 kΩ
1000 pF
332 kΩ
47 µF
47 pF
0.01 µF
3.24 kΩ
29.4 kΩ
68.1 kΩ
Figure 8-1. Typical Application Schematic
8.2.1 Design Requirements
For this design example, use the values listed in Table 8-1 as the input parameters
Table 8-1. Design Parameters
14
Design Parameter
Example Value
Input voltage range
7 to 28 V
Output voltage
3.3 V
Input ripple voltage
300 mV
Output ripple voltage
30 mV
Output current rating
3A
Operating frequency
570 kHz
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8.2.2 Detailed Design Procedure
The following design procedure can be used to select component values for the TPS54331 device. Alternately,
the WEBENCH software can be used to generate a complete design. The WEBENCH software uses an iterative
design procedure and accesses a comprehensive database of components when generating a design. This
section presents a simplified discussion of the design process.
8.2.2.1 Custom Design with WEBENCH Tools
Click here to create a custom design using the WEBENCH® Power Designer.
1. Start by entering your VIN, VOUT and IOUT requirements.
2. Optimize your design for key parameters like efficiency, footprint and cost using the optimizer dial and
compare this design with other possible solutions from Texas Instruments.
3. WEBENCH Power Designer provides you with a customized schematic along with a list of materials with real
time pricing and component availability.
4. In most cases, you will also be able to:
• Run electrical simulations to see important waveforms and circuit performance,
• Run thermal simulations to understand the thermal performance of your board,
• Export your customized schematic and layout into popular CAD formats,
• Print PDF reports for the design, and share your design with colleagues.
8.2.2.2 Switching Frequency
The switching frequency for the TPS54331 device is fixed at 570 kHz.
8.2.2.3 Output Voltage Set-Point
The output voltage of the TPS54331 device is externally adjustable using a resistor divider network. As shown in
Figure 8-1, this divider network is comprised of R5 and R6. The relationship of the output voltage to the resistor
divider is given by Equation 4 and Equation 5.
R5 ´ Vref
VOUT - Vref
(4)
é R5 ù
VOUT = Vref ´ ê
+1ú
ë R6 û
(5)
R6 =
Select a value of R5 to be approximately 10 kΩ. Slightly increasing or decreasing the value of R5 can result in
closer output-voltage matching when using standard value resistors. In this design, R4 = 10.2 kΩ and R = 3.24
kΩ, resulting in a 3.31-V output voltage. The 0-Ω resistor, R4, is provided as a convenient location to break the
control loop for stability testing.
8.2.2.4 Input Capacitors
The TPS54331 device requires an input decoupling capacitor and, depending on the application, a bulk input
capacitor. The typical recommended value for the decoupling capacitor is 10 μF. A high-quality ceramic type X5R
or X7R is recommended. The voltage rating must be greater than the maximum input voltage. A smaller value
can be used as long as all other requirements are met, however, a value of 10 μF has been shown to work
well in a wide variety of circuits. Additionally, some bulk capacitance can be required, especially if the TPS54331
circuit is not located within approximately two inches from the input voltage source. The value for this capacitor
is not critical but must be rated to handle the maximum input voltage including ripple voltage, and should filter
the output so that input ripple voltage is acceptable. For this design, two 4.7-μF capacitors are used for the input
decoupling capacitor. The capacitors are X7R dielectric rated for 50 V. The equivalent series resistance (ESR)
is approximately 2 mΩ and the current rating is 3 A. Additionally, a small 0.01-μF capacitor is included for high
frequency filtering.
Use Equation 6 to calculate the input ripple voltage.
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DVIN =
IO(MAX) ´ 0.25
CBULK ´ ƒSW
(
+ IO(MAX) ´ ESRMAX
)
(6)
where
•
•
•
•
IOUT(MAX) is the maximum load current.
ƒSW is the switching frequency.
CBULK is the bulk capacitor value.
ESRMAX is the maximum series resistance of the bulk capacitor.
The maximum RMS ripple current must also be checked. For worst case conditions, use Equation 7 to calculate
the maximum-RMS input ripple current, ICIN(RMS).
ICIN(RMS) =
IO(MAX)
(7)
2
In this case, the input ripple voltage is 143 mV and the RMS ripple current is 1.5 A.
Note
The actual input voltage ripple is greatly affected by parasitics associated with the layout and the
output impedance of the voltage source.
The actual input voltage ripple for this circuit is listed in Table 8-1 and is larger than the calculated value.
This measured value is still below the specified input limit of 300 mV. The maximum voltage across the input
capacitors is VIN(MAX) + ΔVIN / 2. The selected bulk and bypass capacitors are each rated for 50 V and the ripple
current capacity is greater than 3 A, both providing ample margin. The maximum ratings for voltage and current
must not be exceeded under any circumstance.
8.2.2.5 Output Filter Components
Two components must be selected for the output filter, L1 and C2. Because the TPS54331 device is an
externally compensated device, a wide range of filter component types and values can be supported.
8.2.2.5.1 Inductor Selection
To calculate the minimum value of the output inductor, use Equation 8.
LMIN =
VOUT(MAX) ´
(VIN(MAX) - VOUT )
VIN(MAX) ´ KIND ´ IOUT ´ ƒSW
(8)
where
•
KIND is a coefficient that represents the amount of inductor ripple current relative to the maximum output
current.
In general, this value is at the discretion of the designer; however, the following guidelines may be used. For
designs using low-ESR output capacitors, such as ceramics, a value as high as KIND = 0.3 can be used. When
using higher ESR output capacitors, KIND = 0.2 yields better results.
For this design example, use KIND = 0.3 and the minimum inductor value is calculated as 5.7 μH. For this design,
a large value was selected: 6.8 μH.
For the output filter inductor, do not exceed the RMS current and saturation current ratings. Use Equation 9 to
calculate the inductor ripple current (ILPP).
ILPP =
16
(
VOUT × VIN(MAX)
-
VOUT
)
VIN(MAX) × LOUT ´ ƒSW ´ 0.8
(9)
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Use Equation 10 to calculate the RMS inductor current.
IL(RMS) = I2 OUT(MAX) +
1
× I2LPP
12
(10)
Use Equation 11 to calculate the peak inductor current.
IL(PK) = IOUT(MAX) +
ILPP
2
(11)
For this design, the RMS inductor current is 3.01 A and the peak inductor current is 3.47 A. The selected
inductor is a Sumida CDRH103-6R8, 6.8 μH. This inductor has a saturation current rating of 3.84 A and an
RMS current rating of 3.6 A, which meets these requirements. Smaller or larger inductor values can be used
depending on the amount of ripple current the designer wants to allow, so long as the other design requirements
are met. Larger value inductors have lower AC current and result in lower output voltage ripple, while smaller
inductor values increase AC current and output voltage ripple. In general, inductor values for use with the
TPS54331 device are in the range of 6.8 μH to 47 μH.
8.2.2.6 Capacitor Selection
The important design factors for the output capacitor are DC voltage rating, ripple current rating, and equivalent
series resistance (ESR). The DC voltage and ripple current ratings cannot be exceeded. The ESR is important
because along with the inductor current it determines the amount of output ripple voltage. The actual value of the
output capacitor is not critical, but some practical limits do exist. Consider the relationship between the desired
closed-loop crossover frequency of the design and LC corner frequency of the output filter. In general, keeping
the closed-loop crossover frequency at less than 1/5 of the switching frequency is desired. With high switching
frequencies such as the 570-kHz frequency of this design, internal circuit limitations of the TPS54331 device
limit the practical maximum crossover frequency to approximately 25 kHz. In general, the closed-loop crossover
frequency must be higher than the corner frequency determined by the load impedance and the output capacitor.
Use Equation 12 to calculate the limits of the minimum capacitor value.
CO(MIN) = 1/ (2 ´ p ´ RO ´ FCO(MAX) )
(12)
where
•
•
RO is the output load impedance (VO / IO).
FCO(MAX) is the desired crossover frequency.
For a desired maximum crossover of 25 kHz, the minimum value for the output capacitor is approximately 5.8
μF. This value may not satisfy the output ripple voltage requirement. The output ripple voltage consists of two
components: the voltage change because of the charge and discharge of the output filter capacitance and the
voltage change because the ripple current times the ESR of the output filter capacitor. Use Equation 13 to
estimate the output ripple voltage.
é (D - 0.5)
ù
+ RESR ú
VOPP = ILPP ê
ë 4 ´ ƒSW ´ CO
û
(13)
The maximum ESR of the output capacitor can be determined from the amount of allowable output ripple as
specified in the initial design parameters. The contribution to the output ripple voltage because the ESR is the
inductor ripple current times the ESR of the output filter. Therefore, use Equation 14 to calculate the maximum
specified ESR as listed in the capacitor data sheet.
ESRmax =
VOPP(MAX)
ILPP
-
(D
- 0.5 )
4 ´ ƒSW ´ CO
(14)
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where
•
VOPP(MAX) is the desired maximum peak-to-peak output ripple.
Use Equation 15 to calculate the maximum RMS ripple current.
ICOUT(RMS) =
(
) ö÷
æ VOUT × VIN(MAX) - VOUT
× ç
ç VIN(MAX) × LOUT × ƒSW × NC
12
è
1
÷
ø
(15)
where
•
NC is the number of output capacitors in parallel.
For this design example, two 47-μF ceramic output capacitors are selected for C8 and C9. These capacitors are
TDK C3216X5R0J476MT, rated at 6.3 V with a maximum ESR of 2 mΩ and a ripple current rating in excess of
3 A. The calculated total RMS ripple current is 161 mA (80.6 mA each) and the maximum total ESR required
is 43 mΩ. These output capacitors exceed the requirements by a wide margin and result in a reliable, highperformance design.
Note
The actual capacitance in circuit may be less than the catalog value when the output is operating at
the desired output of 3.3 V.
The selected output capacitor must be rated for a voltage greater than the desired output voltage plus half of the
ripple voltage. Any derating amount must also be included. Other capacitor types work well with the TPS54331
device, depending on the needs of the application.
8.2.2.7 Compensation Components
The external compensation used with the TPS54331 device allows for a wide range of output filter
configurations. A large range of capacitor values and types of dielectric are supported. The design example
uses ceramic X5R dielectric output capacitors, but other types are supported.
A Type II compensation scheme is recommended for the TPS54331 device. The compensation components are
selected to set the desired closed-loop crossover frequency and phase margin for output filter components. The
Type II compensation has the following characteristics: a DC gain component, a low frequency pole, and a mid
frequency zero-pole pair.
Use Equation 16 to calculate the DC gain.
GDC =
Vggm ´ VREF
VO
(16)
where
•
•
Vggm is 800.
VREF is 0.8 V.
Use Equation 17 to calculate the low-frequency pole.
VPO = 1/ (2 ´ p ´ ROO ´ CZ )
(17)
Use Equation 18 to calculate the mid-frequency zero.
FZ1 = 1/ (2 ´ p ´ R Z ´ CZ )
(18)
Use Equation 19 to calculate the mid-frequency pole.
18
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FP1 = 1/ (2 ´ p ´ R Z ´ CP )
(19)
The first step is to select the closed-loop crossover frequency. In general, the closed-loop crossover frequency
must be less than 1/8 of the minimum operating frequency. However, for the TPS54331 device, not exceeding
25 kHz for the maximum closed-loop crossover frequency is recommended. The second step is to calculate the
required gain and phase boost of the crossover network. By definition, the gain of the compensation network
must be the inverse of the gain of the modulator and output filter. For this design example, where the ESR zero
is much higher than the closed-loop crossover frequency, the gain of the modulator and output filter can be
approximated by Equation 20.
Gain = - 20 log (2 ´ p ´ RSENSE ´ FCO ´ CO )
(20)
where
•
•
•
RSENSE is 1 Ω / 12.
FCO is the closed-loop crossover frequency.
CO is the output capacitance.
Use Equation 21 to calculate the phase loss.
PL = a tan (2 ´ p ´ FCO ´ RESR ´ CO ) - a tan (2 ´ p ´ FCO ´ RO ´ CO )
(21)
where
•
•
RESR is the equivalent series resistance of the output capacitor.
RO is VO / IO.
The measured overall loop-response for the circuit is given in Figure 8-7. The actual closed-loop crossover
frequency is higher than intended at approximately 25 kHz, which is primarily because variation in the actual
values of the output filter components and tolerance variation of the internal feedforward gain circuitry. Overall,
the design has greater than 60 degrees of phase margin and is completely stable over all combinations of line
and load variability.
Now that the phase loss is known, the required amount of phase boost to meet the phase margin requirement
can be determined. Use Equation 22 to calculate the required phase boost.
PB = (PM - 90 deg ) - PL
(22)
where
•
PM is the desired phase margin.
A zero-pole pair of the compensation network is placed symmetrically around the intended closed-loop
frequency to provide maximum phase boost at the crossover point. The amount of separation can be calculated
with Equation 23. Use Equation 24 and Equation 25 to calculate the resultant zero and pole frequencies.
ö
æ PB
k = tanç
+ 45 deg ÷
ø
è 2
FZ 1 =
(23)
FCO
k
(24)
FP1 = FCO ´ k
(25)
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The low-frequency pole is set so that the gain at the crossover frequency is equal to the inverse of the gain of
the modulator and output filter. Because of the relationships established by the pole and zero relationships, use
Equation 26 to calculate the value of RZ.
2 × p × FCO × VO × CO × ROA
GMICOMP × Vggm × VREF
RZ =
(26)
where
•
•
•
•
•
•
•
VO is the output voltage.
CO is the output capacitance.
FCO is the desired crossover frequency.
ROA is 8 MΩ.
GMCOMP is 12 A/V.
Vggm is 800.
VREF is 0.8 V.
With the value of RZ known, use Equation 27 and Equation 28 to calculate the values of CZ and CP.
CZ =
CP =
1
2 ´ p ´ FZ 1 ´ Rz
(27)
1
2 ´ p ´ FP1 ´ Rz
(28)
For this design, the two 47-μF output capacitors are used. For ceramic capacitors, the actual output capacitance
is less than the rated value when the capacitors have a DC bias voltage applied, which occurs in a DC-DC
converter. The actual output capacitance can be as low as 54 μF. The combined ESR is approximately 0.001 Ω.
Using Equation 20 and Equation 21, the output stage gain and phase loss are equivalent as:
• Gain = –2.26 dB
• PL = –83.52 degrees
For 70 degrees of phase margin, Equation 22 requires 63.52 degrees of phase boost.
Use Equation 23, Equation 24, and Equation 25 to calculate the zero and pole frequencies of the following
values:
• FZ1 = 5883 Hz
• FP1 = 106200 Hz
Use Equation 26, Equation 27, and Equation 28 to calculate the values of RZ, CZ, and CP.
2 ´ p ´ 25000 ´ 3.3 ´ 54 ´ 10-6 ´ 8 ´ 106
= 29.2 kW
12 ´ 800 ´ 0.8
(29)
Cz =
1
= 928 pF
2 ´ p ´ 6010 ´ 29200
(30)
Cp =
1
= 51 pF
2 ´ p ´ 103900 ´ 29200
(31)
Rz =
Referring to Figure 8-1 and using standard values for R3, C6, and C7, the calculated values are as follows:
• R3 = 29.4 kΩ
• C6 = 1000 pF
20
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C7 = 47 pF
8.2.2.8 Bootstrap Capacitor
Every TPS54331 design requires a bootstrap capacitor, C4. The bootstrap capacitor must have a value of 0.1
μF. The bootstrap capacitor is located between the PH pin and BOOT pin. The bootstrap capacitor must be a
high-quality ceramic type with X7R or X5R grade dielectric for temperature stability.
8.2.2.9 Catch Diode
The TPS54331 device is designed to operate using an external catch diode between the PH and GND pins.
The selected diode must meet the absolute maximum ratings for the application. The reverse voltage must be
higher than the maximum voltage at the PH pin, which is VIN(MAX) + 0.5 V. The peak current must be greater
than IOUT(MAX) plus half the peak-to peak-inductor current. The forward-voltage drop must be small for higher
efficiencies. The catch diode conduction time is (typically) longer than the high-side FET on time, so attention
paid to diode parameters can make a marked improvement in overall efficiency. Additionally, check that the
selected device is capable of dissipating the power losses. For this design, a Diodes, Inc. B340A was selected,
with a reverse voltage of 40 V, forward current of 3 A, and a forward-voltage drop of 0.5 V.
8.2.2.10 Output Voltage Limitations
Because of the internal design of the TPS54331 device, any given input voltage has both upper and lower output
voltage limits. The upper limit of the output-voltage set point is constrained by the maximum duty cycle of 91%
and is calculated with Equation 32.
VO(MAX) = 0.91 ×
((V
IN(MIN)
)
- IO(MAX) × RDS(on)max + VD
) - (I
O(MAX)
× RL
) - VD
(32)
where
•
•
•
•
VIN(MIN) is the minimum input voltage.
IO(MAX) is the maximum load current.
VD is the catch diode forward voltage.
RL is the output inductor series resistance.
The equation assumes the maximum on resistance for the internal high-side FET.
The lower limit is constrained by the minimum controllable on time, which can be as high as 130 ns. Use
Equation 33 to calculate the approximate minimum output voltage for a given input voltage and minimum load
current.
VO(MIN) = 0.089 ×
((V
IN(MAX)
)
- IO(MIN) × RDS(on)min + VD
) - (I
O(MIN)
× RL
) - VD
(33)
where
•
•
•
•
VIN(MAX) is the maximum input voltage.
IO(MIN) is minimum load current.
VD is the catch diode forward voltage.
RL is the output inductor series resistance.
The nominal on-resistance for the high-side FET in Equation 33 is assumed. Equation 33 accounts for the worst
case variation of operating-frequency set point. Any design operating near the operational limits of the device
must be carefully checked to ensure proper functionality.
8.2.2.11 Power Dissipation Estimate
The following formulas show how to estimate the device power dissipation under continuous-conduction mode
(CCM) operations. These formulas must not be used if the device is working in the discontinuous-conduction
mode (DCM) or pulse-skipping Eco-mode.
The device power dissipation includes:
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1. Conduction loss:
Pcon = IOUT 2 × RDS(on) × VOUT / VIN
where
• IOUT is the output current (A).
• RDS(on) is the on-resistance of the high-side MOSFET (Ω).
• VOUT is the output voltage (V).
• VIN is the input voltage (V).
2. Switching loss:
Psw = 0.5 × 10-9 × VIN 2 × IOUT × ƒSW
where
• ƒSW is the switching frequency (Hz).
3. Gate charge loss:
Pgc = 22.8 × 10-9 × ƒSW
4. Quiescent current loss
Pq = 0.11 × 10-3 × VIN
Therefore:
Ptot = Pcon + Psw + Pgc + Pq
where
•
Ptot is the total device power dissipation (W).
For given TA :
TJ = TA + Rth × Ptot
where
•
•
•
TJ is the junction temperature (°C).
TA is the ambient temperature (°C).
Rth is the thermal resistance of the package (°C/W).
For given TJMAX = 150°C:
TAMAX = TJMAX – Rth × Ptot
where
•
•
22
TJMAX is maximum junction temperature (°C).
TAMAX is maximum ambient temperature (°C).
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8.2.3 Application Curves
100
100
VIN = 7 V
VIN = 21 V
70
VIN = 14 V
70
Efficiency - %
VIN = 28 V
60
50
40
VIN = 21 V
60
VIN = 28 V
50
40
30
30
20
20
10
10
0
0
0
0.5
2
1.5
IO - Output Current - A
1
0
3
2.5
0.02 0.04 0.06 0.08 0.1 0.12 0.14 0.16 0.18 0.2
IO - Output Current - A
TPS54331 (D package)
TPS54331 (D package)
Figure 8-3. Low Current Efficiency
Figure 8-2. Efficiency
1.004
3.38
1.0035
3.37
1.003
3.36
1.0025
3.35
1.002
VO - Output Voltage - V
VIN = 28 V
1.0015
VIN = 21 V
1.001
VIN = 7 V
1.0005
1
VIN = 14 V
IO = 0 A
IO = 1.5 A
IO = 3 A
3.34
3.33
3.32
3.31
3.3
0.9995
3.29
0.999
3.28
0.9985
0
0.5
1
1.5
2
2.5
IO - Output Current - A
3
3.5
0
5
TPS54331 (D package)
10
15
20
VI - Input Voltage -V
25
30
TPS54331 (D package)
Figure 8-4. Load Regulation
Figure 8-5. Line Regulation
Gain - dB
VOUT
Output Current
70
210
60
180
50
150
40
120
30
90
20
60
10
30
0
0
-10
-30
-20
-60
-30
10
t - Time - 200 ms/div
Figure 8-6. Transient Response
100
1k
10k
f - Frequency - Hz
100k
Phase - deg
Output Regulation - %
VIN = 14 V
80
80
Efficiency - %
VIN = 7 V
90
90
-90
1M
Figure 8-7. Loop Response
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VOUT
VIN
PH
PH
t - Time - 1 ms/div
t - Time - 1 ms/div
Figure 8-8. Output Ripple
Figure 8-9. Input Ripple
ENA
VIN
VOUT
VOUT
t - Time - 5 ms/div
t - Time - 5 ms/div
Figure 8-10. Start-Up
Figure 8-11. Start-Up Relative to Enable
1.75
30
25
VO - Output Voltage - V
VO - Output Voltage - V
1.5
IO = 2 A
1.25
1
IO = 2 A
20
15
10
IO = 3 A
0.75
5
IO = 3 A
0.5
0
3
8
13
18
VI - Input Volatage - V
23
28
Figure 8-12. Typical Minimum Output Voltage vs
Input Voltage
24
3
8
13
18
VI - Input Voltage - V
23
28
Figure 8-13. Typical Maximum Output Voltage vs
Input Voltage
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150
150
125
125
TJ - Junction Temperature - °C
TJ - Junction Temperature - °C
www.ti.com
100
75
50
25
0
0.2
0.4
0.6
0.8
1
1.2
100
75
50
25
0
0.2
0.4
0.6
0.8
1
1.2
1.4
PD - Power Dissipation - W
PD - Power Dissipation - W
TPS54331 (D package)
TPS54331 (D package)
Figure 8-14. Maximum Power Dissipation vs
Junction Temperature
1.6
1.8
2
Figure 8-15. Maximum Power Dissipation vs
Junction Temperature
9 Power Supply Recommendations
The devices are designed to operate from an input-voltage supply range between 3.5 V and 28 V. This input
supply must be well regulated. If the input supply is located more than a few inches from the converter, additional
bulk capacitance can be required in addition to the ceramic bypass capacitors. An electrolytic capacitor with a
value of 100 μF is a typical choice.
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10 Layout
10.1 Layout Guidelines
The VIN pin must be bypassed to ground with a low-ESR ceramic bypass capacitor. Take care to minimize
the loop area formed by the bypass capacitor connections, the VIN pin, and the anode of the catch diode.
The typical recommended bypass capacitor is 10-μF ceramic with a X5R or X7R dielectric and the optimum
placement is closest to the VIN pins and the source of the anode of the catch diode. Figure 10-1 shows a PCB
layout example. The GND pin must be tied to the PCB ground plane at the pin of the device. The source of the
low-side MOSFET must be connected directly to the top-side PCB ground area used to tie together the ground
sides of the input and output capacitors as well as the anode of the catch diode. The PH pin must be routed to
the cathode of the catch diode and to the output inductor. Because the PH connection is the switching node, the
catch diode and output inductor must be located very close to the PH pins, and the area of the PCB conductor
minimized to prevent excessive capacitive coupling. For operation at full rated load, the top-side ground area
must provide adequate heat dissipating area. The TPS54331 device uses a fused lead frame so that the GND
pin acts as a conductive path for heat dissipation from the die. Many applications have larger areas of internal or
back-side ground plane available, and the top-side ground area can be connected to these areas using multiple
vias under or adjacent to the device to help dissipate heat. The additional external components can be placed
approximately as shown. Obtaining acceptable performance with alternate layout schemes may be possible,
however this layout has been shown to produce good results and is intended as a guideline.
10.2 Layout Example
OUTPUT
FILTER
CAPACITOR
TOPSIDE
GROUND
AREA
Route BOOT CAPACITOR
trace on other layer to provide
wide path for topside ground
Vout
Feedback Trace
OUTPUT
INDUCTOR
CATCH
DIODE
PH
INPUT
BYPASS
CAPACITOR
BOOT
Vin
UVLO
RESISTOR
DIVIDER
VIN
GND
EN
COMP
SS
VSENSE
SLOW START
CAPACITOR
Thermal VIA
BOOT
CAPACITOR
PH
RESISTOR
DIVIDER
COMPENSATION
NETWORK
Signal VIA
Figure 10-1. TPS54331 Device D Board Layout
26
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10.3 Electromagnetic Interference (EMI) Considerations
As EMI becomes a rising concern in more and more applications, the internal design of the TPS54331 device
includes features to reduce the EMI. The high-side MOSFET gate drive is designed to reduce the PH pin voltage
ringing. The internal IC rails are isolated to decrease the noise sensitivity. A package bond wire scheme is used
to lower the parasitics effects.
To achieve the best EMI performance, external component selection and board layout are equally important.
Follow the steps listed in Section 8.2.2 to prevent potential EMI issues.
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11 Device and Documentation Support
11.1 Device Support
11.1.1 Development Support
11.1.1.1 Custom Design with WEBENCH Tools
Click here to create a custom design using the WEBENCH® Power Designer.
1. Start by entering your VIN, VOUT and IOUT requirements.
2. Optimize your design for key parameters like efficiency, footprint and cost using the optimizer dial and
compare this design with other possible solutions from Texas Instruments.
3. WEBENCH Power Designer provides you with a customized schematic along with a list of materials with real
time pricing and component availability.
4. In most cases, you will also be able to:
• Run electrical simulations to see important waveforms and circuit performance,
• Run thermal simulations to understand the thermal performance of your board,
• Export your customized schematic and layout into popular CAD formats,
• Print PDF reports for the design, and share your design with colleagues.
11.2 Support Resources
TI E2E™ support forums are an engineer's go-to source for fast, verified answers and design help — straight
from the experts. Search existing answers or ask your own question to get the quick design help you need.
Linked content is provided "AS IS" by the respective contributors. They do not constitute TI specifications and do
not necessarily reflect TI's views; see TI's Terms of Use.
11.3 Receiving Notification of Documentation Updates
To receive notification of documentation updates, navigate to the device product folder on ti.com. Click on
Subscribe to updates to register and receive a weekly digest of any product information that has changed. For
change details, review the revision history included in any revised document.
11.4 Trademarks
PowerPAD™ and TI E2E™ are trademarks of Texas Instruments.
WEBENCH® is a registered trademark of Texas Instruments.
All trademarks are the property of their respective owners.
11.5 Electrostatic Discharge Caution
This integrated circuit can be damaged by ESD. Texas Instruments recommends that all integrated circuits be handled
with appropriate precautions. Failure to observe proper handling and installation procedures can cause damage.
ESD damage can range from subtle performance degradation to complete device failure. Precision integrated circuits may
be more susceptible to damage because very small parametric changes could cause the device not to meet its published
specifications.
11.6 Glossary
TI Glossary
This glossary lists and explains terms, acronyms, and definitions.
12 Mechanical, Packaging, and Orderable Information
The following pages include mechanical, packaging, and orderable information. This information is the most
current data available for the designated devices. This data is subject to change without notice and revision of
this document. For browser-based versions of this data sheet, refer to the left-hand navigation.
28
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PACKAGE OPTION ADDENDUM
www.ti.com
10-Dec-2020
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
(4/5)
(6)
TPS54331D
ACTIVE
SOIC
D
8
75
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 150
54331
TPS54331DDA
ACTIVE SO PowerPAD
DDA
8
75
RoHS & Green
NIPDAU
Level-2-260C-1 YEAR
-40 to 150
54331
TPS54331DDAR
ACTIVE SO PowerPAD
DDA
8
2500
RoHS & Green
NIPDAU
Level-2-260C-1 YEAR
-40 to 150
54331
TPS54331DG4
ACTIVE
SOIC
D
8
75
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 150
54331
TPS54331DR
ACTIVE
SOIC
D
8
2500
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 150
54331
TPS54331DRG4
ACTIVE
SOIC
D
8
2500
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 150
54331
TPS54331GDR
ACTIVE
SOIC
D
8
2500
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 150
54331
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of
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