TPS54362-HT
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SLVSBI9 – SEPTEMBER 2012
1-A, 48-V STEP DOWN DC/DC CONVERTER WITH LOW Iq
Check for Samples: TPS54362-HT
FEATURES
1
•
2
•
•
•
•
•
•
•
•
•
•
•
•
•
•
•
Withstands Transients up to 60 V With an
Operating Range of 3.6 V to 48 V
Asynchronous Switch Mode Regulator With
External Components (L and C), Load Current
up to 1 A (Maximum)
0.8 V ±1.5% Voltage Reference
200-kHz to 2.2-MHz Switching Frequency
High Voltage Tolerant Enable Input for ON/OFF
State
Soft Start on Enable Cycle
Slew Rate Control on Internal Power Switch
External Clock Input for Synchronization
Pulse Skip Mode (PFM) During Light Output
Loads With Quiescent Current = 65 μA Typical
(LPM Operation)
External Compensation for Wide Bandwidth
Error Amplifier
Internal Undervoltage Lock Out UVLO
Programmable Reset Power on Delay
Reset Function Filter Time for Fast Negative
Transients
Programmable Overvoltage Output Monitoring
Programmable Undervoltage Output
Monitoring, Issuance of Reset if Output Falls
Below Threshold
Switch Current Limit Protection
•
•
Short Circuit and Overcurrent Protection of
FET
Package: 20-pin HTSSOP PowerPAD™
APPLICATIONS
•
•
Down-hole Drilling
High Temperature Environments
SUPPORTS EXTREME TEMPERATURE
APPLICATIONS
•
•
•
•
•
•
•
•
Controlled Baseline
One Assembly and Test Site
One Fabrication Site
Available in Extreme (–55°C to 175°C)
Temperature Range
Extended Product Life Cycle
Extended Product-Change Notification
Product Traceability
Texas Instruments' high temperature products
utilize highly optimized silicon (die) solutions
with design and process enhancements to
maximize performance over extended
temperatures.
DESCRIPTION
The TPS54362 is a step down switch mode power supply with voltage supervisor. Integrated input voltage line
feed forward topology improves line transient regulation of the voltage mode buck regulator. The regulator has a
cycle-by cycle current limit. A pulse skip mode operation under no load reduces the supply current to 65 μA.
Using the enable pin, the supply shutdown current is reduced to 1 μA.
An open drain reset signal indicates when the nominal output drops below the threshold set by an external
resistor divider network. The output voltage start up ramp is controlled by a soft start capacitor. There is an
internal undervoltage shut down which is activated when the input supply ramps down to 2.6 V.
1
2
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
PowerPAD is a trademark of Texas Instruments.
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of the Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
Copyright © 2012, Texas Instruments Incorporated
TPS54362-HT
SLVSBI9 – SEPTEMBER 2012
VIN
LPM
Supply
VIN
www.ti.com
95%
90%
Rslew
Rslew = 5kW
EN
SS
90%
85%
VIN = 6 V
SYNC
TPS54362
80%
VReg
BOOT
PH
Vreg
Rslew = 35kW
75%
VIN = 12 V,
VReg = 5 V,
Fsw = 500 kHz,
L = 22 mH
CO = 100 mF,
TA = 25oC
70%
Cdly
VSENSE
COMP
RST_TH
GND
65%
2
80%
VIN = 20 V
VReg = 5 V,
Fsw = 500 kHz,
L = 22 mH
CO = 100 mF
75%
70%
Rslew = 5 kW,
o
TA = 25 C
65%
60%
60%
0
OV_TH
(1)
Efficency (%)
RT
Efficency (%)
nRST
85%
0.5
1
1.5
2
IL - Load Current - A
2.5
3
0
0.5
1
1.5
2
2.5
3
IL - Load Current - A
Load Current is limited to 1 A for 175°C applications.
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This integrated circuit can be damaged by ESD. Texas Instruments recommends that all integrated circuits be handled with
appropriate precautions. Failure to observe proper handling and installation procedures can cause damage.
ESD damage can range from subtle performance degradation to complete device failure. Precision integrated circuits may be more
susceptible to damage because very small parametric changes could cause the device not to meet its published specifications.
ORDERING INFORMATION (1)
(1)
TJ
PACKAGE
PART NUMBER
TOP-SIDE MARKING
–55°C to 175°C
PWP
TPS54362HPWP
54632H
For the most current package and ordering information, see the Package Option Addendum at the end of this document, or see the TI
web site at www.ti.com.
ABSOLUTE MAXIMUM RATINGS (1)
Input voltage
Output voltage
VALUE
UNIT
EN
–0.3 to 60
V
VIN
–0.3 to 60
VReg
–0.3 to 20
LPM
–0.3 to 5.5
OV_TH
–0.3 to 5.5
RST_TH
–0.3 to 5.5
SYNC
–0.3 to 5.5
VSENSE
–0.3 to 5.5
BOOT
–0.3 to 65
PH
V
–0.3 to 60
-2 for 30 ns
-1 for 200 ns
-0.85 at TA= -55°C
-0.5 at TA= 175°C
Temperature
Electrostatic discharge HBM
(1)
(2)
RT
–0.3 to 5.5
RST
–0.3 to 5.5
Cdly
–0.3 to 8
SS
–0.3 to 8
COMP
–0.3 to 7
Operating virtual junction temperature range, TJ
–55 to 185
Storage temperature range, TS
–55 to 185
°C
2
kV
(2)
°C
Stresses beyond those listed under absolute maximum ratings may cause permanent damage to the device. These are stress ratings
only and functional operation of the device at these or any other conditions beyond those indicated under recommended operating
conditions is not implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability. All voltages
are with respect to ground.
The human body model is a 100pF capacitor discharged through a 1.5kΩ resistor into each pin
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RECOMMENDED OPERATING CONDITIONS
MIN
MAX
VI
Unregulated Buck supply input voltage (VIN, EN)
3.6
48
V
VReg
Regulator voltage range
0.9
18
V
VReg
Power up in Low Power Mode (LPM) or Discontinuous Mode (DCM)
0.9
5.5
V
Bootstrap Capacitor (BOOT)
3.6
56
V
Switched outputs (PH)
3.6
48
V
0
5.25
V
Logic level inputs (RST, VSENSE, OV_TH, RST_TH, Rslew, SYNC, RT)
Logic level inputs (SS, Cdly, COMP)
Operating junction temperature range (1)
TJ
(1)
UNIT
0
6.5
V
–55
175
°C
This assumes TA = TJ – Power dissipation × θJA (Junction to Ambient).
THERMAL INFORMATION
TPS54362
THERMAL METRIC (1)
PWP
UNITS
20 PINS
θJA
Junction-to-ambient thermal resistance (2)
37.5
θJCtop
Junction-to-case (top) thermal resistance (3)
21.4
(4)
θJB
Junction-to-board thermal resistance
ψJT
Junction-to-top characterization parameter (5)
ψJB
Junction-to-board characterization parameter (6)
θJCbot
(1)
(2)
(3)
(4)
(5)
(6)
(7)
4
Junction-to-case (bottom) thermal resistance
18.5
0.5
(7)
°C/W
18.2
1.1
For more information about traditional and new thermal metrics, see the IC Package Thermal Metrics application report, SPRA953.
The junction-to-ambient thermal resistance under natural convection is obtained in a simulation on a JEDEC-standard, high-K board, as
specified in JESD51-7, in an environment described in JESD51-2a.
The junction-to-case (top) thermal resistance is obtained by simulating a cold plate test on the package top. No specific JEDECstandard test exists, but a close description can be found in the ANSI SEMI standard G30-88.
The junction-to-board thermal resistance is obtained by simulating in an environment with a ring cold plate fixture to control the PCB
temperature, as described in JESD51-8.
The junction-to-top characterization parameter, ψJT, estimates the junction temperature of a device in a real system and is extracted
from the simulation data for obtaining θJA, using a procedure described in JESD51-2a (sections 6 and 7).
The junction-to-board characterization parameter, ψJB, estimates the junction temperature of a device in a real system and is extracted
from the simulation data for obtaining θJA , using a procedure described in JESD51-2a (sections 6 and 7).
The junction-to-case (bottom) thermal resistance is obtained by simulating a cold plate test on the exposed (power) pad. No specific
JEDEC standard test exists, but a close description can be found in the ANSI SEMI standard G30-88.
Spacer
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SLVSBI9 – SEPTEMBER 2012
DC ELECTRICAL CHARACTERISTICS
VIN = 7 V to 48 V, EN = VIN, TJ = –55°C to 175°C (unless otherwise noted)
PARAMETER
TEST CONDITIONS
TJ = –40°C to 125°C
TJ = –55°C to 175°C
MIN
MAX
MIN
48
3.6
TYP
TYP
UNIT
MAX
INPUT POWER SUPPLY
Normal mode–Buck mode after initial start up
3.6
48
Low power mode:
VIN
Supply voltage on VIN line
Falling threshold (LPM disabled)
8
8
Rising threshold (LPM activated)
8.5
8.5
High voltage threshold (LPM disabled)
Iq-Normal
Quiescent current normal
mode
Open loop test – max duty cycle
VIN = 7 V to 48 V
Iq-LPM
Quiescent current; low
power mode
ILOAD < 1 mA, VIN = 12 V
ISD
Shutdown
25
27
30
5
65
ILOAD < 1 mA, VIN = 24 V
25
V
27
30
10
5
10
mA
75
65
88
μA
98
μA
85
EN = 0 V, device is OFF, VIN = 24 V
5.5
10
8.5
16
EN = 0 V, device is OFF, VIN = 12 V
1
4
2.5
7.5
μA
TRANSITION TIMES (LOW POWER – NORMAL MODES) (1)
td1
Transition delay between
normal mode to low power
mode
VIN = 12 V, VReg = 5 V, ILOAD = 1 A to 1 mA
100
100
μs
td2
Transition delay between
low power mode to normal
mode
VIN = 12 V, VReg = 5 V ILOAD = 1 mA to 1 A
5
5
μs
SWITCH MODE SUPPLY; VReg
VReg
Regulator output
VSENSE = 0.8 V ref
VSENSE
Feedback voltage
VReg = 0.9 V to 18 V, VIN = 7 V to 48 V
RDS(on)
Internal switch resistance
Measured across VIN and PH, ILoad = 500 mA
ICL
Switch current limit cycle by
cycle
VIN = 12 V
tON-Min
Duty cycle pulse width
0.9
0.788
Switch mode frequency
4
Set using external resistor on RT pin
Internal oscillator frequency
ISink
Start up condition
OV_TH = 0 V, VReg = 10 V
ILimit
Prevent overshoot
0 V < OV_TH < 0.8 V, VReg = 10 V
(1)
18
0.9
0.812
0.770
18
0.8
500
tOFF-Min
fSW
0.8
6
8
0.830
600
4
6
100
100
200
200
8
V
V
mΩ
A
ns
0.2
2.2
0.2
2.2
–10
10
–22
22
MHz
%
1
1
mA
80
80
mA
This test is for characterization only
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DC ELECTRICAL CHARACTERISTICS
VIN = 7 V to 48 V, EN = VIN, TJ = –55°C to 175°C (unless otherwise noted)
PARAMETER
TEST CONDITIONS
TJ = –40°C to 125°C
TJ = –55°C to 175°C
MIN
MIN
TYP
MAX
TYP
UNIT
MAX
ENABLE (EN)
VIL
Low input threshold
VIH
High input threshold
Ilkg
Leakage into EN terminal
0.7
1.7
0.7
V
1.7
V
EN = 60 V
100
135
110
200
μA
EN = 12 V
8
15
10
20
μA
2
2.6
2
2.8
μA
RESET DELAY (Cdly)
IO
External capacitor charge
current
EN = high
VThreshold
Switching threshold
Output voltage in regulation
VIL
Low input threshold
VIN = 12 V
VIH
High input threshold
VIN = 12 V
Ilkg
Leakage into LPM terminal LPM = 5 V
1.4
1.3
2
2
0.7
1.7
V
0.7
V
95
μA
ms
1.7
65
95
V
65
RESET OUTPUT (RST)
trdly
POR delay timer
Based on Cdly capacitor, Cdly = 4.7 nF
RST_TH
Reset threshold for VReg
Check RST output
tnRSTdly
Filter time
Delay before RST is asserted low
ISS
Soft-start source current
3.6
7
3.3
8
0.768
0.832
0.768
0.832
V
10
20
35
9.3
20
35
μs
40
50
60
40
50
60
μA
SYNCHRONIZATION (SYNC) (1)
VIL
VSYNC
0.7
VIH
Ilkg
SYNC
SYNCtrans
SYNCCLK
1.7
0.7
1.7
Leakage
SYNC = 5 V
Input clock
VIN = 12 V, VReg = 5 V,
fSW < fext < 2 × fSW
65
External clock to internal
clock
No external clock, VIN = 12 V, VReg = 5 V
32
32
Internal clock to external
clock
External clock = 1 MHz, VIN = 12 V,
VReg = 5 V
2.5
2.5
Min duty cycle
180
95
2200
65
180
V
95
μA
2200
kHz
μs
30
30
Max duty cycle
70
70
%
Rslew
IRslew
Rslew = 50 kΩ
20
20
Rslew = 10 kΩ
100
100
μA
OVERVOLTAGE SUPERVISORS (OV_TH)
OV_TH
(1)
6
Threshold for VReg during
overvoltage
Internal switch is turned OFF
VReg = 5 V
Internal pull down on VReg, with
OV_TH = 1 V
0.768
0.832
70
0.768
0.832
70
V
mA
The SYNC input clock can have a maximum frequency of 2x the programmed clock frequency up to a maximum value of 1.1MHz.
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10000000
Life (Hrs)
1000000
100000
10000
1000
120
140
160
180
200
Operating Junction Temperature ( °C)
(1)
See data sheet for absolute maximum and minimum recommended operating conditions.
(2)
Silicon operating life design goal is 10 years at 105°C junction temperature (does not include package interconnect
life).
(3)
The predicted operating lifetime vs. junction temperature is based on reliability modeling using electromigration as the
dominant failure mechanism affecting device wearout for the specific device process and design characteristics.
(4)
This device is qualified for 1000 hours of continuous operation at maximum rated temperature.
Figure 1. Electromigration Fail Mode Derating Chart
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DEVICE INFORMATION
PWP 20-PIN PACKAGE
TOP VIEW
NC
NC
SYNC
LPM
EN
RT
Rslew
RST
Cdly
GND
1
20
2
19
3
18
4
17
5
16
6
15
7
14
8
13
9
12
10
11
BOOT
VIN
VIN
PH
VReg
COMP
VSENSE
RST_TH
OV_TH
SS
PIN FUNCTIONS
PIN
NAME
NO.
I/O
DESCRIPTION
NC
1
NC
Connect to ground
NC
2
NC
Connect to ground
SYNC
3
I
External synchronization clock input to override the internal oscillator clock. An internal pull down
resistor of 62kΩ (typical) is connected to the ground.
LPM
4
I
Low-power mode control using digital input signal. An internal pull down resistor of 62kΩ (typical) is
connected to the ground.
EN
5
I
Enable pin, internally pulled up. Must be externally pulled up or down to enable/disable the device.
RT
6
O
External resistor to ground to program the internal oscillator frequency.
Rslew
7
O
External resistor to ground to control the slew rate of internal switching FET.
RST
8
O
Active low, open drain reset output connected to external bias voltage through a resistor, asserted
high after the device starts regulating.
Cdly
9
O
External capacitor to ground to program power on reset delay.
GND
10
O
Ground pin, must be electrically connected to the exposed pad on the PCB for proper thermal
performance.
SS
11
O
External capacitor to ground to program soft start time.
OV_TH
12
I
Sense input for overvoltage detection on regulated output, an external resisitor network is connected
between VReg and ground to program the overvoltage threshold.
RST_TH
13
I
Sense input for overvoltage detection on regulated output, an external resisitor network is connected
between VReg and ground to program the reset and undervoltage threshold.
VSENSE
14
I
Inverting node of error amplifier for voltage mode control
COMP
15
O
Error amplifier output to connect external compensation components.
VReg
16
I
Internal low-side FET to load output during startup or limit overshoot.
PH
17
O
Source of the internal switching FET
VIN
18
I
Unregulated input voltage. Pin 18 and pin 19 must be connected externally.
VIN
19
I
Unregulated input voltage. Pin 18 and pin 19 must be connected externally.
BOOT
20
O
External bootstrap capacitor to PH to drive the gate of the internal switching FET.
8
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FUNCTIONAL BLOCK DIAGRAM
20
Bandgap
ref
LPM 4
D1
VIN
Vsupply
18
R11
C1
7
R7
Internal
supply
19
16
Gate Drive with
Over-Current Limit
for Internal Switch
5
R10
Rslew
0.8 V ref
0.2 V ref
Internal
Voltage
Rail
EN
BOOT
VIN
VReg
C3
L
PH
Vreg
17
RT
R8
6
Selectable
Oscillator
Thermal
Sensor
D2
Error
amp
SYNC 3
Cdly
14
+
0.8 V ref
9
C2
R9
-
11
15
R12
RST
+ 0.8 V ref
-
8
Reset with
Delay Timer
GND
10
Voltage
comp
+
-
0.82 V ref
R4
VSENSE
SS
C6
Vreg
C4
C7
ref
R5
C8
COMP
RST_ TH
C5
R6
R1
13
+ 0.8 V ref
12
OV _ TH
C10
R2
R3
C9
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TYPICAL CHARACTERISTICS
Efficiency Data of Power Supply
FET SWITCHING (SLOW SLEW RATE)
FAST SLEW RATE ON SWITCHING FET
90%
95%
Rslew = 5kW
90%
85%
85%
Efficency (%)
80%
Efficency (%)
VIN = 6 V
Rslew = 35kW
75%
VIN = 12 V,
VReg = 5 V,
Fsw = 500 kHz,
L = 22 mH
CO = 100 mF,
o
TA = 25 C
70%
65%
0.5
1
1.5
2
VIN = 20 V
75%
VReg = 5 V,
Fsw = 500 kHz,
L = 22 mH
CO = 100 mF
70%
65%
60%
0
80%
2.5
Rslew = 5 kW,
TA = 25oC
60%
3
0
0.5
IL - Load Current - A
1
1.5
2
2.5
3
IL - Load Current - A
Figure 2.
Figure 3.
Output Voltage Drop Out
LOAD CURRENT > 100 mA
LOAD CURRENT < 100 mA
7
7
6.5
No Load
10 mA
3A
5.5
VI - Intput Voltage - V
VI - Intput Voltage - V
6
5
4.5
200 mA
4
536 mA
3.5
5
50 mA
4
100 mA
1.4 A
3
3
2.5
Input Voltage Ramp Down Tracking
(Load at 5 V Output and fsw = 500 kHz)
Input Voltage Ramp Down Tracking
(Load at 5 V Output and fsw = 500 kHz)
2
0
1
5
4
2
3
VO - Output Voltage - V
(1) Load Current is limited to 1 A for 175°C applications.
Figure 4.
10
6
2
2.5
3
3.5
4
4.5
VO - Output Voltage - V
5
5.5
(1) Load Current is limited to 1 A for 175°C applications.
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TYPICAL CHARACTERISTICS (continued)
OUTPUT VOLTAGE TRACKING
8
VReg = 5 V at Free Air
7
Start (power up)
VI - Input Voltage - V
6
5
Tracking(power down)
4
3
2
1
0
0
0.05
0.1
0.15
IL - Load Current - A
0.2
Figure 6.
NOTE
Tracking: The input voltage at which the output voltage drops approximately -0.7 V of the
regulated voltage or for low input voltages (tracking function) over the load range.
Start: The input voltage required to achieve the 5V regulation on power up with the stated
load currents.
LPM, QUIESCENT CURRENT VARIATION
WITH TEMPERATURE
LPM, QUIESCENT CURRENT VARIATION
WITH TEMPERATURE
62
62
VI = 24 V
61
61
60
60
59
59
Quiescent Current (µA)
Quiescent Current (µA)
VI = 12 V
58
57
56
55
58
57
56
55
54
54
53
53
52
−55
2
60
118
Free-Air Temperature (°C)
175
52
−55
2
60
118
Free-Air Temperature (°C)
G006
Figure 7.
175
G007
Figure 8.
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TYPICAL CHARACTERISTICS (continued)
SHUT DOWN CURRENT
SHUT DOWN CURRENT
3
10
VI = 12 V
VI = 24 V
9
Shut Down Current (µA)
Shut Down Current (µA)
8
2
1
7
6
5
4
3
2
1
0
−55
2
60
118
Free-Air Temperature (°C)
0
−55
175
2
60
118
Free-Air Temperature (°C)
175
G008
G009
Figure 9.
Figure 10.
INTERNAL REFERENCE VOLTAGE
VOLTAGE DROP ON Rslew FOR CURRENT REFERENCE;
(SLEW RATE / Rslew)
1006
801
VI = 12 V
1004
Voltage Drop on Rslew - mV
Internal Reference Voltage (mV)
800
799
798
797
796
1000
998
996
795
794
−55
1002
2
60
118
Free-Air Temperature (°C)
175
994
-40
-20
G010
Figure 11.
12
0
20 40 60 80 100 120 140
TA - Free-Air Temperature - °C
Figure 12.
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TYPICAL CHARACTERISTICS (continued)
CURRENT CONSUMPTION WITH TEMPERATURE
5.6
EN = High,
VI = 12 V
IIN - Current Consumption - mA
5.55
5.5
5.45
5.4
5.35
5.3
5.25
-50
-30 -10 10 30 50 70 90 110 130 150
TA - Free-Air Temperature - °C
Figure 13.
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OVERVIEW
The TPS54362 is a 60V, 1A dc/dc step down (buck) converter using voltage-control mode scheme. The device
features supervisory function for power-on-rest during system power on. Once the output voltage has exceeded
the threshold set by RST_TH, a delay of 1ms/nF (based on capacitor value on RSTDLY terminal) is invoked
before RST line is released high. Conversely on power down, once the output voltage falls below the same set
threshold, the RST is pulled low only after a de-glitch filter of approximately 20μs (typ) expires. This is
implemented to prevent RST from being triggered due to fast transient line noise on the output supply.
An overvoltage monitor function, is used to limit output voltage to the threshold set by OV_TH. Both the RST_TH
and OV_TH monitoring voltages are set to be a pre-scale of the output voltage, and thresholds based on the
internal bias voltages of the voltage comparators (0.8V typical).
Detection of undervoltage on the output is based on the RST_TH setting and will invoke RST line to be asserted
low. Detection of over-voltage on the output is based on the OV_TH setting and will NOT invoke the RST line to
be asserted low. However, the internal switch is commanded to turn OFF.
In systems where power consumption is critical, low power mode is implemented to reduce the non-switching
quiescent current during light load conditions. The PFM operation is determined when the system enters
discontinuous current mode (DCM) for at least 100μs. The operation of when the device enters discontinuous
mode is dependent on the selection of external components.
DETAILED DESCRIPTION
The TPS54362 is a DC/DC Converter using a voltage-control mode scheme with an input voltage feed-forward
technique. The device can be programmed for a range of output voltages with a wide input voltage range. Below
are details with regard to the pin functionality.
INPUT VOLTAGE
The VIN pin is the input power source for the TPS54362. This pin must be externally protected against voltage
level greater than 60V and reverse battery. In Buck Mode the input current drawn from this pin is pulsed, with
fast rise and fall times. Therefore, this input line requires a filter capacitor to minimize noise. Additionally, for EMI
considerations, an input filter inductor may also be required.
FUNCTION MODE
FUNCTION
OPERATING VOLTAGE
RANGE
Buck
3.6V to 48V
OUTPUT CURRENT CAPABILITY
VReg = 0.9V to 18V and ILoad Up to 1A;
however, at higher output power the part
is derated for max temperature rating
COMMENTS
Optimum performance: VIN/VReg ratios
should always be set such that min required
duty cycle pulse (ton min) >150ns. The min off
time is 250ns for ALL conditions.
OUTPUT VOLTAGE VReg
The output voltage VReg is generated by the converter supplied from the battery voltage VIN and the external
components (L, C). The output is sensed through an external resistor divider and compared with an internal
reference voltage.
The value of the adjustable output voltage in Buck Mode is selectable between 0.9V and 18V by choosing the
external resistors, according to the relationship:
VReg = Vref (1 + R4/R5)
(1)
Where R5 and R4 are feedback resistors.
Vref = 0.8V (typical)
The internal reference voltage has a ±1.5% tolerance. The overall output voltage tolerance will be dependent on
the external feedback resistors. To determine the overall output voltage tolerance, use the following relationship:
tolVReg = tolVref + (R4/(R4 + R5)) × (tolR4 + tolR5)
(2)
Where R4 and R5 are feedback resistors.
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Vref = 0.8V (typical)
The VReg pin is also internally connected to a load of 100Ω, which is turned ON in the following conditions:
• During startup conditions, when the device is powered up with no-load, or whenever EN is toggled, the
internal load connected to VReg pin is turned ON for about 100 µs to charge the bootstrap capacitor to
provide gate drive voltage to the switching transistor.
• During normal operating conditions, when the regulated output voltage exceeds the overvoltage threshold
(preset by external resisitors R1, R2, and R3), the internal load is turned ON, and this pin is pulled down to
bring the regulated output voltage down.
Typically an output capacitor within the range of 10-400μF is used. This terminal will have a filter capacitor with
low ESR characteristics in order to minimize output ripple voltage.
OSCILLATOR FREQUENCY: (RT)
Oscillator frequency is selectable by means of a resistor placed at the RT pin. The switching frequency (Fsw) can
be set in the range 200 kHz – 2200 kHz. In addition, the switching frequency can be imposed externally by a
clock signal (Fext) at the SYNC pin with Fsw < Fext< 2×Fsw. In this case the external clock overrides the
switching frequency determined by the RT pin and the internal oscillator is clocked by the external
synchronization clock.
2000
fsw - Switching Frequency - kHz
1800
1600
1400
1200
40 V
1000
24 V
800
8V
14 V
600
400
200
0
100
200
300
Resistor on RT - kW
400
500
600
Figure 14. Switching Frequency vs Resistor Value
SYNCHRONIZATION (SYNC)
This is an external input signal to synchronize the switching frequency using an external clock signal. The
synchronization input will over-ride the internally fixed oscillator signal. The synchronization signal has to be valid
for approximately 2 clock cycles (pulses) before the transition is made for synchronization with the external
frequency input. If the external clock input does NOT transition low or high for 32μS (typ), the system will default
to the internal clock set by the RT pin. The SYNC input clock can have a maximum frequency of 2X the
programmed clock frequency up to a maximum value of 2.2MHz
ENABLE / SHUTDOWN:(EN)
The Enable pin provides electrical on/off control of the regulator. Once the Enable pin voltage exceeds the
threshold voltage, the regulator starts operation and the internal soft start begins to ramp. If the Enable pin
voltage is pulled below the threshold voltage, the regulator stops switching and the internal soft start resets.
Connecting the pin to ground or to any voltage less than 0.7V disables the regulator and activate the shutdown
mode. This pin must have an external pull up or pull down to change the state of the device.
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RESET DELAY (Cdly)
The Reset delay pin sets the desired delay time to assert the RESET pin high after the supply has exceeded the
programmed VReg_RST voltage. The delay may be programmed in the range of 2.2ms to 200ms using
capacitors in the range of 2.2nF to 200nF. The delay time is calculated using the following equation:
PORdly = 1ms / nF × C, Where C = capacitor on Cdly pin
(3)
RESET PIN (nRST)
The RESET pin is an open-drain output. The power-on reset output is asserted low until the output voltage
exceeds the programmed VReg_RST voltage threshold and the reset delay timer has expired. Additionally,
whenever the ENABLE pin is low or open, RESET is immediately asserted low regardless of the output voltage.
There is a reset filter timer to prevent reset being invoked due to short negative transients on the output line.
Power On Condition/ Reset Line
Power Down Condition/ Reset Line
VIN
VIN
Css
Css
VReg
0.92 x VReg
VReg
Set by RST_TH
Terminal
Cdly
Cdly
tdelay
RST
RST
20 ms
(Typ-Deglitch Time)
BOOST CAPACITOR (BOOT)
This capacitor provides the gate drive voltage for the Internal MOSFET switch. X7R or X5R grade dielectrics are
recommended due to their stable values over temperature. Boost cap may need to be tweaked lower for low
Vreg and/or high frequencies applications. The cap may need to be tweaked higher for high Vreg and/or low
frequencies applications. (e.g. 100nF for 500kHz/5V and 220n for 500kHz/8V.)
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SOFT START (SS)
On
1.
2.
3.
4.
powerup or after a short circuit event , the following conditions are recommended:
VIN – VReg > 2.5V
Load current < 1A, until RST goes high.
In discontinuous mode or LPM (i.e., light loads), in addition to 1), Vreg < 5.5V also applies.
Equation 4 should be satisfied. This condition also applies when there is a short circuit on the output.
1.55 × Css
50 × 10
-6
<
30 × 10-6
×
D × ILOAD
CO
L
(4)
Where:
D = VO/VIN duty cycle.
(5)
Css = 1 nF to 220 nF, providing the above equations are satisfied.
L is inductance of inductor.
LOW POWER MODE (LPM)
The TPS54362 enters automatically low power mode once the regulation goes into discontinuous mode. The
internal control circuitry for any transition from Low Power Mode to High Power Mode occurs within 5μs (typ). In
low power mode, the converter operates as a hysteretic controller with the threshold limits set by VReg_UV =
0.82 x (R1 + R2 +R3 / (R2 + R3), for the lower limit and ~VReg for the upper limit. To ensure tight regulation in
the low power mode, R2 and R3 values are set accordingly.
The device operates with both automatic and digital controlled low power mode. The digital low power mode can
over-ride the automatic low power mode function by applying the appropriate signal on the LPM terminal. The
part goes into active or normal mode for at least 100μs, whenever RST_TH or VREG_UV is tripped. In active
mode or normal mode, ALL blocks including OV function are enabled.
In LPM mode, OV function is disabled.
Active or Normal Mode: When part is in DCM with LPM=High or in CCM with LPM=High or Low
LPM: When part is in DCM with LPM = Low
Automatic and Digital
LPM high:
device forced normal mode, fixed frequency, even at light load current (part will do
pulse skipping to keep output voltage in regulation at light loads)
LPM low or open:
device will automatically change between normal and low power mode dependent on
load current
BUCK MODE LOW POWER MODE OPERATION
When operating in low power mode (Buck reg), and if the output is shorted to ground, a reset is asserted.
The low power mode operation is initiated once the converter enters discontinuous mode of operation.
EXTERNAL LPM OPERATION
The low power mode (LPM) is active low, if there is an open on this terminal the IC enters the low power mode
(internal pull down).
To allow low power mode operation, the load current has to be low and the LPM terminal is set to ground.
To inhibit low power mode, the microcontroller has to drive the terminal high, and the converter is not in
discontinuous mode of operation.
Part can ONLY power-up in LPM/DCM if, VReg < 5.5V AND VIN-VReg > 2.5V.
In active mode. the part powers-up when VIN > 3.6V (min).
Note: In LPM, the OV_TH circuit is not enabled.
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Active or Normal Mode: When the device is in CCM or DCM with LPM = High
LPM: When the device is in DCM with LPM = Low
SHORT CIRCUIT PROTECTION
The TPS54362 features an output short-circuit protection. Short-circuit conditions are detected by monitoring the
RST_TH, and when the voltage on this node drops below 0.2V, the switching frequency is decreased and current
limit is folded back to protect the device. The switching frequency is fold back to approximately 25kHz and the
current limit is reduced to 30% of the current limit typical value.
OVERCURRENT PROTECTION
Overcurrent protection is implemented by sensing the current through the NMOS switch FET. The sensed
current is then compared to a current reference level representing the overcurrent threshold limit. If the sensed
current exceeds the overcurrent threshold limit, the overcurrent indicator is set true. The system will ignore the
overcurrent indicator for the leading edge blanking time at the beginning of each cycle to avoid any turn-on noise
glitches.
Once overcurrent indicator is set true, overcurrent protection is triggered. The MOSFET is turned off for the rest
of the cycle after a propagation delay. The overcurrent protection scheme is called cycle-by-cycle current limiting.
If the sensed current continues to increase during cycle-by-cycle current limiting, the temperature of part will start
rising, the TSD will kick in and shut down switching until part cools down.
SLEW RATE CONTROL (Rslew)
This pin controls the switching slew rate of the internal power NMOS. The slew rate will be set by an external
resistor with a slew rate range shown below for rise and fall times. The range of rise time tr = 15ns to 35ns, and
fall time tf = 15ns to 200ns, with Rslew range of 10k to 50k (see plots below).
35
350
14 V
8V
30
40 V
300
tf - Fall Time - ns
tr - Rise Time - ns
24 V
25
20
15
40 V
10
250
24 V
200
14 V
150
8V
100
50
5
0
10
20
30
40
50
60
70
0
10
20
30
40
50
60
70
Rslew - Slew Resistor - kW
Rslew - Slew Resistor - kW
Figure 15. FET Rise Time
Figure 16. FET Fall Time
REGULATION VOLTAGE (VSENSE)
This pin is used to program the regulated output voltage based on a resistor feedback network monitoring the VO
output voltage. The selected ratio of R4 to R5 will set the VReg voltage.
RESET THRESHOLD (RST_TH)
This pin is programmable for setting the output accuracy for the low power mode (LPM) to set the undervoltage
monitoring of the regulated output voltage (VReg_UV), and the voltage to initiate a rest output signal
(VReg_RST). The resistor combination of R1 to R3 is used to program the threshold for detection of
undervoltage. Voltage bias on R2 + R3 sets the Reset threshold.
Undervoltage for transient and Low Power Mode Operation:
VReg_UV = 0.82V × (R1 + R2 +R3 / (R2 + R3)
Reset Threshold = VReg_RST = 0.8V × (R1 + R2 + R3 / (R2 + R3)
(6)
(7)
Recommended range: 70% to 92% of the regulation voltage.
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OVERVOLTAGE SUPERVISOR for VReg (OV_TH)
This pin is programmable to set the overvoltage monitoring of the regulated output voltage. The resistor
combination of R1 to R3 is used to program the threshold for detection of overvoltage. The bias voltage of R3
sets the OV threshold and the output voltage accuracy in hysteretic mode during transient events.
Overvoltage ref = VReg _OV = 0.8V x (R1 +R2 + R3) / (R3),
(8)
Recommended range: 106% to 110% of the regulation voltage
NOISE FILTER ON RST_TH AND OV_TH TERMINALS
There is some noise sensitivity on the RST_TH and OV_TH pins and capacitance is added to filter this noise.
The noise is more pronounced with fast falling edges on the PH pin. So the smaller the Rslew resistor (minimum
recommended value is 10kΩ) the more capacitance may be required on RST_TH and OV_TH. Users should use
the smallest capacitance necessary, because larger values will increase the loop response time and degrade
short circuit protection and transient response. The upper limit is determined by the 2μs maximum time constant
seen on the OVTH/RSTTH when VReg = 0 V (i.e. [R2 + R3]×[C9 + C10] < 2μs). The noise in the RST_TH /
OV_TH resistor chain may change with PCB layout or application set-up, so the RST_TH and/or the OVTH
capacitor may not be needed in all applications. Users can place the footprint and only populate it, if necessary.
Example
VReg
R1 = 36K
R2 = 600
R3 = 6.6k
RST _TH
C4
R1
VReg_RST = 0.8 × (43.2k) /7.2k) = 4.8V
R2
VReg_OV = 0.8 × (43.2k) /6.6k) = 5.24V
C10
OV_TH
R3
C9
Typical cap values for RST_TH/OV_TH caps are between 10 pf to 100 pf range for total resistance on
RSTH/OVTH divider of < 200 kΩ.
OUTPUT TOLERANCES BASED ON MODES OF OPERATION
VReg_OV
VReg (UL)
VReg Output
VReg (LL)
Load reg/Line reg in
Hysteretic mode
VReg_UV = 0.82 x (R1 + R2 + R3) / (R2 + R3)
VReg_RST = 0.8 x (R1 + R2 + R3) / (R2 + R3)
LPM
Active (Normal)
Modes Of Operation
empty paragraph for space between the illustration and table
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Mode Of Operation
VO - Lower Limit
VO – Upper limit
Comments
Hysteretic Mode
0.82V × (R1 + R2 + R3)/(R2 + R3)
0.8V × (R1+R2+R3)/(R3)
Min to max ripple on output
Low power Mode
0.82V × (R1 + R2 + R3)/(R2 + R3)
VReg + tolVReg
Min to max ripple on output
Active (Normal)
VReg – tolVReg
VReg + tolVReg
Min to max ripple on output
empty paragraph for space between the two tables
Supervisor Thresholds
VO - Typical value
Tolerance
Comments
Overvoltage
0.8V × (R1 + R2 + R3)/(R3)
± (tolVref + (R1 + R2/[R1 + R2 + R3]) ×
(tolR1 + tolR2 + tolR3)
Overvoltage threshold
setting
Reset
0.8V × (R1 + R2 + R3)/(R2 + R3)
± (tolVref + (R1/[R1 + R2 + R3]) ×
(tolR1 + tolR2 + tolR3)
Reset threshold setting
Load reg/Line reg in Hysteretic Mode
This mode of operation is when a load or line transient step occurs in the application. The converter will go into a
hysteretic mode of operation until the error amplifier stabilizes and controls the output regulation to a tighter
output tolerance. During the load step the regulator upper threshold is set by the VReg_OV and the lower
threshold is set by the VReg_UV limit.
The converter enters this mode of operation during load or line transient events if the main control loop cannot
respond to regulate within the specified tolerances. The regulator exits this mode once the main control loop
responds.
Internal Undervoltage Lock Out (UVLO)
The IC is enabled on power up once the internal bandgap and bias currents are stabile, this is typically at
VI = 3.4V (min). On power down, the internal circuitry is disabled at VI = 2.6V (max).
Loop Control Frequency Compensation
L
VReg
C8
C7
C ESR
R4
C4
R6
C5
R9
VSENSE
Error
AMP
R5
COMP
Vref = 0.8 V
Type 3
Compensation
Figure 17. Type 3 Compensation
Type III Compensation
fc = fsw × 0.1 (the cut off frequency, when the gain is 1 is called the unity gain frequency).
The fc is typically 1/5 to 1/10 of the switching frequency, double pole frequency response due to the LC output
filter
The modulator break frequencies as a function of the output LC filter is derived from Equation 9 and Equation 10.
The LC output filter gives a “Double Pole” which has a –180 degree phase shift
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fLC =
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1
1/2
2p (LCO )
(9)
The ESR of the output capacitor C gives a “ZERO” that has a 90 degree phase shift
1
fESR =
(2p CO ´ ESR)
(R4 + R5)
Vreg = Vref ´
R5
Vreg
(R4 + R5)
=
0.8V
R5
(10)
(11)
(12)
The VIN/Vr modulator gain is about 10 for 8V
x
(Farads)
8 ´ fsw
æ VO-Ripple ö
ç
÷
ç IRipple ÷
è
ø
(25)
The most critical condition based on the calculations above indicates that the output capacitance has to be a
minimum of 157μF to keep the output voltage in regulation during load transients.
Additional capacitance de-ratings for temperature, aging and dc bias has to be factored, and so a value of 220μF
with ESR calculated using Equation 26 of less than 100mΩ should be used on the output stage.
Maximum ESR of the out capacitor based on output ripple voltage specification.
VO-Ripple
RESR <
(Ohms)
IRipple
(26)
Output capacitor root mean square (RMS) ripple current. This is to prevent excess heating or failure due to high
ripple currents. This parameter is sometimes specified by the manufacturers.
(
VO x VI_max - VO
IO_RMS =
)
(Apms)
12 x VI_max x LO x fsw
(27)
FLYBACK SCHOTTKY DIODE
The TPS54362 requires an external Schottky diode connected between the PH and power ground termination.
The absolute voltage at PH pin should not go beyond the values mentioned in Absolute Maximum Ratings table
on page 2 of this document. The schottky diode conducts the output current during the off state of the internal
power switch. This schottky diode must have a reverse breakdown higher then the maximum input voltage of the
application. A schottky diode is selected for its lower forward voltage. The schottky diode is selected based on
the appropriate power rating, which factors in the DC conduction losses and the AC losses due to the high
switching frequencies; this is determined by Equation 28.
Pdiode
(
)
æ VI_max - VO x IO x V fd
= ç
ç
VI- max
è
ö
÷+
÷
ø
æ V- V
ç I fd
ç
ç
è
(
2
)
x fsw x CJ ö÷
÷ (Watts)
2
÷
ø
(28)
Where:
Vfd = forward conducting voltage of Schottky diode
Cj = junction capacitance of the Schottky diode
The recommended part numbers are PDS360 and SBR8U60P5.
INPUT CAPACITOR, CI
The requires an input ceramic de-coupling capacitor type X5R or X7R and bulk capacitance to minimize input
ripple voltage. The dc voltage rating of this input capacitance must be greater than the maximum input voltage.
The capacitor must have an input ripple current rating higher than the maximum input ripple current of the
converter for the application; this is determined by Equation 29.
The input capacitors for power regulators are chosen to have reasonable capacitance to volume ratio and fairly
stable over temperature. The value of the input capacitance also determines the input ripple voltage of the
regulator, shown by Equation 30.
II_RMS IO x
24
(
VI_min - VO
VO
x
VI_min
VI_min
)
(Amps)
(29)
I
x 0.25
DVI = O-max
(Volts)
CI x fsw
(30)
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OUTPUT VOLTAGE AND FEEDBACK RESISTOR SELECTION
In the design example, 187kΩ was selected for R4, using Equation 1, R4 is calculated as 59kΩ. To minimize the
effect of leakage current on the VSENSE terminal, the current flowing through the feedback network should be
greater than 5μA in order to maintain output accuracy. Higher resistor values help improve the converter
efficiency at low output currents, but may introduce noise immunity problems.
OVERVOLTAGE RESISTOR SELECTION
Using Equation 8, the value of R3 is determined to set the overvoltage threshold at 1.06 × 3.3V. The total resistor
network from VReg output to ground is approximately 100kΩ (this is R1 + R2 +R3). Then R3 is calculated to be
22.87kΩ. Use the nearest standard value, which is 22.6kΩ. A noise decoupling capacitor may be required on this
terminal to ensure proper operation; the value chosen for this design is 56pF.
RESET THRESHOLD RESISTOR SELECTION
Then using Equation 7 the value of R2 + R3 is calculated, and then knowing R3 from the OV_TH setting, R2 is
determined. The value of R2 + R3 yielded 26.35kΩ, which means R2 is approximately 3.48kΩ. This will set the
reset threshold at 0.92 × 3.3V. A noise decoupling capacitor may be required on this terminal to ensure proper
operation; the value chosen for this design is 15pF. R1 is determined to be 73.6kΩ.
LOW POWER MODE THRESHOLD
To obtain an approximation of the output load current at which the converter is operating in discontinuous mode,
use Equation 31. The values used in the equation for minimum and maximum input voltage will affect the duty
cycle and the overall discontinuous mode load current. With a maximum input voltage of 28V, the output load
current for DCM is 165.8mA, and for minimum input voltage of 8V the DCM mode load current is 111.7mA.
These are nominal values and other factors are not taken into consideration like external component variations
with temperature and aging.
(1 - D ) ´ VO
IL _DISCONT = IL_LPM =
(Amperes) (with ± 30% hysteresis)
2 ´ ¦ SW ´ L
(31)
UNDERVOLTAGE THRESHOLD FOR LOW POWER MODE AND LOAD TRANSIENT OPERATION
This threshold is set above the reset threshold to ensure the regulator operates within the specified tolerances
during output load transient of low load to high load and during discontinuous conduction mode. Using
Equation 6 the typical voltage threshold is determined.
In this design, the value for this threshold is 0.95 × 3.3V.
SOFTSTART CAPACITOR
The soft start capacitor determines the minimum time to reach the desired output voltage during a power up
cycle. This is useful when a load requires a controlled voltage slew rate, and helps to limit the current draw from
the input voltage supply line. Equation 4 has to be satisfied in addition to the other conditions stated in the soft
start section of this document. In this design, a 4.7nF capacitor is required to meet these criteria.
BOOTSTRAP CAPACITOR SELECTION
A 0.1μF ceramic capacitor must be connected between the PH and BOOT terminals for the converter to operate
and regulate the desired output voltage. It is recommended to use a capacitor with X5R or better grade dielectric
material, and the voltage rating on this capacitor of at least 25V to allow for derating.
COMPENSATION
Guidelines for Compensation Components
1
Make the two zeroes close to the double pole (LC), e.g fz1 ≈ fz2 ≈ 2 x p LCO
1. Make first zero below the filter double pole (approx 50% to 75% of fLC)
2. Make second zero at filter double pole (fLC)
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Make the two poles above the cross-over frequency fc,
1. Make first pole at the ESR frequency (fESR)
2. Make the second pole at 0.5 the switching frequency (0.5 × fsw)
Select R4 = 187kW
(R4 ´ 0.8 )
R5 =
(VO - 0.8)
(32)
fc ´ Vramp ´ R4
R6 =
(fLC ´ VI)
(33)
Calculate C5 based on placing a zero at 50% to 75% of the output filter double pole frequency.
1
C5 =
p ´ R6 ´ fLC
(34)
Calculate C8 by placing the first pole at the ESR zero frequency.
C5
C8 =
(2 p ´ R6 ´ C5 ´ fESR - 1)
(35)
Set the second pole at 0.5 the switching frequency and also set the second zero at the output filter double pole
frequency.
R4
R9 =
æ æ fsw ö
ö
çç ç
÷ - 1÷÷
è è 2 ´ fLC ø
ø
(36)
1
C7 =
p ´ R9 ´ fsw
(37)
Calculate the Loop Compensation
DC modulator gain (Amod) = 8 /Vr
Vr = 0.8
Amod (dB) = 20 log (10) = 20 dB
Output filter due to LCO poles and CO ESR zeros from Equation 9 and Equation 10.
fLC = 2.3 kHz for LCO = 22µH, CO = 220µF
fESR = 7.23 kHz for CO = 220µF, ESR = 100mΩ
Choose R4 = 187kΩ
26
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SLVSBI9 – SEPTEMBER 2012
The poles and zeros for a type III network are calculated using equations Equation 32 to Equation 37.
R5 = 59.8k (use standard value 59k)
R6 = 326.9k (use standard value 324k)
C5 = 425.5pF (use standard value 430pF)
C8 = 79.9pF (use standard value 43pF)
R9 = 2.16k (use standard value 2.15K)
C7 = 367.7pF (use standard value 360pF)
The poles and zeros based on these compensation values can be calculated using Equation 14 to Equation 17.
Power Dissipation
The power dissipation losses are applicable for continuous conduction mode operation (CCM)
æV ö
PCON = IO2 × RDS(on) ´ ç O ÷ (Conduction losses)
è VI ø
PSW = 1/2 ´ VI ´ IO ´ (tr + t f ) ´ fSW
(38)
(Switching losses)
(39)
-9
PGate = Vdrive ´ Qg ´ fsw (Gate drive losses) where Qg = 1 ´ 10
(nC)
(40)
PIC = VI ´ Iq-normal (Supply losses)
(41)
PTotal = PCON + PSW + PGate + PIC (Watts)
(42)
Where:
VO = Output voltage
VI = Input voltage
IO = Output current
tr = FET switching rise time (tr max = 40ns)
tf = FET switching fall time
Vdrive = FET gate drive voltage (typically Vdrive = 6V and Vdrive max = 8V)
fsw = Switching frequency
For given operating ambient temperature TA
TJ = TAmb + Rth ´ PTotal
(43)
For a given max junction temperature TJ-Max = 150°C
TAmb-Max = TJ-Max - Rth ´ PTotal
(44)
Where:
PTotal = Total power dissipation (Watts)
TAmb = Ambient Temperature in °C
TJ = Junction Temperature in °C
TAmb-Max = Maximum Ambient Temperature in °C
TJ-Max = Maximum junction temperature in °C
Rth = Thermal resistance of package in (°C/W)
Other factors NOT included in the information above which affect the overall efficiency and power losses are
Inductor ac and dc losses.
Trace resistance and losses associated with the copper trace routing connection
Flyback catch diode
The output current rating for the regulator may have to be derated for ambient temperatures above 85°C. The
de-rate value will depend on calculated worst case power dissipation and the thermal management
implementation in the application.
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Product Folder Links : TPS54362-HT
27
TPS54362-HT
SLVSBI9 – SEPTEMBER 2012
www.ti.com
LAYOUT
The following guidelines are recommended for PCB layout of the TPS54362 device.
INDUCTOR
Use a low EMI inductor with a ferrite type shielded core. Other types of inductors may be used; however, they
must be low EMI characteristics and located away from the low power traces and components in the circuit.
INPUT FILTER CAPACITORS
Input ceramic filter capacitors should be located in the close proximity of the VIN terminal. Surface mount
capacitors are recommended to minimize lead length and reduce noise coupling.
FEEDBACK
Route the feedback trace such that there is minimum interaction with any noise sources associated with the
switching components. Recommended practice is to ensure the inductor is placed away from the feedback trace
to prevent EMI noise source.
TRACES AND GROUND PLANE
All power (high current) traces should be thick and short as possible. The inductor and output capacitors should
be as close to each other as possible. This will reduce EMI radiated by the power traces due to high switching
currents.
In a two sided PCB, it is recommended to have ground planes on both sides of the PCB to help reduce noise
and ground loop errors. The ground connection for the input and output capacitors and IC ground should be
connected to this ground plane.
In a multilayer PCB, the ground plane is used to separate the power plane (high switching currents and
components are placed) from the signal plane (where the feedback trace and components are) for improved
performance.
Also, arrange the components such that the switching current loops curl in the same direction. Place the high
current components such that during conduction the current path is in the same direction. This will prevent
magnetic field reversal caused by the traces between the two half cycles, helping to reduce radiated EMI.
Output
Capacitor
Topside Supply Area
Input Capacitor
Ground
Plane
Catch Diode
NC
BOOT
NC
VIN
Output
Inductor
VIN
SYNC
LPM
PH
EN
VReg
RT
COMP
Rslew
VSENSE
RST
RST_TH
Cdly
OV_TH
GND
Compensation Network
Supervisor Network
SS
Resistor
Divider
Signal via to
Ground Plane
Topside Ground Area
Thermal Via
Signal Via
Figure 19. PCB Layout Example
28
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Product Folder Links : TPS54362-HT
PACKAGE OPTION ADDENDUM
www.ti.com
24-Feb-2022
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
(4/5)
(6)
TPS54362HPWP
ACTIVE
HTSSOP
PWP
20
70
RoHS & Green
NIPDAU
Level-4-260C-72 HR
-55 to 175
54362H
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of
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