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TPS54560B-Q1
SLVSDP8 – FEBRUARY 2017
TPS54560B-Q1 4.5 V to 60 V Input, 5 A, Step Down DC-DC Converter with Eco-mode™
1 Features
3 Description
•
The TPS54560B-Q1 is a 60 V, 5 A, step down
regulator with an integrated high side MOSFET. The
device survives load dump pulses up to 65V per ISO
7637. Current mode control provides simple external
compensation and flexible component selection. A
low ripple pulse skip mode reduces the no load
supply current to 146 μA. Shutdown supply current is
reduced to 2 μA when the enable pin is pulled low.
1
•
•
•
•
•
•
•
•
•
•
High Efficiency at Light Loads with Pulse Skipping
Eco-mode™
92-mΩ High-Side MOSFET
146 μA Operating Quiescent Current and
2 μA Shutdown Current
100 kHz to 2.5 MHz Fixed Switching Frequency
Synchronizes to External Clock
Low Dropout at Light Loads with Integrated BOOT
Recharge FET
Adjustable UVLO Voltage and Hysteresis
0.8 V 1% Internal Voltage Reference
8-Terminal HSOP with PowerPAD™ Package
–40°C to 150°C TJ Operating Range
Create a Custom Design using the
TPS54560B-Q1 with the WEBENCH® Power
Designer
Undervoltage lockout is internally set at 4.3 V but can
be increased using the enable pin. The output voltage
start up ramp is internally controlled to provide a
controlled start up and eliminate overshoot.
A wide switching frequency range allows either
efficiency or external component size to be optimized.
Output current is limited cycle-by-cycle. Frequency
foldback and thermal shutdown protects internal and
external components during an overload condition.
The TPS54560B-Q1 is available in an 8-terminal
thermally enhanced HSOP PowerPAD™ package.
2 Applications
•
•
•
•
Industrial Automation and Motor Control
Vehicle Accessories: GPS, Entertainment
USB Dedicated Charging Ports and Battery
Chargers
12 V, 24 V and 48 V Industrial, Automotive and
Communications Power Systems
Device Information(1)
ORDER NUMBER
PACKAGE
BODY SIZE
TPS54560B-Q1
HSOP (8)
4.89 mm x 3.9 mm
(1) For all available packages, see the orderable addendum at
the end of the data sheet.
spacer
Simplified Schematic
Efficiency vs Load Current
100
VIN
VIN
36 V to 12 V
BOOT
95
EN
SW
VOUT
COMP
Efficiency (%)
90
85
12 V to 3.3 V
80
12 V to 5 V
75
70
VOUT = 12 V, fsw = 800 kHz
VOUT = 5 V and 3.3 V, fsw = 400 kHz
65
RT/CLK
FB
60
0
0.5
1
1.5
2
2.5
3
3.5
IO - Output Current (A)
4
4.5
5
C024
GND
Copyright © 2016, Texas Instruments Incorporated
1
An IMPORTANT NOTICE at the end of this data sheet addresses availability, warranty, changes, use in safety-critical applications,
intellectual property matters and other important disclaimers. PRODUCTION DATA.
TPS54560B-Q1
SLVSDP8 – FEBRUARY 2017
www.ti.com
Table of Contents
1
2
3
4
5
6
7
Features ..................................................................
Applications ...........................................................
Description .............................................................
Revision History.....................................................
Pin Configuration and Functions .........................
Specifications.........................................................
1
1
1
2
3
4
6.1
6.2
6.3
6.4
6.5
6.6
6.7
4
4
4
4
5
6
6
Absolute Maximum Ratings ......................................
ESD Ratings..............................................................
Recommended Operating Conditions.......................
Thermal Information ..................................................
Electrical Characteristics...........................................
Timing Requirements ................................................
Typical Characteristics ..............................................
8
8.1
8.2
8.3
8.4
Overview .................................................................
Functional Block Diagram .......................................
Feature Description ................................................
Device Functional Modes........................................
Application Information............................................
Typical Application ..................................................
Inverting Power .......................................................
Split Rail Power Supply...........................................
23
23
36
36
9 Power Supply Recommendation ........................ 37
10 Layout................................................................... 38
10.1 Layout Guidelines ................................................. 38
10.2 Layout Examples................................................... 38
11 Device and Documentation Support ................. 39
11.1
11.2
11.3
11.4
11.5
11.6
Detailed Description ............................................ 10
7.1
7.2
7.3
7.4
Application and Implementation ........................ 23
10
11
11
22
Device Support......................................................
Receiving Notification of Documentation Updates
Community Resources..........................................
Trademarks ...........................................................
Electrostatic Discharge Caution ............................
Glossary ................................................................
39
39
39
39
39
39
12 Mechanical, Packaging, and Orderable
Information ........................................................... 39
4 Revision History
NOTE: Page numbers for previous revisions may differ from page numbers in the current version.
2
DATE
REVISION
INITIAL
February 2017
*
Initial release
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5 Pin Configuration and Functions
HSOP PACKAGE
(TOP VIEW)
BOOT
1
VIN
2
8
SW
7
GND
PowerPAD
EN
3
6
COMP
RT/CLK
4
5
FB
Pin Functions
PIN
NAME
NO.
I/O
DESCRIPTION
BOOT
1
O
A bootstrap capacitor is required between BOOT and SW. If the voltage on this capacitor is below the
minimum required to operate the high side MOSFET, the output is switched off until the capacitor is
refreshed.
VIN
2
I
Input supply voltage with 4.5 V to 60 V operating range.
EN
3
I
Enable terminal, with internal pull-up current source. Pull below 1.2 V to disable. Float to enable. Adjust the
input undervoltage lockout with two resistors. See the Enable and Adjusting Undervoltage Lockout section.
RT/CLK
4
I
Resistor Timing and External Clock. An internal amplifier holds this terminal at a fixed voltage when using an
external resistor to ground to set the switching frequency. If the terminal is pulled above the PLL upper
threshold, a mode change occurs and the terminal becomes a synchronization input. The internal amplifier is
disabled and the terminal is a high impedance clock input to the internal PLL. If clocking edges stop, the
internal amplifier is re-enabled and the operating mode returns to resistor frequency programming.
FB
5
I
Inverting input of the transconductance (gm) error amplifier.
COMP
6
O
Error amplifier output and input to the output switch current (PWM) comparator. Connect frequency
compensation components to this terminal.
GND
7
–
Ground
SW
8
I
The source of the internal high-side power MOSFET and switching node of the converter.
–
GND terminal must be electrically connected to the exposed pad on the printed circuit board for proper
operation.
Thermal Pad
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6 Specifications
6.1 Absolute Maximum Ratings (1)
over operating free-air temperature range (unless otherwise noted)
MIN
MAX
VIN
–0.3
65
EN
–0.3
8.4
BOOT
Input voltage
73
–0.3
COMP
–0.3
3
RT/CLK
–0.3
3.6
–0.6
65
3
–2
65
–40
150
°C
–65
150
°C
8
SW
SW, 10-ns Transient
Operating junction temperature
Storage temperature range
(1)
V
FB
BOOT-SW
Output voltage
UNIT
TSTG
V
Stresses beyond those listed under absolute maximum ratings may cause permanent damage to the device. These are stress ratings
only and functional operation of the device at these or any other conditions beyond those indicated under recommended operating
conditions is not implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
6.2 ESD Ratings
MIN
(1)
VESD
(1)
(2)
(3)
Human Body Model (HBM) ESD Stress Voltage
(2)
Charged Device Model (HBM) ESD Stress Voltage
(3)
MAX
UNIT
±2000
V
±500
V
Electrostatic discharge (ESD) to measure device sensitivity and immunity to damage caused by assembly line electrostatic discharges
into the device.
Level listed above is the passing level per ANSI/ESDA/JEDEC JS-001. JEDEC document JEP155 states that 500V HBM allows safe
manufacturing with a standard ESD control process. terminals listed as 1000V may actually have higher performance.
Level listed above is the passing level per EIA-JEDEC JESD22-C101. JEDEC document JEP157 states that 250V CDM allows safe
manufacturing with a standard ESD control process. terminals listed as 250V may actually have higher performance.
6.3 Recommended Operating Conditions
over operating free-air temperature range (unless otherwise noted)
MIN
MAX
VOUT + VDO (1)
60
V
Output voltage
0.8
58.8
V
IO
Output current
0
5
A
TJ
Junction Temperature
–40
150
°C
VIN
Supply input voltage
VO
(1)
UNIT
See
6.4 Thermal Information
THERMAL METRIC (1) (2)
TPS54560B-Q1
DDA (8 PINS)
UNIT
θJA
Junction-to-ambient thermal resistance (standard board)
42.0
°C/W
ψJT
Junction-to-top characterization parameter
5.9
°C/W
ψJB
Junction-to-board characterization parameter
23.4
°C/W
θJCtop
Junction-to-case(top) thermal resistance
45.8
°C/W
θJCbot
Junction-to-case(bottom) thermal resistance
3.6
°C/W
θJB
Junction-to-board thermal resistance
23.4
°C/W
(1)
(2)
4
For more information about traditional and new thermal metrics, see the Semiconductor and IC Package Thermal Metrics application
report.
Power rating at a specific ambient temperature TA should be determined with a junction temperature of 150°C. This is the point where
distortion starts to substantially increase. See power dissipation estimate in application section of this data sheet for more information.
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6.5 Electrical Characteristics
TJ = –40°C to 150°C, VIN = 4.5 V to 60 V (unless otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
60
V
4.3
4.48
V
SUPPLY VOLTAGE (VIN TERMINAL)
Operating input voltage
Internal undervoltage lockout threshold
4.5
Rising
4.1
Internal undervoltage lockout threshold
hysteresis
325
mV
Shutdown supply current
EN = 0 V, 25°C, 4.5 V ≤ VIN ≤ 60 V
2.25
4.5
Operating: nonswitching supply current
FB = 0.9 V, TA = 25°C
146
175
1.2
1.3
μA
ENABLE AND UVLO (EN TERMINAL)
Enable threshold voltage
Input current
No voltage hysteresis, rising and falling
1.1
Enable threshold +50 mV
Enable threshold –50 mV
Hysteresis current
–4.6
V
μA
–0.58
–1.2
-1.8
–2.2
–3.4
-4.5
μA
0.792
0.8
0.808
V
92
190
VOLTAGE REFERENCE
Voltage reference
HIGH-SIDE MOSFET
On-resistance
VIN = 12 V, BOOT-SW = 6 V
mΩ
ERROR AMPLIFIER
Input current
Error amplifier dc gain
VFB = 0.8 V
Min unity gain bandwidth
Error amplifier source/sink
V(COMP) = 1 V, 100 mV overdrive
COMP to SW current transconductance
50
nA
10,000
V/V
2500
kHz
±30
μA
17
A/V
CURRENT LIMIT
Current limit test
All VIN and temperatures, Open Loop (1)
6.3
7.9
9.5
All temperatures, VIN = 12 V, Open Loop (1)
6.3
7.9
9.5
VIN = 12 V, TA = 25°C, Open Loop (1)
7.0
7.9
8.8
A
THERMAL SHUTDOWN
Thermal shutdown
Thermal shutdown hysteresis
176
°C
12
°C
TIMING RESISTOR AND EXTERNAL CLOCK (RT/CLK TERMINAL)
Switching frequency range using RT mode
fSW
Switching frequency
100
RT = 200 kΩ
Switching frequency range using CLK mode
450
RT/CLK high threshold
1.55
RT/CLK low threshold
(1)
500
160
0.5
2500
kHz
550
kHz
2300
kHz
2
1.2
V
V
Open Loop current limit measured directly at the SW terminal and is independent of the inductor value and slope compensation.
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6.6 Timing Requirements
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
ENABLE AND UVLO (EN TERMINAL)
Enable to COMP active
VIN = 12 V, TA = 25°C
340
µs
INTERNAL SOFT-START TIME
Soft-Start Time
fSW = 500 kHz, 10% to 90%
2.1
ms
Soft-Start Time
fSW = 2.5 MHz, 10% to 90%
0.42
ms
–2 μA < ICOMP < 2 μA, VCOMP = 1 V
350
μs
77
μs
60
ns
15
ns
ERROR AMPLIFIER
Error amplifier transconductance (gM)
Error amplifier transconductance (gM) during
–2 μA < ICOMP < 2 μA, VCOMP = 1 V, VFB = 0.4 V
soft-start
CURRENT LIMIT
Current limit threshold delay
TIMING RESISTOR AND EXTERNAL CLOCK (RT/CLK TERMINAL)
Minimum CLK input pulse width
RT/CLK falling edge to SW rising edge
delay
Measured at 500 kHz with RT resistor in series
55
ns
PLL lock in time
Measured at 500 kHz
78
μs
6.7 Typical Characteristics
0.814
VFB - Voltage Referance ( V)
RDSON - On-State Resistance ( )
0.25
0.2
0.15
0.1
0.05
BOOT-SW = 3 V
0.809
0.804
0.799
0.794
0.789
BOOT-SW = 6 V
0
0.784
±50
±25
0
25
50
75
100
125
±50
150
TJ - Junction Temperature (ƒC)
0
25
50
75
100
125
150
TJ - Junction Temperature (ƒC)
VIN = 12 V
C026
VIN = 12 V
Figure 1. On Resistance vs Junction Temperature
Figure 2. Voltage Reference vs Junction Temperature
9.5
9
4.5
12
60
9
High Side Switch Current (A)
High Side Switch Current (A)
±25
C025
8.5
8
7.5
7
6.5
6
-40
8.5
8
7.5
7
-40 qC
25 qC
150 qC
6.5
6
-10
20
50
80
110
Temperature Junction (Tj)
140
170
0
10
D001
20
30
40
Input Voltage (V)
50
60
D002
VIN = 12 V
Figure 3. Switch Current Limit vs Junction Temperature
6
Figure 4. Switch Current Limit vs Input Voltage
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550
500
540
450
FSW - Switching Frequency (kHz)
FS - Switching Frequency (kHz)
Typical Characteristics (continued)
530
520
510
500
490
480
470
460
450
350
300
250
200
150
100
50
0
±50
±25
0
25
50
75
100
125
TJ - Junction Temperature (ƒC)
VIN = 12 V
150
200
300
400
500
600
700
800
900
RT/CLK - Resistance (k )
C029
1000
C030
ƒsw (kHz) = 92471 x RT (kΩ)-0.991
RT (kΩ) = 101756 x ƒsw (kHz)-1.008
RT = 200 kΩ
Figure 5. Switching Frequency vs Junction Temperature
Figure 6. Switching Frequency vs RT/CLK Resistance
Low Frequency Range
2500
500
2300
450
2100
1900
400
gm (µA/V)
FSW - Switching Frequency (kHz)
400
1700
1500
1300
350
300
1100
900
250
700
500
200
0
50
100
150
±50
200
RT/CLK - Resistance (k )
±25
0
25
50
75
100
125
TJ - Junction Temperature (ƒC)
C031
150
C032
VIN = 12 V
Figure 7. Switching Frequency vs RT/CLK Resistance
High Frequency Range
Figure 8. EA Transconductance vs Junction Temperature
120
110
EN - Threshold (V)
100
gm (µA/V)
90
80
70
60
50
40
30
20
±50
±25
0
25
50
75
100
TJ - Junction Temperature (ƒC)
125
150
1.3
1.29
1.28
1.27
1.26
1.25
1.24
1.23
1.22
1.21
1.2
1.19
1.18
1.17
1.16
1.15
±50
±25
VIN = 12 V
0
25
50
75
100
125
TJ - Junction Temperature (ƒC)
C033
150
C034
VIN = 12 V
Figure 9. EA Transconductance During Soft-Start vs
Junction Temperature
Figure 10. EN Terminal Voltage vs Junction Temperature
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±3.5
±0.5
±3.7
±0.7
±3.9
±0.9
±4.1
±1.1
±4.3
±1.3
IEN (µA)
IEN (uA)
Typical Characteristics (continued)
±4.5
±4.7
±1.7
±4.9
±1.9
±5.1
±2.1
±5.3
±2.3
±5.5
±2.5
±50
±25
0
25
50
75
100
125
VIN = 12 V
±50
150
TJ - Junction Temperature (ƒC)
±25
IEN = Threshold +50 mV
VIN = 12 V
% of Nominal Switching Frequency
±3.1
±3.3
±3.5
±3.7
±3.9
±4.1
±4.3
50
75
100
125
150
C036
IEN = Threshold –50 mV
100
±2.9
25
Figure 12. EN Terminal Current vs Junction Temperature
±2.5
±2.7
0
TJ - Junction Temperature (ƒC)
C035
Figure 11. EN Terminal Current vs Junction Temperature
IEN - Hysteresis (µA)
±1.5
Series2
VSENSE
Falling
VSENSE
Rising
Series4
75
50
25
0
±4.5
±50
±25
0
25
50
75
100
125
0.0
150
TJ - Junction Temperature (ƒC)
0.1
0.2
0.3
0.4
0.5
0.6
0.7
VSENSE (V)
C037
0.8
C038
VIN = 12 V
3
3
2.5
2.5
2
2
1.5
1.5
1
1
0.5
0.5
0
0
±50
±25
0
25
50
75
100
125
150
TJ - Junction Temperature (ƒC)
VIN = 12 V
0
10
20
30
40
VIN - Input Voltage (V)
C039
50
60
C040
TA = 25°C
Figure 15. Shutdown Supply Current vs Junction
Temperature
8
Figure 14. Switching Frequency vs VSENSE
IVIN (µA)
IVIN (µA)
Figure 13. EN Terminal Current Hysteresis vs Junction
Temperature
Figure 16. Shutdown Supply Current vs Input Voltage (VIN)
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210
210
190
190
170
170
IVIN (µA)
IVIN (µA)
Typical Characteristics (continued)
150
130
150
130
110
110
90
90
70
70
±50
±25
0
25
50
75
100
125
150
TJ - Junction Temperature (ƒC)
0
40
50
60
C042
Figure 18. VIN Supply Current vs Input Voltage
4.5
BOOT-PH UVLO Falling
BOOT-PH UVLO Rising
UVLO Start Switching
UVLO Stop Switching
4.4
2.4
4.3
2.3
4.2
VIN (V)
VI - BOOT-PH (V)
30
TJ = 25°C
Figure 17. VIN Supply Current vs Junction Temperature
2.5
20
VIN - Input Voltage (V)
VIN = 12 V
2.6
10
C041
2.2
4.1
2.1
4.0
2.0
3.9
1.9
3.8
3.7
1.8
±50
±25
0
25
50
75
100
125
±50
150
TJ - Junction Temperature (ƒC)
±25
0
25
50
75
100
125
TJ - Junction Temperature (ƒC)
C043
Figure 19. BOOT-SW UVLO vs Junction Temperature
150
C044
Figure 20. Input Voltage UVLO vs Junction Temperature
10
9
Soft-Start Time (ms)
8
7
6
5
4
3
2
1
0
2500
2300
2100
1900
1700
1500
1300
1100
900
700
500
300
100
C045
Switching Frequency (kHz)
VIN = 12 V
TJ = 25°C
Figure 21. Soft-Start Time vs Switching Frequency
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7 Detailed Description
7.1 Overview
The TPS54560B-Q1 is a 60 V, 5 A, step-down (buck) regulator with an integrated high side n-channel MOSFET.
The device implements constant frequency, current mode control which reduces output capacitance and
simplifies external frequency compensation. The wide switching frequency range of 100 kHz to 2500 kHz allows
either efficiency or size optimization when selecting the output filter components. The switching frequency is
adjusted using a resistor to ground connected to the RT/CLK terminal. The device has an internal phase-locked
loop (PLL) connected to the RT/CLK terminal that will synchronize the power switch turn on to a falling edge of
an external clock signal.
The TPS54560B-Q1 has a default input start-up voltage of approximately 4.3 V. The EN terminal can be used to
adjust the input voltage undervoltage lockout (UVLO) threshold with two external resistors. An internal pull up
current source enables operation when the EN terminal is floating. The operating current is 146 μA under no load
condition (not switching). When the device is disabled, the supply current is 2 μA.
The integrated 92mΩ high side MOSFET supports high efficiency power supply designs capable of delivering 5
amperes of continuous current to a load. The gate drive bias voltage for the integrated high side MOSFET is
supplied by a bootstrap capacitor connected from the BOOT to SW terminals. The TPS54560B-Q1 reduces the
external component count by integrating the bootstrap recharge diode. The BOOT terminal capacitor voltage is
monitored by a UVLO circuit which turns off the high side MOSFET when the BOOT to SW voltage falls below a
preset threshold. An automatic BOOT capacitor recharge circuit allows the TPS54560B-Q1to operate at high
duty cycles approaching 100%. Therefore, the maximum output voltage is near the minimum input supply voltage
of the application. The minimum output voltage is the internal 0.8 V feedback reference.
Output overvoltage transients are minimized by an Overvoltage Transient Protection (OVP) comparator. When
the OVP comparator is activated, the high side MOSFET is turned off and remains off until the output voltage is
less than 106% of the desired output voltage.
The TPS54560B-Q1 includes an internal soft-start circuit that slows the output rise time during start-up to reduce
in-rush current and output voltage overshoot. Output overload conditions reset the soft-start timer. When the
overload condition is removed, the soft-start circuit controls the recovery from the fault output level to the nominal
regulation voltage. A frequency foldback circuit reduces the switching frequency during start-up and overcurrent
fault conditions to help maintain control of the inductor current.
10
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7.2 Functional Block Diagram
EN
VIN
Thermal
Shutdown
UVLO
Enable
Comparator
OV
Shutdown
Shutdown
Logic
Enable
Threshold
Boot
Charge
Voltage
Reference
Boot
UVLO
Minimum
Clamp
Pulse
Skip
Error
Amplifier
Current
Sense
PWM
Comparator
FB
BOOT
Logic
Shutdown
6
Slope
Compensation
SW
COMP
Frequency
Foldback
Reference
DAC for
Soft- Start
Maximum
Clamp
Oscillator
with PLL
8/8/ 2012 A 0192789
GND
POWERPAD
RT/ CLK
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7.3 Feature Description
7.3.1 Fixed Frequency PWM Control
The TPS54560B-Q1 uses fixed frequency, peak current mode control with adjustable switching frequency. The
output voltage is compared through external resistors connected to the FB terminal to an internal voltage
reference by an error amplifier. An internal oscillator initiates the turn on of the high side power switch. The error
amplifier output at the COMP terminal controls the high side power switch current. When the high side MOSFET
switch current reaches the threshold level set by the COMP voltage, the power switch is turned off. The COMP
terminal voltage will increase and decrease as the output current increases and decreases. The device
implements current limiting by clamping the COMP terminal voltage to a maximum level. The pulse skipping Ecomode is implemented with a minimum voltage clamp on the COMP terminal.
7.3.2 Slope Compensation Output Current
The TPS54560B-Q1 adds a compensating ramp to the MOSFET switch current sense signal. This slope
compensation prevents sub-harmonic oscillations at duty cycles greater than 50%. The peak current limit of the
high side switch is not affected by the slope compensation and remains constant over the full duty cycle range.
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Feature Description (continued)
7.3.3 Pulse Skip Eco-mode
The TPS54560B-Q1 operates in a pulse skipping Eco-mode at light load currents to improve efficiency by
reducing switching and gate drive losses. If the output voltage is within regulation and the peak switch current at
the end of any switching cycle is below the pulse skipping current threshold, the device enters Eco-mode. The
pulse skipping current threshold is the peak switch current level corresponding to a nominal COMP voltage of
600 mV.
When in Eco-mode, the COMP terminal voltage is clamped at 600 mV and the high side MOSFET is inhibited.
Since the device is not switching, the output voltage begins to decay. The voltage control loop responds to the
falling output voltage by increasing the COMP terminal voltage. The high side MOSFET is enabled and switching
resumes when the error amplifier lifts COMP above the pulse skipping threshold. The output voltage recovers to
the regulated value, and COMP eventually falls below the Eco-mode pulse skipping threshold at which time the
device again enters Eco-mode. The internal PLL remains operational when in Eco-mode. When operating at light
load currents in Eco-mode, the switching transitions occur synchronously with the external clock signal.
During Eco-mode operation, the TPS54560B-Q1 senses and controls peak switch current, not the average load
current. Therefore the load current at which the device enters Eco-mode is dependent on the output inductor
value. The circuit in Figure 33 enters Eco-mode at about 25.3 mA output current. As the load current approaches
zero, the device enters a pulse skip mode during which it draws only 146 μA input quiescent current.
7.3.4 Low Dropout Operation and Bootstrap Voltage (BOOT)
The TPS54560B-Q1 provides an integrated bootstrap voltage regulator. A small capacitor between the BOOT
and SW terminals provides the gate drive voltage for the high side MOSFET. The BOOT capacitor is refreshed
when the high side MOSFET is off and the external low side diode conducts. The recommended value of the
BOOT capacitor is 0.1 μF. A ceramic capacitor with an X7R or X5R grade dielectric with a voltage rating of 10 V
or higher is recommended for stable performance over temperature and voltage.
When operating with a low voltage difference from input to output, the high side MOSFET of the TPS54560B-Q1
will operate at 100% duty cycle as long as the BOOT to SW terminal voltage is greater than 2.1 V. When the
voltage from BOOT to SW drops below 2.1V, the high side MOSFET is turned off and an integrated low side
MOSFET pulls SW low to recharge the BOOT capacitor. To reduce the losses of the small low side MOSFET at
high output voltages, it is disabled at 24 V output and re-enabled when the output reaches 21.5 V.
Since the gate drive current sourced from the BOOT capacitor is small, the high side MOSFET can remain on for
many switching cycles before the MOSFET is turned off to refresh the capacitor. Thus the effective duty cycle of
the switching regulator can be high, approaching 100%. The effective duty cycle of the converter during dropout
is mainly influenced by the voltage drops across the power MOSFET, the inductor resistance, the low side diode
voltage and the printed circuit board resistance.
During high duty cycle (low dropout) conditions, inductor current ripple increases when the BOOT capacitor is
being recharged resulting in an increase in output voltage ripple. Increased ripple occurs when the off time
required to recharge the BOOT capacitor is longer than the high side off time associated with cycle by cycle
PWM control.
Equation 1 calculates the minimum input voltage required to regulate the output voltage and ensure proper
operation of the device. This calculation must include tolerance of the component specifications and the variation
of these specifications at their maximum operating temperature in the application
V IN (min)
V OUT
VF
R dc u I OUT
D
R DS(on) u I OUT
VF
where
•
•
•
•
12
VF = Schottky diode forward voltage
RDC = Total DC resistance of inductor and PCB
RDS(on) = High-side MOSFET resistance
D = 0.99 effective duty cycle
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Feature Description (continued)
At heavy loads, the minimum input voltage must be increased to ensure a monotonic startup. Equation 2 can be
used to calculate the minimum input voltage for this condition.
V OUT(max) = D (max) x (V IN(min) - I OUT(max) x R DS(on) + VF) - VF + I OUT(max) x R dc
where
•
•
•
•
•
•
D(max) ≥ 0.9
IB2SW = 100 µA
TSW = 1 / Fsw
VB2SW = VBOOT + VF
VBOOT = (1.41 x VIN - 0.554 - VF / TSW - 1.847 x 103 x IB2SW) / (1.41 + 1 / Tsw)
RDS(on) = 1 / (-0.3 x VB2SW2 + 3.577 x VB2SW - 4.246)
(2)
7.3.5 Error Amplifier
The TPS54560B-Q1 voltage regulation loop is controlled by a transconductance error amplifier. The error
amplifier compares the FB terminal voltage to the lower of the internal soft-start voltage or the internal 0.8 V
voltage reference. The transconductance (gm) of the error amplifier is 350 μA/V during normal operation. During
soft-start operation, the transconductance is reduced to 78 μA/V and the error amplifier is referenced to the
internal soft-start voltage.
The frequency compensation components (capacitor, series resistor and capacitor) are connected between the
error amplifier output COMP terminal and GND terminal.
7.3.6 Adjusting the Output Voltage
The internal voltage reference produces a precise 0.8 V ±1% voltage reference over the operating temperature
and voltage range by scaling the output of a bandgap reference circuit. The output voltage is set by a resistor
divider from the output node to the FB terminal. It is recommended to use 1% tolerance or better divider
resistors. Select the low side resistor RLS for the desired divider current and use Equation 3 to calculate RHS. To
improve efficiency at light loads consider using larger value resistors. However, if the values are too high, the
regulator will be more susceptible to noise and voltage errors from the FB input current may become noticeable.
æ Vout - 0.8V ö
RHS = RLS ´ ç
÷
0.8 V
è
ø
(3)
7.3.7 Enable and Adjusting Undervoltage Lockout
The TPS54560B-Q1 is enabled when the VIN terminal voltage rises above 4.3 V and the EN terminal voltage
exceeds the enable threshold of 1.2 V. The TPS54560B-Q1 is disabled when the VIN terminal voltage falls below
4 V or when the EN terminal voltage is below 1.2 V. The EN terminal has an internal pull-up current source, I1, of
1.2 μA that enables operation of the TPS54560B-Q1 when the EN terminal floats.
If an application requires a higher undervoltage lockout (UVLO) threshold, use the circuit shown in Figure 22 to
adjust the input voltage UVLO with two external resistors. When the EN terminal voltage exceeds 1.2 V, an
additional 3.4 μA of hysteresis current, IHYS, is sourced out of the EN terminal. When the EN terminal is pulled
below 1.2 V, the 3.4 μA Ihys current is removed. This additional current facilitates adjustable input voltage UVLO
hysteresis. Use Equation 4 to calculate RUVLO1 for the desired UVLO hysteresis voltage. Use Equation 5 to
calculate RUVLO2 for the desired VIN start voltage.
In applications designed to start at relatively low input voltages (that is, from 4.5 V to 9 V) and withstand high
input voltages (that is, from 40 V to 60 V), the EN terminal may experience a voltage greater than the absolute
maximum voltage of 8.4 V during the high input voltage condition. To avoid exceeding this voltage when using
the EN resistors, the EN terminal is clamped internally with a 5.8 V zener diode that will sink up to 150 μA.
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Feature Description (continued)
VIN
TPS54560B-Q1
i1
VIN
ihys
RUVLO1
RUVLO1
EN
EN
10 kW
Node
VEN
RUVLO2
RUVLO2
Copyright © 2016, Texas Instruments Incorporated
Figure 22. Adjustable Undervoltage Lockout
(UVLO)
Copyright © 2016, Texas Instruments Incorporated
Figure 23. Low Input Voltages Applications
- VSTOP
V
RUVLO1 = START
IHYS
RUVLO2 =
5.8 V
(4)
VENA
VSTART - VENA
+ I1
RUVLO1
(5)
7.3.8 Internal Soft-Start
The TPS54560B-Q1 has an internal digital soft-start that ramps the reference voltage from zero volts to its final
value in 1024 switching cycles. The internal soft-start time (10% to 90%) is calculated using Equation 6.
1024
tSS (ms) =
fSW (kHz)
(6)
If the EN terminal is pulled below the stop threshold of 1.2 V, switching stops and the internal soft-start resets.
The soft-start also resets in thermal shutdown.
7.3.9 Constant Switching Frequency and Timing Resistor (RT/CLK) Terminal)
The switching frequency of the TPS54560B-Q1 is adjustable over a wide range from 100 kHz to 2500 kHz by
placing a resistor between the RT/CLK terminal and GND terminal. The RT/CLK terminal voltage is typically 0.5
V and must have a resistor to ground to set the switching frequency. To determine the timing resistance for a
given switching frequency, use Equation 7 or Equation 8 or the curves in Figure 5 and Figure 6. To reduce the
solution size one would typically set the switching frequency as high as possible, but tradeoffs of the conversion
efficiency, maximum input voltage and minimum controllable on time should be considered. The minimum
controllable on time is typically 135 ns which limits the maximum operating frequency in applications with high
input to output step down ratios. The maximum switching frequency is also limited by the frequency foldback
circuit. A more detailed discussion of the maximum switching frequency is provided in the next section.
101756
RT (kW) =
f sw (kHz)1.008
(7)
f sw (kHz) =
14
92417
RT (kW)0.991
(8)
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Feature Description (continued)
7.3.10 Accurate Current Limit Operation and Maximum Switching Frequency
The TPS54560B-Q1 implements peak current mode control in which the COMP terminal voltage controls the
peak current of the high side MOSFET. A signal proportional to the high side switch current and the COMP
terminal voltage are compared each cycle. When the peak switch current intersects the COMP control voltage,
the high side switch is turned off. During overcurrent conditions that pull the output voltage low, the error
amplifier increases switch current by driving the COMP terminal high. The error amplifier output is clamped
internally at a level which sets the peak switch current limit. The TPS54560B-Q1 provides an accurate current
limit threshold with a typical current limit delay of 60 ns. With smaller inductor values, the delay will result in a
higher peak inductor current. The relationship between the inductor value and the peak inductor current is shown
in Figure 24.
Inductor Current (A)
Peak Inductor Current
ΔCLPeak
Open Loop Current Limit
ΔCLPeak = VIN/L x tCLdelay
tCLdelay
tON
Figure 24. Current Limit Delay
To protect the converter in overload conditions at higher switching frequencies and input voltages, the
TPS54560B-Q1 implements a frequency foldback. The oscillator frequency is divided by 1, 2, 4, and 8 as the FB
terminal voltage falls from 0.8 V to 0 V. The TPS54560B-Q1 uses a digital frequency foldback to enable
synchronization to an external clock during normal start-up and fault conditions. During short-circuit events, the
inductor current can exceed the peak current limit because of the high input voltage and the minimum
controllable on time. When the output voltage is forced low by the shorted load, the inductor current decreases
slowly during the switch off time. The frequency foldback effectively increases the off time by increasing the
period of the switching cycle providing more time for the inductor current to ramp down.
With a maximum frequency foldback ratio of 8, there is a maximum frequency at which the inductor current can
be controlled by frequency foldback protection. Equation 10 calculates the maximum switching frequency at
which the inductor current will remain under control when VOUT is forced to VOUT(SC). The selected operating
frequency should not exceed the calculated value.
Equation 9 calculates the maximum switching frequency limitation set by the minimum controllable on time and
the input to output step down ratio. Setting the switching frequency above this value will cause the regulator to
skip switching pulses to achieve the low duty cycle required at maximum input voltage.
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Feature Description (continued)
æ I ´R + V
dc
OUT + Vd
´ç O
ç VIN - IO ´ RDS(on ) + Vd
è
ö
÷
÷
ø
fDIV æç ICL ´ Rdc + VOUT(sc ) + Vd
´
tON ç VIN - ICL ´ RDS(on ) + Vd
è
ö
÷
÷
ø
fSW (max skip ) =
fSW(shift) =
1
tON
(9)
where
•
•
•
•
•
•
•
•
•
•
IO — Output current
ICL — Current limit
Rdc — inductor resistance
VIN — maximum input voltage
VOUT — output voltage
VOUTSC — output voltage during short
Vd — diode voltage drop
RDS(on) — switch on resistance
tON — controllable on time
ƒDIV — frequency divide equals (1, 2, 4, or 8)
(10)
7.3.11 Synchronization to RT/CLK Terminal
The RT/CLK terminal can receive a frequency synchronization signal from an external system clock. To
implement this synchronization feature connect a square wave to the RT/CLK terminal through either circuit
network shown in Figure 25. The square wave applied to the RT/CLK terminal must switch lower than 0.5 V and
higher than 1.7 V and have a pulsewidth greater than 15 ns. The synchronization frequency range is 160 kHz to
2300 kHz. The rising edge of the SW will be synchronized to the falling edge of RT/CLK terminal signal. The
external synchronization circuit should be designed such that the default frequency set resistor is connected from
the RT/CLK terminal to ground when the synchronization signal is off. When using a low impedance signal
source, the frequency set resistor is connected in parallel with an ac coupling capacitor to a termination resistor
(e.g., 50 Ω) as shown in Figure 25. The two resistors in series provide the default frequency setting resistance
when the signal source is turned off. The sum of the resistance should set the switching frequency close to the
external CLK frequency. It is recommended to ac couple the synchronization signal through a 10 pF ceramic
capacitor to RT/CLK terminal.
The first time the RT/CLK is pulled above the PLL threshold the TPS54560B-Q1 switches from the RT resistor
free-running frequency mode to the PLL synchronized mode. The internal 0.5 V voltage source is removed and
the RT/CLK terminal becomes high impedance as the PLL starts to lock onto the external signal. The switching
frequency can be higher or lower than the frequency set with the RT/CLK resistor. The device transitions from
the resistor mode to the PLL mode and locks onto the external clock frequency within 78 microseconds. During
the transition from the PLL mode to the resistor programmed mode, the switching frequency will fall to 150 kHz
and then increase or decrease to the resistor programmed frequency when the 0.5 V bias voltage is reapplied to
the RT/CLK resistor.
The switching frequency is divided by 8, 4, 2, and 1 as the FB terminal voltage ramps from 0 to 0.8 volts. The
device implements a digital frequency foldback to enable synchronizing to an external clock during normal startup and fault conditions. Figure 26, Figure 27 and Figure 28 show the device synchronized to an external system
clock in continuous conduction mode (CCM), discontinuous conduction (DCM), and pulse skip mode (Eco-Mode).
SPACER
16
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Feature Description (continued)
TPS54560B-Q1
TPS54560B-Q1
RT/CLK
RT/CLK
PLL
PLL
RT
Hi-Z
Clock
Source
Clock
Source
RT
Copyright © 2016, Texas Instruments Incorporated
Figure 25. Synchronizing to a System Clock
SW
SW
EXT
EXT
IL
IL
Figure 26. Plot of Synchronizing in CCM
Figure 27. Plot of Synchronizing in DCM
SW
EXT
IL
Figure 28. Plot of Synchronizing in Eco-mode
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Feature Description (continued)
7.3.12 Overvoltage Protection
The TPS54560B-Q1 incorporates an output overvoltage protection (OVP) circuit to minimize voltage overshoot
when recovering from output fault conditions or strong unload transients in designs with low output capacitance.
For example, when the power supply output is overloaded the error amplifier compares the actual output voltage
to the internal reference voltage. If the FB terminal voltage is lower than the internal reference voltage for a
considerable time, the output of the error amplifier will increase to a maximum voltage corresponding to the peak
current limit threshold. When the overload condition is removed, the regulator output rises and the error amplifier
output transitions to the normal operating level. In some applications, the power supply output voltage can
increase faster than the response of the error amplifier output resulting in an output overshoot.
The OVP feature minimizes output overshoot when using a low value output capacitor by comparing the FB
terminal voltage to the rising OVP threshold which is nominally 109% of the internal voltage reference. If the FB
terminal voltage is greater than the rising OVP threshold, the high side MOSFET is immediately disabled to
minimize output overshoot. When the FB voltage drops below the falling OVP threshold which is nominally 106%
of the internal voltage reference, the high side MOSFET resumes normal operation.
7.3.13 Thermal Shutdown
The TPS54560B-Q1 provides an internal thermal shutdown to protect the device when the junction temperature
exceeds 176°C. The high side MOSFET stops switching when the junction temperature exceeds the thermal trip
threshold. Once the die temperature falls below 164°C, the device reinitiates the power up sequence controlled
by the internal soft-start circuitry.
7.3.14 Small Signal Model for Loop Response
Figure 29 shows an equivalent model for the TPS54560B-Q1 control loop which can be simulated to check the
frequency response and dynamic load response. The error amplifier is a transconductance amplifier with a gmEA
of 350 μA/V. The error amplifier can be modeled using an ideal voltage controlled current source. The resistor Ro
and capacitor Co model the open loop gain and frequency response of the amplifier. The 1mV ac voltage source
between the nodes a and b effectively breaks the control loop for the frequency response measurements.
Plotting c/a provides the small signal response of the frequency compensation. Plotting a/b provides the small
signal response of the overall loop. The dynamic loop response can be evaluated by replacing RL with a current
source with the appropriate load step amplitude and step rate in a time domain analysis. This equivalent model is
only valid for continuous conduction mode (CCM) operation.
SW
VO
Power Stage
gmps 17 A/V
a
b
RESR
R1
RL
COMP
c
0.8 V
CO
R3
C2
RO
FB
COUT
gmea
R2
350 mA/V
C1
Copyright © 2016, Texas Instruments Incorporated
Figure 29. Small Signal Model for Loop Response
18
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Feature Description (continued)
7.3.15 Simple Small Signal Model for Peak Current Mode Control
Figure 30 describes a simple small signal model that can be used to design the frequency compensation. The
TPS54560B-Q1 power stage can be approximated by a voltage-controlled current source (duty cycle modulator)
supplying current to the output capacitor and load resistor. The control to output transfer function is shown in
Equation 11 and consists of a dc gain, one dominant pole, and one ESR zero. The quotient of the change in
switch current and the change in COMP terminal voltage (node c in Figure 29) is the power stage
transconductance, gmPS. The gmPS for the TPS54560B-Q1 is 17 A/V. The low-frequency gain of the power stage
is the product of the transconductance and the load resistance as shown in Equation 12.
As the load current increases and decreases, the low-frequency gain decreases and increases, respectively. This
variation with the load may seem problematic at first glance, but fortunately the dominant pole moves with the
load current (see Equation 13). The combined effect is highlighted by the dashed line in the right half of
Figure 30. As the load current decreases, the gain increases and the pole frequency lowers, keeping the 0-dB
crossover frequency the same with varying load conditions. The type of output capacitor chosen determines
whether the ESR zero has a profound effect on the frequency compensation design. Using high ESR aluminum
electrolytic capacitors may reduce the number frequency compensation components needed to stabilize the
overall loop because the phase margin is increased by the ESR zero of the output capacitor (see Equation 14).
VO
Adc
VC
RESR
fp
RL
gmps
COUT
fz
Figure 30. Simple Small Signal Model and Frequency Response for Peak Current Mode Control
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Feature Description (continued)
æ
s
ç1 +
2p ´ fZ
VOUT
= Adc ´ è
VC
æ
s
ç1 +
2p ´ fP
è
Adc = gmps ´ RL
fP =
ö
÷
ø
ö
÷
ø
(11)
(12)
1
COUT ´ RL ´ 2p
(13)
1
fZ =
COUT ´ RESR ´ 2p
(14)
7.3.16 Small Signal Model for Frequency Compensation
The TPS54560B-Q1 uses a transconductance amplifier for the error amplifier and supports three of the
commonly-used frequency compensation circuits. Compensation circuits Type 2A, Type 2B, and Type 1 are
shown in Figure 31. Type 2 circuits are typically implemented in high bandwidth power-supply designs using low
ESR output capacitors. The Type 1 circuit is used with power-supply designs with high-ESR aluminum
electrolytic or tantalum capacitors. Equation 15 and Equation 16 relate the frequency response of the amplifier to
the small signal model in Figure 31. The open-loop gain and bandwidth are modeled using the RO and CO shown
in Figure 31. See the application section for a design example using a Type 2A network with a low ESR output
capacitor.
Equation 15 through Equation 24 are provided as a reference. An alternative is to use WEBENCH software tools
to create a design based on the power supply requirements.
VO
R1
FB
gmea
COMP
Type 2A
Type 2B
Type 1
Vref
R2
RO
CO
R3
C2
R3
C2
C1
C1
Copyright © 2016, Texas Instruments Incorporated
Figure 31. Types of Frequency Compensation
20
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Feature Description (continued)
Aol
A0
P1
Z1
P2
A1
BW
Figure 32. Frequency Response of the Type 2A and Type 2B Frequency Compensation
Aol(V/V)
gmea
gmea
=
2p ´ BW (Hz)
Ro =
CO
(15)
(16)
æ
ö
s
ç1 +
÷
2p ´ fZ1 ø
è
EA = A0 ´
æ
ö æ
ö
s
s
ç1 +
÷ ´ ç1 +
÷
2p ´ fP1 ø è
2p ´ fP2 ø
è
A0 = gmea
A1 = gmea
P1 =
Z1 =
P2 =
(17)
R2
´ Ro ´
R1 + R2
R2
´ Ro| | R3 ´
R1 + R2
(18)
(19)
1
2p ´ Ro ´ C1
(20)
1
2p ´ R3 ´ C1
(21)
1
type 2a
2p ´ R3 | | RO ´ (C2 + CO )
(22)
1
P2 =
type 2b
2p ´ R3 | | RO ´ CO
P2 =
2p ´ R O
(23)
1
type 1
´ (C2 + C O )
(24)
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7.4 Device Functional Modes
7.4.1 Operation with VIN < 4.5 V (Minimum VIN)
The device is recommended to operate with input voltages above 4.5 V. The typical VIN UVLO threshold is 4.3 V
and the device may operate at input voltages down to the UVLO voltage. At input voltages below the actual
UVLO voltage, the device will not switch. If EN is externally pulled up to VIN or left floating, when VIN passes the
UVLO threshold the device will become active. Switching is enabled the soft start sequence is initiated. The
TPS54560B-Q1 will start at the soft start time determined by the internal soft start time.
7.4.2 Operation with EN Control
The enable threshold voltage is 1.2 V typical. With EN held below that voltage the device is disabled and
switching is inhibited even if VIN is above its UVLO threshold. The IC quiescent current is reduced in this state. If
the EN voltage is increased above the threshold while VIN is above its UVLO threshold, the device becomes
active. Switching is enabled, and the soft start sequence is initiated. The TPS54560B-Q1 will start at the soft
start time determined by the internal soft start time.
22
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8 Application and Implementation
NOTE
Information in the following applications sections is not part of the TI component
specification, and TI does not warrant its accuracy or completeness. TI’s customers are
responsible for determining suitability of components for their purposes. Customers should
validate and test their design implementation to confirm system functionality.
8.1 Application Information
The TPS54560B-Q1 is a 60 V, 5 A, step down regulator with an integrated high side MOSFET. Idea applications
are: 12 V, 24 V and 48 V Industrial, automotive and communications power systems
8.2 Typical Application
L1
7.2uH
5V, 5A VOUT
C4 0.1uF
U1
TPS54560B-Q1
7V to 60V
2
3
C10
C3
C1
C2
2.2uF
2.2uF
2.2uF
2.2uF
R1
442k
4
SW
BOOT
VIN
GND
EN
COMP
RT/CLK
PWRPD
1
VIN
9
R2
90.9k
R3
243k
FB
8
D1
C6
C7
C9
B560C
47uF
47uF
47uF
R5
53.6k
7
6
5
FB
FB
R4
16.9k
C8
R6
10.2k
47pF
C5
4700pF
Copyright © 2016, Texas Instruments Incorporated
Figure 33. 5 V Output TPS54560B-Q1 Design Example
8.2.1 Design Requirements
This guide illustrates the design of a high frequency switching regulator using ceramic output capacitors. A few
parameters must be known in order to start the design process. These requirements are typically determined at
the system level. Calculations can be done with the aid of WEBENCH or the excel spreadsheet (SLVC452)
located on the product page. For this example, start with the following known parameters:
Table 1. Design Parameters
DESIGN PARAMETERS
EXAMPLE VALUES
Output Voltage
5V
Transient Response 1.25 A to 3.75 A load step
ΔVOUT = 4 %
Maximum Output Current
5A
Input Voltage
12 V nom. 7 V to 60 V
Output Voltage Ripple
0.5% of VOUT
Start Input Voltage (rising VIN)
6.5 V
Stop Input Voltage (falling VIN)
5V
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8.2.2 Detailed Design Procedure
8.2.2.1 Custom Design with WEBENCH® Tools
Click here to create a custom design using the TPS54560B-Q1 device with the WEBENCH® Power Designer.
1. Start by entering your VIN, VOUT, and IOUT requirements.
2. Optimize your design for key parameters like efficiency, footprint and cost using the optimizer dial and
compare this design with other possible solutions from Texas Instruments.
3. The WEBENCH Power Designer provides you with a customized schematic along with a list of materials with
real time pricing and component availability.
4. In most cases, you will also be able to:
– Run electrical simulations to see important waveforms and circuit performance
– Run thermal simulations to understand the thermal performance of your board
– Export your customized schematic and layout into popular CAD formats
– Print PDF reports for the design, and share your design with colleagues
5. Get more information about WEBENCH tools at www.ti.com/WEBENCH.
8.2.2.2 Selecting the Switching Frequency
The first step is to choose a switching frequency for the regulator. Typically, the designer uses the highest
switching frequency possible since this produces the smallest solution size. High switching frequency allows for
lower value inductors and smaller output capacitors compared to a power supply that switches at a lower
frequency. The switching frequency that can be selected is limited by the minimum on-time of the internal power
switch, the input voltage, the output voltage and the frequency foldback protection.
Equation 9 and Equation 10 should be used to calculate the upper limit of the switching frequency for the
regulator. Choose the lower value result from the two equations. Switching frequencies higher than these values
results in pulse skipping or the lack of overcurrent protection during a short circuit.
The typical minimum on time, tonmin, is 135 ns for the TPS54560B-Q1. For this example, the output voltage is 5 V
and the maximum input voltage is 60 V, which allows for a maximum switch frequency up to 708 kHz to avoid
pulse skipping from Equation 9. To ensure overcurrent runaway is not a concern during short circuits use
Equation 10 to determine the maximum switching frequency for frequency foldback protection. With a maximum
input voltage of 60 V, assuming a diode voltage of 0.7 V, inductor resistance of 11 mΩ, switch resistance of 92
mΩ, a current limit value of 6 A and short circuit output voltage of 0.1 V, the maximum switching frequency is 855
kHz.
For this design, a lower switching frequency of 400 kHz is chosen to operate comfortably below the calculated
maximums. To determine the timing resistance for a given switching frequency, use Equation 7 or the curve in
Figure 6. The switching frequency is set by resistor R3 shown in Figure 33. For 400 kHz operation, the closest
standard value resistor is 243 kΩ.
1
æ 5 A x 11 mW + 5 V + 0.7 V ö
´ ç
fSW(max skip) =
÷ = 708 kHz
135ns
è 60 V - 5 A x 92 mW + 0.7 V ø
(25)
8
æ 6 A x 11 mW + 0.1 V + 0.7 V ö
´ ç
÷ = 855 kHz
135 ns
è 60 V - 6 A x 92 mW + 0.7 V ø
101756
RT (kW) =
= 242 kW
400 (kHz)1.008
fSW(shift) =
(26)
(27)
8.2.2.3 Output Inductor Selection (LO)
To calculate the minimum value of the output inductor, use Equation 28.
KIND is a ratio that represents the amount of inductor ripple current relative to the maximum output current. The
inductor ripple current is filtered by the output capacitor. Therefore, choosing high inductor ripple currents
impacts the selection of the output capacitor since the output capacitor must have a ripple current rating equal to
or greater than the inductor ripple current. In general, the inductor ripple value is at the discretion of the designer,
however, the following guidelines may be used.
24
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For designs using low ESR output capacitors such as ceramics, a value as high as KIND = 0.3 may be desirable.
When using higher ESR output capacitors, KIND = 0.2 yields better results. Since the inductor ripple current is
part of the current mode PWM control system, the inductor ripple current should always be greater than 150 mA
for stable PWM operation. In a wide input voltage regulator, it is best to choose relatively large inductor ripple
current. This provides sufficienct ripple current with the input voltage at the minimum.
For this design example, KIND = 0.3 and the inductor value is calculated to be 7.6 μH. The nearest standard value
is 7.2 μH. It is important that the RMS current and saturation current ratings of the inductor not be exceeded. The
RMS and peak inductor current can be found from Equation 30 and Equation 31. For this design, the RMS
inductor current is 5 A and the peak inductor current is 5.8 A. The chosen inductor is a WE 7447798720, which
has a saturation current rating of 7.9 A and an RMS current rating of 6 A.
As the equation set demonstrates, lower ripple currents will reduce the output voltage ripple of the regulator but
will require a larger value of inductance. Selecting higher ripple currents will increase the output voltage ripple of
the regulator but allow for a lower inductance value.
The current flowing through the inductor is the inductor ripple current plus the output current. During power up,
faults or transient load conditions, the inductor current can increase above the peak inductor current level
calculated above. In transient conditions, the inductor current can increase up to the switch current limit of the
device. For this reason, the most conservative design approach is to choose an inductor with a saturation current
rating equal to or greater than the switch current limit of the TPS54560B-Q1 which is nominally 7.5 A.
VIN(max ) - VOUT
VOUT
60 V - 5 V
5V
´
=
´
= 7.6 mH
LO(min ) =
IOUT ´ KIND
VIN(max ) ´ fSW
5 A x 0.3
60 V ´ 400 kHz
(28)
spacer
IRIPPLE =
VOUT ´ (VIN(max ) - VOUT )
VIN(max ) ´ LO ´ fSW
=
5 V x (60 V - 5 V)
= 1.591 A
60 V x 7.2 mH x 400 kHz
(29)
spacer
(
æ
1 ç VOUT ´ VIN(max ) - VOUT
2
IL(rms ) = (IOUT ) +
´
12 çç
VIN(max ) ´ LO ´ fSW
è
)
2
ö
÷
÷ =
÷
ø
2
(5 A )
2
æ
5 V ´ (60 V - 5 V ) ö
1
+
´ ç
÷ =5A
ç
÷
12
è 60 V ´ 7.2 mH ´ 400 kHz ø
(30)
spacer
IL(peak ) = IOUT +
IRIPPLE
1.591 A
= 5A +
= 5.797 A
2
2
(31)
8.2.2.4 Output Capacitor
There are three primary considerations for selecting the value of the output capacitor. The output capacitor
determines the modulator pole, the output voltage ripple, and how the regulator responds to a large change in
load current. The output capacitance needs to be selected based on the most stringent of these three criteria.
The desired response to a large change in the load current is the first criteria. The output capacitor needs to
supply the increased load current until the regulator responds to the load step. The regulator does not respond
immediately to a large, fast increase in the load current such as transitioning from no load to a full load. The
regulator usually needs two or more clock cycles for the control loop to sense the change in output voltage and
adjust the peak switch current in response to the higher load. The output capacitance must be large enough to
supply the difference in current for 2 clock cycles to maintain the output voltage within the specified range.
Equation 32 shows the minimum output capacitance necessary, where ΔIOUT is the change in output current, ƒsw
is the regulators switching frequency and ΔVOUT is the allowable change in the output voltage. For this example,
the transient load response is specified as a 4% change in VOUT for a load step from 1.25 A to 3.75 A. Therefore,
ΔIOUT is 3.75 A - 1.25 A = 2.5 A and ΔVOUT = 0.04 × 5 = 0.2 V. Using these numbers gives a minimum
capacitance of 62.5 μF. This value does not take the ESR of the output capacitor into account in the output
voltage change. For ceramic capacitors, the ESR is usually small enough to be ignored. Aluminum electrolytic
and tantalum capacitors have higher ESR that must be included in load step calculations.
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The output capacitor must also be sized to absorb energy stored in the inductor when transitioning from a high to
low load current. The catch diode of the regulator can not sink current so energy stored in the inductor can
produce an output voltage overshoot when the load current rapidly decreases. A typical load step response is
shown in Figure 34. The excess energy absorbed in the output capacitor will increase the voltage on the
capacitor. The capacitor must be sized to maintain the desired output voltage during these transient periods.
Equation 33 calculates the minimum capacitance required to keep the output voltage overshoot to a desired
value, where LO is the value of the inductor, IOH is the output current under heavy load, IOL is the output under
light load, Vf is the peak output voltage, and Vi is the initial voltage. For this example, the worst case load step
will be from 3.75 A to 1.25 A. The output voltage increases during this load transition and the stated maximum in
our specification is 4 % of the output voltage. This makes Vf = 1.04 × 5 = 5.2. Vi is the initial capacitor voltage
which is the nominal output voltage of 5 V. Using these numbers in Equation 33 yields a minimum capacitance of
44.1 μF.
Equation 34 calculates the minimum output capacitance needed to meet the output voltage ripple specification,
where ƒsw is the switching frequency, VORIPPLE is the maximum allowable output voltage ripple, and IRIPPLE is the
inductor ripple current. Equation 34 yields 19.9 μF.
Equation 35 calculates the maximum ESR an output capacitor can have to meet the output voltage ripple
specification. Equation 35 indicates the ESR should be less than 15.7 mΩ.
The most stringent criteria for the output capacitor is 62.5 μF required to maintain the output voltage within
regulation tolerance during a load transient.
Capacitance de-ratings for aging, temperature and dc bias increases this minimum value. For this example, 3 x
47 μF, 10 V ceramic capacitors with 5 mΩ of ESR will be used. The derated capacitance is 87.4 µF, well above
the minimum required capacitance of 62.5 µF.
Capacitors are generally rated for a maximum ripple current that can be filtered without degrading capacitor
reliability. Some capacitor data sheets specify the Root Mean Square (RMS) value of the maximum ripple
current. Equation 36 can be used to calculate the RMS ripple current that the output capacitor must support. For
this example, Equation 36 yields 459 mA.
2 ´ DIOUT
2 ´ 2.5 A
COUT >
=
= 62.5 mF
fSW ´ DVOUT 400 kHz x 0.2 V
(32)
((I ) - (I ) ) = 7.2 mH x (3.75 A - 1.25 A ) = 44.1 mF
x
(5.2 V - 5 V )
((V ) - (V ) )
2
OH
COUT > LO
2
2
2
OL
2
f
2
2
2
I
1
1
1
1
´
=
= 19.9 mF
COUT >
x
8 ´ fSW æ VORIPPLE ö 8 x 400 kHz
æ 25 mV ö
ç 1.591 A ÷
ç
÷
è
ø
è IRIPPLE ø
V
25 mV
RESR < ORIPPLE =
= 15.7 mW
IRIPPLE
1.591 A
ICOUT(rms) =
(
VOUT ´ VIN(max ) - VOUT
)=
12 ´ VIN(max ) ´ LO ´ fSW
5V ´
(60 V
(33)
(34)
(35)
- 5 V)
12 ´ 60 V ´ 7.2 mH ´ 400 kHz
= 459 mA
(36)
8.2.2.5 Catch Diode
The TPS54560B-Q1 requires an external catch diode between the SW terminal and GND. The selected diode
must have a reverse voltage rating equal to or greater than VIN(max). The peak current rating of the diode must be
greater than the maximum inductor current. Schottky diodes are typically a good choice for the catch diode due
to their low forward voltage. The lower the forward voltage of the diode, the higher the efficiency of the regulator.
Typically, diodes with higher voltage and current ratings have higher forward voltages. A diode with a minimum of
60 V reverse voltage is preferred to allow input voltage transients up to the rated voltage of the TPS54560B-Q1.
For the example design, the B560C-13-F Schottky diode is selected for its lower forward voltage and good
thermal characteristics compared to smaller devices. The typical forward voltage of the B560C-13-F is 0.70 volts
at 5 A.
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The diode must also be selected with an appropriate power rating. The diode conducts the output current during
the off-time of the internal power switch. The off-time of the internal switch is a function of the maximum input
voltage, the output voltage, and the switching frequency. The output current during the off-time is multiplied by
the forward voltage of the diode to calculate the instantaneous conduction losses of the diode. At higher
switching frequencies, the ac losses of the diode need to be taken into account. The ac losses of the diode are
due to the charging and discharging of the junction capacitance and reverse recovery charge. Equation 37 is
used to calculate the total power dissipation, including conduction losses and ac losses of the diode.
The B560C-13-F diode has a junction capacitance of 300 pF. Using Equation 37, the total loss in the diode at the
maximum input voltage is 3.43 Watts.
If the power supply spends a significant amount of time at light load currents or in sleep mode, consider using a
diode which has a low leakage current and slightly higher forward voltage drop.
PD =
(V
IN(max ) - VOUT
)´ I
OUT
+
VIN(max )
(60 V
2
´ Vf d
- 5 V ) ´ 5 A x 0.7 V
60 V
+
C j ´ fSW ´ (VIN + Vf d)
=
2
300 pF x 400 kHz x (60 V + 0.7 V)2
= 3.43 W
2
(37)
8.2.2.6 Input Capacitor
The TPS54560B-Q1 requires a high quality ceramic type X5R or X7R input decoupling capacitor with at least 3
μF of effective capacitance. Some applications will benefit from additional bulk capacitance. The effective
capacitance includes any loss of capacitance due to dc bias effects. The voltage rating of the input capacitor
must be greater than the maximum input voltage. The capacitor must also have a ripple current rating greater
than the maximum input current ripple of the TPS54560B-Q1. The input ripple current can be calculated using
Equation 38.
The value of a ceramic capacitor varies significantly with temperature and the dc bias applied to the capacitor.
The capacitance variations due to temperature can be minimized by selecting a dielectric material that is more
stable over temperature. X5R and X7R ceramic dielectrics are usually selected for switching regulator capacitors
because they have a high capacitance to volume ratio and are fairly stable over temperature. The input capacitor
must also be selected with consideration for the dc bias. The effective value of a capacitor decreases as the dc
bias across a capacitor increases.
For this example design, a ceramic capacitor with at least a 60 V voltage rating is required to support the
maximum input voltage. Common standard ceramic capacitor voltage ratings include 4 V, 6.3 V, 10 V, 16 V, 25
V, 50 V or 100 V. For this example, four 2.2 μF, 100 V capacitors in parallel are used. Table 2 shows several
choices of high voltage capacitors.
The input capacitance value determines the input ripple voltage of the regulator. The input voltage ripple can be
calculated using Equation 39. Using the design example values, IOUT = 5 A, CIN = 8.8 μF, ƒsw = 400 kHz, yields
an input voltage ripple of 355 mV and a rms input ripple current of 2.26 A.
ICI(rms ) = IOUT x
VOUT
x
VIN(min )
(V
IN(min ) - VOUT
VIN(min )
) = 5A
5V
´
7V
(7 V
- 5 V)
7V
= 2.26 A
(38)
I
´ 0.25
5 A ´ 0.25
DVIN = OUT
=
= 355 mV
CIN ´ fSW
8.8 mF ´ 400 kHz
(39)
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Table 2. Capacitor Types
VALUE (μF)
1 to 2.2
1 to 4.7
1
1 to 2.2
1 to 1.8
1 to 1.2
1 to 3.9
1 to 1.8
1 to 2.2
1.5 to 6.8
1 to 2.2
1 to 3.3
1 to 4.7
1
1 to 4.7
1 to 2.2
EIA Size
VOLTAGE
DIALECTRIC
100 V
1210
GRM32 series
50 V
100 V
1206
COMMENTS
GRM31 series
50 V
50 V
2220
100 V
VJ X7R series
50 V
2225
100 V
100 V
1812
X7R
C series C4532
50 V
100 V
1210
C series C3225
50 V
50 V
1210
100 V
X7R dielectric series
50 V
1812
100 V
8.2.2.7 Bootstrap Capacitor Selection
A 0.1-μF ceramic capacitor must be connected between the BOOT and SW terminals for proper operation. A
ceramic capacitor with X5R or better grade dielectric is recommended. The capacitor should have a 10 V or
higher voltage rating.
8.2.2.8 Undervoltage Lockout Set Point
The Undervoltage Lockout (UVLO) can be adjusted using an external voltage divider on the EN terminal of the
TPS54560B-Q1. The UVLO has two thresholds, one for power up when the input voltage is rising and one for
power down or brown outs when the input voltage is falling. For the example design, the supply should turn on
and start switching once the input voltage increases above 6.5 V (UVLO start). After the regulator starts
switching, it should continue to do so until the input voltage falls below 5 V (UVLO stop).
Programmable UVLO threshold voltages are set using the resistor divider of RUVLO1 and RUVLO2 between Vin and
ground connected to the EN terminal. Equation 4 and Equation 5 calculate the resistance values necessary. For
the example application, a 442 kΩ between VIN and EN (RUVLO1) and a 90.9 kΩ between EN and ground (RUVLO2)
are required to produce the 6.5 V and 5 V start and stop voltages.
(40)
RUVLO2 =
VENA
1.2 V
=
= 90.9 kW
VSTART - VENA
6.5 V - 1.2 V
+
m
1.2
A
+ I1
442 kW
RUVLO1
(41)
8.2.2.9 Output Voltage and Feedback Resistors Selection
The voltage divider of R5 and R6 sets the output voltage. For the example design, 10.2 kΩ was selected for R6.
Using Equation 3, R5 is calculated as 53.5 kΩ. The nearest standard 1% resistor is 53.6 kΩ. Due to the input
current of the FB terminal, the current flowing through the feedback network should be greater than 1 μA to
maintain the output voltage accuracy. This requirement is satisfied if the value of R6 is less than 800 kΩ.
Choosing higher resistor values decreases quiescent current and improves efficiency at low output currents but
may also introduce noise immunity problems.
V
- 0.8 V
æ 5 V - 0.8 V ö
= 10.2 kW x ç
RHS = RLS x OUT
÷ = 53.5 kW
0.8 V
0.8 V
è
ø
(42)
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8.2.2.10 Minimum Input Voltage, VIN
To ensure proper operation of the device and to keep the output voltage in regulation, the input voltage at the
device must be above the value calculated with Equation 43. Using the typical values for the RDS(on), Rdc and VF
in this application example, the minimum input voltage is 5.71 V. The BOOT-SW = 3 V curve in Figure 1 was
used for RHS = 0.12 Ω because the device will be operating with low drop out. When operating with low dropout,
the BOOT-SW voltage is regulated at a lower voltage because the BOOT-SW capacitor is not refreshed every
switching cycle. In the final application, the values of RDS(on), Rdc and VF used in this equation must include
tolerance of the component specifications and the variation of these specifications at their maximum operating
temperature in the application
V IN (min)
V IN (min)
Vo VF
R dc u Io
R DS(on) u Io V F
D
5 V 0.5 V 0.0113 : u 5 A
0.12 : u 5 A 0.5 V
0.99
5.71 V
(43)
8.2.2.11 Compensation
There are several methods to design compensation for DC-DC regulators. The method presented here is easy to
calculate and ignores the effects of the slope compensation that is internal to the device. Since the slope
compensation is ignored, the actual crossover frequency will be lower than the crossover frequency used in the
calculations. This method assumes the crossover frequency is between the modulator pole and the ESR zero
and the ESR zero is at least 10 times greater the modulator pole.
To get started, the modulator pole, ƒp(mod), and the ESR zero, ƒz1 must be calculated using Equation 44 and
Equation 45. For COUT, use a derated value of 87.4 μF. Use equations Equation 46 and Equation 47 to estimate
a starting point for the crossover frequency, ƒco. For the example design, ƒp(mod) is 1821 Hz and ƒz(mod) is 1100
kHz. Equation 45 is the geometric mean of the modulator pole and the ESR zero and Equation 47 is the mean of
modulator pole and half of the switching frequency. Equation 46 yields 44.6 kHz and Equation 47 gives 19.1 kHz.
Use the geometric mean value of Equation 46 and Equation 47 for an initial crossover frequency. For this
example, after lab measurement, the crossover frequency target was increased to 30 kHz for an improved
transient response.
Next, the compensation components are calculated. A resistor in series with a capacitor is used to create a
compensating zero. A capacitor in parallel to these two components forms the compensating pole.
IOUT(max )
5A
fP(mod) =
=
= 1821 Hz
2 ´ p ´ VOUT ´ COUT 2 ´ p ´ 5 V ´ 87.4 mF
(44)
f Z(mod) =
1
=
2 ´ p ´ RESR ´ COUT
fco1 =
fp(mod) x f z(mod) =
fco2 =
fp(mod) x
fSW
2
=
1
= 1100 kHz
2 ´ p ´ 1.67 mW ´ 87.4 mF
1821 Hz x 1100 kHz
1821 Hz x
400 kHz
2
= 44.6 kHz
(45)
(46)
= 19.1 kHz
(47)
To determine the compensation resistor, R4, use Equation 48. Assume the power stage transconductance,
gmps, is 17 A/V. The output voltage, VO, reference voltage, VREF, and amplifier transconductance, gmea, are 5
V, 0.8 V and 350 μA/V, respectively. R4 is calculated to be 16.8 kΩ and a standard value of 16.9 kΩ is selected.
Use Equation 49 to set the compensation zero to the modulator pole frequency. Equation 49 yields 5172 pF for
compensating capacitor C5. 4700 pF is used for this design.
ö
VOUT
æ 2 ´ p ´ fco ´ COUT ö æ
ö
5V
æ 2 ´ p ´ 29.2 kHz ´ 87.4 mF ö æ
R4 = ç
÷ = ç
÷ x ç
÷ x ç 0.8 V x 350 mA / V ÷ = 16.8 kW
gmps
V
x
gmea
17
A
/
V
è
ø è
ø
è
ø è REF
ø
(48)
1
1
C5 =
=
= 5172 pF
2 ´ p ´ R4 x fp(mod)
2 ´ p ´ 16.9 kW x 1821 Hz
(49)
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A compensation pole can be implemented if desired by adding capacitor C8 in parallel with the series
combination of R4 and C5. Use the larger value calculated from Equation 50 and Equation 51 for C8 to set the
compensation pole. The selected value of C8 is 47 pF for this design example.
C
x RESR
87.4 mF x 1.67 mW
=
= 8.64 pF
C8 = OUT
R4
16.9 kW
(50)
1
1
C8 =
=
= 47.1 pF
R4 x f sw x p
16.9 kW x 400 kHz x p
(51)
8.2.2.12 Discontinuous Conduction Mode and Eco-mode Boundary
With an input voltage of 12 V, the power supply enters discontinuous conduction mode when the output current
is less than 408 mA. The power supply enters Eco-mode when the output current is lower than 25.3 mA. The
input current draw is 257 μA with no load.
8.2.2.13 Power Dissipation Estimate
The following formulas show how to estimate the TPS54560B-Q1 power dissipation under continuous conduction
mode (CCM) operation. These equations should not be used if the device is operating in discontinuous
conduction mode (DCM).
The power dissipation of the IC includes conduction loss (PCOND), switching loss (PSW), gate drive loss (PGD) and
supply current (PQ). Example calculations are shown with the 12 V typical input voltage of the design example.
æV
ö
5V
2
PCOND = (IOUT ) ´ RDS(on ) ´ ç OUT ÷ = 5 A 2 ´ 92 mW ´
= 0.958 W
V
12
V
è IN ø
(52)
spacer
PSW = VIN ´ fSW ´ IOUT ´ trise = 12 V ´ 400 kHz ´ 5 A ´ 4.9 ns = 0.118 W
(53)
spacer
PGD = VIN ´ QG ´ fSW = 12 V ´ 3nC ´ 400 kHz = 0.014 W
(54)
spacer
PQ = VIN ´ IQ = 12 V ´ 146 mA = 0.0018 W
where
•
•
•
•
•
•
•
•
IOUT is the output current (A).
RDS(on) is the on-resistance of the high-side MOSFET (Ω)
VOUT is the output voltage (V).
VIN is the input voltage (V).
fsw is the switching frequency (Hz)
trise is the SW terminal voltage rise time and can be estimated by trise = VIN x 0.16 ns/V + 3 ns
QG is the total gate charge of the internal MOSFET
IQ is the operating nonswitching supply current
(55)
Therefore,
PTOT = PCOND + PSW + PGD + PQ = 0.958 W + 0.118 W + 0.014 W + 0.0018 W = 1.092 W
(56)
For given TA,
TJ = TA + RTH ´ PTOT
(57)
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For given TJMAX = 150°C
TA (max ) = TJ(max ) - RTH ´ PTOT
where
•
•
•
•
•
•
Ptot is the total device power dissipation (W)
TA is the ambient temperature (°C).
TJ is the junction temperature (°C).
RTH is the thermal resistance of the package (°C/W)
TJMAX is maximum junction temperature (°C)
TAMAX is maximum ambient temperature (°C).
(58)
There will be additional power losses in the regulator circuit due to the inductor ac and dc losses, the catch diode
and PCB trace resistance impacting the overall efficiency of the regulator.
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1 A/div
10 V/div
8.2.3 Application Curves
VIN
10 mV/div
200 mV/div
IOUT
VOUT ±5V offset
VOUT ±5V offset
Time = 4 ms/div
Time = 100 Ps/div
Figure 35. Line Transient (8 V to 40 V)
2 V/div
EN
4 V/div
4 V/div
VIN
VIN
1 V/div
5 V/div
5 V/div
Figure 34. Load Transient
VOUT
EN
VOUT
Time = 2 ms/div
Time = 2 ms/div
Figure 37. Start-up With EN
Figure 36. Start-up With VIN
10 V/div
500 mA/div
IL
SW
IL
10 mV/div
10 mV/div
1 A/div
10 V/div
SW
VOUT ± AC Coupled
VOUT ± AC Coupled
Time = 4 Ps/div
Time = 4 Ps/div
IOUT = 100 mA
Figure 38. Output Ripple CCM
32
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Figure 39. Output Ripple DCM
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10 V/div
1 m\A/div
IL
200 mV/div
10 V/div
IL
10 mV/div
200 mA/div
SW
SW
VOUT ± AC Coupled
VIN ± AC Coupled
Time = 1 ms/div
Time = 4 Ps/div
No Load
Figure 40. Output Ripple PSM
Figure 41. Input Ripple CCM
10 V/div
2 V/div
SW
SW
IL
VOUT
20 mV/div
10 mV/div
200 mA/div
500 mA/div
IL
VIN ± AC Coupled
Time = 4 Ps/div
Time = 40 Ps/div
IOUT = 100 mA
No Load
Figure 42. Input Ripple DCM
EN Floating
Figure 43. Low Dropout Operation
100
100
90
95
80
70
Efficiency (%)
Efficiency (%)
90
85
80
75
60
50
40
30
70
20
VIN =Series4
7V
VIN =36V
36 V
65
60
0
0.5
1
1.5
2
VIN =12V
12 V
VIN =24V
24 V
VIN =48V
48 V
VIN =60V
60 V
2.5
3
3.5
4
IO - Output Current (A)
VOUT = 5 V
4.5
VIN =7V
7V
VIN =36V
36 V
10
5
0
0.001
0.00
0.01
24VV
VIN = 24
VIN = 60
60VV
0.10
IO - Output Current (A)
C024
VOUT = 5 V
ƒsw = 400 kHz
VIN =12V
12 V
VIN = 48V
48 V
Figure 44. Efficiency vs Load Current
Product Folder Links: TPS54560B-Q1
C024
ƒsw = 400 kHz
Figure 45. Light Load Efficiency
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100
100
95
90
80
70
Efficiency (%)
85
80
75
60
50
VIN = 6V
6V
VIN =12V
12 V
VIN =24v
24 V
VIN =36v
36 V
VIN =48V
48 V
VIN =60V
60 V
40
30
70
20
VIN =6V
6V
VIN =36V
36 V
65
60
0
0.5
1
1.5
2
2.5
VIN =12V
12 V
VIN =24V
24 V
VIN =48V
48 V
VIN =60V
60 V
3
3.5
4
4.5
IO - Output Current (A)
VOUT = 3.3 V
10
0
0.001
0.00
5
0.01
ƒsw = 400 kHz
VOUT = 3.3 V
Figure 46. Efficiency vs Load Current
1.00
C024
ƒsw = 400 kHz
Figure 47. Light Load Efficiency
60
100
Gain (dB)
85
80
75
VIN 18in
= 18 V
VINSeries1
= 24 V
VINSeries3
= 36 V
VINSeries6
= 48 V
VINSeries8
= 60 V
70
65
60
0
0.5
1
1.5
2
2.5
3
3.5
4
4.5
IO - Output Current (A)
150
Phase
40
90
180
Gain
50
95
Efficiency (%)
0.10
IO - Output Current (A)
C024
120
30
90
20
60
10
30
0
0
±10
±30
±20
±60
±30
±90
VIN = 12 V
VOUT = 5 V
IOUT = 5 A
±40
±50
±120
±150
±60
±180
10
5
100
1k
10k
100k
1M
Frequency (Hz)
C024
VIN = 12 V
V = 12 V
Phase (ƒ)
Efficiency (%)
90
C001
VOUT = 5 V
IOUT = 5 A
Figure 49. Overall Loop Frequency Response
Figure 48. Efficiency vs Output Current
0.6
0.3
Output Voltage Normalized (%)
Output Voltage Normalized (%)
0.5
0.4
0.3
0.2
0.1
±0.0
±0.1
±0.2
±0.3
±0.4
0.2
0.1
0.0
±0.1
±0.2
±0.5
±0.6
0.0
0.5
1.0
1.5
2.0
2.5
3.0
3.5
4.0
IO - Output Current (A)
VIN = 12 V
VOUT = 5 V
5.0
±0.3
5
10
15
ƒsw = 400 kHz
20
25
30
35
40
45
50
VI - Input Voltage (V)
C024
Figure 50. Regulation vs Load Current
34
4.5
VOUT = 5 V
IOUT = 5 A
55
60
C024
ƒsw = 400 kHz
Figure 51. Regulation vs Input Voltage
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8.2.3.1 Safe Operating Area
90
90
80
80
70
70
60
60
TA (ƒC)
TA (ƒC)
The safe operating area (SOA) of the device is shown in Figure 52, through Figure 55 for 3.3 V, 5 V and 12 V
outputs and varying amounts of forced air flow. The temperature derating curves represent the conditions at
which the internal and external components are at or below the manufacturer’s maximum operating
temperatures. Derating limits apply to devices soldered directly to a double-sided PCB with 2 oz. copper, similar
to the EVM. Careful attention must be paid to the other components chosen for the design, especially the catch
diode. In most of these test conditions, the thermal performance is limited by the catch diode. When operating at
high duty cycles or at higher switching frequency the TPS54560B-Q1 thermal performance can become the
limiting factor.
50
6V
12 V
24 V
36 V
48 V
60 V
40
30
20
0.0
0.5
50
8V
12 V
24 V
36 V
48 V
60 V
40
30
20
1.0
1.5
2.0
2.5
3.0
3.5
4.0
4.5
IOUT (Amps)
5.0
0.0
0.5
1.0
80
80
70
70
60
60
TA (ƒC)
TA (ƒC)
90
50
18 V
24 V
36 V
48 V
60 V
20
0.0
0.5
1.0
2.5
3.0
3.5
4.0
4.5
5.0
C048
Figure 53. 5 V Outputs
90
30
2.0
IOUT (Amps)
Figure 52. 3.3 V Outputs
40
1.5
C047
50
400 LFM
40
200 LFM
30
100 LFM
Nat Conv
20
1.5
2.0
2.5
3.0
3.5
IOUT (Amps)
ƒsw = 800 kHz
4.0
4.5
5.0
0.0
0.5
1.0
1.5
2.0
2.5
3.0
3.5
4.0
4.5
IOUT (Amps)
C048
5.0
C048
ƒsw = 800 kHz
Figure 54. 12 V Outputs
Figure 55. Air Flow Conditions
VIN = 36 V, VO = 12 V
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8.3 Inverting Power
The TPS54560B-Q1can be used to convert a positive input voltage to a negative output voltage. Idea
applications are amplifiers requiring a negative power supply. For a more detailed example see SLVA317.
VIN
+
Cin
Cboot
Lo
VIN
Cd
BOOT
GND
SW
R1
+
GND
Co
R2
TPS54560B-Q1
FB
VOUT
EN
COMP
Rcomp
RT/CLK
Czero
RT
Cpole
Copyright © 2016, Texas Instruments Incorporated
Figure 56. TPS54560B-Q1 Inverting Power Supply from SLVA317 Application Note
8.4 Split Rail Power Supply
The TPS54560B-Q1 can be used to convert a positive input voltage to a split rail positive and negative output
voltage by using a coupled inductor. Idea applications are amplifiers requiring a split rail positive and negative
voltage power supply. For a more detailed example see SLVA369.
VOPOS
+
VIN
Copos
+
Cin
VIN
Cboot
BOOT
GND
SW
Lo
Cd
R1
GND
+
Coneg
R2
TPS54560B-Q1
VONEG
FB
EN
COMP
Rcomp
RT/CLK
RT
Czero
Cpole
Copyright © 2016, Texas Instruments Incorporated
Figure 57. TPS54560B-Q1 Split Rail Power Supply
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9 Power Supply Recommendation
The devices are designed to operate from an input voltage supply range between 4.5 V and 60 V. If the input
supply is located more than a few inches from the TPS54560B-Q1 converter additional bulk capacitance may be
required in addition to the ceramic bypass capacitors. An electrolytic capacitor with a value of 100 μF is a typical
choice.
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10 Layout
10.1 Layout Guidelines
Layout is a critical portion of good power supply design. There are several signal paths that conduct fast
changing currents or voltages that can interact with stray inductance or parasitic capacitance to generate noise
or degrade performance.
• To reduce parasitic effects, the VIN terminal should be bypassed to ground with a low ESR ceramic bypass
capacitor with X5R or X7R dielectric.
• Care should be taken to minimize the loop area formed by the bypass capacitor connections, the VIN
terminal, and the anode of the catch diode.
• The GND terminal should be tied directly to the power pad under the IC and the PowerPAD™.
• The PowerPAD™ should be connected to internal PCB ground planes using multiple vias directly under the
IC.
• The SW terminal should be routed to the cathode of the catch diode and to the output inductor.
• Since the SW connection is the switching node, the catch diode and output inductor should be located close
to the SW terminals, and the area of the PCB conductor minimized to prevent excessive capacitive coupling.
• For operation at full rated load, the top side ground area must provide adequate heat dissipating area.
• The RT/CLK terminal is sensitive to noise so the RT resistor should be located as close as possible to the IC
and routed with minimal lengths of trace.
• The additional external components can be placed approximately as shown.
• It may be possible to obtain acceptable performance with alternate PCB layouts, however this layout has
been shown to produce good results and is meant as a guideline.
10.2 Layout Examples
Vout
Output
Capacitor
Topside
Ground
Area
Input
Bypass
Capacitor
Vin
UVLO
Adjust
Resistors
Output
Inductor
Route Boot Capacitor
Trace on another layer to
provide wide path for
topside ground
BOOT
Catch
Diode
SW
VIN
GND
EN
COMP
RT/CLK
Frequency
Set Resistor
FB
Compensation
Network
Resistor
Divider
Thermal VIA
Signal VIA
Figure 58. PCB Layout Example
10.2.1 Estimated Circuit Area
Boxing in the components in the design of Figure 33 the estimated printed circuit board area is 1.025 in2 (661
mm2). This area does not include test points or connectors.
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11 Device and Documentation Support
11.1 Device Support
11.1.1 Custom Design with WEBENCH® Tools
Click here to create a custom design using the TPS54560B-Q1 device with the WEBENCH® Power Designer.
1. Start by entering your VIN, VOUT, and IOUT requirements.
2. Optimize your design for key parameters like efficiency, footprint and cost using the optimizer dial and
compare this design with other possible solutions from Texas Instruments.
3. The WEBENCH Power Designer provides you with a customized schematic along with a list of materials with
real time pricing and component availability.
4. In most cases, you will also be able to:
– Run electrical simulations to see important waveforms and circuit performance
– Run thermal simulations to understand the thermal performance of your board
– Export your customized schematic and layout into popular CAD formats
– Print PDF reports for the design, and share your design with colleagues
5. Get more information about WEBENCH tools at www.ti.com/WEBENCH.
11.2 Receiving Notification of Documentation Updates
To receive notification of documentation updates, navigate to the device product folder on ti.com. In the upper
right corner, click on Alert me to register and receive a weekly digest of any product information that has
changed. For change details, review the revision history included in any revised document.
11.3 Community Resources
The following links connect to TI community resources. Linked contents are provided "AS IS" by the respective
contributors. They do not constitute TI specifications and do not necessarily reflect TI's views; see TI's Terms of
Use.
TI E2E™ Online Community TI's Engineer-to-Engineer (E2E) Community. Created to foster collaboration
among engineers. At e2e.ti.com, you can ask questions, share knowledge, explore ideas and help
solve problems with fellow engineers.
Design Support TI's Design Support Quickly find helpful E2E forums along with design support tools and
contact information for technical support.
11.4 Trademarks
Eco-mode, PowerPAD, E2E are trademarks of Texas Instruments.
WEBENCH is a registered trademark of Texas Instruments.
11.5 Electrostatic Discharge Caution
These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
11.6 Glossary
SLYZ022 — TI Glossary.
This glossary lists and explains terms, acronyms, and definitions.
12 Mechanical, Packaging, and Orderable Information
The following pages include mechanical packaging and orderable information. This information is the most
current data available for the designated devices. This data is subject to change without notice and revision of
this document. For browser-based versions of this data sheet, refer to the left-hand navigation.
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PACKAGE OPTION ADDENDUM
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10-Dec-2020
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
(4/5)
(6)
TPS54560BQDDAQ1
ACTIVE SO PowerPAD
DDA
8
75
RoHS & Green
NIPDAUAG
Level-2-260C-1 YEAR
-40 to 125
5456BQ
TPS54560BQDDARQ1
ACTIVE SO PowerPAD
DDA
8
2500
RoHS & Green
NIPDAUAG
Level-2-260C-1 YEAR
-40 to 125
5456BQ
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of
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