TPS55383,, TPS55386
www.ti.com ......................................................................................................................................................................................... SLUS818 – SEPTEMBER 2008
3-A DUAL NON-SYNCHRONOUS CONVERTER WITH INTEGRATED HIGH-SIDE MOSFET
AND EXTERNAL COMPENSATION
FEATURES
CONTENTS
1
•
•
•
•
23
•
•
•
•
•
•
•
•
•
•
•
4.5-V to 28-V Input Range
Output Voltage 0.8 V to 90% of Input Voltage
Output Current Up to 3 A
Two Fixed Switching Frequency Versions:
– TPS55383: 300 kHz
– TPS55386: 600 kHz
Three Selectable Levels of Overcurrent
Protection (Output 2)
0.8-V 1.75% Voltage Reference
2.1-ms Internal Soft Start
Dual PWM Outputs 180° Out-of-Phase
Ratiometric or Sequential Startup Modes
Configurable as Dual Output or Two-Channel
Single Output Multiphase for 6 amp Capability
85-mΩ Internal High-Side MOSFETs
Current Mode Control with External
Compensation
Pulse-by-Pulse Overcurrent Protection
Thermal Shutdown Protection at +148°C
16-Pin PowerPAD™ HTSSOP package
Device Ratings
Electrical Characteristics
4
Device Information
10
Application Information
13
Design Examples
30
Additional References
39
DESCRIPTION
The TPS55383 and TPS55386 are dual output,
non-synchronous buck converters capable of
supporting 3-A output applications that operate from a
4.5-V to 28-V input supply voltage, and require output
voltages between 0.8 V and 90% of the input voltage.
With an internally-determined operating frequency
and soft start time, these converters provide many
features with a minimum of external components. The
outputs of the two error amplifiers are accessible
allowing user optimization of the feedback loop under
a wide range of output filter characteristics.
Channel 1 overcurrent protection is set at 4.5 A, while
Channel 2 overcurrent protection level is selected by
connecting a pin to ground, to BP, or left floating. The
setting levels are used to allow for scaling of external
components for applications that do not need the full
load capability of both outputs.
APPLICATIONS
•
•
•
•
2
Set Top Box
Digital TV
Power for DSP
Consumer Electronics
The outputs may be enabled independently, or
configured to allow either ratiometric or sequential
startup sequencing. Additionally, the two outputs may
be powered from different sources.
VIN
TPS55383
1
PVDD1
PVDD2 16
2
BOOT1
BOOT2 15
3
SW1
OUTPUT1
OUTPUT2
SW2 14
4
GND
BP 13
5
EN1
SEQ 12
6
EN2
ILIM2 11
7
FB1
8
COMP1
FB2 10
COMP2
9
UDG-08045
1
2
3
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas
Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
PowerPAD is a trademark of Texas Instruments.
All other trademarks are the property of their respective owners.
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of the Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
Copyright © 2008, Texas Instruments Incorporated
TPS55383,, TPS55386
SLUS818 – SEPTEMBER 2008 ......................................................................................................................................................................................... www.ti.com
These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
ORDERING INFORMATION (1)
DEVICE NUMBER
OPERATING FREQUENCY (kHz)
PACKAGE
TPS55383PWP
MEDIA
UNITS (Pieces)
Tube
90
Tape and Reel
2000
300
TPS55383PWPR
Plastic 16-Pin HTSSOP
TPS55386PWP
Tube
90
Tape and Reel
2000
600
TPS55386PWPR
(1)
For the most current package and ordering information see the Package Option Addendum at the end of this document, or see the TI
web site at www.ti.com.
DEVICE RATINGS
ABSOLUTE MAXIMUM RATINGS (1)
VALUE
PVDD1, PVDD2, EN1, EN2
Input voltage range
BOOT1, BOOT2
VSW+ 7
SW1, SW2
–2 to 30
SW1, SW2 transient (< 50ns)
–3 to 31
BP
V
6.5
SEQ, ILIM2
–0.3 to 6.5
COMP1, COMP2
–0.3 to 3.5
FB1, FB2
–0.3 to 3
SW1, SW2 output current
7
A
BP load current
35
mA
Tstg
Storage temperature
–55 to +165
TJ
Operating temperature
–40 to +150
Soldering temperature
+260
(1)
UNIT
30
°C
Permanent device damage may occur if Absolute Maximum Ratings are exceeded. Functional operation should be limited to the
Recommended DC Operating Conditions detailed in this data sheet. Exposure to conditions beyond the operational limits for extended
periods of time may affect device reliability.
RECOMMENDED OPERATING CONDITIONS
MIN
MAX
UNIT
VPVDD2
Input voltage
4.5
28
V
TJ
Operating junction
temperature
–40
+125
°C
2
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ELECTROSTATIC DISCHARGE (ESD) PROTECTION
MIN
Human body model
UNIT
2k
CDM
1.5k
Machine Model
250
V
PACKAGE DISSIPATION RATINGS (1) (2) (3)
(1)
(2)
(3)
(4)
PACKAGE
THERMAL IMPEDANCE
JUNCTION-TO-THERMAL
PAD (°C/W)
TA = +25°C
POWER RATING (W)
TA = +85°C
POWER RATING (W)
Plastic 16-Pin HTSSOP (PWP)
2.07 (4)
1.6
1.0
For more information on the PWP package, refer to TI Technical Brief (SLMA002A).
TI device packages are modeled and tested for thermal performance using printed circuit board designs outlined in JEDEC standards
JESD 51-3 and JESD 51-7.
For application information, see the Power Derating section.
TJ-A = +40°C/W.
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ELECTRICAL CHARACTERISTICS
–40°C ≤ TJ ≤ +125°C, VPVDD1 = VPVDD2 = 12 V, unless otherwise noted.
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
INPUT SUPPLY (PVDD)
VPVDD1
Input voltage range
VPVDD2
4.5
28
V
µA
IDDSDN
Shutdown
VEN1 = VEN2 = VPVDD2
70
150
IDDQ
Quiescent, non-switching
VFB = 0.9 V, Outputs OFF
1.8
3.0
IDDSW
Quiescent, while-switching
SW node unloaded; Measured as BP sink
current
VUVLO
Minimum turn-on voltage
PVDD2 only
VUVLO(hys)
Hysteresis
tSTART (1) (2)
Time from startup to softstart begin
mA
5
3.8
CBP = 10 µF, EN1 and EN2 go low
simultaneously
4.1
4.4
V
400
600
mV
2
ms
ENABLE (EN)
VEN1, VEN2
Enable threshold
0.9
Enable threshold hysteresis (1)
1.2
1.5
50
IEN1. IEN2
Enable pull-up current
VEN1 = VEN2 = 0 V
tEN (1)
Time from enable to soft-start begin
Other EN pin = GND
6
V
mV
12
µA
µs
10
BP REGULATOR (BP)
BP
Regulator voltage
8 V < PVDD2 < 28 V
BPLDO
Dropout voltage
PVDD2 = 4.5 V; switching, no external load on
BP
5
IBP (1)
Regulator external load
IBPS
Regulator short circuit
5.25
5.6
V
400
550
mV
2
4.5 V < PVDD2 < 28 V
10
20
30
TPS55383
255
310
375
TPS55386
510
630
750
mA
OSCILLATOR
fSW
Switching frequency
tDEAD (1)
Clock dead time
140
kHz
ns
ERROR AMPLIFIER (EA) and VOLTAGE REFERENCE (REF)
VFB1, VFB2
Feedback input voltage
IFB1, IFB2
Feedback input bias current
gM1, gM2 (1)
Error Amplifier transconductance
fp1, fp2 (1)
Error Amplifier dominant pole frequency
ISINK(COMP1),
ISINK(COMP2)
Error Amplifier sink current capability
ISRC(COMP1),
ISRC(COMP2)
Error Amplifier source current capability
0°C < TJ < +85°C
786
–40°C < TJ < +125°C
784
800
812
812
mV
3
50
nA
220
315
420
µS
5
6
VFB1 = VFB2 = 0.9V, VCOMP = 2 V
15
30
40
µA
VFB1 = VFB2 = 0.7V, VCOMP = 0 V
15
30
40
µA
1.5
2.1
2.7
ms
kHz
SOFT START (SS)
TSS1, TSS2
(1)
(2)
4
Soft start time
Ensured by design. Not production tested.
When both outputs are started simultaneously, a 20-mA current source charges the BP capacitor. Faster times are possible with a lower
BP capacitor value. More information can be found in the Input UVLO and Startup section.
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ELECTRICAL CHARACTERISTICS (continued)
–40°C ≤ TJ ≤ +125°C, VPVDD1 = VPVDD2 = 12 V, unless otherwise noted.
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
OVERCURRENT PROTECTION
ICL1
Current limit Channel 1
ICL2
Current limit Channel 2
3.6
4.5
5.6
VILIM2 = VBP
3.6
4.5
5.6
VILIM2 = (floating)
2.4
3.0
3.6
1.15
1.50
1.75
670
730
VILIM2 = GND
VUV1
Low-level output threshold to declare a fault
VUV2
THICCUP
tON1(oc)
(3)
Measured at feedback pin.
Hiccup timeout
10
Minimum overcurrent pulse width
90
A
mV
ms
(3)
tON2(oc) (3)
150
ns
BOOTSTRAP
RBOOT1,
RBOOT2
Bootstrap switch resistance
From BP to BOOT1 or BP to BOOT2,
IEXT = 50 mA
18
TJ = +25°C, VPVDD2 = 8 V
85
–40°C < TJ < +125°C, VPVDD2 = 8 V
85
165
100
200
ns
0
%
Ω
OUTPUT STAGE (Channel 1 and Channel 2)
RDS(on) (3)
MOSFET on resistance plus bond wire resistance
tON(min) (3)
Minimum controllable pulse width
ISWx peak current > 1 A (4)
DMIN
Minimum Duty Cycle
VFB = 0.9 V
DMAX
Maximum Duty Cycle
ISW
Switching node leakage current (sourcing)
TPS55383 fSW = 300 kHz
90
95
TPS55386 fSW = 600 kHz
85
90
Outputs OFF
2
mΩ
%
%
12
µA
THERMAL SHUTDOWN
TSD (3)
Shutdown temperature
TSD(hys) (3)
Hysteresis
(3)
(4)
148
20
°C
Ensured by design. Not production tested.
See Figure 14 for ISWx peak current
enable threshold voltage
Active
Tie EN1 to > enable threshold voltage
for low quiescent current (BP inactive)
when VEN2 > enable threshold voltage
Ignored by the device.when VEN1 <
enable threshold voltage
GND
Sequential, Output 1 then Output 2
Tie EN2 to < enable threshold voltage
for BP to be active when VEN1 >
enable threshold voltage
Active
Tie EN2 to > enable threshold voltage
for low quiescent current (BP inactive)
when VEN1 > enable threshold voltage
(floating)
Independent or Ratiometric, Output 1
and Output 2
Active. EN1 and EN2 must be tied
together for Ratio-metric startup.
Active. EN1 and EN2 must be tied
together for Ratio-metric startup.
If the SEQ pin is connected to BP, then when Output 2 is enabled, Output 1 is allowed to start approximately 400
µs after Output 2 has reached regulation; that is, sequential startup where Output 1 is slave to Output 2. If EN2 is
allowed to go high after the outputs have been operating, then both outputs are disabled immediately, and the
output voltages decay according to the load that is present.
If the SEQ pin is connected to GND, then when Output 1 is enabled, Output 2 is allowed to start approximately
400 µs after Output 1 has reached regulation; that is, sequential startup where Output 2 is slave to Output 1. If
EN1 is allowed to go high after the outputs have been operating, then both outputs are disabled immediately,
and the output voltages decay according to the load that is present.
SEQ = BP
Sequential
CH2 then CH1
SEQ = GND
Sequential
CH1 then CH2
5-V VOUT1
(2 V/div)
5-V VOUT1
(2 V/div)
3.3-V VOUT2
(2 V/div)
3.3-V VOUT2
(2 V/div)
T - Time - 1 ms/div
T - Time - 1 ms/div
Figure 18. SEQ Pin TIed to BP
Figure 19. SEQ Pin Tied to GND
NOTE:
An R-C network connected to the ENx pin may be used in addition to the SEQ pin in
sequential mode to delay the startup of the first output voltage. This approach may be
necessary in systems with a large number of output voltages and elaborate voltage
sequencing requirements. See Enable and Timed Turn On of the Outputs.
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If the SEQ pin is left floating, Output 1 and Output 2 each start ratiometrically when both outputs are enabled at
the same time. Output 1 and Output 2 soft start at a rate that is determined by the respective final output
voltages and enter regulation at the same time. If the EN1 and EN2 pins are allowed to operate independently,
then the two outputs also operate independently.
5-V VOUT1
(2 V/div)
3.3-V VOUT2
(2 V/div)
T - Time - 1 ms/div
Figure 20. SEQ Pin Floating
Soft Start
Each output has a dedicated soft-start circuit. The soft-start voltage is an internal digital reference ramp to one of
two noninverting inputs of the error amplifier. The other input is the (internal) precision 0.8-V reference. The total
ramp time for the FB voltage to charge from 0 V to 0.8 V is about 2.1 ms. During a soft-start interval, the
TPS5538x output slowly increases the voltage to the noninverting input of the error amplifier. In this way, the
output voltage ramps up slowly until the voltage on the noninverting input to the error amplifier reaches the
internal 0.8-V reference voltage. At that time, the voltage at the noninverting input to the error amplifier remains
at the reference voltage.
During the soft-start interval, pulse-by-pulse current limiting is in effect. If an overcurrent pulse is detected, six
PWM pulses are skipped to allow the inductor current to decay before another PWM pulse is applied. (See the
Output Overload Protection section.) There is no pulse skipping if a current limit pulse is not detected.
DESIGN HINT
If the rate of rise of the input voltage (PVDDx) is such that the input voltage is too low
to support the desired regulation voltage by the time soft-start has completed, then
the output UV circuit may trip and cause a hiccup in the output voltage. In this case,
use a timed delay startup from the ENx pin to delay the startup of the output until the
PVDDx voltage has the capability of supporting the desired regulation voltage. See
Operating Near Maximum Duty Cycle and Maximum Output Capacitance for related
information.
16
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Output Voltage Regulation
Each output has a dedicated feedback loop comprised of a voltage setting divider, an error amplifier, a pulse
width modulator, and a switching MOSFET. The regulation output voltage is determined by a resistor divider
connecting the output node, the FBx pin, and GND (see Figure 21). Assuming the value of the upper voltage
setting divider is known, the value of the lower divider resistor for a desired output voltage is calculated by
Equation 2.
VREF
R2 = R1´
VOUT - VREF
(2)
where
•
VREF is the internal 0.8-V reference voltage
TPS5538x
1
PVDD1
PVDD2 16
2
BOOT1
BOOT2 15
3
SW1
SW2 14
4
GND
BP 13
5
EN1
SEQ 12
6
EN2
ILIM2 11
7
FB1
FB2 10
8
COMP1
OUTPUT1
R1
R2
COMP2
9
UDG-08041
Figure 21. Voltage Setting Divider Network for Channel 1
DESIGN HINT
There is a leakage current of up to 12 µA out of the SW pin when a single output of
the TPS5538x is disabled. Keeping the series impedance of R1 + R2 less than 50 kΩ
prevents the output from floating above the referece voltage while the controller output
is in the OFF state.
Feedback Loop Compensation Component Selection
In the feedback signal path, the output voltage setting divider is followed by an internal gM-type error amplifier
with a typical transconductance of 315 µS. An external series connected R-C circuit from the gM amplifier output
(COMPx pin) to ground serves as the compensation network for the converter. The signal from the error amplifier
output is then buffered and combined with a slope compensation signal before it is mirrored to be referenced to
the SW node. Here, it is compared with the current feedback signal to create a pulse-width-modulated (PWM)
signal-fed to drive the upper MOSFET switch. A simplified equivalent circuit of the signal control path is depicted
in Figure 22.
NOTE:
Noise coupling from the SWx node to internal circuitry of BOOTx may impact narrow
pulse width operation, especially at load currents less than 1 A. See SW Node
Ringing for further information on reducing noise on the SWx node.
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BOOT
TPS5538x
ICOMP – ISLOPE
FB
0.8 VREF
PWM to
Switch
x2
Error Amplifier
ISLOPE
+
+
ICOMP
Offset
f(IDRAIN)
COMP
SW
11.5 kW
RCOMP
CCOMP
UDG-08040
Figure 22. Feedback Loop Equivalent Circuit
A more conventional small-signal equivalent block diagram is shown in Figure 23. Here, the full closed-loop
signal path is shown. Because the TPS5538x contains internal slope compensation, the external L-C filter must
be selected appropriately so that the resulting control loop meets criteria for stability.
VIN
VC
+
VOUT
+
Modulator
VREF
_
_
Filter
Current
Feedback
Network
Compensation
Network
Figure 23. Small Signal Equivalent Block Diagram
18
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Inductor Selection
Calculate the inductance value so that an output ripple current between 300 mA and 900 mA results. Lower
ripple current results in discontinuous mode (DCM) operation at a lower DC load current, while higher ripple
current generally allows for higher closed loop bandwidth.
V -V
L = IN OUT
DIOUT
(3)
NOTE:
For wide input range converters, highest input voltage results in the highest ripple
current.
NOTE:
The load current at which the overcurrent protection (OCP) engages is dependent on
the amount of ripple current, because it is the peak current in the switch that is
monitored. See Output Overload Protection.
Maximum Output Capacitance
With internal pulse-by-pulse current limiting and a fixed soft-start time, there is a maximum output capacitance
which may be used before startup problems begin to occur. If the output capacitance is large enough so that the
device enters a current-limit protection mode during startup, then there is a possibility that the output never
reaches regulation. Instead, the TPS5538x simply shuts down and attempts a restart as if the output were
short-circuited to ground. The maximum output capacitance (including bypass capacitance distributed at the
load) is given by Equation 4:
COUT(max ) =
tSS
VOUT
æ
ö
æ1
ö
ç ICLx - ç ´ IRIPPLE ÷ - ILOAD ÷
2
è
ø
è
ø
(4)
Minimum Output Capacitance
Ensure the value of capacitance selected for closed-loop stability is compatible with the requirements of Soft
Start.
Compensation For The Feedback Loop
To determine the components necessary for compensating the feedback loop, the controller frequency response
characteristics must be understood and the desired crossover frequency selected. The best results are obtained
if 10% of the switching frequency is used as this closed loop crossover frequency. In some cases, up to 20% of
the switching frequency is also possible.
With the output filter components selected, the next step is to calculate the DC gain of the modulator. For the
TPS55386:
FmTPS55386 =
600000
é
ù
1.5 ´ 106 ´ t ON
ê
-6 ´ æ VIN - VOUT ö ú
´
+
´
19.7
e
50
10
çç
÷÷ ú
ê
L
è
øú
ê
ë
û
(
)
(5)
The gain of the TPS55383 modulator is approximated by:
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300000
FmTPS55383 =
é
5.6 ´ 105 ´ tON
æ V - VOUT
ê
+ 50 ´ 10-6 ´ ç IN
ê19.7 ´ e
ç
L
è
ê
ë
(
)
ù
öú
÷÷ ú
øú
û
(6)
The overall DC gain of the of the converter control-to-output transfer function is approximated by:
fc =
VIN ´ Fm ´ 2 ´ 10
-4
æ æ V ´ Fm ´ 50 ´ 106 ö ö
ç 1 + ç IN
÷÷
RLOAD
ç ç
÷÷
è è
øø
(7)
The next step is to find the desired gain of the error amplifier at the desired crossover frequency. Assuming a
single pole roll off, evaluate the following expression at the desired crossover frequency.
æ
ö
fc
KEA = -20 ´ log ç
÷
è 1 + 2p ´ fCO ´ RLOAD ´ COUT ø
(8)
TPS5538x
Output1
COUT
L
ZUPPER
C1
(optional)
C2
(optional)
R1
R2
CCOMP
1
PVDD1
PVDD2 16
2
BOOT1
BOOT2 15
3
SW1
SW2 14
4
GND
BP 13
5
EN1
SEQ 12
6
EN2
ILIM2 11
7
FB1
FB2 10
8
COMP1
RCOMP
COMP2
9
ZLOWER
UDG-08042
Figure 24. Loop Compensation Components
If operating at wide duty cycles (over 50%), a capacitor may be necessary across the upper resistor of the
voltage setting divider. (Ref Figure 24) If duty cycles are less than 50%, this capacitor may be omitted.
C1 =
L ´ COUT
R1
(9)
If a high ESR capacitor is used in the output filter, a zero appears in the loop response that could lead to
instability. To compensate, a small capacitor is placed in parallel with the lower voltage setting divider resistor
(Ref Figure 24). The value of the capacitor is determined such that a pole is placed at the same frequency as the
ESR zero. If low ESR capacitors are used, this capacitor may be omitted.
20
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C2 = C OUT ´
R ESR ´ (R2 + R1)
R2 ´ R1
(10)
Next, calculate the value of the error amplifier gain setting resistor and capacitor.
KEA
10 20 ´ (ZLOWER + ZUPPER )
RCOMP =
gM ´ ZLOWER
CCOMP =
(11)
1
2p ´ fPOLE ´ RCOMP
(12)
where
fPOLE =
1
2p ´ RLOAD ´ COUT
(13)
NOTE:
Once the filter and compensation component values have been established,
laboratory measurements of the physical design should be performed to confirm
converter stability.
Bootstrap for the N-Channel MOSFET
A bootstrap circuit provides a voltage source higher than the input voltage and of sufficient energy to fully
enhance the switching MOSFET each switching cycle. The PWM duty cycle is limited to a maximum of 90%,
allowing an external bootstrap capacitor to charge through an internal synchronous switch (between BP and
BOOTx) during every cycle. When the PWM switch is commanded to turn ON, the energy used to drive the
MOSFET gate is derived from the voltage on this capacitor.
To allow the bootstrap capacitor to charge each switching cycle, an internal pulldown MOSFET (from SW to
GND) is turned ON for approximately 140 ns at the beginning of each switching cycle. In this way, if, during light
load operation, there is insufficient energy for the SW node to drive to ground naturally, this MOSFET forces the
SW node toward ground and allow the bootstrap capacitor to charge.
Because this is a charge transfer circuit, care must be taken in selecting the value of the bootstrap capacitor. It
must be sized such that the energy stored in the capacitor on a per cycle basis is greater than the gate charge
requirement of the MOSFET being used.
DESIGN HINT
For the bootstrap capacitor, use a ceramic capacitor with a value between 22 nF and
82 nF.
NOTE:
For 5-V input applications, connect PVDDx to BP directly. This connection bypasses
the internal control circuit regulator and provides maximum voltage to the gate drive
circuitry. In this configuration, shutdown mode IDDSDNis the same as quiescent IDDQ.
Operating Near Maximum Duty Cycle
If the TPS5538x operates at maximum duty cycle, and if the input voltage is insufficient to support the output
voltage (at full load or during a load current transient), then there is a possibility that the output voltage will fall
from regulation and trip the output UV comparator. If this should occur, the TPS5538x protection circuitry
declares a fault and enter a shut down-and-restart cycle.
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DESIGN HINT
Ensure that under ALL conditions of line and load regulation, there is sufficient duty
cycle to maintain output voltage regulation.
To calculate the operating duty cycle, use Equation 14.
d=
VOUT + VDIODE
VIN + VDIODE
(14)
where
•
VDIODE is the forward voltage drop of the rectifier diode
Light Load Operation
There is no special circuitry for pulse skipping at light loads. The normal characteristic of a nonsynchronous
converter is to operate in the discontinuous conduction mode (DCM) at an average load current less than
one-half of the inductor peak-to-peak ripple current. Note that the amplitude of the ripple current is a function of
input voltage, output voltage, inductor value, and operating frequency, as shown in Equation 15.
1 VIN - VOUT
IDCM = ´
´ d ´ TS
L
2
(15)
During discontinuous mode operation the commanded pulse width may become narrower than the capability of
the converter to resolve. To maintain the output voltage within regulation, skipping switching pulses at light load
conditions is a natural by-product of that mode. This condition may occur if the output capacitor is charged to a
value greater than the output regulation voltage and there is insufficient load to discharge the capacitor. A
by-product of pulse skipping is an increase in the peak-to-peak output ripple voltage.
SW Waveform
SW Waveform
VOUT
Ripple
VOUT
Ripple
Skipping
VIN = 12 V
VOUT = 5 V
Inductor
Current
Steady State
VIN = 12 V
VOUT = 5 V
Inductor
Current
Figure 25. Steady State
Figure 26. Skipping
DESIGN HINT
If additional output capacitance is required to reduce the output voltage ripple during
DCM operation, be sure to recheck the Maximum Output Capacitance section.
SW Node Ringing
A portion of the control circuitry is referenced to the SW node. To ensure jitter-free operation, it is necessary to
decrease the voltage waveform ringing at the SW node to less than 5-V peak and of a duration of less than
30-ns. In addition to following good printed circuit board (PCB) layout practices, there are a couple of design
techniques for reducing ringing and noise.
22
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SW Node Snubber
Voltage ringing at the SW node is caused by fast switching edges and parasitic inductance and capacitance. If
the ringing results in excessive voltage on the SW node, or erratic operation of the converter, an R-C snubber
may be used to dampen the ringing and ensure proper operation over the full load range.
DESIGN HINT
A series-connected R-C snubber (C = between 330 pF and 1 nF, R = 10 Ω)
connected from SW to GND reduces the ringing on the SW node.
Bootstrap Resistor
A small resistor in series with the bootstrap capacitor reduces the turn-on time of the internal MOSFET, thereby
reducing the rising edge ringing of the SW node.
DESIGN HINT
A resistor with a value between 1 Ω and 3 Ω may be placed in series with the
bootstrap capacitor to reduce ringing on the SW node.
DESIGN HINT
Placeholders for these components should be placed on the initial prototype PCBs in
case they are needed.
Output Overload Protection
In the event of an overcurrent during soft-start on either output (such as starting into an output short),
pulse-by-pulse current limiting and PWM frequency division are in effect for that output until the internal soft-start
timer ends. At the end of the soft-start time, a UV fault is declared. During this fault, both PWM outputs are
disabled and the small pulldown MOSFETs (from SWx to GND) are turned ON. This process ensures that both
outputs discharge to GND in the event that overcurrent is on one output while the other is not loaded. The
converter then enters a hiccup mode timeout before attempting to restart. Frequency Division describes a
condition when an overcurrent pulse is detected and six clock cycles are skipped before a next PWM pulse is
initiated, effectively dividing the operating frequency by six and preventing excessive current build up in the
inductor.
In the event of an overcurrent condition on either output after the output reaches regulation, pulse-by-pulse
current limit is in effect for that output. In addition, an output undervoltage (UV) comparator monitors the FBx
voltage (that follows the output voltage) to declare a fault if the output drops below 85% of regulation. During this
fault condition, both PWM outputs are disabled and the small pulldown MOSFETs (from SWx to GND) are turned
ON. This design ensures that both outputs discharge to GND, in the event that overcurrent is on one output while
the other is not loaded. The converter then enters a hiccup mode timeout before attempting to restart.
The overcurrent threshold for Output 1 is set nominally at 4.5 A. The overcurrent level of Output 2 is determined
by the state of the ILIM2 pin. The ILIM setting of Output 2 is not latched in place and may be changed during
operation of the converter.
Table 2. Current Limit Threshold Adjustment for
Output 2
ILIM2 Connection
OCP Threshold for Output 2
BP
4.5 A nominal setting
(floating)
3.0 A nominal setting
GND
1.5 A nominal setting
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DESIGN HINT
The OCP threshold refers to the peak current in the internal switch. Be sure to add
one-half of the peak inductor ripple current to the dc load current in determining how
close the actual operating point is to the OCP threshold.
Dual Supply Operation
It is possible to operate a TPS5538x from two supply voltages. If this application is desired, then the sequencing
of the supplies must be such that PVDD2 is above the UVLO voltage before PVDD1 begins to rise. This level
requirement ensures that the internal regulator and the control circuitry are in operation before PVDD1 supplies
energy to the output. In addition, Output 1 must be held in the disabled state (EN1 high) until there is sufficient
voltage on PVDD1 to support Output 1 in regulation. (See the Operating Near Maximum Duty Cycle section.)
The preferred sequence of events is:
1. PVDD2 rises above the input UVLO voltage
2. PVDD1 rises with Output 1 disabled until PVDD1 rises above level to support Output 1 regulation.
With these two conditions satisfied, there is no restriction on PVDD2 to be greater than, or less than PVDD1.
DESIGN HINT
An R-C delay on EN1 may be used to delay the startup of Output 1 for a long enough
period of time to ensure that PVDD1 can support Output 1 load.
Cascading Supply Operation
It is possible to source PVDD1 from Output 2 as depicted in Figure 27 and Figure 28. This configuration may be
preferred if the input voltage is high, relative to the voltage on Output 1.
VIN
TPS55383
1
PVDD1
PVDD2 16
2
BOOT1
BOOT2 15
3
SW1
SW2 14
4
GND
BP 13
5
EN1
SEQ 12
6
EN2
ILIM2 11
7
FB1
FB2 10
8
COMP1
OUTPUT2
OUTPUT1
COMP2
9
UDG-08043
Figure 27. Schematic Showing Cascading PVDD1 from Output 2
24
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PVDD2
Output2
PVDD1
Output1
T - Time
Figure 28. Waveforms Resulting from Cascading PVDD1 from Output 2
In this configuration, the following conditions must be maintained:
1. Output 2 must be of a voltage high enough to maintain regulation of Output 1 under all load conditions.
2. The sum of the current drawn by Output 2 load plus the current into PVDD1 must be less than the overload
protection current level of Output 2.
3. The method of output sequencing must be such that the voltage on Output 2 is sufficient to support Output 1
before Output 1 is enabled. This requrement may be accomplished by:
a. a delay of the enable function
b. selecting sequential sequencing of Output 1 starting after Output 2 is in regulation
Multiphase Operation
The TPS5538x may be configured to operate as a two-channel multiphase converter capable of delivering up to
6 A. Figure 29 indicates the recommended pin connections. In this configuration, FB2 must be tied to BP for the
maximum current configuration and the two output filter inductors must be the same value. Calculate RCOMP and
CCOMP as outlined for a single channel output, then use one-half the RCOMP value and two times the CCOMP value
as the compensation components. Contact the factory for further support.
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VIN
TPS55383
1
PVDD1
PVDD2 16
2
BOOT1
BOOT2 15
3
SW1
SW2 14
4
GND
BP 13
5
EN1
SEQ 12
6
EN2
ILIM2 11
7
FB1
FB2 10
8
COMP1
Output
COMP2
9
UDG-08123
Figure 29. Multiphase Operation Schematic
Bypass and FIltering
As with any integrated circuit, supply bypassing is important for jitter-free operation. To improve the noise
immunity of the converter, ceramic bypass capacitors must be placed as close to the package as possible.
1. PVDD1 to GND: Use a 10-µF ceramic capacitor
2. PVDD2 to GND: Use a 10-µF ceramic capacitor
3. BP to GND: Use a 4.7-µF to 10-µF ceramic capacitor
Overtemperature Protection and Junction Temperature Rise
The overtemperature thermal protection limits the maximum power to be dissipated at a given operating ambient
temperature. In other words, at a given device power dissipation, the maximum ambient operating temperature is
limited by the maximum allowable junction operating temperature. The device junction temperature is a function
of power dissipation, and the thermal impedance from the junction to the ambient. If the internal die temperature
should reach the thermal shutdown level, the TPS5538x shuts off both PWMs and remains in this state until the
die temperature drops below the hysteresis value, at which time the device restarts.
The first step to determine the device junction temperature is to calculate the power dissipation. The power
dissipation is dominated by the two switching MOSFETs and the BP internal regulator. The power dissipated by
each MOSFET is composed of conduction losses and output (switching) losses incurred while driving the
external rectifier diode. To find the conduction loss, first find the RMS current through the upper switch MOSFET.
2
æ
æ (D I
2
OUTPUTx )
IRMS(outputx) = D ´ ç (IOUTPUTx ) + ç
çç
ç
12
è
è
öö
÷÷
÷ ÷÷
øø
(16)
where
•
•
•
26
D is the duty cycle
IOUTPUTx is the dc output current
ΔIOUTPUTx is the peak ripple current in the inductor for Outputx
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Notice the impact of the operating duty cycle on the result.
Multiplying the result by the RDS(on) of the MOSFET gives the conduction loss.
PD(cond) = IRMS(outputx)2 ´ RDS(on)
(17)
The switching loss is approximated by:
2
(VIN) ´ CJ ´ fS
PD(SW) =
2
(18)
where
•
•
where CJ is the prallel capacitance of the rectifier diode and snubber (if any)
fS is the switching frequency
The total power dissipation is found by summing the power loss for both MOSFETs plus the loss in the internal
regulator.
PD = PD(cond)output1 + PD(SW )output1 + PD(cond)output2 + PD(SW )output2 + VIN ´ Iq
(19)
The temperature rise of the device junction depends on the thermal impedance from junction to the mounting pad
(See the Package Dissipation Ratings table), plus the thermal impedance from the thermal pad to ambient. The
thermal impedance from the thermal pad to ambient depends on the PCB layout (PowerPAD interface to the
PCB, the exposed pad area) and airflow (if any). See the PCB Layout Guidelines, Additional References section.
The operating junction temperature is shown in Equation 20.
TJ = TA + PD ´ qTH(pkg) + qTH(pad-amb)
(
)
(20)
Power Derating
The TPS5538x delivers full current at ambient temperatures up to +85°C if the thermal impedance from the
thermal pad maintains the junction temperature below the thermal shutdown level. At higher ambient
temperatures, the device power dissipation must be reduced to maintain the junction temperature at or below the
thermal shutdown level. Figure 30 illustrates the power derating for elevated ambient temperature under various
airflow conditions. Note that these curves assume that the PowerPAD is properly soldered to the recommended
thermal pad. (See the References section for further information.)
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POWER DISSIPATION
vs
AMBIENT TEMPERATURE
1.8
LFM = 250
1.6
LFM = 500
PD - Power Dissipation - W
1.4
LFM = 0
1.2
LFM = 150
1.0
0.8
0.6
LFM
0
150
250
500
0.4
0.2
0
0
20
40
60
80
100
120
TA - Ambient Temperature - °C
140
Figure 30. Power Derating Curves
PowerPAD Package
The PowerPAD package provides low thermal impedance for heat removal from the device. The PowerPAD
derives its name and low thermal impedance from the large bonding pad on the bottom of the device. The circuit
board must have an area of solder-tinned-copper underneath the package. The dimensions of this area depend
on the size of the PowerPAD package. Thermal vias connect this area to internal or external copper planes and
should have a drill diameter sufficiently small so that the via hole is effectively plugged when the barrel of the via
is plated with copper. This plug is needed to prevent wicking the solder away from the interface between the
package body and the solder-tinned area under the device during solder reflow. Drill diameters of 0.33 mm (13
mils) work well when 1-oz. copper is plated at the surface of the board while simultaneously plating the barrel of
the via. If the thermal vias are not plugged when the copper plating is performed, then a solder mask material
should be used to cap the vias with a diameter equal to the via diameter of 0.1 mm minimum. This capping
prevents the solder from being wicked through the thermal vias and potentially creating a solder void under the
package. (See the Additional References section.)
28
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PCB Layout Guidelines
The layout guidelines presented here are illustrated in the PCB layout examples given in Figure 31 and
Figure 32.
• Power pad must be connected to low current ground with available surface copper to dissipate heat.
Recommend extending ground land beyond device package area.
• Connect the GND pin to the PowerPAD through a 10-mil (.010 in, or 0.0254 mm) wide trace.
• Place the ceramic input capacitors close to PVDD1 and PVDD2; Connect ceramic input capacitor ground to
PowerPad with min 50mil wide trace.
• Maintain tight loop of wide traces from SW1 or SW2 through switch node, inductor, output capacitor and
rectifier diode. Avoid using vias in this loop.
• Use wide ground connection from input capacitor to rectifier diode as close to power path as possible.
Recommend directly under diode and switch node.
• Locate bootstrap capacitor close to BOOT pin to minimize gate drive loop.
• Locate feedback and compensation components over GND and away from switch node and rectifier diode to
input capacitor ground connection.
• Locate snubber components close to rectifier diode with minimize loop area.
• Locate BP bypass capacitor very close to device. Recommend minimal loop area.
• Locate output ceramic capacitor close to inductor output terminal between inductor and electrolytic capacitors
if used.
Figure 31. Top Layer Copper Layout and Component
Placement
Figure 32. Bottom Layer Copper Layout
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DESIGN EXAMPLES
Example 1: Detailed Design of a 12-V to 5-V and 3.3-V Converter
DESIGN EXAMPLE 1 GENERAL DESCRIPTION
The following example illustrates a design process and component selection for a 12-V to 5-V and 3.3-V dual
non-synchronous buck regulator using the TPS55386 converter. Design Example, and List of Materials is found
at the end of this section.
PARAMETER
NOTES AND CONDITIONS
MIN
NOM
MAX
UNIT
9.6
V
INPUT CHARACTERISTICS
VIN
Input Voltage
12.0
13.2
IIN
Input Current
VIN = Nom, IOUT1 = IOUT2 = Max
2.4
2.6
A
No Load Input Current
VIN = Nom, IOUT = 0 A
12
20
mA
Input UVLO
IOUT = Min to Max
4.0
4.2
4.4
V
VIN_UVLO
OUTPUT CHARACTERISTICS
VOUT1
Output Voltage 1
VIN = Nom, IOUT = Nom
4.80
5.0
5.20
V
VOUT2
Output Voltage 2
VIN = Nom, IOUT = Nom
3.20
3.3
3.40
V
Line Regulation
VIN = Min to Max
1%
Load Regulation
IOUT = Min to Max
1%
VOUT_ripple
Output Voltage Ripple
VIN = Nom, IOUT = Max
50
mVpp
IOUT1
Output Current 1
VIN = Min to Max
0
3.0
A
IOUT2
Output Current 2
VIN = Min to Max
0
3.0
A
IOCP1
Output Over Current Channel 1
VIN = Nom, VOUT = VOUT1–5%
3.3
4.2
5.2
A
IOCP2
Output Over Current Channel 2
VIN = Nom, VOUT = VOUT2–5%
3.3
4.2
5.2
A
Transient Response
ΔVout from load transient
ΔIOUT = 1 A at 3 A/µs
Settling Time
To 1% of Vout
200
mV
1
ms
SYSTEM CHARACTERISTICS
fSW
Switching Frequency
ηpk
Peak Efficiency
VIN = Nom, IOUT1 = IOUT2
500
η
Full Load Efficiency
VIN = Nom, IOUT1 = IOUT2 = Max
Top
Operating Temperature Range
VIN = Min to Max, IOUT = Min to Max
600
700
kHz
60
°C
93%
86%
0
25
Figure 33. Design Example Schematic
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The bill of materials for this application is shown below in Table 3. The efficiency, line and load regulation
measurements from boards built using this design are shown in Figure 34 and Figure 35.
DESIGN EXAMPLE 1 STEP-BY-STEP DESIGN PROCEDURE
Duty Cycle Estimation
The duty cycle of the main switching FET of each channel is estimated by:
VOUT1 + VFD
5.0 + 0.4
=
= 0.540
VIN(min ) + VFD 9.6 + 0.4
(21)
VOUT2 + VFD
3.3 + 0.4
=
= 0.370
VIN(min ) + VFD 9.6 + 0.4
(22)
VOUT1 + VFD
5.0 + 0.4
=
= 0.397
VIN(max ) + VFD 13.2 + 0.4
(23)
VOUT2 + VFD
3.3 + 0.4
=
= 0.272
VIN(max ) + VFD 13.2 + 0.4
(24)
DMAX1 »
DMAX2 »
DMIN1»
DMIN2 »
Inductor Selection
The peak-to-peak ripple is to be limited to 25% of the max output current, so that
ILrip(max ) = 0.25 ´ IOUT(max ) = 0.25 ´ 3.0 A = 0.750 A
(25)
The minimum inductor size is estimated by:
L min1 »
Lmin 2 »
VIN(max ) - VOUT1
ILrip1(max)
´ D min1 ´
VIN(max ) - VOUT2
ILrip2(max)
1
13.2 - 5.0
1
=
´ 0.397 ´
= 7.23 mH
fSW
0.75 A
600 kHz
´ Dmin 2 ´
1
fSW
=
(26)
13.2 - 3.3
1
´ 0.272 ´
= 6.0 mH
0.75 A
600kHz
(27)
The standard inductor value of 8.2 µH is selected for both Channel 1 and Channel 2. The resulting ripple
currents are estimated by:
IRIPPLE1 »
IRIPPLE2 »
VIN(max ) - VOUT1
L1
´ Dmin1 ´
VIN(max ) - VOUT2
L2
1
fSW
=
13.2 - 5.0
1
´ 0.397 ´
= 0.661A
8.2 mH
600kHz
1
13.2 - 3.3
1
=
´ 0.272 ´
= 0.547 A
fSW
8.2 mH
600kHz
´ Dmin 2 ´
(28)
(29)
RMS current through the inductor is approximated by:
2
IL1(rms ) =
(I ( ) ) +
L1 avg
2
1
12 IRIPPLE1
(
)
»
2
(I
OUT1(max
)) +
2
1
12 IRIPPLE1
(
)
(3.0 )2 + 112 (0.661)2 A = 3.0 A
=
(30)
IL2(rms ) =
(I
L2(avg)
2
)+
2
1
12 IRIPPLE2
(
)
»
(I
OUT2(max )
2
)+
2
1
12 IRIPPLE2
(
)
=
(3.0 )2 + 112 (0.547 )2 A = 3.0 A
(31)
The RMS inductor current is 3.0 for both channels.
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A DC current with 30% peak to peak ripple has an RMS current approximately 0.4% above the average current.
The peak inductor current is estimated by:
IL1(peak ) » IOUT1(max ) + 12 IRIPPLE = 3.0 A + 12 0.661A = 3.3 A
1
(32)
1
IL2(peak ) » IOUT2(max ) + 2 IRIPPLE = 3.0 A + 2 0.547 A = 3.3 A
(33)
An 8.2-µH inductor with a minimum RMS current rating of 3.0 A and minimum saturation current rating of 3.3 A
must be selected. A Coilcraft MSS1048-822ML 8.2-µH, 4.38-A inductor is chosen for both outputs.
Rectifier Diode Selection
A low forward voltage drop schottky diode is used as a rectifier diode to minimize power dissipation and
maximize efficiency.
V(BR )R(min ) ³
VIN(max )
0.8
= 1.25 ´ VIN(max ) = 1.25 ´ 13.2 V = 16.5 V
(34)
Allowing 20% over VIN for ringing on the switch node, the rectifier diode’s minimum reverse break-down voltage
is given by:
ID1(avg) » IOUT1(max ) ´ (1 - DMIN1 ) = 3.0 A ´ (1 - 0.397) = 1.81A
(35)
ID2(avg) » IOUT2(max ) ´ (1 - DMIN2 ) = 3.0 A ´ (1 - 0.272) = 2.18 A
(36)
ID(peak ) = IL(peak )
(37)
Reviewing 20-V and 30-V schottky diodes, the MBRS330T3, 30-V, 3-A diodes in an SMC package are selected
for both channels. This diode has a forward voltage drop of 0.4 V at 3 A, so the conduction power dissipation is:
PD1(max ) » VFM ´ ID1(avg) » 0.4 V ´ 1.81 = 0.72 W
(38)
PD2(max) » VFM ´ ID2(avg) » 0.4V ´ 2.18 = 0.87W
(39)
For this design, the maximum power dissipation is estimated as 0.72 W and 0.87 W respectively.
Output Capacitor Selection
Output capacitors are selected to support load transients and output ripple current. The minimum output
capacitance to meet the transient specification is given by:
2
COUT1(min )
(I
=
TRAN(MAX )
2
) ´ L = (1A ) ´ 8.2 mH = 8.2 mF
(VOUT1 )´ VOVER
5.0 V ´ 0.2 V
(40)
2
COUT2(min )
32
(I
=
TRAN(MAX )
2
) ´ L = (1A ) ´ 8.2 mH = 12.4 mF
(VOUT2 )´ VOVER
3.3 V ´ 0.2 V
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The maximum ESR to meet the ripple specification is given by:
æ
ö
IRIPPLE1
æ
ö
0.661A
VRIPPLE1(total) - ç
÷ 0.050 V - ç
÷
è 8 ´ C OUT1 ´ fSW ø =
è 8 ´ 8.2 mF ´ 600 kHz ø = 0.024 W F
ESR1(max ) =
IRIPPLE1
0.661A
(42)
æ
ö
IRIPPLE
æ
ö
0.547 A
VRIPPLE(total) - ç
÷ 0.050 V - ç
÷
8
C
f
´
´
8
12.4
F
600kHz
´
m
´
OUT1 SW ø
è
è
ø = 0.033 WF
ESR(max ) =
=
IRIPPLE
0.547 A
(43)
A single 22-µF ceramic capacitor with approximately 2.5 mΩ of ESR is selected to provide sufficient margin for
capacitance loss due to DC voltage bias.
Input Capacitor Selection
The TPS55386 datasheet recommends a 10µF (minimum) ceramic bypass capacitor on each PVDD pin. While
out of phase operation reduces input RMS current, the input capacitors must be sized to support the greater of
the two input RMS currents, or 1.5A to allow operation when one channel is at maximum load and the other is
un-loaded. The ceramic capacitor must handle the RMS input ripple current of the converter.
The RMS current in the input capacitors is estimated by:
IRMS _ CIN = IOUT ´ D ´ (1 - D ) = 3 A ´ 0.5 ´ (1 - 0.5 ) = 1.5 A
(44)
One 1210 size 10-µF, 25-V, X5R ceramic capacitor with a 2-mΩ ESR and a 2-A RMS current rating are selected
to bypass each PVDD input. Higher voltage capacitors minimize capacitance loss under DC bias voltage,
ensuring the capacitors have sufficient capacitance at their working voltage.
Voltage Feedback
The primary feedback divider resistor (RFB) from VOUT to FB should be selected between 10 kΩ and 100 kΩ to
maintain a balance between power dissipation and noise sensitivity. For a 3.3-V and 5-V output, 20.5 kΩ is
selected, so the lower resistor is given by:
RBIAS =
VFB ´ RFB
VOUT - VFB
(45)
For RFB = R2 = R9 = 20.5 kΩ and VFB = 0.80V, RBIAS1 = 3.90kΩ and RBIAS2 = 6.5kΩ (R4 = 3.83kΩ and
R7 = 6.49 kΩ selected) for 5.0 V and 3.3 V respectively.
Compensation Components
The TPS55386 controller uses an internal transconductance error amplifier, which compares the feedback
voltage to the internal 0.80-V reference and sources a current proportional to the resulting error out of the COMP
pin. A series resistor and capacitor to ground generate an integrator with zero while a high frequency capacitor
provides a second pole to reduce the high frequency gain. The compensation loop components are selected by
the following equations with the 5.0-V output used in example calculations:
Calculate the modulator gain at DC:
FM1 =
600000
600000
6
(1.5´10 ´t ) + 50 ´ 10-6 ´ æ VIN - VOUT1 ö
19.7 ´ e
ON
ç
è
L
÷
ø
=
6
(1.5´10 ´6.68´10-7 )
19.7 ´ e
+ 50 ´ 10
-6
æ 13.2 - 5.0 ö
´ç
÷
è 8.2 mH ø
= 5.82 ´ 103
(46)
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Then calculate the converter gain at DC:
-4
fc1 =
3
VIN ´ Fm ´ 2 ´ (10 )
æ V ´ Fm ´ 50 ´ (10 )-6
1 + ç IN
ç
RLOAD1
è
=
ö
÷
÷
ø
-4
13.2 ´ 5.82 ´ (10 ) ´ 2 ´ (10 )
æ 13.2 ´ 5.82 ´ (10 )3 ´ 50 ´ (10 )-6
1+ ç
ç
1.67W
è
= 4.63
ö
÷
÷
ø
(47)
Calculate the required error amplifier gain at the desired crossover frequency of 35 kHz:
æ
ö
æ
ö
fc1
4.65
K EA1 = - 20 ´ log ç
÷ = - 20 ´ log ç
÷ = 5.80 dB
1
2
f
R
C
1
2
35
kHz
1.67
22
F
+
p
´
´
´
+
p
´
´
W
´
m
è
ø
CO
LOAD1
OUT1 ø
è
(48)
Then compensation resistor at the output of the error amplifier is:
KEA
5.80 dB
10 20 ´ (ZLOWER + ZUPPER ) 10 20 ´ (3.83kW + 20.5kW )
=
= 38.5kW Þ R15 = 38.3kW
RCOMP1 =
gM ´ ZLOWER
315 mS ´ 3.83kW
(49)
Calculate the required compensation zero frequency:
fZERO1 =
1
2p ´ COUT1 ´ RLOAD1
=
1
= 4.4kHz
2p ´ 22 mF ´ 1.67 W
(50)
Then calculate the compensation capacitor:
CCOMP1 =
1
1
=
= 967pF Þ C21 = 1nF
2p ´ fPOLE1 ´ RCOMP1 2p ´ 4.4kHz ´ 3.83kW
(51)
The high-frequency pole is placed at eight times the crossover frequency:
CHF1 =
1
2p ´ 4 ´ fCO ´ RCOMP
=
1
= 29.6pF Þ C23 = 33pF
2p ´ 4 ´ 35kHz ´ 38.3kW
(52)
Boot-Strap Capacitor
To ensure proper charging of the high-side FET gate and limit the ripple voltage on the boost capacitor, a 47-nF
boot strap capacitor is used.
ILIM2
The current limit must be set above the peak inductor current ILpeak. Comparing ILpeak to the available minimum
current limits, ILIM is connected to BP for a 3.6-A minimum current limit.
SEQ
The SEQ pin is left floating, leaving the enable pins to function independently. If the enable pins are tied
together, the two supplies start-up ratio-metrically. SEQ could also be connected to BP or GND to provide
sequential start-up.
34
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Power Dissipation
The power dissipation in the TPS55386 is from FET conduction losses, switching losses and regulator losses.
Conduction losses are estimated by:
2
PCON1 = RDS(on )´ IQSW (RMS )
(
) »R
DS(on )´ IOUT
(
2
PCON2 = RDS(on )´ IQSW (RMS )
(
) »R
DS(on )´ IOUT
(
)2 ´
2
D = 0.085 W ´ (3 A ) ´ 0.540 = 0.562 W
(53)
)2 ´
2
D = 0.085 W ´ (3 A ) ´ 0.370 = 0.465 W
(54)
The switching losses are estimated by:
2
PSW1 = PSW 2
(V
»
IN(max )
) ´ (C
Dj
+ COSS ) ´ fSW
=
2
(13.2 )2 ´ (200pF + 250pF) ´ 600kHz
2
= 23.5mW
(55)
The regulator losses are estimated by:
PREG » IDD ´ VIN(max ) + IBP ´ VIN(max ) - VBP = 5mA ´ 13.2 V = 66mW
)
(
(56)
Total power dissipation in the device is the sum of conduction and switching losses for both channels plus
regulator losses, and are estimated to total 1.2 W.
DESIGN EXAMPLE 1 TEST RESULTS
EFFICIENCY
vs
LOAD CURRENT
EFFICIENCY
vs
LOAD CURRENT
100
100
VIN = 8 V
VOUT= 5 V
VOUT= 3.3 V
95
95
90
VIN = 12 V
85
h – Efficiency – %
h – Efficiency – %
VIN = 8 V
VIN = 14 V
80
VIN (V)
14
12
8
75
70
0.15
0.65
1.15
1.65
2.15
2.65
ILOAD – Load Current – A
90
85
VIN = 12 V
VIN (V)
14
12
8
75
3.15
VIN = 14 V
80
70
0.15
0.65
Figure 34.
1.15
1.65
2.15
2.65
ILOAD – Load Current – A
3.15
Figure 35.
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Table 3. TPS55386 Design Example List of Materials
QTY
36
REFERENC
E
DESIGNAT
OR
VALUE
DESCRIPTION
SIZE
PART NUMBER
MFR
2
C2, C14
22 µF
Capacitor, Ceramic, 6.3V, X5R, 20%
1206
C3216X5R0J226M
TDK
2
C3, C13
470 pF
Capacitor, Ceramic, 25V, X7R, 20%
0603
Std
Std
2
C4, C11
0.047 µF
Capacitor, Ceramic, 25V, X7R, 20%
0603
Std
Std
2
C5, C10
10 µF
Capacitor, Ceramic, 25V, X5R, 20%
1210
C3225X5R1E106M
TDK
1
C12
4.7 µF
Capacitor, Ceramic, 10V, X5R, 20%
0805
Std
Std
2
C9, C6
1.0 nF
Capacitor, Ceramic, 25V, X7R, 20%
0603
Std
Std
1
C8
47 pF
Capacitor, Ceramic, 25V, X7R, 20%
0603
Std
Std
1
C7
33 pF
Capacitor, Ceramic, 25V, X7R, 20%
0603
Std
Std
2
D1, D2
MBRS330T3
Diode, Schottky, 3-A, 30-V
SMC
MBRS330T3
OnSemi
2
L1, L2
8.2 µH
Inductor, SMT, 4.38A, 20milliohm
0.402 x 0.394 inch
MSS1048-822L
Coilcraft
1
R7
23.7 kΩ
Resistor, Chip, 1/16W, 1%
0603
Std
Std
1
R6
38.3 kΩ
Resistor, Chip, 1/16W, 1%
0603
Std
Std
2
R3, R12
20.5 kΩ
Resistor, Chip, 1/16W, 1%
0603
Std
Std
2
R2, R11
10 Ω
Resistor, Chip, 1/16W, 5%
0603
Std
Std
1
R4
3.83 kΩ
Resistor, Chip, 1/16W, 1%
0603
Std
Std
1
R10
6.49 kΩ
Resistor, Chip, 1/16W, 1%
0603
Std
Std
1
U1
TPS55386PWP
IC, Dual 600kHz Non-Sync BUCK with Interal
FET
HTSSOP-16
TPS55386PWP
TI
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Example 2: Cascading Configuration: 24 V to 12 V at 2 A then 3.3 V at 2 A
This example illustrates a cascaded configuration. To accommodate the low duty cycle of a 24-V to 3.3-V supply,
PVDD1 is connected to VOUT2, a 12-V output. VOUT2 is used as the source supply for VOUT1. The sequence
pin is connected to BP, ensuring the 12-V supply is in regulation before the 3.3-V is allowed to turn on.
U
Figure 36. Design Example 2, TPS55386 in a Cascaded Configuration
EFFICIENCY
vs
LOAD CURRENT
100
90
VOUT= 3.3 V
80
h – Efficiency – %
70
VOUT= 12 V
60
50
40
30
20
VOUT (V)
3.3
12
10
0
0
0.5
1.0
1.5
2.0
ILOAD – Load Current – A
Figure 37.
2.5
Figure 38. Design Example 2 Outputs and Switch Nodes
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37
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Example 3: Multiphase 12 V to 5.0 V at 6 A
The combination of current mode control and a transconductance amplifier allows the TPS55386 to serve as a
single-output 2-phase supply. This configuration allows this part to serve as a 6-A non-synchronous converter at
an effective 1.2 MHz. COMP2 is connected to COMP1 and FB2 is connected to BP. While not implemented in
this example, EN2 could be used to disable Channel 2 at light load, improving efficiency.
Figure 39. Design Example 3, TPS55386 as a Phase Non-Synchronous Buck Converter
EFFICIENCY
vs
LOAD CURRENT
100
90
80
h – Efficiency – %
70
60
50
40
30
20
10
VOUT= 5 V
0
0
1
3
4
2
5
ILOAD – Load Current – A
Figure 40.
38
6
7
Figure 41. Design Example 3, Output and Switch Nodes
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ADDITIONAL REFERENCES
Related Devices
The following devices have characteristics similar to the TPS55383/TPS55386 and may be of interest.
Table 4. Devices Related to the TPS55383 and TPS55386
TI LITERATURE
NUMBER
DEVICE
SLUS642
TPS40222
5-V Input, 1.6-A Non-Synchronous Buck Converter
SLUS749
TPS54283 /
TPS54286
2-A Dual Non-Synchronous Converter with Integrated High-Side MOSFET
SLUS774
TPS54383 /
TPS54386
3-A Dual Non-Synchronous Converter with Integrated High-Side MOSFET
SLVS839
TPS54331
3.5 V to 28 V, Single 3-A Non-Synchronous Buck Converter with Integrated High-Side
MOSFET
DESCRIPTION
References
These references, design tools and links to additional references, including design software, may be found at
http:www.power.ti.com
Table 5. References
TI LITERATURE
NUMBER
DESCRIPTION
SLMA002
PowerPAD Thermally Enhanced Package Application Report
SLMA004
PowerPAD™ Made Easy
SLUP206
Under The Hood Of Low Voltage DC/DC Converters. SEM1500 Topic 5, 2002 Seminar Series
SLVA057
Understanding Buck Power Stages in Switchmode Power Supplies
SLUP173
Designing Stable Control Loops. SEM 1400, 2001 Seminar Series
Package Outline and Recommended PCB Footprint
The following pages outline the mechanical dimensions of the 16-Pin PWP package and provide
recommendations for PCB layout.
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39
PACKAGE MATERIALS INFORMATION
www.ti.com
5-Jan-2022
TAPE AND REEL INFORMATION
*All dimensions are nominal
Device
Package Package Pins
Type Drawing
SPQ
Reel
Reel
A0
Diameter Width (mm)
(mm) W1 (mm)
TPS55383PWPR
HTSSOP
PWP
16
2000
330.0
12.4
TPS55386PWPR
HTSSOP
PWP
16
2000
330.0
12.4
Pack Materials-Page 1
B0
(mm)
K0
(mm)
P1
(mm)
W
Pin1
(mm) Quadrant
6.9
5.6
1.6
8.0
12.0
Q1
6.9
5.6
1.6
8.0
12.0
Q1
PACKAGE MATERIALS INFORMATION
www.ti.com
5-Jan-2022
*All dimensions are nominal
Device
Package Type
Package Drawing
Pins
SPQ
Length (mm)
Width (mm)
Height (mm)
TPS55383PWPR
HTSSOP
PWP
16
2000
350.0
350.0
43.0
TPS55386PWPR
HTSSOP
PWP
16
2000
350.0
350.0
43.0
Pack Materials-Page 2
PACKAGE MATERIALS INFORMATION
www.ti.com
5-Jan-2022
TUBE
*All dimensions are nominal
Device
Package Name
Package Type
Pins
SPQ
L (mm)
W (mm)
T (µm)
B (mm)
TPS55383PWP
PWP
HTSSOP
16
90
530
10.2
3600
3.5
TPS55386PWP
PWP
HTSSOP
16
90
530
10.2
3600
3.5
Pack Materials-Page 3
PACKAGE OUTLINE
PWP0016C
TM
PowerPAD TSSOP - 1.2 mm max height
SCALE 2.500
SMALL OUTLINE PACKAGE
6.6
TYP
6.2
A
C
PIN 1 INDEX
AREA
0.1 C
SEATING
PLANE
14X 0.65
16
1
2X
5.1
4.9
NOTE 3
4.55
8
9
4.5
4.3
B
16X
0.30
0.19
0.1
C A B
SEE DETAIL A
(0.15) TYP
2X 0.95 MAX
NOTE 5
4X (0.3)
8
9
2X 0.23 MAX
NOTE 5
2.31
1.75
17
0.25
GAGE PLANE
16
1
0 -8
1.2 MAX
0.15
0.05
0.75
0.50
DETAIL A
A 20
THERMAL
PAD
2.46
1.75
TYPICAL
4224559/B 01/2019
PowerPAD is a trademark of Texas Instruments.
NOTES:
1. All linear dimensions are in millimeters. Any dimensions in parenthesis are for reference only. Dimensioning and tolerancing
per ASME Y14.5M.
2. This drawing is subject to change without notice.
3. This dimension does not include mold flash, protrusions, or gate burrs. Mold flash, protrusions, or gate burrs shall not
exceed 0.15 mm per side.
4. Reference JEDEC registration MO-153.
5. Features may differ or may not be present.
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EXAMPLE BOARD LAYOUT
PWP0016C
TM
PowerPAD TSSOP - 1.2 mm max height
SMALL OUTLINE PACKAGE
(3.4)
NOTE 9
(2.46)
16X (1.5)
METAL COVERED
BY SOLDER MASK
SYMM
1
16X (0.45)
16
(1.2) TYP
(R0.05) TYP
SYMM
(2.31)
17
(0.6)
(5)
NOTE 9
14X (0.65)
( 0.2) TYP
VIA
9
8
SOLDER MASK
DEFINED PAD
(1) TYP
SEE DETAILS
(5.8)
LAND PATTERN EXAMPLE
EXPOSED METAL SHOWN
SCALE: 10X
SOLDER MASK
OPENING
METAL UNDER
SOLDER MASK
METAL
SOLDER MASK
OPENING
EXPOSED METAL
EXPOSED METAL
0.05 MAX
ALL AROUND
0.05 MIN
ALL AROUND
NON-SOLDER MASK
DEFINED
SOLDER MASK
DEFINED
SOLDER MASK DETAILS
15.000
4224559/B 01/2019
NOTES: (continued)
6. Publication IPC-7351 may have alternate designs.
7. Solder mask tolerances between and around signal pads can vary based on board fabrication site.
8. This package is designed to be soldered to a thermal pad on the board. For more information, see Texas Instruments literature
numbers SLMA002 (www.ti.com/lit/slma002) and SLMA004 (www.ti.com/lit/slma004).
9. Size of metal pad may vary due to creepage requirement.
10. Vias are optional depending on application, refer to device data sheet. It is recommended that vias under paste be filled, plugged
or tented.
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EXAMPLE STENCIL DESIGN
PWP0016C
TM
PowerPAD TSSOP - 1.2 mm max height
SMALL OUTLINE PACKAGE
(2.46)
BASED ON
0.125 THICK
STENCIL
16X (1.5)
1
METAL COVERED
BY SOLDER MASK
16
16X (0.45)
(R0.05) TYP
SYMM
(2.31)
BASED ON
0.125 THICK
STENCIL
17
14X (0.65)
9
8
SYMM
(5.8)
SEE TABLE FOR
DIFFERENT OPENINGS
FOR OTHER STENCIL
THICKNESSES
SOLDER PASTE EXAMPLE
BASED ON 0.125 mm THICK STENCIL
SCALE: 10X
STENCIL
THICKNESS
SOLDER STENCIL
OPENING
0.1
0.125
0.15
0.175
2.75 X 2.58
2.46 X 2.31 (SHOWN)
2.25 X 2.11
2.08 X 1.95
4224559/B 01/2019
NOTES: (continued)
11. Laser cutting apertures with trapezoidal walls and rounded corners may offer better paste release. IPC-7525 may have alternate
design recommendations.
12. Board assembly site may have different recommendations for stencil design.
www.ti.com
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