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UCC28722
SLUSBL7B – DECEMBER 2013 – REVISED OCTOBER 2015
UCC28722 Constant-Voltage, Constant-Current Controller
With Primary-Side Regulation, BJT Drive
1 Features
3 Description
•
•
The UCC28722 flyback power supply controller
provides isolated-output constant-voltage (CV) and
constant-current (CC) output regulation without the
use of an optical coupler. The device processes
information from the primary power switch and an
auxiliary flyback winding for precise control of output
voltage and current.
1
•
•
•
•
•
•
•
•
< 50-mW No-Load Power
Primary-Side Regulation (PSR) Eliminates OptoCoupler
Dynamic BJT Drive
± 5% Voltage and Current Regulation Across Line
and Load
80-kHz Maximum Switching Frequency Enables
High-Power Density Charger Designs
Quasi-Resonant Valley-Switching Operation for
Highest Overall Efficiency
Wide VDD Range Allows Small Bias Capacitor
Output Overvoltage, Low-Line, and Overcurrent
Protection Functions
Programmable Cable Compensation
SOT23-6 Package
Dynamically-controlled operating states and a tailored
modulation profile support high-efficiency operation at
all load levels without sacrificing output transient
response.
Control algorithms in the UCC28722 device allow
operating efficiencies to meet or exceed applicable
standards. The output drive interfaces to a bipolar
transistor power switch, enabling lower-cost converter
design. Discontinuous conduction mode (DCM) with
valley switching reduces switching losses, while
modulation of switching frequency and primary
current peak amplitude (FM and AM) keeps the
conversion efficiency high across the entire load and
line ranges.
2 Applications
•
•
•
USB-Compliant Adapters and Chargers for
Consumer Electronics
– Smart Phones
– Tablet Computers
– Cameras
Standby Supply for TVs and Desktops
White Goods
The controller has a maximum switching frequency of
80 kHz and always maintains control of the peakprimary current in the transformer. Output overvoltage
and overcurrent as well as input undervoltage
protection features help keep primary and secondary
component stresses in check. The UCC28722 also
allows compensation for voltage drop in the cable to
be programmed with an external resistor.
Device Information(1)
PART NUMBER
UCC28722
PACKAGE
DBV (6)
BODY SIZE (NOM)
2.90 mm × 1.60 mm
(1) For all available packages, see the orderable addendum at
the end of the data sheet.
+
Simplified Application Diagram
Ns
VOUT
–
VIN
Np
Na
UCC28722
1
CBC
2 VDD
3
DRV
VS 6
GND
5
CS
4
1
An IMPORTANT NOTICE at the end of this data sheet addresses availability, warranty, changes, use in safety-critical applications,
intellectual property matters and other important disclaimers. PRODUCTION DATA.
UCC28722
SLUSBL7B – DECEMBER 2013 – REVISED OCTOBER 2015
www.ti.com
Table of Contents
1
2
3
4
5
6
7
Features ..................................................................
Applications ...........................................................
Description .............................................................
Revision History.....................................................
Pin Configuration and Functions .........................
Specifications.........................................................
1
1
1
2
3
3
6.1
6.2
6.3
6.4
6.5
6.6
3
4
4
4
5
6
Absolute Maximum Ratings ......................................
ESD Ratings ............................................................
Recommended Operating Conditions.......................
Thermal Information ..................................................
Electrical Characteristics...........................................
Typical Characteristics ..............................................
Detailed Description .............................................. 8
7.1 Overview ................................................................... 8
7.2 Functional Block Diagram ......................................... 8
7.3 Feature Description................................................... 9
7.4 Device Functional Modes........................................ 11
8
Application and Implementation ........................ 16
8.1 Application Information............................................ 16
8.2 Typical Application ................................................. 16
9 Power Supply Recommendations...................... 22
10 Layout................................................................... 23
10.1 Layout Guidelines ................................................. 23
10.2 Layout Example .................................................... 23
11 Device and Documentation Support ................. 25
11.1
11.2
11.3
11.4
11.5
11.6
Device Support ....................................................
Documentation Support .......................................
Community Resources..........................................
Trademarks ...........................................................
Electrostatic Discharge Caution ............................
Glossary ................................................................
25
26
27
27
27
27
12 Mechanical, Packaging, and Orderable
Information ........................................................... 27
4 Revision History
NOTE: Page numbers for previous revisions may differ from page numbers in the current version.
Changes from Revision A (January 2014) to Revision B
•
Page
Added ESD Rating table, Feature Description section, Device Functional Modes, Application and Implementation
section, Power Supply Recommendations section, Layout section, Device and Documentation Support section, and
Mechanical, Packaging, and Orderable Information section. ................................................................................................ 1
Changes from Original (December 2013) to Revision A
Page
•
Changed Simplified Application Diagram. .............................................................................................................................. 1
•
Changed Supply current, fault values from 95 µA and 170 µA to 2.00 mA and 2.65 mA...................................................... 5
•
Changed Bias Supply Current vs VDD Voltage image........................................................................................................... 6
•
Changed Operating Current vs Junction Temperature image................................................................................................ 6
•
Changed Functional Block Diagram. ...................................................................................................................................... 8
•
Changed Simplified Flyback Convertor image. .................................................................................................................... 11
•
Changed CDD equation. ........................................................................................................................................................ 20
2
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SLUSBL7B – DECEMBER 2013 – REVISED OCTOBER 2015
5 Pin Configuration and Functions
DBV Package
6-Pin SOT-23
Top View
CBC
1
6
VS
VDD
2
5
GND
DRV
3
4
CS
Pin Functions
PIN
NAME
NO.
I/O
DESCRIPTION
CBC
1
I
Cable compensation is a programming pin for compensation of cable voltage drop. Cable
compensation is programmed with a resistor to GND.
CS
4
I
Current sense input connects to a ground-referenced current-sense resistor in series with the power
switch. The resulting voltage is used to monitor and control the peak primary current. A series resistor
can be added to this pin to compensate the peak switch current levels as the AC-mains input varies.
DRV
3
O
Drive is an output used to drive the base of an external high voltage NPN transistor.
GND
5
—
The ground pin is both the reference pin for the controller and the low-side return for the drive output.
Special care should be taken to return all AC decoupling capacitors as close as possible to this pin and
avoid any common trace length with analog signal return paths.
VDD
2
I
VDD is the bias supply input pin to the controller. A carefully-placed bypass capacitor to GND is
required on this pin.
I
Voltage sense is an input used to provide voltage and timing feedback to the controller. This pin is
connected to a voltage divider between an auxiliary winding and GND. The value of the upper resistor
of this divider is used to program the AC-mains run and stop thresholds and line compensation at the
CS pin.
VS
6
6 Specifications
6.1 Absolute Maximum Ratings
over operating free-air temperature range (unless otherwise noted) (1)
MIN
MAX
UNIT
VVDD
Bias supply voltage, VDD
38
V
IDRV
Continuous base current sink
50
mA
IDRV
Continuous base current source
Self-limiting
mA
IVS
Peak current, VS
VDRV
Base drive voltage at DRV
Voltage
TJ
–1.2
mA
–0.5
Self-limiting
V
VS
–0.75
7
V
CS, CBC
–0.5
5
V
–55
150
°C
260
°C
150
°C
Operating junction temperature
Lead temperature 0.6 mm from case for 10 seconds
Tstg
(1)
Storage temperature
–65
Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. These are stress ratings
only, which do not imply functional operation of the device at these or any other conditions beyond those indicated under Recommended
Operating Conditions. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
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SLUSBL7B – DECEMBER 2013 – REVISED OCTOBER 2015
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6.2 ESD Ratings
V(ESD)
(1)
(2)
Electrostatic discharge
VALUE
UNIT
Human-body model (HBM), per ANSI/ESDA/JEDEC JS-001 (1)
±2000
V
Charged-device model (CDM), per JEDEC specification JESD22C101 (2)
±500
V
JEDEC document JEP155 states that 500-V HBM allows safe manufacturing with a standard ESD control process.
JEDEC document JEP157 states that 250-V CDM allows safe manufacturing with a standard ESD control process.
6.3 Recommended Operating Conditions
over operating free-air temperature range (unless otherwise noted)
MIN
NOM
MAX
UNIT
VDD
Bias supply operating voltage
9
35
V
CVDD
VDD bypass capacitor
1.0
10
µF
RCBC
Cable-compensation resistance
10
IVS
VS pin current
TJ
Operating junction temperature
kΩ
−1
mA
−40
125
°C
6.4 Thermal Information
UCC28722
THERMAL METRIC (1)
DBV (SOT-23)
UNIT
6 PINS
RθJA
Junction-to-ambient thermal resistance
180.0
°C/W
RθJC(top)
Junction-to-case (top) thermal resistance
72.2
°C/W
RθJB
Junction-to-board thermal resistance
44.4
°C/W
ψJT
Junction-to-top characterization parameter
5.1
°C/W
ψJB
Junction-to-board characterization parameter
43.8
°C/W
RθJC(bot)
Junction-to-case (bottom) thermal resistance
NA
°C/W
(1)
4
For more information about traditional and new thermal metrics, see the Semiconductor and IC Package Thermal Metrics application
report, SPRA953.
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6.5 Electrical Characteristics
over operating free-air temperature range, VVDD = 25 V, HV = open, RCBC = open, TA = -40°C to 125°C, TA = TJ
(unless otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
BIAS SUPPLY INPUT
IRUN
Supply current, run
IDRV = 0, run state
2.00
2.65
mA
IWAIT
Supply current, wait
IDRV = 0, wait state
95
170
µA
ISTART
Supply current, start
IDRV = 0, VVDD = 18 V, start state, IHV = 0
1.0
1.5
µA
IFAULT
Supply current, fault
IDRV = 0, fault state
2.00
2.65
mA
UNDERVOLTAGE LOCKOUT
VVDD(on)
VDD turnon threshold
VVDD low to high
19
21
23
V
VVDD(off)
VDD turnoff threshold
VVDD high to low
7.2
7.7
8.3
V
VVSR
Regulating level
Measured at no-load condition, TJ = 25°C (1)
3.99
4.05
4.11
V
VVSNC
Negative clamp level
IVS = –300 µA, volts below ground
190
250
325
mV
IVSB
Input bias current
VVS = 4 V
–0.25
0
0.25
µA
VS INPUT
CS INPUT
VCST(max) Max CS threshold voltage
VVS = 3.7 V
730
780
820
mV
VCST(min)
Min CS threshold voltage
VVS = 4.35 V
170
190
220
mV
KAM
AM control ratio
VCST(max) / VCST(min)
3.6
4.0
4.4
V/V
VCCR
Constant current regulating level
CC regulation constant
314
330
347
mV
KLC
Line compensation current ratio
IVSLS = –300 µA, IVSLS / current out of CS pin
24.0
25.0
28.6
A/A
TCSLEB
Leading-edge blanking time
DRV output duration, V CS = 1 V
230
290
355
ns
IDRS(max)
Maximum DRV source current
VDRV = 2 V, VVDD = 9 V, VVS = 3.85 V
31
37
42
mA
IDRS(min)
Minimum DRV source current
VDRV = 2 V, VVDD = 9 V, VVS = 4.30 V
15
19
23
mA
RDRVLS
DRV low-side drive resistance
IDRV = 10 mA
1
2.4
Ω
VDRCL
DRV clamp voltage
VVDD = 35 V
5.9
7
V
RDRVSS
DRV pulldown in start state
20
25
kΩ
DRIVER
TIMING
fSW(max)
Maximum switching frequency
VVS = 3.7 V
72
80
89
kHz
fSW(min)
Minimum switching frequency
VVS = 4.35 V
570
650
750
Hz
tZTO
Zero-crossing timeout delay
2.4
3.1
3.7
µs
4.49
4.60
4.75
V
PROTECTION
VOVP
Overvoltage threshold
At VS input, TJ = 25°C (1)
VOCP
Overcurrent threshold
At CS input
1.4
1.5
1.6
V
IVSL(run)
VS line-sense run current
Current out of VS pin increasing
188
225
277
µA
IVSL(stop)
VS line-sense stop current
Current out of VS pin decreasing
KVSL
VS line sense ratio
IVSL(run) / IVSL(stop)
TJ(stop)
Thermal shutdown temperature
Internal junction temperature
70
80
100
µA
2.45
2.80
3.05
A/A
165
°C
CABLE COMPENSATION
VCBC(max)
Cable compensation maximum
voltage
Voltage at CBC at full load
2.9
3.1
3.5
V
VCVS(min)
Minimum compensation at VS
VCBC = open, change in VS regulating level at full
load
–55
–15
25
mV
VCBC = 0 V, change in VS regulating level at full
load
270
320
385
mV
VCVS(max) Maximum compensation at VS
(1)
The regulating level and over voltage at VS decreases with temperature by 0.8 mV/˚C. This compensation is included to reduce the
power supply output voltage variance over temperature.
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6.6 Typical Characteristics
VDD = 25 V, unless otherwise noted.
10
10
Run State
IRUN, VDD = 25 V
1
IVDD − Bias Supply Current (mA)
IVDD − Bias Supply Current (mA)
1
Wait State
0.1
0.01
VDD Turn−Off
0.001
VDD Turn−On
IWAIT, VDD = 25 V
0.1
0.01
0.001
0.0001
ISTART, VDD = 18 V
Start State
0.00001
0
5
10
15
20
25
VDD − Bias Supply Voltage (V)
30
35
0.0001
−25
0
25
50
75
TJ − Temperature (°C)
100
125
G001
Figure 1. Bias Supply Current vs VDD Voltage
G002
Figure 2. Operating Current vs Junction Temperature
250
VS Pin Current Threshold (µA)
VS Regulating Voltage (V)
4.2
4.15
4.1
4.05
4
3.95
150
100
-50
-25
0
25
50
75
100
Junction Temperature (°C)
-50
125
VVS = 3.7 V
600
500
400
VVS = 4.35 V
200
100
0
-50
-25
0
25
50
75
100
Junction Temperature (°C)
50
75
100
125
C004
125
340
335
330
325
320
-50
-25
0
25
50
75
Junction Temperature (°C)
C005
Figure 5. Current Sense Threshold
vs Temperature
25
Junction Temperature (°C)
Constant Current Regulating Level (mV)
800
300
0
Figure 4. VS Pin Start and Stop Thresholds
vs Junction Temperature
900
700
-25
C003
Figure 3. VS Pin Regulation Voltage
vs Junction Temperature
Minimum Current Sense Threshold (mV)
Stop
50
0
3.9
6
Start
200
100
125
C006
Figure 6. Constant Current Regulation Level
vs Junction Temperature
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Typical Characteristics (continued)
VDD = 25 V, unless otherwise noted.
85
Maximum Switching Frequency (kHz)
Minimum Switching Frequency (Hz)
700
690
680
670
660
650
640
630
620
610
600
84
83
82
81
80
79
78
77
76
75
-50
-25
0
25
50
75
100
Junction Temperature (°C)
125
-50
-25
0
25
50
75
100
Junction Temperature (°C)
C007
Figure 7. Minimum Switching Frequency
vs Junction Temperature
125
C012
Figure 8. Maximum Switching Frequency
vs Junction Temperature
1.6
40
VVS = 3.7 V
Driver Low-6LGH 5HVLVWDQFH
Driver Source Current (mA)
1.5
35
30
25
20
VVS = 4.35 V
15
10
1.4
1.3
1.2
1.1
1
0.9
0.8
0.7
0.6
-50
-25
0
25
50
75
100
Junction Temperature (°C)
125
-50
-25
0
25
50
75
100
Junction Temperature (°C)
C008
Figure 9. Driver Output Source Current
vs Junction Temperature
125
C009
Figure 10. Driver Pull Down Resistance
vs Junction Temperature
Overvoltage Protection Threshold (V)
4.68
4.66
4.64
4.62
4.6
4.58
4.56
4.54
-50
-25
0
25
50
75
Junction Temperature (°C)
100
125
C010
Figure 11. Over Voltage Protection Threshold
vs Junction Temperature
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UCC28722
SLUSBL7B – DECEMBER 2013 – REVISED OCTOBER 2015
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7 Detailed Description
7.1 Overview
The UCC28722 is a flyback power supply controller that provides accurate voltage and constant current
regulation with primary-side feedback, eliminating the need for opto-coupler feedback circuits. The controller
operates in discontinuous conduction mode with valley-switching to minimize switching losses. The modulation
scheme is a combination of frequency and primary peak current modulation to provide high conversion efficiency
across the load range. The control law provides a wide-dynamic operating range of output power, which allows
the power designer to achieve less than 75-mW of stand-by power.
During low-power operating ranges, the device has power-management features to reduce the device operating
current at operating frequencies below 28 kHz. Accurate voltage and constant current regulation, fast dynamic
response, and fault protection are achieved with primary-side control. A complete charger solution can be
realized with a straightforward design process, low cost and low component count.
7.2 Functional Block Diagram
VDD
2
GND
5
OC FAULT
OV FAULT
TSD/SD FAULT
LINE FAULT
POWER
& FAULT
MANAGEMENT
UVLO
21 V / 8 V
VDD
5V
4.05 V + VCVS
35mA
+
CONTROL
LAW
E/A
VS
6
SAMPLER
+
VALLEY
SWITCHING
4
CS
20 kO
S
Q
R
Q
+
SECONDARY
TIMING
DETECT
DRV
1 / fSW
OV FAULT
VOVP
3
6V
VCST
CURRENT
REGULATION
VCST
LEB
IVSLS
LINE
SENSE
IVSLS / KLC
IVSLS
+
10 kO
+
LINE
FAULT
2.2 V / 0.80 V
OC FAULT
1.5 V
VCVS = ICBC x 3 kO
CABLE
COMPENSATION
0 V-3 V
+
ICBC
CBC
8
28 kO
1
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7.3 Feature Description
7.3.1 Device Bias Voltage Supply (VDD)
The VDD pin is connected to a bypass capacitor to ground. The VDD turnon UVLO threshold is 21 V and turnoff
UVLO threshold is 7.7 V, with an available operating range up to 35 V on VDD. The USB charging specification
requires the output current to operate in constant-current mode from 5 V to a minimum of 2 V, which is easily
achieved with a nominal VDD of approximately 22 V. The additional VDD headroom (up to 35 V) allows for VDD
to rise due to the leakage energy delivered to the VDD capacitor in high-load conditions.
NOTE
It is possible for the start-up resistor to supply more current to the VDD node than the IC
will consume at higher bulk input voltages. A Zener diode clamp is required on the VDD
pin to keep the VDD pin voltage within limits if this is the case.
7.3.2 Ground (GND)
There is one ground reference external to the device for the base drive current and analog signal reference. TI
recommends placing the VDD bypass capacitor close to GND and VDD with short traces to minimize noise on
the VS and CS signal pins.
7.3.3 Voltage-Sense (VS)
The VS pin is connected to a resistor divider from the auxiliary winding to ground. The output-voltage feedback
information is sampled at the end of the transformer secondary current demagnetization time to provide an
accurate representation of the output voltage. Timing information for achieving valley-switching and to control the
duty cycle of the secondary transformer current is determined by the waveform on the VS pin. Avoid placing a
filter capacitor on this input because it would interfere with accurate sensing of this waveform.
The VS pin also senses the bulk capacitor voltage to provide for AC-input run and stop thresholds, and to
compensate the current-sense threshold across the AC-input range. During the transistor on-time, the VS pin is
clamped to approximately 250 mV below GND and the current out of the VS pin is sensed. For the AC-input run
and stop function, the run threshold on VS is 225 µA and the stop threshold is 80 µA. The values for the auxiliary
voltage divider upper-resistor (RS1) and lower-resistor (RS2) can be determined by Equation 1 and Equation 2.
RS1 =
VIN(run) ´ 2
NPA ´ IVSL(run)
where
•
•
•
RS2 =
NPA is the transformer primary-to-auxiliary turns ratio.
VIN(run) is the AC RMS voltage to enable turnon of the controller (run).
IVSL(run) is the run-threshold for the current pulled out of the VS pin during the switch on-time (see Electrical
Characteristics).
(1)
RS1 ´ VVSR
NAS ´ (VOCV + VF ) - VVSR
where
•
•
•
•
•
VOCV is the converter regulated output voltage.
VF is the output rectifier forward drop at near-zero current.
NAS is the transformer auxiliary to secondary turns ratio.
RS1 is the VS divider high-side resistance.
VVSR is the CV regulating level at the VS input (see Electrical Characteristics).
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Feature Description (continued)
7.3.4 Base Drive (DRV)
The DRV pin is connected to the NPN transistor base pin. The driver provides a base drive signal limited to 7 V.
The turn-on characteristic of the driver is a 19-mA to 37-mA current source that is scaled with the current sense
threshold dictated by the operating point in the control scheme. When the minimum current sense threshold is
being used, the base drive current is also at its minimum value. As the current sense threshold is increased to
the maximum, the base drive current scales linearly to its maximum of 35-mA typical. The turn-off current is
determined by the low-side driver RDS(on).
7.3.5 Current Sense (CS)
The current-sense pin is connected through a series resistor (RLC) to the current-sense resistor (RCS). The
current-sense threshold is 0.78 V for IPP(max) and 0.19 V for IPP(min). The series resistor RLC provides the function
of feedforward line compensation to eliminate change in IPP due to change in di/dt and the propagation delay of
the internal comparator and NPN transistor turn-off time. There is an internal leading-edge blanking time of
approximately 300 ns to eliminate sensitivity to the turnon current spike. It should not be necessary to place a
bypass capacitor on the CS pin. The value of RCS is determined by the target output current in constant current
(CC) regulation. The values of RCS and RLC can be determined by Equation 3 and Equation 4. The term ηXFMR is
intended to account for the energy stored in the transformer but not delivered to the secondary, which includes
transformer resistance and core loss, bias power, and primary-to-secondary leakage ratio.
7.3.5.1 Example
With a transformer core and winding loss of 5%, primary-to-secondary leakage inductance of 3.5%, and bias
power-to-output power ratio of 1.5%, the ηXFMR value is approximately: 1 – 0.05 – 0.035 – 0.015 = 0.9.
V
´ NPS
RCS = CCR
´ hXFMR
2IOCC
where
•
•
•
•
VCCR is a current regulation constant (see Electrical Characteristics).
NPS is the transformer primary-to-secondary turns ratio (a ratio of 13 to 15 is recommended for 5-V output).
IOCC is the target output current in constant-current regulation.
ηXFMR is the transformer efficiency.
(3)
spacer
RLC =
KLC ´ RS1 ´ RCS ´ tD ´ NPA
LP
where
•
•
•
•
•
•
10
RS1 is the VS pin high-side resistor value.
RCS is the current-sense resistor value.
t D is the current-sense delay including NPN transistor turn-off delay, add approximately 50 ns to transistor
delay.
NPA is the transformer primary-to-auxiliary turns ratio.
LP is the transformer primary inductance.
KLC is a current-scaling constant (see Electrical Characteristics).
(4)
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Feature Description (continued)
7.3.6 Cable Compensation (CBC)
The cable compensation pin is connected to a resistor to ground to program the amount of output voltage
compensation to offset cable resistance. The cable compensation block provides a 0-V to 3-V voltage level on
the CBC pin corresponding to IOCC(max) output current. Connecting a resistance from CBC to GND programs a
current that is summed into the VS feedback divider, increasing the regulation voltage as IOUT increases. There is
an internal series resistance of 28 kΩ to the CBC pin that sets a maximum cable compensation of a 5-V output to
400 mV when CBC is shorted to ground. The CBC resistance value can be determined by Equation 5.
RCBC =
VCBC(max) ´ 3 kW ´ (VOCV + VF )
VVSR ´ VOCBC
- 28 kW
where
•
•
•
•
•
VOCV is the regulated output voltage.
VF is the diode forward voltage in V.
VOCBC is the target cable compensation voltage at the output terminals.
VCBC(max) is the maximum voltage at the cable compensation pin at the maximum converter output current (see
Electrical Characteristics).
VVSR is the CV regulating level at the VS input (see Electrical Characteristics).
(5)
7.4 Device Functional Modes
7.4.1 Primary-Side Voltage Regulation
Figure 12 illustrates a simplified flyback convertor with the main voltage regulation blocks of the device shown.
The power train operation is the same as any DCM flyback circuit but accurate output voltage and current
sensing is the key to primary-side control.
Bulk Voltage
Secondary
Winding
Primary Winding
VOUT
COUT RLOAD
Timing
Aux
Winding
RS1
VS
RS2
Discriminator
and
Sampler
VCL
Zero Crossings
Control
Law
Minimum
Period
and Peak
Primary
Current
GD
DRV
CS
RCS
UDG-13093
The main voltage regulation blocks are shown.
Figure 12. Simplified Flyback Convertor
In primary-side control, the output voltage is sensed on the auxiliary winding during the transfer of transformer
energy to the secondary. As shown in Figure 13, it is clear there is a down slope representing a decreasing total
rectifier (VF) and resistance voltage drop (ISRS) as the secondary current decreases to zero. To achieve an
accurate representation of the secondary output voltage on the auxiliary winding, ensure that the discrimantor:
• Reliably recognizes the leakage inductance reset, ringing, and ingores
• Continuously samples the auxiliary voltage during the down slope after the ringing is diminished
• Captures the error signal at the time the secondary winding reaches zero current
The internal reference on VS is 4.05 V. Temperature compensation on the VS reference voltage of –0.8-mV/°C
offsets the change in the output rectifier forward voltage with temperature. The resistor divider is selected as
outlined in the VS pin description.
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Device Functional Modes (continued)
VS Sample
(VOUT
+ VF + IS ´ RS ) ´ NA
NS
0V
- (VBLK ) ´ NA
NP
Figure 13. Auxiliary Winding Voltage
The UCC28722 includes a VS signal sampler that uses discrimination methods to ensure an accurate sample of
the output voltage from the auxiliary winding. There are some conditions that must be met on the auxiliary
winding signal to ensure reliable operation. These conditions are the reset time of the leakage inductance and
the duration of any subsequent leakage inductance ring. Refer to Figure 14 for a detailed illustration of waveform
criteria to ensure a reliable sample on the VS pin. The first detail to examine is the duration of the leakage
inductance reset pedestal, tLK_RESET in Figure 14. Because this can mimic the waveform of the secondary current
decay followed by a sharp downslope, it is important to keep the leakage reset time less than 600 ns for IPRI
minimum, and less than 2.2 µs for IPRI maximum. The second detail is the amplitude of ringing on the VAUX
waveform following tLK_RESET. The peak-to-peak voltage at the VS pin should be less than approximately 100
mVp-p at least 200 ns before the end of the demagnetization time, tDM. If there is a concern with excessive
ringing, it usually occurs during light or no-load conditions, when tDM is at the minimum. The tolerable ripple on
VS scales up when measured at the auxiliary winding by RS1 and RS2, and is equal to 100 mV × (RS1 + RS2) /
RS2 when measured directly at the auxiliary winding.
tLK_RESET
tSMPL
VS Ring (p-p)
tDM
UDG-12202
Figure 14. Auxiliary Waveform Details
During voltage regulation, the controller operates in frequency modulation mode and amplitude modulation mode
as illustrated in Figure 15. The internal operating frequency limits of the device are 80 kHz, fSW(max) and 650 Hz,
fSW(min). The transformer primary inductance and primary peak current chosen sets the maximum operating
frequency of the converter. The output preload resistor and efficiency at low power determines the converter
minimum operating frequency. There is no stability compensation required for the UCC28722.
12
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Device Functional Modes (continued)
Control Law Profile in Constant Voltage (CV) Mode
80 kHz
fSW(max)
IPP
IPP(Peak Primary Current)
æ 1 ö
fSW(max) ç
÷
è MINP ø
IPP(max)
fSW
28 kHz
FM
AM
IPP(max)/4
FM
fSW(min)
3.3 kHz
0.75 V 1.3 V
2.2 V
3.55 V
Control Voltage, E/A Output (VCL)
5V
UDG-13095
Figure 15. Frequency and Amplitude Modulation Modes During Voltage Regulation
7.4.2 Primary-Side Current Regulation
Timing information at the VS pin and current information at the CS pin allow accurate regulation of the secondary
average current. The control law dictates that as power is increased in CV regulation and approaching CC
regulation the primary-peak current is at IPP(max). Referring to Figure 16, the primary-peak current, turns ratio,
secondary demagnetization time (tDM), and switching period (tSW) determine the secondary average output
current. Ignoring leakage inductance effects, the average output current is given by Equation 6. When the
average output current reaches the regulation reference in the current control block, the controller operates in
frequency modulation mode to control the output current at any output voltage at or below the voltage regulation
target as long as the auxiliary winding can keep VDD above the UVLO turnoff threshold.
IS ´ NS
NP
IPP
tON
tDM
tSW
UDG-12203
Figure 16. Transformer Currents
I
N
t
IOUT = PP ´ P ´ DM
2 NS tSW
(6)
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Device Functional Modes (continued)
5
VOCV
5.25 V
Output Voltage (V)
4.75 V
4
±5%
3
2
1
IOCC
Output Current
UDG-12201
Figure 17. Typical Target Output V-I Characteristic
7.4.3 Valley Switching
The UCC28722 utilizes valley switching to reduce switching losses in the transistor, to reduce induced-EMI, and
to minimize the turnon current spike at the sense resistor. The controller operates in valley-switching in all load
conditions unless the collector voltage (VC) ringing has subsided.
Referring to Figure 18, the UCC28722 operates in a valley-skipping mode in most load conditions to maintain an
accurate voltage or current regulation point and still switch on the lowest available VC.
VC
VDRV
UDG-13091
Figure 18. Valley-Skipping Mode
7.4.4 Start-Up Operation
An external resistor connected from the bulk capacitor voltage (VBLK) to the VDD pin charges the VDD capacitor.
The amount of startup current that is available to charge the VDD capacitor is dependent on the value of this
external startup resistor. Larger values supply less current and increase startup time but at the expense of
increasing standby power and decreasing efficiency at high input voltage and light loading. When VDD reaches
the 21-V UVLO turnon threshold, the controller is enabled, the converter starts switching. The initial three cycles
are limited to IPP(min). After the initial three cycles at minimum IPP(min), the controller responds to the condition
dictated by the control law. The converter will remain in discontinuous mode during charging of the output
capacitors, maintaining a constant output current until the output voltage is in regulation.
14
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Device Functional Modes (continued)
NOTE
It is possible for the startup resistor to supply more current to the VDD node than the IC
will consume at higher bulk input voltages. A Zener diode clamp will be required on the
VDD pin to keep the VDD pin voltage within limits if this is the case.
7.4.5 Fault Protection
The UCC28722 provides comprehensive fault protection. Protection functions include the following:
• Output over-voltage fault
• Input under-voltage fault
• Internal over-temperature fault
• Primary over-current fault
• CS pin fault
• VS pin fault
A UVLO reset and restart sequence applies for all fault protection events.
The output over-voltage function is determined by the voltage feedback on the VS pin. If the voltage sample on
VS exceeds 115% of the nominal VOUT, the device stops switching and the internal current consumption is IFAULT
which discharges the VDD capacitor to the UVLO turnoff threshold. After that, the device returns to the start state
and a start-up sequence ensues.
The UCC28722 always operates with cycle-by-cycle primary peak current control. The normal operating range of
the CS pin is 0.78 V to 0.195 V. There is additional protection if the CS pin reaches 1.5 V. This results in a UVLO
reset and restart sequence.
The line input run and stop thresholds are determined by current information at the VS pin during the transistor
on-time. While the VS pin is clamped close to GND during the transistor on-time, the current through RS1 is
monitored to determine a sample of the bulk capacitor voltage. A wide separation of run and stop thresholds
allows clean start-up and shut-down of the power supply with the line voltage. The run current threshold is
225 µA and the stop current threshold is 80 µA.
The internal over-temperature protection threshold is 165°C. If the junction temperature reaches this threshold
the device initiates a UVLO reset cycle. If the temperature is still high at the end of the UVLO cycle, the
protection cycle repeats.
Protection is included in the event of component failures on the VS pin. If complete loss of feedback information
on the VS pin occurs, the controller stops switching and restarts.
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8 Application and Implementation
NOTE
Information in the following applications sections is not part of the TI component
specification, and TI does not warrant its accuracy or completeness. TI’s customers are
responsible for determining suitability of components for their purposes. Customers should
validate and test their design implementation to confirm system functionality.
8.1 Application Information
The UCC28722 flyback power supply controller provides constant voltage (CV) and constant current (CC) output
regulation to help meet USB-compliant adaptors and charger requirements. This device uses the information
obtained from auxiliary winding sensing (VS) to control the output voltage and does not require optocoupler or
TL431 feedback circuitry. Not requiring optocoupler feedback reduces the component count and makes the
design more cost effective and efficient.
8.2 Typical Application
+ VF -
VBLK
+
CB1
CB2
Np
Ns
COUT
RPL
VOUT
–
RST
- VFA +
VAC
VAUX
RS1
RCBC
Na
RS2
1
CBC
2 VDD
VS 6
GND
5
CS
4
CDD
30V
3
DRV
RLC
RCS
Figure 19. Design Procedure Application Example
16
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Typical Application (continued)
8.2.1 Design Requirements
The design parameters are listed in Table 1.
Table 1. Design Parameters
PARAMETER
TEST CONDITIONS
MIN
NOM
MAX
UNIT
100 (VIN(MIN)) 115/230
240
V
47
64
Hz
INPUT CHARACTERISTICS
VIN
RMS Input Voltage
fLINE
Line Frequency
VIN(RUN)
Brownout Voltage
IOUT = Nom
50/60
70
V
OUTPUT CHARACTERISTICS
VOCV
Output Voltage
VIN = Nom, IOUT = NOM
VRIPPLE
Output Voltage Ripple
VIN = Nom, IO = Max
IOUT
Output Current
VIN = Min to Max
Output OVP
IOUT = Min to Max
4.75
5
1
5.25
V
0.15
V
1.05
A
5.75
V
Transient Response
Load Step (ITRAN = 0.6 A)
(0.1 to 0.6 A) or (0.6 to 0.1 A)
VOΔ= 0.9 V for Calculations
4.1
5
6
V
SYSTEMS CHARACTERISTICS
fMAX
Switching Frequency
ƞ
Full Load Efficiency (115/230 V RMS
input)
70
IOUT = 1 A
kHz
75%
8.2.2 Detailed Design Procedure
This procedure outlines the steps to design a constant-voltage, constant-current flyback converter using the
UCC28722 controller. Refer to Figure 19 for component names and network locations. The design procedure
equations use terms that are defined in Stand-by Power Estimate through Startup Resistance and Startup Time.
8.2.2.1
Stand-by Power Estimate
Assuming no-load stand-by power is a critical design parameter, determine estimated no-load power based on
target converter maximum switching frequency and output power rating.
The followingEquation 7 estimates the stand-by power of the converter.
POUT ´ fMIN
PSB _ CONV =
hSB ´ K AM2 ´ fMAX
(7)
For a typical USB charger application, the bias power during no-load is approximately 2.5 mW. This is based on
25-V VDD and 100-µA bias current. The output preload resistor can be estimated by VOCV and the difference in
the converter stand-by power and the bias power. Equation 8 shows output preload resistance accounts for bias
power estimated at 2.5 mW.
RPL =
VOCV 2
PSB _ CONV - 2.5 mW
(8)
Typical startup resistance values for RSTR range from 1 MΩ to 5MΩ to achieve 2 s startup time. The capacitor
bulk voltage for the loss estimation is the highest voltage for the stand-by power measurement, typically 325 VDC,
see Equation 9.
V 2
PRSTR = BLK
RSTR
(9)
For the total stand-by power estimation add an estimated 2.5 mW for snubber loss to the converter stand-by
power loss, see Equation 10 and Equation 11.
PSB = PSB _ CONV + 2.5 mW
(10)
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PSB = PSB _ CONV + PRSTR + 2.5 mW
(11)
8.2.2.2 Input Bulk Capacitance and Minimum Bulk Voltage
Determine the minimum voltage on the input capacitance, CB1 and CB2 total, in order to determine the maximum
Np to Ns turns ratio of the transformer. The input power of the converter based on target full-load efficiency,
minimum input RMS voltage, and minimum AC input frequency are used to determine the input capacitance
requirement.
Maximum input power is determined based on VOCV, IOCC, and the full-load efficiency target, see Equation 12.
V
´I
PIN = OCV OCC
h
(12)
Equation 13 provides an accurate solution for input capacitance based on a target minimum bulk capacitor
voltage. To target a given input capacitance value, iterate the minimum capacitor voltage to achieve the target
capacitance.
CBULK
æ
æ VBULK(min)
1
2PIN ´ ç 0.25 +
´ arcsin ç
ç 2 ´ VIN(min)
2P
ç
è
è
=
2
2VIN(min) - VBULK(min)2 ´ fLINE
(
)
öö
÷÷
÷÷
øø
(13)
8.2.2.3 Transformer Turns Ratio, Inductance, Primary-Peak Current
The maximum primary-to-secondary turns ratio can be determined by the target maximum switching frequency at
full load, the minimum input capacitor bulk voltage, and the estimated DCM quasi-resonant time.
Initially determine the maximum available total duty cycle of the on time and secondary conduction time based on
target switching frequency and DCM resonant time. For DCM resonant time, assume 500 kHz if you do not have
an estimate from previous designs. For the transition mode operation limit, the period required from the end of
secondary current conduction to the first valley of the VCE voltage is 1/2 of the DCM resonant period, or 1 µs
assuming 500-kHz resonant frequency. DMAX can be determined using Equation 14.
æt
ö
DMAX = 1 - ç R ´ fMAX ÷ - DMAGCC
è 2
ø
(14)
Once DMAX is known, the maximum turns ratio of the primary to secondary can be determined with the equation
below. DMAGCC is defined as the secondary diode conduction duty cycle during constant-current, CC, operation. It
is set internally by the UCC28722 at 0.425. The total voltage on the secondary winding needs to be determined;
which is the sum of VOCV, the secondary rectifier VF, and the cable compensation voltage (VOCBC). For the 5-V
USB charger applications, a turns ratio range of 13 to 15 is typically used, see Equation 15.
DMAX ´ VBULK(min)
NPS(max) =
DMAGCC ´ (VOCV + VF + VOCBC )
(15)
Once an optimum turns ratio is determined from a detailed transformer design, use this ratio for the following
parameters.
The UCC28722 constant-current regulation is achieved by maintaining a maximum DMAG duty cycle of 0.425 at
the maximum primary current setting. The transformer turns ratio and constant-current regulating voltage
determine the current sense resistor for a target constant current.
Since not all of the energy stored in the transformer is transferred to the secondary, a transformer efficiency term
is included in Equation 16. This efficiency number includes the core and winding losses, leakage inductance
ratio, and bias power ratio to rated output power. For a 5-V, 1-A charger example, bias power of 1.5% is a good
estimate. An overall transformer efficiency of 0.9 is a good estimate to include 3.5% leakage inductance, 5% core
and winding loss, and 1.5% bias power.
V
´ NPS
RCS = CCR
´ hXFMR
2IOCC
(16)
18
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The primary transformer inductance can be calculated using the standard energy storage equation for flyback
transformers. Primary current, maximum switching frequency and output and transformer power losses are
included in Equation 17 and Equation 18. Initially determine transformer primary current.
Primary current is simply the maximum current sense threshold divided by the current sense resistance.
VCST(max)
IPP(max) =
RCS
LP =
(17)
2 (VOCV + VF + VOCBC ) ´ IOCC
hXFMR ´ IPP(max)2 ´ fMAX
(18)
The secondary winding to auxiliary winding transformer turns ratio (NAS) is determined by the lowest target
operating output voltage in constant-current regulation and the VDD UVLO of the UCC28722. There is additional
energy supplied to VDD from the transformer leakage inductance energy which allows a lower turns ratio to be
used in many designs Equation 19
VDD(off ) + VFA
NAS =
VOCC + VF
(19)
8.2.2.4 Transformer Parameter Verification
The transformer turns ratio selected affects the transistor VC and secondary rectifier reverse voltage so these
should be reviewed. The UCC28722 does require a minimum on time of the transistor (tON) and minimum DMAG
time (tDMAG) of the secondary rectifier in the high line, minimum load condition. The selection of fMAX, LP and RCS
affects the minimum tON and tDMAG.
The secondary rectifier and transistor voltage stress can be determined by Equation 20.
VREV =
VIN(max) ´ 2
NPS
+ VOCV + VOCBC
(20)
For the transistor VC voltage stress, Equation 21, an estimated leakage inductance voltage spike (VLK) needs to
be included.
VCPK = ( VIN(max) ´ 2 ) + ( VOCV + VF + VOCBC ) ´ NPS + VLK
(21)
Equation 22 and Equation 23 are used to determine if the minimum tON target of 300 ns and minimum tDMAG
target of 1.2 µs is achieved.
IPP(max ) ´ VCST(min )
LP
tON(min ) =
´
VCST(max )
VIN(max ) ´ 2
(22)
tDMAG(min ) =
tON ´ VIN(max ) ´ 2
NPS ´ (VOCV + VF )
(23)
8.2.2.5 Output Capacitance
The output capacitance value is typically determined by the transient response requirement from no-load. For
example, in some USB charger applications there is a requirement to maintain a minimum VO of 4.1 V with a
load-step transient of 0 mA to 500 mA . Equation 24 assumes that the switching frequency can be at the
UCC28722 minimum of fSW(min).
COUT
æ
ö
1
ITRAN ç
+ 150 ms ÷
ç fSW(min)
÷
è
ø
=
VOD
(24)
Another consideration of the output capacitor is the ripple voltage requirement which is reviewed based on
secondary peak current and ESR. A margin of 20% is added to the capacitor ESR requirement in Equation 25.
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VRIPPLE ´ 0.8
IPP(max) ´ NPS
(25)
8.2.2.6 VDD Capacitance, CDD
The capacitance on VDD needs to supply the device operating current until the output of the converter reaches
the target minimum operating voltage in constant-current regulation. At this time, the auxiliary winding can
sustain the voltage to the UCC28722. The total output current available to the load and to charge the output
capacitors is the constant-current regulation target, IOCC. Equation 26 assumes all the output current of the
flyback is available to charge the output capacitance from 0 V to VOCC. If the converter is going to be loaded
during the time the output is ramping from 0 V to VOCC, that load current must be subtracted for the available
output current limit value, IOCC. There is 1 V of margin added to VDD in the calculation.
COUT ´ VOCC
IRUN + IDRS(max) ´ 1 - Dmagcc ) ´
IOCC
CDD =
(VDD(on) - VDD(off ) ) - 1 V
(26)
(
(
))
NOTE
The typical ceramic capacitor of sufficient ratings for use here varies considerably in
effective capacitance as the voltage across the capacitor changes. As the capacitor
voltage increases beyond 25% of its rated voltage, the effective capacitance can become
significantly less than the nominal capacitance at zero bias. This equation calculated the
effective capacitance needed over the 8V to 21V range, not the nominal zero bias
capacitance required. Evaluation of the particular capacitor chosen for this function is
strongly recommended to ensure adequate capacitance over the 8V to 21V range.
8.2.2.7 VS Resistor Divider, Line Compensation, and Cable Compensation
The VS divider resistors determine the output voltage regulation point of the flyback converter, also the high-side
divider resistor (RS1) determines the line voltage at which the controller enables continuous DRV operation. RS1
is initially determined based on transformer auxiliary to primary turns ratio and desired input voltage operating
threshold in Equation 27.
RS1 =
VIN(run) ´ 2
NPA ´ IVSL(run)
(27)
The low-side VS pin resistor is selected based on desired VO regulation voltage in Equation 28.
RS1 ´ VVSR
RS2 =
NAS ´ (VOCV + VF ) - VVSR
(28)
The UCC28722 can maintain tight constant-current regulation over input line by utilizing the line compensation
feature. The line compensation resistor (RLC) value is determined by current flowing in RS1 and expected base
drive and transistor turnoff delay in Equation 29. Assume a 50-ns internal delay in the UCC28722.
K ´ RS1 ´ RCS ´ tD ´ NPA
RLC = LC
LP
(29)
The UCC28722 has adjustable cable drop compensation. The resistance for the desired compensation level at
the output terminals can be determined using Equation 30.
RCBC =
20
VCBC(max) ´ 3 kW ´ (VOCV + VF )
VVSR ´ VOCBC
- 28 kW
(30)
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8.2.2.8 Startup Resistance and Startup Time
When the VDD capacitor is known, there is a tradeoff to be made between startup time and overall standby input
power to the converter. Faster startup time requires a smaller startup resistance, which results in higher standby
input power in Equation 31.
RSTR =
2 ´ VIN(min)
VDD(on) ´ CDD
ISTART +
TSTR
(31)
8.2.3 Application Curves
78%
76%
Efficiency
74%
72%
70%
68%
66%
Efficiency at 115 V RMS
Efficiency at 230 V RMS
64%
10% 20% 30% 40% 50% 60% 70% 80% 90% 100%
Output Power
C001
Figure 20. Efficiency
Figure 21. Output at Start-up 115-V RMS, No Load
Figure 22. Output at Start-up 115-V RMS, 5-Ω Load
Figure 23. Output at Start-up 230-V RMS, No Load
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SPACE
Figure 24. Output at Start-up 230-V RMS, 5-Ω Load
CH1 = VOCV with 5-V offset, CH2 = IOUT
Figure 26. Load Transient (0.6- to 0.1-A Load Step)
CH1 = VOCV with 5-V offset, CH2 = IOUT
Figure 25. Load Transients (0.1- to 0.6-A Load Step)
Figure 27. Output Ripple CH1 = VOCV at Supply Output
9 Power Supply Recommendations
The UCC28722 is intended for AC/DC adapters and chargers with input voltage range of 85 VAC(rms) to 265
VAC(rms) using Flyback topology. This device can be used in other applications and converter topologies with
different input voltages. Ensure that all voltages and currents are within the Recommended Operating Conditions
and Absolute Maximum Ratings of the device. To maintain output current regulation over the entire input voltage
range, design the converter to operate close to fMAX when in full-load conditions. To improve thermal
performance, increase the copper area connected to GND pins.
22
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10 Layout
10.1 Layout Guidelines
•
•
•
•
•
•
•
•
•
•
•
•
High frequency bypass Capacitor C7 should be placed arcoss Pin 2 and 5 as close as you can get it to the
pins.
Resistor R15 and C7 form a low pass filter and the connection of R15 and C7 should be as close to the VDD
pin as possible.
C9 should be put as close to CS pin and R10 as possible. This forms a low pass filter with R10.
The connection for C9 and R10 should be as close to the CS pin as possible.
C9 may not be required in all designs. However, it is wise to put a place holder for it in your design.
The VS pin controls the output voltage through the transformer turns ratio and the voltage divider of R7 and
R9. The trace with between the R7, R9 and VS pin should be as short as possible to reduce and eliminate
possible EMI coupling.
The IC ground and power ground should meet at the return of the bulk capacitors (C4 and C5). Ensure that
high frequency and high current from the power stage does not go through the signal ground
– The high frequency and high current path that you need to be cautious of on the primary is C4, C5 +,
T1(P1,P2), Q1e, Q1c, R13 to the return of C4 and C5.
Keep all high current loops as short as possible.
Keep all high current and high frequency traces away from or perpendicular to other traces in the design.
Traces on the voltage clamp formed by D1, R1, D4 and C4 as short as possible.
C4 return needs to be as close to the bulk capacitor supply as possible. This reduces the magnitude of dv/dt
caused by large di/dt.
Avoid mounting semiconductors under magnetics.
D4
C1
TP4
VOUT-
R4
10.2 Layout Example
C5
C3
C8
R2
R15
R11
D3
U1
J1
R3
C6
R9
C7
R8
Q1
C9
R7
TP3
D2
R12
JMP2
VOUT+
D5
C4
R6
R13
D1
T1
Line
R5
R1
JMP3
J2
C2
TP1
L1
TP3
Nuetral
R10
Figure 28. PCB Layout Example
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Figure 29. 5-W USB Adapter Schematic
24
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11 Device and Documentation Support
11.1 Device Support
11.1.1 Device Nomenclature
11.1.1.1 Definition of Terms
11.1.1.1.1 Capacitance Terms in Farads
•
•
•
CBULK: total input capacitance of CB1 and CB2.
CDD: minimum required capacitance on the VDD pin.
COUT: minimum output capacitance required.
11.1.1.1.2 Duty Cycle Terms
•
•
DMAGCC: secondary diode conduction duty cycle in CC, 0.425.
DMAX: transistor on-time duty cycle.
11.1.1.1.3 Frequency Terms in Hertz
•
•
•
•
fLINE: minimum line frequency.
fMAX: target full-load maximum switching frequency of the converter.
fMIN: minimum switching frequency of the converter, add 15% margin over the fSW(min) limit of the device.
fSW(min): minimum switching frequency (see Electrical Characteristics).
11.1.1.1.4 Current Terms in Amperes
•
•
•
•
•
•
IOCC: converter output constant-current target.
IPP(max): maximum transformer primary current.
ISTART: start-up bias supply current (see Electrical Characteristics).
ITRAN : required positive load-step current.
IVSL(run): VS pin run current (see Electrical Characteristics).
IDRS: Driver source current (see Electrical Characteristics).
11.1.1.1.5 Current and Voltage Scaling Terms
•
•
KAM: maximum-to-minimum peak primary current ratio (see Electrical Characteristics).
KLC: current-scaling constant (see Electrical Characteristics).
11.1.1.1.6 Transformer Terms
•
•
•
•
LP: transformer primary inductance.
NAS: transformer auxiliary-to-secondary turns ratio.
NPA: transformer primary-to-auxiliary turns ratio.
NPS: transformer primary-to-secondary turns ratio.
11.1.1.1.7 Power Terms in Watts
•
•
•
•
•
PIN: converter maximum input power.
POUT: full-load output power of the converter.
PRSTR: VDD start-up resistor power dissipation.
PSB: total stand-by power.
PSB_CONV: PSB minus start-up resistor and snubber losses.
11.1.1.1.8 Resistance Terms in Ω
•
•
•
•
RCS: primary current programming resistance.
RESR: total ESR of the output capacitor(s).
RPL: preload resistance on the output of the converter.
RS1: high-side VS pin resistance.
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Device Support (continued)
•
•
RS2: low-side VS pin resistance.
RSTR : startup resistance.
11.1.1.1.9 Timing Terms in Seconds
•
•
•
•
•
tD: current-sense delay including transistor turnoff delay; add 50 ns to transistor delay.
tDMAG(min): minimum secondary rectifier conduction time.
tON(min): minimum transistor on time.
tR: resonant frequency during the DCM (discontinuous conduction mode) time.
tST: startup time
11.1.1.1.10 Voltage Terms in Volts
•
•
•
•
•
•
•
•
•
•
•
•
•
•
•
•
•
•
•
VBLK: highest bulk capacitor voltage for stand-by power measurement.
VBULK(min): minimum voltage on CB1 and CB2 at full power.
VOCBC: target cable compensation voltage at the output terminals.
VCBC(max): maximum voltage at the CBC pin at the maximum converter output current (see Electrical
Characteristics).
VCCR: constant-current regulating voltage (see Electrical Characteristics).
VCST(max): CS pin maximum current-sense threshold (see Electrical Characteristics).
VCST(min): CS pin minimum current-sense threshold (see Electrical Characteristics).
VDD(off): UVLO turnoff voltage (see Electrical Characteristics).
VDD(on): UVLO turnon voltage (see Electrical Characteristics).
VOΔ: output voltage drop allowed during the load-step transient.
VCPK: peak transistor collector to emitter voltage at high line.
VF: secondary rectifier forward voltage drop at near-zero current.
VFA: auxiliary rectifier forward voltage drop.
VLK: estimated leakage inductance energy reset voltage.
VOCV: regulated output voltage of the converter.
VOCC: target lowest converter output voltage in constant-current regulation.
VREV: peak reverse voltage on the secondary rectifier.
VRIPPLE: output peak-to-peak ripple voltage at full-load.
VVSR: CV regulating level at the VS input (see Electrical Characteristics).
11.1.1.1.11 AC Voltage Terms in VRMS
•
•
•
VIN(max): maximum input voltage to the converter.
VIN(min): minimum input voltage to the converter.
VIN(run): converter input start-up (run) voltage.
11.1.1.1.12 Efficiency Terms
•
•
•
ηSB: estimated efficiency of the converter at no-load condition, not including start-up resistance or bias losses.
ηSB: For a 5-V USB charger application, 60% to 65% is a good initial estimate.
η: converter overall efficiency.
ηXFMR: transformer primary-to-secondary power transfer efficiency.
11.2 Documentation Support
11.2.1 Related Documentation
See the following: UCC28722/UCC28720 5W USB BJT Flyback Design Example, SLUA700
26
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11.3 Community Resources
The following links connect to TI community resources. Linked contents are provided "AS IS" by the respective
contributors. They do not constitute TI specifications and do not necessarily reflect TI's views; see TI's Terms of
Use.
TI E2E™ Online Community TI's Engineer-to-Engineer (E2E) Community. Created to foster collaboration
among engineers. At e2e.ti.com, you can ask questions, share knowledge, explore ideas and help
solve problems with fellow engineers.
Design Support TI's Design Support Quickly find helpful E2E forums along with design support tools and
contact information for technical support.
11.4 Trademarks
E2E is a trademark of Texas Instruments.
All other trademarks are the property of their respective owners.
11.5 Electrostatic Discharge Caution
These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
11.6 Glossary
SLYZ022 — TI Glossary.
This glossary lists and explains terms, acronyms, and definitions.
12 Mechanical, Packaging, and Orderable Information
The following pages include mechanical, packaging, and orderable information. This information is the most
current data available for the designated devices. This data is subject to change without notice and revision of
this document. For browser-based versions of this data sheet, refer to the left-hand navigation.
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PACKAGE OPTION ADDENDUM
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10-Dec-2020
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
(4/5)
(6)
UCC28722DBVR
ACTIVE
SOT-23
DBV
6
3000
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 125
U722
UCC28722DBVT
ACTIVE
SOT-23
DBV
6
250
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 125
U722
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of
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