EVALUATION KIT AVAILABLE
MAX16993
Step-Down Controller with
Dual 2.1MHz Step-Down DC-DC Converters
General Description
The MAX16993 power-management integrated circuit
(PMIC) is a 2.1MHz, multichannel, DC-DC converter designed for automotive applications. The device
integrates three supplies in a small footprint. The device
includes one high-voltage step-down controller (OUT1)
designed to run directly from a car battery and two lowvoltage step-down converters (OUT2/OUT3) cascaded
from OUT1. Under no-load conditions, the MAX16993
consumes only 30µA of quiescent current, making it ideal
for automotive applications.
The high-voltage synchronous step-down DC-DC
controller (OUT1) operates from a voltage up to 36V
continuous and is protected from load-dump transients up
to 42V. There is a pin-selectable frequency option of either
2.1MHz or a factory-set frequency for 1.05MHz, 525kHz,
420kHz, or 350kHz. The low-voltage, synchronous stepdown DC-DC converters run directly from OUT1 and can
supply output currents up to 3A.
The device provides a spread-spectrum enable input
(SSEN) to provide quick improvement in electromagnetic
interference when needed. There is also a SYNC
input for providing an input to synchronize to
an external clock source (see the Selector Guide).
The device includes overtemperature shutdown and
overcurrent limiting. The device also includes individual RESET_ outputs and individual enable inputs.
The individual RESET_ outputs provide voltage
monitoring for all output channels.
Benefits and Features
● High-Efficiency Voltage DC-DC Controller Saves
Power
• 3.5V to 36V Operating Supply Voltage
• Output Voltage: Pin Selectable, Fixed, or
Resistor-Divider Adjustable
• 350kHz to 2.1MHz Operation
• 30μA Quiescent Current with DC-DC
Controller Enabled
● Dual 2.1MHz DC-DC Converters with Integrated
FETs Save Space
• OUT2 and OUT3 are Cascaded from OUT1, Improving Efficiency
• 3A Integrated FETs
• 0.8V to 3.95V Output Voltage
• Fixed or Resistor-Divider-Adjustable Output Voltage
• 180° Out-of-Phase Operation
• Robust for the Automotive Environment
● Current-Mode Architecture with Forced-PWM and
Skip Modes of Operation
• Frequency Synchronization Input/Output Reduces
System Noise
• Individual Enable Inputs and RESET_ Outputs
• Overtemperature and Short-Circuit Protection
• AECQ-100 Qualified
• 32-Pin TQFN-EP (5mm x 5mm x 0.75mm) and
Side-Wettable QFND-EP (5mm x 5mm x 0.8mm)
• -40°C to +125°C Operating Temperature Range
The MAX16993 is available in a 32-pin TQFN/sidewettable QFND-EP package and is specified for operation
over the -40°C to +125°C automotive temperature range.
Applications
● Automotive
● Industrial
19-6684; Rev 14; 12/16
Ordering Information and Selector Guide appear at end of
data sheet.
MAX16993
Step-Down Controller with
Dual 2.1MHz Step-Down DC-DC Converters
Absolute Maximum Ratings
VSUP, EN1 to GND................................................-0.3V to +45V
PV_ to GND..........................................................-0.3V to +6.0V
PV_ to GND..........................................................-0.3V to +6.0V
PV2 to GND, PV2 to PGND2................................-0.3V to +6.0V
PV3 to GND, PV3 to PGND3................................-0.3V to +6.0V
PGND2–PGND3 to GND......................................-0.3V to +0.3V
LX1 to GND................................................-6.0V to VSUP + 6.0V
BST1 to LX1 (Note 1)............................................-0.3V to +6.0V
DH1 to LX1 (Note 1)..................................-0.3V to BST1 + 0.3V
BIAS to GND.........................................................-0.3V to +6.0V
DL1 to GND (Note 1)...................................-0.3V to PV1 + 0.3V
LX2 to PGND2.............................................-0.3V to PV2 + 0.3V
LX3 to PGND3.............................................-0.3V to PV3 + 0.3V
OUT1, CS1, OUT2, OUT3 to GND.......................-0.3V to +6.0V
SYNC to GND..............................................-0.3V to PV_ + 0.3V
FB1, EN2, EN3 to GND........................................-0.3V to +6.0V
RESET_, ERR to GND..........................................-0.3V to +6.0V
CS1 to OUT1.........................................................-0.3V to +0.3V
CSEL1, SSEN to GND..........................................-0.3V to +6.0V
COMP1 to GND..............................................-0.3V to PV + 0.3V
LX2, LX3 Output Short-Circuit Duration.....................Continuous
Continuous Power Dissipation (TA = +70ºC)
Side-Wettable QFND (derate 27mW/ºC above +70ºC)......2160mW
TQFN (derate 34.5mW/ºC above +70ºC)...............2758.6mW
Operating Temperature Range...........................-40ºC to +125°C
Junction Temperature.......................................................+150°C
Storage Temperature Range..............................-65ºC to +150°C
Lead Temperature (soldering, 10s).................................. +300°C
Soldering Temperature (reflow)........................................+260°C
Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these
or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to absolute maximum rating conditions for extended periods may affect
device reliability.
Package Thermal Characteristics (Note 2)
Side-Wettable QFND
Junction-to-Ambient Thermal Resistance (θJA).......... 37°C/W
Junction-to-Case Thermal Resistance (θJC)............. 2.8°C/W
TQFN
Junction-to-Ambient Thermal Resistance (θJA).......... 29°C/W
Junction-to-Case Thermal Resistance (θJC)............. 1.7°C/W
Note 1: Self-protected against transient voltages exceeding these limits for ≤ 50ns under normal operation and loads up to the
maximum rated output current.
Note 2: Package thermal resistances were obtained using the method described in JEDEC specification JESD51-7, using a four-layer
board. For detailed information on package thermal considerations, refer to www.maximintegrated.com/thermal-tutorial.
Electrical Characteristics
(VSUP = 14V, VPV1 = VBIAS, VPV2 = VPV3 = VOUT1; TA = TJ = -40°C to +125°C, unless otherwise noted. Typical values are at
TA = +25°C under normal conditions, unless otherwise noted.) (Note 3)
PARAMETER
SYMBOL
Supply Voltage Startup
Threshold
VSUP,STARTUP
Supply Voltage Range
VSUP
Supply Current
ISUP
Oscillator Frequency
fSW
CONDITIONS
MIN
TYP
MAX
UNITS
VSUP rising
4.25
4.5
4.75
V
Normal operation, after Buck 1 startup
3.5
36
V
VEN1 = VEN2 = VEN3 = 0V
4
15
VEN1 = 5V, VEN2 = VEN3 = 0V (no load)
20
40
2.1
2.2
MHz
2.4
MHz
2.0
SYNC Input Frequency
Range
1.7
Spread-Spectrum Range
BIAS Regulator Voltage
PV_ POR
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VBIAS
VSSEN = VGND
0
VSSEN = VBIAS
+6
%
6V ≤ VSUP ≤ 42V, no switchover
4.6
5.0
5.4
VBIAS falling
2.5
2.7
2.9
Hysteresis
0.45
µA
V
V
Maxim Integrated │ 2
MAX16993
Step-Down Controller with
Dual 2.1MHz Step-Down DC-DC Converters
Electrical Characteristics (continued)
(VSUP = 14V, VPV1 = VBIAS, VPV2 = VPV3 = VOUT1; TA = TJ = -40°C to +125°C, unless otherwise noted. Typical values are at
TA = +25°C under normal conditions, unless otherwise noted.) (Note 3)
PARAMETER
SYMBOL
CONDITIONS
MIN
TYP
MAX
UNITS
OUT1: HIGH-VOLTAGE SYNCHRONOUS STEP-DOWN DC-DC CONTROLLER
OUT1 Switching Frequency
Voltage
fSW1
VOUT1
FB1 Regulation Voltage
Internally generated
(see the Selector
Guide)
Fixed option
(see the Selector
Guide)
VCSEL1 = VGND
2100
VCSEL1 = VBIAS
(factory option)
1050
VCSEL1 = VBIAS
(factory option)
525
VCSEL1 = VBIAS
(factory option)
420
VCSEL1 = VBIAS
(factory option)
350
VFB1 = VGND
3.3
VFB1 = VBIAS
(factory option)
5.0
VFB1 = VBIAS
(factory option)
3.15
Adjustable option (see the Selector Guide)
Error Amplifier
Transconductance
gMEA
Voltage Accuracy
VOUT1
5.5V ≤ VSUP ≤ 18V, 0 < VLIM1 < 75mV,
PWM mode
kHz
V
0.985
1.0
1.019
V
300
700
1200
µS
+2.5
%
-2.0
DC Load Regulation
PWM mode
0.02
%/A
DC Line Regulation
PWM mode
0.03
%/V
OUT1 Discharge Resistance
VEN1 = VGND or VSUP
100
200
High-Side Output Drive
Resistance
VDH1 rising, IDH1 = 100mA
2
4
VDH1 falling, IDH1 = 100mA
1
4
Low-Side Output Drive
Resistance
VDL1 rising, IDL1 = 100mA
2.5
5
VDL1 falling, IDL1 = 100mA
1.5
3
Ω
Ω
Ω
Output Current-Limit
Threshold
VLIM1
CSI – OUT1
100
120
150
mV
Skip Current Threshold
ISKIP
CS1 – OUT1, no load
10
35
60
mV
Soft-Start Ramp Time
LX_ Leakage Current
VLX1 = VSUP
Duty-Cycle Range
PWM mode
Minimum On-Time
OUT1 OV Threshold
www.maximintegrated.com
107
4
ms
0.01
µA
97.2
%
60
75
ns
110
113
%
Maxim Integrated │ 3
MAX16993
Step-Down Controller with
Dual 2.1MHz Step-Down DC-DC Converters
Electrical Characteristics (continued)
(VSUP = 14V, VPV1 = VBIAS, VPV2 = VPV3 = VOUT1; TA = TJ = -40°C to +125°C, unless otherwise noted. Typical values are at
TA = +25°C under normal conditions, unless otherwise noted.) (Note 3)
PARAMETER
SYMBOL
CONDITIONS
MIN
TYP
MAX
UNITS
5.5
V
5
µA
OUT2 AND OUT3: LOW-VOLTAGE SYNCHRONOUS STEP-DOWN DC-DC CONVERTERS
Supply Voltage Range
VSUP
2.7
Supply Current
IPV_
VEN_ = 5V, no load
VOUT
0A ≤ ILOAD ≤ IMAX, PWM mode
-3.0
Adjustable mode, IOUT2 = 0mA
0.806
0.815
0A ≤ ILOAD ≤ IMAX (PWM mode)
-1.5
-1.0
0A ≤ ILOAD ≤ IMAX (PWM mode, low gain,
see the Selector Guide)
-2.5
-1.7
0.1
Skip Mode Peak Current
Voltage Accuracy
0.2 x ILMAX
Feedback-Voltage Accuracy
Load Regulation
mA
+3.0
%
0.824
V
%
LX_ On-Resistance High
ILX_ = -800mA
70
110
mΩ
LX_ On-Resistance Low
ILX_ = 800mA
50
90
mΩ
Current-Limit Threshold
LX_ Rise/Fall Time
ILMAX
IMAX = 3.0A option (see the Selector Guide)
5.0
5.6
IMAX = 1.5A option (see the Selector Guide)
2.5
3.0
PV2 = PV3 = 3.3V, IOUT_ = 2A
Soft-Start Ramp Time
LX_ Leakage Current
Duty-Cycle Range
OUT2, OUT3 Active
Timeout Period
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ns
2.5
ms
15
RESET_
OUT1 Active Timeout Period
4
0.01
PWM mode
LX_ Discharge Resistance
Reset Threshold
A
µA
100
%
22
48
Ω
Rising (relative to nominal output voltage)
92
95
98
Falling (relative to nominal output voltage)
90
92
95
See the Selector Guide
(16,384 clocks)
7.8
See the Selector Guide
(8192 clocks)
3.9
See the Selector Guide
(4096 clocks)
1.9
See the Selector Guide
(256 clocks)
0.1
See the Selector Guide
(16,384 clocks)
7.8
See the Selector Guide
(8192 clocks)
3.9
See the Selector Guide
(4096 clocks)
1.9
See the Selector Guide
(256 clocks)
0.1
%
ms
ms
Maxim Integrated │ 4
MAX16993
Step-Down Controller with
Dual 2.1MHz Step-Down DC-DC Converters
Electrical Characteristics (continued)
(VSUP = 14V, VPV1 = VBIAS, VPV2 = VPV3 = VOUT1; TA = TJ = -40°C to +125°C, unless otherwise noted. Typical values are at
TA = +25°C under normal conditions, unless otherwise noted.) (Note 3)
PARAMETER
Output Low Level
Propagation Time
ERR
Output Low Level
SYMBOL
CONDITIONS
MIN
ISINK = 3mA
TYP
MAX
UNITS
0.1
0.2
V
OUT1, 5% below threshold
5
10
20
µs
OUT2/OUT3, 5% below threshold
2
4
8
µs
0.1
0.2
V
ISINK = 3mA
THERMAL OVERLOAD
Thermal-Warning
Temperature
+150
°C
Thermal-Shutdown
Temperature
+170
°C
Thermal-Shutdown
Hysteresis
15
°C
ENABLE INPUTS (EN_)
Input High
VEN_ rising
1.6
VEN_ = 5V
0.5
Input High
SYNC input option
(see the Selector Guide)
1.8
Input Low
SYNC input option
(see the Selector Guide)
Input Current
SYNC input option (see the Selector
Guide); VSYNC = 5V
Hysteresis
EN Input Current
1.8
2.0
V
2.0
µA
0.2
1.0
V
SYNCHRONIZATION I/O (SYNC)
V
50
Pulldown Resistance
0.8
V
80
µA
100
kΩ
LOGIC INPUTS (CSEL1, SSEN)
Input High
1.4
Input Low
Input Current
TA = +25°C
V
0.5
V
2
µA
Note 3: All units are 100% production tested at TA = +25°C. All temperature limits are guaranteed by design.
www.maximintegrated.com
Maxim Integrated │ 5
MAX16993
Step-Down Controller with
Dual 2.1MHz Step-Down DC-DC Converters
Typical Operating Characteristics
(VSUP = 14V, TA = +25°C, unless otherwise noted)
40
30
PWM MODE
5.015
TA = +25ºC
5.010
5.005
4.990
0
1.00E-06
1.00E-04
1.00E-02
1.00E+00
0
1
2
3
TA = +25ºC
100.1
100.0
99.9
99.8
TA = -40ºC
99.7
15
20
25
30
35
100.0
99.8
99.6
99.0
40
IOUT1 = 3.75A
5.010
5.005
5.000
4.995
4.990
0
5
10
15
20
25
100.3
100.1
50
100
TEMPERATURE (ºC)
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30
35
99.9
99.5
40
BUCK 2 EFFICIENCY
90
80
70
150
60
50
PWM MODE
40
30
0
1.00E-06
3.19
0
5
10
15
20
fSW = 2.1MHz,
VSUP = 14V,
VPV2 = 5.0V,
VOUT2 = 3.15V
1.00E-04
25
30
35
40
1.00E-02
IOUT3 (A)
1.00E+00
BUCK 2 LOAD REGULATION (PWM MODE)
VPV2 = 5.0V, IMAX = 1.5A, VOUT2 = 3.15V
3.18
3.17
3.16
SKIP MODE
10
0
6
VSUP (V)
20
4.985
5
99.7
100
EFFICIENCY (%)
VOUT1 (V)
5.015
4
100.5
VSUP (V)
VOUT1 vs. TEMPERATURE
-50
3
100.7
VOUT2 (V)
10
2
VOUT1 = 3.3V
100.9
99.4
5.020
4.980
MAX16993 toc05
100.2
MAX16993 toc07
5.025
VOUT1 = 5.0V
MAX16993 toc08
5
1
BUCK 1 LINE REGULATION (SKIP MODE)
99.2
0
0
IOUT1 (A)
100.4
VSUP (V)
5.030
4.90
6
5
100.6
99.6
99.5
4
VOUT1 (% NOMINAL)
TA = +125ºC
TA = -40ºC
4.94
BUCK 1 LINE REGULATION (SKIP MODE)
100.8
VOUT1 (% NOMINAL)
VOUT1 (% NOMINAL)
100.3
100.2
101.0
MAX16993 toc04
VOUT1 = 5.0V
4.98
IOUT1 (A)
BUCK 1 LINE REGULATION (PWM MODE)
100.4
5.00
4.92
IOUT1 (A)
100.5
TA = +25ºC
5.02
4.96
TA = -40ºC
4.995
10
TA = +125ºC
5.04
5.000
20
5.06
MAX16993 toc06
SKIP MODE
5.08
VOUT1 (V)
50
MAX16993 toc03
5.020
70
60
TA = +125ºC
5.025
VOUT1 (V)
EFFICIENCY (%)
80
BUCK 1 LOAD REGULATION (SKIP)
5.10
MAX16993 toc09
MAX16993 toc01
90
BUCK 1 LOAD REGULATION (PWM)
5.030
MAX16993 toc02
BUCK 1 EFFICIENCY
100
TA = +125ºC
3.15
3.14
3.13
TA = +25ºC
3.12
3.11
TA = -40ºC
3.10
3.09
3.08
0
0.2
0.4
0.6
0.8
1.0
1.2
1.4
1.6
IOUT2 (A)
Maxim Integrated │ 6
MAX16993
Step-Down Controller with
Dual 2.1MHz Step-Down DC-DC Converters
Typical Operating Characteristics (continued)
(VSUP = 14V, TA = +25°C, unless otherwise noted)
3.325
3.320
3.135
3.130
TA = +25ºC
99.6
99.4
3.315
1.5
2.0
2.5
3.0
3.5
IOUT2 (A)
BUCK 3 EFFICIENCY
90
80
60
SKIP MODE
PWM MODE
40
30
10
1.00E-02
VPV3 = 5.0V, IMAX = 1.5A, VOUT3 = 1.8V
1.82
TA = +25ºC
1.80
1.77
0.2
0.4
0.8
1.0
TA = +25ºC
99.8
1.224
1.222
1.220
1.2
1.4
1.6
0
0.5
1.0
1.810
TA = -40ºC
4.3
VPV3 (V)
4.8
2.0
2.5
3.0
3.5
IOUT3 (A)
VOUT3 vs. TEMPERATURE
IOUT3 = 1.125A
1.805
1.795
1.790
99.7
99.6
1.5
1.800
100.0
99.9
VPV3 = 5.0V
IMAX = 3A
VOUT3 = 1.2V
IOUT3 (A)
TA = +125ºC
100.1
150
1.216
MAX16993 toc16
100.2
100
1.218
VOUT3 (V)
VOUT3 (% NOMINAL)
0.6
50
BUCK 3 LOAD REGULATION (PWM MODE)
1.226
1.214
0
100.3
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0
-50
1.228
TA = -40ºC
1.78
1.00E+00
3.8
1.230
TA = +125ºC
VOUT3 = 1.8V
3.3
3.100
5.7
TEMPERATURE (ºC)
BUCK 3 LINE REGULATION (PWM MODE)
100.4
99.5
5.2
BUCK 3 LOAD REGULATION (PWM MODE)
IOUT3 (A)
100.5
4.7
1.79
fSW = 2.1MHz,
VSUP = 14V,
VPV3 = 5.0V,
VOUT3 = 1.8V
20
1.00E-04
4.2
1.81
50
0
1.00E-06
1.83
VOUT3 (V)
70
3.7
VPV2 (V)
MAX16993 toc13
100
3.2
2.7
MAX16993 toc17
1.0
3.105
VOUT3 (V)
0.5
3.120
3.110
MAX16993 toc14
0
3.125
3.115
TA = -40ºC
99.2
99.0
MAX16993 toc12
3.140
100.2
99.8
IOUT2 = 1.125A
3.145
100.4
100.0
VOUT2 vs. TEMPERATURE
3.150
MAX16993 toc15
VOUT2 (% NOMINAL)
VOUT2 (V)
TA = +125ºC
100.6
3.330
EFFICIENCY (%)
VOUT2 = 3.15V
100.8
3.335
3.310
BUCK 2 LINE REGULATION (PWM MODE)
VOUT2 (V)
VPV2 = 5.0V
IMAX = 3A
VOUT2 = 3.3V
3.340
101.0
MAX16993 toc11
BUCK 2 LOAD REGULATION (PWM MODE)
MAX16993 toc10
3.345
5.3
1.785
1.780
-50
0
50
100
150
TEMPERATURE (ºC)
Maxim Integrated │ 7
MAX16993
Step-Down Controller with
Dual 2.1MHz Step-Down DC-DC Converters
Typical Operating Characteristics (continued)
(VSUP = 14V, TA = +25°C, unless otherwise noted)
MAX16993 toc18
120
SUPPLY CURRENT (µA)
5V/div
5V/div
VOUT1
VRESET1
5V/div
5V/div
VOUT2
VRESET2
80
40
TA = +25ºC
20
5V/div
VRESET3
TA = +125ºC
60
5V/div
VOUT3
VFB = VGND
SKIP MODE
ALL THREE BUCKS ENABLED
MEASURED AT VSUP
100
5V/div
VEN1
SUPPLY CURRENT vs. SUPPLY VOLTAGE
MAX16993 toc19
STARTUP SEQUENCE
(VEN2 = VEN3 = VOUT1)
TA = -40ºC
0
2ms/div
0
5
10
15
20
25
30
35
40
VSUP (V)
VOUT1 = 5.0V, SKIP MODE
ONLY BUCK CONTROLLER ENABLED
60
SUPPLY CURRENT (µA)
LOAD TRANSIENT RESPONSE (PWM MODE)
SUPPLY CURRENT vs. SUPPLY VOLTAGE
50
MAX16993 toc21
MAX16993 toc20
70
TA = +125ºC
VOUT1
100mV/div
40
TA = +25ºC
30
20
10
0
IOUT1
TA = -40ºC
0
5
10
15
20
25
30
1A/div
200µs/div
40
35
VSUP (V)
101
100
99
98
97
8
7
TA = +125ºC
6
5
4
3
TA = -40ºC
2
-50
0
50
100
TEMPERATURE (ºC)
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150
0
0
5
10
15
40
SS DISABLED
SS ENABLED
30
20
10
0
TA = +25ºC
1
50
SPECTRAL ENERGY DENSITY
MAX16993 toc24
VEN1 = VEN2 = VEN3 = VGND
MEASURED AT VSUP
9
60
OUTPUT SPECTRUM (dBµV)
102
MAX16993 toc23
fSW = 2.1MHz
SHUTDOWN CURRENT
vs. SUPPLY VOLTAGE
10
SHUTDOWN CURRENT (µA)
MAX16993 toc22
SWITCHING FREQUENCY (% NOMINAL)
103
fSW vs. TEMPERATURE
20
VSUP (V)
25
30
35
40
-10
1.90 1.95 2.00 2.05 2.10 2.15 2.20 2.25 2.30
FREQUENCY (MHz)
Maxim Integrated │ 8
MAX16993
Step-Down Controller with
Dual 2.1MHz Step-Down DC-DC Converters
PV2
LX2
PGND2
PGND3
LX3
PV3
RESET3
TOP VIEW
RESET2
Pin Configuration
24
23
22
21
20
19
18
17
OUT2 25
16
OUT3
CSEL1 26
15
EN3
SSEN 27
14
EN2
13
OUT1
12
CS1
11
FB1
RESET1 28
MAX16993
GND 29
COMP1 30
ERR 31
EP = GND
DL1
GND
5
6
7
8
EN1
PV1
4
VSUP
3
BST1
2
LX1
1
DH1
+
SYNC 32
10
PV
9
BIAS
TQFN/SIDE-WETTABLE QFND
Pin Description
PIN
NAME
FUNCTION
1
PV1
Supply Input for Buck 1 Low-Side Gate Drive. Connect a ceramic bypass capacitor of at least 0.1µF from PV1
to GND.
2
DL1
Low-Side Gate-Drive Output for Buck 1. DL1 output voltage swings from VGND to VPV1.
3
GND
Power Ground for Buck 1
4
LX1
Inductor Connection for Buck 1. Connect LX1 to the switched side of the inductor. LX1 serves as the lower
supply rail for the DH1 high-side gate drive.
5
DH1
High-Side Gate-Drive Output for Buck 1. DH1 output voltage swings from VLX1 to VBST1.
6
BST1
Bootstrap Capacitor Connection for High-Side Gate Drive of Buck 1. Connect a high-voltage diode between
BIAS and BST1. Connect a ceramic capacitor between BST1 and LX1. See the High-Side Gate-Drive Supply
(BST1) section.
7
VSUP
Supply Input. Bypass VSUP with a minimum 0.1µF capacitor as close as possible to the device.
8
EN1
High-Voltage Tolerant, Active-High Digital Enable Input for Buck 1. Driving EN1 high enables Buck 1.
9
BIAS
5V Internal Linear Regulator Output. Bypass BIAS to GND with a low-ESR ceramic capacitor of
2.2µF minimum value. BIAS provides the power to the internal circuitry. See the Linear Regulator (BIAS)
section.
10
PV
Analog Supply. Connect PV to BIAS through a 10Ω resistor and connect a 1µF ceramic capacitor from PV to
ground.
FB1
Feedback Input for Buck 1. For the fixed output-voltage option, connect FB1 to BIAS for the factory-trimmed
(3.0V to 3.75V or 4.6V to 5.35V) fixed output. Connect FB1 to GND for the 3.3V fixed output. For the resistordivider adjustable output-voltage option, connect FB1 to a resistive divider between OUT1 and GND to adjust
the output voltage between 3.0V and 5.5V. In adjustable mode, FB1 regulates to 1.0V (typ). See the OUT1
Adjustable Output-Voltage Option section.
11
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Maxim Integrated │ 9
MAX16993
Step-Down Controller with
Dual 2.1MHz Step-Down DC-DC Converters
Pin Description (continued)
PIN
NAME
FUNCTION
12
CS1
13
OUT1
14
EN2
Active-High Digital Enable Input for Buck 2. Driving EN2 high enables Buck 2.
15
EN3
Active-High Digital Enable Input for Buck 3. Driving EN3 high enables Buck 3.
16
OUT3
17
RESET3
18
PV3
Buck 3 Voltage Input. Connect a 2.2µF or larger ceramic capacitor from PV3 to PGND3. Connect PV3 to OUT1.
19
LX3
Buck 3 Switching Node. LX3 is high impedance when the device is off.
20
PGND3
Power Ground for Buck 3
21
PGND2
Power Ground for Buck 2
22
LX2
Buck 2 Switching Node. LX2 is high impedance when the device is off.
23
PV2
Buck 2 Voltage Input. Connect a 2.2µF or larger ceramic capacitor from PV2 to PGND2. Connect PV2 to OUT1.
24
RESET2
25
OUT2
Buck Converter 2 Voltage-Sense Input. Connect OUT2 to the output of Buck 2. Connect OUT2 to an external
feedback divider when setting DC-DC2 voltage externally. See the OUT2/OUT3 Adjustable Output-Voltage
Option section.
26
CSEL1
Buck 1 Clock Select. Connect CSEL1 to GND for 2.1MHz operation. Connect CSEL1 to BIAS for an OTPprogrammable divide-down operation. See the Selector Guide for the fSW1 divide ratio.
27
SSEN
Spread-Spectrum Enable. Connect SSEN to GND for standard oscillator operation. Connect SSEN to BIAS to
enable the spread-spectrum oscillator.
28
RESET1
29
GND
30
COMP1
31
ERR
32
SYNC
—
EP
Positive Current-Sense Input for Buck 1. Connect CS1 to the positive terminal of the current-sense resistor.
See the Current-Limit/Short-Circuit Protection and Current-Sense Measurement sections.
Output Sense and Negative Current-Sense Input for Buck 1. The buck uses OUT1 to sense the output
voltage. Connect OUT1 to the negative terminal of the current-sense resistor.
See the Current-Limit/Short-Circuit Protection and Current-Sense Measurement sections.
Buck Converter 3 Voltage-Sense Input. Connect OUT3 to the output of Buck 3. Connect OUT3 to an external
feedback divider when setting DC-DC3 voltage externally. See the OUT2/OUT3 Adjustable Output-Voltage
Option section.
Open-Drain Buck 3 Reset Output. RESET3 remains low for a fixed time after the output of Buck 3 has
reached its regulation level (see the Selector Guide). To obtain a logic signal, pull up RESET3 with an
external resistor connected to a positive voltage lower than 5V.
Open-Drain Buck 2 Reset Output. This output remains low for a fixed time after the output of Buck 2 has
reached its regulation level (see the Selector Guide). To obtain a logic signal, pull up RESET2 with an
external resistor connected to a positive voltage lower than 5V.
Open-Drain Buck 1 Reset Output. RESET1 remains low for a fixed time after the output of Buck 1 has
reached its regulation level (see the Selector Guide). To obtain a logic signal, pull up RESET1 with an
external resistor connected to a positive voltage lower than 5V.
Analog Ground
Compensation for Buck 1. See the Compensation Network section.
Open-Drain Error-Status Output. ERR signals a thermal-warning/shutdown condition. To obtain a logic signal,
pull up ERR with an external resistor connected to a positive voltage lower than 5V.
Synchronization Input. SYNC allows the device to synchronize to other supplies. Connect SYNC to GND or
leave unconnected to enable skip-mode operation under light loads. Connect SYNC to BIAS or an external
clock to enable fixed-frequency forced-PWM-mode operation.
Exposed Pad. Connect the exposed pad to ground. Connecting the exposed pad to ground does not remove
the requirement for proper ground connections to PGND2–PGND3 and GND. The exposed pad is attached
with epoxy to the substrate of the die, making it an excellent path to remove heat from the IC.
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Maxim Integrated │ 10
MAX16993
Step-Down Controller with
Dual 2.1MHz Step-Down DC-DC Converters
Typical Operating Circuit
BIAS
GND
LINEAR
REGULATOR
MAX16993
BIAS
PV1
BST1
PV
VSUP
VBATP
PV3
N
VOUT1
DH1
P
LX1
N
STEP-DOWN
PWM
OUT3
DL1
GND
CS1
STEP-DOWN
CONTROLLER
OUT1
OUT1
FB1
COMP1
PWM
EN
LX3
PGND3
OUT3
0.8V TO 3.95V
1.5A TO 3.0A
PWM
EN
RESET1
VOUT1
P
LX2
RESET2
STEP-DOWN
PWM
OUT2
RESET3
EN1
EN3
VOUT3
N
PV2
EN2
VOUT1
POR
GENERATION
AND
CONTROL
0.8V TO 3.95V
1.5A TO 3.0A
VOUT2
N
PGND2
OUT2
ERR
SSEN
PWM
EN
CSEL1
SYNC
EP
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Maxim Integrated │ 11
MAX16993
Step-Down Controller with
Dual 2.1MHz Step-Down DC-DC Converters
Detailed Description
Enable Inputs (EN_)
The 2.1MHz, high-voltage buck controller operates with
a 3.5V to 36V input voltage range and is protected from
load-dump transients up to 42V. The high-frequency
operation eliminates AM band interference and reduces
the solution footprint. It can provide an output voltage
between 3.0V and 5.5V set at the factory or with external
resistors. Each device has two frequency options that
are pin selectable: 2.1MHz or a lower frequency based
on factory setting. Available factory-set frequencies are
1.05MHz, 525kHz, 420kHz, or 350kHz. Under no-load
conditions, the device consumes only 30µA of quiescent
current with OUT1 enabled.
Reset Outputs (RESET_)
The MAX16993 power-management integrated circuit
(PMIC) is a 2.1MHz, multichannel, DC-DC converter
designed for automotive applications. The device includes
one high-voltage step-down controller (OUT1) designed
to run directly from a car battery and two low-voltage stepdown converters (OUT2/OUT3) cascaded from OUT1.
The dual buck converters can deliver 1.5A or 3.0A of
load current per output. They operate directly from OUT1
and provide 0.8V to 3.95V output voltage range. Factory
trimmed output voltages achieve ±3% output error over
load, line, and temperature without using expensive
±0.1% resistors. In addition, adjustable output-voltage
versions can be set to any desired values between 0.8V
and 3.6V using an external resistive divider. On-board
low RDS(ON) switches help minimize efficiency losses
at heavy loads and reduce critical/parasitic inductance,
making the layout a much simpler task with respect to
discrete solutions. Following a simple layout and footprint
ensures first-pass success in new designs (see the PCB
Layout Guidelines section).
The device features a SYNC input (see the Synchronization
(SYNC) section and the Selector Guide). An optional
spread-spectrum frequency modulation minimizes radiated electromagnetic emissions due to the switching
frequency, and a factory-programmable synchronization
I/O (SYNC) allows better noise immunity. Additional features include a 4ms fixed soft-start for OUT1 and 2.5ms
for OUT2/OUT3, individual RESET_ outputs, overcurrent,
and overtemperature protections. See the Selector Guide
for the available options.
www.maximintegrated.com
All three regulators have their own enable input. When
EN1 exceeds the EN1 high threshold, the internal
linear regulator is switched on. When VSUP exceeds the
VSUP,STARTUP threshold, Buck 1 is enabled and OUT1
starts to ramp up with a 4ms soft-start. Once the Buck 1
soft-start is complete, Buck 2 and Buck 3 can be enabled.
When either Buck 2 or Buck 3 is enabled, the corresponding output ramps up with a 2.5ms soft-start. When an
enable input is pulled low, the converter is switched off
and the corresponding OUT_ and RESET_ are driven
low. If EN1 is low, all regulators are disabled.
The device features individual open-drain RESET_ outputs for each buck output that asserts when the buck
output voltage drops 6% below the regulated voltage.
RESET_ remains asserted for a fixed timeout period after
the buck output rises up to its regulated voltage. The
fixed timeout period is programmable between 0.1ms and
7.4ms (see the Selector Guide). To obtain a logic signal,
pull up RESET_ with an external resistor connected to a
positive voltage lower than 5V.
Linear Regulator (BIAS)
The device features a 5V internal linear regulator (BIAS).
Connect BIAS to PV, which acts as a supply for internal
circuitry. Also connect BIAS to PV1, which acts as a
supply for the low-side gate driver of Buck 1. Bypass BIAS
as close as possible to the device with a 2.2µF or larger
ceramic capacitor. BIAS can provide up to 100mA (max),
but is not designed to supply external loads. After OUT1
completes soft-start, BIAS LDO is turned off and the BIAS
pin is shorted to the OUT1 pin internally to power the
internal circuits (e.g., if OUT1 is set to 3.3V, BIAS transitions from 5V to 3.3V after soft-start).
Internal Oscillator
Buck 1 Clock Select (CSEL1)
The device offers a Buck 1 clock-select input. Connect
CSEL1 to GND for 2.1MHz operation. Connect CSEL1 to
BIAS to divide down the Buck 1 clock frequency by 2, 4, 5,
or 6 (see the Selector Guide). Buck 2 and Buck 3 switch
at 2.1MHz (typ) and are not controlled by CSEL1.
Maxim Integrated │ 12
MAX16993
Step-Down Controller with
Dual 2.1MHz Step-Down DC-DC Converters
fSW + 6%
INTERNAL OSCILLATOR
FREQUENCY
fSW
t
t + 250µs
t + 500µs
t + 750µs
TIME
Figure 1. Effect of Spread Spectrum on Internal Oscillator
Spread-Spectrum Enable (SSEN)
The device features a spread-spectrum enable (SSEN)
input that can quickly enable spread-spectrum operation
to reduce radiated emissions. Connect SSEN to BIAS to
enable the spread-spectrum oscillator. Connect SSEN
to GND for standard oscillator operation. When spread
spectrum is enabled, the internal oscillator frequency
is varied between fSW and (fSW + 6%). The change in
frequency has a sawtooth shape and a frequency of 4kHz
(see Figure 1). This function does not apply to externally
applied oscillation frequency. See the Selector Guide for
available options.
Synchronization (SYNC)
SYNC is factory-programmable I/O. See the Selector
Guide for available options. When SYNC is configured as
an input, a logic-high on SYNC enables fixed-frequency,
forced-PWM mode. Apply an external clock on the SYNC
input to synchronize the internal oscillator to an external
clock. The SYNC input accepts signal frequencies in the
range of 1.7MHz < fSYNC < 2.4MHz. The external clock
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should have a duty cycle of 50%. A logic-low at the SYNC
input enables the device to enter a low-power skip mode
under light-load conditions.
Common Protection Features
Undervoltage Lockout
The device offers an undervoltage-lockout feature.
Undervoltage detection is performed on the PV input. If
VSUP decreases to the point where Buck 1 is in dropout, PV begins to decrease. If PV falls below the UVLO
threshold (2.7V, typ), all three converters switch off and
the RESET_ outputs assert low. Once the device has
been switched off, VSUP must exceed the VSUP,STARTUP
threshold before Buck 1 turns back on.
Output Overvoltage Protection
The device features overvoltage protection on the buck
converter outputs. If the FB1 input exceeds the output
overvoltage threshold, a discharge current is switched on
at OUT1 and RESET1 asserts low.
Maxim Integrated │ 13
MAX16993
Step-Down Controller with
Dual 2.1MHz Step-Down DC-DC Converters
Soft-Start
The device includes a 4ms fixed soft-start time on OUT1
and 2.5ms fixed soft-start time on OUT2/OUT3. Soft-start
time limits startup inrush current by forcing the output
voltage to ramp up towards its regulation point. If OUT1
is prebiased above 1.25V, all three buck converters do
not start up until the prebias has been removed. Once the
prebias has been removed, OUT1 self-discharges to GND
and then goes into soft-start.
Thermal Warning and Overtemperature
Protection
The device features an open-drain, thermal-warning
indicator (ERR). ERR asserts low when the junction
temperature exceeds +150°C (typ). The hysteresis on
the thermal warning is 15°C (typ). For a logic signal,
connect a pullup resistor from ERR to a supply less than
or equal to 5V. When the junction temperature exceeds
+170°C (typ), an internal thermal sensor shuts down the
buck converters, allowing the device to cool. The thermal
sensor turns the device on again after the junction
temperature cools by 15°C (typ).
Buck 1 (OUT1)
Buck controller 1 uses a PWM current-mode control
scheme. An internal transconductance amplifier establishes an integrated error voltage. The heart of the PWM
controller is an open-loop comparator that compares the
integrated voltage-feedback signal against the amplified
current-sense signal plus the slope-compensation ramp,
which are summed into the main PWM comparator to
preserve inner-loop stability and eliminate inductor staircasing. At each rising edge of the internal clock, the highside MOSFET turns on until the PWM comparator trips or
the maximum duty cycle is reached, or the peak current
limit is reached. During this on-time, current ramps up
through the inductor, storing energy in a magnetic field
and sourcing current to the output. The current-mode
feedback system regulates the peak inductor current as a
function of the output-voltage error signal. The circuit acts
as a switch-mode transconductance amplifier and pushes
the output LC filter pole normally found in a voltage-mode
PWM to a higher frequency.
During the second half of the cycle, the high-side
MOSFET turns off and the low-side MOSFET turns on.
The inductor releases the stored energy as the current
ramps down, providing current to the output. The output capacitor stores charge when the inductor current
exceeds the required load current and discharges when
the inductor current is lower, smoothing the voltage
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across the load. Under soft-overload conditions, when the
peak inductor current exceeds the selected current limit
(see the Current-Limit/Short-Circuit Protection section),
the high-side MOSFET is turned off immediately and the
low-side MOSFET is turned on and remains on to let the
inductor current ramp down until the next clock cycle.
PWM/Skip Modes
The device features a synchronization input that puts all
the buck regulators either in skip mode or forced-PWM
mode of operation (see the Synchronization (SYNC)
section). In the PWM mode of operation, the regulator
switches at a constant frequency with variable on-time.
In the skip mode of operation, the regulator’s switching
frequency is load dependent until the output load reaches
a certain threshold. At higher load current, the switching frequency does not change and the operating mode
is similar to the PWM mode. Skip mode helps improve
efficiency in light-load applications by allowing the regulator to turn on the high-side switch only when the output
voltage falls below a set threshold. As such, the regulator
does not switch MOSFETs on and off as often as is the
case in the PWM mode. Consequently, the gate charge
and switching losses are much lower in skip mode.
Minimum On-Time and Duty Cycle
The high-side gate driver for Buck 1 has a minimum ontime of 75ns (max). This helps ensure no skipped pulses
when operating the device in PWM mode at 2.1MHz with
supply voltage up to 18V and output voltage down to
3.3V. Pulse skipping can occur if the on-time falls below
the minimum allowed (see the Electrical Characteristics).
Current-Limit /Short-Circuit Protection
OUT1 offers a current-limit feature that protects Buck 1
against short-circuit and overload conditions on the buck
controller. Buck 1 offers a current-limit sense input (CS1).
Place a sense resistor in the path of the channel 1 current
flow. Connect CS1 to the high side of the sense resistor
and OUT1 to the low side of the sense resistor. Currentlimit protection activates once the voltage across the
sense resistor increases above the 120mV (typ) currentlimit threshold. In the event of a short-circuit or overload
condition, the high-side MOSFET remains on until the
inductor current reaches the current-limit threshold. The
converter then turns on the low-side MOSFET and the
inductor current ramps down. The converter allows the
high-side MOSFET to turn on only when the voltage
across the current-sense resistor ramps down to below
120mV (typ). This cycle repeats until the short or overload
condition is removed.
Maxim Integrated │ 14
MAX16993
Step-Down Controller with
Dual 2.1MHz Step-Down DC-DC Converters
Current-Sense Measurement
For the best current-sense accuracy and overcurrent protection, use a 1% tolerance current-sense resistor between
the inductor and output, as shown in Figure 2. This configuration constantly monitors the inductor current, allowing accurate current-limit protection. Use low-inductance
current-sense resistors for accurate measurement.
High-Side Gate-Drive Supply (BST1)
The high-side MOSFET is turned on by closing an inter
nal switch between BST1 and DH1 and transferring the
bootstrap capacitor’s (at BST1) charge to the gate of the
high-side MOSFET. This charge refreshes when the highside MOSFET turns off and the LX1 voltage drops down
to ground potential, taking the negative terminal of the
capacitor to the same potential. At this time, the bootstrap
diode recharges the positive terminal of the bootstrap
capacitor. The selected n-channel high-side MOSFET
determines the appropriate boost capacitance values
(CBST1 in the Typical Operating Circuit) according to the
following equation:
QG
C BST 1 =
∆VBST 1
where QG is the total gate charge of the high-side
MOSFET and ΔVBST1 is the voltage variation allowed
on the high-side MOSFET driver after turn-on. Choose
ΔVBST1 such that the available gate-drive voltage is not
significantly degraded (e.g., ΔVBST1 = 100mV to 300mV)
when determining CBST1. Use a Schottky diode when
efficiency is most important, as this maximizes the gatedrive voltage. If the quiescent current at high temperature
is important, it may be necessary to use a low-leakage
switching diode.
The boost capacitor should be a low-ESR ceramic
capacitor. A minimum value of 100nF works in most
cases. A minimum value of 470nF is recommended when
using a Schottky diode.
Dropout
When OUT1 input voltage is lower than the desired output
voltage, the converter is in dropout mode. Buck 1 continuously draws current from the bootstrap capacitor when the
high-side switch is on. Therefore, the bootstrap capacitor
needs to be refreshed periodically. When in dropout, the
Buck 1 high-side gate drive shuts off every 8µs, at which
point the low-side gate drive turns on for 120ns.
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MAX16993
DH1
CIN
N
LX1
DL1
L1
VSUP
RCS
COUT
N
GND
CS1
OUT1
OUTPUT SERIES RESISITOR SENSING
Figure 2. Current-Sense Configuration
Buck 2 and Buck 3 (OUT2 and OUT3)
Buck converters 2 and 3 are high-efficiency, lowvoltage converters with integrated FETs. They use a
PWM current-mode control scheme that is operated at
2.1MHz to optimize component size and efficiency, while
eliminating AM band interference. The buck converters
can be configured to deliver 1.5A or 3.0A per channel.
They operate directly from OUT1 and have either fixed
or resistor-programmable (see the Selector Guide) output
voltages that range from 0.8V to 3.95V. Buck 2 and Buck 3
feature low on-resistance internal FETs that contribute to
high efficiency and smaller system cost and board space.
Integration of the p-channel high-side FET enables both
channels to operate with 100% duty cycle when the input
voltage falls to near the output voltage. They feature a
programmable active timeout period (see the Selector
Guide) that adds a fixed delay before the corresponding
RESET_ can go high.
FPWM/Skip Modes
The MAX16993 features an input (SYNC) that puts the
converter either in skip mode or forced PWM (FPWM)
mode of operation. See the Internal Oscillator section.
In FPWM mode, the converter switches at a constant
frequency with variable on-time. In skip mode, the converter’s switching frequency is load-dependent until the
output load reaches a certain threshold. At higher load
current, the switching frequency does not change and the
operating mode is similar to the FPWM mode.
Maxim Integrated │ 15
MAX16993
Step-Down Controller with
Dual 2.1MHz Step-Down DC-DC Converters
Skip mode helps improve efficiency in light-load applications by allowing the converters to turn on the highside switch only when the output voltage falls below a
set threshold. As such, the converter does not switch
MOSFETs on and off as often as is the case in the FPWM
mode. Consequently, the gate charge and switching
losses are much lower in skip mode.
VOUT1
OUT1
FB1
Current-Limit/Short-Circuit Protection
Buck converters 2 and 3 feature current limit that protects
the device against short-circuit and overload conditions at
their outputs. The current limit value is dependent on the
version selected, 1.5A or 3.0A maximum DC current. See
the Selector Guide for the current limit value of the chosen
option and the Electrical Characteristics table for the corresponding current limit. In the event of a short-circuit or
overload condition at an output, the high-side MOSFET
remains on until the inductor current reaches the highside MOSFET’s current-limit threshold. The converter
then turns on the low-side MOSFET and the inductor current ramps down.
The converter allows the low-side MOSFET to turn off
only when the inductor current ramps down to the lowside MOSFET’s current threshold. This cycle repeats until
the short or overload condition is removed.
Applications Information
OUT1 Adjustable Output-Voltage Option
The device’s adjustable output-voltage version (see
the Selector Guide for details) allows the customer to
set OUT1 voltage between 3.0V and 5.5V. Connect a
resistive divider from OUT1 to FB1 to GND to set the
output voltage (Figure 3). Select R2 (FB1 to GND resistor)
less than or equal to 100kΩ. Calculate R1 (VOUT1 to FB1
resistor) with the following equation:
VOUT 1
=
R 1 R 2
− 1
VFB 1
where VFB1 = 1.0V (see the Electrical Characteristics).
The external feedback resistive divider must be frequency
compensated for proper operation. Place a capacitor
across R1 in the resistive divider network. Use the following equation to determine the value of the capacitor:
if R2/R1 > 1, C1 = C(R2/R1)
else, C1 = C, where C = 10pF.
For fixed output options, connect FB1 to BIAS for the
factory-programmed, fixed output voltage. Connect FB1
to GND for a fixed 3.3V output voltage.
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C1
R1
MAX16993
R2
Figure 3. Adjustable OUT1 Voltage Configuration
OUT1 Current-Sense Resistor Selection
Choose the current-sense resistor based on the maximum
inductor current ripple (KINDMAX) and minimum current-limit
threshold across current-sense resistor (VLIM1MIN = 0.1V).
The formula for calculating the current-sense resistor is:
Rcs MAX =
VLIM1MIN
I OUTMAX × ( 1 +
K INDMAX
2
)
where IOUTMAX is the maximum load current for Buck 1
and KINDMAX is the maximum inductor current ripple.
The maximum inductor current ripple is a function of the
inductor chosen, as well as the operating conditions, and
is typically chosen between 0.3 and 0.4:
K INDMAX =
( VSUP − VOUT ) × D
I OUTMAX × f SW 1 [MHz] × L [µH]
where D is the duty cycle. Below is a numerical example to calculate the current-sense resistor in Figure 2.
The maximum inductor current ripple is chosen at the
maximum supply voltage (36V) to be 0.4:
Rcs MAX =
0.1
K INDMAX
I OUTMAX × 1 +
2
0.1
=
= 0.0166 Ω
0.4
5 × 1 +
2
OUT1 Inductor Selection
Three key inductor parameters must be specified for
operation with the device: inductance value (L), inductor
saturation current (ISAT), and DC resistance (RDCR). Use
Maxim Integrated │ 16
MAX16993
Step-Down Controller with
Dual 2.1MHz Step-Down DC-DC Converters
the following formulas to determine the minimum inductor
value:
VOUT 1
( VSUPMAX − VOUT 1 ) ×
VSUPMAX
L MIN 1 [ H
=] 1.3 ×
1
×
f
SW 1 × I OUTMAX × K INDMAX
where fSW1 is the operating frequency and 1.3 is a
coefficient that accounts for inductance initial precision.
or:
L MIN 2 [ H ] =×
1.3
× A V_CS ×
VOUT1
0.8 V
Use the following formula to determine the minimum output capacitor for Buck 1:
I OUT1(MAX)
C OUT ≥
∆VOUT1
2π × f CO ×
× VOUT1
VOUT1
where fCO is the crossover frequency set by RC and CC,
and ΔVOUT1 is the allowable change in voltage during a
load transient condition.
For proper functionality, ceramic capacitors must be
used. Make sure that the self-resonance of the ceramic
capacitors is above 1MHz to avoid instability.
Buck 1 MOSFET Selection
× R CS
Buck 1 drives two external logic-level n-channel MOSFETs
as the circuit switch elements. The key selection parameters to choose these MOSFETs are:
2.1× 10 6
f SW1
● ● On-resistance (RDS(ON))
where AV_CS is current-sense amplifier gain (8V/V, typ).
For proper operation, the chosen inductor value must be
greater than or equal to LMIN1 and LMIN2. The maximum
inductor value recommended is twice the chosen value
from the above formulas.
Table 1 lists some of the inductor values for 5A output
current and several switching frequencies and output
voltages.
Buck 1 Input Capacitor
The device is designed to operate with a single 0.1µF
capacitor on the VSUP input and a single 0.1µF capacitor on
the PV1 input. Place these capacitors as close as possible to
their corresponding inputs to ensure the best EMI and jitter
performance.
OUT1 Output Capacitor
The primary purpose of the OUT1 output capacitor is
to reduce the change in VOUT1 during load transient
conditions. The minimum capacitor depends on the output
voltage, maximum current, and load regulation accuracy.
● ● Maximum drain-to-source voltage (VDS(MAX))
● ● Minimum threshold voltage (VTH(MIN))
● ● Total gate charge (QG)
● ● Reverse transfer capacitance (CRSS)
● ● Power dissipation
Both n-channel MOSFETs must be logic-level types with
guaranteed on-resistance specifications at VGS = 4.5V
when VOUT1 is set to 5V or VGS = 3V when VOUT1 is set
to 3.3V. The conduction losses at minimum input voltage
should not exceed MOSFET package thermal limits or
violate the overall thermal budget. Also, ensure that the
conduction losses plus switching losses at the maximum
input voltage do not exceed package ratings or violate the
overall thermal budget. In particular, check that the dV/dt
caused by DH1 turning on does not pull up the DL1 gate
through its drain-to-gate capacitance. This is the most
frequent cause of cross-conduction problems.
Gate-charge losses are dissipated by the driver and do
not heat the MOSFET. Therefore, the power dissipation
in the device due to drive losses must be checked. Both
MOSFETs must be selected so that their total gate charge
Table 1. Inductor Values vs. (VSUPMAX, VOUT1)
VSUPMAX to VOUT1 (V)
VSUPMAX = 36V, VOUT1 = 5V
VSUPMAX = 36V, VOUT1 = 3.3V
fSW1 (MHz)
2.1
1.05
0.525
0.420
0.350
2.1
1.05
0.525
0.420
0.350
INDUCTOR (µH), ILOAD = 5A
1.5
3.3
5.6
6.8
8.2
1.0
2.2
4.7
4.7
6.8
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Maxim Integrated │ 17
MAX16993
Step-Down Controller with
Dual 2.1MHz Step-Down DC-DC Converters
is low enough; therefore, PV1/ VOUT1 can power both
drivers without overheating the device:
PDRIVE = VOUT1 x (QGTOTH + QGTOTL) x fSW1
where QGTOTL is the low-side MOSFET total gate charge
and QGTOTH is the high-side MOSFET total gate charge.
Select MOSFETs with a QG_ total of less than 10nC. The
selected MOSFET must have an input capacitance (CISS)
less than 900pF (typ) to prevent possible damage to the
device.
The n-channel MOSFETs must deliver the average
current to the load and the peak current during switching.
Dual MOSFETs in a single package can be an economical
solution. To reduce switching noise for smaller MOSFETs,
use a series resistor in the DH1 path and additional gate
capacitance. Contact the factory for guidance using gate
resistors.
Compensation Network
The device uses a current-mode-control scheme that
regulates the output voltage by forcing the required
current through the external inductor, so the controller
uses the voltage drop across the DC resistance of the
inductor or the alternate series current-sense resistor
to measure the inductor current. Current-mode control
eliminates the double pole in the feedback loop caused
by the inductor and output capacitor, resulting in a smaller
phase shift and requiring less elaborate error-amplifier
compensation than voltage-mode control. A single series
resistor (RC) and capacitor (CC) is all that is required
to have a stable, high-bandwidth loop in applications
where ceramic capacitors are used for output filtering
(see Figure 4). For other types of capacitors, due to the
higher capacitance and ESR, the frequency of the zero
created by the capacitance and ESR is lower than the
desired closed-loop crossover frequency. To stabilize a
nonceramic output capacitor loop, add another compensation capacitor (CF) from COMP1 to GND to cancel this
ESR zero.
The basic regulator loop is modeled as a power modulator, output feedback divider, and an error amplifier
(see Figure 4). The power modulator has a DC gain set by
gmc x RLOAD, with a pole and zero pair set by RLOAD, the
output capacitor (COUT), and its ESR. The loop response
is set by the following equation:
In a current-mode step-down converter, the output capacitor and the load resistance introduce a pole at the following frequency:
1
f pMOD =
2 π × C OUT × R LOAD
The unity-gain frequency of the power stage is set by
COUT and gmc:
f UGAINpMOD =
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2 π × C OUT
The output capacitor and its ESR also introduce a zero at:
1
2 π × ESR × C OUT
f zMOD =
When COUT is composed of “n” identical capacitors in
parallel, the resulting COUT = n x COUT(EACH), and ESR
= ESR(EACH) /n. Note that the capacitor zero for a parallel
combination of like-value capacitors is the same as for an
individual capacitor.
The feedback voltage-divider has a gain of GAINFB =
VFB/VOUT, where VFB is 1V (typ).
The transconductance error amplifier has a DC gain
of GAINEA(DC) = gm,EA x ROUT,EA, where gm,EA is
the error amplifier transconductance, which is 660µS
(typ), and ROUT,EA is the output resistance of the error
amplifier, which is 30MΩ (typ).
A dominant pole (fdpEA) is set by the compensation capacitor (CC) and the amplifier output resistance (ROUT,EA). A
zero (fZEA) is set by the compensation resistor (RC) and
the compensation capacitor (CC). There is an optional
pole (fPEA) set by CF and RC to cancel the output
gmc = 1/(AVCS x RDC)
CS_
OUT_
RESR
COUT
CURRENT-MODE
POWER MODULATION
gMEA = 660µS
R1
FB_
R2
VREF
ERROR
AMP
GAINMOD(dc) = gmc x RLOAD
where RLOAD = VOUT /ILOUT(MAX) in Ω and gmc =
1/(AV_CS x RDC) in S. AV_CS is the voltage gain of the
current-sense amplifier and is typically 8V/V. RDC is the
DC resistance of the inductor or the current-sense resistor
in Ω.
g mc
COMP_
30MΩ
RC
CF
CC
Figure 4. Compensation Network
Maxim Integrated │ 18
MAX16993
Step-Down Controller with
Dual 2.1MHz Step-Down DC-DC Converters
capacitor ESR zero if it occurs near the crossover
frequency (fC, where the loop gain equals 1 (0dB)).
Thus:
f dpEA =
1
2 π × C C × ( R OUT,EA + R C )
1
f zEA =
2 π × CC × RC
1
f pEA =
2 π × CF × R C
VOUT
5
GAIN
=
MOD ( f C ) GAINMOD ( dc ) ×
GAINMOD (f ) ×
C
VFB
VOUT
f pMOD
fC
× g m,EA × R C =
1
VOUT
g m,EA × VFB × GAINMOD (f )
C
Set the error-amplifier compensation zero formed by
RC and CC at the fpMOD. Calculate the value of CC as
follows:
1
CC =
2 π × f pMOD × R C
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VOUT = 5V
RLOAD = VOUT/IOUT(MAX) = 5V/6A = 0.833Ω
COUT = 4 x 47µF = 188µF
ESR = 9mΩ/4 = 2.25mΩ
GAINMOD(dc) = 5.68 x 0.833 = 4.73
1
=
≈ 1kHz
f pMOD
2 π × 188 µF × 0.833
f SW
f pMOD
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