IR3876TR1PbF

IR3876MBF SupIRBuck Features             TM 12A HIGHLY INTEGRATED WIDE-INPUT VOLTAGE, SYNCHRONOUS BUCK REGULATOR Description The IR3876 SupIRBuckTM is an easy-to-use, fully integrated and highly efficient DC/DC voltage regulator. The onboard constant on time hysteretic controller and MOSFETs make IR3876 a space-efficient solution that delivers up to 12A of precisely controlled output voltage in 60°C ambient temperature applications without airflow. Programmable switching frequency, soft start, and over current protection allows for a very flexible solution suitable for many different applications and an ideal choice for battery powered applications. Additional features include pre-bias startup, very precise 0.5V reference, over/under voltage shut down, power good output, and enable input with voltage monitoring capability. Input Voltage Range: 3V to 21V Output Voltage Range: 0.5V to 12V Continuous 12A Load Capability Constant On-Time control Excellent Efficiency at very low output current levels Compensation Loop not Required Programmable switching frequency, soft start, and over current protection Power Good Output Precision Voltage Reference (0.5V, +/-1%) Pre-bias Start Up Under/Over Voltage Fault Protection Ultra small, low profile 5mm x 6mm QFN Package Applications     Notebook and desktop computers Game consoles Consumer electronics – STB, LCD, TV, printers General purpose POL DC-DC converters 1 IR3876MBF ABSOLUTE MAXIMUM RATINGS (Voltages referenced to GND unless otherwise specified) • • • • • • • • • • • • VIN. FF ………………………………………………. -0.3V to 25V VCC, PGood, EN ………………….…....……..….… -0.3V to 8.0V Boot ……………………………………..………….…. -0.3V to 33V PHASE ……………………………………………...... -0.3V to 25V(DC), -5V(100ns) Boot to PHASE…..…………………………….…..….. -0.3V to 8V ISET …………………………………………..……. -0.3V to 30V PGND to GND ……………...……………………….... -0.3V to +0.3V All other pins ……………...……………………….….. -0.3V to 3.9V Storage Temperature Range ................................... -65°C To 150°C Junction Temperature Range ................................... -10°C To 150°C ESD Classification …………………………….……… JEDEC Class 1C Moisture sensitivity level ..……………...…………….. JEDEC Level 3@260 °C Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only and functional operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications are not implied. PACKAGE INFORMATION 5mm x 6mm POWER QFN θJA = 35 o C / W θJ -PCB = 2 o C / W ORDERING INFORMATION PKG DESIG M M PACKAGE DESCRIPTION IR3876MTRPbF IR3876TR1PbF PIN COUNT 17 17 PARTS PER REEL 4000 750 2 IR3876MBF Block Diagram 3 IR3876MBF Pin Description NAME NC ISET PGOOD GND FB SS NC 3VCBP NC VCC PGND PHASE VIN BOOT FF EN NUMBER 1 2 3 4,17 5 6 7 8 9 10 11 12 13 14 15 16 5V Reference 3.3V 3.3V ----3.3V ----5V Reference VIN VIN VIN +VCC VIN 5V I/O LEVEL ----DESCRIPTION No connection Connecting resistor to PHASE pin sets over current trip point Power good – pull up to 3.3V Bias return and signal reference Inverting input to PWM comparator, OVP / PGood sense Set soft start slew-rate with a capacitor to GND No connection LDO output. A minimum of 1.0 µF ceramic capacitor is required No connection Gate drive supply Power return Phase node (or switching node) of MOSFET half bridge Input voltage for the system. Bootstrapped gate drive supply – connect a capacitor to PHASE Input voltage feed forward – sets on-time with a resistor to VIN Enable 4 IR3876MBF Recommended Operating Conditions Symbol VIN VCC VOUT IOUT Fs TJ Definition Input Voltage Supply Voltage Output Voltage Output Current Switching Frequency Junction Temperature Min 3 4.5 0.5 0 N/A 0 Max 21* 7.5 12 12 1000 125 Unit V A kHz oC * Note: PHASE pin must not exceed 25V. Electrical Specifications PARAMETER BIAS SUPPLIES VCC Turn-on Threshold VCC Turn-off Threshold VCC Threshold Hysteresis VCC Operating Current VCC Shutdown Current FF Shutdown Current VIN Shutdown Current INTERNAL LDO OUTPUT LDO Output Voltage Range Output Current CONTROL LOOP Reference Accuracy, VREF On-Time Accuracy Min Off Time Soft-Start Current Zero Current Threshold FAULT PROTECTION ISET Pin Output Current Under Voltage Threshold Under Voltage Hysteresis Over Voltage Threshold Over Voltage Hysteresis PGOOD Delay Threshold (VSS) Unless otherwise specified, these specification apply over VIN = 12V, VCC = 5V, 0oC ≤ TJ ≤ 125oC. NOTE TEST CONDITION MIN 3.9 3.6 RFF = 200K, EN = HIGH, Fs = 300kHz EN = LOW EN = LOW EN = LOW CLDO = 1µF 3.1 TYP 4.2 3.9 150 9.2 35 2 1 3.3 MAX 4.5 4.2 UNIT V V mV mA µA µA µA V mA V ns ns µA mV µA V mV 0.66 V mV V 50 3.5 8 VREF RFF = 180K, TJ = 65oC 1 1 EN = HIGH Measure at VPHASE 0.495 280 8 -5 18 0.37 0.58 0.5 300 400 10 -2.4 20 0.4 10 0.62 10 1 0.505 320 12 0 22 0.43 Falling VFB & Monitor PGOOD 1 1 Rising VFB Rising VFB & Monitor PGOOD Falling VFB 5 IR3876MBF Electrical Specifications (continued) PARAMETER GATE DRIVE Dead Time BOOTSTRAP PFET Forward Voltage UPPER MOSFET Static Drain-to-Source OnResistance LOWER MOSFET Static Drain-to-Source OnResistance LOGIC INPUT AND OUTPUT EN High Logic Level EN Low Logic Level EN Input Current PGOOD Pull Down Resistance NOTE 1 Unless otherwise specified, these specification apply over VIN = 12V, VCC = 5V, 0oC ≤ TJ ≤ 125oC. TEST CONDITION Monitor body diode conduction on PHASE pin I(BOOT) = 10mA VCC = 5V, ID = 12A, TJ = 25oC MIN 5 TYP MAX 30 UNIT ns 100 7 200 12 300 16 mV mΩ VCC = 5V, ID = 12A, TJ = 25oC 4 5.3 7 mΩ 2 EN = 3.3V 11 25 0.6 50 V V µA Ω Note1: Guaranteed by design, not tested in production 6 IR3876MBF TYPICAL OPERATING DATA Tested with demoboard shown in Figure 7, VIN = 12.6V, VCC = 5V, Vout = 1.05V, Fs = 300kHz, TA = 25oC, no airflow, unless otherwise specified 95% 85% Efficiency Efficiency 75% 65% 55% 45% 35% 0.01 16VIN 12.6VIN 7VIN 100% 90% 80% 1.05VOUT; L = 1.2uH, 2.9m Ω 70% 60% 50% 40% 0.01 1.5V OUT; L = 1.2uH, 2.9m Ω 3.3V OUT; L = 2.2uH, 4.2m Ω 0.1 1 Output Current (A) 10 100 0.1 1 Output Current (A) 10 100 Figure 1. Efficiency vs. Output Current for VOUT = 1.05V, L = 1.2µH (2.9mΩ) Figure 2. Efficiency vs. Output Current for VIN = 12.6V 350 300 Frequency (kHz) 250 200 150 100 50 0 0 2 4 6 8 10 12 Output Current (A) 1200 1000 RFF (kOhm) 800 600 400 200 0 200 5.0 Vout 4.0 3.0 2.0 1.0 4.5 3.5 2.5 1.5 0.5 400 600 800 Frequency (kHz) 1000 Figure 3. Switching Frequency vs. Output Current Figure 4. Frequency vs. RFF 1.058 Output Voltage (V) 1.057 1.056 VOUT @ 7VIN 1.055 1.054 0 2 4 6 8 10 12 Output Current (A) Output Voltage (V) VOUT @ 16VIN VOUT @ 12.6VIN 1.058 1.057 1.056 1.055 1.054 6 7 8 9 10 11 12 13 14 15 16 17 Input Voltage (V) Figure 5. Output Voltage Regulation vs. Output Current Figure 6. Output Voltage Regulation vs. Input Voltage at IOUT = 12A 7 IR3876MBF TYPICAL APPLICATION CIRCUIT Demoboard Schematic: VOUT = 1.05V, Fs = 300kHz Figure 7. Typical Application Circuit for VOUT = 1.05V, Fs = 300kHz Bill of Materials 8 IR3876MBF TYPICAL OPERATING DATA Tested with demoboard shown in Figure 7, VIN = 12.6V, VCC = 5V, Vout = 1.05V, Fs = 300kHz, TA = 25oC, no airflow, unless otherwise specified EN PGOOD SS VOUT EN PGOOD SS VOUT 5V/div 5V/div 1V/div 500mV/div 5ms/div 5V/div 5V/div 1V/div 500mV/div 200µs/div Figure 8: Startup Figure 9: Shutdown VOUT PHASE VOUT PHASE iL iL 20mV/div 5V/div 2A/div 10µs/div 20mV/div 5V/div 5A/div 2µs/div Figure 10: DCM (IOUT = 0.1A) Figure 11: CCM (IOUT = 12A) PGOOD SS PGOOD FB VOUT iL VOUT IOUT 5V/div 1V/div 1V/div 10A/div 2ms/div 5V/div 1V/div 500mV/div 2A/div 50µs/div Figure 12: Over Current Protection (tested by shorting VOUT to PGND) Figure 13: Over Voltage Protection (tested by shorting FB to VOUT) 9 IR3876MBF TYPICAL OPERATING DATA Tested with demoboard shown in Figure 7, VIN = 12.6V, VCC = 5V, Vout = 1.05V, Fs = 300kHz, TA = 25oC, no airflow, unless otherwise specified VOUT PHASE VOUT PHASE iL iL 50mV/div 10V/div 5A/div 20µs/div 50mV/div 10V/div 5A/div 20µs/div Figure 14: Load Transient 0-8A Figure 15: Load Transient 4-12A Figure 16: Thermal Image at IOUT = 12A (IR3876: 81oC, Inductor: 58oC, PCB: 54oC) 10 IR3876MBF CIRCUIT DESCRIPTION PWM COMPARATOR The PWM comparator initiates a SET signal (PWM pulse) when the FB pin falls below the reference (VREF) or the soft start (SS) voltage. ON-TIME GENERATOR The PWM on-time duration is programmed with an external resistor (RFF) from the input supply (VIN) to the FF pin. The simplified equation for RFF is shown in equation 1. The FF pin is held to an internal reference after EN goes HIGH. A copy of the current in RFF charges a timing capacitor, which sets the on-time duration, as shown in equation 2. GATE DRIVE LOGIC The gate drive logic features adaptive dead time, diode emulation, and a minimum lower gate interval. An adaptive dead time prevents the simultaneous conduction of the upper and lower MOSFETs. The lower gate voltage (LGATE) must be below approximately 1V after PWM goes HIGH before the upper MOSFET can be gated on. Also, the upper gate voltage (UGATE), the difference voltage between UGATE and PHASE, must be below approximately 1V after PWM goes LOW before the lower MOSFET can be gated on. The control MOSFET is gated on after the adaptive delay for PWM = HIGH and the synchronous MOSFET is gated on after the adaptive delay for PWM = LOW. The lower MOSFET is driven ‘off’ when the signal ZCROSS indicates that the inductor current has reversed as detected by the PHASE voltage crossing the zero current threshold. The synchronous MOSFET stays ‘off’ until the next PWM falling edge. When the lower peak of inductor current is above zero, a forced continuous current condition is selected. The control MOSFET is gated on after the adaptive delay for PWM = HIGH, and the synchronous MOSFET is gated on after the adaptive delay for PWM = LOW. The synchronous MOSFET gate is driven on for a minimum duration. This minimum duration allows time to recharge the bootstrap capacitor and allows the current monitor to sample the PHASE voltage. RFF  VOUT (1) 1V  20 pF  FSW TON  RFF 1V  20 pF (2) VIN SOFT START An internal 10µA current source charges the external capacitor on the SS pin to set the output voltage slew rate during the soft start interval. The output voltage reaches regulation when the FB pin is above the under voltage threshold and the UV# = HIGH. Once the voltage on the SS pin is above the PGOOD delay threshold, the combination of the SSDelay and UV# signals release the PGOOD pin. With EN = LOW, the capacitor voltage and SS pin is held to the FB pin voltage. CONTROL LOGIC The control logic monitors input power sources, sequences the converter through the soft-start and protective modes, and indicates output voltage status on the PGOOD pin. VCC and 3VCBP pins are continuously monitored, and the IR3876 will be disabled if the voltage of either pin drops below the falling thresholds. The IR3876 will initiate a soft start when VCC is in the normal range and the EN pin = HIGH. In the event of a sustained overload, a counter keeps track of four soft-start cycles and then disables the IR3876. The overload counter is only reset by EN or UVLO. 11 IR3876MBF CIRCUIT DESCRIPTION OVER CURRENT MONITOR The over current circuitry monitors the output current during each switching cycle. The voltage across the synchronous MOSFET, VPHASE, is monitored for over current and zero crossing. The minimum LGATE interval allows time to sample VPHASE. The over current trip point is programmed with a resistor from the ISET pin to PHASE pin, as shown in equation 3, where Tj is the junction temperature of Q2 at operation conditions, and 0.4 is the temperature coefficient (~4000 ppm/C) of Q2 Rdson. When over current is detected, the output gates are tri-state and SS voltage is pulled to 0V. This initiates a new soft start cycle. If there was a total of three OC events, the IR3876 will disable switching. Toggling VCC or EN will allow the next start up. STABILITY CONSIDERATIONS Constant-on-time control is a fast , ripple based control scheme. Unstable operation can occur if certain conditions are not met. The system instability is usually caused by: • Switching noise coupled to FB input. This causes the PWM comparator to trigger prematurely after the 400ns minimum Q2 ontime. It will result in double or multiple pulses every switching cycle instead of the expected single pulse. Double pulsing can causes higher output voltage ripple, but in most application it will not affect operation. This can usually be prevented by careful layout of the ground plane and the FB sensing trace. • Steady state ripple on FB pin being too small. The PWM comparator in IR3876 requires minimum 7mVp-p ripple voltage to operate stably. Not enough ripple will result in similar double pulsing issue described above. Solving this may require using output capacitors with higher ESR. Another way to solve this is to add a ~10pF ceramic capacitor from Vout to FB to couple more Vout ripple to the FB pin. • ESR loop instability. The stability criteria of constant on-time is: ESR*Cout>Ton/2. If ESR is too small that this criteria is violated then sub-harmonic oscillation will occur. This is similar to the instability problem of peakcurrent-mode control with D>0.5. Increasing ESR is the most effective way to stabilize the system, but the price paid is the larger output voltage ripple. • For applications with all ceramic output capacitors, the ESR is usually too small to meet the stability criteria. In these applications, external slope compensation is necessary to make the loop stable. The ramp injection circuit, composed of R6, C13, and C14, shown in Figure 7 is required. The inductor current ripple sensed by R6 and C13 is AC coupled to the FB pin through C14. The injected ramp slope can be adjusted by changing the RC time constant of R6 and C13, which is similar to the circuit used for DCR current sensing. C14 is typically chosen by C14(R7//R8)=Ton. RSET  RDSON  IOC 20 A  (1  Tj  25 100  0.4) (3) OVER VOLTAGE PROTECTION The IR3876 monitors the voltage at the FB node. If the FB voltage is above the over voltage threshold, the gates are turned off and the PGOOD signal is pulled low. Toggling VCC will allow the next start up. 12 IR3876MBF CIRCUIT DESCRIPTION COMPONENT SELECTION Selection of components for the converter is an iterative process which involves meeting the specifications and trade-offs between performance and cost. The following sections will guide one through the process. Inductor Selection Inductor selection involves meeting the steady state output ripple requirement, minimizing the switching loss of the upper MOSFET, meeting transient response specifications and minimizing the output capacitance. The output voltage includes a DC voltage and a small AC ripple component due to the low pass filter which has incomplete attenuation of the switching harmonics. Neglecting the inductance in series with the output capacitor, the magnitude of the AC voltage ripple is determined by the total inductor ripple current flowing through the total equivalent series resistance (ESR) of the output capacitor bank. Input Capacitor Selection The main function of the input capacitor bank is to provide the input ripple current and fast slew rate current during the load current step up. The input capacitor bank must have adequate ripple current carrying capability to handle the total RMS current. Figure 17 shows a typical input current. Equation 5 shows the RMS input current. The RMS input current contains the DC load current and the inductor ripple current. As shown in equation 4, the inductor ripple current is unrelated to the load current. The maximum RMS input current occurs at the maximum output current. The maximum power dissipation in the input capacitor equals the square of the maximum RMS input current times the input capacitor’s total ESR. ΔI  TON  VIN  VOUT  (4) 2L One can use equation 4 to find the required inductance. ΔI is defined as shown in Figure 17. The main advantage of small inductance is increased inductor current slew rate during a load transient, which leads to a smaller output capacitance requirement as discussed in the Output Capacitor Selection section. The draw back of using smaller inductances is increased switching power loss in upper MOSFET, which reduces the system efficiency and increases the thermal dissipation. Figure 17. Typical Input Current Waveform. 1   f 2 t   dt IIN_RMS  Ts 0 1  ΔI   IOUT  Ton  Fs  1     (5) 3  IOUT  The voltage rating of the input capacitor needs to be greater than the maximum input voltage because of high frequency ringing at the phase node. The typical percentage is 25%. 2 Ts 13 IR3876MBF Output Capacitor Selection Selection of the output capacitor requires meeting voltage overshoot requirements during load removal, and meeting steady state output ripple voltage requirements. The output capacitor is the most expensive converter component and increases the overall system cost. The output capacitor decoupling in the converter typically includes the low frequency capacitor, such as Specialty Polymer Aluminum, and mid frequency ceramic capacitors. The first purpose of output capacitors is to provide current when the load demand exceeds the inductor current, as shown in Figure 18. Equation 6 shows the charge requirement for a certain load. The advantage provided by the IR3876 at a load step is to reduce the delay compared to a fixed frequency control method (in microseconds or (1D)*Ts). If the load increases right after the PWM signal goes low, the longest delay will be equal to the minimum lower gate on as shown in the Electrical Specification table. The IR3876 also reduces the inductor current slew time, the time it takes for the inductor current to reach equality with the output current, by increasing the switching frequency up to 2.5MHz. The result reduces the recovery time. VESR is usually much greater than VESL. The IR3876 requires a total ESR such that the ripple voltage at the FB pin is greater than 7mV. The second purpose of the output capacitor is to minimize the overshoot of the output voltage when the load decreases as shown in Figure 19. By using the law of energy before and after the load removal, equation 7 shows the output capacitance requirement for a load step. C OUT L  ISTEP 2  V OS 2  V OUT 2 (7) Figure 19. Typical Output Voltage Response Waveform. Boot Capacitor Selection The boot capacitor starts the cycle fully charged to a voltage of VB(0). Cg equals 1.2nF in IR3876. Choose a sufficiently small ΔV such that VB(0)-ΔV exceeds the maximum gate threshold voltage to turn on the high side MOSFET.  V (0)  C BOOT  Cg   B  1 (8)  ΔV  Choose a boot capacitor value larger than the calculated CBOOT in equation 8. Equation 8 is based on charge balance at CCM operation. Usually the boot capacitor will be discharged to a much lower voltage when the circuit is operating in DCM mode at light load, due to much longer Q2 off time and the bias current drawn by the IC. Boot capacitance needs to be increased if insufficient turn-on of Q1 is observed at light load, typically larger than 0.1µF is needed. The voltage rating of this part needs to be larger than VB(0) plus the desired derating voltage. Its ESR and ESL needs to be low in order to allow it to deliver the large current and di/dt’s which drive MOSFETs most efficiently. In support of these requirements a ceramic capacitor should be chosen. 14 Figure 18. Charge Requirement during Load Step Q  C  V  0.5  Istep  t COUT  1 (6a)  1 L  ΔIstep 2   (6b) VDROP  2 VIN  VOUT     The output voltage drop, VDROP, initially depends on the characteristic of the output capacitor. VDROP is the sum of the equivalent series inductance (ESL) of the output capacitor times the rate of change of the output current and the ESR times the change of the output current. IR3876MBF DESIGN EXAMPLE Design Criteria: Input Voltage, VIN, = 7V to 16V Output Voltage, VOUT = 1.05V Switching Frequency, Fs = 300KHz Inductor Ripple Current, 2ΔI = 3A Maximum Output Current, IOUT = 12A Over Current Trip, IOC = 18A Overshoot Allowance, VOS = VOUT + 50mV Undershoot Allowance, VDROP = 50mV Find RFF : Choose an inductor with the lowest DCR and AC power loss as possible to increase the overall system efficiency. For instance, choose MPL1055-1R21R manufactured by Delta. The inductance of this part is 1.2µH and has 2.9mΩ DCR. Ripple current needs to be recalculated using the chosen inductor. ΔI  1.05V  16V - 1.05V   1.36 A 2 16V 1.2uH  300K Hz Choose an input capacitor: 1.05V RFF   175 K 1V  20 pF  300KHz Pick a standard value 178 kΩ, 1% resistor. Find RSET : 1.05V 1  1.36 A  IIN_RMS  12 A   1   16V 3  12 A   3.1A 2 RSET  1.4  5.2m  18 A 20A  6.55K The RDSON of the lower MOSFET could be expected to increase by a factor of 1.4 over temperature. Therefore, pick a 6.65kΩ, 1% standard resistor. Find a resistive voltage divider for VOUT = 1.05V: A Panasonic 10µF (ECJ3YB1E106M) accommodates 6 Arms of ripple current at 300KHz. Due to the chemistry of multilayer ceramic capacitors, the capacitance varies over temperature and operating voltage, both AC and DC. One 10µF capacitor is recommended. In a practical solution, one 1µF capacitor is required along with the 10µF. The purpose of the 1µF capacitor is to suppress the switching noise and deliver high frequency current. Choose an output capacitor: To meet the undershoot specification, select a set of output capacitors which has an equivalent ESR of 10mΩ (50mV/5A). To meet the overshoot specification, equation 7 will be used to calculate the minimum output capacitance. As a result, 300µF will be needed for 5A load removal. Combine those two requirements, one can choose a set of output capacitors from manufactures such as Sanyo or Rubycon. A 330µF (2SWZ330M R05) from Rubycon is recommended. This capacitor has 4.5mΩ ESR which leaves margin for the voltage drop of the ESL during load step up. VFB  R2  VOUT  0.5V R 2  R1 R2 = 2.55kΩ, R1 = 2.80kΩ, both 1% standard resistors. Choose the soft start capacitor: Once the soft start time has chosen, such as 1000us to reach to the reference voltage, a 22nF for CSS is used to meet 1000us. Choose an inductor to meet the design specification: L VOUT  VIN  VOUT  VIN  2ΔI  Fs 1.05V  16V - 1.05V   16V  3 A  300KHz  1.1H 15 IR3876MBF LAYOUT RECOMMENDATION Bypass Capacitor: One 1µF high quality ceramic capacitor should be placed as near VCC pin as possible. The other end of capacitor can be connected to a via or connected directly to GND plane. Use a GND plane instead of a thin trace to the GND pin because a thin trace have too much impedance. Boot Circuit: CBOOT should be placed near the BOOT and PHASE pins to reduce the impedance when the upper MOSFET turns on. Power Stage: Figure 20 shows the current paths and their directions for the on and off periods. The on time path has low average DC current and high AC current. Therefore, it is recommended to place the input ceramic capacitor, upper, and lower MOSFET in a tight loop as shown in Figure 20. The purpose of the tight loop from the input ceramic capacitor is to suppress the high frequency (10MHz range) switching noise and reduce Electromagnetic Interference (EMI). If this path has high inductance, the circuit will cause voltage spikes and ringing, and increase the switching loss. The off time path has low AC and high average DC current. Therefore, it should be laid out with a tight loop and wide trace at both ends of the inductor. Lowering the loop resistance reduces the power loss. The typical resistance value of 1-ounce copper thickness is 0.5mΩ per square inch. VIN ON VOUT IR3876 IR3871 OFF COUT CIN Figure 20. Current Path of Power Stage 16 IR3876MBF PCB Metal and Components Placement Lead lands (the 13 IC pins) width should be equal to nominal part lead width. The minimum lead to lead spacing should be ≥ 0.2mm to minimize shorting. Lead land length should be equal to maximum part lead length + 0.3 mm outboard extension. The outboard extension ensures a large toe fillet that can be easily inspected. Pad lands (the 4 big pads) length and width should be equal to maximum part pad length and width. However, the minimum metal to metal spacing should be no less than; 0.17mm for 2 oz. Copper or no less than 0.1mm for 1 oz. Copper or no less than 0.23mm for 3 oz. Copper. 17 IR3876MBF Solder Resist It is recommended that the lead lands are Non Solder Mask Defined (NSMD). The solder resist should be pulled away from the metal lead lands by a minimum of 0.025mm to ensure NSMD pads. The land pad should be Solder Mask Defined (SMD), with a minimum overlap of the solder resist onto the copper of 0.05mm to accommodate solder resist misalignment. Ensure that the solder resist in between the lead lands and the pad land is ≥ 0.15mm due to the high aspect ratio of the solder resist strip separating the lead lands from the pad land. 18 IR3876MBF Stencil Design The Stencil apertures for the lead lands should be approximately 80% of the area of the lead lads. Reducing the amount of solder deposited will minimize the occurrences of lead shorts. If too much solder is deposited on the center pad the part will float and the lead lands will open. The maximum length and width of the land pad stencil aperture should be equal to the solder resist opening minus an annular 0.2mm pull back in order to decrease the risk of shorting the center land to the lead lands when the part is pushed into the solder paste. 19 IR3876MBF DIM A A1 b b1 c D E e e1 e2 MILIMITERS MIN MAX 0.8 1 0 0.05 0.375 0.475 0.25 0.35 0.203 REF. 5.000 BASIC 6.000 BASIC 1.033 BASIC 0.650 BASIC 0.852 BASIC INCHES MIN MAX 0.0315 0.0394 0 0.002 0.1477 0.1871 0.0098 0.1379 0.008 REF. 1.970 BASIC 2.364 BASIC 0.0407 BASIC 0.0256 BASIC 0.0259 BASIC DIM L M N O P Q R S t1, t2, t3 t4 t5 MILIMITERS MIN MAX 0.35 0.45 2.441 2.541 0.703 0.803 2.079 2.179 3.242 3.342 1.265 1.365 2.644 2.744 1.5 1.6 0.401 BASIC 1.153 BASIC 0.727 BASIC INCHES MIN MAX 0.0138 0.0177 0.0962 0.1001 0.0277 0.0314 0.0819 0.0858 0.1276 0.1316 0.0498 0.05374 0.1042 0.1081 0.0591 0.063 0.016 BACIS 0.045 BASIC 0.0286 BASIC IR WORLD HEADQUARTERS: 233 Kansas St., El Segundo, California 90245, USA Tel: (310) 252-7105 TAC Fax: (310) 252-7903 Visit us at www.irf.com for sales contact information Data and specifications subject to change without notice. 09/2010 20