LTC1923
High Efficiency Thermoelectric
Cooler Controller
DESCRIPTIO
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FEATURES
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The LTC®1923 is a pulse width modulator intended for
thermoelectric cooler (TEC) or heater applications requiring either unidirectional or bidirectional drive circuits. All
of the necessary control circuitry and two sets of complementary output drivers are integrated into the LTC1923 to
drive a full bridge, providing an efficient means of bidirectional current flow to the TEC. An accurate temperature
control loop to stabilize the temperature of a laser diode
system is easily achieved with the addition of just a few
external components. Typical temperature setpoint accuracy of 0.1°C is achievable with the LTC1923. Adding an
instrumentation amplifier front end allows setpoint stability of 0.01°C.
High Efficiency, Low Noise Topology
Adjustable Output Slew Rate Reduces EMI
Full-Bridge Controller for Bidirectional
Current Control
Adjustable Pulse-by-Pulse Bidirectional TEC
Current Limit
Open/Shorted Thermistor Indication
Solution Footprint in Less Than 0.6" × 0.8"
(Double-Sided PCB)
Available in 5mm x 5mm QFN and
28-Pin SSOP Packages
TEC Voltage Clamping
TEC Current, Voltage and Heat/Cool Status Outputs
Adjustable/Synchronizable Oscillator Frequency
Reduces Filter Component Size and System Noise
2.5V Reference Voltage Output
2.7V Minimum Operating Voltage
The part features independent adjustable heating and
cooling pulse-by-pulse current limit, current soft-start for
controlled start-up, output slew rate control to reduce
system noise, differential current sense and voltage amplifiers and a host of auxiliary circuits to protect the laser
and provide redundant system monitoring.
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APPLICATIO S
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Laser-Based Fiber Optic Links
Medical Instruments
CPU Temperature Regulators
, LTC and LT are registered trademarks of Linear Technology Corporation.
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TYPICAL APPLICATIO
Laser Temperature Control Loop Achieving Setpoint Stability of 0.01°C
10k
0.1%
10k
PLLLPF
RT
RSLEW
CT
330pF
82k
+
REF
10k
NTC
TMP
CMD
LTC1658
VOUT
REF
LTC2053
A = 10
VDD
–
4.7µF
100k
10M
SDSYNC
VREF
CNTRL
PDRVB
EAOUT
NDRVB
LTC1923
C1, C2: TAIYO YUDEN JMK325BJ226MM-T (X7R)
L1, L2: SUMIDA CDRH6D2B-220NC
*MNA, MPA: SILICONIX Si9801
**MNB, MPB: SILICONIX Si9801
AGND
10µF
NDRVA
ILIM
PDRVA
VSET
CS +
FAULT
CS –
VTHRM
ITEC
TEC +
TEC –
MPB**
MPA*
L1
10µH
1µF
PGND
SS
H/C
VTEC
VDD
VDD
FB
1µF
1µF
VREF
L2
10µH
COOLER
TEC
C2
22µF
C1
MNB** 22µF
MNA*
3
1
RS
4
2
1923 TA01
1923f
1
LTC1923
W W
W
AXI U
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ABSOLUTE
RATI GS
(Note 1)
VDD to GND................................................. – 0.3V to 6V
SDSYNC, RSLEW ......................................... – 0.3V to 6V
FB, CNTRL, VTHRM, ILIM, VSET ..................... – 0.3V to 6V
CS +, CS –, TEC +, TEC – .................................– 0.3V to 6V
FAULT, H/C ................................................. – 0.3V to 6V
Operating Temperature Range (Note 2) .. – 40°C to 85°C
Storage Temperature Range ................. – 65°C to 125°C
Lead Temperature (Soldering, 10 sec).................. 300°C
ORDER PART
NUMBER
1
28 RT
RSLEW
2
27 CT
SDSYNC
3
26 VREF
CNTRL
4
25 PDRVB
22 VDD
EAOUT
5
24 NDRVB
21 VDD
FB
6
23 VDD
LTC1923EUH
32 31 30 29 28 27 26 25
CNTRL 1
24 PDRVB
PIN 1
TOP VIEW
EAOUT 2
23 NDRVB
FB 3
ILIM
9
20 PDRVA
VSET 8
17 CS +
VSET 10
19 CS +
9 10 11 12 13 14 15 16
FAULT 11
18 CS –
VTHRM 12
17 ITEC
CS –
18 PDRVA
ITEC
21 NDRVA
ILIM 7
TEC +
22 PGND
8
TEC –
7
SS
VTEC
AGND
19 NDRVA
H/C
20 PGND
SS 6
VTHRM
NC 5
FAULT
AGND 4
UH PACKAGE
32-LEAD PLASTIC QFN
TJMAX = 125°C, θJA = 34°C/W
EXPOSED PAD IS PGND
(MUST BE SOLDERED TO PCB)
ORDER PART
NUMBER
TOP VIEW
PLLLPF
NC
NC
VREF
CT
RT
PLLLPF
RSLEW
SDSYNC
U
U
W
PACKAGE/ORDER I FOR ATIO
H/C 13
16 TEC +
VTEC 14
15 TEC –
LTC1923EGN
GN PACKAGE
28-LEAD PLASTIC SSOP
TJMAX = 125°C, θJA = 120°C/W
Consult LTC Marketing for parts specified with wider operating temperature ranges.
ELECTRICAL CHARACTERISTICS
The ● denotes specifications which apply over the full operating temperature range, otherwise specifications are TA = 25°C.
VDD = 5V, RSLEW = VDD, SDSYNC = VDD, RT = 10k, CT = 330pF unless otherwise noted.
SYMBOL PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
Input Supply
VDD
Operating Supply Voltage
●
UVLO
Undervoltage Lockout
Low to High Threshold
●
UVHYST
Hysteresis
High to Low
●
IDD
Operating Supply Current
No Output Load, Outputs Not Switching
IDDSHDN
Shutdown IDD
SDSYNC = 0V
SHDNTH
Shutdown Threshold
Measured at PDRVA, PDRVB
2.7
5.5
2.6
50
2
0.3
2.7
130
V
V
mV
4
mA
10
25
µA
0.8
1.4
V
1923f
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LTC1923
ELECTRICAL CHARACTERISTICS
The ● denotes specifications which apply over the full operating temperature range, otherwise specifications are TA = 25°C.
VDD = 5V, RSLEW = VDD, SDSYNC = VDD, RT = 10k, CT = 330pF unless otherwise noted.
SYMBOL PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
2.462
2.450
2.5
2.538
2.550
V
V
Reference
VREF
Reference Output Voltage
No Load
●
VREFGD
VREF Good Threshold
VREF Rising Threshold
2.25
2.45
LDREG
Load Regulation
ILOAD = –1mA to –10mA
10
25
mV
LINEREG Line Regulation
VDD = 2.7V to 5.5V
5
20
mV
VREFISC
VREF = 0V
10
20
190
225
260
kHz
Short-Circuit Current
●
V
mA
Oscillator and Phase-Locked Loop
fOSCI
Initial Oscillator Frequency
RT = 10k, CT = 330pF
fOSC
Frequency Variation
VDD = 2.7V to 5V, CT = 330pF, RT = 10k
165
225
270
kHz
OSCPK
CT Ramp Peak
1.4
1.5
1.6
V
OSCVLY
CT Ramp Valley
0.4
0.5
0.6
V
CTICH
CT Charge Current
CT = 0.3V, RT = 10k
–150
µA
CTIDIS
CT Discharge Current
CT = 1.8V, RT = 10k
150
µA
●
PLLGAIN Gain from PLLLPF to RT
IPLLLPF
– 1.1
Phase Detector Output Current
Sinking
Sourcing
fSYNC < fOSC
fSYNC > fOSC
MSTTH
Master Threshold On PLLLPF Pin
Measured at SDSYNC Pin
SDDLY
Shutdown Delay to Output
– 0.9
– 0.7
µA
µA
12
–12
VDD – 0.7 VDD – 0.4
20
V/V
V
µs
45
Error Amplifier
VOS
Input Offset Voltage
EAOUT = 1V, VCM = 2.5V
–18
18
AOL
Open-Loop Gain
EAOUT = 0.45V to 1.55V, CNTRL = 2.5V
VCM
Common Mode Input Range
EAOUT = 1V
IIB
FB and CNTRL Input Bias Currents
FB = CNTRL = 1.25
VOH
Output High
ILOAD = – 100µA
1.65
VOL
Output Low
ILOAD = 100µA
0.3
0.45
V
ISOURCE
Sourcing Current
EAOUT = 1V, FB = 2.4V, CNTRL = 2.5V
–1.5
– 0.5
mA
ISINK
Sinking Current
EAOUT = 1V, FB = 5V, CNTRL = 2.5V
GBW
Gain-Bandwidth Product
f = 100kHz (Note 3)
80
mV
dB
0.2
VDD + 0.2
V
– 100
100
nA
1
V
2
mA
2
MHz
Current Sense Amplifier
ACS
Amplifier Gain
CSOFF
Amplifier Offset
Measured at ITEC
10
ITECH
Output Sourcing Load Regulation
CS +
ITECL
Output Sinking Load Regulation
CS + - CS – = 100mV, ILOAD = 0 to 50µA
f3dB
–3dB Frequency
(Note 3)
ILIMTH
Current Limit Threshold
Measured at CS+, CS–
ILIMDLY
Current Limit Delay to Output
SSICHG
Soft-Start Charge Current
SS = 0.75V
SSILIM
Soft-Start Current Limit Threshold
ILIM
ILIM Current Limit Threshold
- CS – = 100mV, I
– 15
LOAD = 0 to – 50µA
–2
V/V
10
mV
0.1
0.2
V
0.1
0.2
V
500
kHz
125
145
300
450
ns
– 2.5
– 1.5
– 0.5
µA
SS = 0.5V, Measured at CS+, CS–
50
70
90
mV
ILIM = 0.5V, Measured at CS+, CS–
50
70
90
mV
●
165
mV
1923f
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LTC1923
ELECTRICAL CHARACTERISTICS
The ● denotes specifications which apply over the full operating temperature range, otherwise specifications are TA = 25°C.
VDD = 5V, RSLEW = VDD, SDSYNC = VDD, RT = 10k, CT = 330pF unless otherwise noted.
SYMBOL PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
0.98
1
1.02
V/V
TEC Voltage Amplifier
ATEC
Amplifier Gain
TECOFF
Amplifier Offset
Measured at VTEC, VCM = 2.5V
–7
mV
TECCMR
Common Mode Rejection
0.1V < VCM < 4.9V
60
dB
VTECH
Output High Voltage
ILOAD = –50µA
VTECL
Output Low Voltage
ILOAD = 50µA
f3dB
–3dB Frequency
(Note 3)
4.7
4.9
0.1
V
0.3
1
V
MHz
Output Drivers
OUTH
Output High Voltage
IOUT = – 100mA
OUTL
Output Low Voltage
IOUT = 100mA
4
0.7
tRISE
Output Rise Time
CLOAD = 1nF
20
tFALL
Output Fall Time
CLOAD = 1nF
20
ns
trSLEW
Output Rise Time
CLOAD = 1nF, RSLEW = 10k
20
ns
tfSLEW
Output Fall Time
CLOAD = 1nF, RSLEW = 10k
20
ns
trSLEW
Output Rise Time
CLOAD = 1nF, RSLEW = 100k
90
ns
tfSLEW
Output Slew Fall Time
CLOAD = 1nF, RSLEW = 100k
SLEWVT
RSLEW Disable Threshold
DLY
Output Dead Time
RT = 10k
4.5
V
1.2
V
ns
90
ns
2.75
V
90
ns
Fault
OPENTH
Open Thermistor Threshold
VSET = 5V, Measured with Respect to VSET
– 410
mV
SHRTTH
Shorted Thermistor Threshold
VSET = 5V, Measured with Respect to GND
0.975
V
FLTV
Fault Output Low Voltage
1mA Into FAULT, During Fault
150
300
mV
Direction Comparator
DIRH
Low-to-High Threshold
TEC – = 2.5V, Measured with Respect to TEC –
Sensed When H/C Toggles Low
50
mV
DIRL
High-to-Low Threshold
TEC – = 2.5V, Measured with Respect to TEC –
Sensed When H/C Toggles High
– 50
mV
HCV
H/C Output Low Voltage
1mA Into Pin
150
Note 1: Absolute Maximum Ratings are those values beyond which the life
of a device may be impaired.
Note 2: The LTC1923E is guaranteed to meet specifications from 0°C to
70°C. Specifications over the –40°C to 85°C operating temperature range
are assured by design, characterization and correlation with statistical
process controls.
300
mV
Note 3: Guaranteed by design, not tested in production.
1923f
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LTC1923
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TYPICAL PERFOR A CE CHARACTERISTICS
Oscillator Frequency
vs Temperature
Oscillator Frequency vs RT
1800
225
205
185
2.510
VDD = 2.7V, 5V
TA = 25°C
1600
245
VREF vs Temperature
2.505
1400
1200
CT = 68pF
1000
VREF (V)
CT = 330pF
RT = 10k
OSCILLATOR FREQUENCY (kHz)
OSCILLATOR FREQUENCY (kHz)
265
CT = 150pF
800
2.500
2.495
600
400
2.490
200
CT = 330pF
165
–50
–20
10
40
70
TEMPERATURE (°C)
100
0
130
10
5
2.485
–50
20
15
1923 G01
150
250
TA = 25°C
TA = 25°C
TA = 125°C
2.495
TA = 25°C
2.485
2.480
2.470
5
100
VDD = 5V
75
50
TA = –50°C
0
VDD = 2.7V
RISE TIME (ns)
DEAD TIME (ns)
VREF (V)
200
125
2.505
25
15
10
5
7.5
10
1923 G05
Shorted Thermistor Threshold
vs Temperature
1.00
0
–0.5
100
130
1923 G07
SHORTED THERMISTOR THRESHOLD (V)
OPEN THERMISTOR THRESHOLD (V)
ERROR AMPLIFIER VIO (mV)
VSET = 5V
0.5
300
1923 G06
Open Thermistor Threshold
vs Temperature
1.0
200
RSLEW (kΩ)
4.60
10
40
70
TEMPERATURE (°C)
100
0
RT (kΩ)
Error Amplifier Offset Voltage
vs Temperature
–20
100
0
15
12.5
1923 G04
–1.0
–50
150
50
IREF (mA)
1.5
130
Output Rise/Fall Time vs RSLEW
2.510
2.475
100
1923 G03
Output Dead Time vs RT
2.515
2.490
10
40
70
TEMPERATURE (°C)
1923 G02
VREF vs IREF for Different
Temperatures
2.500
–20
RT (kΩ)
4.59
4.58
4.57
4.56
4.55
–50
–20
10
40
70
TEMPERATURE (°C)
100
130
1923 G08
VSET = 5V
0.99
0.98
0.97
0.96
0.95
–50
–20
10
40
70
TEMPERATURE (°C)
100
130
1923 G09
1923f
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LTC1923
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TYPICAL PERFOR A CE CHARACTERISTICS
System Power Loss
vs TEC Current
0.7
160
0.6
155
150
145
140
135
TEC Clamp Voltage
vs Temperature
2.550
TA = 25°C
TEC CLAMP VOLTAGE (V)
165
POWER LOSS (W)
CURRENT LIMIT THRESHOLD (mV)
Current Limit Threshold
vs Temperature
0.5
0.4
VDD = 5V
0.3
VDD = 3.3V
0.2
2.530
2.510
2.490
2.470
0.1
130
125
–50
2.450
–50
0
–20
40
70
10
TEMPERATURE (°C)
100
130
0
0.5
1
–20
TEC CURRENT (A)
1923 G10
40
70
10
TEMPERATURE (°C)
1923 G11
100
130
1923 G12
Representative Waveforms for NDRVA,
NDRVB, TEC Current and CS + – CS –
CH1: TEC CURRENT (500mA/DIV)
CH2: VOLTAGE ACROSS
0.1Ω SENSE RESISTOR
(CS + – CS –) 100mV/DIV
CH3: NDRVA (5V/DIV)
CH4: NDRVB (5V/DIV)
VDD = 5V
RTEC = 2.5Ω
RS = 0.1Ω
1923 G15.tif
Representative Waveforms for
TEC Current, CS + – CS – and ITEC
CH2: VOLTAGE ACROSS 0.1Ω SENSE
RESISTOR RS (CS + – CS –) 100mV/DIV
CH2
CH3: VOLTAGE ON ITEC PIN EQUAL TO
TEN TIMES THE ABSOLUTE VALUE OF
CH2 (200mV/DIV)
CH1: TEC CURRENT (500mA/DIV)
CH1, CH3
VDD = 5V
RTEC = 2.5Ω
RS = 0.1Ω
1923 G16.tif
1923f
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LTC1923
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TYPICAL PERFOR A CE CHARACTERISTICS
Long-Term Cooling Mode Stability Measured in Environment that Steps 20 Degrees Above Ambient
Every Hour. Data Shows Resulting 0.008°C Peak-to-Peak Variation, Indicating Thermal Gain of
2500. 0.0025°C Baseline Tilt Over Plot Length Derives From Varying Ambient Temperature
1923 G13.tif
Identical Test Conditions as Above, Except in Heating Mode. TEC’s Higher Heating Mode Efficiency
Results in Higher Thermal Gain. 0.002°C Peak-to-Peak Variation Is 4x Stability Improvement.
Baseline Tilt, Just Detectable, Shows Similar 4x Improvement vs Above
1923 G14.tif
1923f
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LTC1923
U
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PI FU CTIO S
(GN Package/UH Package)
PLLLPF (Pin 1/Pin 30): This pin serves as the lowpass
filter for the phase-locked loop when the part is being
synchronized. The average voltage on this pin equally
alters both the oscillator charge and discharge currents,
thereby changing the frequency of operation. Bringing the
voltage on this pin above VDD – 0.4V signifies that the part
will be used as the synchronization master. This allows
multiple devices on the same board to be operated at the
same frequency. The SDSYNC pin will be pulled low during
each CT charging cycle to facilitate synchronization.
RSLEW (Pin 2/Pin 31): Placing a resistor from this pin to
AGND sets the voltage slew rate of the output driver pins.
The minimum resistor value is 10k and the maximum
value is 300k. Slew rate limiting can be disabled by tying
this pin to VDD, allowing the outputs to transition at their
maximum rate.
SDSYNC (Pin 3/Pin 32): This pin can be used to disable the
IC, synchronize the internal oscillator or be the master to
synchronize other devices. Grounding this pin will disable
all internal circuitry and cause NDRVA and NDRVB to be
forced low and PDRVA and PDRVB to be forced to VDD.
EAOUT will be forced low. FAULT will also be asserted low
indicating a fault condition. The pin can be pulled low for
up to 20µs without triggering the shutdown circuitry. The
part can either be slaved to an external clock or can be used
as the master (see Applications Information for a more
detailed explanantion).
CNTRL (Pin 4/Pin 1): Noninverting Input to the Error
Amplifier.
SS (Pin 8/Pin 6): The TEC current can be soft-started by
adding a capacitor from this pin to ground. This capacitor
will be charged by a 1.5µA current source. This pin connects
to one of the inverting inputs of the current limit comparator and allows the TEC current to be linearly ramped up from
zero. The voltage on this pin must be greater than 1.5V to
allow the open/shorted thermistor window comparitor to
signal a fault.
ILIM (Pin 9/Pin 7): A voltage divider from VREF to this pin
sets the current limit threshold for the TEC. If the voltage
on this pin is set higher than 1V, then ILIMIT = 150mV/RS
as that is the internal current limit comparator level. If the
voltage on this pin is set less than 1V, the current limit
value where the comparator trips is:
ILIMIT = [0.15 • RILIM1 • VREF]/[(RILIM1 + RILIM2) • RS]
VSET (Pin 10/Pin 8): This is the input for the setpoint
reference of the temperature sense element divider network or bridge. This pin must be connected to the bias
source for the thermistor divider network.
FAULT (Pin 11/Pin 9): Open-drain output that indicates by
pulling low when the voltage on VTHRM is outside the
specified window, the part is in shutdown, undervoltage
lockout (UVLO), or the reference is not good. When the
voltage on VTHRM is outside the specified window, it
signifies that the thermistor impedance is out of its acceptable range. This signal can be used to flag a microcontroller
to shut the system down or used to disconnect power from
the bridge. See Applications Information for using this
signal for redundant protection.
EAOUT (Pin 5/Pin 2): Output of the Error Amplifier. The
loop compensation network is connected between this pin
and FB. The voltage on this pin is the input to the PWM
comparator and commands anywhere between 0% and
100% duty cycle to control the temperature of the temperature sense element.
VTHRM (Pin 12/Pin 10): Voltage Across the Thermistor. If
the voltage on this pin is outside the range between 410mV
below VSET and 0.2 • VSET, the FAULT pin will be asserted
(and latched) low indicating that the thermistor temperature has moved outside the acceptable range.
FB (Pin 6/Pin 3): The Inverting Input to the Error Amplifier.
This input is connected to EAOUT through a compensating
feedback network.
H/C (Pin 13/Pin 11): This open-drain output provides the
direction information of the TEC current flow. If TEC + is
greater than TEC –, which typically corresponds to the
system cooling, this output will be a logic low. If the
opposite is the case, this pin will pull to a logic high.
AGND (Pin 7/Pin 4): Signal Ground. All voltages are
measured with respect to AGND. Bypass VDD and VREF
with low ESR capacitors to the ground plane near this pin.
1923f
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LTC1923
U
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PI FU CTIO S
(GN Package/UH Package)
VTEC (Pin 14/Pin 12): Output of the differential TEC voltage
amplifier equal to the magnitude of the voltage across
the␣ TEC.
NDRVA, NDRVB (Pins 21, 24/Pins 19, 23): These pushpull outputs are configured to drive the opposite low side
switches in a full-bridge arrangement.
TEC– (Pin 15/Pin 13): Inverting Input to the Differential TEC
Voltage Amplifier. This amplifier has a fixed gain of 1 with its
output being the voltage across the TEC with respect to
AGND. This input, along with TEC+, signifies whether the
TEC is heating or cooling the laser as indicated by the
H/C␣ pin.
PGND (Pin 22/Pin 20): This is the high current ground for
the IC. The external current sense resistor should be
referenced to this point.
TEC + (Pin 16/Pin 14): Noninverting Input to the Differential TEC Voltage Amplifier.
ITEC (Pin 17/Pin 15): Output of the Differential Current
Sense Amplifier. The voltage on this pin is equal to 10 •
(ITEC + IRIPPLE) • RS, where ITEC is the thermoelectric
cooler current, IRIPPLE is the inductor ripple current and RS
is the sense resistor used to sense this current. This
voltage represents only the magnitude of the current and
provides no direction information. Current limit occurs
when the voltage on this pin exceeds the lesser of 1.5
times the voltage on SS, 1.5 times the voltage on ILIM or
1.5V. When this condition is present, the pair of outputs,
which are presently conducting, are immediately turned
off. The current limit condition is cleared when the CT pin
reaches the next corresponding peak or valley (see Current Limit section).
CS – (Pin 18/Pin 16): Inverting Input to the Differential
Current Sense Amplifier.
CS + (Pin 19/Pin 17): Noninverting Input of the Differential
Current Sense Amplifier. The amplifier has a fixed gain
of␣ 10.
PDRVA, PDRVB (Pins 20, 25/Pins 18, 24): These pushpull outputs are configured to drive the opposite high side
PMOS switches in a full-bridge arrangement.
VDD (Pin 23/Pins 21, 22): Positive Supply Rail for the IC.
Bypass this pin to PGND and AGND with > 10µF low ESL,
ESR ceramic capacitors. The turn on voltage level for VDD
is 2.6V with 130mV of hysteresis.
VREF (Pin 26/Pin 27): This is the output of the Reference.
This pin should be bypassed to GND with a 1µF ceramic
capacitor. The reference is able to supply a minimum of
10mA of current and is internally short-circuit current
limited.
CT (Pin 27/Pin 28): The triangular wave oscillator timing
capacitor pin is used in conjunction with RT to set the
oscillator frequency. The equation for calculating frequency is:
fOSC =
0.75
Hz
R T • CT
RT (Pin 28/Pin 29): A single resistor from RT to AGND sets
the charging and discharging currents for the triangle
oscillator. This pin also sets the dead time between turning
one set of outputs off and turning the other set on to ensure
the outputs do not cross conduct. The voltage on this pin
is regulated to 0.5V. For best performance, the current
sourced from the RT pin should be limited to a maximum
150µA. Selecting RT to be 10k is recommended and
provides 90ns of dead time.
1923f
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VTHRM
VSET
EAOUT
CNTRL
FB
CT
RT
IRT
gm
0.5V
SHUTDOWN
COMPARATOR
+
ERROR
AMPLIFIER
–
–
+
–
1 = UVLO
UVLO
VDD
0.2VSET
AGND
VREF
2.5V LDO
REF
VBG
REFGOOD
UVLO
SHDN
TSD
TEC
CLAMP
TSD
VTEC
2.5V
ENABLE
ENABLE
Q
–
R Q
+
ENABLE
VREF GOOD
COMPARATOR
SS
FAULT
1.5V
RAMP = 0.5V – 1.5V
S Q
0.3V
MASTER
COMPARATOR
–
–
1 = REFGOOD
R
S
gm
+
gm
VDD – 0.4V
OSCILLATOR
1 = SHDN
OPEN/SHORTED THERMISTOR
350mV
R
20µs
DELAY
+
SDSYNC
+
–
–
+
–
0.7V
+
+
DIGITAL
PHASE
DETECTOR
PLLLPF
+
–
10
QB
Q
R
S
R
S
ITEC
VTEC
2R
–
–
–
+
R
PDRVA
NDRVA
PDRVB
NDRVA
SLEW LIMITING
1/2 VDD
TEC AMP
CS AMP
1V
+
–
H/C
X1
X10
–
+
–
+
90ns
DELAY
90ns
DELAY
DELAY ∝ IRT
DIRECTIONAL
COMPARATOR
INPUT
SELECT
INPUT
SELECT
1 = NO SLEW LIMITING
SWITCHES OPEN
–
+
TEC –
TEC +
CS –
CS +
ILIM
SS
PGND
1.5µA
1923 BD
VDD
NDRVB
NDRVA
PDRVB
PDRVA
VDD
W
FU CTIO AL DIAGRA
U
U
RSLEW
ISLEW
ENABLE
OSC VALLEY
Q
ENABLE
OSC PEAK
QB
VDD + VBE
CURRENT LIMIT
Q
Q
Q
Q
4
1
VDD
LTC1923
1923f
LTC1923
U
OPERATIO
MAIN CONTROL LOOP
The LTC1923 uses a constant frequency, voltage mode
architecture to control temperature. The relative duty
cycles of two pairs of N-/P-channel external MOSFETs, set
up in a full-bridge (also referred to as an H-bridge)
configuration are adjusted to control the system temperature. The full-bridge architecture facilitates bidirectional
current flow through a thermoelectric cooler (TEC) or
other heating element. The direction of the current flow
determines whether the system is being heated or cooled.
Typically a thermistor, platinum RTD or other appropriate
element is used to sense the system temperature. The
control loop is closed around this sense element and TEC.
The voltage on the output of the error amplifier, EAOUT,
relative to the triangle wave on CT, controls whether the
TEC will be heating or cooling. A schematic of the external
full bridge is shown in Figure 1. The “A” side of the bridge
is comprised of the top left PMOS, MPA, and lower right
NMOS, MNA. The gates of these devices are attached to
the PDRVA and NDRVA outputs of the LTC1923, respectively. The “B” side of the bridge is comprised of PMOS,
MPB and NMOS, MNB. The gates of these MOSFETs are
controlled by the PDRVB and NDRVB outputs of the
LTC1923.
The “A” side of the bridge is turned on (NDRVA is high and
PDRVA is low) when the output of the error amplifier is
less than the voltage on the CT pin as shown in Figure 2.
For this condition, the state of each output driver is as
follows: PDRVA is low, NDRVA is high, PDRVB is high and
NDRVB is low. When the voltage on EAOUT is greater than
the voltage on the CT pin, the “B” side of the bridge is
turned on. The average voltage across the TEC, VTECOOLER,
is approximately:
VTECOOLER = VTEC+ – VTEC– = VDD •␣ (DA – DB)
where
VDD = the full-bridge supply voltage
VTECOOLER = VTEC+ – VTEC–
DA = the duty cycle of the “A” side of the bridge or the
amount of time the “A” side is on divided by the
oscillator period
DB = the duty cycle of the “B” side of the bridge
Duty cycle terms DA and DB are related by the following
equation:
DA = 1 – DB
In steady-state, the polarity of VTECOOLER indicates whether
the system is being heated or cooled. Typically, when
current flows into the TEC + side of the cooler, the system
is being cooled and heated when current flows out of this
terminal. Note: Do not confuse the TEC+ side of the TEC
with the TEC+ input of the LTC1923, although these two
points should be connected together.
VDD
EAOUT
PDRVB
PDRVA
MPA
+
VTECOOLER
TEC
MPB
CT
–
NDRVA
NDRVB
MNB
A
SIDE
ON
B
SIDE
ON
MNA
NDRVA
PDRVA
CS +
3
1923 F01
1
RS
CS –
4
NDRVB
2
PDRVB
TEC +
TEC –
1923 F02
Figure 1. Full-Bridge Schematic
Figure 2. Error Amplifier Output, CT and Output Driver Waveforms
1923f
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OPERATIO
PROTECTION FEATURES
Many protection features have been integrated into the
LTC1923 to ensure that the TEC is not overstressed or the
system does not thermally run away. These features
include pulse-by-pulse current limiting, TEC voltage clamping and open/shorted thermistor detection.
Current Limit
The peak current in the full bridge during each switching
cycle can be limited by placing a sense resistor, RS, from
the common NMOS source connections of MNA and MNB
to ground. The CS + and CS – connections should be made
as shown in Figure 1. Current limit is comprised of a fixed
gain of ten differential amplifier, an attenuator (resistor
divider) and a current limit comparator. A detailed diagram
of the circuitry is shown in Figure 3. The differential
amplifier output, ITEC, is provided to allow the user the
ability to monitor the instantaneous current flowing in the
bridge. If an average current is desired, an external RC
filter can be used to filter the ITEC output. Approximately
50ns of leading edge blanking is also internally integrated
to prevent nuisance tripping of the current sense circuitry.
It relieves the filtering requirements for the CS input pins.
During a switching cycle, current limit occurs when the
voltage on ITEC exceeds the lowest of the following three
conditions: 1) 1.5 times the voltage on the SS pin, 2) 1.5
times the voltage on the ILIM pin or 3) 1.5V. When a
current limit condition is sensed, all four external FETs are
immediately shut off. These devices are turned back on
only after CT reaches the same state (either charging or
discharging) as when the current limit condition occurred. For instance, if CT is charging when current limit
occurs, the outputs are forced off for the remainder of this
charging time, the entire CT discharge time, and are only
re-enabled when CT reaches its valley voltage and begins
charging again. An analogous sequence of events occurs
if current limit is tripped while CT is being discharged.
The full-bridge current can be soft-started (gradually
increased) by placing a capacitor from the SS pin to
ground. A 1.5µA current is sourced from the chip and will
charge the capacitor. This limits the inrush current at startup and allows the current delivered to the TEC to be linearly
increased from zero.
The LTC1923 features a dedicated pin, ILIM, to adjust
current limit. If the voltage placed on ILIM is greater than
1V, the default current limit, ILIMIT, is:
ILIMIT = 150mV/RS
where RS = the current sense resistor.
Utilizing the ILIM pin allows the current limit threshold to
be easily set and adjusted (the current limit threshold can
also be adjusted by changing RS). More importantly, it
facilitates independent setting of the heating and cooling
current limits with the addition of one transistor. Figure 4
shows how to implement this using three resistors and an
external NMOS, M1. In many applications, a higher cooling capability is desired. When TEC + is greater than
TEC –, the H/C output is in a low state signifying that the
system is being cooled (this is typical for most lasers).
TEC + NDRVA
TEC –
NDRVB
CS+
+
CS –
INPUT SELECT
CURRENT SENSE
AMPLIFIER
NDRVA NDRVB
A = 10
–
R
LEB
ITEC
2R
ILIM
1V
+
–
–
–
PULSE-BY-PULSE
CURRENT LIMIT
S
OSCILLATOR
PEAK/VALLEY
SS
Q
SHUT
OUTPUTS
OFF
R
1923 F03
1.5µA
Figure 3. Current Sense Circuitry
1923f
12
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OPERATIO
VREF
VDD
RILIM2
RPULLUP
ILIM
RILIM1
RLIM3
LTC1923
H/C
M1
2N7002
+
–
TEC+
TEC –
parameters including the size of the TEC and how well
heatsinked the device is. The TEC itself dissipates power
to produce the temperature differential, generating heat,
which must also be removed. At a certain level of power
dissipation in the TEC, both sides will begin to heat. This
is because the TEC will not be able to pump the selfgenerated heat to the outside world, which can lead to
thermal runaway. If the device thermally runs away, damage to the TEC and possibly the components whose
temperature is being regulated will occur.
1923 F04
Figure 4. Independently Heating/Cooling Current Limit
Transistor M1 is off and the current limit threshold is given
by:
ILIMIT =
0.15 • RILIM1 • VREF
(RILIM1 + RILIM2 ) • RS
When TEC – is greater than TEC +, the open-drain output,
H/C, pulls high through RPULLUP, causing M1 to turn on.
The current limit value is given by:
ILIMIT =
(
)
0.15 • RILIM1 RILIM3 • VREF
(RILIM2 + RILIM1 RILIM3 ) • RS
The LTC1923 contains two dedicated comparators that
directly monitor the voltage on the thermistor. If this
voltage is outside the valid window, a latch is set and the
FAULT pin is asserted low. The output drivers are not shut
off and the control circuitry is not disabled, meaning the
part will continue to try to regulate temperature. It is up to
the user to use the FAULT signal to disable the appropriate
circuitry. There are a couple of ways to do this. The first
way is to have the FAULT signal a system microprocessor
to shut the system down through the SDSYNC pin. Figure␣ 5 shows another means of protecting the system.
External NMOS M1 and PMOS M2 have been added along
with two pull-up resistors (RP1 and RP2). M1 and RP2
invert the FAULT signal while M2 acts as a switch in series
with bridge. When no fault is present, the gate of M1 is
VDD
VDD
reducing the current limit threshold for heating. If the
heating current limit needs to be greater than the cooling
limit, an extra inversion can be added.
RP1
VDD
RP2
M2
FAULT
M1
Open/Shorted Thermistor Detection
The temperature sense element (NTC thermistor, platinum RTD or other appropriate component) must be properly connected in order for the system to regulate
temperature. If the sense element is incorrectly connected, the system will be unable to control the temperature and the potential exists for the system to thermally run
away.
A TEC by nature produces a temperature differential between opposite sides of the device depending upon how
much current is flowing through it. There is a maximum
limit to the amount of temperature differential that can be
produced, which depends upon a number of physical
PDRVB
PDRVA
TEC
NDRVB
NDRVA
CS +
3
1923 F05
1
RS
CS –
4
2
Figure 5. Redundant Fault Protection
1923f
13
LTC1923
U
OPERATIO
pulled to VDD forcing the gate of M2 low, which allows the
bridge to operate as described earlier. When a fault occurs
and FAULT is asserted low, M1 is shut off, forcing the gate
of M2 high, shutting that device off. The power path is thus
opened, ensuring no current is delivered to the TEC. M2
wants to have low RDS(ON) (less than the value of RS to
minimize the power losses associated with it). RP1 and
RP2 can be selected on the order of 100k.
The lower comparator threshold level is 20% (twenty
percent) of VSET and the upper comparator threshold level
is 350mV below VSET, where VSET is the voltage applied on
the VSET pin. VSET is typically tied to the bias source for the
thermistor divider so that any variations will track out.
The VSET pin has a high input impedance so that a divideddown voltage can be supplied to this pin to modify the
acceptable thermistor impedance range. This is shown in
Figure 6. The voltage applied to the VSET pin must be a
minimum of 2V. The lower thermistor impedance threshold is:
RTH(LOWER) =
0.2 • R1 • R3
R2 + 0.8 • R3
The upper impedance threshold is:
RTH(UPPER) =
R1(R3 – α(R2 + R3))
R2 + α(R2 + R3)
where α = 0.35/VSET.
Changing R1 also changes the valid thermistor impedance
range.
Example: VREF = VSET = 2.5V
R1 = 10k, R2 = 0Ω, R3 = open
RTH = 10k NTC thermistor with a temperature coefficient of – 4.4%/C at 25°C.
The acceptable thermistor impedance range before causing a fault is 2.5kΩ to 61kΩ. This corresponds to a valid
temperature range of between about –10°C and 60°C.
To ensure the part does not power up with a latched fault
at start-up, a fault will not be latched until soft-start has
completed. This corresponds to the voltage on SS reaching 1.5V. For a 1µF soft-start capacitor, this delay is
approximately 1 second. This provides enough time for all
supplies (VDD, setpoint reference and VREF) to settle at
their final values.
TEC Voltage Clamping
An internal clamp circuit is included to protect the TEC
from an overvoltage condition. When the differential voltage across the TEC exceeds 2.5V, the error amplifier
output voltage at the input of the PWM comparator is
limited. This clamps the duty cycle of the output drivers,
and therefore, the voltage across the TEC. The voltage
where clamping occurs can be increased by placing a
resistor divider in parallel with the TEC and by making the
appropriate connections to TEC + and TEC – as shown in
Figure 7. The divider increases the voltage across the TEC,
VTECOOLER, where the clamp activates, to:
RTE1 RTE1
RTE1
+
1+
• 2.5 – VCM
200k
RTE2 100k
VTECOOLER =
R
1 + TE1
200k
VREF
R2
VSET
R1
R3
VTECOOLER
+
VTHRM
RTH
10k
NTC
Figure 6. Modifying the Acceptable Thermistor Range
TEC
TEC +
TEC –
1923 F06
–
RTE2
VCM
RTE1
1923 F07
Figure 7. Increasing Voltage Clamp Threshold
1923f
14
LTC1923
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OPERATIO
The terms containing the fixed resistance values are the
loading errors introduced by the input impedance of the
differential amplifier. A common mode voltage error is
also introduced since the addition of RTE1 and RTE2 change
the fully differential nature of the amplifier. In order to
minimize these errors select RTE1 and RTE2 to be 10k or
less. The above equation reduces to:
R
VTECOOLER ≅ 1 + TE1 2.5
RTE2
The Higher Voltage Applications section shows a fully
differential means to increase the clamp voltage.
This will similarly alter the heating and cooling direction
thresholds by the same factor, increasing the thresholds
to (RTE1 and RTE2 are assumed to be ≤10k):
R
DIRH = 50mV 1 + TE1
RTE2
R
DIRL = – 50mV 1 + TE1
RTE2
The output voltage on the VTEC pin, VVTEC, will be reduced
by the same ratio:
VVTEC =
VTECOOLER
R
1 + TE1
RTE2
Oscillator Frequency
The oscillator determines the switching frequency and the
fundamental positioning of all harmonics. The switching
frequency also affects the size of the inductor that needs
to be selected for a given inductor ripple current (as
opposed to TEC ripple current which is a function of both
the filter inductor and capacitor). A higher switching
frequency allows a smaller valued inductor for a given
ripple current. The oscillator is a triangle wave design. A
current defined by external resistor RT is used to charge
and discharge the capacitor CT. The charge and discharge
rates are equal. The selection of high quality external
components (5% or better multilayer NPO or X7R ceramic
capacitor) is important to ensure oscillator frequency
stability.
The frequency of oscillation is determined by:
fOSC(kHz) = 750 • 106/[RT(kΩ) • CT(pF)]
The LTC1923 can run at frequencies up to 1MHz. The value
selected for RT will also affect the delay time between one
side of the full bridge turning off and the opposite side
turning on. This time is also known as the “break-beforemake” time. The typical value of 10kΩ will produce a 90ns
“break-before-make” time. For higher frequency applications, a smaller value of RT may be required to reduce this
delay time. For applications where significant slew rate
limiting or external gate driver chips are used, a higher
value for RT may necessary, increasing the dead time. The
“break-before-make” time can be approximately calculated by:
tDELAY = RT (kΩ) • 5.75 • 10–9 + 35ns
Phase-Locked Loop
The LTC1923 has an internal voltage-controlled oscillator
(VCO) and phase detector comprising a phase-locked
loop. This allows the oscillator to be synchronized with
another oscillator by slaving it to a master through the
SDSYNC pin. The part can also be designated as the
master by pulling the PLLLPF pin high to VDD. This will
result in the part toggling the SDSYNC pin at its set
oscillator frequency. This signal can then be used to
synchronize additional oscillators.
When being slaved to another oscillator, the frequency
should be set 20% to 30% lower than the target frequency.
The frequency lock range is approximately ±50%.
The phase detector is an edge sensitive digital type, which
provides zero degrees phase shift between the external
and internal oscillators. This detector will not lock up on
input frequencies close to the harmonics of the VCO center
frequency. The VCO hold-in range is equal to the capture
range dfH = dfC = ±0.5fO.
The output of the phase detector is a complementary pair
of current sources charging or discharging the external
filter network on the PLLLPF pin. A simplified block
diagram is shown in Figure 8.
1923f
15
LTC1923
U
OPERATIO
the oscillator is slaved to an external clock. Figure 9c
illustrates how one LTC1923 can be used as a master to
synchronize other LTC1923s or additional devices requiring synchronization. To implement this, determine the
values of RT and CT to obtain the desired free-running
oscillator frequency of the master by using the equation
given in the oscillator frequency section. Tie the master’s
PLLLPF pin to VDD and the SDSYNC pin to VDD through a
resistor RPLL as shown in Figure 9c. RPLL typically can be
set to 10k, but may need to be a lower value if higher
frequency operation is desired (above 250kHz). Set the
slave free-running frequencies to be 20% to 30% less
than this. The SDSYNC pin of the master will switch at its
free-running frequency (with approximately 50% duty
cycle), and this can be used to synchronize the other
devices.
If the external frequency (fPLLIN) is greater than the oscillator frequency, current is sourced continuously out of the
PLLLPF pin. When the external frequency is less than the
oscillator frequency, current is sunk by the PLLLPF pin.
The loop filter components RLP, CLP and CLP2, smooth out
current pulses from the phase detector and provide a
stable input to the VCO. These components also determine
how fast the loop acquires lock. In most instances CLP2
can be omitted, RLP can be set to 1k and CLP can be
selected to be 0.01µF to 0.1µF to stabilize the loop. Make
sure that the low side of filter components is tied to AGND
to keep unwanted switching noise from altering the performance of the PLL.
Figure 9 illustrates three different ways to set the oscillator frequency. In Figure 9a, the oscillator is free running
with the frequency determined by RT and CT. In Figure 9b,
VDD
VDD
RPLL
EXTERNAL
FREQUENCY
RT
SDSYNCB
PLLLPF
DIGITAL
PHASE
FREQUENCY
DETECTOR
CT
OSC
RLP
CLP
CLP2
1923 F08
Figure 8. Phase-Locked Loop Block Diagram
NC
PLLLPF
RT
PLLLPF
RT
CLP2
SDSYNC
RT
CLP
LTC1923
VDD
CT
CLKIN
LTC1923
SDSYNC
CT
1923 F09a
CT
(9b) Slave Operation with External Clock—
Set Oscillator Frequency at 70% to 80% of External Clock
MASTER
PLLLPF
SLAVE
RT
PLLLPF
RT
RPLL
CT
1923 F09b
(9a) Free Running
VDD
RT
RLP
CT
1.2 • RT
LTC1923
SDSYNC
CT
RT
RLP
CLP
LTC1923
SDSYNC
CLP2
CT
1923 F09c
CT
(9c) Master/Slave Operation—Set Oscillator Frequency of Slave at 70% to 80% of Master
Figure 9. Oscillator Frequency Setup: a) Free Running b) Slaved Operation c) Master/Slave Operation
1923f
16
LTC1923
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The peak inductor current is equal to ITEC + ∆I1/2 and is
the current level that trips the current limit comparator.
Keeping the ripple current component small relative to
ITEC keeps the current limit trip level equal to the current
flowing through the TEC.
The thermistor may be isolated from the control circuitry.
It has a relatively high input impedance and is therefore
susceptible to noise pick up. Extreme care should be taken
to ensure this signal is noise free by shielding the line
(coaxially). A lowpass filter can be added between the
thermistor and the input to the LTC2053, but since it is in
the signal path, there are limitations on how much filtering
can be added.
Example: VBRIDGE = 5V, RTEC = 2.5Ω, VTEC = 2.5V,
ITEC = 1A, L = 22µH, fOSC = 250kHz. The peak-to-peak
ripple current using the above equation is:
∆I1 = 170mA
Inductor Ripple Current
The current that flows in the bridge can be separated into
two components, the DC current that flows through the
TEC and the inductor ripple current that is present due to
the switchmode nature of the controller. Although the TEC
current has its own ripple component, proper filtering will
minimize this ripple relative to the inductor ripple current,
validating this assumption that the TEC current is constant
(see TEC Ripple Current section). A simplified half-circuit
of the bridge in steady-state is shown in Figure 10. The
current, IL, through the inductor (L) consists of the ripple
current (∆I1) and static TEC current (ITEC). The ripple
current magnitude, ∆I1, can be calculated using the following equation:
∆I1 = (VBRIDGE 2 – VTEC 2)/(4 • fOSC
The peak inductor current is therefore 1.085A in order to
get 1A of DC TEC current.
TEC Ripple Current
Every TEC has a fundamental limitation (based mainly on
the TEC’s physical characteristics) on the maximum
temperature differential that it can create between sides.
The ability to create this maximum temperature differential is affected by the amount of ripple current that flows
through the device, relative to the DC component. An
approximation of this degradation due to TEC ripple current is given by the following equation:
dT/dTMAX = 1/(1 + N2)
• L • VBRIDGE)
where:
where
dT is the adjusted achievable temperature differential
dTMAX is the maximum possible temperature differential when the TEC is fed strictly by DC current and is
typically specified by the manufacturer
N is the ratio of TEC ripple current to DC current
VBRIDGE is the full-bridge supply voltage (typically VDD)
fOSC is the oscillator frequency
L is the filter inductor value
VTEC is the DC voltage drop across the TEC
TEC manufacturers typically state that N should be no
greater than 10%.
VBRIDGE
PDRVA
MPA
NDRVB
MNB
IL
L
ITEC
ESR
+
TEC
VTEC/2
C
1/2 VBRIDGE
1923 F10
Figure 10. Full-Bridge Half Circuit
1923f
17
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APPLICATIO S I FOR ATIO
In this application, the bridge supply voltage, oscillator
frequency and external filter components determine the
amount of ripple current that flows through the TEC.
Higher valued filter components reduce the amount of
ripple current through the TEC at the expense of increased
board area. Filter capacitor ESR along with inductor ripple
current will determine the peak-to-peak voltage ripple
across the TEC and therefore the ripple current since the
TEC appears resistive.
The ripple current through the TEC, ITEC(RIPPLE), is
approximately equal to:
ITEC(RIPPLE) ≅
VBRIDGE2 – VTEC2
16 • fOSC2 • L • C • RTEC • VBRIDGE
+
(V
2
BRIDGE
)
– VTEC2 • ESR
2 • fOSC • L • VBRIDGE • RTEC
where:
fOSC = the oscillator frequency
L = the filter inductor value
C = the filter capacitor value
RTEC = the resistance of the TEC
VTEC = the DC voltage drop across the TEC
ESR = the equivalent series resistance of the filter
capacitor
VBRIDGE = the full-bridge supply voltage typically equal
to VDD
The equation above shows that there are two components,
which comprise TEC ripple current. The first term is the
increase in voltage from the charging of the filter capacitor. The second term is due to the filter capacitor ESR and
is typically the dominant contributor. Therefore the filter
capacitor selected wants to have a low ESR. This capacitor
can be made of multilevel ceramic, OS-CON electrolytic or
other suitable capacitor. Increasing the oscillator frequency will also reduce the TEC ripple current since both
terms have an inverse relationship to operating frequency.
Example: VBRIDGE = 5V, RTEC = 2.5Ω, VTEC = 2.5V,
L = 22µH, C = 22µF, fOSC = 250kHz, ESR = 100mΩ
ITEC(RIPPLE) = 3.1mA + 13.6mA = 16.7mA
18
For this example the DC current flowing through the TEC
is 1A, making the ripple current equal to approximately 1.7%
(this illustrates why ITEC can be approximated to be DC).
Closing the Feedback Loop
Closing the feedback loop around the TEC and thermistor
(or other temperature sensitive element) involves identifying where the thermal system’s poles are located and
placing electrical pole(s) (and zeroes) to stabilize the
control loop. High DC loop gain is desirable to keep
extremely tight control on the system temperature. Unfortunately the higher the desired loop gain, the larger the
compensation values required to stabilize the system.
Given the inherently slow time constants associated with
thermal systems (on the order of many seconds), this can
lead to unreasonably large component values. Therefore,
the amount of loop gain necessary to maintain the desired
temperature accuracy should be calculated, and after
adding some margin, this should be the target DC loop
gain for the system. A block diagram of the system is
shown in Figure 11. The gain blocks are as follows:
KIA = instrumentation amplifier gain (V/V)
KEA = error amplifier gain (V/V)
KMOD = modulator gain (d/V)
KPWR = power stage gain (V/d)
KTEC = TEC gain (°C/V)
KTHRM = Thermistor Gain (V/°C)
KIA and KEA are the electrical gains associated with the
instrumentation and LTC1923 error amplifier. Switching
regulators are sampled systems that convert voltage to
duty cycle (d), which explains why the KMOD and KPWR
gain terms are expressed as a function of duty cycle and
voltage. The TEC converts voltage to temperature change,
while the thermistor’s impedance and therefore voltage
across it changes with temperature.
The loop gain can be expressed by the following equation:
T (loop gain) = KIA • KEA • KMOD • KPWR • KTEC • KTHRM
And the error introduced by the finite gain of the system,
VE, can be expressed by:
VE = VIN/(1 + T)
1923f
LTC1923
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This voltage error translates back into a temperature
setpoint error.
KMOD • KPWR = 2 • VDD/VCT = 2 • VDD
where VCT = the CT voltage which has a fixed 1V amplitude.
Example:
The TEC gain depends upon the TEC selected and corresponds to the relationship between the voltage across the
device and what temperature differential is created. This
gain term changes with operating temperature, and whether
the TEC is heating or cooling. TECs are inherently more
efficient at heating (and therefore have a higher gain) as
compared to cooling. A worst-case rough estimation of
the gain can be obtained by taking the maximum TEC
voltage required to force a given change in temperature
from the TEC specifications:
RTHRM = 10k
NTC with 4.4%/°C at 25°C
R1 = 10k
VREF = 2.5V
T = 25°C
For this thermistor with a 25°C temperature setpoint, the
change in thermistor voltage with temperature is given by
– 25mV/°C. In order to maintain a 0.01°C temperature
accuracy, this translates into a 250µV error signal, VE. The
minimum loop gain can now be calculated from the above
equation:
KTEC = dT/VTEC(MAX)
The thermistor gain should be linearized around temperature setpoint.
VE = VIN/(1 + T)
Example:
A 25°C setpoint temperature requires VIN = 1.25V for
VREF = 2.5V. The required loop gain is 5000 or 74dB.
Setpoint T = 25°C
VDD = 5V
RTHRM = 10k NTC with 4.4%/°C at 25°C
R1 = 10k
VREF = 2.5V
dT/VTEC(MAX) = 45°C/1.5V = 30°C/V
There are two handles to adjust the loop gain, KIA and KEA,
while the other handles are fixed and depend upon the TEC
and thermistor characteristics (KTEC and KTHRM), VSET and
R1 (KTHRM) and VDD (KMOD and KPWR). The modulator and
power gain product is given by:
KTHRM
KIA
KEA
KMOD
KPWR
KTEC
VTECOOLER
VREF
R1
10k
LTC2053
+
10k
NTC
–
+
CT
+
POWER
STAGE
+
VE
VIN
LTC1923
+
–
RA
ERROR
AMP
TEC
–
1923 F11
–
RF
CF
Figure 11. Simplified Loop Block Diagram
1923f
19
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The linearized thermistor gain around 25°C is – 25mV/°C.
For a minimum loop gain of 5000 as calculated above, the
combined gain of the instrumentation and error amplifiers
can be calculated:
KIA • KEA = T/(KMOD • KPWR • KTEC • KTHRM)
KIA • KEA = 5000/(10 • 30 • 0.025) = 667
A combined gain of 1000 can be selected to provide
adequate margin. The instrumentation amplifier gain should
be set at typically 10, as this attenuates any errors by its
gain factor. The error amplifier gain would then be limited
to the remainder through the gain setting resistors, RF and
RA shown in Figure 11.
RF/RA = KEA – 1
The multiple poles associated with the TEC/thermistor
system makes it difficult to compensate. Compounding
this problem is that there will be significant variations in
thermal time constants for the same system, making
elaborate compensation schemes difficult to reliably implement. The most robust method (i.e., least prone to
oscillation) is to place a dominant pole well below the
thermal system time constant (τ) (anywhere from many
seconds to minutes). This time constant will set the
capacitor value by the following equation:
CF = τ/RF
Please refer to Application Note 89 for more detailed
information on compensating the loop. Ceramic capacitors are not recommended for use as the integrating
capacitor or anywhere in the signal path as they exhibit a
piezoelectric effect which can introduce noise into the
system. The component values shown on the front page of
this data sheet provide a good starting point, but some
adjustment may be required to optimize the response.
Dominant pole compensation does have its limitations. It
provides good loop response over a wide range of laser
module types. It does not provide the fastest transient
response to step changes in temperature. If this is a
necessity, a more complex compensation approach as
shown in Figure 12 may be required. This approach adds
an additional zero into the feedback loop to speed up the
transient response. First note that the LTC2053 inputs
have been swapped as the LTC1923 error amplifier is now
running in an inverting configuration. Capacitor CA is
needed to provide the lead term. Resistor RC is used to
buffer the LTC2053 from capacitive loading and limit the
error amplifier high frequency gain.
Since the system thermal pole locations are not known, a
qualitative compensation approach must be employed.
This entails looking at the transient response when the
TEC is heating (due to the inherent higher gain) for a smallsignal step change in temperature and modifying compensation components to improve the response. A reasonable
starting point is to select components that mimic the
response that will be obtained from the front page of this
data sheet. Therefore RA, RB and CB would be selected to
be 1MΩ, 1MΩ and 0.47µF, respectively. RC should be
selected to be a factor of 100 smaller than RA, or on the
order of 10k. Make sure that the loop is stable prior to the
introduction of capacitor CA. The addition of CA will
provide some phase boost in the loop (in effect, offsetting
one of the poles associated with the thermal system). Start
CC
10k
TMP
CMD
CA
VOUT
LTC1658
+
REF
RC
LTC2053
–
10k
NTC
CB
RB
REF
RA
6
FB
A = 10
4
CNTRL
–
LTC1923
+
5
EAOUT
ERROR
AMPLIFIER
1923 F12
Figure 12. Alternative Compensation Method to Improve Transient Response
1923f
20
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with CA on the order of CB and note its affect on system
response. Adjust the values based on observing whether
the transient response was improved or not with the goal
of reducing CB to improve settling time. As the system
thermal poles can vary between “identical” laser modules
(i.e., same manufacturer and model), care must be taken
to ensure that the values selected provide the desired
response even with these thermal term variations. Compensation should also be tailored for each unique laser
module as thermal terms can vary significantly between
different brands. CC rolls off high frequency gain , minimizing noise in the outputs. It is typically about 25 times
smaller than CB. CA, CB and CC should be film capacitors.
cantly to temperature stability. The relatively mild operating conditions inside the laser module promote good longterm thermistor stability. A high quality, low temperature
coefficient resistor should be selected to bias the thermistor. If the 10k resistor has a 100ppm/°C temperature
coefficient, this translates into a 0.18°C setpoint temperature differential over a 0°C to 70°C ambient for a desired
25°C laser setpoint. Depending upon the temperature
stability requirements of the system, this is very significant. A lower temperature coefficient resistor may therefore be desired. The LTC2053 has maximum offset drift to
50nV/°C which translates into less than 0.001°C change
for a 0°C to 70°C ambient.
Temperature Stability
The offset drift of the LTC1923 error amplifier divided by
the gain of the LTC2053 also affects temperature stability.
The offset drift of the LTC1923 (see characteristic curves)
is typically 1mV over a 0°C to 70°C ambient. After attenuation by the LTC2053 gain, this translates into a temperature setpoint variation of 0.004°C. Neither of these offsets
drifts significantly with aging. Depending upon the setpoint
temperature stability requirements of the system, the
LTC2053 instrumentation amplifier may not be necessary.
Figure 13 shows a simplified schematic with the LTC2053
omitted.
It is important to differentiate between temperature accuracy and stability. Since each laser’s output maximizes at
some temperature, temperature setpoint is typically
incremented until this peak is achieved. After this, only
temperature stability is required. The predominant parameters which affect temperature stability are the thermistor,
the thermistor biasing resistor and any offset drift of the
front-end electrical circuitry. Sufficient loop gain ensures
that any downstream variations do not contribute signifi-
10k
CNTRL
4
10k
NTC
TMP
CMD
REF
100k
FB
6
VOUT
LTC1658
10M
+
LTC1923
–
ERROR
AMPLIFIER
4.7µF
EAOUT
5
1923 F13
Figure 13. Simplifed Temperature Control Loop Omitting
the LTC2053 Instrumentation Amplifier Front End
1923f
21
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Noise and Slew Rate Control
One disadvantage of switching regulators is that the
switching creates wideband harmonic energy. The high
frequency content can pose problems to associated circuitry. To combat this issue, the LTC1923 offers a pin
called RSLEW that controls the slew rate of the output drive
waveforms. Slowing down the transition interval reduces
the harmonic frequency content by spreading out the
energy over a longer time period. The additional transition
time causes some efficiency loss (on the order of 2% to
3%) but significantly improves the high frequency noise
reflected onto the input supply.
Slew rate control is engaged by placing a resistor from
RSLEW to AGND. If slew rate control is not desired, the RSLEW
pin should be tied to VDD allowing the output drivers to
transition at their fastest rate. The resistor value should be
set between 10k (fastest transition) and 300k (slowest transition). This provides about a 10:1 slew rate range to optimize noise performance. The “break-before-make” time
may need to be increased if slew control is implemented,
especially for slower transition rates. Adjustment can be
done by increasing the value of RT (CT can be reduced to
maintain the same frequency of operation), to ensure that
the bridge MOSFETs receive nonoverlapping drive.
Power MOSFET Selection
Four external MOSFETs must be selected for use with the
LTC1923; a pair of N-channel MOSFETs for the bottom of
the bridge and a pair of P-channel MOSFETs for the top
diagonals of the bridge. The MOSFETs should be selected
for their RDS(ON), gate charge and maximum VDS, VGS
ratings. A maximum VDS rating of 20V is more than
sufficient for 5V and 12V bridge applications, but as
mentioned in the High Voltage Application section, a 12V
maximum VGS rating is insufficient and higher voltage
MOSFETs must be selected. There is a trade-off between
RDS(ON) and gate charge. The RDS(ON) affects the conduction losses (ITEC2 • RDS(ON)), while gate charge is a
dominant contributor to switching losses. A higher RDS(ON)
MOSFET typically has a smaller gate capacitance and thus
requires less current to charge the gate for the same
BVDSS. For 1A TEC applications, the Si9801DY or Si9928DY
complimentary N- and P-channel MOSFETs provide a
good trade-off between switching and conduction losses.
Above this TEC current level the MOSFETs selected should
have lower RDS(ON) to maintain the high end efficiency.
Efficiency Considerations
Unlike typical voltage regulators, where the output voltage
is fixed, independent of load current, the output voltage of
this regulator changes with load current. This is because
the TEC appears resistive and the current through the TEC
sets the voltage. The output power of the regulator is
defined as:
POUT = ITEC2 • RTEC
The efficiency of a switching regulator is equal to the
output power divided by the input power times 100%.
Often it is useful to analyze individual losses to determine
what is limiting the efficiency and which change would
produce the most significant improvement. Efficiency can
be expressed as:
Efficiency = 100% – (L1 + L2 + L3 + …)
where L1, L2, etc. are the individual losses as a percentage
of input power.
For this application, the main efficiency concern is typically at the high end of output power. A higher power loss
translates into a greater system temperature rise, resulting in the need for heat sinking, increasing both the system
size and cost.
There are three main sources which usually account for
most of the losses in the application shown on the front
page of the data sheet: Input supply current, MOSFET
switching losses and I2R losses.
1) The input supply current is comprised of the quiescent
current draw from the LTC1658, LTC2053, LTC1923 and
any additional circuitry added. The total maximum supply
current for these devices is on the order of 5mA, which
gives a total power dissipation of 25mW. This power loss
is independent of TEC current.
2) The MOSFET driver current results from switching the
gate capacitance of the power MOSFETs. Each time a gate
is switched from low to high to low again, a packet of
charge dQ moves from VDD to ground. The gate charging
current, IGATECHG = 2 • f • (QP + QN), where QP and QN are
1923f
22
LTC1923
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Total series resistance = 0.055 + 0.08 + 2 • 0.1 + 0.1
= 0.435Ω
the total gate charges of the NMOS and PMOS on one side
of the bridge, and f is the oscillator frequency. The factor
of 2 arises from there being two sets of MOSFETs that
make up the full bridge. Note that increasing the switching
frequency will increase the dynamic current and therefore
power dissipation by the same factor. This power loss is
independent of TEC current.
Power Loss = (1A)2 • 0.435Ω = 0.435W
Output Power = (1A)2 • 2.5Ω = 2.5W
This represents a 17% efficiency loss due to conduction
losses. The other two power loss mechanisms comprise
a little more than a 3% efficiency loss at this output power
level. This may sound alarming if electrical efficiency is the
primary concern and can be easily improved by choosing
lower RDS(ON) MOSFETs, lower series resistance inductors and a smaller valued sense resistor. If temperature
rise is the primary concern, this power dissipation may be
acceptable. At higher current levels, this example does
illustrate that lower resistance components should be
selected.
Example: QN = 10nC max, QP = 15nC max, f = 225kHz,
VDD = 5V
Power loss = 2 • f • (QP + QN) • VDD = 56mW
3) The DC resistances of the external bridge MOSFETs,
filter inductors and sense resistor are typically the dominant loss mechanism at the high end TEC current. The
conduction path of the current includes one NMOS, one
PMOS, two inductors and the sense resistor so the DC
resistances associated with the components dissipate
power.
Low Voltage Requirements
All components shown on the front page of this data sheet
will operate with a 2.7V input supply. Minor modifications
are required to guarantee correct operation. The voltage
on the REF input of the LTC2053 should be at least 1V
below VDD. Figure 14 shows how to implement this. By
dividing down the 2.5V reference with 500Ω of impedance, feeding this to the REF input of the LTC2053 and the
integrating resistor of the LTC1923 error amplifier, any
common mode issues will be avoided.
Example:
RDS(ON)NMOS at 5V = 0.055Ω max
RDS(ON)PMOS at 5V = 0.08Ω max
RS = 0.1Ω
RL = 0.1Ω,
ITEC = 1A
RTEC = 2.5Ω
250Ω
10k
1µF
250Ω
10k
NTC
+
REF
REF
CNTRL VREF
LTC2053
LTC1658
VOUT
–
1µF
EAOUT
A = 10
4.7µF
100k
10M
LTC1923GN
FB
VDD
2.7V TO 3.3V
VDD
VSET
VTHRM
1923 F14
Figure 14. Low Input Supply Voltage Circuit
1923f
23
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Higher Voltage Applications
A bank of TECs can be wired in series to minimize board
real estate utilized by the application. A higher voltage
supply may be required depending upon how many TECs
are placed in series and what their maximum voltage
drop␣ is. In other applications, only one high current
supply may be available, with the output voltage of this
supply being greater than the LTC1923’s absolute maximum voltage rating. The absolute maximum input voltage
for the LTC1923 is 6V. Since the current drawn by the
LTC1923 is small, it can be powered from a low current,
5V (or less) supply. A 12V application for driving the full
bridge is shown in Figure 15. Two LTC1693-1 high speed
dual MOSFET drivers are used to step up the lower voltage
produced by the LTC1923 drivers to the higher voltage
levels required to drive the full bridge. The LTC1693
requires proper bypassing and grounding due to its high
switching speed and large AC currents. Mount the low
ESR bypass capacitors as close to the pins as possible,
shortening the leads as much as possible to reduce
inductance. Refer to the LTC1693 data sheet for more
information. Since the LTC1693-1 low-to-high and highto-low propagation delays are almost identical (typically
35ns), there is minimal skew introduced by the addition
of these drivers. Sufficient dead time (typically 50ns)
between one leg of the bridge shutting off and the other
turning on, as set up by the LTC1923, will be maintained.
If this dead time is insufficient, the resistor tied to the R T
pin can be increased to increase this time.
Care must be taken to ensure that the external MOSFETs
are properly selected based on the maximum drain-source
voltage, VDS, gate-source voltage, VGS, and RDS(ON). Many
MOSFETs that have an absolute maximum VDS of 20V
have a maximum VGS of only 12V, which is insufficient for
12V applications. Even the 14V maximum VGS rating of the
Si9801DY may not provide adequate margin for a 12V
bridge supply voltage. Refer to Efficiency Considerations
for more discussion about selecting a MOSFET with
RDS(ON).
Two pairs of resistors, RT1 and RT2, must be added to
ensure that the absolute maximum input voltage is not
exceeded on the TEC + and TEC – inputs. The maximum
voltage on TEC + and TEC – must be less than the VDD input
supply to the LTC1923 which, for this example, is 5V. The
following equation will guarantee this:
VBRIDGE
< VDD
RT 1 RT 1
+
1+
RT 2 100k
where VBRIDGE is the supply voltage to the external bridge
circuitry and VDD is the input supply to the LTC1923.
These additional level shifting resistors affect some parameters in the data sheet. The direction comparator
thresholds are increased to:
(1 + RT1/RT2 + RT1/100k) • 50mV and
(1 + RT1/RT2 + RT1/100k) • – 50mV
The output voltage on the VTEC pin represents the voltage
across the TEC (VTECOOLER) reduced by a factor of
(1 + RT1/RT2 + RT1/100k) or:
VVTEC = VTECOOLER/(1 + RT1/RT2 + RT1/100k)
The term containing 100k is the loading error introduced
by the input impedance of the differential amplifier. Typically this value will be 100k, but can vary due to normal
process tolerances and temperature (up to ±30%). Due to
this variability, it may be desirable to minimize the loading
effect to try to keep a tight tolerance on the TEC clamp
voltage. Although it will increase quiescent current draw,
this can be accomplished by making the value of RT1 as
small as possible.
As a result of this level shifting, the TEC voltage necessary
to activate the clamp is raised. The voltage across the TEC
where the voltage clamp activates will be:
VTECOOLER = (1 + RT1/RT2 + RT1/100k) • 2.5V
One drawback with using the LTC1693 MOSFET drivers is
the inability to adjust the slew rate of the output drivers to
reduce system noise.
1923f
24
LTC1923
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PLLLPF
RT
RSLEW
CT
12V
LTC1693-1
SDSYNCB
VREF
CNTRL
PDRVB
EAOUT
NDRVB
IN1
IN2
FB
VDD
AGND
PGND
LTC1923
SS
NDRVA
ILIM
PDRVA
VSET
CS +
VCC1
VCC2
CS –
VTHRM
ITEC
47µF
4.7µF
GND2 OUT2
0.1µF
5V
1µF
TEC
0.1µF
4.7µF
LTC1693-1
IN1
VCC1
GND1 OUT1
FAULT
10µF
GND1 OUT1
IN2
3
VCC2
1
RS
2
GND2 OUT2
4
H/C
TEC +
VTEC
TEC –
RT1
RT1
RT2
RT2
1923 F15
Figure 15. Higher Voltage Applications with the LTC1923
1923f
25
26
CS/LD
DIN
CLK
C2
0.1µF
8
VCC
6
VDD
D3
LED (RED)
FAULT
Q3
2N7002
RT1
THERMISTOR
10k NTC
5
Q4
2N7002
R20
100k
VDD
NTC –
NTC +
C11
0.1µF
SM
FILM
R7
10k 1%
OPTIONAL FOR VISUAL FAULT INDICATION
R19
200Ω
4
REF
CLK
U2
7
2
LTC1658IMS8
DIN
VOUT
14 BIT
3
CS/LD
DOUT
GND
1
C3
0.1µF
R3
VSET 10k
0.1%
2
3
5
R27
200Ω
D5
LED (GRN)
COOL
R9
9.09K
1%
R25
200Ω
Q6
2N7002
VDD
C19
1µF
R14
1k 1%
C10
0.1µF
SM FILM
U3
1 LTC2053IMS8
7
C4
0.1µF
D4
LED (ORN)
HEAT
R13
100k 1%
R10
10M
1%
0.1µF
VDD
Q5
2N7002
R23
100k
9
8
7
6
5
4
3
TEC –
TEC +
ITEC
CS –
CS +
PDRVA
NDRVA
PGND
15
16
17
18
19
20
21
22
23
24
25
26
27
28
R1 10k
C6 1µF
C14
1µF
VDD
Q1
Si9801DY
VDD
C15
22µF
6.3V
C7
0.1µF
L1 10µH
TEC –
1923 TA02
C16
22µF 6.3V
TEC
TEC +
C17
22µF
6.3V
C8
0.1µF
Q2
Si9801DY
VDD
C9
10µF
6.3V
L2 10µH
C2, C3, C4, C7, C8: AVX 0603YC104KAT1A (X7R, 16V)
C5: AVX 06035A331KAT1A (NPO, 50V)
C6, C14, C18, C19: AVX 06036D105KAT2A (X5R, 6.3V)
C9: TAIYO YUDEN JMK316BJ106ML-T (X7R, 6.3V)
C10, C11: PANASONIC ECP-U1C104MA5 (SM FILM, 16V)
C13: AVX TAJA475M020R (TANTALUM, 20V)
C15, C16, C17: TAIYO YUDEN JMK325BJ226MM-T (X7R, 6.3V)
D3 (OPTIONAL): PANASONIC LN1251-C-TR (RED)
D4 (OPTIONAL): PANASONIC LN1851-C-TR (ORANGE)
D5 (OPTIONAL): PANASONIC LN1351-C-TR (GREEN)
L1, L2: SUMIDA CDRH6D28-100NC
Q1, Q2: SILICONIX Si9801DY
Q3, Q4, Q5, Q6 (OPTIONAL): 2N7002
R3: KOA RN73T2AT1002B O.1%
R15: KOA SR73H2ER100F
ALL RESISTORS 5% TOLERANCE UNLESS NOTED
R10, R13, C13 MAY NEED TO BE CHANGED TO ACHIEVE DESIRED LOOP RESONSE
R15
0.1Ω
0.5W 1%
VSET
C5 330pF
Laser Temperature Control Loop Achieving Set Point Stability of 0.01°C
R26
100k
VTEC
H/C
VTHRM
FAULT
VSET
ILIM
SS
NDRVB
PDRVB
VREF
CT
RT
VDD
U1
LTC1923EGN
AGND
FB
EAOUT
CNTRL
SDSYNC
RSLEW
PLLLPF
VDD
14
13
12
11
10
C18 1µF
C13
4.7µF
20V
VSET
VDD
1
R4 82k 2
10k
OPTIONAL FOR VISUAL HEATING/COOLING INDICATION
VDD
6
4
R20
100k
VDD
VSET
–
+
8
VDD
+
VDD
LTC1923
TYPICAL APPLICATIO
1923f
U
LTC1923
U
PACKAGE DESCRIPTIO
GN Package
28-Lead Plastic SSOP (Narrow .150 Inch)
(Reference LTC DWG # 05-08-1641)
0.386 – 0.393*
(9.804 – 9.982)
28 27 26 25 24 23 22 21 20 19 18 17 1615
0.229 – 0.244
(5.817 – 6.198)
0.150 – 0.157**
(3.810 – 3.988)
1
0.015 ± 0.004
× 45°
(0.38 ± 0.10)
0.0075 – 0.0098
(0.191 – 0.249)
0.033
(0.838)
REF
2 3
4
5 6
7
8
9 10 11 12 13 14
0.053 – 0.069
(1.351 – 1.748)
0.004 – 0.009
(0.102 – 0.249)
0° – 8° TYP
0.016 – 0.050
(0.406 – 1.270)
0.008 – 0.012
(0.203 – 0.305)
0.0250
(0.635)
BSC
* DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH
SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE
** DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD
FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE
GN28 (SSOP) 1098
1923f
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
27
LTC1923
U
PACKAGE DESCRIPTIO
UH Package
32-Lead Plastic QFN (5mm × 5mm)
(Reference LTC DWG # 05-08-1693)
0.57 ±0.05
5.35 ±0.05
4.20 ±0.05
3.45 ±0.05
(4 SIDES)
PACKAGE OUTLINE
0.23 ± 0.05
0.50 BSC
RECOMMENDED SOLDER PAD LAYOUT
5.00 ± 0.10
(4 SIDES)
BOTTOM VIEW—EXPOSED PAD
R = 0.115
TYP
0.75 ± 0.05
0.40 ± 0.10
31 32
0.00 – 0.05
PIN 1
TOP MARK
1
2
3.45 ± 0.10
(4-SIDES)
(UH) QFN 0102
0.23 ± 0.05
0.200 REF
0.50 BSC
NOTE:
1. DRAWING PROPOSED TO INCLUDE JEDEC PACKAGE OUTLINE
M0-220 VARIATION WHHD-(X) (TO BE APPROVED)
2. DRAWING NOT TO SCALE
3. ALL DIMENSIONS ARE IN MILLIMETERS
4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE
MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.20mm ON ANY SIDE
5. EXPOSED PAD SHALL BE SOLDER PLATED
RELATED PARTS
PART NUMBER
DESCRIPTION
COMMENTS
LTC1658
14-Bit Rail-to-Rail Micropower DAC
3V or 5V Single Supply Operation, ICC = 270µA,
8-Lead MSOP Package
LTC1693-1
High Speed Dual N-Channel MOSFET Driver
1.5A Peak Output Current, 1GΩ Electrical Isolation, SO-8 Package
LTC2053
Zero Drift Instrumentation Amp
Max Gain Error 0.01%, Input Offset Drift of 50nV/°C, Input Offset Voltage of 10µV
1923f
28
Linear Technology Corporation
LT/TP 0502 2K • PRINTED IN USA
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900 ● FAX: (408) 434-0507
●
www.linear.com
LINEAR TECHNOLOGY CORPORATION 2001
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湘ICP备2023018690号
