LTC3586EUFE-1-TRPBF

FEATURES Power Manager n High Efficiency Switching PowerPathTM Controller with Bat-TrackTM Adaptive Output Control and “Instant-On” Operation n Programmable USB or Wall Current Limit (100mA/500mA/1A) n Full Featured Li-Ion/Polymer Battery Charger with Float Voltage of 4.1V (LTC3586-1) or 4.2V (LTC3586) with 1.5A Maximum Charge Current n Internal 180mΩ Ideal Diode Plus External Ideal Diode Controller Powers Load in Battery Mode n VUVLO , IVOUT = 0μA VBUS = 0V, IVOUT = 0μA (Ideal Diode Mode) End-of-Charge Indication Current Ratio (Note 6) 3586fa 3 LTC3586/LTC3586-1 ELECTRICAL CHARACTERISTICS SYMBOL RON_CHG TLIM NTC VCOLD VHOT VDIS INTC Ideal Diode VFWD RDROPOUT IMAX_DIODE VLDO3V3 RCL_LDO3V3 ROL_LDO3V3 VIL VIH IPD FAULT Output VFAULT FAULT Pin Output Low Voltage FAULT Delay FBx Voltage Threshold for FAULT (x = 1, 2, 3, 4) Switching Regulators 1, 2, 3 and 4 VIN1,2,3,4 VOUTUVLO Input Supply Voltage VOUT UVLO—VOUT Falling VOUT UVLO—VOUT Rising Oscillator Frequency FBx Input Current VFBx Servo Voltage VFB1,2,3,4 = 0.85V l The l denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. VBUS = 5V, BAT = 3.8V, VIN1 = VIN2 = VIN3 = VIN4 = VOUT3 = 3.8V, VOUT4 = 5V, RPROG = 1k, RCLPROG = 3.01k, unless otherwise noted. PARAMETER Battery Charger Power FET On-Resistance (Between VOUT and BAT) Junction Temperature in Constant Temperature Mode Cold Temperature Fault Threshold Voltage Hot Temperature Fault Threshold Voltage NTC Disable Threshold Voltage NTC Leakage Current Forward Voltage Internal Diode On-Resistance, Dropout Internal Diode Current Limit Regulated Output Voltage Closed-Loop Output Resistance Dropout Output Resistance Logic Low Input Voltage Logic High Input Voltage Pull-Down Current IFAULT = 5mA 1.2 1 65 14 0.736 100 0mA < ILDO3V3 < 20mA Rising Threshold Hysteresis Falling Threshold Hystersis Falling Threshold Hysteresis VNTC = VBUS = 5V VBUS = 0V, IVOUT = 10mA IVOUT = 10mA VBUS = 0V 1.6 3.1 3.3 4 23 0.4 3.5 75.0 33.4 0.7 –50 2 15 0.18 CONDITIONS MIN TYP 0.18 110 MAX UNITS Ω °C 76.5 1.5 34.9 1.5 1.7 50 78.0 36.4 2.7 50 %VBUS %VBUS %VBUS %VBUS %VBUS mV nA mV mV Ω A V Ω Ω V V μA mV ms V Always On 3.3V Supply Logic Input (EN1, EN2, EN3, EN4, MODE, ILIM0, ILIM1, FAULT) 2.7 VIN1,2,3,4 Connected to VOUT Through Low Impedance. Switching Regulators are Disabled in UVLO 2.5 2.6 2.8 2.25 0.80 5.5 2.9 2.7 50 0.82 V V V MHz nA V fOSC IFB1,2,3,4 VFB1,2,3,4 1.8 –50 0.78 3586fa 4 LTC3586/LTC3586-1 ELECTRICAL CHARACTERISTICS SYMBOL IVIN1,2 PARAMETER Pulse-Skip Mode Input Current Burst Mode® Input Current Shutdown Input Current PMOS Switch Current Limit PMOS RDS(ON) NMOS RDS(ON) Maximum Duty Cycle SW1,2 Pull-Down in Shutdown Input Current PWM Mode, IVOUT3 = 0μA Burst Mode Operation, IVOUT3 = 0μA Shutdown For Burst Mode Operation or PWM Mode 5.5 PWM Mode (Note 5) l l l The l denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. VBUS = 5V, BAT = 3.8V, VIN1 = VIN2 = VIN3 = VIN4 = VOUT3 = 3.8V, VOUT4 = 5V, RPROG = 1k, RCLPROG = 3.01k, unless otherwise noted. CONDITIONS IVOUT1,2 = 0μA, (Note 7) IVOUT1,2 = 0μA, (Note 7) IVOUT1,2 = 0μA, (Note 7) Pulse-Skip/Burst Mode Operation (Note 5) 600 MIN TYP 225 35 800 0.6 0.7 100 10 220 13 0 2.65 5.6 2.5 275 0 3 350 30 2 200 –30 50 0.22 0.17 –1 –1 10 PWM Mode PWM Mode l MAX UNITS μA μA μA mA Ω Ω % kΩ Switching Regulators 1 and 2 (Buck) 60 1 1100 ILIM1,2 RP1,2 RN1,2 D1,2 RSW1,2 IVIN3 Switching Regulator 3 (Buck-Boost) 400 20 1 2.75 μA μA μA V V A mA mA mA Ω Ω 1 1 μA μA kΩ % 75 0.5 FB4 = 0V, IVOUT4 = 0μA Shutdown, VOUT4 = 0V FB4 = 0V (Note 5) 2000 5.1 180 1 7.5 2800 5 5.3 0.3 5.5 % ms μA μA mA mA V V V VOUT3(LOW) VOUT3(HIGH) ILIMF3 IPEAK3(BURST) IZERO3(BURST) IMAX3(BURST) RDS(ON)P RDS(ON)N ILEAK(P) ILEAK(N) RVOUT3 DBUCK(MAX) DBOOST(MAX) tSS3 IVIN4 IVOUT4 ILIMF4 VOUT4 VOV4 ΔVOV4 Minimum Regulated Output Voltage Maximum Regulated Output Voltage Forward Current Limit (Switch A) Forward Burst Current Limit (Switch A) Burst Mode Operation Reverse Burst Current Limit (Switch D) Burst Mode Operation Maximum Deliverable Output Current in 2.7V ≤ VIN3 ≤ 5.5V, 2.75V ≤ VOUT3 ≤ 5.5V Burst Mode Operation (Note 8) PMOS RDS(ON) NMOS RDS(ON) PMOS Switch Leakage NMOS Switch Leakage VOUT3 Pull-Down in Shutdown Maximum Buck Duty Cycle Maximum Boost Duty Cycle Soft-Start Time Input Current Q-Current Drawn from Boost Output NMOS Switch Current Limit Output Voltage Adjust Range Overvoltage Shutdown Overvoltage Shutdown Hysteresis Switches A, D Switches B, C Switches A, D Switches B, C 100 Switching Regulator 4 (Boost) Burst Mode is a registered trademark of Linear Technology Corporation. 3586fa 5 LTC3586/LTC3586-1 ELECTRICAL CHARACTERISTICS SYMBOL RDS(ON)P4 RDS(ON)N4 ILEAK(P)4 ILEAK(N)4 RVOUT4 DBOOST(MAX) tSS4 PARAMETER PMOS RDS(ON) NMOS RDS(ON) PMOS Switch Leakage NMOS Switch Leakage VOUT4 Pull-Down in Shutdown Maximum Boost Duty Cycle Soft-Start Time The l denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. VBUS = 5V, BAT = 3.8V, VIN1 = VIN2 = VIN3 = VIN4 = VOUT3 = 3.8V, VOUT4 = 5V, RPROG = 1k, RCLPROG = 3.01k, unless otherwise noted. CONDITIONS Synchronous Switch Main Switch Synchronous Switch Main Switch –1 –1 10 91 0.375 94 MIN TYP 0.25 0.17 1 1 MAX UNITS Ω Ω μA μA kΩ % ms Note 1: Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. Exposure to any Absolute Maximum Rating condition for extended periods may affect device reliability and lifetime. Note 2: The LTC3586E/LTC3586E-1 are guaranteed to meet performance specifications from 0°C to 85°C. Specifications over the –40°C to 85°C operating temperature range are assured by design, characterization and correlation with statistical process controls. Note 3: The LTC3586E/LTC3586E-1 include overtemperature protection that is intended to protect the device during momentary overload conditions. Junction temperature will exceed 125°C when overtemperature protection is active. Continuous operation above the specified maximum operating junction temperature may impair device reliability. Note 4: Total input current is the sum of quiescent current, IVBUSQ, and measured current given by: VCLPROG/RCLPROG • (hCLPROG +1) Note 5: The current limit features of this part are intended to protect the IC from short term or intermittent fault conditions. Continuous operation above the maximum specified pin current rating may result in device degradation or failure. Note 6: hC/10 is expressed as a fraction of measured full charge current with indicated PROG resistor. Note 7: FBx above regulation such that regulator is in sleep. Specification does not include resistive divider current reflected back to VINX. Note 8: Guaranteed by design. TYPICAL PERFORMANCE CHARACTERISTIC Ideal Diode V-I Characteristics 1.0 INTERNAL IDEAL DIODE WITH SUPPLEMENTAL EXTERNAL VISHAY Si2333 PMOS RESISTANCE (Ω) 0.25 (TA = 25°C unless otherwise noted) Output Voltage vs Output Current (Battery Charger Disabled) 4.50 BAT = 4V VBUS = 5V 5x MODE Ideal Diode Resistance vs Battery Voltage 0.8 CURRENT (A) 0.20 INTERNAL IDEAL DIODE 0.15 OUTPUT VOLTAGE (V) 4.25 0.6 INTERNAL IDEAL DIODE ONLY 0.4 4.00 BAT = 3.4V 3.75 0.10 INTERNAL IDEAL DIODE WITH SUPPLEMENTAL EXTERNAL VISHAY Si2333 PMOS 0.2 VBUS = 0V VBUS = 5V 0 0 0.04 0.12 0.16 0.08 FORWARD VOLTAGE (V) 0.20 3586 G01 0.05 3.50 0 2.7 3.25 3.0 3.6 3.9 3.3 BATTERY VOLTAGE (V) 4.2 3586 G02 0 200 600 800 400 OUTPUT CURRENT (mA) 1000 3586 G03 3586fa 6 LTC3586/LTC3586-1 TYPICAL PERFORMANCE CHARACTERISTICS USB Limited Battery Charge Current vs Battery Voltage 700 LTC3586 600 CHARGE CURRENT (mA) 500 400 300 LTC3586-1 200 100 VBUS = 5V RPROG = 1k RCLPROG = 3k CHARGE CURRENT (mA) 125 100 75 LTC3586-1 50 VBUS = 5V RPROG = 1k 25 RCLPROG = 3k 1x USB SETTING, BATTERY CHARGER SET FOR 1A 0 2.7 3.0 3.3 3.6 3.9 BATTERY VOLTAGE (V) 150 LTC3586 BATTERY DRAIN CURRENT (μA) 20 (TA = 25°C unless otherwise noted) Battery Drain Current vs Battery Voltage 25 IVOUT = 0μA VBUS = 0V USB Limited Battery Charge Current vs Battery Voltage 15 10 VBUS = 5V (SUSPEND MODE, RCLPROG = 3.01k) 5 5x USB SETTING, BATTERY CHARGER SET FOR 1A 0 3.0 3.3 3.6 3.9 2.7 BATTERY VOLTAGE (V) 4.2 3586 G04 4.2 3586 G05 0 2.7 3.0 3.6 3.9 3.3 BATTERY VOLTAGE (V) 4.2 3586 G06 PowerPath Switching Regulator Efficiency vs Output Current 100 90 EFFICIENCY (%) 80 70 60 50 40 0.01 BAT = 3.8V 1x MODE 100 5x, 10x MODE Battery Charging Efficiency vs Battery Voltage with No External Load (PBAT/PBUS) 45 40 VBUS QUIESCENT CURRENT (μA) 90 EFFICIENCY (%) 1x CHARGING EFFICIENCY 5x CHARGING EFFICIENCY 35 30 25 20 15 10 5 0 VBUS Current vs VBUS Voltage (Suspend) 80 70 LTC3586 RCLPROG = 3.01k RPROG = 1K IVOUT = 0mA 3 3.5 3.9 3.3 BATTERY VOLTAGE (V) 4.2 3586 G08 0.1 OUTPUT CURRENT (A) 1 3586 G07 60 2.7 0 1 3 2 VBUS VOLTAGE (V) 4 5 3586 G09 Output Voltage vs Output Current in Suspend 5.0 0.5 VBUS Current vs Output Current in Suspend VBUS = 5V BAT = 3.3V RCLPROG = 3.01k OUTPUT VOLTAGE (V) 3.4 3.3V LDO Output Voltage vs Output Current, VBUS = 0V BAT = 3.9V 4.2V , BAT = 3.4V BAT = 3.5V BAT = 3.6V 4.5 OUTPUT VOLTAGE (V) VBUS CURRENT (mA) 0.4 3.2 4.0 0.3 3.0 3.5 0.2 3.0 VBUS = 5V BAT = 3.3V RCLPROG = 3.01k 0 0.1 0.3 0.4 0.2 OUTPUT CURRENT (mA) 0.5 3586 G10 0.1 2.8 2.5 0 0 0.1 0.3 0.4 0.2 OUTPUT CURRENT (mA) 0.5 3586 G11 2.6 BAT = 3V BAT = 3.1V BAT = 3.2V BAT = 3.3V 0 5 15 20 10 OUTPUT CURRENT (mA) 25 3586 G12 3586fa 7 LTC3586/LTC3586-1 TYPICAL PERFORMANCE CHARACTERISTICS Battery Charge Current vs Temperature 600 500 CHARGE CURRENT (mA) 400 THERMAL REGULATION 300 200 100 RPROG = 2k 10x MODE 0 –40 –20 0 4.17 –40 3.60 –40 OUTPUT VOLTAGE (V) 4.20 FLOAT VOLTAGE (V) 3.66 4.21 (TA = 25°C unless otherwise noted) Low-Battery (Instant On) Output Voltage vs Temperature 3.68 BAT = 2.7V IVOUT = 100mA 5x MODE Battery Charger Float Voltage vs Temperature 4.19 3.64 4.18 3.62 20 40 60 80 TEMPERATURE (°C) 100 120 3586 G13 –15 35 10 TEMPERATURE (°C) 60 85 3586 G14 –15 35 10 TEMPERATURE (°C) 60 85 3586 G15 Oscillator Frequency vs Temperature 2.6 VBUS QUIESCENT CURRENT (mA) 15 VBUS Quiescent Current vs Temperature VBUS = 5V IVOUT = 0μA 5x MODE 70 VBUS QUIESCENT CURRENT (μA) VBUS Quiescent Current in Suspend vs Temperature IVOUT = 0μA 2.4 FREQUENCY (MHz) VBUS = 5V 2.2 BAT = 3V VBUS = 0V BAT = 2.7V VBUS = 0V 1.8 –40 –15 35 10 TEMPERATURE (°C) BAT = 3.6V VBUS = 0V 12 60 9 50 1x MODE 6 2.0 40 60 85 3586 G16 3 –40 –15 35 10 TEMPERATURE (°C) 60 85 3586 G17 30 –40 –15 35 10 TEMPERATURE (°C) 60 85 3586 G18 CHRG Pin Current vs Voltage (Pull-Down State) 100 VBUS = 5V BAT = 3.8V 3.3V LDO Step Response (5mA to 15mA) 50 BATTERY DRAIN CURRENT (μA) ILDO3V3 5mA/DIV 0mA VLDO3V3 20mV/DIV AC COUPLED 3586 G20 Battery Drain Current vs Temperature BAT = 3.8V VBUS = 0V ALL REGULATORS OFF CHRG PIN CURRENT (mA) 80 40 60 30 40 20 20 BAT = 3.8V 20μs/DIV 10 0 0 1 3 4 2 CHRG PIN VOLTAGE (V) 5 3586 G19 0 –40 –15 35 10 TEMPERATURE (°C) 60 85 3586 G21 3586fa 8 LTC3586/LTC3586-1 TYPICAL PERFORMANCE CHARACTERISTICS Switching Regulators 1, 2 PulseSkip Mode Quiescent Currents 325 VIN1,2 = 3.8V 1.95 100 90 1.90 INPUT CURRENT (mA) EFFICIENCY (%) VOUT1, 2 = 2.5V (CONSTANT FREQUENCY) 80 VOUT1, 2 = 1.8V EFFICIENCY (%) 70 60 50 40 30 20 10 85 3586 G22 (TA = 25°C unless otherwise noted) Switching Regulators 1, 2 Burst Mode Efficiency 100 90 80 VOUT1, 2 = 2.5V VOUT1, 2 = 1.2V VOUT1, 2 = 1.8V Switching Regulators 1, 2 Pulse-Skip Mode Efficiency VOUT1, 2 = 2.5V VOUT1, 2 = 1.2V QUIESCENT CURRENT (μA) 300 70 60 50 40 30 20 10 275 1.85 250 1.80 225 VOUT1,2 = 1.25V (PULSE SKIPPING) 200 – 40 –15 35 10 TEMPERATURE (°C) 60 1.75 1.70 0 1 VIN1, 2 = 3.8V 10 100 OUTPUT CURRENT (mA) 1000 3586 G23 0 0.1 VIN1, 2 = 3.8V 1 10 100 OUTPUT CURRENT (mA) 1000 3586 G24 Switching Regulators 1, 2 Load Regulation at VOUT1, 2 = 1.2V 1.230 VBUS = 3.8V Burst Mode OPERATION OUTPUT VOLTAGE (V) 1.845 Switching Regulators 1, 2 Load Regulation at VOUT1, 2 = 1.8V VBUS = 3.8V Burst Mode OPERATION PULSE-SKIP MODE 1.800 OUTPUT VOLTAGE (V) 1.823 2.56 Switching Regulators 1, 2 Load Regulation at VOUT1, 2 = 2.5V VBUS = 3.8V OUTPUT VOLTAGE (V) 1.215 2.53 Burst Mode OPERATION PULSE-SKIP MODE 2.50 1.200 PULSE-SKIP MODE 1.185 1.778 2.47 1.170 0.1 1 10 100 OUTPUT CURRENT (mA) 1000 3586 G25 1.755 0.1 1 10 100 OUTPUT CURRENT (mA) 1000 3586 G26 2.44 0.1 1 10 100 OUTPUT CURRENT (mA) 1000 3586 G27 Buck-Boost Regulator Efficiency vs ILOAD 100 90 80 70 EFFICIENCY (%) 60 50 40 30 20 10 0 0.1 1 VOUT3 = 3.3V TYPE 3 COMPENSATION 10 ILOAD (mA) 100 1000 3586 G28 RDS(ON) For Buck-Boost Regulator 0.30 0.40 0.35 NMOS RDS(ON) (Ω) 0.30 NMOS VIN3 = 3V NMOS VIN3 = 3.6V NMOS VIN3 = 4.5V 0.25 0.20 0.15 0.10 5 25 45 65 85 105 125 TEMPERATURE (°C) 3586 G29 Buck-Boost Regulator Forward Current Limit 2600 2550 VIN3 = 3.6V 2500 ILIMF (mA) VIN3 = 4.5V 2450 2400 2350 2300 –55 –35 –15 VIN3 = 3V PMOS RDS(ON) (Ω) Burst Mode OPERATION CURVES VIN3 = 3V VIN3 = 3.6V VIN3 = 4.5V PMOS VIN3 = 3V PMOS VIN3 = 3.6V 0.25 PMOS VIN3 = 4.5V PWM MODE CURVES VIN3 = 3V VIN3 = 3.6V VIN3 = 4.5V 0.20 0.15 0.10 0.05 0 –55 –35 –15 5 25 45 65 85 105 125 TEMPERATURE (°C) 3586 G30 3586fa 9 LTC3586/LTC3586-1 TYPICAL PERFORMANCE CHARACTERISTICS Buck-Boost Regulator Burst Mode Operation Quiescent Current 14.0 13.5 13.0 IQ (μA) 12.5 12.0 11.5 11.0 –55 –35 –15 VIN3 = 4.5V 300 250 200 150 100 50 0 5 25 45 65 85 105 125 TEMPERATURE (°C) 3586 G31 (TA = 25°C unless otherwise noted.) Reduction in Current Deliverability at Low VIN1 STEADY STATE ILOAD START-UP WITH A RESISTIVE LOAD START-UP WITH A CURRENT SOURCE LOAD VOUT3 100mV/DIV AC COUPLED Buck-Boost Step Response VIN1 = 3V VIN3 = 3.6V REDUCTION BELOW 1A (mA) 300mA IVOUT3 200mA/DIV 0 VOUT3 = 3.3V TYPE 3 COMPENSATION 2.7 3.1 3.5 3.9 VIN1 (V) 4.3 4.7 3586 G32 VIN3 = 3.8V VOUT3 = 3.3V 100μs/DIV 3586 G33 Boost Efficiency (VIN4 = 3.8V) 100 90 80 EFFICIENCY (%) EFFICIENCY 0.4 POWER LOSS 0.3 0.2 0.1 0 1 10 100 IVOUT4 (mA) 1000 3586 G23 Boost Efficiency vs VIN4 0.7 0.6 0.5 POWER LOSS (W) 100 90 80 EFFICIENCY (%) 70 60 50 40 30 20 10 0 2.6 3 IVOUT4 = 300mA VOUT4 = 5V 3.4 3.8 4.2 4.6 INPUT VOLTAGE VIN4 (V) 5 5.4 3586 G35 Boost Output Voltage vs Temperature 5.000 4.995 4.990 4.985 VOUT4 (V) 4.980 4.975 4.970 4.965 4.960 4.955 4.950 –45 –30 –15 0 15 30 45 60 TEMPERATURE (ºC) 75 90 SYNCH PMOS OFF VIN4 = 2.7V VOUT4 = 5V 70 60 50 40 30 20 10 0 VIN4 = 4.5V VIN4 = 3.8V 3586 G36 Maximum Deliverable Boost Output Current 2200 2000 OUTPUT CURRENT IVOUT4 (mA) 1800 1600 1400 1200 1000 800 600 400 200 0 2.7 3 3.3 3.6 3.9 VIN4 (V) 4.2 4.5 3586 G37 Maximum Boost Duty Cycle vs VIN4 100 Boost Step Response (50mA to 300mA) VIN3 = 3.8V VOUT3 = 3.3V VOUT4 100mV/DIV AC COUPLED 300mA IVOUT4 125mA/DIV 50mA VIN4 = 3.8V VOUT4 = 5V L = 2.2μH C = 10μF 50μs/DIV 3586 G39 L = 2.2μH VOUT4 = 4.9V (SET FOR 5V) MAXIMUM DUTY CYCLE (%) 95 T = 90ºC 90 T = 25ºC T = –45ºC T = 90ºC T = 25ºC T = –45ºC 85 80 2.7 3 3.3 3.6 3.9 VIN4 (V) 4.2 4.5 3586 G38 3586fa 10 LTC3586/LTC3586-1 PIN FUNCTIONS ILIM0, ILIM1 (Pins 1, 2): Logic Inputs. ILIM0 and ILIM1 control the current limit of the PowerPath switching regulator. See Table 1. Table 1. USB Current Limit Settings (ILIM1) 0 0 1 1 (ILIM0) 0 1 0 1 USB SETTING 1x Mode (USB 100mA Limit) 10x Mode (Wall 1A Limit) Suspend 5x Mode (USB 500mA Limit) Mode 0 1 Buck Pulse-Skip Burst SW4 (Pin 8): Switch Node for the (Boost) Switching Regulator 4. An external inductor connects between this pin and VIN4. MODE (Pin 9): Digital Input. The MODE pin controls different modes of operation for the switching regulators according to Table 2. Table 2. Switching Regulators Mode REGULATION MODE Buck-Boost PWM Burst Boost Pulse-Skip Pulse-Skip LDO3V3 (Pin 3): 3.3V LDO Output Pin. This pin provides a regulated always-on 3.3V supply voltage. LDO3V3 gets its power from VOUT. It may be used for light loads such as a watch dog microprocessor or real time clock. A 1μF capacitor is required from LDO3V3 to ground. If the LDO3V3 output is not used it should be disabled by connecting it to VOUT. CLPROG (Pin 4): USB Current Limit Program and Monitor Pin. A resistor from CLPROG to ground determines the upper limit of the current drawn from the VBUS pin. A fraction of the VBUS current is sent to the CLPROG pin when the synchronous switch of the PowerPath switching regulator is on. The switching regulator delivers power until the CLPROG pin reaches 1.188V. Several VBUS current limit settings are available via user input which will typically correspond to the 500mA and 100mA USB specifications. A multilayer ceramic averaging capacitor is required at CLPROG for filtering. NTC (Pin 5): Input to the Thermistor Monitoring Circuits. The NTC pin connects to a battery’s thermistor to determine if the battery is too hot or too cold to charge. If the battery’s temperature is out of range, charging is paused until it re-enters the valid range. A low drift bias resistor is required from VBUS to NTC and a thermistor is required from NTC to ground. If the NTC function is not desired, the NTC pin should be grounded. VOUT4 (Pins 6, 7): Power Output for the (Boost) Switching Regulator 4. A 10μF MLCC capacitor should be placed as close to the pins as possible. FB4 (Pin 10): Feedback Input for the (Boost) Switching Regulator 4. When the control loop is complete, the voltage on this pin servos to 0.8V. FB3 (Pin 11): Feedback Input for (Buck-Boost) Switching Regulator 3. When regulator 3’s control loop is complete, this pin servos to 0.8V. VC3 (Pin 12): Output of the Error Amplifier and Voltage Compensation Node for (Buck-Boost) Switching Regulator 3. External Type I or Type III compensation (to FB3) connects to this pin. See the Applications Information section for selecting buck-boost compensation components. SWAB3 (Pin 13): Switch Node for (Buck-Boost) Switching Regulator 3. Connected to Internal Power Switches A and B. An external inductor connects between this node and SWCD3. VIN3 (Pins 14, 15): Power Input for (Buck-Boost) Switching Regulator 3. These pins will generally be connected to VOUT. A 1μF MLCC capacitor is recommended on these pins. VOUT3 (Pins 16, 17): Output Voltage for (Buck-Boost) Switching Regulator 3. EN3 (Pin 18): Digital Input. This input enables the buck-boost switching regulator 3. SWCD3 (Pin 19): Switch Node for (Buck-Boost) Switching Regulator 3 Connected to Internal Power Switches C and D. An external inductor connects between this node and SWAB3. 3586fa 11 LTC3586/LTC3586-1 PIN FUNCTIONS EN2 (Pin 20): Digital Input. This input enables the buck switching regulator 2. EN1 (Pin 21): Digital Input. This input enables the buck switching regulator 1. VIN4 (Pin 22): Power Input for Switching Regulator 4 (Boost). This pin will generally be connected to VOUT. A 1μF MLCC capacitor is recommended on this pin. FB2 (Pin 23): Feedback Input for (Buck) Switching Regulator 2. When regulator 2’s control loop is complete, this pin servos to 0.8V. VIN2 (Pin 24): Power Input for (Buck) Switching Regulator 2. This pin will generally be connected to VOUT. A 1μF MLCC capacitor is recommended on this pin. SW2 (Pin 25): Power Transmission Pin for (Buck) Switching Regulator 2. SW1 (Pin 26): Power Transmission Pin for (Buck) Switching Regulator 1. VIN1 (Pin 27): Power Input for (Buck) Switching Regulator 1. This pin will generally be connected to VOUT. A 1μF MLCC capacitor is recommended on this pin. FB1 (Pin 28): Feedback Input for (Buck) Switching Regulator 1. When regulator 1’s control loop is complete, this pin servos to 0.8V. PROG (Pin 29): Charge Current Program and Charge Current Monitor Pin. Connecting a resistor from PROG to ground programs the charge current. If sufficient input power is available in constant-current mode, this pin servos to 1V. The voltage on this pin always represents the actual charge current. CHRG (Pin 30): Open-Drain Charge Status Output. The CHRG pin indicates the status of the battery charger. Four possible states are represented by CHRG: charging, not charging, unresponsive battery and battery temperature out of range. CHRG is modulated at 35kHz and switches between a low and a high duty cycle for easy recognition by either humans or microprocessors. See Table 3. CHRG requires a pull-up resistor and/or LED to provide indication. GATE (Pin 31): Analog Output. This pin controls the gate of an optional external P-channel MOSFET transistor used to supplement the ideal diode between VOUT and BAT. The external ideal diode operates in parallel with the internal ideal diode. The source of the P-channel MOSFET should be connected to VOUT and the drain should be connected to BAT. If the external ideal diode FET is not used, GATE should be left floating. BAT (Pin 32): Single Cell Li-Ion Battery Pin. Depending on available VBUS power, a Li-Ion battery on BAT will either deliver power to VOUT through the ideal diode or be charged from VOUT via the battery charger. EN4 (Pin 33): Digital Input. This input enables the boost switching regulator 4. V OUT ( Pin 34): O utput Voltage of the Switching PowerPath Controller and Input Voltage of the Battery Charger. The majority of the portable product should be powered from VOUT. The LTC3586/LTC3586-1 will partition the available power between the external load on VOUT and the internal battery charger. Priority is given to the external load and any extra power is used to charge the battery. An ideal diode from BAT to VOUT ensures that VOUT is powered even if the load exceeds the allotted power from VBUS or if the VBUS power source is removed. VOUT should be bypassed with a low impedance ceramic capacitor. VBUS (Pins 35, 36): Primary Input Power Pin. These pins deliver power to VOUT via the SW pin by drawing controlled current from a DC source such as a USB port or wall adapter. SW (Pin 37): Power Transmission Pin for the USB Power Path. The SW pin delivers power from VBUS to VOUT via the buck switching regulator. A 3.3μH inductor should be connected from SW to VOUT. FAULT (Pin 38): Bi-directional input/output (open-drain) used to alert or receive information from other power management ICs regarding an electrical fault. Exposed Pad (Pin 39): Ground. The Exposed Pad should be connected to a continuous ground plane on the second layer of the printed circuit board by several vias directly under the LTC3586/LTC3586-1. 3586fa 12 LTC3586/LTC3586-1 BLOCK DIAGRAM VBUS 35, 36 2.25MHz PowerPath SWITCHING REGULATOR SUSPEND LDO 500μA CC/CV CHARGER 0.3V 3.6V 3.3V LDO 37 SW 3 LDO3V3 34 VOUT + CLPROG 4 IDEAL – + – 15mV 31 GATE NTC 5 BATTERY TEMPERATURE MONITOR 1.188V CHRG 30 CHARGE STATUS FAULT 38 FAULT LOGIC EN1 21 EN2 20 EN3 18 EN4 33 MODE 9 ILIM0 1 ILIM1 2 VIN4 22 MASTER LOGIC EN1 400mA 2.25MHz (BUCK) SWITCHING REGULATOR 1 EN2 400mA 2.25MHz (BUCK) SWITCHING REGULATOR 2 EN3 VOUT4 SW4 8 6, 7 800mA 2.25MHz (BOOST) SWITCHING REGULATOR 4 EN4 FB4 10 –+ 14, 15 A 1A 2.25MHz (BUCK-BOOST) SWITCHING REGULATOR 3 B 16, 17 D C 39 GND + + – + – 32 BAT 29 PROG 27 VIN1 26 SW1 28 FB1 24 VIN2 25 SW2 23 FB2 VIN3 13 SWAB3 VOUT3 19 SWCD3 11 FB3 12 VC3 3586 BD 3586fa 13 LTC3586/LTC3586-1 OPERATION Introduction The LTC3586/LTC3586-1 are highly integrated power management ICs which include a high efficiency switch mode PowerPath controller, a battery charger, an ideal diode, an always-on LDO, two 400mA buck switching regulators, a 1A buck-boost switching regulator, and an 800mA boost switching regulator. All of the regulators can be independently controlled via ENABLE pins. Designed specifically for USB applications, the PowerPath controller incorporates a precision average input current buck switching regulator to make maximum use of the allowable USB power. Because power is conserved, the LTC3586/LTC3586-1 allow the load current on VOUT to exceed the current drawn by the USB port without exceeding the USB load specifications. The PowerPath switching regulator and battery charger communicate to ensure that the input current never violates the USB specifications. The ideal diode from BAT to VOUT guarantees that ample power is always available to VOUT even if there is insufficient or absent power at VBUS. An always-on LDO provides a regulated 3.3V from available power at VOUT. Drawing very little quiescent current, this LDO will be on at all times and can be used to supply up to 20mA. Along with constant frequency PWM mode, the buck and the buck-boost switching regulators have a low power burst mode setting for significantly reduced quiescent current under light load conditions. High Efficiency Switching PowerPath Controller Whenever VBUS is available and the PowerPath switching regulator is enabled, power is delivered from VBUS to VOUT via SW. VOUT drives the combination of the external load (including switching regulators 1, 2, 3 and 4) and the battery charger. If the combined load does not exceed the PowerPath switching regulator’s programmed input current limit, VOUT will track 0.3V above the battery (Bat-Track). By keeping the voltage across the battery charger low, efficiency is optimized because power lost to the linear battery charger is minimized. Power available to the external load is therefore optimized. If the combined load at VOUT is large enough to cause the switching PowerPath supply to reach the programmed input current limit, the battery charger will reduce its charge current by that amount necessary to enable the external load to be satisfied. Even if the battery charge current is set to exceed the allowable USB current, the USB specification will not be violated. The PowerPath switching regulator will limit the average input current so that the USB specification is never violated. Furthermore, load current at VOUT will always be prioritized and only excess available power will be used to charge the battery. If the voltage at BAT is below 3.3V, or the battery is not present, and the load requirement does not cause the PowerPath switching regulator to exceed the USB specification, VOUT will regulate at 3.6V, as shown in Figure 1. This “instant-on” feature will allow a portable product to run immediately when power is applied without waiting for the battery to charge. If the load exceeds the current limit at VBUS, VOUT will range between the no-load voltage and slightly below the battery voltage, indicated by the shaded region of Figure 1. For very low-battery voltages, the battery charger acts like a load and, due to limited input power, its current will tend to pull VOUT below the 3.6V “instant-on” voltage. To prevent VOUT from falling below this level, an undervoltage circuit automatically detects that VOUT is falling and reduces the battery charge as needed. This reduction ensures that load current and output voltages are always priortized while allowing as much battery charge current as possible. See Over-Programming the Battery Charger in Applications Information Section. The power delivered from VBUS to VOUT is controlled by a 2.25MHz constant-frequency buck switching regulator. To meet the USB maximum load specification, the switching regulator includes a control loop which ensures that the average input current is below the level programmed at CLPROG. 3586fa 14 LTC3586/LTC3586-1 OPERATION The current at CLPROG is a fraction (hCLPROG–1) of the VBUS current. When a programming resistor and an averaging capacitor are connected from CLPROG to GND, the voltage on CLPROG represents the average input current of the PowerPath switching regulator. When the input current approaches the programmed limit, CLPROG reaches VCLPROG , 1.188V and power out is held constant. The input current limit is programmed by the ILIM0 and ILIM1 pins to limit average input current to one of several possible settings as well as be deactivated (USB Suspend). The input current limit will be set by the VCLPROG servo voltage and the resistor on CLPROG according to the following expression: IVBUS = IVBUSQ + VCLPROG • (hCLPROG + 1 ) RCLPROG If the load current increases beyond the power allowed from the switching regulator, additional power will be pulled from the battery via the ideal diode. Furthermore, if power to VBUS (USB or wall power) is removed, then all of the application power will be provided by the battery via the ideal diode. The transition from input power to battery power at VOUT will be quick enough to allow only the10μF capacitor to keep VOUT from drooping. The ideal diode consists of a precision amplifier that enables a large on-chip P-channel MOSFET transistor whenever the voltage at VOUT is approximately 15mV (VFWD) below the voltage at BAT. The resistance of the internal ideal diode is approximately 180mΩ. If this is sufficient for the application, then no external components are necessary. However, if more conductance is needed, an external P-channel MOSFET transistor can be added from BAT to VOUT. See Figure 2. When an external P-channel MOSFET transistor is present, the GATE pin of the LTC3586/LTC3586-1 drive its gate for automatic ideal diode control. The source of the external P-channel MOSFET should be connected to VOUT and the drain should be connected to BAT. Capable of driving a 1nF load, the GATE pin can control an external P-channel MOSFET transistor having an on-resistance of 40mΩ or lower. 2200 2000 4.2 3.9 CURRENT (mA) VOUT (V) NO LOAD 3.6 300mV 3.3 3.0 2.7 2.4 2.4 1800 1600 1400 1200 1000 800 600 400 200 0 2.7 3.0 3.6 3.3 BAT (V) 3.9 4.2 3586 F01 Figure 1 shows the range of possible voltages at VOUT as a function of battery voltage. Ideal Diode from BAT to VOUT The LTC3586/LTC3586-1 have an internal ideal diode as well as a controller for an optional external ideal diode. The ideal diode controller is always on and will respond quickly whenever VOUT drops below BAT. 4.5 VISHAY Si2333 OPTIONAL EXTERNAL IDEAL DIODE LTC3586/ LTC3586-1 IDEAL DIODE ON SEMICONDUCTOR MBRM120LT3 0 60 120 180 240 300 360 420 480 FORWARD VOLTAGE (mV) (BAT – VOUT) 3586 F02 Figure 1. VOUT vs BAT Figure 2. Ideal Diode Operation 3586fa 15 LTC3586/LTC3586-1 OPERATION Suspend LDO If the LTC3586/LTC3586-1 are configured for USB suspend mode, the switching regulator is disabled and the suspend LDO provides power to the VOUT pin (presuming there is power available to VBUS). This LDO will prevent the battery from running down when the portable product has access to a suspended USB port. Regulating at 4.6V, this LDO only becomes active when the switching converter is disabled (Suspended). To remain compliant with the USB specification, the input to the LDO is current limited so that it will not exceed the 500μA low power suspend specification. If the load on VOUT exceeds the suspend current limit, the additional current will come from the battery via the ideal diode. 3.3V Always-On Supply The LTC3586/LTC3586-1 include a low quiescent current low dropout regulator that is always powered. This LDO can be used to provide power to a system pushbutton controller, standby microcontroller or real-time clock. Designed to deliver up to 20mA, the always-on LDO requires at least a 1μF low impedance ceramic bypass capacitor for compensation. The LDO is powered from VOUT, and therefore will enter dropout at loads less than 20mA as VOUT falls near 3.3V. If the LDO3V3 output is not used, it should be disabled by connecting it to VOUT. VBUS Undervoltage Lockout (UVLO) An internal undervoltage lockout circuit monitors VBUS and keeps the PowerPath switching regulator off until VBUS rises above 4.30V and is about 200mV above the battery voltage. Hysteresis on the UVLO turns off the regulator if VBUS drops below 4.00V or to within 50mV of BAT. When this happens, system power at VOUT will be drawn from the battery via the ideal diode. Battery Charger The LTC3586/LTC3586-1 include a constant-current/constant-voltage battery charger with automatic recharge, automatic termination by safety timer, low voltage trickle charging, bad cell detection and thermistor sensor input for out-of-temperature charge pausing. Battery Preconditioning When a battery charge cycle begins, the battery charger first determines if the battery is deeply discharged. If the battery voltage is below VTRKL, typically 2.85V, an automatic trickle charge feature sets the battery charge current to 10% of the programmed value. If the low voltage persists for more than 1/2 hour, the battery charger automatically terminates and indicates via the CHRG pin that the battery was unresponsive. TO USB OR WALL ADAPTER VBUS 35, 36 SW 37 3.5V TO (BAT + 0.3V) TO SYSTEM LOAD PWM AND GATE DRIVE ISWITCH/ hCLPROG CONSTANT-CURRENT CONSTANT-VOLTAGE BATTERY CHARGER 15mV 0.3V 3.6V IDEAL DIODE VOUT 34 + – + – GATE 31 OPTIONAL EXTERNAL IDEAL DIODE PMOS 4 1.188V AVERAGE INPUT CURRENT LIMIT CONTROLLER AVERAGE OUTPUT VOLTAGE LIMIT CONTROLLER 3586 F03 Figure 3. PowerPath Block Diagram 3586fa 16 + + – + – CLPROG +– BAT 32 + SINGLE CELL Li-Ion LTC3586/LTC3586-1 OPERATION Once the battery voltage is above 2.85V, the battery charger begins charging in full power constant-current mode. The current delivered to the battery will try to reach 1022V/ RPROG . Depending on available input power and external load conditions, the battery charger may or may not be able to charge at the full programmed rate. The external load will always be prioritized over the battery charge current. The USB current limit programming will always be observed and only additional power will be available to charge the battery. When system loads are light, battery charge current will be maximized. Charge Termination The battery charger has a built-in safety timer. When the voltage on the battery reaches the pre-programmed float voltage, the battery charger will regulate the battery voltage and the charge current will decrease naturally. Once the battery charger detects that the battery has reached the float voltage, the four hour safety timer is started. After the safety timer expires, charging of the battery will discontinue and no more current will be delivered. Automatic Recharge After the battery charger terminates, it will remain off drawing only microamperes of current from the battery. If the portable product remains in this state long enough, the battery will eventually self discharge. To ensure that the battery is always topped off, a charge cycle will automatically begin when the battery voltage falls below the recharge threshold which is typically 100mV less than the charger’s float voltage. In the event that the safety timer is running when the battery voltage falls below the recharge threshold, it will reset back to zero. To prevent brief excursions below the recharge threshold from resetting the safety timer, the battery voltage must be below the recharge threshold for more than 1.3ms. The charge cycle and safety timer will also restart if the VBUS UVLO cycles low and then high (e.g., VBUS is removed and then replaced). Charge Current The charge current is programmed using a single resistor from PROG to ground. 1/1022th of the battery charge current is sent to PROG which will attempt to servo to 1.000V. Thus, the battery charge current will try to reach 1022 times the current in the PROG pin. The program resistor and the charge current are calculated using the following equations: RPROG = 1022V 1022V , ICHG = ICHG RPROG In either the constant-current or constant-voltage charging modes, the voltage at the PROG pin will be proportional to the actual charge current delivered to the battery. Therefore, the actual charge current can be determined at any time by monitoring the PROG pin voltage and using the following equation: IBAT = VPROG • 1022 RPROG In many cases, the actual battery charge current, IBAT, will be lower than ICHG due to limited input power available and prioritization with the system load drawn from VOUT. Charge Status Indication The CHRG pin indicates the status of the battery charger. Four possible states are represented by CHRG which include charging, not charging, unresponsive battery, and battery temperature out of range. The signal at the CHRG pin can be easily recognized as one of the above four states by either a human or a microprocessor. An open-drain output, the CHRG pin can drive an indicator LED through a current limiting resistor for human interfacing or simply a pull-up resistor for microprocessor interfacing. 3586fa 17 LTC3586/LTC3586-1 OPERATION To make the CHRG pin easily recognized by both humans and microprocessors, the pin is either LOW for charging, HIGH for not charging, or it is switched at high frequency (35kHz) to indicate the two possible faults, unresponsive battery and battery temperature out of range. When charging begins, CHRG is pulled low and remains low for the duration of a normal charge cycle. When charging is complete, i.e., the BAT pin reaches the float voltage and the charge current has dropped to one tenth of the programmed value, the CHRG pin is released (Hi-Z). If a fault occurs, the pin is switched at 35kHz. While switching, its duty cycle is modulated between a high and low value at a very low frequency. The low and high duty cycles are disparate enough to make an LED appear to be on or off thus giving the appearance of “blinking”. Each of the two faults has its own unique “blink” rate for human recognition as well as two unique duty cycles for machine recognition. The CHRG pin does not respond to the C/10 threshold if the LTC3586/LTC3586-1 are in VBUS current limit. This prevents false end-of-charge indications due to insufficient power available to the battery charger. Table 3 illustrates the four possible states of the CHRG pin when the battery charger is active. Table 3. CHRG Signal STATUS Charging Not Charging NTC Fault Bad Battery FREQUENCY 0Hz 0Hz 35kHz 35kHz MODULATION (BLINK) FREQUENCY DUTY CYCLES 0Hz (Lo-Z) 100% 0Hz (Hi-Z) 0% 1.5Hz at 50% 6.25% to 93.75% 6.1Hz at 50% 12.5% to 87.5% Note that the LTC3586/LTC3586-1 are 3-terminal PowerPath products where system load is always prioritized over battery charging. Due to excessive system load, there may not be sufficient power to charge the battery beyond the trickle charge threshold voltage within the bad battery timeout period. In this case, the battery charger will falsely indicate a bad battery. System software may then reduce the load and reset the battery charger to try again. Although very improbable, it is possible that a duty cycle reading could be taken at the bright-dim transition (low duty cycle to high duty cycle). When this happens the duty cycle reading will be precisely 50%. If the duty cycle reading is 50%, system software should disqualify it and take a new duty cycle reading. NTC Thermistor The battery temperature is measured by placing a negative temperature coefficient (NTC) thermistor close to the battery pack. To use this feature, connect the NTC thermistor, RNTC , between the NTC pin and ground and a resistor, RNOM , from VBUS to the NTC pin. RNOM should be a 1% resistor with a value equal to the value of the chosen NTC thermistor at 25°C (R25). A 100k thermistor is recommended since thermistor current is not measured by the LTC3586/LTC3586-1 and will have to be budgeted for USB compliance. The LTC3586/LTC3586-1 will pause charging when the resistance of the NTC thermistor drops to 0.54 times the value of R25 or approximately 54k. For Vishay “Curve 1” thermistor, this corresponds to approximately 40°C. If the battery charger is in constant voltage (float) mode, the safety timer also pauses until the thermistor indicates a return to a valid temperature. As the temperature drops, the resistance of the NTC thermistor rises. The LTC3586/ LTC3586-1 are also designed to pause charging when the value of the NTC thermistor increases to 3.25 times the value of R25. For Vishay “Curve 1” this resistance, 325k, corresponds to approximately 0°C. The hot and cold comparators each have approximately 3°C of hysteresis to prevent oscillation about the trip point. Grounding the NTC pin disables the NTC charge pausing function. 3586fa An NTC fault is represented by a 35kHz pulse train whose duty cycle varies between 6.25% and 93.75% at a 1.5Hz rate. A human will easily recognize the 1.5Hz rate as a “slow” blinking which indicates the out-of-range battery temperature while a microprocessor will be able to decode either the 6.25% or 93.75% duty cycles as an NTC fault. If a battery is found to be unresponsive to charging (i.e., its voltage remains below 2.85V for 1/2 hour), the CHRG pin gives the battery fault indication. For this fault, a human would easily recognize the frantic 6.1Hz “fast” blink of the LED while a microprocessor would be able to decode either the 12.5% or 87.5% duty cycles as a bad battery fault. 18 LTC3586/LTC3586-1 OPERATION Thermal Regulation To optimize charging time, an internal thermal feedback loop may automatically decrease the programmed charge current. This will occur if the die temperature rises to approximately 110°C. Thermal regulation protects the LTC3586/LTC3586-1 from excessive temperature due to high power operation or high ambient thermal conditions and allows the user to push the limits of the power handling capability with a given circuit board design without risk of damaging the LTC3586/LTC3586-1 or external components. The benefit of the LTC3586/LTC3586-1 thermal regulation loop is that charge current can be set according to actual conditions rather than worst-case conditions with the assurance that the battery charger will automatically reduce the current in worst-case conditions. A flow chart of battery charger operation can be seen in Figure 4. Low Supply Operation The LTC3586/LTC3586-1 incorporate an undervoltage lockout circuit on VOUT which shuts down all four general purpose switching regulators when VOUT drops below VOUTUVLO. This UVLO prevents unstable operation. FAULT Pin FAULT is a bi-directional pin with an open-drain output used to indicate a fault condition on any of the general purpose regulators. If any of the four regulators are enabled, and their corresponding FB pin voltage does not rise to within 8% of the internal reference voltage (0.8V) within 14ms, a fault condition will be reported by FAULT going low. This, in turn, will disable all of the regulators. Alternatively, the regulators can be all disabled simultaneously by driving FAULT low externally. This fault condition can be cleared only if all of the ENABLE inputs are pulled low for at least 3.6μs. Since FAULT is an open-drain output, it requires a pull-up resistor to the input voltage of the monitoring microprocessor or another appropriate power source such as LD03V3. If any of the ENABLE pins is tied high during start-up, the FAULT pin can erroneously report a fault condition. To avoid such an event, the ENABLE pins should be tied high through a lowpass filter (comprised of a 1k resistor and a 0.1μF capacitor) to the same power source to which FAULT pin is pulled up. General Purpose Buck Switching Regulators The LTC3586/LTC3586-1 contain two 2.25MHz constantfrequency current mode buck switching regulators. Each buck regulator can provide up to 400mA of output current. Both buck regulators can be programmed for a minimum output voltage of 0.8V and can be used to power a microcontroller core, microcontroller I/O, memory, disk drive or other logic circuitry. Both buck converters support 100% duty cycle operation (low dropout mode) when their input voltage drops very close to their output voltage. To suit a variety of applications, selectable mode functions can be used to trade-off noise for efficiency. Two modes are available to control the operation of the LTC3586/LTC3586-1’s buck regulators. At moderate to heavy loads, the pulseskip mode provides the least noise switching solution. At lighter loads, Burst Mode operation may be selected. The buck regulators include soft-start to limit inrush current when powering on, short-circuit current protection and switch node slew limiting circuitry to reduce radiated EMI. No external compensation components are required. The operating mode of the buck regulators can be set by the MODE pin. The buck converters can be individually enabled by the EN1 and EN2 pins. Both buck regulators have a fixed feedback servo voltage of 800mV. The buck regulator input supplies VIN1 and VIN2 will generally be connected to the system load pin VOUT. Buck Regulator Output Voltage Programming Both buck regulators can be programmed for output voltages greater than 0.8V. The output voltage for each buck regulator is programmed using a resistor divider from the buck regulator output connected to the feedback pins (FB1 and FB2) such that: ⎛ R1 ⎞ VOUTX = VFBX ⎜ + 1⎟ ⎝ R2 ⎠ where VFB is fixed at 0.8V and X = 1, 2. See Figure 4. Typical values for R1 are in the range of 40k to 1M. The capacitor CFB cancels the pole created by feedback resistors 3586fa 19 LTC3586/LTC3586-1 OPERATION POWER ON CLEAR EVENT TIMER ASSERT CHRG LOW NTC OUT OF RANGE YES INHIBIT CHARGER NO BAT < 2.85V BATTERY STATE BAT > 4.15V CHRG CURRENTLY HIGH-Z YES 2.85V < BAT < 4.15V CHARGE AT 100V/RPROG (C/10 RATE) CHARGE AT 1022V/RPROG RATE CHARGE WITH FIXED VOLTAGE (4.200V) NO INDICATE NTC FAULT AT CHRG RUN EVENT TIMER PAUSE EVENT TIMER RUN EVENT TIMER NO TIMER > 30 MINUTES TIMER > 4 HOURS NO YES YES NO INHIBIT CHARGING STOP CHARGING IBAT < C/10 YES INDICATE BATTERY FAULT AT CHRG BAT RISING THROUGH 4.1V YES RELEASE CHRG HIGH-Z RELEASE CHRG HIGH-Z NO NO BAT > 2.85V BAT FALLING THROUGH 4.1V YES NO BAT < 4.1V YES NO YES 3586 F04 Figure 4. Flow Chart for Battery Charger Operation (LTC3586) 3586fa 20 LTC3586/LTC3586-1 OPERATION VINx SWx LTC3586/ LTC3586-1 FBx L VOUTx CFB R1 COUT X = 1, 2 R2 GND 3586 F05 At high duty cycles (VOUTx > VINx /2) it is possible for the inductor current to reverse, causing the buck regulator to operate continuously at light loads. This is normal and regulation is maintained, but the supply current will increase to several milliamperes due to continuous switching. In Burst Mode operation, the buck regulator automatically switches between fixed frequency PWM operation and hysteretic control as a function of the load current. At light loads, the buck regulators operate in hysteretic mode in which the output capacitor is charged to a voltage slightly higher than the regulation point. The buck converter then goes into sleep mode, during which the output capacitor provides the load current. In sleep mode, most of the regulator’s circuitry is powered down, helping conserve battery power. When the output voltage drops below a predetermined value, the buck regulator circuitry is powered on and the normal PWM operation resumes. The duration for which the buck regulator operates in sleep mode depends on the load current. The sleep time decreases as the load current increases. Beyond a certain load current point (about 1/4 rated output load current) the step-down switching regulators will switch to a low noise constant frequency PWM mode of operation, much the same as pulse-skip operation at high loads. For applications that can tolerate some output ripple at low output currents, Burst Mode operation provides better efficiency than pulse skip at light loads while still providing the full specified output current of the buck regulator. The buck regulators allow mode transition on the fly, providing seamless transition between modes even under load. This allows the user to switch back and forth between modes to reduce output ripple or increase low current efficiency as needed. Buck Regulator in Shutdown The buck regulators are in shutdown when not enabled for operation. In shutdown, all circuitry in the buck regulator is disconnected from the buck regulator input supply leaving only a few nanoamps of leakage current. The buck regulator outputs are individually pulled to ground through a 10k resistor on the switch pins (SW1 and SW2) when in shutdown. 3586fa Figure 5. Buck Converter Application Circuit and the input capacitance of the FBx pin and also helps to improve transient response for output voltages much greater than 0.8V. A variety of capacitor sizes can be used for CFB but a value of 10pF is recommended for most applications. Experimentation with capacitor sizes between 2pF and 22pF may yield improved transient response. Buck Regulator Operating Modes The LTC3586/LTC3586-1’s buck regulators include two possible operating modes to meet the noise/ power needs of a variety of applications. In pulse-skip mode, an internal latch is set at the start of every cycle which turns on the main P-channel MOSFET switch. During each cycle, a current comparator compares the peak inductor current to the output of an error amplifier. The output of the current comparator resets the internal latch which causes the main P-channel MOSFET switch to turn off and the N-channel MOSFET synchronous rectifier to turn on. The N-channel MOSFET synchronous rectifier turns off at the end of the 2.25MHz cycle or if the current through the N-channel MOSFET synchronous rectifier drops to zero. Using this method of operation, the error amplifier adjusts the peak inductor current to deliver the required output power. All necessary compensation is internal to the switching regulator requiring only a single ceramic output capacitor for stability. At light loads, the inductor current may reach zero on each pulse which will turn off the N-channel MOSFET synchronous rectifier. In this case, the switch node (SW1, SW2) goes high impedance and the switch node voltage will “ring”. This is discontinuous mode operation, and is normal behavior for a switching regulator. At very light loads, the buck regulators will automatically skip pulses as needed to maintain output regulation. 21 LTC3586/LTC3586-1 OPERATION Buck Regulator Dropout Operation It is possible for a buck regulator’s input voltage, VINx , to approach its programmed output voltage (e.g., a battery voltage of 3.4V with a programmed output voltage of 3.3V). When this happens, the PMOS switch duty cycle increases until it is turned on continuously at 100%. In this dropout condition, the respective output voltage equals the buck regulator’s input voltage minus the voltage drops across the internal P-channel MOSFET and the inductor. Buck Regulator Soft-Start Operation Soft-start is accomplished by gradually increasing the peak inductor current for each buck regulator over a 500μs period. This allows each output to rise slowly, helping minimize the battery in-rush current. A soft-start cycle occurs whenever a given buck regulator is enabled, or after a fault condition has occurred (thermal shutdown or UVLO). A soft-start cycle is not triggered by changing operating modes. This allows seamless output operation when transitioning between modes. Buck Regulator Switching Slew Rate Control The buck regulators contain new patent pending circuitry to limit the slew rate of the switch node (SW1 and SW2). This new circuitry is designed to transition the switch node over a period of a couple of nanoseconds, significantly reducing radiated EMI and conducted supply noise. BUCK-BOOST DC/DC SWITCHING REGULATOR The LTC3586/LTC3586-1 contain a 2.25MHz constantfrequency voltage-mode buck-boost switching regulator. The regulator provides up to 1A of output load current. The buck-boost can be programmed to a minimum output voltage of 2.5V and can be used to power a microcontroller core, microcontroller I/O, memory, disk drive, or other logic circuitry. The converter is enabled by pulling EN3 high. To suit a variety of applications, a selectable mode function allows the user to trade-off noise for efficiency. Two modes are available to control the operation of the LTC3586/LTC3586-1’s buck-boost regulator. At moderate to heavy loads, the constant frequency PWM mode provides the least noise switching solution. At lighter loads Burst Mode operation may be selected. The output voltage is programmed by a user-supplied resistive divider returned to FB3. An error amplifier compares the divided output voltage with a reference and adjusts the compensation voltage accordingly until the FB3 pin has stabilized to the reference voltage (0.8V). The buck-boost regulator includes a soft-start to limit inrush current and voltage overshoot when powering on, short-circuit current protection, and switch node slew limiting circuitry for reduced radiated EMI. Input Current Limit The input current limit comparator will shut the input PMOS switch off once current exceeds 2.5A (typical). The 2.5A input current limit also protects against a grounded VOUT3 node. Output Overvoltage Protection If the FB3 node were inadvertently shorted to ground, then the output would increase indefinitely with the maximum current that could be sourced from VIN3 . The LTC3586/ LTC3586-1 protect against this by shutting off the input PMOS if the output voltage exceeds 5.6V (typical). Low Output Voltage Operation When the output voltage is below 2.65V (typical) during start-up, Burst Mode operation is disabled and switch D is turned off (allowing forward current through the well diode and limiting reverse current to 0mA). Buck-Boost Regulator PWM Operating Mode In PWM mode the voltage seen at FB3 is compared to the reference voltage (0.8V). From the FB3 voltage an error amplifier generates an error signal seen at VC3 . This error signal commands PWM waveforms that modulate switches A, B, C, and D. Switches A and B operate synchronously as do switches C and D. If VIN3 is significantly greater than the programmed VOUT3 , then the converter will operate in buck mode. In this case switches A and B will be modulated, with switch D always on (and switch C always off), to step-down the input voltage to the programmed output. If VIN3 is significantly less than the programmed VOUT3 , then the converter will operate in boost mode. In this case switches C and D are modulated, with switch A 3586fa 22 LTC3586/LTC3586-1 OPERATION always on (and switch B always off), to step-up the input voltage to the programmed output. If VIN3 is close to the programmed VOUT3 , then the converter will operate in 4-switch mode. In this case the switches sequence through the pattern of AD, AC, BD to either step the input voltage up or down to the programmed output. Buck-Boost Regulator Burst-Mode Operation In Burst Mode operation, the buck-boost regulator uses a hysteretic FB3 voltage algorithm to control the output voltage. By limiting FET switching and using a hysteretic control loop, switching losses are greatly reduced. In this mode output current is limited to 50mA typical. While operating in Burst Mode operation, the output capacitor is charged to a voltage slightly higher than the regulation point. The buck-boost converter then goes into a sleep state, during which the output capacitor provides the load current. The output capacitor is charged by charging the inductor until the input current reaches 250mA typical and then discharging the inductor until the reverse current reaches 0mA typical. This process is repeated until the feedback voltage has charged to 6mV above the regulation point. In the sleep state, most of the regulator’s circuitry is powered down, helping to conserve battery power. When the feedback voltage drops 6mV below the regulation point, the switching regulator circuitry is powered on and another burst cycle begins. The duration for which the regulator sleeps depends on the load current and output capacitor value. The sleep time decreases as the load current increases. The buck-boost regulator will not go to sleep if the current is greater than 50mA, and if the load current increases beyond this point while in Burst Mode operation the output will lose regulation. Burst Mode operation provides a significant improvement in efficiency at light loads at the expense of higher output ripple when compared to PWM mode. For many noise-sensitive systems, Burst Mode operation might be undesirable at certain times (i.e. during a transmit or receive cycle of a wireless device), but highly desirable at others (i.e. when the device is in low power standby mode). The MODE pin is used to enable or disable Burst Mode operation at any time, offering both low noise and low power operation when they are needed. Buck-Boost Regulator Soft-Start Operation Soft-start is accomplished by gradually increasing the maximum VC3 voltage over a 0.5ms (typical) period. Ramping the VC3 voltage limits the duty cycle and thus the VOUT3 voltage minimizing output overshoot during startup. A soft-start cycle occurs whenever the buck-boost is enabled, or after a fault condition has occurred (thermal shutdown or UVLO). A soft-start cycle is not triggered by changing operating modes. This allows seamless output operation when transitioning between Burst Mode operation and PWM mode. SYNCHRONOUS BOOST DC/DC SWITCHING REGULATOR The LTC3586/LTC3586-1 contain a 2.25MHz constantfrequency current mode synchronous boost switching regulator with true output disconnect feature. The regulator provides at least 800mA of output load current and the output voltage can be programmed up to a maximum of 5V. The converter is enabled by pulling EN4 high. The boost regulator also includes soft-start to limit inrush current and voltage overshoot when powering on, short circuit current protection and switch node slew limiting circuitry for reduced radiated EMI. Error Amp The boost output voltage is programmed by a user-supplied resistive divider returned to the FB4 pin. An internally compensated error amplifier compares the divided output voltage with an internal 0.8V reference and adjusts the voltage accordingly until FB4 servos to 0.8V. Current Limit Lossless current sensing converts the NMOS switch current signal to a voltage to be summed with the internal slope compensation signal. The summed signal is then compared to the error amplifier output to provide a peak current control command for the peak comparator. Peak switch current is limited to 2.4A independent of output voltage. 3586fa 23 LTC3586/LTC3586-1 OPERATION Zero Current Comparator The zero current comparator monitors the inductor current to the output and shuts off the synchronous rectifier once the current drops to approximately 65mA. This prevents the inductor current from reversing in polarity thereby improving efficiency at light loads. Antiringing Control The antiringing control circuitry prevents high frequency ringing of the SW pin as the inductor current goes to zero in discontinuous mode. The damping of the resonant circuit formed by L and CSW (capacitance of the SW4 pin) is achieved internally by switching a 150Ω resistor across the inductor. PMOS Synchronous Rectifier To prevent the inductor current from running away, the PMOS synchronous rectifier is only enabled when VOUT > (VIN + 130mV). Output Disconnect and Inrush Limiting The LTC3586/LTC3586-1 boost converter is designed to allow true output disconnect by eliminating body diode conduction of the internal PMOS rectifier. This allows VOUT to go to zero volts during shutdown, drawing zero current from the input source. It also allows for inrush current limiting at start-up, minimizing surge currents seen by the input supply. Note that to obtain the advantage of output disconnect, there must not be an external Schottky diode connected between the SW4 and VOUT4 pin. Short Circuit Protection Unlike most boost converters, the LTC3586/LTC3586-1 boost converter allows its output to be short-circuited due to the output disconnect feature. It incorporates internal features such as current limit foldback and thermal shutdown for protection from an excessive overload or short circuit. VIN > VOUT Operation The LTC3586/LTC3586-1 boost converter will maintain voltage regulation even if the input voltage is above the output voltage. This is achieved by terminating the switching of the synchronous PMOS and applying VIN4 statically on its gate. This ensures that the slope of the inductor current will reverse during the time when current is flowing to the output. Since the PMOS no longer acts as a low impedance switch in this mode, there will be more power dissipation within the IC. This will cause a sharp drop in the efficiency (see Typical Performance Characteristics, Boost Efficiency vs VIN4). The maximum output current should be limited in order to maintain an acceptable junction temperature. Boost Soft-Start The LTC3586/LTC3586-1 boost converter provides softstart by slowly ramping the peak inductor current from zero to a maximum of 2.4A in about 500μs. Ramping the peak inductor current limits transient inrush currents during start-up. A soft-start cycle occurs whenever the boost is enabled, or after a fault condition has occurred (thermal shutdown or UVLO). Boost Overvoltage Protection If the FB4 node were inadvertently shorted to ground, then the boost converter output would increase indefinitely with the maximum current that could be sourced from VIN4 . The LTC3586/LTC3586-1 protects against this by shutting off the main switch if the output voltage exceeds 5.3V. 3586fa 24 LTC3586/LTC3586-1 APPLICATIONS INFORMATION PowerPath CONTROLLER APPLICATIONS SECTION CLPROG Resistor and Capacitor As described in the High Efficiency Switching PowerPath Controller section, the resistor on the CLPROG pin determines the average input current limit when the switching regulator is set to either the 1x mode (USB 100mA), the 5x mode (USB 500mA) or the 10x mode. The input current will be comprised of two components, the current that is used to drive VOUT and the quiescent current of the switching regulator. To ensure that the USB specification is strictly met, both components of input current should be considered. The Electrical Characteristics table gives the worst-case values for quiescent currents in either setting as well as current limit programming accuracy. To get as close to the 500mA or 100mA specifications as possible, a 1% resistor should be used. Recall that IVBUS = IVBUSQ + VCLPROG/RCLPPROG • (hCLPROG +1). An averaging capacitor is required in parallel with the CLPROG resistor so that the switching regulator can determine the average input current. This network also provides the dominant pole for the feedback loop when current limit is reached. To ensure stability, the capacitor on CLPROG should be 0.1μF . Choosing the PowerPath Inductor Because the input voltage range and output voltage range of the power path switching regulator are both fairly narrow, the LTC3586/LTC3586-1 are designed for a specific inductance value of 3.3μH. Some inductors which may be suitable for this application are listed in Table 4. Table 4. Recommended Inductors for PowerPath Controller INDUCTOR L TYPE (μH) LPS4018 D53LC DB318C WE-TPC Type M1 CDRH6D12 CDRH6D38 3.3 3.3 3.3 3.3 3.3 3.3 MAX IDC (A) 2.2 MAX DCR (Ω) 0.08 SIZE IN mm (L × W × H) VBUS and VOUT Bypass Capacitors The style and value of capacitors used with the LTC3586/ LTC3586-1 determine several important parameters such as regulator control-loop stability and input voltage ripple. Because the LTC3586/LTC3586-1 use a buck switching power supply from VBUS to VOUT, its input current waveform contains high frequency components. It is strongly recommended that a low equivalent series resistance (ESR) multilayer ceramic capacitor be used to bypass VBUS . Tantalum and aluminum capacitors are not recommended because of their high ESR. The value of the capacitor on VBUS directly controls the amount of input ripple for a given load current. Increasing the size of this capacitor will reduce the input ripple. To prevent large VOUT voltage steps during transient load conditions, it is also recommended that a ceramic capacitor be used to bypass VOUT. The output capacitor is used in the compensation of the switching regulator. At least 4μF of actual capacitance with low ESR are required on VOUT. Additional capacitance will improve load transient performance and stability. Multilayer ceramic chip capacitors typically have exceptional ESR performance. MLCCs combined with a tight board layout and an unbroken ground plane will yield very good performance and low EMI emissions. There are several types of ceramic capacitors available each having considerably different characteristics. For example, X7R ceramic capacitors have the best voltage and temperature stability. X5R ceramic capacitors have apparently higher packing density but poorer performance over their rated voltage and temperature ranges. Y5V ceramic capacitors have the highest packing density, but must be used with caution, because of their extreme non-linear characteristic of capacitance verse voltage. The actual in-circuit capacitance of a ceramic capacitor should be measured with a small AC signal as is expected in-circuit. Many vendors specify the capacitance verse voltage with a 1V RMS AC test signal and as a result overstate the capacitance that the capacitor will present in the application. Using similar operating conditions as the application, the user must measure or request from the vendor the actual capacitance to determine if the selected capacitor meets the minimum capacitance that the application requires. 3586fa MANUFACTURER 3.9 × 3.9 × 1.7 Coilcraft www.coilcraft.com 2.26 0.034 Toko 5×5×3 1.55 0.070 3.8 × 3.8 × 1.8 www.toko.com 1.95 0.065 4.8 × 4.8 × 1.8 Wurth Elektronik www.we-online.com 2.2 0.0625 6.7 × 6.7 × 1.5 Sumida 3.5 0.020 www.sumida.com 7×7×4 25 LTC3586/LTC3586-1 APPLICATIONS INFORMATION Over-Programming the Battery Charger The USB high power specification allows for up to 2.5W to be drawn from the USB port (5V • 500mA). The PowerPath switching regulator transforms the voltage at VBUS to just above the voltage at BAT with high efficiency, while limiting power to less than the amount programmed at CLPROG. In some cases the battery charger may be programmed (with the PROG pin) to deliver the maximum safe charging current without regard to the USB specifications. If there is insufficient current available to charge the battery at the programmed rate, the PowerPath regulator will reduce charge current until the system load on VOUT is satisfied and the VBUS current limit is satisfied. Programming the battery charger for more current than is available will not cause the average input current limit to be violated. It will merely allow the battery charger to make use of all available power to charge the battery as quickly as possible, and with minimal power dissipation within the battery charger. Alternate NTC Thermistors and Biasing The LTC3586/LTC3586-1 provide temperature qualified charging if a grounded thermistor and a bias resistor are connected to NTC. By using a bias resistor whose value is equal to the room temperature resistance of the thermistor (R25) the upper and lower temperatures are pre-programmed to approximately 40°C and 0°C, respectively (assuming a Vishay “Curve 1” thermistor). The upper and lower temperature thresholds can be adjusted by either a modification of the bias resistor value or by adding a second adjustment resistor to the circuit. If only the bias resistor is adjusted, then either the upper or the lower threshold can be modified but not both. The other trip point will be determined by the characteristics of the thermistor. Using the bias resistor in addition to an adjustment resistor, both the upper and the lower temperature trip points can be independently programmed with the constraint that the difference between the upper and lower temperature thresholds cannot decrease. Examples of each technique are given below. NTC thermistors have temperature characteristics which are indicated on resistance-temperature conversion tables. The Vishay-Dale thermistor NTHS0603N011-N1003F, used in the following examples, has a nominal value of 100k and follows the Vishay “Curve 1” resistance-temperature characteristic. In the explanation below, the following notation is used. R25 = Value of the Thermistor at 25°C RNTC|COLD = Value of thermistor at the cold trip point RNTC|HOT = Value of the thermistor at the hot trip point rCOLD = Ratio of RNTC|COLD to R25 rHOT = Ratio of RNTC|COLD to R25 RNOM = Primary thermistor bias resistor (see Figure 6a) R1 = Optional temperature range adjustment resistor (see Figure 6b) The trip points for the LTC3586/LTC3586-1’s temperature qualification are internally programmed at 0.349 • VBUS for the hot threshold and 0.765 • VBUS for the cold threshold. Therefore, the hot trip point is set when: RNTCHOT | RNOM + RNTCHOT | • VBUS = 0.349 • VBUS and the cold trip point is set when: RNTC|COLD RNOM + RNTC|COLD • VBUS = 0.765 • VBUS Solving these equations for RNTC|COLD and RNTC|HOT results in the following: RNTC|HOT = 0.536 • RNOM and RNTC|COLD = 3.25 • RNOM By setting RNOM equal to R25, the above equations result in rHOT = 0.536 and rCOLD = 3.25. Referencing these ratios to the Vishay Resistance-Temperature Curve 1 chart gives a hot trip point of about 40°C and a cold trip point of about 0°C. The difference between the hot and cold trip points is approximately 40°C. 3586fa 26 LTC3586/LTC3586-1 APPLICATIONS INFORMATION By using a bias resistor, RNOM , different in value from R25, the hot and cold trip points can be moved in either direction. The temperature span will change somewhat due to the non-linear behavior of the thermistor. The following equations can be used to easily calculate a new value for the bias resistor: RNOM = RNOM = rHOT • R25 0.536 rCOLD • R25 3.25 “temperature gain” of the thermistor as absolute temperature increases. The upper and lower temperature trip points can be independently programmed by using an additional bias resistor as shown in Figure 6b. The following formulas can be used to compute the values of RNOM and R1: RNOM = rCOLD – rHOT • R25 2.714 R1 = 0.536 • RNOM – rHOT • R25 For example, to set the trip points to 0°C and 45°C with a Vishay Curve 1 thermistor choose: RNOM = 3.266 – 0.4368 • 100k = 104.2k 2.714 where rHOT and rCOLD are the resistance ratios at the desired hot and cold trip points. Note that these equations are linked. Therefore, only one of the two trip points can be chosen, the other is determined by the default ratios designed in the IC. Consider an example where a 60°C hot trip point is desired. From the Vishay Curve 1 R-T characteristics, rHOT is 0.2488 at 60°C. Using the above equation, RNOM should be set to 46.4k. With this value of RNOM , the cold trip point is about 16°C. Notice that the span is now 44°C rather than the previous 40°C. This is due to the decrease in VBUS VBUS LTC3586/LTC3586-1 NTC BLOCK the nearest 1% value is 105k: R1 = 0.536 • 105k – 0.4368 • 100k = 12.6k the nearest 1% value is 12.7k. The final circuit is shown in Figure 6b and results in an upper trip point of 45°C and a lower trip point of 0°C. VBUS VBUS LTC3586/LTC3586-1 NTC BLOCK RNOM 100k NTC 5 RNTC 100k 0.765 • VBUS – TOO_COLD RNOM 105k NTC 5 0.765 • VBUS – TOO_COLD + + T – TOO_HOT 0.349 • VBUS R1 12.7k 0.349 • VBUS T RNTC 100k – TOO_HOT + + + NTC_ENABLE 0.017 • VBUS + NTC_ENABLE 0.017 • VBUS – 3586 F06a – 3586 F06b (6a) Figure 6. NTC Circuits (6b) 3586fa 27 LTC3586/LTC3586-1 APPLICATIONS INFORMATION USB Inrush Limiting When a USB cable is plugged into a portable product, the inductance of the cable and the high-Q ceramic input capacitor form an L-C resonant circuit. If the cable does not have adequate mutual coupling or if there is not much impedance in the cable, it is possible for the voltage at the input of the product to reach as high as twice the USB voltage (~10V) before it settles out. In fact, due to the high voltage coefficient of many ceramic capacitors, a nonlinearity, the voltage may even exceed twice the USB voltage. To prevent excessive voltage from damaging the LTC3586/LTC3586-1 during a hot insertion, it is best to have a low voltage coefficient capacitor at the VBUS pin to the LTC3586/LTC3586-1. This is achievable by selecting an MLCC capacitor that has a higher voltage rating than that required for the application. For example, a 16V, X5R, 10μF capacitor in a 1206 case would be a better choice than a 6.3V, X5R, 10μF capacitor in a smaller 0805 case. Alternatively, the soft connect circuit (Figure 7) can be employed. In this circuit, capacitor C1 holds MP1 off when the cable is first connected. Eventually C1 begins to charge up to the USB input voltage applying increasing gate support to MP1. The long time constant of R1 and C1 prevent the current from building up in the cable too fast thus dampening out any resonant overshoot. Battery Charger Stability Considerations The LTC3586/LTC3586-1’s battery charger contains both a constantvoltage and a constant-current control loop. The constantvoltage loop is stable without any compensation when a battery is connected with low impedance leads. Excessive lead length, however, may add enough series inductance to require a bypass capacitor of at least 1μF from BAT to GND. Furthermore, when the battery is MP1 Si2333 5V USB INPUT C1 100nF USB CABLE R1 40k C2 10μF VBUS LTC3586/ LTC3586-1 disconnected, a 4.7μF capacitor in series with a 0.2Ω to 1Ω resistor from BAT to GND is required to keep ripple voltage low. High value, low ESR multilayer ceramic chip capacitors reduce the constant-voltage loop phase margin, possibly resulting in instability. Ceramic capacitors up to 22μF may be used in parallel with a battery, but larger ceramics should be decoupled with 0.2Ω to 1Ω of series resistance. In constant-current mode, the PROG pin is in the feedback loop rather than the battery voltage. Because of the additional pole created by any PROG pin capacitance, capacitance on this pin must be kept to a minimum. With no additional capacitance on the PROG pin, the battery charger is stable with program resistor values as high as 25k. However, additional capacitance on this node reduces the maximum allowed program resistor. The pole frequency at the PROG pin should be kept above 100kHz. Therefore, if the PROG pin has a parasitic capacitance, CPROG, the following equation should be used to calculate the maximum resistance value for RPROG: RPROG ≤ 1 2π • 100kHz • CPROG BUCK REGULATOR APPLICATIONS SECTION Buck Regulator Inductor Selection Many different sizes and shapes of inductors are available from numerous manufacturers. Choosing the right inductor from such a large selection of devices can be overwhelming, but following a few basic guidelines will make the selection process much simpler. The buck converters are designed to work with inductors in the range of 2.2μH to 10μH. For most applications a 4.7μH inductor is suggested for both buck regulators. Larger value inductors reduce ripple current which improves output ripple voltage. Lower value inductors result in higher ripple current and improved transient response time. To maximize efficiency, choose an inductor with a low DC resistance. For a 1.2V output, efficiency is reduced about 2% for 100mΩ series resistance at 400mA load current, and about 2% for 300mΩ series resistance at 100mA 3586fa GND 3586 F07 Figure 7. USB Soft Connect Circuit 28 LTC3586/LTC3586-1 APPLICATIONS INFORMATION load current. Choose an inductor with a DC current rating at least 1.5 times larger than the maximum load current to ensure that the inductor does not saturate during normal operation. If output short circuit is a possible condition, the inductor should be rated to handle the maximum peak current specified for the buck converters. Different core materials and shapes will change the size/current and price/current relationship of an inductor. Toroid or shielded pot cores in ferrite or Permalloy materials are small and don’t radiate much energy, but generally cost more than powdered iron core inductors with similar electrical characteristics. Inductors that are very thin or have a very small volume typically have much higher core and DCR losses, and will not give the best efficiency. The choice of which style inductor to use often depends more on the price vs size, performance and any radiated EMI requirements than on what the LTC3586/LTC3586-1 require to operate. The inductor value also has an effect on Burst Mode operations. Lower inductor values will cause the Burst Mode switching frequencies to increase. Table 5 shows several inductors that work well with the LTC3586/LTC3586-1’s buck regulators. These inductors offer a good compromise in current rating, DCR and physical size. Consult each manufacturer for detailed information on their entire selection of inductors. Table 5. Recommended Inductors for Buck Regulators MAX INDUCTOR L IDC TYPE (μH) (A) DE2818C DE2812C CDRH3D16 SD3118 SD3112 LPS3015 *Typical DCR 4.7 4.7 4.7 4.7 4.7 4.7 MAX DCR (Ω) SIZE IN mm (L × W × H) Buck Regulator Input/Output Capacitor Selection Low ESR (equivalent series resistance) MLCC capacitors should be used at both buck regulator outputs as well as at each buck regulator input supply (VIN1 and VIN2). Only X5R or X7R ceramic capacitors should be used because they retain their capacitance over wider voltage and temperature ranges than other ceramic types. A 10μF output capacitor is sufficient for most applications. For good transient response and stability the output capacitor should retain at least 4μF of capacitance over operating temperature and bias voltage. Each buck regulator input supply should be bypassed with a 1μF capacitor. Consult with capacitor manufacturers for detailed information on their selection and specifications of ceramic capacitors. 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