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RT2853A/B
3A, 18V, 650kHz, ACOT® Synchronous Step-Down Converter
General Description
Features
The RT2853A/B are high-performance 650kHz 3A stepdown regulators with internal power switches and
synchronous rectifiers. They feature quick transient
response using their Advanced Constant On-Time
(ACOT®) control architecture that provides stable operation
with small ceramic output capacitors and without
complicated external compensation, among other benefits.
The input voltage range is from 4.5V to 18V and the output
is adjustable from 0.765V to 7V.
The proprietary ACOT®control improves upon other fastresponse constant on-time architectures, achieving nearly
constant switching frequency over line, load, and output
voltage ranges. Since there is no internal clock, response
to transients is nearly instantaneous and inductor current
can ramp quickly to maintain output regulation without
large bulk output capacitance. The RT2853A/B are stable
with and optimized for ceramic output capacitors.
With internal 110mΩ switches and 30mΩ synchronous
rectifiers, the RT2853A/B display excellent efficiency and
good behavior across a range of applications, especially
for low output voltages and low duty cycles. Cycle-bycycle current limit, input under-voltage lock-out, externallyadjustable soft-start, output under- and over-voltage
protection, and thermal shutdown provide safe and smooth
operation in all operating conditions.
Fast Transient Response
Steady 650kHz Switching Frequency
Enhanced Efficiency at Light Load (RT2853A)
Advanced Constant On-Time (ACOT®) Control
Optimized for Ceramic Output Capacitors
4.5V to 18V Input Voltage Range
Internal 110mΩ
Ω Switch and 30mΩ
Ω Synchronous
Rectifier
0.765V to 7V Adjustable Output Voltage
Externally-adjustable, Pre-biased Compatible SoftStart
Cycle-by-Cycle Current Limit
Optional Output Discharge Function
Output Over- and Under-voltage Shut-down
Latched (RT2853ALGQW/RT2853BLGQW Only)
With Hiccup Mode (RT2853AHGQW/
RT2853BHGQW Only)
Applications
Industrial and Commercial Low Power Systems
Computer Peripherals
LCD Monitors and TVs
Point of Load Regulation for High-Performance DSPs,
FPGAs, and ASICs
Fast-Transient Response
The RT2853A and RT2853B are each available in
WQFN-16L 3x3 package, with exposed thermal pads.
VOUT
(50mV/Div)
Simplified Application Circuit
VIN
RT2853A/B
VIN
SW
VCC
Power Good
VREG5
Input Signal
PGOOD
RT2853B
VOUT
BOOT
VS
FB
EN
SS
VREG5
GND PGND
IOUT
(1A/Div)
VIN = 12V, VOUT = 1.05V, IOUT = 0 to 3A
Time (100μs/Div)
Copyright © 2021 Richtek Technology Corporation. All rights reserved.
DS2853A/B-04 March 2021
is a registered trademark of Richtek Technology Corporation.
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RT2853A/B
Ordering Information
Pin Configurations
RT2853A/B
(TOP VIEW)
VS
VCC
VIN
VIN
Package Type
QW : WQFN-16L 3x3 (W-Type)
16 15 14 13
Lead Plating System
G : Green (Halogen Free and Pb Free)
FB
VREG5
SS
GND
H : Hiccup Mode OVP & UVP
L : Latched OVP & UVP
12
2
11
GND
3
10
17
4
5
6
7
9
BOOT
SW
SW
SW
8
PGOOD
EN
PGND
PGND
A : Enhanced Light Load Efficiency
B : Continuous Switching Mode
Note :
Richtek products are :
1
WQFN-16L 3x3
RoHS compliant and compatible with the current requirements of IPC/JEDEC J-STD-020.
Suitable for use in SnPb or Pb-free soldering processes.
Marking Information
RT2853AHGQW
RT2853BHGQW
2K= : Product Code
2K=YM
DNN
YMDNN : Date Code
RT2853ALGQW
2H= : Product Code
2H=YM
DNN
RT2853BLGQW
2J= : Product Code
2J=YM
DNN
YMDNN : Date Code
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YMDNN : Date Code
2G= : Product Code
2G=YM
DNN
YMDNN : Date Code
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RT2853A/B
Absolute Maximum Ratings
(Note 1)
Supply Voltage, VIN, VCC --------------------------------------------------------------------------------------- −0.3V to 21V
Switch Voltage, SW ----------------------------------------------------------------------------------------------- −0.8V to (VVIN + 0.3V)
< 10ns ---------------------------------------------------------------------------------------------------------------- −5V to 25V
BOOT to SW -------------------------------------------------------------------------------------------------------- −0.3V to 6V
VREG5 to VIN or VCC -------------------------------------------------------------------------------------------- −18V to 0.3V
Other Pins Voltage ------------------------------------------------------------------------------------------------- −0.3V to 21V
Power Dissipation, PD @ TA = 25°C
WQFN-16L 3x3 ----------------------------------------------------------------------------------------------------- 2.1W
Package Thermal Resistance (Note 2)
WQFN-16L 3x3, θJA ------------------------------------------------------------------------------------------------ 47.4°C/W
WQFN-16L 3x3, θJC ----------------------------------------------------------------------------------------------- 7.5°C/W
Junction Temperature Range ------------------------------------------------------------------------------------- 150°C
Lead Temperature (Soldering, 10 sec.) ------------------------------------------------------------------------ 260°C
Storage Temperature Range ------------------------------------------------------------------------------------- −65°C to 150°C
ESD Susceptibility (Note 3)
HBM (Human Body Model) --------------------------------------------------------------------------------------- 2kV
Recommended Operating Conditions
(Note 4)
Supply Voltage, VIN ----------------------------------------------------------------------------------------------- 4.5V to 18V
Junction Temperature Range ------------------------------------------------------------------------------------- −40°C to 125°C
Ambient Temperature Range ------------------------------------------------------------------------------------- −40°C to 85°C
Electrical Characteristics
(VIN = 12V, TA = −40°C to 85°C, unless otherwise specified)
Parameter
Symbol
Test Conditions
Min
Typ
Max
Unit
Supply Current
Shutdown Current
ISHDN
TA = 25C, VEN = 0V
--
1
10
A
Quiescent Current
IQ
TA = 25C, VEN = 5V, VFB = 0.8V
--
1
1.3
mA
Logic-High
2
--
18
Logic-Low
--
--
0.4
Logic Threshold
EN Voltage
V
VFB Voltage and Discharge Resistance
Feedback Threshold Voltage
VFB
TA = 25C
0.757
0.765 0.773
TA = 40C to 85C
0.755
--
0.775
V
Feedback Input Current
IFB
VFB = 0.8V, TA = 25C
--
0.01
0.1
A
VS Discharge Resistance
RDIS
VEN = 0V, VS = 0.5V
--
50
100
VREG5
TA = 25C, 6V VIN 18V,
0 < IVREG5 5mA
4.8
5.1
5.4
V
VREG5 Output
VREG5 Output Voltage
Line Regulation
Load Regulation
Output Current
IVREG5
6V VIN 18V, IVREG5 = 5mA
0 IVREG5 5mA
--
--
20
mV
--
--
100
mV
VIN = 6V, VREG5 = 4V, T A = 25C
--
70
--
mA
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RT2853A/B
Parameter
Symbol
Test Conditions
Min
Typ
Max
Unit
RDS(ON)
Switch On
Resistance
High-Side
RDS(ON)_H
TA = 25C (VBOOT VSW) = 5.5V
--
110
--
Low-Side
RDS(ON)_L
TA = 25C
--
30
--
ILIM
LSW = 1.4H
4
4.5
6
TSD
Shutdown Temperature
--
150
--
--
20
--
m
Current Limit
Current Limit
A
Thermal Shutdown
Thermal Shutdown Threshold
Thermal Shutdown Hysteresis TSD
C
On-Time Timer Control
On-Time
tON
VOUT = 1.05V
--
135
--
ns
Minimum Off-Time
tOFF(MIN)
VFB = 0.7V, TA = 25C
--
260
310
ns
Soft-Start
SS Charge Current
VSS = 0V
1.4
2
2.6
A
SS Discharge Current
VSS = 0.5V
0.1
0.2
--
mA
Wake Up VREG5
3.6
3.85
4.1
0.13
0.35
0.47
VFB Rising
85
90
95
VFB Falling
--
85
--
2.5
5
--
mA
115
120
125
%
OVP Prop Delay
--
5
--
s
UVP Trip Threshold
65
70
75
UVP Hysteresis
--
10
--
UVP Prop Delay
--
250
--
s
--
tSS
x 1.7
--
ms
UVLO
UVLO Threshold
Hysteresis
V
Power Good
PGOOD Threshold
PGOOD Sink Current
PGOOD = 0.5V
%
Output Under Voltage and Over Voltage Protection
OVP Trip Threshold
UVP Enable Delay
OVP Detect
tUVPEN
Relative to Soft-Start Time
%
Note 1. Stresses beyond those listed “Absolute Maximum Ratings” may cause permanent damage to the device. These are
stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated in
the operational sections of the specifications is not implied. Exposure to absolute maximum rating conditions may
affect device reliability.
Note 2. θJA is measured at TA = 25°C on a high effective thermal conductivity four-layer test board per JEDEC 51-7. θJC is
measured at the exposed pad of the package.
Note 3. Devices are ESD sensitive. Handling precaution is recommended.
Note 4. The device is not guaranteed to function outside its operating conditions.
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RT2853A/B
Typical Operating Characteristics
Efficiency vs. Load Current
100
RT2853A
90
RT2853B
90
80
80
VOUT = 5V
VOUT = 1.05V
70
Efficiency (%)
Efficiency (%)
Efficiency vs. Load Current
100
60
50
40
30
20
VOUT = 5V
VOUT = 1.05V
70
60
50
40
30
20
10
10
VIN = 12V
0
0.001
0.01
0.1
1
VIN = 12V
0
0.001
10
0.01
Load Current (A)
Output Voltage vs. Load Current
1.10
RT2853A
10
RT2853B
1.08
Output Voltage (V)
Output Voltage (V)
1
Output Voltage vs. Load Current
1.10
1.08
1.06
1.04
1.02
1.06
1.04
1.02
VIN = 12V, VOUT = 1.05V, IOUT = 0 to 3A
VIN = 12V, VOUT = 1.05V, IOUT = 0 to 3A
1.00
1.00
0
0.5
1
1.5
2
2.5
3
0
0.5
Load Current (A)
1
1.5
2
2.5
3
Load Current (A)
Switching Frequency vs. Load Current
800
Switching Frequency vs. Load Current
800
RT2853A
Switching Frequency (kHz)1
Switching Frequency (kHz)1
0.1
Load Current (A)
700
600
500
400
300
200
100
RT2853B
700
600
500
400
300
200
100
VIN = 12V, VOUT = 1.05V, IOUT = 0 to 3A
VIN = 12V, VOUT = 1.05V, IOUT = 0 to 3A
0
0
0
0.5
1
1.5
2
2.5
Load Current (A)
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3
0
0.5
1
1.5
2
2.5
3
Load Current (A)
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RT2853A/B
Feedback Voltage vs. Input Voltage
0.780
0.78
0.772
Feedback Voltage (V)
Feedback Voltage (V)
Feedback Voltage vs. Temperature
0.80
0.76
0.74
0.72
0.764
0.756
0.748
VIN = 12V, VOUT = 0.765V, IOUT = 0.6A
VIN = 12V, VOUT = 0.765V, IOUT = 0.6A
0.70
0.740
-50
-25
0
25
50
75
100
125
4
6
8
Temperature (°C)
VIN = 12V, VOUT = 1.05V, IOUT = 0A
16
18
VIN = 12V, VOUT = 1.05V, IOUT = 0A
950
25
Quiescent Current (µA)
Shutdown Current (µA)1
14
Quiescent Current vs. Temperature
1000
20
15
10
5
900
850
800
750
700
650
0
600
-50
-25
0
25
50
75
100
-50
125
-25
0
Temperature (°C)
25
50
75
100
125
Temperature (°C)
Current Limit vs. Input Voltage
Maximum Output Current vs. Temperature
6.0
Maximum Output Current (A)1
7.0
6.5
Current Limit (A)
12
Input Voltage (V)
Shutdown Current vs. Temperature
30
10
Upper Threshold
6.0
5.5
Lower Threshold
5.0
4.5
4.0
3.5
5.5
VIN = 18V
5.0
VIN = 12V
4.5
4.0
3.5
VIN = 12V, VOUT = 1.05V
3.0
VOUT = 1.05V
3.0
4
6
8
10
12
14
16
Input Voltage (V)
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18
-50
-25
0
25
50
75
100
125
Temperature (°C)
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RT2853A/B
Load Transient Response
Load Transient Response
RT2853B
RT2853A
VOUT
(50mV/Div)
IOUT
(1A/Div)
VOUT
(50mV/Div)
VIN = 12V, VOUT = 1.05V, IOUT = 1A to 3A
IOUT
(1A/Div)
Time (100μs/Div)
Time (100μs/Div)
Output Ripple Voltage
Power On from VIN
VIN
(10V/Div)
VOUT
(10mV/Div)
PGOOD
(5V/Div)
VOUT
(5V/Div)
VSW
(5V/Div)
VIN = 12V, VOUT = 1.05V, IOUT = 3A
IOUT
(2A/Div)
VIN = 12V, VOUT = 1.05V, IOUT = 3A
Time (500ns/Div)
Time (1ms/Div)
Power Off from VIN
Power On from EN
VIN
(10V/Div)
EN
(2V/Div)
VOUT
(5V/Div)
PGOOD
(5V/Div)
PGOOD
(5V/Div)
VOUT
(1V/Div)
I SW
(2A/Div)
VIN = 12V, VOUT = 1.05V, IOUT = 0 to 3A
VIN = 12V, VOUT = 1.05V, IOUT = 3A
Time (5ms/Div)
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IOUT
(2A/Div)
VIN = 12V, VOUT = 1.05V, IOUT = 3A
Time (1ms/Div)
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RT2853A/B
Power Off from EN
EN
(2V/Div)
VOUT
(2V/Div)
PGOOD
(5V/Div)
I SW
(2A/Div)
VIN = 12V, VOUT = 1.05V, IOUT = 3A
Time (5ms/Div)
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RT2853A/B
Functional Pin Description
Pin No.
Pin Name
Pin Function
1
FB
Feedback Input Voltage. Connect FB to the midpoint of the external feedback
resistive divider to sense the output voltage. Place the resistive divider within
5mm from the FB pin. The IC regulates VFB at 0.765V (typical).
2
VREG5
Internal Regulator Output. Connect a 1F capacitor to GND to stabilize
output voltage.
3
SS
Soft-Start Control. Connect an external capacitor between this pin and GND
to set the soft-start time.
4
GND
Ground.
5
PGOOD
Open Drain Power-good Output. PGOOD connects to PGND whenever VFB
is less than 90% of its regulation threshold (typical).
6
EN
Enable Control Input. A logic-high enables the converter; a logic-low forces
the IC into shutdown mode reducing the supply current to less than 10A.
PGND
Power Ground. PGND connects to the source of the internal N-channel
MOSFET synchronous rectifier and to other power ground nodes of the IC.
The exposed pad and the 2 PGND pins should be well soldered to the input
and output capacitors and to a large PCB area for good power dissipation.
SW
Switching Node. SW is the source of the internal N-channel MOSFET switch
and the drain of the internal N-Channel MOSFET synchronous rectifier.
Connect SW to the inductor with a wide short PCB trace and minimize its
area to reduce EMI.
BOOT
Bootstrap Supply for High-Side Gate Driver. Connect a 0.1F capacitor
between BOOT and SW to power the internal gate driver.
13, 14
VIN
Power Input. The input voltage range is from 4.5V to 18V. Must bypass with a
suitably large (10F x 2) ceramic capacitors at this pin.
15
VCC
Internal Linear Regulator Supply Input. VCC supplies power for the internal
linear regulator that powers the IC. Connect VIN to the input voltage and
bypass to ground with a 0.1F ceramic capacitor.
16
VS
Optional Output Voltage Discharge Connection. The open drain output
connects to ground when the device is disabled. If output voltage discharge is
desired, connect VS to the output voltage.
7, 8,
17 (Exposed pad)
9, 10, 11
12
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RT2853A/B
Function Block Diagram
BOOT
VREG5
VCC
Internal Regulator
VREF
Over Current
VREG5
Protection
VIN
VBIAS
UGATE
GND
SW
Driver
PVCC
2µA
Switch
Controller
Ripple
Gen.
LGATE
0.9 VREF
FB
+
SS
FB
-
On-Time
FB
Comparator
PGND
SW
PGOOD
+
-
Under & Over
Voltage
Protection
PGOOD
Comparator
VS
EN
Discharge
EN
Detailed Description
The RT2853A/B are high-performance 650kHz 3A stepdown regulators with internal power switches and
synchronous rectifiers. They feature an Advanced Constant
On-Time (ACOT®) control architecture that provides stable
operation with ceramic output capacitors without
complicated external compensation, among other benefits.
The input voltage range is from 4.5V to 18V and the output
is adjustable from 0.765V to 7V.
The proprietary ACOT® control scheme improves upon
other constant on-time architectures, achieving nearly
constant switching frequency over line, load, and output
voltage ranges. The RT2853A/B are optimized for ceramic
output capacitors. Since there is no internal clock,
response to transients is nearly instantaneous and inductor
current can ramp quickly to maintain output regulation
without large bulk output capacitance.
Constant On-Time (COT) Control
The heart of any COT architecture is the on-time oneshot. Each on-time is a pre-determined “fixed” period
that is triggered by a feedback comparator. This robust
arrangement has high noise immunity and is ideal for low
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duty cycle applications. After the on-time one-shot period,
there is a minimum off-time period before any further
regulation decisions can be considered. This arrangement
avoids the need to make any decisions during the noisy
time periods just after switching events, when the
switching node (SW) rises or falls. Because there is no
fixed clock, the high-side switch can turn on almost
immediately after load transients and further switching
pulses can ramp the inductor current higher to meet load
requirements with minimal delays.
Traditional current mode or voltage mode control schemes
typically must monitor the feedback voltage, current
signals (also for current limit), and internal ramps and
compensation signals, to determine when to turn off the
high-side switch and turn on the synchronous rectifier.
Weighing these small signals in a switching environment
is difficult to do just after switching large currents, making
those architectures problematic at low duty cycles and in
less than ideal board layouts.
Because no switching decisions are made during noisy
time periods, COT architectures are preferable in low duty
cycle and noisy applications. However, traditional COT
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DS2853A/B-04 March 2021
RT2853A/B
control schemes suffer from some disadvantages that
preclude their use in many cases. Many applications require
a known switching frequency range to avoid interference
with other sensitive circuitry. True constant on-time control,
where the on-time is actually fixed, exhibits variable
switching frequency. In a step-down converter, the duty
factor is proportional to the output voltage and inversely
proportional to the input voltage. Therefore, if the on-time
is fixed, the off-time (and therefore the frequency) must
change in response to changes in input or output voltage.
Modern pseudo-fixed frequency COT architectures greatly
improve COT by making the one-shot on-time proportional
to VOUT and inversely proportional to VIN. In this way, an
on-time is chosen as approximately what it would be for
an ideal fixed-frequency PWM in similar input/output
voltage conditions. The result is a big improvement but
the switching frequency still varies considerably over line
and load due to losses in the switches and inductor and
other parasitic effects.
Another problem with many COT architectures is their
dependence on adequate ESR in the output capacitor,
making it difficult to use highly-desirable, small, low-cost,
but low-ESR ceramic capacitors. Most COT architectures
use AC current information from the output capacitor,
generated by the inductor current passing through the
ESR, to function in a way like a current mode control
system. With ceramic capacitors the inductor current
information is too small to keep the control loop stable,
like a current mode system with no current information.
ACOT® Control Architecture
Making the on-time proportional to VOUT and inversely
proportional to VIN is not sufficient to achieve good
constant-frequency behavior for several reasons. First,
voltage drops across the MOSFET switches and inductor
cause the effective input voltage to be less than the
measured input voltage and the effective output voltage to
be greater than the measured output voltage. As the load
changes, the switch voltage drops change causing a
switching frequency variation with load current. Also, at
light loads if the inductor current goes negative, the switch
dead-time between the synchronous rectifier turn-off and
the high-side switch turn-on allows the switching node to
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DS2853A/B-04 March 2021
rise to the input voltage. This increases the effective ontime and causes the switching frequency to drop
noticeably.
One way to reduce these effects is to measure the actual
switching frequency and compare it to the desired range.
This has the added benefit eliminating the need to sense
the actual output voltage, potentially saving one pin
connection. ACOT® uses this method, measuring the
actual switching frequency (at SW) and modifying the ontime with a feedback loop to keep the average switching
frequency in the desired range.
To achieve good stability with low-ESR ceramic capacitors,
ACOT® uses a virtual inductor current ramp generated
inside the IC. This internal ramp signal replaces the ESR
ramp normally provided by the output capacitor's ESR.
The ramp signal and other internal compensations are
optimized for low-ESR ceramic output capacitors.
ACOT® One-shot Operation
The RT2853A/B control algorithm is simple to understand.
The feedback voltage, with the virtual inductor current ramp
added, is compared to the reference voltage. When the
combined signal is less than the reference the on-time
one-shot is triggered, as long as the minimum off-time
one-shot is clear and the measured inductor current
(through the synchronous rectifier) is below the current
limit. The on-time one-shot turns on the high-side switch
and the inductor current ramps up linearly. After the ontime, the high-side switch is turned off and the synchronous
rectifier is turned on and the inductor current ramps down
linearly. At the same time, the minimum off-time one-shot
is triggered to prevent another immediate on-time during
the noisy switching time and allow the feedback voltage
and current sense signals to settle. The minimum off-time
is kept short (260ns typical) so that rapidly-repeated ontimes can raise the inductor current quickly when needed.
Discontinuous Operating Mode (RT2853A Only)
After soft start, the RT2853B operates in fixed frequency
mode to minimize interference and noise problems. The
RT2853A uses variable-frequency discontinuous switching
at light loads to improve efficiency. During discontinuous
switching, the on-time is immediately increased to add
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RT2853A/B
“hysteresis” to discourage the IC from switching back to
continuous switching unless the load increases
substantially.
The IC returns to continuous switching as soon as an ontime is generated before the inductor current reaches zero.
The on-time is reduced back to the length needed for
650kHz switching and encouraging the circuit to remain
in continuous conduction, preventing repetitive mode
transitions between continuous switching and
discontinuous switching.
Current Limit
The RT2853A/B current limit is a cycle-by-cycle “valley”
type, measuring the inductor current through the
synchronous rectifier during the off-time while the inductor
current ramps down. The current is determined by
measuring the voltage between source and drain of the
synchronous rectifier, adding temperature compensation
for greater accuracy. If the current exceeds the upper
current limit, the on-time one-shot is inhibited until the
inductor current ramps down below the upper current limit
plus a wide hysteresis band of about 1A until it drops
below the lower current limit level. Thus, only when the
inductor current is well below the upper current limit is
another on-time permitted. This arrangement prevents the
average output current from greatly exceeding the
guaranteed upper current limit value, as typically occurs
with other valley-type current limits. If the output current
exceeds the available inductor current (controlled by the
current limit mechanism), the output voltage will drop. If it
drops below the output under-voltage protection level (see
next section) the IC will stop switching to avoid excessive
heat.
The RT2853B also includes a negative current limit to
protect the IC against sinking excessive current and
possibly damaging the IC. If the voltage across the
synchronous rectifier indicates the negative current is too
high, the synchronous rectifier turns off until after the next
high-side on-time. The RT2853A does not sink current
and therefore does not need a negative current limit.
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Hiccup Mode
The RT2853AHGQW/ RT2853BHGQW, use hiccup mode
OVP and UVP. When the protection function is triggered,
the IC will shut down for a period of time and then attempt
to recover automatically. Hiccup mode allows the circuit
to operate safely with low input current and power
dissipation, and then resume normal operation as soon
as the overload or short circuit is removed. During hiccup
mode, the shutdown time is determined by the capacitor
at SS. A 0.5μA current source discharges VSS from its
starting voltage (normally VREG5). The IC remains shut
down until VSS reaches 0.2V, about 40ms for a 3.9nF
capacitor. At that point the IC begins to charge the SS
capacitor at 2μA, and a normal start-up occurs. If the fault
remains, OVP and UVP protection will be enabled when
VSS reaches 2.2V (typical). The IC will then shut down
and discharge the SS capacitor from the 2.2V level, taking
about 17ms for a 3.9nF SS capacitor.
Latch-Off Mode
The RT2853ALGQW/ RT2853BLGQW, use latch-off mode
OVP and UVP. When the protection function is triggered
the IC will shut down. The IC stops switching, leaving
both switches open, and is latched off. To restart operation,
toggle EN or power the IC off and then on again.
Input Under-voltage Lock-out
In addition to the enable function, the RT2853A/B feature
an under-voltage lock-out (UVLO) function that monitors
the internal linear regulator output (VREG5). To prevent
operation without fully-enhanced internal MOSFET
switches, this function inhibits switching when VREG5
drops below the UVLO-falling threshold. The IC resumes
switching when VREG5 exceeds the UVLO-rising
threshold.
Shut-down, Start-up and Enable (EN)
The enable input (EN) has a logic-low level of 0.4V. When
VEN is below this level the IC enters shutdown mode and
supply current drops to less than 10μA. When VEN exceeds
its logic-high level of 2V the IC is fully operational.
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RT2853A/B
Between these 2 levels there are 2 thresholds (0.8V typical
and 1.2V typical). When VEN exceeds the lower threshold
the internal bias regulators begin to function and supply
current increases above the shutdown current level.
Switching operation begins when VEN exceeds the upper
threshold. Unlike many competing devices, EN is a high
voltage input that can be safely connected to VIN (up to
18V) for automatic start-up.
Soft-Start (SS)
The RT2853A/B soft-start uses an external pin (SS) to
clamp the output voltage and allow it to slowly rise. After
VEN is high and VREG5 exceeds its UVLO threshold, the
IC begins to source 2μA from the SS pin. An external
capacitor at SS is used to adjust the soft-start timing.
The available capacitance range is from 2.7nF to 220nF.
Do not leave SS unconnected.
During start-up, while the SS capacitor charges, the
RT2853A/B operate in discontinuous mode with very small
pulses. This prevents negative inductor currents and keeps
the circuit from sinking current. Therefore, the output
voltage may be pre-biased to some positive level before
start-up. Once the VSS ramp charges enough to raise the
internal reference above the feedback voltage, switching
will begin and the output voltage will smoothly rise from
the pre-biased level to its regulated level. After VSS rises
above about 2.2V output over-and under-voltage protections
are enabled and the RT2853B begins continuous-switching
operation.
An internal linear regulator (VREG5) produces a 5.1V
supply from VIN that powers the internal gate drivers, PWM
logic, reference, analog circuitry, and other blocks. If VIN
is 6V or greater, VREG5 is guaranteed to provide significant
power for external loads.
External Bootstrap Capacitor
Connect a 0.1μF low ESR ceramic capacitor between
BOOT and SW. This bootstrap capacitor provides the gate
driver supply voltage for the high-side N-channel MOSFET
switch.
Over Temperature Protection
The RT2853A/B includes an over temperature protection
(OTP) circuitry to prevent overheating due to excessive
power dissipation. The OTP will shut down switching
operation when the junction temperature exceeds 150°C.
Once the junction temperature cools down by
approximately 20°C the IC will resume normal operation
with a complete soft-start. For continuous operation,
provide adequate cooling so that the junction temperature
does not exceed 150°C.
Output Discharge Control
When EN pin is low, the RT2853A/B will discharge the
output with an internal 50Ω MOSFET connected between
VS to GND pin.
OVP/UVP Protection
The RT2853A/B detects over and under voltage conditions
by monitoring the feedback voltage on FB pin. The two
functions are enabled after approximately 1.7 times the
soft-start time. When the feedback voltage becomes
higher than 120% of the target voltage, the OVP
comparator will go high to turn off both internal high-side
and low-side MOSFETs. When the feedback voltage is
lower than 70% of the target voltage for 250μs, the UVP
comparator will go high to turn off both internal high-side
and low-side MOSFETs.
PGOOD Comparator
PGOOD is an open drain output controlled by a comparator
connected to the feedback signal. If FB exceeds 90% of
the internal reference voltage, PGOOD will be high
impedance. Otherwise, the PGOOD output is connected
to PGND.
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RT2853A/B
Typical Application Circuit
RT2853A/B
VIN
13, 14
C1
10µF x 2
C2
0.1µF
15
VIN
VCC
Output Signal
VREG5
SW
BOOT
VS
R3 100k
5 PGOOD
6 EN
Input Signal
C5
3.3nF
L1
1.4µH
9, 10, 11
C6
0.1µF
12
VOUT
1.05V/3A
C3
R1
8.25k
16
C7
22µF x 2
FB 1
R2
22.1k
VREG5 2
VREG5
C4
3 SS
1µF
GND PGND
7, 8, 17 (Exposed Pad)
4
Table 1. Suggested Component Values (VIN = 12V)
VOUT (V)
R1 (k)
R2 (k)
C3 (pF)
L1 (H)
C7 (F)
1
6.81
22.1
--
1.4
22 to 68
1.05
8.25
22.1
--
1.4
22 to 68
1.2
12.7
22.1
--
1.4
22 to 68
1.8
30.1
22.1
5 to 22
2
22 to 68
2.5
49.9
22.1
5 to 22
2
22 to 68
3.3
73.2
22.1
5 to 22
2
22 to 68
5
124
22.1
5 to 22
3.3
22 to 68
7
180
22.1
5 to 22
3.3
22 to 68
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RT2853A/B
Design Procedure
Inductor Selection
Selecting an inductor involves specifying its inductance
and also its required peak current. The exact inductor value
is generally flexible and is ultimately chosen to obtain the
best mix of cost, physical size, and circuit efficiency.
Lower inductor values benefit from reduced size and cost
and they can improve the circuit's transient response, but
they increase the inductor ripple current and output voltage
ripple and reduce the efficiency due to the resulting higher
peak currents. Conversely, higher inductor values increase
efficiency, but the inductor will either be physically larger
or have higher resistance since more turns of wire are
required and transient response will be slower since more
time is required to change current (up or down) in the
inductor. A good compromise between size, efficiency,
and transient response is to use a ripple current (ΔIL) about
20-50% of the desired full output load current. Calculate
the approximate inductor value by selecting the input and
output voltages, the switching frequency (f SW), the
maximum output current (IOUT(MAX)) and estimating a ΔIL
as some percentage of that current.
L=
VOUT VIN VOUT
VIN fSW IL
Once an inductor value is chosen, the ripple current (ΔIL)
is calculated to determine the required peak inductor
current.
VOUT VIN VOUT
I
IL =
and IL(PEAK) = IOUT(MAX) L
VIN fSW L
2
To guarantee the required output current, the inductor
needs a saturation current rating and a thermal rating that
exceeds IL(PEAK). These are minimum requirements. To
maintain control of inductor current in overload and shortcircuit conditions, some applications may desire current
ratings up to the current limit value. However, the IC's
output under-voltage shutdown feature make this
unnecessary for most applications.
IL(PEAK) should not exceed the minimum value of IC's upper
current limit level or the IC may not be able to meet the
desired output current. If needed, reduce the inductor ripple
current (ΔIL) to increase the average inductor current (and
the output current) while ensuring that IL(PEAK) does not
exceed the upper current limit level.
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For best efficiency, choose an inductor with a low DC
resistance that meets the cost and size requirements.
For low inductor core losses some type of ferrite core is
usually best and a shielded core type, although possibly
larger or more expensive, will probably give fewer EMI
and other noise problems.
Considering the Typical Operating Circuit for 1.05V output
at 3A and an input voltage of 12V, using an inductor ripple
of 1A (33%), the calculated inductance value is :
L=
1.05V 12V 1.05V
= 1.47μH
12V 650kHz 1A
The ripple current was selected at 1A and, as long as we
use the calculated 1.47μH inductance, that should be the
actual ripple current amount. Typically the exact calculated
inductance is not readily available and a nearby value is
chosen. In this case 1.4μH was available and actually used
in the typical circuit. To illustrate the next calculation,
assume that for some reason is was necessary to select
a 1.8μH inductor (for example). We would then calculate
the ripple current and required peak current as below :
1.05V 12V 1.05V
IL =
= 0.82A
12V 650kHz 1.8μH
and IL(PEAK) = 3A 0.82 = 3.41A
2
For the 1.8μH value, the inductor's saturation and thermal
rating should exceed 3.41A. Since the actual value used
was 1.4μH and the ripple current exactly 1A, the required
peak current is 3.53A.
Input Capacitor Selection
The input filter capacitors are needed to smooth out the
switched current drawn from the input power source and
to reduce voltage ripple on the input. The actual
capacitance value is less important than the RMS current
rating (and voltage rating, of course). The RMS input ripple
current (IRMS) is a function of the input voltage, output
voltage, and load current :
IRMS = IOUT
VOUT VVIN VOUT
VVIN
Ceramic capacitors are most often used because of their
low cost, small size, high RMS current ratings, and robust
surge current capabilities. However, take care when these
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RT2853A/B
capacitors are used at the input of circuits supplied by a
wall adapter or other supply connected through long, thin
wires. Current surges through the inductive wires can
induce ringing at the RT2853A/B's input which could
potentially cause large, damaging voltage spikes VIN. If
this phenomenon is observed, some bulk input capacitance
may be required. Ceramic capacitors (to meet the RMS
current requirement) can be placed in parallel with other
types such as tantalum, electrolytic, or polymer (to reduce
ringing and overshoot).
Choose capacitors rated at higher temperatures than
required. Several ceramic capacitors may be paralleled to
meet the RMS current, size, and height requirements of
the application. The typical operating circuit uses two 10μF
and one 0.1μF low ESR ceramic capacitors on the input.
Output Capacitor Selection
The RT2853A/B are optimized for ceramic output
capacitors and best performance will be obtained using
them. The total output capacitance value is usually
determined by the desired output voltage ripple level and
transient response requirements for sag (undershoot on
positive load steps) and soar (overshoot on negative load
steps).
Output Ripple
Output ripple at the switching frequency is caused by the
inductor current ripple and its effect on the output
capacitor's ESR and stored charge. These two ripple
components are called ESR ripple and capacitive ripple.
Since ceramic capacitors have extremely low ESR and
relatively little capacitance, both components are similar
in amplitude and both should be considered if ripple is
critical.
VRIPPLE = VRIPPLE(ESR) VRIPPLE(C)
VRIPPLE(ESR) = IL RESR
IL
VRIPPLE(C) =
8 COUT fSW
For the Typical Operating Circuit for 1.05V output and an
inductor ripple of 1A, with 2 x 22μF output capacitance
each with about 10mΩ ESR including PCB trace
resistance, the output voltage ripple components are :
VRIPPLE(C) =
1A
= 4.4mV
8 44μF 0.65MHz
VRIPPLE = 5mV 4.4mV = 9.4mV
Output Transient Undershoot and Overshoot
In addition to voltage ripple at the switching frequency,
the output capacitor and its ESR also affect the voltage
sag (undershoot) and soar (overshoot) when the load steps
up and down abruptly. The ACOT® transient response is
very quick and output transients are usually small.
However, the combination of small ceramic output
capacitors (with little capacitance), low output voltages
(with little stored charge in the output capacitors), and
low duty cycle applications (which require high inductance
to get reasonable ripple currents with high input voltages)
increases the size of voltage variations in response to
very quick load changes. Typically, load changes occur
slowly with respect to the IC's 650kHz switching frequency.
But some modern digital loads can exhibit nearly
instantaneous load changes and the following section
shows how to calculate the worst-case voltage swings in
response to very fast load steps.
The output voltage transient undershoot and overshoot each
have two components : the voltage steps caused by the
output capacitor's ESR, and the voltage sag and soar due
to the finite output capacitance and the inductor current
slew rate. Use the following formulas to check if the ESR
is low enough (typically not a problem with ceramic
capacitors) and the output capacitance is large enough to
prevent excessive sag and soar on very fast load step
edges, with the chosen inductor value.
The amplitude of the ESR step up or down is a function of
the load step and the ESR of the output capacitor:
VESR_STEP = IOUT RESR
The amplitude of the capacitive sag is a function of the
load step, the output capacitor value, the inductor value,
the input-to-output voltage differential, and the maximum
duty cycle. The maximum duty cycle during a fast transient
is a function of the on-time and the minimum off-time since
the ACOT® control scheme will ramp the current using
on-times spaced apart with minimum off-times, which is
VRIPPLE(ESR) = 1A 5m = 5mV
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RT2853A/B
as fast as allowed. Calculate the approximate on-time
(neglecting parasitics) and maximum duty cycle for a given
input and output voltage as :
VOUT
tON
tON =
and DMAX =
VIN fSW
tON tOFF(MIN)
The actual on-time will be slightly longer as the IC
compensates for voltage drops in the circuit, but we can
neglect both of these since the on-time increase
compensates for the voltage losses. Calculate the output
voltage sag as :
VSAG =
L (IOUT )2
2 COUT VIN(MIN) DMAX VOUT
The amplitude of the capacitive soar is a function of the
load step, the output capacitor value, the inductor value
and the output voltage :
L (IOUT )
2 COUT VOUT
2
VSOAR =
For the Typical Operating Circuit for 1.05V output, the
circuit has an inductor 1.4μH and 2 x 22μF output
capacitance with 5mΩ ESR each. The ESR step is 3A x
2.5mΩ = 7.5mV which is small, as expected. The output
voltage sag and soar in response to full 0A-3A-0A
instantaneous transients are :
1.05V
tON =
= 135ns
12V 650kHz
and DMAX =
VSAG =
135ns
= 0.34
135ns 260ns
1.4μH (3A)2
= 47mV
2 44μF 12V 0.34 1.05V
1.4μH (3A)
= 136mV
2 44μF 1.05V
2
VSOAR =
The sag is about 4% of the output voltage and the soar is
a full 13% of the output voltage. The ESR step is negligible
here but it does partially add to the soar, so keep that in
mind whenever using higher-ESR output capacitors.
The soar is typically much worse than the sag in highinput, low-output step-down converters because the high
input voltage demands a large inductor value which stores
lots of energy that is all transferred into the output if the
load stops drawing current. Also, for a given inductor, the
soar for a low output voltage is a greater voltage change
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and an even greater percentage of the output voltage. This
is illustrated by comparing the previous to the next
example.
The Typical Operating Circuit for 12V to 3.3V with a 2μH
inductor and 2 x 22μF output capacitance can be used to
illustrate the effect of a higher output voltage. The output
voltage sag and soar in response to full 0A-3A-0A
instantaneous transients are calculated as follows :
t ON =
3.3V
= 423ns
12V 650kHz
and DMAX =
423ns
= 0.62
423ns 260ns
VSAG =
2μH (3A)2
= 49.5mV
2 44μF 12V 0.62 3.3V
VSOAR =
2μH (3A)2
= 62mV
2 44μF 3.3V
In this case the sag is about 1.5% of the output voltage
and the soar is only 2% of the output voltage.
Any sag is always short-lived, since the circuit quickly
sources current to regain regulation in only a few switching
cycles. With the RT2853B, any overshoot transient is
typically also short-lived since the converter will sink
current, reversing the inductor current sharply until the
output reaches regulation again. The RT2853A's
discontinuous operation at light loads prevents sinking
current so, for that IC, the output voltage will soar until
load current or leakage brings the voltage down to normal.
Most applications never experience instantaneous full load
steps and the RT2853A/B's high switching frequency and
fast transient response can easily control voltage regulation
at all times. Also, since the sag and soar both are
proportional to the square of the load change, if load steps
were reduced to 1A (from the 3A examples preceding) the
voltage changes would be reduced by a factor of almost
ten. For these reasons sag and soar are seldom an issue
except in very low-voltage CPU core or DDR memory
supply applications, particularly for devices with high clock
frequencies and quick changes into and out of sleep
modes. In such applications, simply increasing the amount
of ceramic output capacitor (sag and soar are directly
proportional to capacitance) or adding extra bulk
capacitance can easily eliminate any excessive voltage
transients.
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RT2853A/B
In any application with large quick transients, always
calculate soar to make sure that over-voltage protection
will not be triggered. Under-voltage is not likely since the
threshold is very low (70%), that function has a long delay
(250μs), and the IC will quickly return the output to
regulation. Over-voltage protection has a minimum
threshold of 115% and short delay of 5μs and can actually
be triggered by incorrect component choices, particularly
for the RT2853A which does not sink current.
Output Capacitors Stability Criteria
The RT2853A/B's ACOT® control architecture uses an
internal virtual inductor current ramp and other
compensation that ensures stability with any reasonable
output capacitor. The internal ramp allows the IC to operate
with very low ESR capacitors and the IC is stable with
very small capacitances. Therefore, output capacitor
selection is nearly always a matter of meeting output
voltage ripple and transient response requirements, as
discussed in the previous sections. For the sake of the
unusual application where ripple voltage is unimportant
and there are few transients (perhaps battery charging or
LED lighting) the stability criteria are discussed below.
The equations giving the minimum required capacitance
for stable operation include a term that depends on the
output capacitor's ESR. The higher the ESR, the lower
the capacitance can be and still ensure stability. The
equations can be greatly simplified if the ESR term is
removed by setting ESR to zero. The resulting equation
gives the worst-case minimum required capacitance and
it is usually sufficiently small that there is usually no need
for the more exact equation.
The required output capacitance (COUT) is a function of
the inductor value (L) and the input voltage (VIN) :
11
COUT 5.23 10
VIN L
The worst-case high capacitance requirement is for low
VIN and small inductance, so a 5V to 3.3V converter is
used for an example. Using the inductance equation in a
previous section to determine the required inductance :
L=
3.3V 5V 3.3V
= 1.73μH
5V 650kHz 1A
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Therefore, the required minimum capacitance for the 5V
to 3.3V converter is :
11
COUT 5.23 10
= 6μF
5V 1.73μH
Using the 12V to 1.05V typical application as another
example :
11
COUT 5.24 10
= 3.1μF
12V 1.4μH
Any ESR in the output capacitor lowers the required
minimum output capacitance, sometimes considerably.
For the rare application where that is needed and useful,
the equation including ESR is given here :
VOUT
COUT
2 fSW VIN (RESR 13647 L VOUT )
As can be seen, setting RESR to zero and simplifying the
equation yields the previous simpler equation. To allow
for the capacitor's temperature and bias voltage coefficients,
use at least double the calculated capacitance and use a
good quality dielectric such as X5R or X7R with an
adequate voltage rating since ceramic capacitors exhibit
considerable capacitance reduction as their bias voltage
increases.
Feed-forward Capacitor (C3)
The RT2853A/B are optimized for ceramic output
capacitors and for low duty cycle applications. This
optimization makes circuit stability easy to achieve with
reasonable output capacitors. However, the optimization
affects the quality factor (Q) of the circuit and therefore its
transient response. To avoid an under-damped response
(high Q) and its potential ringing, the internal compensation
was chosen to achieve perfect damping for low output
voltages, where the FB divider has low attenuation (VOUT
is close to VREF). For high-output voltages, with high
feedback attenuation, the circuit's response becomes overdamped and transient response can be slowed. In highoutput voltage circuits (VOUT > 1.5V) transient response
is improved by adding a small “feed-forward” capacitor
(C3) across the upper FB divider resistor, to increase the
circuit's Q and reduce damping to speed up the transient
response without affecting the steady-state stability of
the circuit. Choose a capacitor value that gives, together
with the divider impedance at FB, a time-constant between
100ns and 0.5μs. The divider impedance at FB is R1 in
parallel with R2. C3 can be safely left out in low-output
voltage circuits and if fast transient response is not required.
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Applications Information
Current-Sinking Applications (RT2853B)
Soft-Start (SS)
The RT2853B's is not recommended for current sinking
applications even though its continuous switching
operation allows the IC to sink some current. Sinking
enables a fast recovery from output voltage overshoot
caused by load transients and is normally useful for
applications requiring negative currents, such as DDR VTT
The RT2853A/B soft-start uses an external capacitor at
SS to adjust the soft-start timing according to the following
equation :
C
(nF) 0.765V
tSS (ms) = SS
ISS (μA)
The available capacitance range is from 2.7nF to 220nF. If
a 3.9nF capacitor is used, the typical soft-start will be
1.5ms. Do not leave SS unconnected.
bus termination applications and changing-output voltage
applications where the output voltage needs to slew
quickly from one voltage to another. However, the IC's
negative current limit is set low (about 1.6A) and the current
limit behavior latches the synchronous rectifier off until
the high-side switch's next pulse, to prevent the possibility
of IC damage from large negative currents. Therefore,
sinking current is not necessarily available at all times.
If implementing applications where current-sinking may
occur, take care to allow for the current that is delivered
to the input supply. A step-down converter in sinking
operation functions like a backwards step-up converter.
The current that is sunk at its output terminals is delivered
up to its input terminals. If this current has no outlet, the
input voltage will rise.
A good arrangement for long-term sinking applications is
for a sinking supply (supply A) that is sinking current
sourced from supply B, to both be powered by the same
input supply. That way, any current delivered back to the
input by supply A is current that just left the input through
supply B. In this way, the current simply makes a round
trip and the input supply will not rise.
In cases where this is not possible, make sure that there
are sufficient other loads on the input supply to prevent
that supply's voltage from rising high enough to cause
damage to itself or any of its loads. In cases where the
sinking is not long-term, such as output-voltage slewing
applications, make sure there is sufficient input capacitance
to control any input voltage rise. The worst-case voltage
rise is :
C
VOUT
VIN = OUT
CIN
Enable Operation (EN)
For automatic start-up the high-voltage EN pin can be
connected to VIN, either directly or through a 100kΩ
resistor. Its large hysteresis band makes EN useful for
simple delay and timing circuits. EN can be externally
pulled to VIN by adding a resistor-capacitor delay (REN
and CEN in Figure 1). Calculate the delay time using EN's
internal threshold where switching operation begins (1.4V,
typical).
An external MOSFET can be added to implement digital
control of EN when no system voltage above 2V is available
(Figure 2). In this case, a 100kΩ pull-up resistor, REN, is
connected between VIN and the EN pin. MOSFET Q1 will
be under logic control to pull down the EN pin. To prevent
enabling circuit when VIN is smaller than the VOUT target
value or some other desired voltage level, a resistive voltage
divider can be placed between the input voltage and ground
and connected to EN to create an additional input undervoltage lockout threshold (Figure 3).
EN
VIN
REN
EN
RT2853A/B
CEN
GND
Figure 1. External Timing Control
VIN
Enable
REN
100k
EN
Q1
RT2853A/B
GND
Figure 2. Digital Enable Control Circuit
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RT2853A/B
VIN
REN1
External BOOT Bootstrap Diode
EN
REN2
RT2853A/B
GND
Figure 3. Resistor Divider for Lockout Threshold Setting
Output Voltage Setting
Set the desired output voltage using a resistive divider
from the output to ground with the midpoint connected to
FB. The output voltage is set according to the following
equation :
R1
VOUT = 0.765 (1
)
R2
When the input voltage is lower than 5.5V it is
recommended to add an external bootstrap diode between
VIN (or VCC) and the BOOT pin to improve enhancement
of the internal MOSFET switch and improve efficiency.
The bootstrap diode can be a low cost one such as 1N4148
or BAT54.
5V
BOOT
0.1µF
RT2853A/B
SW
VOUT
R1
FB
RT2853A/B
External BOOT Capacitor Series Resistance
R2
GND
Figure 4. Output Voltage Setting
Place the FB resistors within 5mm of the FB pin. Choose
R2 between 10kΩ and 100kΩ to minimize power
consumption without excessive noise pick-up and
calculate R1 as follows :
R2 (VOUT 0.765V)
R1 =
0.765V
For output voltage accuracy, use divider resistors with 1%
or better tolerance.
Under Voltage Lockout Protection
The RT2853A/B feature an under-voltage lock-out (UVLO)
function that monitors the internal linear regulator output
(VREG5) and prevents operation if VREG5 is too low. In
some multiple input voltage applications, it may be
desirable to use a power input that is too low to allow
VREG5 to exceed the UVLO threshold. In this case, if
there is another low-power supply available that is high
enough to operate the VREG5 regulator, connecting that
supply to VCC will allow the IC to operate, using the lowervoltage high-power supply for the DC/DC power path.
Because of the internal linear regulator, any supply
regulated or unregulated) between 4.5V and 18V will
operate the IC.
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Figure 5. External Bootstrap Diode
The internal power MOSFET switch gate driver is
optimized to turn the switch on fast enough for low power
loss and good efficiency, but also slow enough to reduce
EMI. Switch turn-on is when most EMI occurs since VSW
rises rapidly. During switch turn-of, SW is discharged
relatively slowly by the inductor current during the deadtime between high-side and low-switch on-times.
In some cases it is desirable to reduce EMI further, at the
expense of some additional power dissipation. The switch
turn-on can be slowed by placing a small (
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