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SP7661ER-L

SP7661ER-L

  • 厂商:

    SIPEX(迈凌)

  • 封装:

    VFDFN26

  • 描述:

    IC REG BUCK ADJUSTABLE 3A 26DFN

  • 数据手册
  • 价格&库存
SP7661ER-L 数据手册
SP7661 Wide Input Voltage Range 3A, 600kHz, Buck Regulator PowerBlox SP7661 FEATURES DFN PACKAGE 7mm x 4mm (Option 2) ■ 4.75V to 22V Input Voltage Range using Single Supply ■ 3V to 22V Input Voltage Range using Dual Supply LX 26 ■ ±1% 0.8V Reference BOTTOM VIEW LX 25 ■ 3A Output Capability Heatsink Pad 1 Connect to Lx LX 24 ■ Current Limiting using Inductor DCR LX 23 ■ Built in Low RDS(ON) Power Switches VCC ■ 600kHz Fixed Frequency Operation 22 Pin 27 1 PGND 2 PGND 3 PGND 4 PGND 5 GND UVIN 21 6 VFB ■ Short Circuit Protection with Auto-Restart GND 20 7 COMP ■ Wide BW Amp Allows Type II or III Compensation GND 19 8 SS 9 GND ■ Over Temperature Protection ■ Programmable Soft Start VIN 18 ■ Fast Transient Response BST 17 ■ High Efficiency: Greater than 93% Possible LX 16 ■ Nonsynchronous Start-Up into a Pre-Charged Output LX 15 ■ Available in RoHS Compliant, Lead Free Packaging: TM Heatsink Pad 2 Connect to GND Pin 28 10 ISN Heatsink Pad 3 Connect to VINP Pin 29 LX 14 Small 7mm x 4mm DFN 11 ISP 12 SWN 13 VINP ■ U.S. Patent #6,922,041 DESCRIPTION The SP7661 is a synchronous step-down switching regulator optimized for high efficiency. The part is designed for use with a single 4.75V to 22V single supply or 3V to 22V input if an external Vcc is provided. The SP7661 provides a fully integrated buck regulator solution using a fixed 600kHz frequency, PWM voltage mode architecture. Protection features include UVLO, thermal shutdown, output current limit and short circuit protection. The SP7661 is available in the space saving DFN package. TYPICAL APPLICATION CIRCUIT VIN 12V (9.6V-22V)  VOUT 3.30V, 0-3A  C 22uF LX GND  LX 26 LX 25 LX 24 LX 23 PGND RZ2 3 PGND 4.02k 4 PGND 5 GND VCC 22 6 VFB UVIN 2 7 COMP GND 20 8 SS GND 19 9 GND VIN 18 0 ISN BST 7  ISP LX 6 2 SWN LX 15 3 VIN LX 14 CSS 47nF ISN ISP LX SP7661 29 C9 6.8nF U1 VIN PAD CP 56pF CF1 100pF R3 5.11k 2 VFB 6800pF PGND SWN PAD 2 GND PAD 28  CZ2 L1, Wurth -744311220 2.2uH, 14 mOhm, 7x7mm, 9A 27 C2 22uF LX C5 100uF C4 47nF R9 ISP 61.9k RZ3 400 ISN CVCC 4.7uF R 0k CZ3 1500pF NC VFB  4 GND R2 3.16k SD101AWS DBST 22nF CBST Rs2 1 Ohm Mar 1-07 Rev N  3 R4 5.11k Cs2 2.2nF SP7661 Wide Input Voltage Range 3A, 600kHz Buck Regulator  © 2007 Sipex Corporation ABSOLUTE MAXIMUM RATINGS These are stress ratings only and functional operation of the device at these ratings or any other above those indicated in the operation sections of the specifications below is not implied. Exposure to absolute maximum rating conditions for extended periods of time may affect reliability. Storage Temperature.................................................... -65°C to 150°C Power Dissipation.........................................Internally Limited via OTP Lead Temperature (Soldering, 10 sec)........................................300°C ESD Rating............................................................................ 2kV HBM Thermal Resistance θJC............................................................ 5°C/W VCC ...................................................................................................7V VIN .................................................................................................25V BST................................................................................................. 30V LX-BST................................................................................-0.3V to 7V LX.........................................................................................-1V to 30V All other pins........................................................... -0.3V to VCC + 0.3V ELECTRICAL SPECIFICATIONS Specifications are for TAMB = TJ = 25°C, and those denoted by ♦ apply over the full operating range, -40°C< Tj< 125°C. Unless otherwise specified: 4.5V < Vcc < 5.5V, 3V < Vin < 22V, BST = LX + 5V, UVIN = 3V, CVCC = 1µF, CCOMP = 0.1µF, Css = 50nF. PARAMETER MIN TYP MAX UNITS ♦ CONDITIONS QUIESCENT CURRENT Vin Supply Current (No switching) Vin Supply Current (switching) BST Supply Current (No switching) BST Supply Current (switching) ♦ Vfb= 0.9V ♦ Vfb= 0.9V 1.5 3.0 mA 8 14 mA 0.2 0.4 mA 3 6 mA Vcc UVLO Start Threshold 4.00 4.25 4.50 V ♦ Vcc UVLO Hysteresis 100 200 300 mV UVIN Start Threshold 2.30 2.50 2.65 V UVIN Hysteresis 200 300 400 mV µA ♦ ♦ ♦ ♦ PROTECTION: UVLO UVIN Input Current 1.0 UVIN=3.0V ERROR AMPLIFIER REFERENCE 2X Gain Config., Measure Vfb; Vcc=5V Error Amplifier Reference 0.792 0.800 0.808 V Error Amplifier Reference Over Line 0.784 0.800 0.816 V ♦ µA ♦ Vfb=0.9V, COMP=0.9V COMP Sink Current COMP Source Current 70 150 230 -230 -150 -70 µA ♦ Vfb=0.9V, COMP=0.9V 50 200 nA ♦ Vfb=0.8V 3.5 3.8 V Vfb Input Bias Current COMP Clamp 3.2 COMP Clamp Temp. Coefficient Vfb=0.7V, TA=25°C mV/°C -2.0 VCC Linear Regulator VCC Output Voltage Dropout Voltage Mar 1-07 Rev N 4.7 5.0 4.51 4.73 250 500 5.3 750 V mV VIN = 6 to 23V, ILOAD = 0mA to 30mA ♦ VIN = 5V, 20mA Vin-Vout = Dropout voltage when ♦ Vcc regulated drops by 2%. IVCC = 30 mA. ♦ SP7661 Wide Input Voltage Range 3A, 600kHz Buck Regulator  © 2007 Sipex Corporation ELECTRICAL SPECIFICATIONS Specifications are for TAMB = TJ = 25°C, and those denoted by ♦ apply over the full operating range, -40°C< Tj< 125°C. Unless otherwise specified: 4.5V < Vcc < 5.5V, 3V < Vin < 22V, BST = LX + 5V, UVIN = 3V, CVCC = 1µF, CCOMP = 0.1µF, Css = 50nF. PARAMETER MIN TYP MAX UNITS ♦ CONDITIONS CONTROL LOOP: PWM COMPARATOR, RAMP & LOOP DELAY PATH Ramp Amplitude RAMP Offset 0.80 1.00 1.20 V 1.7 2.0 2.3 V Ramp offset Temperature Coefficient -2 GH Minimum Pulse Width 150 Maximum Controllable Duty Ratio 92 ♦ ♦ mV/°C 180 97 ns ♦ % ♦ % ♦ Maximum Duty Ratio 100 Internal Oscillator Ratio 510 600 690 kHZ ♦ SS Charge Current: -16 -10 -4.0 µA ♦ SS Discharge Current: 1.0 2.0 3.0 mA ♦ Valid for 20 cycles TIMERS: SOFTSTART Fault Present, SS=0.2V PROTECTION: SHORT CIRCUIT, OVERCURRENT & THERMAL Short Circuit Threshold Voltage 0.2 0.25 0.3 V ♦ 90 Hiccup Timeout Overcurrent Threshold 54 Voltage ISP, ISN Common Mode 0 Range Thermal Shutdown 135 Temperature Thermal Recovery Temperature Thermal Hysteresis 110 130 ms ♦ 60 66 mV 3.6 V 155 °C 145 135 °C 10 °C Vfb=0.5V Measured ISP - ISN Guaranteed by design OUTPUT: POWER STAGE High Side Switch RDSON 35 75 mΩ Synchronous Low Side Switch RDSON 35 75 mΩ Maximum Output Current Mar 1-07 Rev N 3 A SP7661 Wide Input Voltage Range 3A, 600kHz Buck Regulator  VGS=4.5V; Idrain=4.1A; TAMB=25°C VGS=4.5V; Idrain=4.1A; TAMB=25°C ♦ © 2007 Sipex Corporation CONTROLLER BLOCK DIAGRAM C OMP VC C A S Y NC . S TAR TUP C OMP A R A TOR SS GL HOLD OFF V FB INT 1.6 V BS T P W M LO O P V FB VC C Gm ER R OR A MP LIFIE R RE S E T DOMINA NT VCC 10 uA FA ULT Gm VPOS S OFTS TA R T INP UT R Q 0.1V P OS R E F SS GH S Y NC HR O NO US DR I V E R QP W M S WN S FAULT GL FA ULT 600 kHZ P GN D R AMP =1V C LK C LOC K P ULS E GEN E R A TOR FAULT 1.3 V 2.8 V R E FE R E NC E C OR E V CC 0.8V R E F OK PO W E R FA ULT 4.25 V ON 4.05 V OFF THE R MA L S HUTDOW N SET DOMINA NT 145ºC O N 135ºC O FF S VC C UV LO HIC C UP FAULT Q 0.25V VP OS 5V R S HOR TC IR C UIT DE TE C TION GND V FB INT LINE A R R E GULA TOR 200ms D elay OV E R C UR R E NT DE TE C TION R E F OK 60 mV 140K IS P 2.50 V ON 2.20 V O FF VIN UV LO C LK C LR V IN UV IN C OU NTE R IS N T HE R MA L A ND O V E R C UR R E NT P R O T E C T IO N 50K UV LO C O MP A R AT O R S Note: Note: The The SP7663 usesuses the Sipex PWM controller SP6136. SP7661 the Sipex PWM controller SP6136 Mar 1-07 Rev N SP7661 Wide Input Voltage Range 3A, 600kHz Buck Regulator  © 2007 Sipex Corporation PIN DESCRIPTION SP7661 DFN PACKAGE 7mm x 4mm (Option 2) LX 26 LX 25 LX 24 BOTTOM VIEW Heatsink Pad 1 Connect to Lx LX 23 VCC 2 2 Pin 27 1 PGND 2 PGND 3 PGND 4 PGND 5 GND UVIN 21 6 VFB GND 20 7 COMP 8 SS 9 GND GND 19 Heatsink Pad 2 Connect to GND VIN 18 Pin 28 10 ISN BST 17 LX 16 LX 15 Heatsink Pad 3 Connect to VINP Pin 29 LX 14 11 ISP 12 SWN 13 VINP Pin Pin # Name Description 1-4 PGND Ground connection for the synchronous rectifier. Ground Pin. The control circuitry of the IC and lower power driver are referenced to this 5, 9, GND 19, 20 pin. Return separately from other ground traces to the (-) terminal of Cout. 6 7 8 10 VFB Feedback Voltage and Short Circuit Detection pin. It is the inverting input of the Error Amplifier and serves as the output voltage feedback point for the Buck Converter. The output voltage is sensed and can be adjusted through an external resistor divider. Whenever VFB drops 0.25V below the positive reference, a short circuit fault is detected and the IC enters hiccup mode. Output of the Error Amplifier. It is internally connected to the inverting input of the COMP PWM comparator. An optimal filter combination is chosen and connected to this pin and either ground or VFB to stabilize the voltage mode loop. Soft Start. Connect an external capacitor between SS and GND to set the soft start SS rate based on the 10µA source current. The SS pin is held low via a 1mA (min) current during all fault conditions. ISN Current sense negative input. Rail-to-rail input for overcurrent detection. 11 ISP 12 SWN Lower supply rail for the GH high-side gate driver. Connect this pin to the switching node as close as possible to pins 23- 27. Do not connect this pin to pins 14 – 16. Input connection to the high side N-channel MOSFET. 13 VINP 14-16, 23-26 LX 17 BST 18 VIN 21 UVIN 22 VCC Mar 1-07 Rev N Current sense positive input. Rail-to-rail input for overcurrent detection. Connect an inductor between this pin and VOUT. High side driver supply pin. Connect BST to the external boost diode and capacitor as shown in the Typical Application Circuit on page 1. The high side driver is connected between BST pin and SWN pin. Vin connection for internal LDO and PWM Controller. UVLO input for Vin voltage. Connect a resistor divider between Vin and UVIN to set minimum operating voltage. Use resistor values below 20kΩ to override internal resistor divider. Output of internal regulator. May be exterinally biased if Vin < 5V. SP7661 Wide Input Voltage Range 3A, 600kHz Buck Regulator  © 2007 Sipex Corporation THEORY OF OPERATION General Overview Soft Start The SP7661 is a fixed frequency, voltage mode, synchronous PWM regulator optimized for high efficiency. The part has been specifically designed for single supply operation from a 5V to 22V input. “Soft Start” is achieved when a power converter ramps up the output voltage while controlling the magnitude of the input supply source current. In a modern step down converter, ramping up the positive terminal of the error amplifier controls soft start. As a result, excess source current can be defined as the current required to charge the output capacitor. The heart of the SP7661 is a wide bandwidth transconductance amplifier designed to accommodate Type II and Type III compensation schemes. A precision 0.8V reference, present on the positive terminal of the error amplifier, permits the programming of the output voltage down to 0.8V via the VFB pin. The output of the error amplifier, COMP, is compared to a 1.1V peak-to-peak ramp, which is responsible for trailing edge PWM control. This voltage ramp and PWM control logic are governed by the internal oscillator that accurately sets the PWM frequency to 600kHz. IVIN = Cout • (∆Vout / ∆Tsoft-start) The SP7661 provides the user with the option to program the soft start rate by tying a capacitor from the SS pin to GND. The selection of this capacitor is based on the 10µA pull up current present at the SS pin and the 0.8V reference voltage. Therefore, the excess source can be redefined as: IVIN = Cout • [∆Vout •10µA / (Css • 0.8V)] Under Voltage Lock Out (UVLO) The SP7661 contains two unique control features that are very powerful in distributed applications. First, nonsynchronous driver control is enabled during startup, to prohibit the low side switch from pulling down the output until the high side switch has attempted to turn on. Second, a 100% duty cycle timeout ensures that the low side switch is periodically enhanced during extended periods at 100% duty cycle. This guarantees the synchronized refreshing of the BST capacitor during very large duty ratios. The SP7661 has two separate UVLO comparators to monitor the bias (Vcc) and Input (Vin) voltages independently. The Vcc UVLO is internally set to 4.25V. The Vin UVLO is programmable through UVIN pin. When UVIN pin is greater than 2.5V the SP7661 is permitted to start up pending the removal of all other faults. A pair of internal resistors is connected to UVIN as shown in the figure below. SP7661 The SP7661 also contains a number of valuable protection features. Programmable VIN UVLO allows the user to set the exact value at which the conversion voltage can safely begin down-conversion, and an internal VCC UVLO which ensures that the controller itself has enough voltage to properly operate. Other protection features include thermal shutdown and short-circuit detection. In the event that either a thermal, short-circuit, or UVLO fault is detected, the SP7661 is forced into an idle state where the output drivers are held off for a finite period before a restart is attempted. Mar 1-07 Rev N VIN R6 140KΩ UVIN + 2.5V ON 2.2V OFF R7 50KΩ GND Internal and external bias of UVIN SP7661 Wide Input Voltage Range 3A, 600kHz Buck Regulator  © 2007 Sipex Corporation THEORY OF OPERATION Therefore without external biasing the Vin start threshold is 9.5V. A small capacitor may be required between UVIN and GND to filter out noise. For applications with Vin of 5V or 3.3V, connect UVIN directly to Vin. To program the Vin start threshold, use a pair of external resistors as shown. If external resistors are an order of magnitude smaller than internal resistors, then the Vin start threshold is given by: across the inductor. Over-current is detected by monitoring a differential voltage across the output inductor as shown in the next figure. SP7661 SWN L = 2.7uH, DCR = 4.1mΩ R3 5.11KΩ Vin(start) = 2.5 • (R6+R7)/R7 For example, if it is required to have a Vin start threshold of 7V, then let R7 = 5KΩ and using the Vin start threshold equation we get R6 = 9.09KΩ. VOUT R4 5.11KΩ ISP ISN CSP 6.8nF CS 0.1uF Thermal and Short-Circuit Protection Because the SP7661 is designed to drive large output current, there is a chance that the power converter will become too hot. Therefore, an internal thermal shutdown (145°C) has been included to prevent the IC from malfunctioning at extreme temperatures. Over-current detection circuit Inputs to an over-current detection comparator, set to trigger at 60 mV nominal, are connected to the inductor as shown. Since the average voltage sensed by the comparator is equal to the product of inductor current and inductor DC resistance (DCR), then Imax = 60mV / DCR. Solving this equation for the specific inductor in circuit 1, Imax = 14.6A. When Imax is reached, a 220 ms time-out is initiated, during which top and bottom drivers are turned off. Following the time-out, a restart is attempted. If the fault condition persists, then the time-out is repeated (referred to as hiccup). A short-circuit detection comparator has also been included in the SP7661 to protect against an accidental short at the output of the power converter. This comparator constantly monitors the positive and negative terminals of the error amplifier, and if the VFB pin falls more than 250mV (typical) below the positive reference, a short-circuit fault is set. Because the SS pin overrides the internal 0.8V reference during soft start, the SP7661 is capable of detecting short-circuit faults throughout the duration of soft start as well as in regular operation. Increasing the Current Limit If it is desired to set Imax > {60mV / DCR} (in this case larger than 14.6A), then a resistor R9 should be added as shown in the next figure. R9 forms a resistor divider and reduces the voltage seen by the comparator. (Imax • DCR) Since: 60mV = R9 {R3 + R4 + R9} Over-Current Protection The Over-current protection feature can only be used on output voltages ≤ 3.3 volts. It is limited by the common mode rating of the op-amp used to sense the voltage Mar 1-07 Rev N SP7661 Wide Input Voltage Range 3A, 600kHz Buck Regulator  © 2007 Sipex Corporation THEORY OF OPERATION Solving for R9 we get: R8 = R9 = [60mV • (R3 + R4)] [(Imax • DCR) – 60mV] R4 • [VOUT - 60mV + (IMAX • DCR)] 60mV - (IMAX • DCR) As an example: if desired Imax is 17A, then R9 = 63.4KΩ. As an example: for Imax of 4A and Vout of 3.3V, calculated R8 is 381kΩ. SP7661 SP7661 SWN L = 2.7uH, DCR = 4.1mΩ R3 5.11KΩ SWN L = 2.7uH, DCR = 4.1mΩ R4 5.11KΩ R3 5.11KΩ VOUT R4 5.11KΩ R9 63.4KΩ ISP ISN VOUT ISP CSP 6.8nF ISN CSP 6.8nF CS 0.1uF R8 1.5MΩ CS 0.1uF Over-current detection circuit for Imax < {60mV / DCR} Over-current detection circuit for Imax > 60mV / DCR Decreasing the Current Limit Handling of Faults If it is required to set Imax < {60mV / DCR, a resistor is added as shown in the following figure. R8 increases the net voltage detected by the current-sense comparator. Voltage at the positive and negative terminal of comparator is given by: Upon the detection of power (UVLO), thermal, or short-circuit faults, the SP7661 is forced into an idle state where the SS and COMP pins are pulled low and both switches are held off. In the event of UVLO fault, the SP7661 remains in this idle state until the UVLO fault is removed. Upon the detection of a thermal or short-circuit fault, an internal 100ms timer is activated. In the event of a short-circuit fault, a restart is attempted immediately after the 100ms timeout expires. Whereas, when a thermal fault is detected the 100ms delay continuously recycles and a restart cannot be attempted until the thermal fault is removed and the timer expires. VSP = Vout + (Imax • DCR) VSN = Vout • {R8 / (R4 +R8)} Since the comparator is triggered at 60mV: VSP-VSN = 60 mV Combining the above equations and solving for R8: Mar 1-07 Rev N SP7661 Wide Input Voltage Range 3A, 600kHz Buck Regulator  © 2007 Sipex Corporation APPLICATIONS INFORMATION Error Amplifier and Voltage Loop The second feature is a 100% duty cycle timeout that ensures synchronized refreshing of the BST capacitor at very high duty ratios. In the event that the high side NFET is on for 20 continuous clock cycles, a reset is given to the PWM flip flop half way through the 21st cycle. This forces GL to rise for the cycle, in turn refreshing the BST capacitor. The boost capacitor is used to generate a high voltage drive supply for the high side switch, which is Vcc above VIN. The heart of the SP7661 voltage error loop is a high performance, wide bandwidth transconductance amplifier. Because of the amplifier’s current limited (+/-150µA) transconductance, there are many ways to compensate the voltage loop or to control the COMP pin externally. If a simple, single-pole, single-zero response is desired, then compensation can be as simple as an RC circuit to Ground. If a more complex compensation is required, then the amplifier has enough bandwidth (45° at 4 MHz), and enough gain (60dB) to run Type III compensation schemes with adequate gain and phase margins at crossover frequencies greater than 50kHz. Power MOSFETs The SP7661 contains a pair of integrated low resistance N-channel switches designed to drive up to 3A of output current. Care should be taken to de-rate the output current based on the thermal conditions in the system such as ambient temperature, airflow and heat sinking. Maximum output current could be limited by thermal limitations of a particular application by taking advantage of the integrated-over-temperature protective scheme employed in the SP7661. The SP7661 incorporates a built-in overtemperature protection to prevent internal overheating. The common mode output of the error amplifier is 0.9V to 2.2V. Therefore, the PWM voltage ramp has been set between 1.1V and 2.2V to ensure proper 0% to 100% duty cycle capability. The voltage loop also includes two other very important features. One is a nonsynchronous startup mode. Basically, the synchronous rectifier cannot turn on unless the high side switch has attempted to turn on or the SS pin has exceeded 1.7V. This feature prevents the controller from “dragging down” the output voltage during startup or in fault modes. Setting Output Voltages The SP7661 can be set to different output voltages. The relationship in the following formula is based on a voltage divider from the output to the feedback pin VFB, which is set to an internal reference voltage of 0.80V. Standard 1% metal film resistors of surface mount size 0603 are recommended. VBST GH Voltage VSWN V(VCC) GL Voltage Vout = 0.80V [R1 / R2 + 1 ] => R2 = R1 / [ ( Vout / 0.80V ) – 1 ] 0V V(VIN) SWN Voltage Where R1 = 10KΩ and for Vout = 0.80V setting, simply remove R2 from the board. Furthermore, one could select the value of the R1 and R2 combination to meet the exact output voltage setting by restricting the R1 resistance range such that 10KΩ < R1 < 100KΩ for overall system loop stability. -0V -V(Diode) V V(VIN)+V(VCC) BST Voltage V(VCC) TIME Mar 1-07 Rev N SP7661 Wide Input Voltage Range 3A, 600kHz Buck Regulator  © 2007 Sipex Corporation APPLICATIONS INFORMATION and provide low core loss at the high switching frequency. Low cost powderediron cores have a gradual saturation characteristic but can introduce considerable AC core loss, especially when the inductor value is relatively low and the ripple current is high. Ferrite materials, although more expensive, have an abrupt saturation characteristic with the inductance dropping sharply when the peak design current is exceeded. Nevertheless, they are preferred at high switching frequencies because they present very low core loss while the designer is only required to prevent saturation. In general, ferrite or molypermalloy materials are a better choice for all but the most cost sensitive applications. Inductor Selection There are many factors to consider in selecting the inductor including core material, inductance vs. frequency, current handling capability, efficiency, size and EMI. In a typical SP7661 circuit, the inductor is chosen primarily for value, saturation current and DC resistance. Increasing the inductor value will decrease output voltage ripple, but degrade transient response. Low inductor values provide the smallest size, but cause large ripple currents, poor efficiency and require more output capacitance to smooth out the larger ripple current. The inductor must be able to handle the peak current at the switching frequency without saturating, and the copper resistance in the winding should be kept as low as possible to minimize resistive power loss. A good compromise between size, loss and cost is to set the inductor ripple current to be within 20% to 40% of the maximum output current. Optimizing Efficiency The power dissipated in the inductor is equal to the sum of the core and copper losses. To minimize copper losses, the winding resistance needs to be minimized, but this usually comes at the expense of a larger inductor. Core losses have a more significant contribution at low output current where the copper losses are at a minimum, and can typically be neglected at higher output currents where the copper losses dominate. Core loss information is usually available from the magnetics vendor. Proper inductor selection can affect the resulting power supply efficiency by more than 15%! The switching frequency and the inductor operating point determine the inductor value as follows: L = . Vout • (Vin(max) - Vout) Vin(max) • ƒs • Kr • Iout(max) where: ƒs = switching frequency Kr = ratio of the AC inductor ripple current to the maximum output current The copper loss in the inductor can be calculated using the following equation: The peak-to-peak inductor ripple current is: IPP = Vout • (Vin(max) - Vout) . Vin(max) • ƒs •L 2 PL(Cu) = I L(RMS) • Rwinding where IL(RMS) is the RMS inductor current that can be calculated as follows: Once the required inductor value is selected, the proper selection of core material is based on peak inductor current and efficiency requirements. The core must be large enough not to saturate at the peak inductor current Mar 1-07 Rev N Ipeak = Iout(max) + IL(RMS) = Iout(max) • IPP 2 SP7661 Wide Input Voltage Range 3A, 600kHz Buck Regulator 10 √ ( ) 1+1 3 . IPP 2 Iout(max) © 2007 Sipex Corporation APPLICATIONS INFORMATION Output Capacitor Selection ∆VOUT = The required ESR (Equivalent Series Resistance) and capacitance drive the selection of the type and quantity of the output capacitors. The ESR must be small enough that both the resistive voltage deviation due to a step change in the load current and the output ripple voltage do not exceed the tolerance limits expected on the output voltage. During an output load transient, the output capacitor must supply all the additional current demanded by the load until the SP7661 adjusts the inductor current to the new value. where: ƒs • COUT + (IPP •RESR) 2 D = Duty Cycle COUT = output capacitance value Input Capacitor Selection The input capacitor should be selected for ripple current rating, capacitance and voltage rating. The input capacitor must meet the ripple current requirement imposed by the switching current. In continuous conduction mode, the source current of the high-side MOSFET is approximately a square wave of duty cycle VOUT/VIN. More accurately, the current wave form is trapezoidal, given a finite turn-on and turn-off, switch transition slope. Most of this current is supplied by the input bypass capacitors. The RMS current handling capability of the input capacitors is determined at maximum output current and under the assumption that the peak-to-peak inductor ripple current is low, it is given by: The ESR of the output capacitor, combined with the inductor ripple current, is typically the main contributor to output voltage ripple. The maximum allowable ESR required to maintain a specified output voltage ripple can be calculated by: ) 2 ƒs = Switching Frequency In order to maintain VOUT,the capacitance must be large enough so that the output voltage is held up while the inductor current ramps to the value corresponding to the new load current. Additionally, the ESR in the output capacitor causes a step in the output voltage equal to the current. Because of the fast transient response and inherent 100% to 0% duty cycle capability provided by the SP7661 when exposed to output load transients, the output capacitor is typically chosen for ESR, not for capacitance value. √( IPP • (1 – D) ICIN(RMS) = Iout(max) • √D(1 - D) The worst case occurs when the duty cycle D is 50% and gives an RMS current value equal to Iout/2. Select input capacitors with adequate ripple current rating to ensure reliable operation. RESR ≤ ∆VOUT Ipk-pk The power dissipated in the input capacitor is: ∆Vout = peak-to-peak output voltage ripple Ipk-pk = peak-to-peak inductor ripple Current PCIN = I CIN(RMS) • RESR(CIN) The total output ripple is a combination of the ESR and the output capacitance value and can be calculated as follows: This can become a significant part of power losses in a converter and hurt the overall energy transfer efficiency. The input voltage ripple primarily depends on the input Mar 1-07 Rev N 2 SP7661 Wide Input Voltage Range 3A, 600kHz Buck Regulator 11 © 2007 Sipex Corporation APPLICATIONS INFORMATION Loop Compensation Design capacitor ESR and capacitance. Ignoring the inductor ripple current, the input voltage ripple can be determined by: The open loop gain of the whole system can be divided into the gain of the error amplifier, PWM modulator, buck converter output stage, and feedback resistor divider. In order to cross over at the desired frequency cut-off (fco), the gain of the error amplifier must compensate for the attenuation caused by the rest of the loop at this frequency. The goal of loop compensation is to manipulate loop frequency response such that its crossover gain at 0db, results in a slope of -20db/decade. ∆VIN = Iout(max)•RESR(CIN) + Iout(max)•Vout•(Vin - Vout) V2in • ƒs • CIN The capacitor type suitable for the output capacitors can also be used for the input capacitors. However, exercise extra caution when tantalum capacitors are used. Tantalum capacitors are known for catastrophic failure when exposed to surge current, and input capacitors are prone to such surge current when power supplies are connected “live” to low impedance power sources. Although tantalum capacitors have been successfully employed at the input, it is generally not recommended. The first step of compensation design is to pick the loop crossover frequency. High crossover frequency is desirable for fast transient response, but often jeopardizes the power supply stability. Crossover frequency should be higher than the ESR zero but less than 1/5 of the switching frequency or Type III Voltage Loop Compensation GAMP (s) Gain Block VREF (Volts) PWM Stage GPWM Gain Block VIN (SRESRCOUT+ 1) VRAMP_PP [S^2LCOUT+S(RESR+RDC) COUT+1] (SRz2Cz2+1)(SR1Cz3+1) SR1Cz2(SRz3Cz3+1)(SRz2Cp1+1) Output Stage GOUT (s) Gain Block VOUT (Volts) Notes: RESR = Output Capacitor Equivalent Series Resistance. RDC = Output Inductor DC Resistance. VRAMP_PP = SP7662 Internal Ramp Amplitude Peak-to-Peak Voltage. Voltage Feedback GFBK Gain Block Condition: Cz2 >> Cp1 & R1 >> Rz3 Output Load Resistance >> RESR & RDC R2 VFBK (Volts) (R1 + R2) or VREF VOUT SP7661 Voltage Mode Control Loop with Loop Dynamic Mar 1-07 Rev N SP7661 Wide Input Voltage Range 3A, 600kHz Buck Regulator 12 © 2007 Sipex Corporation APPLICATIONS INFORMATION 1 ƒP(LC) = 2π • √ L • Cout 120kHz. The ESR zero is contributed by the ESR associated with the output capacitors and can be determined by: ƒ = z(esr) . 1 When the output capacitors are of a Ceramic Type, the SP7661 Evaluation Board requires a Type III compensation circuit to give a phase boost of 180° in order to counteract the effects of an underdamped resonance of the output filter at the double pole frequency. 2π • Cout • Resr The next step is to calculate the complex conjugate poles contributed by the LC output filter, Gain (dB) Error Amplifier Gain Bandwidth Product Condition: C22 >> CP1, R1 >> RZ3 1/6.28 (RZ3) (CZ3) 1/6.28 (RZ2) (CP1) 1/6.28 (R1) (CZ2) 1/6.28 (R1) (CZ3) 1/6.28(R22) (CZ2) 20 Log (RZ2/R1) Frequency (Hz) Bode Plot of Type III Error Amplifier Compensation. CP1 RZ3 CZ3 CZ2 VOUT R1 68.1kΩ, 1% RSET = VFB + RSET 54.48 (kΩ) (VOUT -0.8) + - 0.8V RZ2 COMP CF1 Type III Error Amplifier Compensation Circuit Mar 1-07 Rev N SP7661 Wide Input Voltage Range 3A, 600kHz Buck Regulator 13 © 2007 Sipex Corporation APPLICATIONS INFORMATION VOUT 3.30V, 0-3A L1, Wurth -744311220 2.2uH, 14 mOhm, 7x7mm, 9A LX 1 2 3 RZ2 CZ2 6800pF 4 4.02k 5 CP1 56pF 6 7 CSS CF1 100pF 47nF 8 9 R8 NP 10 11 C9 6.8nF 12 13 27 GND PAD PGND PGND C4 LX PGND U1 LX PGND SP7661 LX GND VCC VFB UVIN COMP GND SS GND GND VIN ISN BST ISP LX SWN LX VIN R11 LX 47nF C5 26 R9 100uF 61.9k C6 C7 NP NP 25 ISP 24 RZ3 ISN 400 23 CVCC 4.7uF 22 1 R1 10k 4 GND R12 NP CZ3 21 1500pF 20 R7 NP 19 R6 NP 18 C8 NP R14 R2 NP 3160 17 16 RBST 0 Ohm SD101AWS DBST 15 14 22nF CBST 29 ISN R4 5.11K 0 Ohm LX VIN PAD 1 SWN PAD 28 3 R3 5.11K ISP LX LX VIN 12V (9.6V-22V) Rs1 Cs1 NP NP Rs2 1 Ohm Cs2 2.2nF 1 1 C3 NP C2 22uF C1 22uF 1 2 GND Evaluation Board Schematic Parts shown for 9.6V-22V input, 3.3V Output Mar 1-07 Rev N SP7661 Wide Input Voltage Range 3A, 600kHz Buck Regulator 14 © 2007 Sipex Corporation typical performance characteristics Efficiency vs. Load at 22VIN 90 90 80 80 70 Vout=5.0V 60 Vout=3.3V 70 Vout=5.0V Vout=3.3V 60 Vout=2.5V Vout=1.8V Vout=2.5V 50 Efficiency vs Load at 12VIN 00 Efficiency (%) Efficiency (%) 00 50 Vout=1.5V Vout=1.8V Vout=1.2V 40 40 0.5 1.0 1.5 2.0 2.5 0.5 3.0 1.0 Output Load (A) Efficie ncy v s. Load at 5.0VIN 95 95 90 90 85 Vout=3.3V Vout=2.5V Vout=1.8V Vout=1.5V Vout=1.2V Vout=0.8V 80 75 1.0 1.5 2.0 3.0 Vout=2.5V 80 Vout=1.8V Vout=1.5V Vout=1.2V Vout=0.8V 2.5 3.0 Output Load (A) Mar 1-07 Rev N 2.5 85 75 70 0.5 2.0 Efficiency vs. Load at 3.3VIN 00 Efficiency (%) Efficiency (%) 00 1.5 Output Load (A) 70   2 2 3 3 Output Load (A) SP7661 Wide Input Voltage Range 3A, 600kHz Buck Regulator 15 © 2007 Sipex Corporation typical performance characteristics Transient Response: CH1: Vout. CH4: Iout 1A/div. Transient Response: CH1: Vout. CH4: Iout 1A/div. Zoom showing ~1.5A/us Irate Transient Response: CH1: Vout. CH4: Iout 1A/div. Short Circuit Protection: CH1:Vout. CH2:Sstart. CH4:Iout(10A/div). Short Circuit Protection: CH1:Vout. CH2: Sstart. CH4:Input Current (5A/div). Short Circuit Protection: CH1:Vout. CH2:Sstart. CH4: Input Current (5A/div). 104ms restart rate shown . Mar 1-07 Rev N SP7661 Wide Input Voltage Range 3A, 600kHz Buck Regulator 16 © 2007 Sipex Corporation typical performance characteristics Start-up in to 3A Load: CH1:Vout. CH2:SS. CH3:Vin. CH4: Iout (2A/div). Current Limit: CH1: Vout. CH2:SoftStart. CH4 Iout (2A/div). Current is reaching approximtely 6A before a shutdown & restart Current Limit: CH1: Vout. CH2:SoftStart. CH4 Iout (2A/div). OCP Repeat rate is 107ms 12Vin Output Ripple: CH1:Vout. CH2: Switch Node taken at 3A Out. 12Vin Output Ripple: CH1: Vout. CH2: Switch Node taken at 0A Out. 22Vin Output Ripple: CH1: Vout. CH2: Switch Node taken at 0A out. Mar 1-07 Rev N SP7661 Wide Input Voltage Range 3A, 600kHz Buck Regulator 17 © 2007 Sipex Corporation typical performance characteristics Operation from exernal bias: CH2: Switchnode @ 2.34Vin. 22Vin Transient Response CH1: Vout. CH4: Iout (1A/div). Operation from exernal bias: CH2: Switchnode @ 3Vin, Iout = 3A. Mar 1-07 Rev N SP7661 Wide Input Voltage Range 3A, 600kHz Buck Regulator 18 © 2007 Sipex Corporation Package: 26 Pin dfn Mar 1-07 Rev N SP7661 Wide Input Voltage Range 3A, 600kHz Buck Regulator 19 © 2007 Sipex Corporation ORDERING INFORMATION Part Number Junction Temperature Package SP7661ER/TR..................................-40°C to +125°C............ .......................................26 Pin 7 X 4 DFN (Option 2) SP7661ER-L/TR...............................-40°C to +125°C......................... (Lead Free) 26 Pin 7 X 4 DFN (Option 2) /TR = Tape and Reel Pack quantity is 3,000 26 pin DFN. Sipex Corporation Headquarters and Sales Office 233 South Hillview Drive Milpitas, CA 95035 TEL: (408) 934-7500 FAX: (408) 935-7600 Sipex Corporation reserves the right to make changes to any products described herein. Sipex does not assume any liability arising out of the application or use of any product or circuit described herein; neither does it convey any license under its patent rights nor the rights of others. Mar 1-07 Rev N SP7661 Wide Input Voltage Range 3A, 600kHz Buck Regulator 20 © 2007 Sipex Corporation
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