TS1002IM8TP

TS1002/TS1004 THE ONLY 0.8V/0.6µA RAIL-TO-RAIL DUAL/QUAD OP AMPS DESCRIPTION FEATURES Single 0.65V to 2.5V Operation Supply current: 0.6μA per amplifier (typ) Offset voltage: 0.5mV (typ) Low TCVOS: 10µV/°C (typ) AVOL Driving 100kΩ Load: 90dB (min) Unity Gain Stable Rail-to-rail Input and Output No Output Phase Reversal Packaging: TS1002 – 8-pin MSOP TS1004 – 14-pin TSSOP The TS1002 and the TS1004 are the industry’s first and only dual and quad single-supply, precision CMOS operational amplifiers fully specified to operate at 0.8V while consuming less than 0.6µA supply current per amplifier. Optimized for ultra-long-life battery-powered applications, the TS1002 and the TS1004 join Touchstone’s TS1001 operational amplifier in the “NanoWatt Analog™” highperformance analog integrated circuits portfolio. Both op amps exhibit a typical offset voltage of 0.5mV, a typical input bias current of 25pA, and rail-to-rail input and output stages. The TS1002 and the TS1004 can operate from single-supply voltages from 0.65V to 2.5V. The TS1002/TS1004’s combined features make either an excellent choice in applications where very low supply current and low operating supply voltage translate into very long equipment operating time. Applications include: nanopower active filters, wireless remote sensors, battery and powerline current sensors, portable gas monitors, and handheld/portable POS terminals. The TS1002 and the TS1004 are fully specified at VDD = 0.8V and over the industrial temperature range (−40°C to +85°C). The TS1002 is available in PCB-space saving 8-lead MSOP surface-mount packages. The TS1004 is available in a 14-pin TSSOP package. APPLICATIONS Battery/Solar-Powered Instrumentation Portable Gas Monitors Low-voltage Signal Processing Nanopower Active Filters W ireless Remote Sensors Battery-powered Industrial Sensors Active RFID Readers Powerline or Battery Current Sensing Handheld/Portable POS Terminals TYPICAL APPLICATION CIRCUIT Supply Current Distribution 30% VDD = 0.8V 25% Percent of Units - % 20% 15% 10% 5% A NanoWatt 2-Pole Sallen-Key Low-Pass Filter Patent(s) Pending NanoWatt Analog and the Touchstone Semiconductor logo are registered trademarks of Touchstone Semiconductor, Incorporated. 0% 0.53 0.58 0.63 0.68 0.73 Supply Current per Amplifier - µA Page 1 © 2012 Touchstone Semiconductor, Inc. All rights reserved. TS1002/TS1004 ABSOLUTE MAXIMUM RATINGS Total Supply Voltage (VDD to VSS) ........................... +2.75 V Voltage Inputs (IN+, IN-) ........... (VSS - 0.3V) to (VDD + 0.3V) Differential Input Voltage .......................................... ±2.75 V Input Current (IN+, IN-) .............................................. 20 mA Output Short-Circuit Duration to GND .................... Indefinite Continuous Power Dissipation (TA = +70°C) 8-Pin MSOP (Derate 7mW/°C above +70°C) ...... 450 mW 14-pin TSSOP (Derate 8.3mW/°C above +70°C) ............................................................................ 500 mW Operating Temperature Range .................... -40°C to +85°C Junction Temperature .............................................. +150°C Storage Temperature Range ..................... -65°C to +150°C Lead Temperature (soldering, 10s) ............................. +300° Electrical and thermal stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only and functional operation of the device at these or any other condition beyond those indicated in the operational sections of the specifications is not implied. Exposure to any absolute maximum rating conditions for extended periods may affect device reliab ility and lifetime. PACKAGE/ORDERING INFORMATION TAPE & REEL PART PACKAGE TAPE & REEL PART PACKAGE ORDER NUMBER MARKING QUANTITY ORDER NUMBER MARKING QUANTITY TS1002IM8TP TADJ TS1002IM8T 3000 TS1004IT14T ----TS1004IT14TP T1004I 3000 ----- Lead-free Program: Touchstone Semiconductor supplies only lead-free packaging. Consult Touchstone Semiconductor for products specified with wider operating temperature ranges. Page 2 TS1002_4DS r1p0 RTFDS TS1002/TS1004 ELECTRICAL CHARACTERISTICS VDD = +0.8V, VSS = 0V, VINCM = VSS; RL = 100kΩ to (VDD-VSS)/2; TA = -40°C to +85°C, unless otherwise noted. Typical values are at TA = +25°C. See Note 1 Parameters Supply Voltage Range Supply Current Symbol VDD-VSS ISY TS1004; RL = Open circuit Input Offset Voltage Input Offset Voltage Drift Input Bias Current Input Offset Current Input Voltage Range Common-Mode Rejection Ratio Power Supply Rejection Ratio Output Voltage High VOS TCVOS IIN+, IINIOS IVR CMRR PSRR VOH VIN+, VIN- = (VDD - VSS)/2 TA = 25°C -40°C ≤ TA ≤ 85°C TA = 25°C Specified as IIN+ - IINVIN+, VIN- = (VDD - VSS)/2 -40°C ≤ TA ≤ 85°C Guaranteed by Input Offset Voltage Test 0V ≤ VIN(CM) ≤ 0.4V 0.65V ≤ (VDD - VSS) ≤ 2.5V TA = 25°C Specified as VDD - VOUT, RL = 100kΩ to VSS -40°C ≤ TA ≤ 85°C TA = 25°C Specified as VDD - VOUT, RL = 10kΩ to VSS -40°C ≤ TA ≤ 85°C TA = 25°C Specified as VOUT - VSS, RL = 100kΩ to VDD -40°C ≤ TA ≤ 85°C TA = 25°C Specified as VOUT - VSS, RL = 10kΩ to VDD -40°C ≤ TA ≤ 85°C TA = 25°C VOUT = VSS -40°C ≤ TA ≤ 85°C TA = 25°C VOUT = VDD -40°C ≤ TA ≤ 85°C TA = 25°C VSS+50mV ≤ VOUT ≤ VDD-50mV -40°C ≤ TA ≤ 85°C RL = 100kΩ to VSS, CL = 20pF Unity-gain Crossover, RL = 100kΩ to VSS, CL = 20pF RL = 100kΩ to VSS, AVCL = +1V/V FPBW = SR/(π • VOUT,PP); VOUT,PP = 0.7VPP f = 1kHz f = 1kHz VIN = VSS or VDD Conditions TS1002; RL = Open circuit TA = 25°C -40°C ≤ TA ≤ 85°C TA = 25°C -40°C ≤ TA ≤ 85°C TA = 25°C -40°C ≤ TA ≤ 85°C Min 0.65 Typ 0.8 1.2 2.4 0.5 10 0.025 20 0.01 VSS 50 50 2 VDD 74 74 1.2 10 0.4 5 0.5 0.3 4.5 3 90 85 1.5 11 104 4 70 1.5 680 0.6 10 mA Max 2.5 1.6 2 3.2 4 3 5 Units V µA µA mV µV/°C nA nA V dB dB mV Output Voltage Low VOL 2 2.5 16 20 0.6 1 7 10 mV ISC+ Short-circuit Current ISCOpen-loop Voltage Gain Gain-Bandwidth Product Phase Margin Slew Rate Full-power Bandwidth Input Voltage Noise Density Input Current Noise Density AVOL GBWP φM SR FPBW en in dB kHz degrees V/ms Hz µV/ pA/ Note 1: All specifications are 100% tested at TA = +25°C. Specification limits over temperature (TA = TMIN to TMAX) are guaranteed by device characterization, not production tested. TS1002_4DS r1p0 Page 3 RTFDS TS1002/TS1004 TYPICAL PERFORMANCE CHARACTERISTICS Total Supply Current vs Supply Voltage 2.8 +85°C, TS1004 Total Supply Current vs Input Common-Mode Voltage 2 TS1004 SUPPLY CURENT - µA 1.78 TA = +25°C SUPPLY CURENT - µA 2.4 1.9 +25°C, TS1004 -40°C, TS1004 1.56 1.5 +85°C, TS1002 1.34 1 -40°C, TS1002 +25°C, TS1002 1.12 TS1002 0.9 0.6 0.65 1.11 1.58 2.04 2.5 0 0.2 0.4 0.6 0.8 SUPPLY VOLTAGE - Volt INPUT COMMON-MODE VOLTAGE - Volt Total Supply Current vs Input Common-Mode Voltage 2 INPUT OFFSET VOLTAGE - mV TS1004 SUPPLY CURENT - µA 1.78 TA = +25°C 0.65 Input Offset Voltage vs Supply Voltage TA = +25°C VINCM = VDD 0.6 1.56 0.55 1.34 1.12 TS1002 0.9 0 0.5 1 1.5 2 2.5 INPUT COMMON-MODE VOLTAGE - Volt 0.55 VINCM = 0V 0.5 0.5 1 1.5 2 2.5 SUPPLY VOLTAGE - Volt Input Offset Voltage vs Input Common-Mode Voltage 1 INPUT OFFSET VOLTAGE - mV INPUT OFFSET VOLTAGE - mV VDD =0.8V TA = +25°C 0.5 Input Offset Voltage vs Input Common-Mode Voltage 1 VDD = 2.5V TA = +25°C 0.5 0 0 -0.5 -0.5 -1 0 0.2 0.4 0.6 0.8 -1 0 0.5 1 1.5 2 2.5 INPUT COMMON-MODE VOLTAGE - Volt INPUT COMMON-MODE VOLTAGE - Volt Page 4 TS1002_4DS r1p0 RTFDS TS1002/TS1004 TYPICAL PERFORMANCE CHARACTERISTICS Input Bias Current (IIN+, IIN-) vs Input Common-Mode Voltage 100 INPUT BIAS CURRENT - pA 75 50 25 0 -25 -50 0 0.2 0.4 0.6 0.8 0 0.5 1 1.5 2 2.5 INPUT COMMON-MODE VOLTAGE - Volt INPUT COMMON-MODE VOLTAGE - Volt VDD =0.8V TA = +25°C Input Bias Current (IIN+, IIN-) vs Input Common-Mode Voltage 250 INPUT BIAS CURRENT - pA 200 150 100 50 0 -50 VDD = 2.5V TA = +25°C OUTPUT SATURATION VOLTAGE - mV 4 3.5 3 2.5 2 1.5 1 0.5 0 -40 RL = 100kΩ VDD = 2.5V OUTPUT SATURATION VOLTAGE - mV Output Voltage High (VOH) vs Temperature, RLOAD =100kΩ 4.5 Output Voltage Low (VOL) vs Temperature, RLOAD =100kΩ 1.8 1.6 1.4 1.2 1 0.8 0.6 0.4 0.2 0 -40 RL = 100kΩ +25 TEMPERATURE - °C +85 VDD = 0.8V VDD = 2.5V VDD = 0.8V +25 TEMPERATURE - °C +85 Output Voltage High (VOH) vs Temperature, RLOAD =10kΩ OUTPUT SATURATION VOLTAGE - mV RL = 10kΩ VDD = 2.5V OUTPUT SATURATION VOLTAGE - mV 35 30 25 20 15 VDD = 0.8V 10 5 0 -40 +25 TEMPERATURE - °C +85 Output Voltage Low (VOL) vs Temperature, RLOAD =10kΩ 20 16 VDD = 2.5V 12 8 VDD = 0.8V 4 RL = 10kΩ 0 -40 +25 TEMPERATURE - °C +85 TS1002_4DS r1p0 Page 5 RTFDS TS1002/TS1004 TYPICAL PERFORMANCE CHARACTERISTICS OUTPUT SHORT-CIRCUIT CURRENT - mA VOUT = 0V 20 VDD = 2.5V OUTPUT SHORT-CIRCUIT CURRENT - mA Output Short Circuit Current, ISC+ vs Temperature 25 Output Short Circuit Current, ISC- vs Temperature 70 VOUT = VDD 60 VDD = 2.5V 50 40 30 20 VDD = 0.8V 10 0 15 10 5 VDD = 0.8V 0 -40 +25 TEMPERATURE - °C +85 -40 +25 TEMPERATURE - °C +85 Gain and Phase vs. Frequency 60 PHASE 70° GAIN GAIN - dB 20 VDD = 0.8V TA = +25°C RL = 100kΩ CL = 20pF AVCL = 1000 V/V 10 100 1k -50 VOUT(N) - 100µV/DIV PHASE - Degrees 40 50 150 0.1Hz to 10Hz Output Voltage Noise 130µVPP 4kHz -150 0 -20 10k FREQUENCY - Hz -250 100k 1 Second/DIV Large-Signal Transient Response VDD = 2.5V, VSS = GND, RLOAD = 100kΩ, CLOAD = 15pF Small-Signal Transient Response VDD = 2.5V, VSS = GND, RLOAD = 100kΩ, CLOAD = 15pF INPUT OUTPUT OUTPUT INPUT 200µs/DIV 2ms/DIV Page 6 TS1002_4DS r1p0 RTFDS TS1002/TS1004 PIN FUNCTIONS TS1002 MSOP 1, 7 4 3, 5 2, 6 8 Pin TS1004 TSSOP 1, 7, 8, 14 7 3, 5, 10, 12 2, 6, 9, 13 14 Label OUT VSS +IN -IN VDD Function Amplifier Outputs: A, B – TS1002; A, B, C, D – TS1004 Negative Supply or Analog GND. If applying a negative voltage to this pin, connect a 0.1µF capacitor from this pin to analog GND. Amplifier Non-inverting Inputs: A, B – TS1002; A, B, C, D – TS1004 Amplifier Inverting Inputs: A, B – TS1002; A, B, C, D – TS1004 Positive Supply Connection. Connect a 0.1µF bypass capacitor from this pin to analog GND. THEORY OF OPERATION The TS1002 and the TS1004 are fully functional for input signals from the negative supply (VSS or GND) to the positive supply (VDD). Their input stages consist of two differential amplifiers, a p-channel CMOS stage and an n-channel CMOS stage that are active over different ranges of the input common mode voltage. The p-channel input pair is active for input common mode voltages, VINCM, between the negative supply to approximately 0.4V below the positive supply. As the common-mode input voltage moves closer towards VDD, an internal current mirror activates the n-channel input pair differential pair. The p-channel input pair becomes inactive for the balance of the input common mode voltage range up to the positive supply. Because both input stages have their own offset voltage (VOS) characteristic, the offset voltage of these amplifiers is a function of the applied input common-mode voltage, VINCM. The VOS has a crossover point at ~0.4V from VDD (Refer to the VOS vs. VCM curve in the Typical Operating Characteristics section). Caution should be taken in applications where the input signal amplitude is comparable to the amplifiers’ VOS value and/or the design requires high accuracy. In these situations, it is necessary for the input signal to avoid the crossover point. In addition, amplifier parameters such as PSRR and CMRR which involve the input offset voltage will also be affected by changes in the input common-mode voltage across the differential pair transition region. The amplifiers’ second stage is a folded-cascode transistor arrangement that converts the input stage differential signals into a single-ended output. A complementary drive generator supplies current to the output transistors that swing rail to rail. The amplifiers’ output stage voltage swings within 1.2mV from the rails at 0.8V supply when driving an output load of 100kΩ - which provides the maximum possible dynamic range at the output. This is particularly important when operating on low supply voltages. When driving a stiffer 10k Ω load, the amplifiers’ output swings within 10mV of VDD and within 5mV of VSS (or GND). APPLICATIONS INFORMATION Portable Gas Detection Sensor Amplifier Gas sensors are used in many different industrial and medical applications. Gas sensors generate a current that is proportional to the percentage of a particular gas concentration sensed in an air sample. This output current flows through a load resistor and the resultant voltage drop is amplified. Depending on the sensed gas and sensitivity of the sensor, the output current can be in the range of tens of microamperes to a few milliamperes. Gas sensor datasheets often specify a recommended load resistor value or a range of load resistors from which to choose. There are two main applications for oxygen sensors – applications which sense oxygen when it is abundantly present (that is, in air or near an oxygen tank) and those which detect traces of oxygen in parts-per-million concentration. In medical applications, oxygen sensors are used when air quality or oxygen delivered to a patient needs to be monitored. In fresh air, the concentration of oxygen TS1002_4DS r1p0 Page 7 RTFDS TS1002/TS1004 is 20.9% and air samples containing less than 18% oxygen are considered dangerous. In industrial applications, oxygen sensors are used to detect the absence of oxygen; for example, vacuum -packaging of food products is one example. The circuit in Figure 1 illustrates a typical implementation used to amplify the output of an oxygen detector. Either amplifier makes an Figure 2: A Simple, Single-pole Active Low-Pass Filter. If additional attenuation is needed, a two-pole Sallen-Key filter can be used to provide the additional attenuation as shown in Figure 3. For best results, the filter’s cutoff frequency should be 8 to 10 times lower than the amplfier’s crossover Figure 1: A Nanopower, Precision Oxygen Gas Sensor Amplifier. excellent choice for this application as it only draws 0.6µA of supply current per amplifier and operates on supply voltages down to 0.65V. With the components shown in the figure, the circuit consumes less than 0.7 μA of supply current ensuring that small form -factor single- or button-cell batteries (exhibiting low mAh charge ratings) could last beyond the operating life of the oxygen sensor. The precision specifications of these amplifiers, such as their low offset voltage, low TCVOS, low input bias current, high CMRR, and high PSRR are other factors which make these amplifiers excellent choices for this application. Since oxygen sensors typically exhibit an operating life of one to two years, an oxygen sensor amplifier built around one of these amplifiers can operate from a conventionallyavailable single 1.5-V alkaline AA battery for over 290 years! At such low power consumption from a single cell, the oxygen sensor could be replaced over 150 times before the battery requires replacing! NanoWatt, Buffered Single-pole Low-Pass Filters W hen receiving low-level signals, limiting the bandwidth of the incoming signals into the system is often required. As shown in Figure 2, the simplest way to achieve this objective is to use an RC filter at the noninverting terminal of the amplifier. Figure 3: A Nanopower 2-Pole Sallen-Key Low-Pass Filter. frequency. Additional operational amplifier phase margin shift can be avoided if the amplifier bandwidth-to-signal bandwidth ratio is greater than 8. The design equations for the 2-pole Sallen-Key lowpass filter are given below with component values selected to set a 400Hz low-pass filter cutoff frequency: R1 = R2 = R = 1MΩ C1 = C2 = C = 400pF Q = Filter Peaking Factor = 1 f–3dB = 1/(2 x π x RC) = 400 Hz R3 = R4/(2-1/Q); with Q = 1, R3 = R4. A Single +1.5 V Supply, Two Op Amp Instrumentation Amplifier The amplifiers’ ultra-low supply current and ultra-low voltage operation make them ideal for batterypowered applications such as the instrumentation amplifier shown in Figure 4 using a TS1002. Page 8 TS1002_4DS r1p0 RTFDS TS1002/TS1004 response is observed as there will appear noticeable peaking/ringing in the output transient response. If any amplifier is used in an application that requires driving larger capacitive loads, an isolation resistor between the output and the capacitive load should be used as illustrated in Figure 5. Figure 4: A Two Op Amp Instrumentation Amplifier. The circuit utilizes the classic two op amp instrumentation amplifier topology with four resistors to set the gain. The equation is simply that of a noninverting amplifier as shown in the figure. The two resistors labeled R1 should be closely matched to each other as well as both resistors labeled R2 to ensure acceptable common-mode rejection performance. Resistor networks ensure the closest matching as well as matched drifts for good temperature stability. Capacitor C1 is included to limit the bandwidth and, therefore, the noise in sensitive applications. The value of this capacitor should be adjusted depending on the desired closed-loop bandwidth of the instrumentation amplifier. The RC combination creates a pole at a frequency equal to 1/(2 π × R1C1). If the AC-CMRR is critical, then a matched capacitor to C1 should be included across the second resistor labeled R1. Because these amplifiers accept rail-to-rail inputs, their input common mode range includes both ground and the positive supply of 1.5V. Furthermore, their rail-to-rail output range ensures the widest signal range possible and maximizes the dynamic range of the system. Also, with their low supply current of 0.6μA per amplifier, this circuit consumes a quiescent current of only ~1.3μA, yet it still exhibits a 1-kHz bandwidth at a circuit gain of 2. Driving Capacitive Loads W hile the amplifiers’ internal gain-bandwidth product is 4kHz, both are capable of driving capacitive loads up to 50pF in voltage follower configurations without any additional components. In many applications, however, an operational amplifier is required to drive much larger capacitive loads. The amplifier’s output impedance and a large capacitive load create additional phase lag that further reduces the amplifier’s phase margin. If enough phase delay is introduced, the amplifier’s phase margin is reduced. The effect is quite evident when the transient Figure 5: Using an External Resistor to Isolate a CLOAD from the Amplifer’s Output. Table 1 illustrates a range of RISO values as a function of the external CLOAD on the output of these amplifiers. The power supply voltage applied on the these amplifiers at which these resistor values were determined empirically was 1.8V. The oscilloscope capture shown in Figure 6 illustrates a typical transient response obtained with a C LOAD = 500pF and an RISO = 50kΩ. Note that as CLOAD is increased a smaller RISO is needed for optimal transient response. External Capacitive Load, CLOAD 0-50pF 100pF 500pF 1nF 5nF 10nF External Output Isolation Resistor, RISO Not Required 120kΩ 50kΩ 33kΩ 18kΩ 13kΩ TS1002_4DS r1p0 Page 9 RTFDS TS1002/TS1004 In the event that an external RLOAD in parallel with CLOAD appears in the application, the use of an R ISO results in gain accuracy loss because the external series RISO forms a voltage-divider with the external load resistor RLOAD. Figure 8: Analog Comparator Hysteresis Band and Output Switching Points. VIN VOUT (VREF) for the circuit at ½ the supply voltage, or 0.75V, while keeping the current drawn by this resistor divider low. Capacitor C1 is used to filter any extraneous noise that could couple into the amplifer’s inverting input. In this application, the desired hysteresis band was set to 100mV (VHYB) with a desired high trip-point (VHI) set at 1V and a desired low trip-point (VLO) set at 0.9V. Figure 6: TS1002/TS1004 Transient Response for RISO = 50kΩ and CLOAD = 500pF. Configuring the TS1002 or the TS1004 into a Nanowatt Analog Comparator Although optimized for use as an operational amplifier, these amplifiers can also be used as a railto-rail I/O comparator as illustrated in Figure 7. Since these amplifers draw very little supply current (0.6µA per amplifier, typical), it is desired that the design of an analog comparator using these amplfiers should also use as little current as practical. The first step in the design, therefore, was to set the feedback resistor R3: R3 = 10MΩ Calculating a value for R1 is given by the following expression: R1 = R3 x (VHYB/VDD) Substituting VHYB = 100mV, VDD = 1.5V, and R3 = 10MΩ into the equation above yields: R1 = 667kΩ The following expression was then used to calculate a value for R2: R2 = 1/[VHI/(VREF x R1) – (1/R1) – (1/R3)] Substituting VHI = 1V, VREF = 0.75V, R1 = 667kΩ, and R3 = 10MΩ into the above expression yields: R2 = 2.5MΩ Figure 7: A NanoWatt Analog Comparator with UserProgrammable Hysteresis. External hysteresis can be employed to minimize the risk of output oscillation. The positive feedback circuit causes the input threshold to change when the output voltage changes state. The diagram in Figure 8 illustrates the amplifiers’ analog comparator hysteresis band and output transfer characteristic. The design of an analog comparator using the TS1002 or the TS1004 is straightforward. In this application, a 1.5-V power supply (VDD) was used and the resistor divider network formed by RD1 and RD2 generated a convenient reference voltage Page 10 TS1002_4DS r1p0 RTFDS TS1002/TS1004 Printed Circuit Board Layout Considerations Even though these amplifiers operate from a single 0.65V to 2.5V power supply and consume very little supply current, it is always good engineering practice to bypass the power supplies pins with a 0.1μF ceramic capacitor placed in close proximity to the VDD and VSS (or GND) pins. Good pcb layout techniques and analog ground plane management improve the performance of any analog circuit by decreasing the amount of stray capacitance that could be introduced at the op amp's inputs and outputs. Excess stray capacitance can easily couple noise into the input leads of the op amp and excess stray capacitance at the output will add to any external capacitive load. Therefore, PC board trace lengths and external component leads should be kept a short as practical to any of the amplifiers’ package pins. Second, it is also good engineering practice to route/remove any analog ground plane from the inputs and the output pins of these amplifiers. TS1002_4DS r1p0 Page 11 RTFDS TS1002/TS1004 PACKAGE OUTLINE DRAWING 8-Pin MSOP Package Outline Drawing (N.B., Drawings are not to scale) 3.10 Max 2.90 Min 0.65 REF 8 0.38 Max 0.28 Min 0.127 0.27 REF 3.10 Max 2.90 Min 5.08 Max 4.67 Min 0.23 Max 0.13 Min GAUGE PLANE 0' -- 6' 1 2 0.25 0.38 Max 0.28 Min 0.70 Max 0.40 Min DETAIL “A” DETAIL ‘A’ 1.10 Max 0.95 Max 0.75 Min SEATING PLANE 0.10 Max 0.15 Max 0.05 Min 0.23 max 0.13 Min NOTE: 1. PACKAGE LENGTH DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS. 2. PACKAGE WIDTH DOES NOT INCLUDE INTERLEAD FLASH OR PROTUSIONS. 3. CONTROLLING DIMENSION IN MILIMETERS. 4. THIS PART IS COMPLIANT WITH JEDEC MO-187 VARIATIONS AA 5. LEAD SPAN/STAND OFF HEIGHT/COPLANARITY ARE CONSIDERED AS SPECIAL CHARACTERISTIC. Page 12 TS1002_4DS r1p0 RTFDS TS1002/TS1004 PACKAGE OUTLINE DRAWING 14-Pin TSSOP Package Outline Drawing (N.B., Drawings are not to scale) 5.10 Max 4.90 Min ( D) N 0.65 REF GAUGE PLANE 0.25 R = R’ = 0.09 Min 6.60 Max 6.20 Min 4.5 Max 4.3 Min (E) R’ SEATING PLANE 0' – 8' Pin 1 ID MARK 0.75 Max 0.45 Min 1 2 0.30 Max 0.19 Min DETAIL ‘A’ 1.05 Max 0.80 Min 1.20 Max 12' TYP 1.00 REF ALL SIDES SEATING PLANE 0.15 Max O.05 Min 0.20 Max 0.09 Min 0.30 Max 0.19 Min 0.16 Max 0.09 Min 0.25 Max 0.19 Min 0.20 Max 0.09 Min 0.10 Max DETAIL ‘A’ NOTE: 1. “D” AND “E” ARE REFERENCE DATUMS AND DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS. MOLD FLASH, PROTRUSIONS OR GATE BURRS SHALL NOT EXCEED 0.15 mm PER SIDE. 2. “N” IS THE NUMBER OF TERMINAL POSITIONS. 3. CONTROLLING DIMENSION IN MILIMETERS AND ANGLES IN DEGREES. 4. THIS PART IS COMPLIANT WITH JEDEC SPECIFICATION MO-153 AB-1 5. LEAD SPAN/STAND OFF HEIGHT/COPLANARITY ARE CONSIDERED AS SPECIAL CHARACTERISTIC. Information furnished by Touchstone Semiconductor is believed to be accurate and reliable. However, Touchstone Semiconductor does not assume any responsibility for its use nor for any infringements of patents or other rights of third parties that may result from its use , and all information provided by Touchstone Semiconductor and its suppliers is provided on an AS IS basis, WITHOUT WARRANTY OF ANY KIN D. Touchstone Semiconductor reserves the right to change product specifications and product descriptions at any time without any advance notice. No license is granted by implication or otherwise under any patent or patent rights of Touchstone Semiconductor. Touchstone Semiconductor assumes no liability for applications assistance or customer product design. Customers are responsible for thei r products and applications using Touchstone Semiconductor components. To minimize the risk associated with customer products and applications, customers should provide adequate design and operating safeguards. Trademarks and registered trademarks are the property of t heir respective owners. Touchstone Semiconductor, Inc. 630 Alder Drive, Milpitas, CA 95035 +1 (408) 215 - 1220 ▪ www.touchstonesemi.com Page 13 TS1002_4DS r1p0 RTFDS