HV9912
Switch-Mode LED Driver IC with High Current Accuracy
and Hiccup Mode Protection
Features
General Description
• Switch-mode Controller for Single-switch Drivers:
- Buck
- Boost
- Buck-boost
- SEPIC
• Works with High-side Current Sensors
• Closed-loop Control of Output Current
• High Pulse-Width Modulation (PWM) Dimming
Ratio
• Internal 90V Linear Regulator (can be extended
using external Zener Diodes)
• Internal 2% Voltage Reference (0°C < TA < 85°C)
• Constant Frequency or Constant Off-time
Operation
• Programmable Slope Compensation
• Linear and PWM Dimming
• +0.2A/–0.4A Gate Driver
• Hiccup Mode Protection for both Short-circuit and
Open-circuit Conditions
• Output Overvoltage Protection
• Synchronization Capability
• Pin Compatible with HV9911
HV9912 is an LED driver IC designed to control
single-switch PWM converters (buck, boost,
buck-boost and SEPIC) in a Constant Frequency or
Constant Off-time mode. The controller uses a peak
Current Mode control scheme with programmable
slope compensation and includes an internal
transconductance amplifier to control the output current
in closed loop, enabling high output current accuracy.
In the case of buck and buck-boost converters, the
output current can be sensed using a high-side current
sensor like the HV7800. In the Constant Frequency
mode, multiple HV9912 ICs can be synchronized with
each other or with an external clock, using the SYNC
pin. Programmable MOSFET current limit enables
current limiting during Input Undervoltage and Output
Overload conditions. The IC also includes a 0.2A
source and 0.4A sink gate driver that makes the
HV9912 suitable for high-power applications. An
internal 90V linear regulator powers the IC, eliminating
the need for a separate power supply for the IC. The IC
also provides a FAULT output, which can be used to
disconnect the LEDs in case of a Fault condition using
an external disconnect FET. HV9912 also provides a
TTL-compatible, low-frequency PWM dimming input
that can accept an external control signal with a duty
ratio of 0-100% and a frequency of up to a few kilohertz.
The HV9912 includes hiccup protection from both short
and open circuits, with automatic recovery after the
Fault condition is cleared.
Applications
• RGB Backlight Applications
• General LED Lighting Applications
• Battery-powered LED Lamps
The HV9912 is a pin-compatible replacement for
HV9911. It can be used with existing HV9911 designs,
which have input voltages of less than 90V, by
changing ROVP1, ROVP, and RT.
Package Type
16-lead SOIC
(Top View)
See Table 2-1 for pin information.
2016 Microchip Technology Inc.
VIN
1
16
FDBK
VDD
2
15
IREF
GATE
3
14
COMP
GND
4
13
PWMD
CS
5
12
OVP
SC
6
11
FAULT
RT
7
10
REF
SYNC
8
9
CLIM
DS20005583A-page 1
HV9912
Functional Block Diagram
Linear Regulator
VIN
VDD
5.60/6.10V
CLIM
Vbg
+
-
REF
SS
+
POR
GATE
Blanking
TBLANK
CS
1:2
+
-
+
-
ramp
R
Q
S
SC
POR
OVD
SCD
-
Hiccup/Dimming
Block
PWMD
FDBK
IREF
5V rising
4.5V falling
OVPD
13R
COMP
FAULT
SS
+
OVP
R
TBLANK,SC
SS
+
-
GM
+
SCD
One Shot
SYNC
RT
2
PWMD
PWMD
DS20005583A-page 2
GND
2016 Microchip Technology Inc.
HV9912
Typical Application Circuit
L1
D2
VIN
D1
Q1
CIN
1
VIN
GATE 3
CDD
RSC
2
VDD
4
GND
ROVP1
CS
5
RCS
ROVP2
CO
OVP 12
HV9912
RSLOPE
6
SC
FAULT 11
7
RT
FDBK 16
Q2
RT
CC
CREF
10 REF
COMP 14
CLIM
PWMD 13
15 IREF
SYNC 8
9
RS
RL1
RL2
2016 Microchip Technology Inc.
RR1
RR2
DS20005583A-page 3
HV9912
1.0
ELECTRICAL CHARACTERISTICS
Absolute Maximum Ratings †
VIN to GND ............................................................................................................................................... –0.5 to +100V
VDD to GND............................................................................................................................................–0.3V to +13.5V
CS to GND ........................................................................................................................................ –0.3V to VDD+0.3V
PWMD to GND .................................................................................................................................. –0.3V to VDD+0.3V
GATE to GND.................................................................................................................................... –0.3V to VDD+0.3V
All Other Pins to GND ....................................................................................................................... –0.3V to VDD+0.3V
Continuous Power Dissipation (TA= +25°C)..................................................................................................... 1200 mW
Operating Junction Temperature Range .............................................................................................. –40°C to +125°C
Storage Temperature Range ................................................................................................................ –65°C to +150°C
† Notice: Stresses above those listed under “Absolute Maximum Ratings” may cause permanent damage to the
device. This is a stress rating only, and functional operation of the device at those or any other conditions above those
indicated in the operational sections of this specification is not intended. Exposure to maximum rating conditions for
extended periods may affect device reliability.
ELECTRICAL CHARACTERISTICS
Electrical Specifications: TA = 25°C and VIN = 12V unless otherwise specified.
Parameters
Sym.
Min.
Typ.
Max.
Units
Conditions
Input DC Supply Voltage Range
VINDC
Note 1
—
90
V
DC input voltage (Note 2)
Shutdown Mode Supply Current
IINSD
—
—
1.5
mA
PWMD connected to GND
(Note 2)
VDD
7.25
7.75
8.25
V
VIN = 9V–90V; PWMD connected to GND (Note 2)
VDD Undervoltage Lockout
Threshold
UVLORISE
6.5
—
7
V
VDD rising
VDD Undervoltage Lockout
Hysteresis
UVLOHYST
—
500
—
mV
VDD falling
INPUT
INTERNAL REGULATOR
Internally Regulated Voltage
REFERENCE
1.225
REF Pin Voltage
Note 1:
2:
3:
1.285
VREF
V
1.225
Line Regulation of Reference Voltage
1.25
VREFLINE
0
1.25
—
REF bypassed with a 0.1 µF
capacitor to GND; IREF = 0;
PWMD = GND;
0°C < TA < +85°C
1.29
REF bypassed with a 0.1 µF
capacitor to GND; IREF = 0;
PWMD = GND;
–40°C < TA < 125°C
20
REF bypassed with a 0.1 µF
capacitor to GND; IREF = 0;
VDD = 7.25V–12V;
PWMD = GND
mV
See Section 3.3 “Minimum Input Voltage at VIN Pin” for the minimum input voltage.
The specifications which apply over the full operating temperature range at
–40°C < TA < +85°C are guaranteed by design and characterization.
For design guidance only
DS20005583A-page 4
2016 Microchip Technology Inc.
HV9912
ELECTRICAL CHARACTERISTICS (CONTINUED)
Electrical Specifications: TA = 25°C and VIN = 12V unless otherwise specified.
Parameters
Load Regulation of Reference
Voltage
Sym.
Min.
Typ.
Max.
Units
Conditions
REF bypassed with a 0.1 µF
capacitor to GND;
IREF = 0 µA–500 µA;
PWMD = GND
VREFLOAD
0
—
10
mV
PWMD Input Low Voltage
VPWMD(LO)
—
—
0.8
V
Note 2
PWMD Input High Voltage
VPWMD(HI)
2
—
—
V
Note 2
RPWMD
50
100
150
kΩ
VPWMD = 5V
PWM DIMMING
PWMD Pull-down Resistance
GATE
GATE Short-circuit Current
ISOURCE
0.2
—
—
A
VGATE = 0V
GATE Sinking Current
ISINK
0.4
—
—
A
VGATE = VDD
GATE Output Rise Time
TRISE
—
50
85
ns
CGATE = 1 nF
GATE Output Fall Time
TFALL
—
25
45
ns
CGATE = 1 nF
VOVP,RISING
4.75
5
5.25
V
OVP rising
VOVP,HYST
—
0.5
—
V
OVP falling
100
—
280
100
—
330
OVERVOLTAGE PROTECTION
Overvoltage Rising Trip Point
Overvoltage Hysteresis
CURRENT SENSE
ns
0°C < TA < +85°C
Leading Edge Blanking
TBLANK
Delay to Output of COMP Comparator
TDELAY1
—
—
200
ns
COMP = VDD; CLIM = REF;
CSENSE = 0 mV to 600 mV
(step up)
Delay to Output of CLIMIT Comparator
TDELAY2
—
—
200
ns
COMP = VDD; CLIM = 300 mV;
CSENSE = 0 mV to 400 mV
(step up)
Comparator Offset Voltage
VOFFSET
–10
—
10
mV
GBW
—
1
—
MHz
AV
60
—
—
dB
Output open
VCM
–0.3
—
3
V
Note 3
VO
0.7
—
6.75
V
Note 3
Transconductance
gM
450
550
650
µA/V
Input Offset Voltage
VOFFSET
–5
—
5
mV
IBIAS
—
0.5
1
nA
Note 3
fOSC1
99
106
118
kHz
RT = 500 kΩ (Note 2)
fOSC2
510
580
650
kHz
RT = 96 kΩ (Note 2)
DMAX
87
—
93
%
–40°C < TA < +125°C
INTERNAL TRANSCONDUCTANCE OPAMP
Gain Bandwidth Product
Open-loop DC Gain
Input Common Mode Range
Output Voltage Range
Input Bias Current
75 pF capacitance at OP pin
(Note 3)
OSCILLATOR
Oscillator Frequency
Maximum Duty Cycle
Note 1:
2:
3:
See Section 3.3 “Minimum Input Voltage at VIN Pin” for the minimum input voltage.
The specifications which apply over the full operating temperature range at
–40°C < TA < +85°C are guaranteed by design and characterization.
For design guidance only
2016 Microchip Technology Inc.
DS20005583A-page 5
HV9912
ELECTRICAL CHARACTERISTICS (CONTINUED)
Electrical Specifications: TA = 25°C and VIN = 12V unless otherwise specified.
Parameters
Sym.
Min.
Typ.
Max.
Units
Conditions
SYNC Input High
VSYNCH
2
—
—
V
SYNC Input Low
VSYNCL
—
—
0.8
V
IOUTSYNC
—
18
—
µA
GSC
1.9
2
2.1
V
0.125
—
0.25
0.125
—
0.26
–40°C < TA < +125°C;
IREF = GND
SYNC Output Current
OUTPUT SHORT-CIRCUIT
Gain for Short-circuit Comparator
Minimum Output Voltage of the Gain
Stage
Propagation Time for Short-circuit
Detection
VOMIN
V
0°C < TA < +85°C;
IREF = GND
TOFF
—
—
250
ns
PWMD = VDD; IREF = 400 mA;
FDBK step from
0 mV to 900 mV; FAULT goes
from high to low
Fault Output Rise Time
TRISE,FAULT
—
—
300
ns
330 pF capacitor at FAULT pin
Fault Output Fall Time
TFALL,FAULT
—
—
300
ns
330 pF capacitor at FAULT pin
Blanking Time
TBLANK,SC
480
—
900
ns
IHICCUP
—
5
—
µA
Current Sourced Out of SC Pin
ISLOPE
0
—
100
µA
Internal Current Mirror Ratio
GSLOPE
1.8
2
2.26
Current Source at COMP Pin used for
Hiccup Mode Protection
SLOPE COMPENSATION
Note 1:
2:
3:
Note 2
ISLOPE = 50 µA;
RSC = 1 kΩ
See Section 3.3 “Minimum Input Voltage at VIN Pin” for the minimum input voltage.
The specifications which apply over the full operating temperature range at
–40°C < TA < +85°C are guaranteed by design and characterization.
For design guidance only
TEMPERATURE SPECIFICATIONS
Parameters
Sym.
Min.
Typ.
Max.
Units
Operating Junction Temperature
TJ
–40
—
+125
°C
Storage Temperature
Ts
–65
—
+150
°C
JA
—
83
—
°C/W
Conditions
TEMPERATURE RANGES
PACKAGE THERMAL RESISTANCE
16-lead SOIC
DS20005583A-page 6
2016 Microchip Technology Inc.
HV9912
2.0
PIN DESCRIPTION
Table 2-1 shows the pin description details of HV9912.
TABLE 2-1:
PIN DESCRIPTION TABLE
Pin Number
Name
1
VIN
This pin is the input of a 90V high-voltage regulator.
2
VDD
This is a power supply pin for all internal circuits. It must be bypassed with a
low-ESR capacitor to GND (at least 0.1 µF).
3
GATE
This pin is the output gate driver for an external N-channel power MOSFET.
4
GND
This is the ground return for all the low-power analog internal circuitry. This pin must
be connected to the return path from the input.
5
CS
This pin is used to sense the source current of the external power FET. It includes a
built-in 100 ns (minimum) blanking time.
6
SC
This pin is used to set the slope compensation.
7
RT
This pin sets the frequency of the power circuit. A resistor between RT and GND will
program the circuit in Constant Frequency mode.
8
SYNC
This I/O pin may be connected to the SYNC pin of other HV9912 circuits and will
cause the oscillators to lock to the highest frequency oscillator.
9
CLIM
This pin provides a programmable input current limit for the converter. The current
limit can be set using a resistor divider from the REF pin.
10
REF
This pin provides 2% accurate reference voltage. It must be bypassed with a
0.01 μF–0.1 μF capacitor to GND.
11
FAULT
This pin is pulled to ground when there is an Output Short-circuit condition or Output
Overvoltage condition. This pin can be used to drive an external MOSFET (in the
case of boost converters) to disconnect the load from the source.
12
OVP
This pin provides the overvoltage protection for the converter. When the voltage at
this pin exceeds 5V, the GATE output of the HV9912 is turned off, and the FAULT
goes low. The IC will turn on when the voltage at the pin goes below 4.5V.
13
PWMD
When this pin is pulled to GND (or left open), switching of the HV9912 is disabled.
When an external TTL high level is applied to it, switching will resume.
14
COMP
Stable Closed-loop control can be accomplished by connecting a compensation network between COMP and GND. This capacitor also controls the hiccup time.
15
IREF
The voltage at this pin sets the output current level. The current reference can be set
using a resistor divider from the REF pin.
16
FDBK
This pin provides output current feedback to the HV9912 by using a current sense
resistor.
2016 Microchip Technology Inc.
Description
DS20005583A-page 7
HV9912
3.0
DETAILED DESCRIPTION
3.1
Power Topology
The HV9912 is a Switch-mode converter LED driver
designed to control a Continuous Conduction mode
buck or boost in a Constant Frequency or Constant
Off-time mode. The IC includes an internal linear
regulator, which operates from input voltages up to
90V, eliminating the need for an external power supply
for the IC. The IC includes features typically required in
LED drivers, such as open LED protection, output
short-circuit protection, linear and PWM dimming,
programmable input current limiting and accurate
control of the LED current. A high-current gate drive
output enables the controller to be used in high-power
converters.
The HV9912 is an enhanced version of the HV9911
with hysteretic overvoltage protection and Hiccup
mode short-circuit protection. The IC includes a
blanking network controlled by the PWMD input to
prevent the short-circuit protection from triggering
prematurely during PWM dimming due to the parasitic
capacitance of the LED string. It also allows the IREF
pin to be pulled all the way down to GND without
triggering the short-circuit protection. It is a
pin-compatible replacement for the HV9911.
3.2
3.3
Minimum Input Voltage at VIN Pin
The minimum input voltage at which the converter will
start and stop depends on the minimum voltage drop
required for the linear regulator. The internal linear
regulator will control the voltage at the VDD pin when
VIN is between 9V and 90V. However, when VIN is less
than 9V, the converter will still function as long as VDD
is greater than the undervoltage lockout. Thus, the
converter might be able to start at input voltages lower
than 9V. The start/stop voltages at the VIN pin can be
determined using the minimum voltage drop across the
linear regulator as a function of the current drawn. This
data is shown in Figure 3-1 for ambient temperatures of
25°C and 85°C.
Assume an ambient temperature of 85°C. Provided
that the IC is driving a 15 nC gate charge FET at
200 kHz, the total input current is estimated to be
4.5 mA when Equation 3-1 is used. At this input
current, the minimum voltage drop from Figure 3-1
would be around VDROP = 1.25V. However, before the
IC starts switching, the current drawn would have been
1.5 mA. At this current level, the voltage drop would be
approximately VDROP1 = 0.3V. Thus, the start/stop VIN
voltages could be computed as demonstrated in
Equation 3-2 and Equation 3-3 below:
EQUATION 3-2:
Linear Regulator
The HV9912 can be powered directly from its VIN pin
that withstands a voltage of up to 90V. When a voltage
is applied to the VIN pin, the HV9912 tries to maintain a
constant 7.75V (typical) at the VDD pin. The regulator
also has a built-in undervoltage lockout which shuts off
the IC if the voltage at the VDD pin falls below the UVLO
threshold.
V IN START = UVLO MAX + V DROP1
= 7V + 0.3V
= 7.3V
EQUATION 3-3:
V IN STOP = UVLO MAX – UVLO + V DROP
= 7V – 0.5V + 1.25V
= 7.75V
The VDD pin must be bypassed by a low-ESR capacitor
(≥0.1 µF) to provide a low-impedance path for the
high-frequency current of the output gate driver.
EQUATION 3-1:
I IN = 1.5mA + Q G f S
In the above equation, fS is the switching frequency,
and QG is the external FET’s gate charge, which can be
obtained from the data sheet of the FET.
Minimum Voltage Drop vs. I IN
3
Minimum Voltage Drop (V)
The input current drawn from the VIN pin is the sum of
the 1.5 mA current drawn by the internal circuit and the
current drawn by the gate driver, which in turn depends
on the switching frequency and the gate charge of the
external FET. See Equation 3-1.
2.5
2
TA = 85OC
1.5
TA = 25OC
1
0.5
0
0
2
4
6
8
10
I IN (mA)
FIGURE 3-1:
Headroom vs. Input Current.
In this case, the gate driver draws too much current and
VINSTART is less than VINSTOP. When this happens, the
IC will oscillate between ON and OFF if the input
DS20005583A-page 8
2016 Microchip Technology Inc.
HV9912
voltage is between the start and stop voltages.
Therefore, it is recommended that the input voltage be
kept higher than VINSTOP.
3.4
Reference
HV9912 includes a 2% accurate 1.25V reference,
which can be used as the reference for the output
current as well as to set the switch current limit. The
reference is buffered so that it can deliver a maximum
of 500 µA external current to drive the external circuitry.
The reference should be bypassed with at least a 10 nF
low-ESR capacitor.
Note:
3.5
To avoid abnormal Startup conditions, the
bypass capacitor at the REF pin should
not exceed 0.1 µF.
3.7
For Continuous Conduction mode converters operating
in the Constant Frequency mode, slope compensation
becomes necessary to ensure stability of the Peak
Current mode controller if the operating duty cycle is
greater than 50%. Choosing a slope compensation
which is one half of the down slope of the inductor
current ensures that the converter will be stable for all
duty cycles.
Slope compensation can be programmed by two
resistors RSLOPE and RSC. Assuming a down slope of
DS (A/µs) for the inductor current, the slope
compensation resistors can be computed as illustrated
in Equation 3-5.
EQUATION 3-5:
Oscillator
Connecting the resistor between RT and GND will
program the time period.
In both cases, resistor RT sets the current, which
charges an internal oscillator capacitor. The capacitor
voltage ramps up linearly. When the voltage increases
beyond the internal set voltage, a comparator triggers
the set input of the internal SR flip-flop. This starts the
next switching cycle. The time period of the oscillator
can be computed as shown in Equation 3-4.
6
R SLOPE DS 10 T S R CS
R SC = -------------------------------------------------------------------------10
Where RCS is the current sense resistor which
senses the switching FET current
Note:
EQUATION 3-4:
T S R T 18pF
3.6
Synchronization
The SYNC pin is an input/output (I/O) port to a
fault-tolerant peer-to-peer and/or master clock
synchronization circuit. For synchronization, the SYNC
pins of multiple HV9912-based converters can be
connected together and may also be connected to the
open drain output of a master clock. When connected
in this manner, the oscillators will lock to the device with
the highest operating frequency. When synchronizing
multiple ICs, it is recommended that the same timing
resistor (corresponding to the switching frequency) be
used in all the HV9912 circuits.
On rare occasions, given the length of the connecting
lines for the SYNC pins, a resistor between SYNC and
GND may be required to damp any ringing due to
parasitic capacitances. It is recommended that the
resistor chosen be greater than 300 kΩ.
When synchronized in this manner, a permanent High
or Low condition on the SYNC pin will result in a loss of
synchronization, but the HV9912-based converters will
continue to operate at their individually set operating
frequencies. Since loss of synchronization will not
result in total system failure, the SYNC pin is
considered fault tolerant.
2016 Microchip Technology Inc.
Slope Compensation
3.8
The maximum current that can be sourced
out of the SC pin is 100 µA. This limits the
minimum value of the RSLOPE resistor to
25 kΩ. If the equation for slope
compensation produces a RSLOPE less
than this value, then RSC would have to be
reduced accordingly. It is recommended
that RSLOPE be chosen within the range of
25 kΩ to 50 kΩ.
Current Sense
The current sense input of the HV9912 includes a
built-in 100 ns (minimum) blanking time to prevent
spurious turn-off due to the initial current spike when
the FET turns on.
The HV9912 includes two high-speed comparators—
one is used during normal operation and the other is
used to limit the maximum input current during Input
Undervoltage or Overload conditions.
The IC includes an internal resistor divider network,
which steps down the voltage at the COMP pin by a
factor of 15. This stepped-down voltage is given to one
of the comparators as the current reference. The
reference to the other comparator, which acts to limit
the maximum inductor current, is given externally.
It is recommended that the sense resistor RCS be
chosen so as to provide about 250 mV current sense
signal.
DS20005583A-page 9
HV9912
3.9
Current Limit
Current limit has to be set by a resistor divider from the
1.25V reference available on the IC. Assuming a
maximum operating inductor current Ipk (including
ripple current), the maximum voltage at the CLIM pin
can be set as shown in Equation 3-6.
EQUATION 3-6:
V CLIM 1.2 I PK R CS + 5 R CS R SLOPE 0.9
Note that this equation assumes a current limit at 120%
of the maximum input current. Also, if VCLIM is greater
than 450 mV, the saturation of the internal opamp will
determine the limit on the input current rather than the
CLIM pin. In such a case, the sense resistor RCS should
be reduced until VCLIM reduces below 550 mV.
It is recommended that no capacitor be connected
between CLIM and GND.
3.10
Internal 1 MHz Transconductance
Amplifier
capacitor maintains the voltage across it. The GATE is
disabled, so the converter stops switching and the
FAULT pin goes low, turning off the disconnect switch.
The output capacitor of the converter determines the
converter’s PWM dimming response because the
capacitor has to get charged and discharged whenever
the PWMD signal goes high or low. In the case of a
buck converter, since the inductor current is
continuous, a very small capacitor is used across the
LEDs. This minimizes the effect of the capacitor on the
converter’s PWM dimming response. However, in the
case of a boost converter, the output current is
discontinuous, and a very large output capacitor is
required to reduce the ripple in the LED current. Thus,
this capacitor will have a significant impact on the PWM
dimming response. By turning off the disconnect switch
when PWMD goes low, the output capacitor is
prevented from being discharged. This dramatically
improves the boost converter’s PWM dimming
response.
Note:
HV9912 includes a built-in 1 MHz transconductance
amplifier with tri-state output, which can be used to
close the feedback loop. The output current sense
signal is connected to the FDBK pin and the current
reference is connected to the IREF pin.
The output of the opamp is controlled by the signal
applied to the PWMD pin. When PWMD is high, the
output of the opamp is connected to the COMP pin.
When PWMD is low, the output is left open. This
enables the integrating capacitor to hold the charge
when the PWMD signal has turned off the gate drive.
When the IC is enabled, the voltage on the integrating
capacitor will force the converter into Steady state
almost instantaneously.
The output of the opamp is buffered and connected to
the current sense comparator using a 15:1 divider. The
buffer helps to prevent the integrator capacitor from
discharging during the PWM Dimming state.
3.11
PWM Dimming
PWM dimming can be achieved by driving the PWMD
pin with a TTL-compatible square wave source. The
PWM signal is connected internally to three different
nodes—the transconductance amplifier, the FAULT
output and the GATE output.
When the PWMD signal is high, the GATE and FAULT
pins are enabled and the transconductance opamp’s
output is connected to the external compensation
network. Thus, the internal amplifier controls the output
current. When the PWMD signal goes low, the output of
the transconductance amplifier is disconnected from
the compensation network. Therefore the integrating
DS20005583A-page 10
3.12
In case of Continuous Conduction mode
boost converters, disconnecting the
capacitor might cause a sudden spike in
the capacitor voltage as the energy in the
inductor is dumped into the capacitor. This
increase in the capacitor voltage might
cause the OVP comparator to trip if the
OVP point is set too close to the maximum
operating voltage. Thus, either the capacitor has to be larger to absorb this energy
without increasing the capacitor voltage
significantly or the OVP set point has to be
increased.
False Triggering of the
Short-Circuit Comparator During
PWM Dimming
During PWM dimming, the parasitic capacitance of the
LED string causes a spike in the output current when
the disconnect FET is turned on. With the HV9911, this
parasitic spike in the output current makes the IC
falsely detect an Overcurrent condition and shut down.
To prevent this false shutdown, an R-C filter is used at
the FDBK pin to filter this spike.
To prevent false triggering in the HV9912, there is a
built-in 500 ns blanking network for the short-circuit
comparator, which eliminates the need for the external
R-C low-pass filter. This blanking network is activated
when the PWMD input goes high. Thus, the
short-circuit comparator will not see the spike in the
LED current during the PWM Dimming turn-on
transition. Once the blanking timer is completed, the
short-circuit comparator will start monitoring the output
current. Thus, the total delay time for detecting a
short-circuit will depend on the condition of the PWMD
input.
2016 Microchip Technology Inc.
HV9912
If the output short-circuit exists before the PWMD
signal goes high, the total detection can be computed
as shown in Equation 3-7:
EQUATION 3-7:
This equation assumes that the voltage drop across RZ
can be neglected compared to the voltage swing at the
COMP pin, which is true in most cases (RZ < 100 kΩ).
The POR time (tPOR) for the HV9912 is 10 μs.
t detect = t blank SC max + t delay max 900 + 250
VIN
1150ns max
If the short-circuit occurs when the PWMD signal is
already high, the time to detect is determined through
Equation 3-8:
POR
EQUATION 3-8:
t detect1 = t delay max 250ns max
COMP
Pull-up
with 5.0µA
Pull-down
with 5.0µA
Gm control
5.0V
3.13
Hiccup Timer
HV9912 reuses the compensation network on the
COMP pin to create a timer which is activated upon
startup or when a detected Fault has been cleared.
When a Fault is detected (either open-circuit or
short-circuit) or upon startup, the COMP pin is
disconnected from the gM amplifier and the GATE and
FAULT pins are pulled low, disabling the LED driver.
When the Fault has cleared, a 5 µA current source is
activated which pulls the COMP network up to 5V.
Once the voltage at the COMP network reaches 5V, the
5 µA sourcing current is disconnected and a 5 µA
sinking current is activated which pulls the COMP pin
low. When the voltage at the COMP pin reaches 1V, the
sinking current is disconnected and the gM amplifier is
reconnected to the COMP pin. The FAULT pin goes
high and the GATE pin would be allowed to switch. The
closed-loop control then takes over the control of the
LED current.
3.14
Startup Condition
The startup waveforms are shown in Figure 3-2.
Assuming a pole-zero R-C network at the COMP pin
(series combination of RZ and CZ in parallel with CC),
the start-up delay time can be approximately computed
as shown in Equation 3-9.
EQUATION 3-9:
9V
t delay t POR + C C + C Z ---------5A
1.0V
tPOR
FLT
tDELAY
FIGURE 3-2:
3.15
Waveforms during Startup.
Fault Condition
In the case of a Fault condition (either open-circuit or
short-circuit), the same sequence is repeated, and the
only difference is that the COMP pin voltage does not
start from zero but from its Steady-state condition.
3.16
Short-Circuit Protection
When a Short-circuit condition is detected (output
current becomes higher than twice the Steady-state
current), the GATE and FAULT outputs are pulled low.
As soon as the disconnect FET is turned off, the output
current goes to zero and the Short-circuit condition
disappears. At this time, the hiccup timer is started.
(See Figure 3-3.) Once the timing is complete, the
converter attempts to restart. If the Fault condition still
persists, the converter shuts down and goes through
the cycle again. If the Fault condition is cleared due to
a momentary output short, the converter will start
regulating the output current normally. This allows the
LED driver to recover from accidental shorts without
having to reset the IC.
The hiccup time will depend on the Steady-state
voltage of the COMP pin (VCOMP). This is typically in
the range of 3V–4V. The hiccup time can be
approximately computed with Equation 3-10.
2016 Microchip Technology Inc.
DS20005583A-page 11
HV9912
EQUATION 3-10:
9V – V COMP
t HICCUP C C + C Z ------------------------------5A
the LED string voltage, at which point no Fault will be
detected and the normal operation of the circuit will
commence. (See Figure 3-4.)
The various delay times can be determined as shown
in Equation 3-11, Equation 3-12 and Equation 3-13:
EQUATION 3-11:
Output Current
Short Circuit
Occurs
t RC 0.1 R OVP1 + R OVP2 C O
Normal Operation
Resumes
EQUATION 3-12:
9V – V COMP
t HICCUP1 C C + C Z ------------------------------5A
FLT
Hiccup Time
EQUATION 3-13:
9V
t HICCUP2 – n C C + C Z ---------5A
COMP
5.0V
1.0V
Note:
The number of hiccup cycles might be
more than two.
tHICCUP
FIGURE 3-3:
3.17
Short-circuit Protection.
Overvoltage Protection
The HV9912 provides hysteretic overvoltage
protection, allowing the IC to recover in case the LED
load is disconnected momentarily.
When the load is disconnected in a boost converter, the
output voltage rises as the output capacitor starts
charging. When the output voltage reaches the OVP
rising threshold, the HV9912 detects an Overvoltage
condition and turns off the converter. The converter is
turned back on only when the output voltage falls below
the falling OVP threshold, which is 10% lower than the
rising threshold. This time is mostly dictated by the R-C
time constant of the output capacitor CO and the
resistor network used to sense overvoltage (ROVP1 +
ROVP2). In case of a persistent Open-circuit condition,
this cycle keeps repeating, maintaining the output
voltage within a 10% band.
In most designs, the lower threshold voltage of the
overvoltage protection is more than the LED string
voltage when the converter is turned on. Thus, when
the LED load is reconnected to the output of the
converter, the voltage differential between the actual
output voltage and the LED string voltage will cause a
spike in the output current when the FAULT signal goes
high. This causes a short-circuit to be detected and the
HV9912 will go into short-circuit protection. This
continues until the output voltage becomes lower than
DS20005583A-page 12
3.18
Linear Dimming
Linear dimming can be achieved by varying the voltage
at the IREF pin because the output current is
proportional to the voltage at the pin. This can be done
either by using a potentiometer from the IREF pin or
applying an external voltage source to the pin.
In the HV9911, due to the offset voltage of the
short-circuit comparator as well as the non-linearity of
the X2 gain stage, pulling the IREF pin very close to
GND will cause the internal short-circuit comparator to
trigger and shut down the IC.
To overcome this in the HV9912, the minimum output
of the gain stage is limited to 125 ~ 250mV, allowing the
IREF pin to be pulled all the way to 0V without triggering
the short-circuit comparator.
Note:
Since this control IC is a Peak Current
mode controller, pulling the IREF pin to
zero will not cause the LED current to
become zero. The converter will still be
operating at its minimum on time, causing
a very small current to flow through the
LEDs. To get zero LED current, the
PWMD input has to be pulled to GND.
2016 Microchip Technology Inc.
HV9912
Output Cap
Voltage 100V
OVP ON
LED string reconnects
LED string voltage
90V
80V
OVP OFF
LED string disconnects
FLT
tRC
tHICCUP1
tHICCUP2
Output Current
COMP
5V
1V
FIGURE 3-4:
Open-circuit Protection.
2016 Microchip Technology Inc.
DS20005583A-page 13
HV9912
4.0
PACKAGING INFORMATION
4.1
Package Marking Information
16-lead SOIC
XXXXXXXXX e3
YYWWNNN
Legend: XX...X
Y
YY
WW
NNN
e3
*
Note:
DS20005583A-page 14
Example
HV9912NG e3
1612389
Product Code or Customer-specific information
Year code (last digit of calendar year)
Year code (last 2 digits of calendar year)
Week code (week of January 1 is week ‘01’)
Alphanumeric traceability code
Pb-free JEDEC® designator for Matte Tin (Sn)
This package is Pb-free. The Pb-free JEDEC designator ( e3 )
can be found on the outer packaging for this package.
In the event the full Microchip part number cannot be marked on one line, it will
be carried over to the next line, thus limiting the number of available
characters for product code or customer-specific information. Package may or
not include the corporate logo.
2016 Microchip Technology Inc.
HV9912
16-Lead SOIC (Narrow Body) Package Outline (NG)
9.90x3.90mm body, 1.75mm height (max), 1.27mm pitch
D
16
θ1
E1 E
Note 1
(Index Area
D/2 x E1/2)
L2
1
L
Top View
View B
View
B
A
h
A A2
h
Seating
Plane
e
A1
Seating
Plane
θ
L1
Gauge
Plane
Note 1
b
Side View
View A-A
A
Note: For the most current package drawings, see the Microchip Packaging Specification at www.microchip.com/packaging.
Note:
1. 7KLVFKDPIHUIHDWXUHLVRSWLRQDO,ILWLVQRWSUHVHQWWKHQD3LQLGHQWL¿HUPXVWEHORFDWHGLQWKHLQGH[DUHDLQGLFDWHG7KH3LQLGHQWL¿HUFDQEH
DPROGHGPDUNLGHQWL¿HUDQHPEHGGHGPHWDOPDUNHURUDSULQWHGLQGLFDWRU
Symbol
MIN
Dimension
NOM
(mm)
MAX
A
A1
A2
b
D
1.35*
0.10
1.25
0.31
9.80*
-
-
-
-
1.75
0.25
1.65*
0.51
9.90
E
E1
e
5.80* 3.80*
6.00
3.90
10.00* 6.20* 4.00*
1.27
BSC
h
L
0.25
0.40
-
-
0.50
1.27
L1
L2
1.04 0.25
REF BSC
ș
ș
0O
5O
-
-
8O
15O
JEDEC Registration MS-012, Variation AC, Issue E, Sept. 2005.
7KLVGLPHQVLRQLVQRWVSHFL¿HGLQWKH-('(&GUDZLQJ
Drawings are not to scale.
2016 Microchip Technology Inc.
DS20005583A-page 15
HV9912
NOTES:
DS20005583A-page 16
2016 Microchip Technology Inc.
HV9912
APPENDIX A:
REVISION HISTORY
Revision A (July 2016)
• Converted Supertex Doc# DSFP-HV9912 to
Microchip DS20005583A.
• Made minor text changes throughout the document.
DS20005583A-page 17
2016 Microchip Technology Inc.
HV9912
PRODUCT IDENTIFICATION SYSTEM
To order or obtain information, e.g., on pricing or delivery, contact your local Microchip representative or sales office.
XX
PART NO.
-
Package
Options
Device
X
-
Environmental
X
Media Type
Device:
HV9912
=
Switch-Mode LED Driver IC with High
Current Accuracy and Hiccup Mode
Protection
Package:
NG
=
16-lead SOIC
Environmental:
G
=
Lead (Pb)-free/RoHS-compliant Package
Media Types:
(blank)
=
45/Tube for an NG Package
M901
=
2600/Reel for an NG Package
M934
=
2600/Reel for an NG Package
Examples:
a)
HV9912NG-G:
Switch-Mode LED Driver IC with
High Current Accuracy and Hiccup Mode Protection, 16-lead
SOIC Package, 45/Tube
b) HV9912NG-G-M901: Switch-Mode LED Driver IC with
High Current Accuracy and Hiccup Mode Protection, 16-lead
SOIC Package, 2600/Reel
c) HV9912NG-G-M934: Switch-Mode LED Driver IC with
High Current Accuracy and Hiccup Mode Protection, 16-lead
SOIC Package, 2600/Reel
Note: For media types M901 and M934, the base quantity for tape and reel
was standardized to 2600/reel. Both options will result in delivery of the
same number of parts/reel.
2016 Microchip Technology Inc.
DS20005583A-page 18
Note the following details of the code protection feature on Microchip devices:
•
Microchip products meet the specification contained in their particular Microchip Data Sheet.
•
Microchip believes that its family of products is one of the most secure families of its kind on the market today, when used in the
intended manner and under normal conditions.
•
There are dishonest and possibly illegal methods used to breach the code protection feature. All of these methods, to our
knowledge, require using the Microchip products in a manner outside the operating specifications contained in Microchip’s Data
Sheets. Most likely, the person doing so is engaged in theft of intellectual property.
•
Microchip is willing to work with the customer who is concerned about the integrity of their code.
•
Neither Microchip nor any other semiconductor manufacturer can guarantee the security of their code. Code protection does not
mean that we are guaranteeing the product as “unbreakable.”
Code protection is constantly evolving. We at Microchip are committed to continuously improving the code protection features of our
products. Attempts to break Microchip’s code protection feature may be a violation of the Digital Millennium Copyright Act. If such acts
allow unauthorized access to your software or other copyrighted work, you may have a right to sue for relief under that Act.
Information contained in this publication regarding device
applications and the like is provided only for your convenience
and may be superseded by updates. It is your responsibility to
ensure that your application meets with your specifications.
MICROCHIP MAKES NO REPRESENTATIONS OR
WARRANTIES OF ANY KIND WHETHER EXPRESS OR
IMPLIED, WRITTEN OR ORAL, STATUTORY OR
OTHERWISE, RELATED TO THE INFORMATION,
INCLUDING BUT NOT LIMITED TO ITS CONDITION,
QUALITY, PERFORMANCE, MERCHANTABILITY OR
FITNESS FOR PURPOSE. Microchip disclaims all liability
arising from this information and its use. Use of Microchip
devices in life support and/or safety applications is entirely at
the buyer’s risk, and the buyer agrees to defend, indemnify and
hold harmless Microchip from any and all damages, claims,
suits, or expenses resulting from such use. No licenses are
conveyed, implicitly or otherwise, under any Microchip
intellectual property rights unless otherwise stated.
Microchip received ISO/TS-16949:2009 certification for its worldwide
headquarters, design and wafer fabrication facilities in Chandler and
Tempe, Arizona; Gresham, Oregon and design centers in California
and India. The Company’s quality system processes and procedures
are for its PIC® MCUs and dsPIC® DSCs, KEELOQ® code hopping
devices, Serial EEPROMs, microperipherals, nonvolatile memory and
analog products. In addition, Microchip’s quality system for the design
and manufacture of development systems is ISO 9001:2000 certified.
QUALITY MANAGEMENT SYSTEM
CERTIFIED BY DNV
== ISO/TS 16949 ==
2016 Microchip Technology Inc.
Trademarks
The Microchip name and logo, the Microchip logo, AnyRate,
dsPIC, FlashFlex, flexPWR, Heldo, JukeBlox, KeeLoq,
KeeLoq logo, Kleer, LANCheck, LINK MD, MediaLB, MOST,
MOST logo, MPLAB, OptoLyzer, PIC, PICSTART, PIC32 logo,
RightTouch, SpyNIC, SST, SST Logo, SuperFlash and UNI/O
are registered trademarks of Microchip Technology
Incorporated in the U.S.A. and other countries.
ClockWorks, The Embedded Control Solutions Company,
ETHERSYNCH, Hyper Speed Control, HyperLight Load,
IntelliMOS, mTouch, Precision Edge, and QUIET-WIRE are
registered trademarks of Microchip Technology Incorporated
in the U.S.A.
Analog-for-the-Digital Age, Any Capacitor, AnyIn, AnyOut,
BodyCom, chipKIT, chipKIT logo, CodeGuard, dsPICDEM,
dsPICDEM.net, Dynamic Average Matching, DAM, ECAN,
EtherGREEN, In-Circuit Serial Programming, ICSP, Inter-Chip
Connectivity, JitterBlocker, KleerNet, KleerNet logo, MiWi,
motorBench, MPASM, MPF, MPLAB Certified logo, MPLIB,
MPLINK, MultiTRAK, NetDetach, Omniscient Code
Generation, PICDEM, PICDEM.net, PICkit, PICtail,
PureSilicon, RightTouch logo, REAL ICE, Ripple Blocker,
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Endurance, TSHARC, USBCheck, VariSense, ViewSpan,
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SQTP is a service mark of Microchip Technology Incorporated
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GestIC is a registered trademarks of Microchip Technology
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All other trademarks mentioned herein are property of their
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© 2016, Microchip Technology Incorporated, Printed in the
U.S.A., All Rights Reserved.
ISBN: 978-1-5224-0798-0
DS20005583A-page 19
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2016 Microchip Technology Inc.