MCP654-E/ST

MCP651/1S/2/3/4/5/9 50 MHz, 200 µV Op Amps with mCal Features: Description: • • • • • • The Microchip Technology Inc. MCP651/1S/2/3/4/5/9 family of high bandwidth and high slew rate operational amplifiers features low offset. At power-up, these op amps are self-calibrated using mCal. Some package options also provide a Calibration/Chip Select pin (CAL/CS) that supports a Low-Power mode of operation, with offset calibration at the time normal operation is re-started. These amplifiers are optimized for high speed, low noise and distortion, single-supply operation with rail-to-rail output and an input that includes the negative rail. • • • • • • Gain-Bandwidth Product: 50 MHz Slew Rate: 30 V/µs Low Input Offset: ±200 µV (maximum) Low Input Bias Current: 6 pA (typical) Noise: 7.5 nV/Hz, at 1 MHz Ease-of-Use: - Unity-Gain Stable - Rail-to-Rail Output - Input Range incl. Negative Rail - No Phase Reversal Supply Voltage Range: +2.5V to +5.5V High Output Current: ±100 mA Supply Current: 6.0 mA/Ch (typical) Low-Power Mode: 5 µA/Ch Small Packages: SOT23-5, DFN Extended Temperature Range: -40°C to +125°C This family is offered in single (MCP651 and MCP651S), single with CAL/CS pin (MCP653), dual (MCP652), dual with CAL/CS pins (MCP655), quad (MCP654) and quad with CAL/CS pins (MCP659). All devices are fully specified from -40°C to +125°C. Typical Application Circuit Driving A/D Converters Fast Low-side Current Sensing Power Amplifier Control Loops Optical Detector Amplifier Barcode Scanners Multi-Pole Active Filter Consumer Audio VDD/2 VOUT RL 1 k 100 k High Gain Amplifier (G = 101V/V) 35% Design Aids: • • • • SPICE Macro Models FilterLab® Software Microchip Advanced Part Selector (MAPS) Analog Demonstration and Evaluation Boards - MCP651EV-VOS • Application Notes Percentage of Occurrences • • • • • • • MCP65X VIN Typical Applications: 30% 25% 80 Samples TA = +25°C VDD = 2.5V and 5.5V Calibrated at +25°C 20% 15% 10% 5% 0% -100 -80 -60 -40 -20 0 20 40 60 Input Offset Voltage (µV) 80 100 High Gain-Bandwidth Op Amp Portfolio Model Family Channels/Package Gain-Bandwidth VOS (max.) IQ/Ch (typ.) MCP621/1S/2/3/4/5/9 1, 2, 4 20 MHz 0.2 mV 2.5 mA MCP631/2/3/4/5/9 1, 2, 4 24 MHz 8.0 mV 2.5 mA MCP651/1S/2/3/4/5/9 1, 2, 4 50 MHz 0.2 mV 6.0 mA 1, 2, 3, 4 60 MHz 8.0 mV 6.0 mA MCP660/1/2/3/4/5/9  2009-2014 Microchip Technology Inc. DS20002146D-page 1 MCP651/1S/2/3/4/5/9 Package Types MCP651 2x3 TDFN * MCP651 SOIC NC 1 8 CAL/CS NC 1 VIN– 2 7 VDD VIN– 2 VIN+ 3 6 VOUT VIN+ 3 VSS 4 5 VCAL VSS 4 8 CAL/CS EP 9 MCP654 SOIC, TSSOP MCP651S SOT-23-5 7 VDD VSS 6 VOUT 5 VDD VOUT 1 2 VIN+ 3 5 VCAL 4 VIN– VOUTA 1 14 VOUTD VINA- 2 VINA+ 3 VDD 4 13 VIND- VINB+ 5 VINB- 6 10 VINC+ 12 VIND+ 11 VSS 9 VINC- VOUTB 7 5 VINB+ VSS 4 6 VINB– 5 VINB+ MCP655 3x3 DFN * VOUTA 1 VINA– 2 VINA+ 3 VSS 4 CALA/CSA 5 EP 11 VSS 5 CAL/CS 2 VIN+ 3 MCP655 MSOP 10 VDD VOUTA 1 10 VDD 9 VOUTB VINA– 2 VINA+ 3 VSS 4 9 VOUTB 8 VINB– 7 VINB+ 8 VINB– 7 VINB+ 6 CALB/CSB CALA/CSA 5 6 CALB/CSB * Includes Exposed Thermal Pad (EP); see Table 3-1. DS20002146D-page 2 4 VIN– VIND- VINA+ 3 6 VDD VOUTD 6 VINB– VOUT 1 16 15 14 13 12 VIND+ VINA- 1 VINA+ 2 11 VSS EP 17 VDD 3 10 VINC+ 9 VINC- VINB+ 4 5 6 7 8 VOUTC VINA– 2 MCP659 4x4 QFN* CALBC/CSBC VSS 4 7 VOUTB 8 VDD 7 VOUTB VOUTA VINA+ 3 EP 9 VOUTA 1 VINB- VINA– 2 8 VDD VOUTB VOUTA 1 MCP653 SOT-23-6 CALAD/CSAD MCP652 SOIC MCP652 3x3 DFN * 8 VOUTC  2009-2014 Microchip Technology Inc. MCP651/1S/2/3/4/5/9 1.0 ELECTRICAL CHARACTERISTICS 1.1 Absolute Maximum Ratings † † Notice: Stresses above those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. This is a stress rating only and functional operation of the device at those or any other conditions above those indicated in the operational listings of this specification is not implied. Exposure to maximum rating conditions for extended periods may affect device reliability. VDD – VSS .......................................................................6.5V Current at Input Pins ....................................................±2 mA Analog Inputs (VIN+ and VIN–) †† . VSS – 1.0V to VDD + 1.0V All other Inputs and Outputs .......... VSS – 0.3V to VDD + 0.3V Difference Input voltage ...................................... |VDD – VSS| Output Short Circuit Current ................................ Continuous Current at Output and Supply Pins ..........................±150 mA Storage Temperature ...................................-65°C to +150°C Max. Junction Temperature ........................................ +150°C ESD protection on all pins (HBM, MM)  1 kV, 200V 1.2 †† See Section 4.2.2 “Input Voltage and Current Limits”. Specifications TABLE 1-1: DC ELECTRICAL SPECIFICATIONS Electrical Characteristics: Unless otherwise indicated, TA = +25°C, VDD = +2.5V to +5.5V, VSS = GND, VCM = VDD/3, VOUT  VDD/2, VL = VDD/2, RL = 1 k to VL and CAL/CS = VSS (refer to Figure 1-2). Parameters Sym. Min. Typ. Max. Units Conditions Input Offset VOS -200 — +200 µV VOSTRM — 37 200 µV VOS/TA — ±2.5 — PSRR 61 76 — IB — 6 — pA IB — 130 — pA TA= +85°C TA= +125°C Input Offset Voltage Input Offset Voltage Trim Step Input Offset Voltage Drift Power Supply Rejection Ratio After calibration (Note 1) µV/°C TA= -40°C to +125°C dB Input Current and Impedance Input Bias Current Across Temperature IB — 1700 5,000 pA Input Offset Current IOS — ±1 — pA Common Mode Input Impedance ZCM — 1013||9 — ||pF Differential Input Impedance ZDIFF — 1013||2 — ||pF Common Mode Input Voltage Range VCMR VSS  0.3 — VDD  1.3 V (Note 2) Common Mode Rejection Ratio CMRR 65 81 — dB VDD = 2.5V, VCM = -0.3 to 1.2V CMRR 68 84 — dB VDD = 5.5V, VCM = -0.3 to 4.2V AOL 88 114 — dB VDD = 2.5V, VOUT = 0.3V to 2.2V AOL 94 123 — dB VDD = 5.5V, VOUT = 0.3V to 5.2V VOL, VOH VSS + 25 — VDD  25 mV VDD = 2.5V, G = +2, 0.5V Input Overdrive VOL, VOH VSS + 50 — VDD  50 mV VDD = 5.5V, G = +2, 0.5V Input Overdrive ISC ±50 ±95 ±145 mA VDD = 2.5V (Note 3) ISC ±50 ±100 ±150 mA VDD = 5.5V (Note 3) Across Temperature Common Mode Open-Loop Gain DC Open-Loop Gain (large signal) Output Maximum Output Voltage Swing Output Short-Circuit Current Note 1: 2: 3: Describes the offset (under the specified conditions) right after power-up, or just after the CAL/CS pin is toggled. Thus, 1/f noise effects (an apparent wander in VOS; see Figure 2-35) are not included. See Figure 2-6 and Figure 2-7 for temperature effects. The ISC specifications are for design guidance only; they are not tested.  2009-2014 Microchip Technology Inc. DS20002146D-page 3 MCP651/1S/2/3/4/5/9 TABLE 1-1: DC ELECTRICAL SPECIFICATIONS (CONTINUED) Electrical Characteristics: Unless otherwise indicated, TA = +25°C, VDD = +2.5V to +5.5V, VSS = GND, VCM = VDD/3, VOUT  VDD/2, VL = VDD/2, RL = 1 k to VL and CAL/CS = VSS (refer to Figure 1-2). Parameters Sym. Min. Typ. Max. Units mV Conditions Calibration Input VCALRNG VSS + 0.1 — VDD – 1.4 Internal Calibration Voltage VCAL 0.31VDD 0.33VDD 0.35VDD Input Impedance ZCAL — 100 || 5 — VDD 2.5 — 5.5 V IQ 3 6 9 mA POR Input Threshold, Low VPRL 1.15 1.40 — V POR Input Threshold, High VPRH — 1.40 1.65 V Calibration Input Voltage Range VCAL pin externally driven VCAL pin open k||pF Power Supply Supply Voltage Quiescent Current per Amplifier Note 1: 2: 3: IO = 0 Describes the offset (under the specified conditions) right after power-up, or just after the CAL/CS pin is toggled. Thus, 1/f noise effects (an apparent wander in VOS; see Figure 2-35) are not included. See Figure 2-6 and Figure 2-7 for temperature effects. The ISC specifications are for design guidance only; they are not tested. TABLE 1-2: AC ELECTRICAL SPECIFICATIONS Electrical Characteristics: Unless otherwise indicated, TA = 25°C, VDD = +2.5V to +5.5V, VSS = GND, VCM = VDD/2, VOUT  VDD/2, VL = VDD/2, RL = 1 k to VL, CL = 20 pF and CAL/CS = VSS (refer to Figure 1-2). Parameters Sym. Min. Typ. Max. Units GBWP — 50 — MHz PM — 65 — ° ROUT — 20 —  THD+N — 0.0012 — % Conditions AC Response Gain-Bandwidth Product Phase Margin Open-Loop Output Impedance G = +1 AC Distortion Total Harmonic Distortion plus Noise G = +1, VOUT = 4VP-P, f = 1 kHz, VDD = 5.5V, BW = 80 kHz Step Response tr — 6 — ns SR — 30 — V/µs G = +1 Input Noise Voltage Eni — 17 — µVP-P f = 0.1 Hz to 10 Hz Input Noise Voltage Density eni — 7.5 — nV/Hz f = 1 MHz Input Noise Current Density ini 4 — fA/Hz Rise Time, 10% to 90% Slew Rate G = +1, VOUT = 100 mVP-P Noise DS20002146D-page 4 f = 1 kHz  2009-2014 Microchip Technology Inc. MCP651/1S/2/3/4/5/9 TABLE 1-3: DIGITAL ELECTRICAL SPECIFICATIONS Electrical Characteristics: Unless otherwise indicated, TA = 25°C, VDD = +2.5V to +5.5V, VSS = GND, VCM = VDD/2, VOUT  VDD/2, VL = VDD/2, RL = 1 k to VL, CL = 20 pF and CAL/CS = VSS (refer to Figure 1-1 and Figure 1-2). Parameters Sym. Min. Typ. Max. Units Conditions CAL/CS Logic Threshold, Low VIL VSS — 0.2VDD V CAL/CS Input Current, Low ICSL — 0 — nA CAL/CS Logic Threshold, High VIH 0.8VDD VDD V CAL/CS Input Current, High ICSH — 0.7 — µA CAL/CS = VDD ISS -3.5 -1.8 — µA Single, CAL/CS = VDD = 2.5V ISS -8 -4 — µA Single, CAL/CS = VDD = 5.5V ISS -5 -2.5 — µA Dual, CAL/CS = VDD = 2.5V Dual, CAL/CS = VDD = 5.5V CAL/CS Low Specifications CAL/CS = 0V CAL/CS High Specifications GND Current ISS -10 -5 — µA RPD — 5 — M IO(LEAK) — 50 — nA CAL/CS = VDD VDD Low to Amplifier Off Time (output goes High Z) tPOFF — 200 — ns G = +1 V/V, VL = VSS, VDD = 2.5V to 0V step to VOUT = 0.1 (2.5V) VDD High to Amplifier On Time (including calibration) tPON 100 200 300 ms G = +1 V/V, VL = VSS, VDD = 0V to 2.5V step to VOUT = 0.9 (2.5V) CAL/CS Input Hysteresis VHYST — 0.25 — V CAL/CS Setup Time (between CAL/CS edges) tCSU 1 — — µs G = +1 V/V, VL = VSS (Notes 2, 3, 4) CAL/CS = 0.8VDD to VOUT = 0.1 (VDD/2) CAL/CS High to Amplifier Off Time (output goes High Z) tCOFF — 200 — ns G = +1 V/V, VL = VSS, CAL/CS = 0.8VDD to VOUT = 0.1 (VDD/2) CAL/CS Low to Amplifier On Time (including calibration) tCON — 3 4 ms G = +1 V/V, VL = VSS, MCP651 and MCP655, CAL/CS = 0.2VDD to VOUT = 0.9 (VDD/2) tCON — 6 8 ms G = +1 V/V, VL = VSS, MCP659, CAL/CS = 0.2VDD to VOUT = 0.9 (VDD/2) CAL/CS Internal Pull-Down Resistor Amplifier Output Leakage POR Dynamic Specifications CAL/CS Dynamic Specifications Note 1: 2: 3: 4: The MCP652 single, MCP653 single, MCP655 dual and MCP659 quad have their CAL/CS inputs internally pulled down to VSS (0V). This time ensures that the internal logic recognizes the edge. However, for the rising edge case, if CAL/CS is raised before the calibration is complete, the calibration will be aborted and the part will return to Low-Power mode. For the MCP655 dual, there is an additional constraint. CALA/CSA and CALB/CSB can be toggled simultaneously (within a time much smaller than tCSU) to make both op amps perform the same function simultaneously. If they are toggled independently, then CALA/CSA (CALB/CSB) cannot be allowed to toggle while op amp B (op amp A) is in Calibration mode; allow more than the maximum tCON time (4 ms) before the other side is toggled. For the MCP659 quad, there is an additional constraint. CALAD/CSAD and CALBC/CSBC can be toggled simultaneously (within a time much smaller than tCSU) to make all four op amps perform the same function simultaneously, and the maximum tCON time is approximately doubled (8 ms). If they are toggled independently, then CALAD/CSAD (CALBC/CSBC) cannot be allowed to toggle while op amps B and C (op amps A and D) are in Calibration mode; allow more than the maximum tCON time (8 ms) before the other side is toggled.  2009-2014 Microchip Technology Inc. DS20002146D-page 5 MCP651/1S/2/3/4/5/9 TABLE 1-4: TEMPERATURE SPECIFICATIONS Electrical Characteristics: Unless otherwise indicated, all limits are specified for: VDD = +2.5V to +5.5V, VSS = GND. Parameters Sym. Min. Typ. Max. Units TA -40 — +125 °C Operating Temperature Range TA -40 — +125 °C Storage Temperature Range TA -65 — +150 °C Conditions Temperature Ranges Specified Temperature Range (Note 1) Thermal Package Resistances Thermal Resistance, 5L-2×3 SOT JA — 220.7 — °C/W Thermal Resistance, 6L-2×3 SOT JA — 190.5 — °C/W Thermal Resistance, 8L-2×3 TDFN JA — 52.5 — °C/W Thermal Resistance, 8L-3×3 DFN JA — 63 — °C/W Thermal Resistance, 8L-SOIC JA — 163 — °C/W Thermal Resistance, 10L-3×3 DFN JA — 71 — °C/W Thermal Resistance, 10L-MSOP JA — 202 — °C/W Thermal Resistance, 14L-SOIC JA — 95.3 — °C/W Thermal Resistance, 14L-TSSOP JA — 100 — °C/W Thermal Resistance, 16L-4x4-QFN JA — 46 — °C/W (Note 2) (Note 2) Operation must not cause TJ to exceed Maximum Junction Temperature specification (150°C). Measured on a standard JC51-7, four-layer printed circuit board with ground plane and vias. Note 1: 2: 1.3 (Note 2) Timing Diagram CAL/CS VDD VPRH tPON tCSU tCOFF VOUT High Z On ISS -3 µA (typical) -6 mA (typical) ICS 0 nA (typical) Note: VIL VIH High Z -3 µA (typical) 0.7 µA (typical) VPRL tCON tPOFF On -6 mA (typical) High Z -3 µA (typical) 0 nA (typical) For the MCP655 dual and the MCP659 quad, there is an additional constraint on toggling the two CAL/CS pins close together; see the TCON specification in Table 1-3. FIGURE 1-1: DS20002146D-page 6 Timing Diagram.  2009-2014 Microchip Technology Inc. MCP651/1S/2/3/4/5/9 1.4 Test Circuits The circuit used for most DC and AC tests is shown in Figure 1-2. This circuit can independently set VCM and VOUT; see Equation 1-1. Note that VCM is not the circuit’s Common mode voltage ((VP + VM)/2), and that VOST includes VOS plus the effects (on the input offset error, VOST) of temperature, CMRR, PSRR and AOL. CF 6.8 pF RG 10 k VP EQUATION 1-1: VIN+ G DM = R F  R G MCP65X V CM =  VP + VDD  2   2 V OUT =  V DD  2  +  V P – V M  + V OST  1 + G DM  Where: GDM = Differential Mode Gain (V/V) VCM = Op Amp’s Common Mode Input Voltage (V)  2009-2014 Microchip Technology Inc. VDD CB1 100 nF VDD/2 CB2 2.2 µF VIN– V OST = V IN– – V IN+ VOST = Op Amp’s Total Input Offset Voltage RF 10 k (mV) VM RG 10 k RL 1 k RF 10 k CF 6.8 pF VOUT CL 20 pF VL FIGURE 1-2: AC and DC Test Circuit for Most Specifications. DS20002146D-page 7 MCP651/1S/2/3/4/5/9 2.0 TYPICAL PERFORMANCE CURVES Note: The graphs and tables provided following this note are a statistical summary based on a limited number of samples and are provided for informational purposes only. The performance characteristics listed herein are not tested or guaranteed. In some graphs or tables, the data presented may be outside the specified operating range (e.g., outside specified power supply range) and therefore outside the warranted range. Note: Unless otherwise indicated, TA = +25°C, VDD = +2.5V to 5.5V, VSS = GND, VCM = VDD/3, VOUT = VDD/2, VL = VDD/2, RL = 1 kto VL, CL = 20 pF, and CAL/CS = VSS. DC Signal Inputs Percentage of Occurrences 35% 30% 25% 700 80 Samples TA = +25°C VDD = 2.5V and 5.5V Calibrated at +25°C Input Offset Voltage (µV) 2.1 20% 15% 10% 5% 0% FIGURE 2-1: 400 300 100 0 -8 -6 -4 1.5 2.0 2.5 3.0 3.5 4.0 4.5 5.0 5.5 6.0 6.5 Power Supply Voltage (V) FIGURE 2-4: Input Offset Voltage vs. Power Supply Voltage. Input Offset Voltage. 80 Samples VDD = 2.5V and 5.5V TA = -40°C to +125°C Calibrated at +25°C -10 -2 0 2 4 6 8 50 40 30 20 10 0 -10 -20 -30 -40 -50 10 Input Offset Voltage Drift. VDD = 5.5V 0.0 80 Samples TA = +25°C VDD = 2.5V and 5.5V No Change (includes noise) Calibration Changed VDD = 2.5V FIGURE 2-5: Output Voltage. Calibration Changed Low Input Common Mode Headroom (V) Percentage of Occurrences FIGURE 2-2: Representative Part 0.0 0.5 1.0 1.5 2.0 2.5 3.0 3.5 4.0 4.5 5.0 5.5 Output Voltage (V) Input Offset Voltage Drift (µV/°C) 55% 50% 45% 40% 35% 30% 25% 20% 15% 10% 5% 0% +125°C +85°C +25°C -40°C 200 80 100 Input Offset Voltage (µV) Percentage of Occurrences 500 -100 -100 -80 -60 -40 -20 0 20 40 60 Input Offset Voltage (µV) 20% 18% 16% 14% 12% 10% 8% 6% 4% 2% 0% Representative Part Calibrated at VDD = 6.5V 600 -0.1 Input Offset Voltage vs. 1 Lot Low (VCMR_L – VSS) -0.2 VDD = 2.5V -0.3 VDD = 5.5V -0.4 -0.5 -100 -80 -60 -40 -20 0 20 40 60 80 100 Input Offset Voltage Repeatability (µV) FIGURE 2-3: Input Offset Voltage Repeatability (repeated calibration). DS20002146D-page 8 -50 -25 0 25 50 75 100 Ambient Temperature (°C) 125 FIGURE 2-6: Low-Input Common Mode Voltage Headroom vs. Ambient Temperature.  2009-2014 Microchip Technology Inc. MCP651/1S/2/3/4/5/9 Note: Unless otherwise indicated, TA = +25°C, VDD = +2.5V to 5.5V, VSS = GND, VCM = VDD/3, VOUT = VDD/2, VL = VDD/2, RL = 1 kto VL, CL = 20 pF, and CAL/CS = VSS. 1 Lot High (VDD – VCMR_H) CMRR, PSRR (dB) High Input Common Mode Headroom (V) 1.4 1.3 VDD = 2.5V 1.2 1.1 VDD = 5.5V 1.0 -50 -25 0 25 50 75 100 Ambient Temperature (°C) 125 CMRR, VDD = 5.5V CMRR, VDD = 2.5V -50 -25 FIGURE 2-8: Input Offset Voltage vs. Common Mode Voltage with VDD = 2.5V. 120 115 VDD = 2.5V 110 105 100 95 -50 10,000 Input Bias, Offset Currents (pA) Input Common Mode Voltage (V) FIGURE 2-9: Input Offset Voltage vs. Common Mode Voltage with VDD = 5.5V.  2009-2014 Microchip Technology Inc. 5.0 4.5 4.0 3.5 3.0 2.5 2.0 1.5 1.0 0.5 -40°C +25°C +85°C +125°C 0.0 125 -25 0 25 50 75 Ambient Temperature (°C) 100 125 FIGURE 2-11: DC Open-Loop Gain vs. Ambient Temperature. VDD = 5.5V Representative Part -0.5 Input Offset Voltage (µV) 100 VDD = 5.5V 125 Input Common Mode Voltage (V) 1000 800 600 400 200 0 -200 -400 -600 -800 -1000 0 25 50 75 Ambient Temperature (°C) FIGURE 2-10: CMRR and PSRR vs. Ambient Temperature. DC Open-Loop Gain (dB) 2.0 1.8 1.6 1.4 1.2 1.0 0.8 0.6 0.4 0.2 0.0 -0.2 -40°C +25°C +85°C +125°C -0.4 PSRR 130 VDD = 2.5V Representative Part -0.6 Input Offset Voltage (µV) FIGURE 2-7: High-Input Common Mode Voltage Headroom vs. Ambient Temperature. 1000 800 600 400 200 0 -200 -400 -600 -800 -1000 110 105 100 95 90 85 80 75 70 65 60 VDD = 5.5V VCM = VCMR_H 1,000 IB 100 10 -IOS 1 25 45 65 85 105 Ambient Temperature (°C) 125 FIGURE 2-12: Input Bias and Offset Currents vs. Ambient Temperature with VDD = +5.5V. DS20002146D-page 9 MCP651/1S/2/3/4/5/9 Note: Unless otherwise indicated, TA = +25°C, VDD = +2.5V to 5.5V, VSS = GND, VCM = VDD/3, VOUT = VDD/2, 160 140 120 100 80 60 40 20 0 -20 -40 -60 1.E-03 1m TA = +85°C VDD = 5.5V 100µ 1.E-04 10µ 1.E-05 IB 1µ 1.E-06 100n 1.E-07 10n 1.E-08 IOS 1n 1.E-09 100p 1.E-10 10p 1.E-11 1p 1.E-12 0.0 0.5 1.0 1.5 2.0 2.5 3.0 3.5 4.0 4.5 5.0 5.5 Common Mode Input Voltage (V) FIGURE 2-13: Input Bias and Offset Currents vs. Common Mode Input Voltage with TA = +85°C. 2000 Input Bias, Offset Currents (pA) Input Current Magnitude (A) Input Bias, Offset Currents (pA) VL = VDD/2, RL = 1 kto VL, CL = 20 pF, and CAL/CS = VSS. 1500 +125°C +85°C +25°C -40°C -1.0 -0.9 -0.8 -0.7 -0.6 -0.5 -0.4 -0.3 -0.2 -0.1 0.0 Input Voltage (V) FIGURE 2-15: Input Bias Current vs. Input Voltage (below VSS). TA = +125°C VDD = 5.5V IB 1000 500 0 IOS -500 -1000 0.0 0.5 1.0 1.5 2.0 2.5 3.0 3.5 4.0 4.5 5.0 5.5 Common Mode Input Voltage (V) FIGURE 2-14: Input Bias and Offset Currents vs. Common Mode Input Voltage with TA = +125°C. DS20002146D-page 10  2009-2014 Microchip Technology Inc. MCP651/1S/2/3/4/5/9 Note: Unless otherwise indicated, TA = +25°C, VDD = +2.5V to 5.5V, VSS = GND, VCM = VDD/3, VOUT = VDD/2, VL = VDD/2, RL = 1 kto VL, CL = 20 pF, and CAL/CS = VSS. Other DC Voltages and Currents 14 8 VDD = 5.5V 12 7 VOL – VSS -IOUT 10 Supply Current (mA/amplifier) 8 6 4 VDD – VOH IOUT VDD = 2.5V 2 2 RL = 1 kΩ 6.5 6.0 5.5 5.0 4.5 4.0 3.5 3.0 2.5 2.0 FIGURE 2-19: Supply Voltage. Supply Current vs. Power 9 VOL – VSS 8 8 6 VDD – VOH VDD = 2.5V 1.5 Power Supply Voltage (V) VDD = 5.5V 10 1.0 100 Supply Current (mA/amplifier) 2 7 VDD = 5.5V 6 5 VDD = 2.5V 4 3 2 1 5.5 5.0 4.5 4.0 3.5 Common Mode Input Voltage (V) FIGURE 2-17: Output Voltage Headroom vs. Ambient Temperature. 100 80 60 40 20 0 -20 -40 -60 -80 -100 3.0 125 2.5 100 2.0 0 25 50 75 Ambient Temperature (°C) 1.5 -25 0.0 -50 1.0 0 0 0.5 Output Headroom (mV) +125°C +85°C +25°C -40°C 3 0.5 10 Output Current Magnitude (mA) 12 FIGURE 2-20: Supply Current vs. Common Mode Input Voltage. POR Trip Voltages (V) 1.8 +125°C +85°C +25°C -40°C 1.6 1.4 VPRH 1.2 1.0 VPRL 0.8 0.6 0.4 0.2 6.5 6.0 5.5 5.0 4.5 4.0 3.5 3.0 2.5 2.0 1.5 1.0 0.5 0.0 0.0 Output Short Circuit Current (mA) 4 0 0 FIGURE 2-16: Ratio of Output Voltage Headroom to Output Current. 4 5 1 1 14 6 0.0 Ratio of Output Headroom to Output Current (mV/mA) 2.2 Power Supply Voltage (V) FIGURE 2-18: Output Short-Circuit Current vs. Power Supply Voltage.  2009-2014 Microchip Technology Inc. -50 -25 0 25 50 75 Ambient Temperature (°C) 100 125 FIGURE 2-21: Power-On Reset Voltages vs. Ambient Temperature. DS20002146D-page 11 MCP651/1S/2/3/4/5/9 Note: Unless otherwise indicated, TA = +25°C, VDD = +2.5V to 5.5V, VSS = GND, VCM = VDD/3, VOUT = VDD/2, 25% 144 Samples VDD = 2.5V and 5.5V 20% 15% 10% 5% Normalized Internal Calibration Voltage; VCAL/VDD FIGURE 2-22: Normalized Internal Calibration Voltage. DS20002146D-page 12 33.52% 33.48% 33.44% 33.40% 33.36% 33.32% 33.28% 33.24% 0% Internal V CAL Resistance (kΩ) 30% 33.20% Percentage of Occurrences VL = VDD/2, RL = 1 kto VL, CL = 20 pF, and CAL/CS = VSS. 140 120 100 80 60 40 20 0 -50 -25 FIGURE 2-23: Temperature. 0 25 50 75 Ambient Temperature (°C) 100 125 VCAL Input Resistance vs.  2009-2014 Microchip Technology Inc. MCP651/1S/2/3/4/5/9 Note: Unless otherwise indicated, TA = +25°C, VDD = +2.5V to 5.5V, VSS = GND, VCM = VDD/3, VOUT = VDD/2, VL = VDD/2, RL = 1 kto VL, CL = 20 pF, and CAL/CS = VSS. Frequency Response 0 100 -30 80 -60 AOL -90 -120 | AOL | 20 -150 0 -180 -20 -210 50 40 40 50 30 40 20 30 -50 -25 0 25 50 75 100 Ambient Temperature (°C) 10 125 FIGURE 2-26: Gain-Bandwidth Product and Phase Margin vs. Ambient Temperature.  2009-2014 Microchip Technology Inc. Open-Loop Output Impedance (Ω) GBWP VDD = 5.5V VDD = 2.5V Phase Margin (°) Gain Bandwidth Product (MHz) 50 60 70 GBWP 30 60 VDD = 2.5V 20 10 PM 50 40 30 FIGURE 2-28: Gain-Bandwidth Product and Phase Margin vs. Output Voltage. 60 70 80 VDD = 5.5V 0.0 0.5 1.0 1.5 2.0 2.5 3.0 3.5 4.0 4.5 5.0 5.5 Output Voltage (V) Open-Loop Gain vs. PM 90 0 70 80 6.0 60 Frequency (Hz) 90 5.5 FIGURE 2-27: Gain-Bandwidth Product and Phase Margin vs. Common Mode Input Voltage. 1.E+1 1.E+5 1.E+6 1.E+7 100M 1.E+8 1.E+9 10 1.E+2 100 1.E+3 1k 1.E+4 10k 100k 1M 10M 1G FIGURE 2-25: Frequency. 5.0 10 4.5 30 Common Mode Input Voltage (V) 120 40 20 10M 1.E+7 CMRR and PSRR vs. 60 40 4.0 1M 1.E+6 30 GBWP Phase Margin (°) Open-Loop Gain (dB) FIGURE 2-24: Frequency. 10k 100k 1.E+4 1.E+5 Frequency (Hz) 50 3.5 1k 1.E+3 Open-Loop Phase (°) 0 100 1.E+2 50 40 3.0 10 60 2.5 20 VDD = 5.5V VDD = 2.5V 1.5 30 70 1.0 40 60 0.5 50 80 0.0 60 70 PM -0.5 PSRR+ PSRR- CMRR Gain Bandwidth Product (MHz) CMRR, PSRR (dB) 70 Gain Bandwidth Product (MHz) 90 80 Phase Margin (°) 90 2.0 2.3 1000 100 10 G = 101 V/V G = 11 V/V G = 1 V/V 1 0.1 1k 10k 100k 1.0E+06 1M 10M 1.0E+08 100M 1.0E+03 1.0E+04 1.0E+05 1.0E+07 Frequency (Hz) FIGURE 2-29: Closed-Loop Output Impedance vs. Frequency. DS20002146D-page 13 MCP651/1S/2/3/4/5/9 Note: Unless otherwise indicated, TA = +25°C, VDD = +2.5V to 5.5V, VSS = GND, VCM = VDD/3, VOUT = VDD/2, 10 9 8 7 6 5 G = 1 V/V G = 2 V/V 4 G • 4 V/V 3 2 1 0 10p 100p 1.0E-10 1n 10n 1.0E-11 1.0E-09 Normalized Capacitive Load; CL/G (F) FIGURE 2-30: Gain Peaking vs. Normalized Capacitive Load. DS20002146D-page 14 Channel-to-Channel Separation (dB) Gain Peaking (dB) VL = VDD/2, RL = 1 kto VL, CL = 20 pF, and CAL/CS = VSS. 150 140 130 120 110 100 90 80 RS = 0Ω RS = 1 kΩ 70 RS = 10 kΩ 60 RS = 100 kΩ 50 1k 10k 1.E+03 1.E+04 RTI VCM = VDD/2 G = +1 V/V 100k 1M 1.E+05 1.E+06 Frequency (Hz) 10M 1.E+07 FIGURE 2-31: Channel-to-Channel Separation vs. Frequency.  2009-2014 Microchip Technology Inc. MCP651/1S/2/3/4/5/9 Note: Unless otherwise indicated, TA = +25°C, VDD = +2.5V to 5.5V, VSS = GND, VCM = VDD/3, VOUT = VDD/2, VL = VDD/2, RL = 1 kto VL, CL = 20 pF, and CAL/CS = VSS. Input Noise and Distortion 1.E+4 10µ Input Offset + Noise; VOS + e ni(t) (µV) 20 1.E+3 1µ 1.E+2 100n 1.E+1 10n 15 Representative Part NPBW = 0.1 Hz 10 5 0 -5 -10 -15 -20 1n 1.E+0 0.1 1.E-1 1 1.E+0 10 1.E+1 160 Input Noise Voltage Density THD + Noise (%) 100 VDD = 5.5V 80 60 40 20 25 30 Time (min) 35 40 45 50 VDD = 5.0V 0.1 BW = 22 Hz to > 500 kHz G = 1 V/V G = 11 V/V 0.01 0.001 BW = 22 Hz to 80 kHz 6.0 5.5 5.0 4.5 4.0 3.5 3.0 2.5 2.0 1.0 0.5 0.0 0.0001 1.E+2 100 Common Mode Input Voltage (V) FIGURE 2-33: Input Noise Voltage Density vs. Input Common Mode Voltage with f = 100 Hz. FIGURE 2-36: 1.E+3 1k 1.E+4 10k Frequency (Hz) 1.E+5 100k THD+N vs. Frequency. VDD = 2.5V VDD = 5.5V 5.5 5.0 4.5 4.0 3.5 3.0 2.5 2.0 1.5 1.0 0.5 0.0 f = 1 MHz -0.5 Input Noise Voltage Density (nV/¥Hz) 15 f = 100 Hz 0 12 11 10 9 8 7 6 5 4 3 2 1 0 10 1 120 20 5 FIGURE 2-35: Input Noise plus Offset vs. Time with 0.1 Hz Filter. VDD = 2.5V 140 -0.5 Input Noise Voltage Density (nV/Hz) FIGURE 2-32: vs. Frequency. 0 100 1.E+3 1k 1.E+4 10k 100k 1M 1.E+7 10M 1.E+2 1.E+5 1.E+6 Frequency (Hz) 1.5 Input Noise Voltage Density (nV/¥Hz) 2.4 Common Mode Input Voltage (V) FIGURE 2-34: Input Noise Voltage Density vs. Input Common Mode Voltage with f = 1 MHz.  2009-2014 Microchip Technology Inc. DS20002146D-page 15 MCP651/1S/2/3/4/5/9 Note: Unless otherwise indicated, TA = +25°C, VDD = +2.5V to 5.5V, VSS = GND, VCM = VDD/3, VOUT = VDD/2, VL = VDD/2, RL = 1 kto VL, CL = 20 pF, and CAL/CS = VSS. 2.5 Time Response VIN 0 Output Voltage (V) Output Voltage (10 mV/div) VDD = 5.5V G=1 VOUT 20 40 60 5.5 5.0 4.5 4.0 3.5 3.0 2.5 2.0 1.5 1.0 0.5 0.0 80 100 120 140 160 180 200 Time (ns) Non-inverting Small Signal VIN VOUT 100 200 FIGURE 2-40: Response. 300 400 500 Time (ns) 600 700 Input, Output Voltages (V) VIN VOUT 800 Inverting Large Signal Step 7 VDD = 5.5V G=1 VDD = 5.5V G=2 VIN 6 5 VOUT 4 3 2 1 0 -1 0 100 200 Output Voltage (10 mV/div) FIGURE 2-38: Step Response. 300 400 500 Time (ns) 600 700 0 800 Non-inverting Large Signal VDD = 5.5V G = -1 RF = 499Ω VOUT 50 100 FIGURE 2-39: Response. DS20002146D-page 16 150 200 250 Time (ns) 300 350 400 1 2 3 4 5 6 Time (ms) 7 8 9 10 FIGURE 2-41: The MCP651/1S/2/3/4/5/9 family shows no input phase reversal with overdrive. VIN 0 VDD = 5.5V G = -1 RF = 499Ω 0 Slew Rate (V/µs) Output Voltage (V) FIGURE 2-37: Step Response. 5.5 5.0 4.5 4.0 3.5 3.0 2.5 2.0 1.5 1.0 0.5 0.0 60 55 50 45 40 35 30 25 20 15 10 5 0 Falling Edge VDD = 5.5V VDD = 2.5V Rising Edge -50 -25 0 25 50 75 Ambient Temperature (°C) 100 125 Inverting Small Signal Step FIGURE 2-42: Temperature. Slew Rate vs. Ambient  2009-2014 Microchip Technology Inc. MCP651/1S/2/3/4/5/9 Note: Unless otherwise indicated, TA = +25°C, VDD = +2.5V to 5.5V, VSS = GND, VCM = VDD/3, VOUT = VDD/2, VL = VDD/2, RL = 1 kto VL, CL = 20 pF, and CAL/CS = VSS. Maximum Output Voltage Swing (VP-P) 10 VDD = 5.5V VDD = 2.5V 1 0.1 100k 1.E+05 1M 10M 1.E+06 1.E+07 Frequency (Hz) 100M 1.E+08 FIGURE 2-43: Maximum Output Voltage Swing vs. Frequency.  2009-2014 Microchip Technology Inc. DS20002146D-page 17 MCP651/1S/2/3/4/5/9 Note: Unless otherwise indicated, TA = +25°C, VDD = +2.5V to 5.5V, VSS = GND, VCM = VDD/3, VOUT = VDD/2, VL = VDD/2, RL = 1 kto VL, CL = 20 pF, and CAL/CS = VSS. Calibration and Chip Select Response 1.1 1.0 0.9 0.8 0.7 0.6 0.5 0.4 0.3 0.2 0.1 0.0 0.40 CAL/CS = VDD CAL/CS Hysteresis (V) CAL/CS Current (µA) 2.6 0.35 0.30 0.20 0.10 0.05 0.00 -50 IDD Op Amp turns off Op Amp turns on Calibration starts CAL/CS VOUT 0 1 2 3 4 5 6 Time (ms) 7 8 9 8 6 4 2 0 -2 -4 -6 -8 -10 -12 Calibration starts Op Amp turns off Op Amp turns on CAL/CS VOUT 0 1 2 3 4 5 6 Time (ms) 7 8 9 Power Supply Current; IDD (mA) IDD 10 FIGURE 2-46: CAL/CS Voltage, Output Voltage and Supply Current (for Side A) vs. Time with VDD = 5.5V. DS20002146D-page 18 100 125 3.5 3.0 2.5 2.0 1.5 1.0 0.5 0.0 -50 10 8 6 4 2 0 -2 -4 -6 -8 -10 -12 -14 VDD = 5.5V G=1 VL = 0V 0 25 50 75 Ambient Temperature (°C) 4.0 10 FIGURE 2-45: CAL/CS Voltage, Output Voltage and Supply Current (for Side A) vs. Time with VDD = 2.5V. 11 10 9 8 7 6 5 4 3 2 1 0 -1 CAL/CS Turn On Time (ms) VDD = 2.5V G=1 VL = 0V -25 FIGURE 2-47: CAL/CS Hysteresis vs. Ambient Temperature. -25 0 25 50 75 Ambient Temperature (°C) 100 125 FIGURE 2-48: CAL/CS Turn-On Time vs. Ambient Temperature. 8 CAL/CS Pull-down Resistor (MΩ) 9 8 7 6 5 4 3 2 1 0 -1 CAL/CS Current vs. Power Power Supply Current; IDD (mA) FIGURE 2-44: Supply Voltage. CAL/CS, V OUT (V) VDD = 2.5V 0.15 0.0 0.5 1.0 1.5 2.0 2.5 3.0 3.5 4.0 4.5 5.0 5.5 Power Supply Voltage (V) CAL/CS, V OUT (V) VDD = 5.5V 0.25 Representative Part 7 6 5 4 3 2 1 0 -50 -25 0 25 50 75 Ambient Temperature (°C) 100 125 FIGURE 2-49: CAL/CS’s Pull-Down Resistor (RPD) vs. Ambient Temperature.  2009-2014 Microchip Technology Inc. MCP651/1S/2/3/4/5/9 Note: Unless otherwise indicated, TA = +25°C, VDD = +2.5V to 5.5V, VSS = GND, VCM = VDD/3, VOUT = VDD/2, VL = VDD/2, RL = 1 kto VL, CL = 20 pF, and CAL/CS = VSS. 1.E-06 CAL/CS = VDD -1 Output Leakage Current (A) Negative Power Supply Current; ISS (µA) 0 -2 -3 -4 +125°C +85°C +25°C -40°C -5 -6 Power Supply Voltage (V) FIGURE 2-50: Quiescent Current in Shutdown vs. Power Supply Voltage.  2009-2014 Microchip Technology Inc. 6.5 6.0 5.5 5.0 4.5 4.0 3.5 3.0 2.5 2.0 1.5 1.0 0.5 0.0 -7 CAL/CS = VDD = 5.5V 1.E-07 +125°C 1.E-08 +85°C 1.E-09 1.E-10 +25°C 1.E-11 0.0 0.5 1.0 1.5 2.0 2.5 3.0 3.5 4.0 4.5 5.0 5.5 6.0 Output Voltage (V) FIGURE 2-51: Output Voltage. Output Leakage Current vs. DS20002146D-page 19 MCP651/1S/2/3/4/5/9 3.0 PIN DESCRIPTIONS Descriptions of the pins are listed in Table 3-1. TABLE 3-1: PIN FUNCTION TABLE MCP651 MCP651S MCP652 MCP653 SOIC TDFN SOT SOIC DFN SOT MCP654 MCP655 SOIC TSSOP MSOP DFN MCP659 QFN Symbol Description 6 6 1 1 1 1 1 1 1 1 16 VOUT, VOUTA Output (op amp A) 2 2 4 2 2 4 2 2 2 2 1 VIN–, VINA– Inverting Input (op amp A) 3 3 3 3 3 3 3 3 3 3 2 VIN+, VINA+ Non-inverting Input (op amp A) 4 4 2 4 4 2 11 11 4 4 11 VSS Negative Power Supply 8 8 — — — 5 — — 5 5 — CAL/CS, CALA/CSA Calibrate/Chip Select Digital Input (op amp A) — — — — — — — — 6 6 — CALB/CSB Calibrate/Chip Select Digital Input (op amp B) — — — — — — — — — — 15 CALAD/ CSAD Calibrate/Chip Select Digital Input (op amps A and D) — — — — — — — — — — 7 CALBC/ CSBC Calibrate/Chip Select Digital Input (op amps B and C) — — — 5 5 — 5 5 7 7 4 VINB+ Non-inverting Input (op amp B) — — — 6 6 — 6 6 8 8 5 VINB– Inverting Input (op amp B) — — — 7 7 — 7 7 9 9 6 VOUTB Output (op amp B) — — — — — — 10 10 — — 10 VINC+ Non-inverting input (op amp C) — — — — — — 9 9 — — 9 VINC- Inverting Input (op amp C) — — — — — — 8 8 — — 8 VOUTC Output (op amp C) — — — — — — 12 12 — — 12 VIND+ Non-inverting Input (op amp D) — — — — — — 13 13 — — 13 VIND- Inverting Input (op amp D) — — — — — — 14 14 — — 14 VOUTD Output (op amp D) 7 7 5 8 8 6 4 4 10 10 3 VDD Positive Power Supply 5 5 — — — — — — — — — VCAL Calibration Common Mode Voltage Input 1 1 — — — — — — — — — NC No Internal Connection — 9 — — 9 — — — — 11 17 EP Exposed Thermal Pad (EP); must be connected to VSS DS20002146D-page 20  2009-2014 Microchip Technology Inc. MCP651/1S/2/3/4/5/9 3.1 Analog Outputs The analog output pins (VOUT) are low-impedance voltage sources. 3.2 Analog Inputs The non-inverting and inverting inputs (VIN+, VIN–, …) are high-impedance CMOS inputs with low bias currents. 3.3 Power Supply Pins The positive power supply (VDD) is 2.5V to 5.5V higher than the negative power supply (VSS). For normal operation, the other pins are between VSS and VDD. Typically, these parts are used in a single (positive) supply configuration. In this case, VSS is connected to ground and VDD is connected to the supply. VDD will need bypass capacitors. 3.4 Calibration Common Mode Voltage Input A low-impedance voltage placed at this input (VCAL) will set the op amps’ Common mode input voltage during calibration. If this pin is left open, the Common mode input voltage during calibration is approximately VDD/3. The internal resistor divider is disconnected from the supplies whenever the part is not in calibration.  2009-2014 Microchip Technology Inc. 3.5 Calibrate/Chip Select Digital Input This input (CAL/CS, …) is a CMOS, Schmitt-Triggered input that affects the Calibration and Low-Power modes of operation. When this pin goes high, the part is placed into a Low-Power mode and the output is High Z. When this pin goes low, a calibration sequence is started (which corrects VOS). At the end of the calibration sequence, the output becomes low-impedance and the part resumes normal operation. An internal POR triggers a calibration event when the part is powered on, or when the supply voltage drops too low. Thus, the MCP652 parts are calibrated, even though they do not have a CAL/CS pin. 3.6 Exposed Thermal Pad (EP) There is an internal connection between the Exposed Thermal Pad (EP) and the VSS pin; they must be connected to the same potential on the Printed Circuit Board (PCB). This pad can be connected to a PCB ground plane to provide a larger heat sink. This improves the package thermal resistance (JA). DS20002146D-page 21 MCP651/1S/2/3/4/5/9 4.0 APPLICATIONS The MCP651/1S/2/3/4/5/9 family of self-zeroed op amps is manufactured using Microchip’s state-of-theart CMOS process. It is designed for low-cost, lowpower and high-precision applications. Its low supply voltage, low quiescent current and wide bandwidth makes the MCP651/1S/2/3/4/5/9 ideal for batterypowered applications. 4.1 Calibration and Chip Select These op amps include circuitry for dynamic calibration of the offset voltage (VOS). 4.1.1 mCal CALIBRATION CIRCUITRY The internal mCal circuitry, when activated, starts a delay timer (to wait for the op amp to settle to its new bias point), then calibrates the input offset voltage (VOS). The mCal circuitry is triggered at power-up (and after some power brown-out events) by the internal POR, and by the memory’s Parity Detector. The powerup time, when the mCal circuitry triggers the calibration sequence, is 200 ms (typical). 4.1.2 CAL/CS PIN The CAL/CS pin gives the user a means to externally demand a Low-Power mode of operation, then to calibrate VOS. Using the CAL/CS pin makes it possible to correct VOS as it drifts over time (1/f noise and aging; see Figure 2-35) and across temperature. The CAL/CS pin performs two functions: it places the op amp(s) in a Low-Power mode when it is held high, and starts a calibration event (correction of VOS) after a rising edge. While in the Low-Power mode, the quiescent current is quite small (ISS = -3 µA, typical). The output is also in a High Z state. During the calibration event, the quiescent current is near, but smaller than, the specified quiescent current (6 mA, typical). The output continues in the High Z state, and the inputs are disconnected from the external circuit, to prevent internal signals from affecting circuit operation. The op amp inputs are internally connected to a Common mode voltage buffer and feedback resistors. The offset is corrected (using a digital state machine, logic and memory), and the calibration constants are stored in memory. Once the calibration event is completed, the amplifier is reconnected to the external circuitry. The turn-on time, when calibration is started with the CAL/CS pin, is 3 ms (typical). There is an internal 5 M pull-down resistor tied to the CAL/CS pin. If the CAL/CS pin is left floating, the amplifier operates normally. For the MCP655 dual and the MCP659 quad, there is an additional constraint on toggling the two CAL/CS pins close together; see the tCON specification in Table 1-3. If the two pins are toggled simultaneously, or if they are toggled separately with an adequate delay between them (greater than tCON), then the CAL/CS inputs are accepted as valid. If one of the two pins toggles while the other pin’s calibration routine is in progress, then an invalid input occurs and the result is unpredictable. 4.1.3 INTERNAL POR This part includes an internal Power-On Reset (POR) to protect the internal calibration memory cells. The POR monitors the power supply voltage (VDD). When the POR detects a low VDD event, it places the part into the Low-Power mode of operation. When the POR detects a normal VDD event, it starts a delay counter, then triggers an calibration event. The additional delay gives a total POR turn-on time of 200 ms (typical); this is also the power-up time (since the POR is triggered at power-up). 4.1.4 PARITY DETECTOR A parity error detector monitors the memory contents for any corruption. In the rare event that a parity error is detected (e.g., corruption from an alpha particle), a POR event is automatically triggered. This will cause the input offset voltage to be re-corrected, and the op amp will not return to normal operation for a period of time (the POR turn-on time, tPON). 4.1.5 CALIBRATION INPUT PIN A VCAL pin is available in some options (e.g., the single MCP651) for those applications that need the calibration to occur at an internally driven Common mode voltage other than VDD/3. Figure 4-1 shows the reference circuit that internally sets the op amp’s Common mode reference voltage (VCM_INT) during calibration (the resistors are disconnected from the supplies at other times). The 5 k resistor provides over-current protection for the buffer. 300 k VCM_INT 5 k VCAL BUFFER 150 k VSS FIGURE 4-1: Input Circuitry. DS20002146D-page 22 To op amp during calibration VDD Common-Mode Reference’s  2009-2014 Microchip Technology Inc. MCP651/1S/2/3/4/5/9 When the VCAL pin is left open, the internal resistor divider generates a VCM_INT of approximately VDD/3, which is near the center of the input Common mode voltage range. It is recommended that an external capacitor from VCAL to ground be added to improve noise immunity. When the VCAL pin is driven by an external voltage source, which is within its specified range, the op amp will have its input offset voltage calibrated at that Common mode input voltage. Make sure that VCAL is within its specified range. It is possible to use an external resistor voltage divider to modify VCM_INT; see Figure 4-2. The internal circuitry at the VCAL pin looks like 100 k tied to VDD/3. The parallel equivalent of R1 and R2 should be much smaller than 100 k to minimize differences in matching and temperature drift between the internal and external resistors. Again, make sure that VCAL is within its specified range. VDD MCP65X R1 VCAL C1 R2 VSS FIGURE 4-2: Resistors. 4.2.1 Input PHASE REVERSAL The input devices are designed to not exhibit phase inversion when the input pins exceed the supply voltages. Figure 2-41 shows an input voltage exceeding both supplies with no phase inversion. 4.2.2 VDD Bond Pad VIN+ Bond Pad INPUT VOLTAGE AND CURRENT LIMITS The ESD protection on the inputs can be depicted as shown in Figure 4-3. This structure was chosen to protect the input transistors, and to minimize input bias current (IB). The input ESD diodes clamp the inputs when they try to go more than one diode drop below VSS. They also clamp any voltages that go too far  2009-2014 Microchip Technology Inc. Bond VIN– Pad Input Stage VSS Bond Pad FIGURE 4-3: Structures. Simplified Analog Input ESD In order to prevent damage and/or improper operation of these amplifiers, the circuit must limit the currents (and voltages) at the input pins (see Section 1.1 “Absolute Maximum Ratings †”). Figure 4-4 shows the recommended approach to protecting these inputs. The internal ESD diodes prevent the input pins (VIN+ and VIN–) from going too far below ground, and the resistors R1 and R2 limit the possible current drawn out of the input pins. Diodes D1 and D2 prevent the input pins (VIN+ and VIN–) from going too far above VDD, and dump any currents onto VDD. When implemented as shown, resistors R1 and R2 also limit the current through D1 and D2. Setting VCM with External For instance, a design goal to set VCM_INT = 0.1V when VDD = 2.5V could be met with: R1 = 24.3 k, R2 = 1.00 k and C1 = 100 nF. This will keep VCAL within its range for any VDD, and should be close enough to 0V for ground-based applications. 4.2 above VDD; their breakdown voltage is high enough to allow normal operation, and low enough to bypass quick ESD events within the specified limits. VDD D1 V1 V2 R1 D2 MCP65X VOUT R2 VSS – (minimum expected V1) 2 mA VSS – (minimum expected V2) R2 > 2 mA R1 > FIGURE 4-4: Inputs. Protecting the Analog It is also possible to connect the diodes to the left of the resistor R1 and R2. In this case, the currents through the diodes D1 and D2 need to be limited by some other mechanism. The resistors then serve as in-rush current limiters; the DC current into the input pins (VIN+ and VIN–) should be very small. DS20002146D-page 23 When operating at very low non-inverting gains, the output voltage is limited at the top by the VCM range (< VDD – 1.3V); see Figure 4-5. +I SC Limited RL = 100Ω RL = 10Ω FIGURE 4-6: 120 100 80 60 40 0 20 -20 -40 -60 VOL Limited -80 The input stage of the MCP651/1S/2/3/4/5/9 op amps uses a differential PMOS input stage. It operates at low Common mode input voltage (VCM), with VCM up to VDD – 1.3V and down to VSS – 0.3V. The input offset voltage (VOS) is measured at VCM = VSS – 0.3V and VDD – 1.3V to ensure proper operation. See Figure 2-6 and Figure 2-7 for temperature effects. RL = 1 kΩ -ISC Limited NORMAL OPERATION VOH Limited (VDD = 5.5V) -100 4.2.3 6.0 5.5 5.0 4.5 4.0 3.5 3.0 2.5 2.0 1.5 1.0 0.5 0.0 -0.5 -120 A significant amount of current can flow out of the inputs (through the ESD diodes) when the Common mode voltage (VCM) is below ground (VSS); see Figure 2-15. Applications that are high-impedance may need to limit the usable voltage range. VOUT (V) MCP651/1S/2/3/4/5/9 IOUT (mA) Output Current. VDD VIN MCP65X VOUT V SS  V IN V OUT  VDD – 1.3V FIGURE 4-5: Unity-Gain Voltage Limitations for Linear Operation. 4.3 4.3.0.1 Rail-to-Rail Output Maximum Output Voltage The Maximum Output Voltage (see Figure 2-16 and Figure 2-17) describes the output range for a given load. For instance, the output voltage swings to within 15 mV of the negative rail with a 1 k load tied to VDD/2. 4.3.0.2 Output Current Figure 4-6 shows the possible combinations of output voltage (VOUT) and output current (IOUT). IOUT is positive when it flows out of the op amp into the external circuit. DS20002146D-page 24  2009-2014 Microchip Technology Inc. MCP651/1S/2/3/4/5/9 4.3.0.3 Power Dissipation Since the output short circuit current (ISC) is specified at ±100 mA (typical), these op amps are capable of both delivering and dissipating significant power. Two common loads, and their impact on the op amp’s power dissipation, will be discussed. Figure 4-7 shows a capacitive load (CL), which is driven by a sine wave with DC offset. The capacitive load causes the op amp to output higher currents at higher frequencies. Because the output rectifies IOUT, the op amp’s dissipated power increases (even though the capacitor does not dissipate power). Figure 4-7 shows a resistive load (RL) with a DC output voltage (VOUT). VL is RL’s ground point, VSS is usually ground (0V) and IOUT is the output current. The input currents are assumed to be negligible. VDD IDD IOUT VOUT MCP65X VDD IDD CL ISS IOUT VSS VOUT MCP65X RL ISS FIGURE 4-8: Diagram for Capacitive Load Power Calculations. The output voltage is assumed to be: VSS VL FIGURE 4-7: Diagram for Resistive Load Power Calculations. The DC currents are: EQUATION 4-4: V OUT = V DC + V AC sin   t  Where: VDC = DC offset (V) VAC = Peak output swing (VPK) EQUATION 4-1: VOUT – VL IOUT = -------------------------RL IDD  IQ + max  0, I OUT  ISS  – I Q + min  0, I OUT  Where: IQ = Quiescent supply current for one op amp (mA/amplifier) VOUT = A DC value (V)  = Radian frequency (2 f) (rad/s) The op amp’s currents are: EQUATION 4-5: dVOUT I OUT = C L  ----------------- = V AC  C L cos   t  dt I DD  I Q + max  0, I OUT  I SS  – I Q + min  0, I OUT  Where: The DC op amp power is: IQ = Quiescent supply current for one op amp (mA/amplifier) EQUATION 4-2: P OA = I DD  VDD – V OUT  + I SS  V SS – V OUT  The maximum op amp power, for resistive loads at DC, occurs when VOUT is halfway between VDD and VL or halfway between VSS and VL: The op amp’s instantaneous power, average power and peak power are: EQUATION 4-6: P OA = I DD  V DD – V OUT  + I SS  V SS – V OUT  EQUATION 4-3: max  P OA  = I DD  VDD – V SS  2 max  V DD – V L VL – VSS  + -----------------------------------------------------------------4R L  2009-2014 Microchip Technology Inc. 4VAC fCL ave  POA  =  V DD – V SS   IQ + ------------------------    max  P OA  =  V DD – V SS   IQ + 2V AC fC L  The power dissipated in a package depends on the powers dissipated by each op amp in that package: DS20002146D-page 25 MCP651/1S/2/3/4/5/9 EQUATION 4-7: n PPKG =  POA k=1 Where: n = Number of op amps in package (1 or 2) When driving large capacitive loads with these op amps (e.g., > 20 pF when G = +1), a small series resistor at the output (RISO in Figure 4-9) improves the feedback loop’s phase margin (stability) by making the output load resistive at higher frequencies. The bandwidth will be generally lower than the bandwidth with no capacitive load. The maximum ambient to junction temperature rise (TJA) and junction temperature (TJ) can be calculated using the maximum expected package power (PPKG), ambient temperature (TA) and the package thermal resistance (JA) found in Table 1-4: EQUATION 4-8: T JA = PPKG  JA T J = T A + T JA The worst-case power de-rating for the op amps in a particular package can be easily calculated: EQUATION 4-9: RG VOUT RN TA = Ambient temperature (°C) Several techniques are available to reduce TJA for a given package: • Reduce JA - Use another package - Improve the PCB layout (ground plane, etc.) - Add heat sinks and air flow • Reduce max(PPKG) - Increase RL - Decrease CL - Limit IOUT using RISO (see Figure 4-9) - Decrease VDD 4.4 4.4.1 Improving Stability CAPACITIVE LOADS Driving large capacitive loads can cause stability problems for voltage feedback op amps. As the load capacitance increases, the feedback loop’s phase margin decreases and the closed-loop bandwidth is reduced. This produces gain peaking in the frequency response, with overshoot and ringing in the step response. See Figure 2-30. A unity-gain buffer (G = +1) is the most sensitive to capacitive loads, though all gains show the same general behavior. DS20002146D-page 26 MCP65X FIGURE 4-9: Output Resistor, RISO Stabilizes Large Capacitive Loads. Figure 4-10 gives recommended RISO values for different capacitive loads and gains. The x-axis is the normalized load capacitance (CL/GN), where GN is the circuit’s noise gain. For non-inverting gains, GN and the Signal Gain are equal. For inverting gains, GN is 1+|Signal Gain| (e.g., -1 V/V gives GN = +2 V/V). 100 Recommended R ISO (Ω) TJmax = Absolute maximum junction temperature (°C) RISO CL T Jmax – T A P PKG  ------------------------- JA Where: RF 10 GN = +1 GN  +2 1 10p 1.E-11 100p 1n 1.E-10 1.E-09 Normalized Capacitance; CL/GN (F) 10n 1.E-08 FIGURE 4-10: Recommended RISO Values for Capacitive Loads. After selecting RISO for your circuit, double check the resulting frequency response peaking and step response overshoot. Modify RISO’s value until the response is reasonable. Bench evaluation and simulations with the MCP651/1S/2/3/4/5/9 SPICE macro model are helpful. 4.4.2 GAIN PEAKING Figure 4-11 shows an op amp circuit that represents non-inverting amplifiers (VM is a DC voltage and VP is the input) or inverting amplifiers (VP is a DC voltage and VM is the input). The capacitances CN and CG represent the total capacitance at the input pins; they include the op amp’s Common mode input capacitance (CCM), board parasitic capacitance and any capacitor placed in parallel.  2009-2014 Microchip Technology Inc. MCP651/1S/2/3/4/5/9 EQUATION 4-10: RN VP CN Given: G N1 = 1 + R F  R G MCP65X G N2 = 1 + C G  C F VOUT VM RG FIGURE 4-11: Capacitance. CG f F = 1   2  RF C F  f Z = f F  G N1  G N2  We need: f F  f GBWP   2G N2  , G N1 < GN2 RF f F  f GBWP   4G N1  , G N1 > GN2 Amplifier with Parasitic CG acts in parallel with RG (except for a gain of +1 V/V), which causes an increase in gain at high frequencies. CG also reduces the phase margin of the feedback loop, which becomes less stable. This effect can be reduced by either reducing CG or RF. CN and RN form a low-pass filter that affects the signal at VP. This filter has a single real pole at 1/(2RNCN). The largest value of RF that should be used depends on noise gain (see GN in Section 4.4.1 “Capacitive Loads”) and CG. Figure 4-12 shows the maximum recommended RF for several CG values. Maximum Recommended RF (Ω) 1.E+05 100k 4.5 Power Supply With this family of operational amplifiers, the Power Supply pin (VDD for single supply) should have a local bypass capacitor (i.e., 0.01 µF to 0.1 µF) within 2 mm for good high-frequency performance. Surface mount, multilayer ceramic capacitors, or their equivalent, should be used. These op amps require a bulk capacitor (i.e., 2.2 µF or larger) within 50 mm to provide large, slow currents. Tantalum capacitors, or their equivalent, may be a good choice. This bulk capacitor can be shared with other nearby analog parts as long as crosstalk through the supplies does not prove to be a problem. GN > +1 V/V CG = 10 pF CG = 32 pF CG = 100 pF CG = 320 pF CG = 1 nF 1.E+04 10k 4.6 1k 1.E+03 1.E+02 100 1 FIGURE 4-12: RF vs. Gain. 10 Noise Gain; GN (V/V) 100 Maximum Recommended Figure 2-37 and Figure 2-38 show the small signal and large signal step responses at G = +1 V/V. The unitygain buffer usually has RF = 0 and RG open. Figure 2-39 and Figure 2-40 show the small signal and large signal step responses at G = -1 V/V. Since the noise gain is 2 V/V and CG  10 pF, the resistors were chosen to be RF = RG = 499 and RN = 249. It is also possible to add a capacitor (CF) in parallel with RF to compensate for the de-stabilizing effect of CG. This makes it possible to use larger values of RF. The conditions for stability are summarized in Equation 410.  2009-2014 Microchip Technology Inc. High-Speed PCB Layout These op amps are fast enough that a little extra care in the PCB (Printed Circuit Board) layout can make a significant difference in performance. Good PCB layout techniques will help achieve the performance shown in the specifications and Typical Performance Curves; it will also help minimize EMC (Electro-Magnetic Compatibility) issues. Use a solid ground plane. Connect the bypass local capacitor(s) to this plane with minimal length traces to cut down inductive and capacitive crosstalk. Separate digital from analog, low-speed from highspeed, and low-power from high-power. This will reduce interference. Keep sensitive traces short and straight. Separate them from interfering components and traces. This is especially important for high-frequency (low rise time) signals. Sometimes, it helps to place guard traces next to victim traces. They should be on both sides of the victim trace, and as close as possible. Connect guard traces to ground plane at both ends, and in the middle for long traces. Use coax cables, or low inductance wiring, to route signal and power to and from the PCB. Mutual and self inductance of power wires is often a cause of crosstalk and unusual behavior. DS20002146D-page 27 MCP651/1S/2/3/4/5/9 4.7 Typical Applications 4.7.1 4.7.3 POWER DRIVER WITH HIGH GAIN Figure 4-13 shows a power driver with high gain (1 + R2/R1). The MCP651/1S/2/3/4/5/9 op amp’s shortcircuit current makes it possible to drive significant loads. The calibrated input offset voltage supports accurate response at high gains. R3 should be small, and equal to R1||R2, in order to minimize the bias current induced offset. R1 VDD/2 R2 H-BRIDGE DRIVER Figure 4-15 shows the MCP652 dual op amp used as a H-bridge driver. The load could be a speaker or a DC motor. ½ MCP652 VIN RF RL RGT VOUT RGB RL R3 VOT RF RF VOB VIN MCP65X FIGURE 4-13: 4.7.2 VDD/2 OPTICAL DETECTOR AMPLIFIER Figure 4-14 shows a transimpedance amplifier, using the MCP651 op amp, in a photo detector circuit. The photo detector is a capacitive current source. The op amp’s input Common mode capacitance (5 pF, typical) acts in parallel with CD. RF provides enough gain to produce 10 mV at VOUT. CF stabilizes the gain and limits the transimpedance bandwidth to about 1.1 MHz. RF’s parasitic capacitance (e.g., 0.2 pF for a 0805 SMD) acts in parallel with CF. CF 1.5 pF Photo Detector ID 100 nA ½ MCP652 Power Driver. RF 100 k CD 30 pF H-Bridge Driver. This circuit automatically makes the noise gains (GN) equal, when the gains are set properly, so that the frequency responses match well (in magnitude and in phase). Equation 4-11 shows how to calculate RGT and RGB so that both op amps have the same DC gains; GDM needs to be selected first. EQUATION 4-11: V OT – V OB G DM  --------------------------------  2 V/V VIN – VDD  2 RF RGT = -------------------------------- G DM  2  – 1 RF RGB = ------------------G DM  2 VOUT MCP651 VDD/2 FIGURE 4-14: Transimpedance Amplifier for an Optical Detector. DS20002146D-page 28 FIGURE 4-15: Equation 4-12 gives the resulting Common mode and Differential mode output voltages. EQUATION 4-12: VOT + V OB VDD --------------------------- = ----------2 2 VDD V OT – VOB = GDM  V IN – -----------  2   2009-2014 Microchip Technology Inc. MCP651/1S/2/3/4/5/9 5.0 DESIGN AIDS Microchip provides the basic design aids needed for the MCP651/1S/2/3/4/5/9 family of op amps. 5.1 SPICE Macro Model The latest SPICE macro model for the MCP651/1S/2/3/4/5/9 op amps is available on the Microchip web site at www.microchip.com. This model is intended to be an initial design tool that works well in the op amp’s linear region of operation over the temperature range. See the model file for information on its capabilities. Bench testing is a very important part of any design and cannot be replaced with simulations. Also, simulation results using this macro model need to be validated by comparing them to the data sheet specifications and characteristic curves. 5.2 FilterLab® Software Microchip’s FilterLab® software is an innovative software tool that simplifies analog active filter (using op amps) design. Available at no cost from the Microchip web site at www.microchip.com/filterlab, the FilterLab design tool provides full schematic diagrams of the filter circuit with component values. It also outputs the filter circuit in SPICE format, which can be used with the macro model to simulate actual filter performance. 5.3 Microchip Advanced Part Selector (MAPS) MAPS is a software tool that helps efficiently identify Microchip devices that fit a particular design requirement. Available at no cost from the Microchip web site at www.microchip.com/maps, the MAPS is an overall selection tool for Microchip’s product portfolio that includes Analog, Memory, MCUs and DSCs. Using this tool, a customer can define a filter to sort features for a parametric search of devices and export side-by-side technical comparison reports. Helpful links are also provided for data sheets, purchase and sampling of Microchip parts. 5.4 Some boards that are especially useful are: • • • • • • MCP6XXX Amplifier Evaluation Board 1 MCP6XXX Amplifier Evaluation Board 2 MCP6XXX Amplifier Evaluation Board 3 MCP6XXX Amplifier Evaluation Board 4 Active Filter Demo Board Kit 8-Pin SOIC/MSOP/TSSOP/DIP Evaluation Board, P/N SOIC8EV 5.5 Application Notes The following Microchip Application Notes are available on the Microchip web site at www.microchip. com/appnotes and are recommended as supplemental reference resources. • ADN003: “Select the Right Operational Amplifier for your Filtering Circuits” (DS21821) • AN722: “Operational Amplifier Topologies and DC Specifications” (DS00722) • AN723: “Operational Amplifier AC Specifications and Applications” (DS00723) • AN884: “Driving Capacitive Loads With Op Amps” (DS00884) • AN990: “Analog Sensor Conditioning Circuits – An Overview” (DS00990) • AN1177: “Op Amp Precision Design: DC Errors” (DS01177) • AN1228: “Op Amp Precision Design: Random Noise” (DS01228) • AN1332: “Current Sensing Circuit Concepts and Fundamentals” (DS01332) Some of these application notes, and others, are listed in the design guide: • “Signal Chain Design Guide” (DS21825) Analog Demonstration and Evaluation Boards Microchip offers a broad spectrum of Analog Demonstration and Evaluation Boards that are designed to help customers achieve faster time to market. For a complete listing of these boards and their corresponding user’s guides and technical information, visit the Microchip web site at www.microchip.com/analog tools.  2009-2014 Microchip Technology Inc. DS20002146D-page 29 MCP651/1S/2/3/4/5/9 6.0 PACKAGING INFORMATION 6.1 Package Marking Information Example: 5-Lead SOT-23 (2x3) (MCP651S) XXNN YW25 6-Lead SOT-23 (2x3) (MCP653) Example: XXNN JD25 Example: 8-Lead TDFN(2x3) (MCP651) AAZ 124 25 Example: 8-Lead DFN (3x3) (MCP652) XXXX YYWW NNN Legend: XX...X Y YY WW NNN e3 * Note: DS20002146D-page 30 Device MCP652 Code DABP Note: Applies to 8-Lead 3x3 DFN DABP 1124 256 Customer-specific information Year code (last digit of calendar year) Year code (last 2 digits of calendar year) Week code (week of January 1 is week ‘01’) Alphanumeric traceability code Pb-free JEDEC® designator for Matte Tin (Sn) This package is Pb-free. The Pb-free JEDEC designator ( e3 ) can be found on the outer packaging for this package. In the event the full Microchip part number cannot be marked on one line, it will be carried over to the next line, thus limiting the number of available characters for customer-specific information.  2009-2014 Microchip Technology Inc. MCP651/1S/2/3/4/5/9 6.2 Package Marking Information 8-Lead SOIC (150 mil) (MCP651, MCP652) XXXXXXXX XXXXYYWW NNN 10-Lead DFN (3x3) (MCP655) XXXX YYWW NNN 10-Lead MSOP (MCP655) XXXXXX YWWNNN 14-Lead SOIC (MCP654) XXXXXXXXXXX XXXXXXXXXXX YYWWNNN 14-Lead TSSOP (MCP654) XXXXXXXX YYWW NNN 16-Lead QFN (4x4) (MCP659) Example: MCP651E SN e3 1124 256 Example: BAFC 1124 256 Example: 655EUN 124256 Example: MCP654 E/SL e3 1124256 Example: 654E/ST 1124 256 Example: 659 XXXXXXX XXXXXXX YWWNNN  2009-2014 Microchip Technology Inc. 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