MCP6D11-E/MG

MCP6D11 Low-Noise, Precision, 90 MHz Differential I/O Amplifier Features General Description • Low Power - IQ: 1.4 mA - Supply Voltage Range: 2.5V to 5.5V • Gain-Bandwidth Product: 90 MHz • Slew Rate: 25V/µs • Low Noise: 5.0 nV/√Hz, f = 10 kHz • Low Distortion (2Vp-p, 10 kHz): - HD2: -138 dBc - HD3: -137 dBc • Fast Settling: 200 ns to 0.01% • Low Offset: 150 µV Max. • Power-Down Function • Input Vcm-Range Includes Negative Rail • Rail-to-Rail Output • Small Packages: MSOP-8, 3 x 3 mm QFN-16 • Extended Temperature Range: -40°C to +125°C The MCP6D11 from Microchip Technology is a lownoise, low-distortion Differential I/O Amplifier optimized for driving high-performance 14-, and 16-Bit SAR ADCs, such as the MCP331x1D ADC family. Featuring a low 5.0 nV/√Hz input-referred voltage noise and distortion of less than -116 dBc with an input signal of up to 100 kHz (2Vpp), the MCP6D11 consumes only 3.5 mW of quiescent power on a 2.5V supply. For power sensitive applications, a Power-Down function reduces the power consumption to less than 13 µW. Typical Applications • Precision ADC Driver: - 14/16/18-bit SAR ADCs - Delta-Sigma ADCs • Single-Ended to Differential Conversion • Differential Active Filter • Line Drivers Through its VOCM pin, the MCP6D11 allows easy control of its output common-mode voltage, which can be set independently of the input common-mode voltage. This, coupled with an input common-mode range that extends below the negative supply rail, and a near rail-to-rail output swing capability, results in a simple driver amplifier solution for a variety of ADCs. The MCP6D11 is the ideal interface solution for converting single-ended, ground-referenced signal sources into a fully differential output signal required to preserve the high performance of today’s ultra-low distortion, single-supply ADCs. The MCP6D11 is specified for the -40°C to +125°C temperature range and is available in QFN-16 (3 x 3 mm) and MSOP-8 package options. MCP6D11 Harmonic Distortion with a 10 kHz, 2Vpp Signal Design Aids • Microchip Advanced Part Selector (MAPS) • Application Notes Related Parts • MCP331x1D SAR ADCs  2019 Microchip Technology Inc. DS20006162A-page 1 MCP6D11 NOTES: DS20006162A-page 2  2019 Microchip Technology Inc. MCP6D11 1.0 ELECTRICAL CHARACTERISTICS Absolute Maximum Ratings † VDD – VSS .................................................................................................................................................................6.0V Supply Voltage Turn-on/off max. dV/dt at +25°C .............................................................................................. ±0.35V/µs Supply Voltage Turn-on/off max. dV/dt at +125°C .............................................................................................. ±0.2V/µs Current at all Inputs (continuous) ........................................................................................................................ ±10 mA Voltage at all Inputs and Outputs ...............................................................................................VSS – 0.3V to VDD+0.3V Differential Input Voltage ......................................................................................................................................... ±1.0V Current at Output and Supply Pins (continuous)...................................................................................................±20 mA Storage Temperature .............................................................................................................................-65°C to +150°C Maximum Junction Temperature .......................................................................................................................... +155°C ESD protection on all pins (HBM, CDM)  4 kV, 2 kV † Notice: Stresses above those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. This is a stress rating only and functional operation of the device at those or any other conditions above those indicated in the operational listings of this specification is not implied. Exposure to maximum rating conditions for extended periods may affect device reliability. AC ELECTRICAL CHARACTERISTICS Electrical Characteristics: Unless otherwise indicated, TA = 25°C, VDD = +2.5V to 5.5V, VSS = 0V, PD\ = VDD, VOCM = open, VICM = VDD/2, Single-ended input, G = 1V/V, RF = RG = 1 k, RL = 1 k between differential outputs. Parameters Sym. Min. Typ. Max. Units Test Level (Note 1) Conditions AC Response Gain-Bandwidth Product Bandwidth (Small-signal, -3 dB) Bandwidth (Large-signal, -3 dB) GBWP — 90 — MHz C G = 10V/V BW — 82 — MHz C G = 1V/V, VOUT_DM = 20mVpp, VDD = 5V — 80 — G = 1V/V, VOUT_DM = 20mVpp, VDD = 3V — 48 — G = 2V/V, VOUT_DM = 20mVpp, VDD = 5V — 10 — — 6.4 — MHz C G = 1V/V, VOUT_DM = 2Vpp, VDD = 5V — 4 — MHz C G = 1V/V, VOUT_DM = 2Vpp, VDD = 5V — 27 — V/µs C G = 1V/V, VOUT_DM = 2V step, VDD = 5V — 26 — G = 2V/V, VOUT_DM = 2V step, VDD = 5V — 24 — G = 1V/V, VOUT_DM = 2V step, VDD = 3V — 23 — — 160 — BW Bandwidth for 0.1 dB Gain Flatness Slew Rate (differential) Settling Time to 0.1% SR tS Settling Time to 0.01% Rise and Fall Times Overdrive Recovery Time G = 10V/V, VOUT_DM = 20mVpp, VDD = 5V G = 2V/V, VOUT_DM = 2V step, VDD = 3V ns C G = 1V/V, VOUT_DM = 2V step, VDD = 5V — 170 — G = 1V/V, VOUT_DM = 2V step, VDD = 3V — 200 — G = 1V/V, VOUT_DM = 2V step, VDD = 5V G = 1V/V, VOUT_DM = 2V step, VDD = 3V — 215 — tR, tF — 34 — ns C VOUT_DM = 1Vpp step trec — 150 — ns C VDD = 5V, 0.5V overdrive — 200 — VDD = 3V, 0.5V overdrive Closed-Loop Output Impedance Rout — 0.1 —  C f = 1 MHz, differential Output Balance Error BAL — 85 — dB C At DC Note 1: “Test Level” designation: A = 100% production tested at 25°C; B = not production tested, limits set by characterization and/or simulation, C = values for information only (based on characterization or design).  2019 Microchip Technology Inc. DS20006162A-page 3 MCP6D11 AC ELECTRICAL CHARACTERISTICS (CONTINUED) Electrical Characteristics: Unless otherwise indicated, TA = 25°C, VDD = +2.5V to 5.5V, VSS = 0V, PD\ = VDD, VOCM = open, VICM = VDD/2, Single-ended input, G = 1V/V, RF = RG = 1 k, RL = 1 k between differential outputs. Parameters Sym. Min. eni — Units Test Level (Note 1) — nV/√Hz C Typ. Max. Conditions Noise and Distortion Input Noise Voltage Density Input Noise Current Density 2nd Order Harmonic Distortion 3rd Order Harmonic Distortion Note 1: 5.0 at f > 10 kHz ini — 0.6 — pA/√Hz C at f ≥ 100 kHz HD2 — -137 — dBc C f = 10 kHz, 2Vpp, VDD = 3V — -138 — f = 10 kHz, 2Vpp, VDD = 5V — -118 — f = 100 kHz, 2Vpp, VDD = 3V HD3 f = 100 kHz, 2Vpp, VDD = 5V — -120 — — -137 — — -137 — f = 10 kHz, 2Vpp, VDD = 5V — -116 — f = 100 kHz, 2Vpp, VDD = 3V — -116 — f = 100 kHz, 2Vpp, VDD = 5V dBc C f = 10 kHz, 2Vpp, VDD = 3V “Test Level” designation: A = 100% production tested at 25°C; B = not production tested, limits set by characterization and/or simulation, C = values for information only (based on characterization or design). DC ELECTRICAL CHARACTERISTICS Electrical Characteristics: Unless otherwise indicated, TA = 25°C, VDD = +2.5V to 5.5V, VSS = 0V, PD\ = VDD, VOCM = open, VICM = VDD/2, Single-ended input, G = 1V/V, RF = RG = 1 kRL = 1 kbetween differential outputs. Parameters Sym. Min. Typ. Max. Units Test Level (Note 1) Conditions Input Offset VOS -150 ±25 +150 µV A Input Offset Voltage Drift VOS/TA -2.0 ±0.5 +2.0 µV/°C B Power Supply Rejection PSRR 100 117 — dB A IB+, IB- -1.7 -0.7 — µA A (Note 3) -1.9 -0.8 -0.4 µA B TA = +85°C Input Offset Voltage TA = -40°C to +125°C (Note 4) Input Bias Current and Impedance Input Bias Current Input Bias Current, across Temperature Input Bias Current Drift Input Offset Current Input Offset Current Drift Differential Input Impedance Note 1: 2: 3: 4: IB/TA -2.1 -1.1 -0.5 µA B TA = +125°C -3.6 ±2.75 +3.6 nA/°C B TA = -40°C to +125°C (QFN) (Note 4) IOS -60 ±10 +60 nA A IOS/TA -130 ±40 +130 pA/°C B ZDIFF — 88||1.0 — k||pF C TA = -40°C to +125°C (QFN) (Note 4) “Test Level” designation: A = 100% production tested at 25°C; B = not production tested, limits set by characterization and/or simulation, C = values for information only (based on characterization or design). The VICM spec is supported by the CMRR tests. Negative polarity sign indicates current flowing out of node. Based on data taken at the temperature range end-points (-40°C, +125°C) and calculated deltas are divided by the temperature range. The Max./Min. specifications are set using +/-4 standard deviations on the device distribution. DS20006162A-page 4  2019 Microchip Technology Inc. MCP6D11 DC ELECTRICAL CHARACTERISTICS (CONTINUED) Electrical Characteristics: Unless otherwise indicated, TA = 25°C, VDD = +2.5V to 5.5V, VSS = 0V, PD\ = VDD, VOCM = open, VICM = VDD/2, Single-ended input, G = 1V/V, RF = RG = 1 kRL = 1 kbetween differential outputs. Sym. Min. Typ. Max. Units Test Level (Note 1) Common-Mode Input Range, high VICM_H VDD - 1.0 VDD - 0.9 — V A VDD - 1.0 — B TA = -40°C to + 125°C Common-Mode Input Range, low VICM_L — VSS - 0.25 VSS - 0.15 V A (Note 2) — VSS - 0.1 VSS B TA = -40°C to + 125°C Common-Mode Rejection Ratio CMRR 95 112 — dB A VDD = 5.5V 95 110 — B VDD = 5.5V TA = -40°C to + 125°C 90 107 — A VDD = 2.5V 90 105 — B VDD = 2.5V TA = -40°C to + 125°C 106 124 — A VDD = 5.5V VOUT = 0.4V to VDD – 0.4V 102 118 — B VDD = 5.5V, TA = -40°C to +125°C 106 121 — A VDD = 2.5V, VOUT = 0.25V to VDD – 0.25V 102 114 — B VDD = 2.5V, TA = -40°C to +125°C Internal Feedback Trace Resistance — 3 —  C QFN package only; pins 10-4, 11-1 Internal Feedback Trace Resistance Mismatch — 0.05 —  C QFN package only; pins 10-4, 11-1 — VSS + 75 VSS + 100 mV A VDD = 5.5V, 0.5V overdrive — VSS + 33 VSS + 50 VDD = 2.5V, 0.5V overdrive VDD - 150 VDD - 100 — VDD = 5.5V, 0.5V overdrive VDD - 75 VDD - 50 — VDD = 2.5V, 0.5V overdrive — ±66 — — ±75 — Parameters Conditions Common-Mode VDD - 1.2 (Note 2) Open Loop Gain DC Open Loop Gain AOL dB Output Maximum Output Voltage Swing VOL VOH Output Short Circuit Current Note 1: 2: 3: 4: ISC mA C VDD = 2.5V VDD = 5.5V “Test Level” designation: A = 100% production tested at 25°C; B = not production tested, limits set by characterization and/or simulation, C = values for information only (based on characterization or design). The VICM spec is supported by the CMRR tests. Negative polarity sign indicates current flowing out of node. Based on data taken at the temperature range end-points (-40°C, +125°C) and calculated deltas are divided by the temperature range. The Max./Min. specifications are set using +/-4 standard deviations on the device distribution.  2019 Microchip Technology Inc. DS20006162A-page 5 MCP6D11 DC ELECTRICAL CHARACTERISTICS (CONTINUED) Electrical Characteristics: Unless otherwise indicated, TA = 25°C, VDD = +2.5V to 5.5V, VSS = 0V, PD\ = VDD, VOCM = open, VICM = VDD/2, Single-ended input, G = 1V/V, RF = RG = 1 kRL = 1 kbetween differential outputs. Parameters Min. Typ. Max. Units Test Level (Note 1) VCM 1.0 0.9 to VDD - 1.0 VDD - 1.1 V A GOCM 0.99 1 1.01 V/V B VOS -5 ±0.8 +5 mV A VOCM pin driven to (VDD/2) -10 ±2 +10 VOS/TA -45 ±15 +30 µV/°C B TA = -40°C to +125°C, VDD = 2.5V (QFN) (Note 4) -25 ±6.9 +25 B TA = -40°C to +125°C, VDD = 5.5V (QFN) (Note 4) Sym. Conditions Output Common-Mode Voltage Control (VOCM) Input Voltage Range Gain Input Offset Voltage Input Offset Voltage Drift Input Bias Current Input Impedance Bandwidth (Small-Signal, -3 dB) Slew Rate VOCM pin floating IB — ±0.01 +1.5 µA B ZCM — 46k||2 — ||pF B For internal VDD/2 reference BWss — 45 — MHz C VOCM = 100mVpp SR — 14 — V/µs C 1V step VDD 2.5 — 5.5 V A VSS = 0V IQ 1.0 1.49 1.8 mA A VDD = 5.5V, IO = 0 0.9 — 2.1 B VDD = 5.5V, TA = -40°C to +125°C Power Supply Supply Voltage Quiescent Current Quiescent Current Drift Power-Up Time 1.0 1.4 1.8 A VDD = 2.5V, IO = 0 0.8 — 2.0 B VDD = 2.5V, TA = -40°C to +125°C IQ/TA — 3.8 — µA/°C C TA = -40°C to +125°C (Note 4) tup — 30 — µs C Power-Down (PD\) IQ_PD — 5 7 µA A Input Voltage, Logic High VIH 0.8 VDD — VDD V A Input Voltage, Logic Low VIL VSS — 0.2 VDD V A Input Current, Logic High IIH — +5 — nA B Input Current, Logic Low IIL — -5 — nA B PD\ = VSS Turn-on Time ton — 1.0 1.5 µs B VDD = 5.5V, Vout = 90% of final value — 2.5 3 — 0.04 0.05 — 0.05 0.06 Quiescent Current Turn-off Time Note 1: 2: 3: 4: toff PD\ = VSS PD\ = VDD VDD = 2.5V, Vout = 90% of final value µs B VDD = 5.5V, Vout = 10% of final value VDD = 2.5V, Vout = 10% of final value “Test Level” designation: A = 100% production tested at 25°C; B = not production tested, limits set by characterization and/or simulation, C = values for information only (based on characterization or design). The VICM spec is supported by the CMRR tests. Negative polarity sign indicates current flowing out of node. Based on data taken at the temperature range end-points (-40°C, +125°C) and calculated deltas are divided by the temperature range. The Max./Min. specifications are set using +/-4 standard deviations on the device distribution. DS20006162A-page 6  2019 Microchip Technology Inc. MCP6D11 TEMPERATURE SPECIFICATIONS Electrical Characteristics: Unless otherwise indicated, all limits are specified for VDD - VSS = 2.5V to 5.5V Parameters Sym Min Typ Max Units Conditions Temperature Ranges Specified Temperature Range TA -40 — +125 °C Operating Temperature Range TA -40 — +125 °C Storage Temperature Range TA -65 — +150 °C Thermal Resistance, 8L-MSOP JA — 170 — °C/W Thermal Resistance, 16L-QFN JA — 60 — °C/W (Note 1) Thermal Package Resistances Note 1: The MCP6D11 operates over this temperature range, but the Junction Temperature (TJ) must not exceed the Absolute Maximum specification of +155°C.  2019 Microchip Technology Inc. DS20006162A-page 7 MCP6D11 NOTES: DS20006162A-page 8  2019 Microchip Technology Inc. MCP6D11 2.0 PIN DESCRIPTIONS TABLE 2-1: PIN FUNCTION TABLE MCP6D11 Symbol Description MSOP-8 QFN-16 1 3 IN- Negative input (Summing Junction) 2 9 VOCM Output common-mode voltage; a high impedance input 3 5,6,7,8 4 10 VDD Positive Power Supply OUT+ Positive output OUT- Negative output 5 11 6 13,14,15,16 VSS Negative Power Supply 7 12 PD\ Power-Down (Low = VSS = Low Power mode; High = VDD = normal operation) Positive Input (Summing Junction) 8 2 IN+ -- 1 FB- Negative feedback; same signal as negative output (OUT-) -- 4 FB+ Positive feedback; same signal as positive output (OUT+) 17 EP ‘Exposed Thermal Pad’ on bottom side of QFN package only (Note 1) -Note 1: The exposed thermal pad should be soldered to a low-noise ground or power plane. This pad is electrically isolated from the die (using non-conductive die attach), however the pad must be connected to a power or ground and can not be left floating. Package Types MCP6D11, MSOP-8 13 VSS 14 VSS 15 VSS 16 VSS MCP6D11, QFN-16 IN- 1 8 IN+ VOCM 2 7 PD FB- 1 6 VSS IN+ 2 11 OUT- 5 OUT- IN- 3 10 OUT+ FB+ 4 9 VOCM 12 PD VDD 8 VDD 7 VDD 5  2019 Microchip Technology Inc. VDD 6 VDD 3 OUT+ 4 EP(17) DS20006162A-page 9 MCP6D11 NOTES: DS20006162A-page 10  2019 Microchip Technology Inc. MCP6D11 3.0 TYPICAL PERFORMANCE CURVES Note: The graphs and tables provided following this note are a statistical summary based on a limited number of samples and are provided for informational purposes only. The performance characteristics listed herein are not tested or guaranteed. In some graphs or tables, the data presented may be outside the specified operating range (e.g., outside specified power supply range) and therefore outside the warranted range. Note: Unless otherwise indicated, TA = 25°C, VDD - VSS = 2.5V to 5.5V, VOCM = open, VICM = mid-supply, PD\ = VDD, single-ended input, 50 input match, G = 1V/V, RF = RG = 1 kCL = 0 pF and RL = 1 k between the differential outputs. 3.1 Frequency Response 12 3 VDD = 3.0V VOUT = 20mVpp VOUT = 100mVpp VDD = 5.0V Normalized Gain (dB) 9 G = 0.1V/V 0 6 Gain (dB) 3 0 -3 -6 -3 -6 G = 1V/V G = 2V/V G = 10V/V -9 -9 -12 100k 1M VOUT = 1Vpp VOUT = 2Vpp VOUT = 4Vpp 100 -12 100M 100k 1M Frequency (Hz) FIGURE 3-1: Small-Signal Frequency Response vs. Gain, VOUTDM = 20mVpp. 100M FIGURE 3-4: Small-Signal Frequency Response vs. VOUTDM (Gain = 1V/V). 9 3 VDD = 5.0V VDD = 3.0V 2 6 1 G = 0.1V/V 0 3 VOCM = 0.90V VOCM = 1.25V VOCM = 1.50V VOCM = 1.75V VOCM = 2.00V -1 Gain (dB) Normalized Gain (dB) 100 Frequency (Hz) 0 -3 -6 100 1M 1M -4 -5 -7 -8 -12 100k -3 -6 G = 1V/V G = 2V/V G = 10V/V -9 -2 -9 100M 100k 1M Frequency (Hz) 100 100M Frequency (Hz) FIGURE 3-2: Small-Signal Frequency Response vs. Gain, VOUTDM = 20mVpp. FIGURE 3-5: Small-Signal Frequency Response vs VOCM, Gain = 1V/V. 3 3 VOUT = 100mVpp VOUT = 20mVpp VDD = 3.0V 2 VDD = 5.0V 1 0 Gain (dB) Gain (dB) 0 -1 -3 -6 -9 VOUT = 1Vpp VOUT = 2Vpp VOUT = 4Vpp -2 -3 -4 VOCM = 0.90V VOCM = 1.50V VOCM = 2.50V VOCM = 3.50V VOCM = 4.00V -5 -6 -7 -8 -12 100k -9 1M 10M 100M 100k FIGURE 3-3: Small-Signal Frequency Response vs. VOUTDM (Gain = 1V/V).  2019 Microchip Technology Inc. 1M 100 100M Frequency (Hz) Frequency (Hz) FIGURE 3-6: Small-Signal Frequency Response vs VOCM, Gain = 1V/V. DS20006162A-page 11 MCP6D11 Note: Unless otherwise indicated, TA = 25°C, VDD - VSS = 2.5V to 5.5V, VOCM = open, VICM = mid-supply, PD\ = VDD, single-ended input, 50 input match, G = 1V/V, RF = RG = 1 kCL = 0 pF and RL = 1 k between the differential outputs. 9 6 VDD = 5.0V 6 3 3 0 Gain (dB) Gain (dB) VDD = 3.0V 0 RL = 50: RL = 100: RL = 200: RL = 500: RL = 1 k: -3 -6 -3 CL = 2 pF, RISO = 0: CL = 10 pF, RISO = 191: CL = 47 pF, RISO = 75: CL = 100 pF, RISO = 62: CL = 330 pF, RISO = 33: -6 -9 -12 -9 100k 1M 100 100k 100M 1M 1M 100M Frequency (Hz) Frequency (Hz) FIGURE 3-7: Small-Signal Frequency Response vs. R-Loads, Gain = 1V/V. FIGURE 3-10: Small-Signal Frequency Response vs. C-Loads. 9 3 VDD = 5.0V 2 6 1 3 -1 VDD = 3.0V Gain (dB) Gain (dB) 0 0 RL = 50: RL = 100: RL = 200: RL = 500: RL = 1 k: -3 -6 -2 -3 -4 RL = 50: RL = 100: RL = 200: RL = 500: RL = 1 k: -5 -6 -7 -8 -9 -9 100k 1M 100 100k 100M 1M 1M 100M Frequency (Hz) Frequency (Hz) FIGURE 3-8: Small-Signal Frequency Response vs R-Loads, Gain = 1V/V. FIGURE 3-11: Large-Signal Frequency Response vs. R-Loads, VOUTDM = 2Vpp. 3 6 VDD = 3.0V 2 3 1 0 -1 VDD = 5.0V Gain (dB) Gain (dB) 0 -3 -6 CL = 2 pF, RISO = 0: CL = 10 pF, RISO = 191: CL = 47 pF, RISO = 93: CL = 100 pF, RISO = 62: CL = 330 pF, RISO = 36: -9 -2 -3 -4 RL = 50: RL = 100: RL = 200: RL = 500: RL = 1 k: -5 -6 -7 -8 -9 -12 100k 1M 1M 100M Frequency (Hz) FIGURE 3-9: Small-Signal Frequency Response vs C-Loads. DS20006162A-page 12 100k 1M 100 100M Frequency (Hz) FIGURE 3-12: Large-Signal Frequency Response vs. R-Loads, VOUTDM = 2Vpp.  2019 Microchip Technology Inc. MCP6D11 6 Open-loop Gain (dB) 3 Gain (dB) 0 -3 -6 -9 100mVpp, 2.5V 100mVpp, 5.5V 1Vpp, 2.5V 1Vpp, 5.5V -12 -15 0 140 -30 120 -60 100 -90 80 -120 60 -150 Gain 5.5V 40 1M 10M -270 1 100M 100 10N 100M 10 )UHquency(H]) FIGURE 3-16: Differential Open-Loop Gain and Phase vs. Frequency (Simulation). 90 110 85 Output Balance (dB) 120 100 CMRR (dB) -240 Phase 2.5V -20 FIGURE 3-13: VOCM Small- and Large-Signal Frequency Response, Gain = 1V/V. 90 80 70 60 CMRR 2.5V CMRR 5.5V 80 75 70 65 60 55 VDD = 5.V VDD = V 50 45 40 40 10k 100k 1M 10M 10k 100M 100k FIGURE 3-14: (Simulation). 1M 10M Frequency (Hz) Frequency (Hz) FIGURE 3-17: Output Balance vs. Frequency, VOUTDM = 100mVpp, Gain = 1V/V (Simulation). CMRR vs. Frequency 10000 140 100 80 60 40 10k 100k 1M 10M 100M Frequency (Hz) FIGURE 3-15: (Simulation). +PSRR, -PSRR vs. Frequency,  2019 Microchip Technology Inc. VDD = 2.5V Output Impedance (Ohm) PSRR+ 2.5V PSRR+ 5.5V PSRR- 2.5V PSRR- 5.5V 120 PSRR (dB) -210 Phase 5.5V Frequency (Hz) 50 -180 Gain 2.5V 20 0 -18 100k 160 Open-loop Phase (deg) Note: Unless otherwise indicated, TA = 25°C, VDD - VSS = 2.5V to 5.5V, VOCM = open, VICM = mid-supply, PD\ = VDD, single-ended input, 50 input match, G = 1V/V, RF = RG = 1 kCL = 0 pF and RL = 1 k between the differential outputs. 1000 100 10 1 G = 1V/V G = 2V/V G = 5V/V 0.1 0.01 0.001 0.0001 100 1k 10k 10k 1M 10M 100M Frequency (Hz) FIGURE 3-18: Closed-Loop, Differential Output Impedance vs. Frequency (Simulation). DS20006162A-page 13 MCP6D11 Note: Unless otherwise indicated, TA = 25°C, VDD - VSS = 2.5V to 5.5V, VOCM = open, VICM = mid-supply, PD\ = VDD, single-ended input, 50 input match, G = 1V/V, RF = RG = 1 kCL = 0 pF and RL = 1 k between the differential outputs. 10000 Output Impedance (Ohm) VDD = 5.5V 1000 100 10 1 G = 1V/V G = 2V/V G = 5V/V 0.1 0.01 0.001 0.0001 100 1k 10k 100k 1M 10M 100M Frequency (Hz) FIGURE 3-19: Closed-Loop, Differential Output Impedance vs. Frequency (Simulation). DS20006162A-page 14  2019 Microchip Technology Inc. MCP6D11 Note: Unless otherwise indicated, TA = 25°C, VDD - VSS = 2.5V to 5.5V, VOCM = open, VICM = mid-supply, PD\ = VDD, single-ended input, 50 input match, G = 1V/V, RF = RG = 1 kCL = 0 pF and RL = 1 k between the differential outputs. 3.2 Input Noise and Distortion -80 100 Single-ended Input MSOP package -90 Distortion (dBc) Voltage Noise (nV/√Hz), Current Noise (pA/√Hz) VDD = 5.5V VDD = 2.5V 10 V-Noise 1 I-Noise -100 -110 -120 HD2, 3V HD2, 5V -130 -140 0.1 10 HD3, 3V HD3, 5V -150 100 1k 10k 100k 1M 1k 10k Frequency (Hz) FIGURE 3-23: VOUTDM = 2Vpp. FIGURE 3-20: Input Noise Voltage and Current Density vs. Frequency. 10000 HD2 and HD3 vs. Frequency, -60 Single-ended input QFN-16 package -70 VDD = 3.0V -80 Distortion (dBc) VOCM Voltage Noise (nV/—Hz) 100k Frequency (Hz) VOCM Pin floating VOCM Pin driven 1000 100 VDD = 5.5V VDD = 2.5V HD3, 100 kHz HD2, 100 kHz -90 -100 -110 -120 -130 HD3, 10 kHz HD2, 10 kHz -140 -150 10 10 100 1N 10N 100k 0 1M 1 2 3 4 5 6 7 8 9 10 Gain (V/V) )requency (Hz) FIGURE 3-21: VOCM Noise Voltage Density vs. Frequency,; a) Pin Floating, b) Pin Driven. FIGURE 3-24: VOUTDM = 2Vpp. HD2, HD3 vs Gain, -60 -80 Single-ended input QFN-16 package -90 Single-ended input QFN-16 package -70 VDD = 5.0V -110 -120 HD3, 5V HD3, 3V -130 HD2, 3V HD2, 5V Distortion (dBc) Distortion (dBc) -80 -100 HD3, 100 kHz HD2, 100 kHz -90 -100 -110 -120 -130 -140 HD3, 10 kHz HD2, 10 kHz -140 -150 1k 10k 100k )UHquency (Hz) FIGURE 3-22: VOUTDM = 2Vpp. HD2, HD3 vs Frequency,  2019 Microchip Technology Inc. -150 0 1 2 3 4 5 6 7 8 9 10 Gain (V/V) FIGURE 3-25: VOUTDM = 2Vpp. HD2, HD3 vs Gain, DS20006162A-page 15 MCP6D11 Note: Unless otherwise indicated, TA = 25°C, VDD - VSS = 2.5V to 5.5V, VOCM = open, VICM = mid-supply, PD\ = VDD, single-ended input, 50 input match, G = 1V/V, RF = RG = 1 kCL = 0 pF and RL = 1 k between the differential outputs. -60 -80 VDD = 3.0V Single-ended input QFN-16 package -90 Single-ended input QFN-16 package -70 VDD = 5.0V Distortion (dBc) Distortion (dBc) -80 -100 HD3, 100 kHz HD2, 100 kHz -110 -120 HD3, 10 kHz HD2, 10 kHz -130 -90 HD2, 100 kHz HD3, 100 kHz -100 -110 -120 HD2, 10 kHz HD3, 10 kHz -130 -140 -140 -150 200 400 600 800 -150 1000 0 Differential Load Resistance (:) FIGURE 3-26: VOUTDM = 2Vpp. HD2, HD3 vs RL, 4 FIGURE 3-29: G = 1V/V. 6 8 HD2, HD3 vs VOUTDM, -60 -80 VDD = 5.0V Single-ended input QFN-16 package -90 Single-ended input QFN-16 package -70 VDD = 3.0V -80 -100 Distortion (dBc) Distortion (dBc) 2 Differential Output Voltage (Vpp) HD3, 100 kHz HD2, 100 kHz -110 -120 HD3, 10 kHz HD2, 10 kHz -130 -140 -90 -100 HD3, 100kHz HD2, 100kHz -110 -120 -130 -140 HD2, 10kHz HD3, 10kHz -150 -150 200 400 600 800 1000 -160 Differential Load Resistance (: :) 0.5 0.7 0.9 1.1 1.3 1.5 1.7 1.9 2.1 External VOCM Voltage (V) FIGURE 3-27: VOUTDM = 2Vpp. FIGURE 3-30: HD2, HD3 vs VOCM, VOUTDM = 2Vpp, Gain = 1V/V. HD2, HD3 vs RL, -60 -60 Single-ended input QFN-16 package -70 VDD = 3.0V VDD = 5.0V -80 Distortion (dBc) -80 Distortion (dBc) Single-ended input QFN-16 package -70 -90 HD3, 100 kHz HD2, 100 kHz -100 -110 -120 -90 -100 -110 HD3, 100 kHz HD2, 100 kHz -120 -130 -140 -130 HD3, 10 kHz HD2, 10 kHz -140 HD2, 10 kHz HD3, 10 kHz -150 -160 -150 0 1 2 3 4 5 Differential Output Voltage (Vpp) FIGURE 3-28: G = 1V/V. DS20006162A-page 16 HD2, HD3 vs VOUTDM, 6 0.5 1 1.5 2 2.5 3 3.5 4 4.5 External VOCM Voltage (V) FIGURE 3-31: HD2, HD3 vs VOCM, VOUTDM = 2Vpp, Gain = 1V/V.  2019 Microchip Technology Inc. MCP6D11 Note: Unless otherwise indicated, TA = 25°C, VDD - VSS = 2.5V to 5.5V, VOCM = open, VICM = mid-supply, PD\ = VDD, single-ended input, 50 input match, G = 1V/V, RF = RG = 1 kCL = 0 pF and RL = 1 k between the differential outputs. FIGURE 3-32: G = 1V/V, VDD = 3V. 10 kHz FFT, VOUTDM = 2Vpp, FIGURE 3-34: G = 1V/V, VDD = 3V. 100 kHz FFT, VOUTDM = 2Vpp, FIGURE 3-33: G = 1V/V, VDD = 5V. 10 kHz FFT, VOUTDM = 2Vpp, FIGURE 3-35: G = 1V/V, VDD = 5V. 100 kHz FFT, VOUTDM = 2Vpp,  2019 Microchip Technology Inc. DS20006162A-page 17 MCP6D11 Note: Unless otherwise indicated, TA = 25°C, VDD - VSS = 2.5V to 5.5V, VOCM = open, VICM = mid-supply, PD\ = VDD, single-ended input, 50 input match, G = 1V/V, RF = RG = 1 kCL = 0 pF and RL = 1 k between the differential outputs. Time Response 2.5 Differential Output Voltage (V) 0.08 0.06 0.04 0.02 0.00 -0.02 VDD = 5.5V -0.04 -0.06 VDD = 2.5V -0.08 Differential Output Voltage (V) 3.3 1.0 0.5 0.0 -0.5 -1.0 -1.5 -2.0 FIGURE 3-39: G = 1V/V. Large Signal Step Response, 2.5 VDD = 3.0V 0.10 0.05 CL = 2 pF CL = 10 pF CL = 15 pF -0.05 -0.10 Differential Output Voltage (V) 0.15 -0.15 VDD = 5.0V 2.0 1.5 1.0 0.5 0.0 -0.5 -1.0 -1.5 -2.0 VOUT = 1Vpp VOUT = 2Vpp VOUT = 4Vpp -2.5 Time (50 ns/Div.) Time (100 ns/Div.) FIGURE 3-37: Small Signal Step Response vs. C-Load, VOUTDM = 200mVpp, G = 1V/V. FIGURE 3-40: G = 1V/V. 0.15 Large Signal Step Response, 6 VDD = 5.0V VDD = 3.0V 0.10 4 Differential,QSXW [ Differential Output 0.05 0.00 -0.05 CL = 2 pF CL = 10 pF CL = 15 pF -0.10 2 0 -2 -4 -0.15 Time (100 ns/Div.) FIGURE 3-38: Small Signal Step Response vs. C-Load, VOUTDM = 200mVpp, G = 1V/V. DS20006162A-page 18 Amplitude (V) Differential Output Voltage (V) VOUT = 1Vpp VOUT = 2Vpp VOUT = 4Vpp Time (50 ns/Div.) FIGURE 3-36: Small Signal Step Response, VOUTDM = 100mVpp, G = 1V/V. Differential Output Voltage (V) 1.5 -2.5 Time (20 ns/Div) 0.00 VDD = 3.0V 2.0 -6 Time (1 μs/Div) FIGURE 3-41: Time, G = 2V/V. Output Overdrive Recovery vs.  2019 Microchip Technology Inc. MCP6D11 Note: Unless otherwise indicated, TA = 25°C, VDD - VSS = 2.5V to 5.5V, VOCM = open, VICM = mid-supply, PD\ = VDD, single-ended input, 50 input match, G = 1V/V, RF = RG = 1 kCL = 0 pF and RL = 1 k between the differential outputs. 0.20 8 VDD = 5.0V Differential Input[ Differential Output 0.10 2 Error (%) Amplitude (V) 4 VDD = 5.5V G = 1V/V 0.15 6 0 -2 0.05 0.00 -0.05 VStep = 0.1Vpp VStep = 1.0Vpp VStep = 2.0Vpp -0.10 -4 -0.15 -6 -0.20 -8 0 30 FIGURE 3-42: Time, G = 2V/V. 60 90 120 150 180 210 240 270 300 'Time (ns), from 50% of Input Edge Time (1μs/Div) Output Overdrive Recovery vs. FIGURE 3-45: (Simulation). Settling Time vs VOUTDM 3.0 0.50 Amplitde (V) Output Common-mode Voltage 0.75 0.25 0.00 VDD = 3.0V VDD = 5.0V -0.25 2.0 VPD\ at 5VDD 1.0 VPD\ at 3VDD 0.0 -1.0 -2.0 -0.50 VOUT at 5VDD VOUT at 3VDD -3.0 -0.75 Time (0.5 μs/Div) Time (100 ns/Div) FIGURE 3-43: VOCM Small- (0.2V Step) and Large (1V step) Signal Step Response. FIGURE 3-46: PD\ Turn-On Transient Response, Input Signal: 1 MHz Sine, 2Vpp. 0.20 VDD = 2.5V G = 1V/V 0.15 3.0 2.0 0.05 Amplitde (V) Error (%) 0.10 0.00 -0.05 VStep = 0.1Vpp VStep = 1.0Vpp VStep = 2.0Vpp -0.10 1.0 -1.0 -2.0 -0.15 -0.20 VOUT at 5VDD 0.0 VOUT at 3VDD VPD\ at 3VDD VPD\ at 5VDD -3.0 0 30 60 90 120 150 180 210 240 270 'Time (ns), from 50% of Input Edge FIGURE 3-44: (Simulation). Settling Time vs. VOUTDM  2019 Microchip Technology Inc. 300 Time (0.5 μs/Div) FIGURE 3-47: PD\ Turn-Off Transient Response, Input Signal: 1 MHz Sine, 2Vpp. DS20006162A-page 19 MCP6D11 Note: Unless otherwise indicated, TA = 25°C, VDD - VSS = 2.5V to 5.5V, VOCM = open, VICM = mid-supply, PD\ = VDD, single-ended input, 50 input match, G = 1V/V, RF = RG = 1 kCL = 0 pF and RL = 1 k between the differential outputs. 3.4 DC Precision FIGURE 3-48: Input Offset Voltage Histogram (Factory Trimmed), VDD = 2.5V. FIGURE 3-51: Input Offset Voltage Drift Histogram, VDD = 5.5V. 150 Offset Voltage (μV) 100 50 0 -50 -100 -150 -40 -20 0 20 40 60 80 100 120 Temperature (qC) FIGURE 3-49: Input Offset Voltage Histogram (Factory Trimmed), VDD = 5.5V. FIGURE 3-52: Input Offset Voltage vs Temperature (45 Units), VDD = 2.5V and 5.5V. 150 Offset Voltage (μV) VDD = 2.5V +125 qC +85 qC +25 qC -40 qC 100 50 0 -50 Specified Min./Max.Range over Temperature -100 Representative Part -150 -0.4 -0.2 0 0.2 0.4 0.6 0.8 1 1.2 1.4 1.6 1.8 2 Input Common-mode Voltage (V) FIGURE 3-50: Input Offset Voltage Drift histogram; VDD = 2.5V. DS20006162A-page 20 FIGURE 3-53: Input Offset Voltage vs. Input Common-Mode Voltage.  2019 Microchip Technology Inc. MCP6D11 Note: Unless otherwise indicated, TA = 25°C, VDD - VSS = 2.5V to 5.5V, VOCM = open, VICM = mid-supply, PD\ = VDD, single-ended input, 50 input match, G = 1V/V, RF = RG = 1 kCL = 0 pF and RL = 1 k between the differential outputs. 150 Offset Voltage (μV) VDD = 5.5V +125qC +85qC +25qC -40qC 100 50 0 -50 Specified Min./Max.Range over Temperature -100 Representative Part -150 -0.4 0 0.4 0.8 1.2 1.6 2 2.4 2.8 3.2 3.6 4 4.4 4.8 Input Common-mode Voltage (V) FIGURE 3-54: Input Offset Voltage vs. Input Common-Mode Voltage. FIGURE 3-57: VDD = 2.5V. Input Offset Current Histogram, FIGURE 3-58: VDD = 5.5V. Input Offset Current Histogram, 3.0 VDD = 5.5V Input Bias Current (μA) 2.5 2.0 1.5 1.0 0.5 0.0 +125 qC +25 qC - 40 qC -0.5 -1.0 -1.5 -2.0 -0.5 0 0.5 1 1.5 2 2.5 3 3.5 4 4.5 5 Input Common-mode Voltage (V) FIGURE 3-55: Input Bias Current vs. Input Common-Mode Voltage. 10.0 250 6.0 Population (Units) 4.0 2.0 0.0 -2.0 +125 qC +25 qC - 40 qC -4.0 -6.0 -8.0 608 Samples QFN-16 package VDD = 5.5V 200 VDD = 2.5V 150 100 50 4.5 5  2019 Microchip Technology Inc. 0.3 0.1 -0.1 Input Offset Current Drift (nA/qqC) Input Common-mode Voltage (V) FIGURE 3-56: Input Offset Current vs. Input Common-mode Voltage. 0.26 4 0.22 3.5 0.18 3 0.14 2.5 0.06 2 0.02 1.5 -0.02 1 -0.06 0.5 -0.14 0 -0.18 -0.5 -0.22 -0.3 0 -10.0 -0.26 Input Offset Current (nA) 300 VDD = 5.5V 8.0 FIGURE 3-59: Histogram. Input Offset Current Drift DS20006162A-page 21 MCP6D11 Note: Unless otherwise indicated, TA = 25°C, VDD - VSS = 2.5V to 5.5V, VOCM = open, VICM = mid-supply, PD\ = VDD, single-ended input, 50 input match, G = 1V/V, RF = RG = 1 kCL = 0 pF and RL = 1 k between the differential outputs. 600 400 350 Population (Units) Population (Units) 500 400 300 200 100 0 -10 300 250 200 150 100 50 0 -5 0 5 10 -4 -3 -2 -1 0 1 2 3 Common Mode Offset Voltage (mV) Common Mode Offset Voltage (mV) FIGURE 3-60: VOCM Offset Voltage Histogram, VOCM Pin Floating, VDD = 2.5V. FIGURE 3-62: VOCM Offset Voltage Histogram, VOCM Pin Driven to Mid-Supply, VDD = 2.5V. 600 400 350 Population (Units) Population (Units) 500 400 300 200 100 0 -10 4 300 250 200 150 100 50 0 -5 0 5 10 -4 -3 -2 -1 0 1 2 3 Common Mode Offset Voltage (mV) Common Mode Offset Voltage (mV) FIGURE 3-61: VOCM Offset Voltage Histogram, VOCM Pin Floating, VDD = 5.5V. FIGURE 3-63: VOCM Offset Voltage Histogram, VOCM Pin Driven to Mid-Supply, VDD = 5.5V. DS20006162A-page 22 4  2019 Microchip Technology Inc. MCP6D11 Note: Unless otherwise indicated, TA = 25°C, VDD - VSS = 2.5V to 5.5V, VOCM = open, VICM = mid-supply, PD\ = VDD, single-ended input, 50 input match, G = 1V/V, RF = RG = 1 kCL = 0 pF and RL = 1 k between the differential outputs. Other DC Voltages and Currents 2.0 2.0 1.8 1.8 1.6 Qiescent Current (mA) Qiescent Current (mA) 3.5 +125 qC +25 qC -40 qC 1.4 1.2 1.0 0.8 0.6 0.4 1.6 1.4 VDD = 5.5V 1.2 1.0 0.8 0.6 VDD = 2.5V 0.4 0.2 0.2 0.0 0.0 0 0.5 1 1.5 2 2.5 3 3.5 4 4.5 5 0 5.5 0.5 1 Supply Voltage (V) FIGURE 3-66: Supply Current vs. Power Quiescent current (mA) 2.0 1.8 1.6 VDD = 5.5V 1.4 VDD = 2.5V 1.2 1.0 0.8 -40 -20 0 20 40 60 80 Supply Current vs.  2019 Microchip Technology Inc. 2.5 3 3.5 4 4.5 5 5.5 100 120 Supply Current vs. PD\ Voltage. 400 350 300 250 VOH 200 VDD = 5.5V 150 VOL 100 VDD = 2.5V 50 0 0 5 Temperature (qC) FIGURE 3-65: Temperature. 2 Voltage at PD\ Pin (V) Output Voltage Headroom (mV) FIGURE 3-64: Supply Voltage. 1.5 10 15 20 25 30 Output Current (+/-mA) FIGURE 3-67: Output Current. Output Voltage Headroom vs. DS20006162A-page 23 MCP6D11 NOTES: DS20006162A-page 24  2019 Microchip Technology Inc. MCP6D11 4.0 FUNCTIONAL DESCRIPTION 4.1 Overview Differential Input/Output amplifiers, also called Fully Differential Amplifiers (FDA), have become common driver amplifier for Precision ADCs (SAR, Delta-Sigma) as well as High-Speed ADCs. Compared to more discrete driver circuits built from standard op amps, integrated Differential I/O amplifiers have a number of advantages: • They allow the signal path to be DC coupled. Other, passive solutions may rely on RF-transformers that are noiseless, but effectively have a band-pass frequency response. Simple AC-coupling creates a high-pass response. • They provide superior common-mode rejection performance. • The input and output common-mode operating points are largely independent of the signal gain setting. • They suppress even-order harmonic distortion. • They allow the output common-mode voltage to be set independently of the input common-mode voltage, which provides for design flexibility when interfacing to ADCs requiring a certain Vcm for best dynamic performance. • They increase the dynamic range by a factor of 2 (6 dB) due to their differential, complementary output signals. • They allow the gain to be set in a wide range, including attenuation (G < 1V/V). An integrated, differential I/O amplifier is very similar in architecture to a standard, voltage-feedback operational amplifier, with a few differences. Both types of amplifiers have differential inputs. Differential I/O amplifiers have balanced differential outputs, while a standard operational amplifier's output is single-ended. In a differential I/O amplifier the output common-mode voltage can be controlled independently of the differential output voltage through an additional input, the VOCM pin. The purpose of the VOCM input is to set the output common-mode voltage for the two differential output pins. In a standard operational amplifier with single-ended output, the output common-mode voltage and the signal are the same. VDD * 3Ω MCP6D11 * FB+ 0.7 pF OUT+ 93 kΩ 26 kΩ IN- -  pF CIN_DM Diff I/O Amp. Commonmode Feedback Loop + - + IN+ VOCM 93 kΩ 26 kΩ OUT- * 3Ω 0.7 pF * FB- PD VSS FIGURE 4-1: MCP6D11 Block Diagram; (*) Internal Metal-Trace Impedances (3) and Pins FB+/Apply to QFN-16 Package Only.  2019 Microchip Technology Inc. DS20006162A-page 25 MCP6D11 The differential feedback, set with external resistors, controls only the differential output voltage. The common-mode feedback controls only the common-mode output voltage. This architecture makes it easy to arbitrarily set the output common-mode level in level shifting applications. It is forced, by internal common-mode feedback, to be equal to the voltage applied to the VOCM input pin, without affecting the differential output voltage. The result is nearly perfectly balanced differential outputs of identical amplitude and exactly 180° apart in phase over a wide frequency range. The circuit can be used with either a differential or a single-ended input, and the voltage gain is equal to the ratio of RF to RG. As a general rule, differential I/O amplifier circuits will benefit from matched feedback networks (RF/RG) to eliminate any common-mode input signal to differential output conversion. However, even if the external feedback networks are mismatched, the internal common-mode feedback loop will still force the outputs to remain balanced. The amplitudes of the signals at each output will remain equal and 180° out of phase. The input-to-output differential-mode gain will vary proportionately to the feedback mismatch, but the output balance will be unaffected. Ratio matching errors in the external resistors will result in a degradation of the circuit's ability to reject input common-mode signals, similar to a four-resistor difference amplifier made from a conventional op amp. The output common-mode voltage is generated at the mid point of the internal resistor string that is between VOUT+ and VOUT-. This voltage is fed into the common-mode feedback loop amplifier and compared to the reference voltage, VOCM. The VOCM input pin connects to an internal resistor divider (2 x 93 k). If the VOCM pin is left open, a voltage approximately halfway between VDD and VSS will develop due to this internal resistor network. An externally applied voltage at the VOCM pin can be used to overdrive the internal bias voltage if better accuracy or flexibility is desired. For a standard operational amplifier, there is typically one feedback path from the output to the negative input. The fully differential amplifier operates with two feedback paths as established with the feedback resistors RF, each from one of the outputs to its respective input. The input stage of the MCP6D11 uses bipolar devices for superior noise performance compared to CMOS devices with the trade-off that the input bias current is higher, typically less than 1 µA. The MCP6D11 has a differential input capacitance of about 1 pF (CIN_DM), which interacts with the feedback resistor, typically 1 k. To optimize the frequency response two internal 0.7 pF feedback capacitors were added to the design as a default compensation. The MCP6D11 uses a proportional to absolute temperature (PTAT) biasing circuit, which is designed such that the device's quiescent current (IQ) increases with an increase in temperature (see Figure 3-65). 4.2 Terminology and Definitions The basic representation of a differential I/O amplifier with its two feedback networks, output load and with its associated voltage nodes is shown in Figure 4-2. The differential signal source is represented showing the signal as VIN- and VIN+, which combine to form the differential input signal VIN_DM. An associated common-mode signal is given as VICM. Being an ideal source, the source impedance is zero and therefore not shown. VOCM RG1 RF1 RL/2 VINVSJ- VICM VOUT+ INOUT+ VIN_DM VOCM VOCM (VIN_CM) VSJ+ IN+ VOUT_DM VOUT_CM OUTVOUT- VIN+ RL/2 RG2 RF2 VOCM FIGURE 4-2: DS20006162A-page 26 Differential I/O Amplifier (Basic Representation).  2019 Microchip Technology Inc. MCP6D11 As described earlier, the voltage on the VOCM pin sets up the DC output common-mode voltage, and the AC signal will swing around this VOCM voltage, as illustrated in Figure 4-2. The internal common-mode feedback loop forces the VOUT+ and VOUT- outputs to be balanced, i.e. the signals at the two outputs are equal in amplitude but 180° out of phase. 4.2.1 DIFFERENTIAL I/O VOLTAGES The differential input voltage is the voltage applied between the VIN+ and VIN- inputs and the differential output voltage is the voltage seen across the OUT+ and OUT- pins. Equations for input and output differential voltages are listed below: EQUATION 4-1: V IN_DM 4.2.2 SIGNAL GAIN, NOISE GAIN While more complex, any circuit analysis of differential I/O amplifiers follows essentially the same rules as standard op-amp analysis. One of the important elements of an analysis is to identify the feedback factor (beta, ); in the case of the differential I/O amplifier there are two: 1 and 2. Equations 4-5 and 4-6 show the expressions for each of the feedback factors based on the circuit configuration of Figure 4-2. EQUATION 4-5: R G1  1 = ---------------------------R G1 +R F1 EQUATION 4-6: = V IN+ –V IN- RG2 R G2 + R F2  EQUATION 4-2: RF V OUT_DM =  V OUT+ – V OUT-   V IN_DM  -------R G Aside from describing the source-related input voltages, it is important to also consider the amplifier's summing junction input voltages, VSJ- and VSJ+. Note that these depend on both the input voltage and the output voltage, as shown in Equations 4-3 and 4-4: EQUATION 4-3: R F2 R G2 V SJ+ = V IN+  ----------------------------- + V OUT-  ----------------------------R +R R +R F2 G2 F2 G2 EQUATION 4-4: R F1 R G1 V SJ- = V IN-  ---------------------------- + V OUT+  ---------------------------R F1 + R G1 R F1 + R G1  2 = ---------------------------- Note that any source impedance present in an actual circuit will add a resistor term in series to the RG values. Here, RG represents the total DC impedance seen by the respective amplifier input (IN+, IN-) back to the source or a DC reference (e.g. ground). These feedback divider ratios will become useful for any output referred noise or error calculation and will be helpful in simplifying the algebra. Most circuit designs for differential I/O amplifiers, for example A/D converter drivers, will require matched feedback factors, 1 = 2, to maintain optimum AC and DC performance (see Section 5.1.3 “Mismatches and DC Errors”). Other beta related terms that can be useful for error calculations are AVG and , with AVG being defined as the average feedback factor and  defined as the difference in the feedback factors; see Equations 4-7 and 4-8: EQUATION 4-7:  1 + 2  AVG = -------------------2 EQUATION 4-8:  =  1 –  2  2019 Microchip Technology Inc. DS20006162A-page 27 MCP6D11 The differential-mode signal gain of the differential I/O amplifier is given in Equation 4-9, which includes the finite frequency dependent open-loop gain of the amplifier A(S) and the average feedback factor. EQUATION 4-9:   RF  V  1 OUT_DM Gain(G) = ---------------------------- = --------   ------------------------------------ 1 R V IN_DM  G 1 + --------------------------   A  (S) AVG Recall that A(S) x  is the loop gain of a single-ended amplifier; for the differential I/O amplifier the loop gain is based on AVG. Setting the open loop gain to infinite (A(S) → ∞) simplifies the equation, resulting in the expression for the ideal closed loop gain of the differential I/O amplifier. Equation 4-10 is used to select the feedback network resistors RF and RG and set the closed loop gain of the amplifier circuit for the differential input and output signal configuration. In the case of the single-ended input to differential output configuration, the equation becomes more complex as additional terms need to be considered; Section 5.1.2 “Interfacing to a Single-Ended Source”. EQUATION 4-10: RF V OUT_DM G = ---------------------------- = -------RG V IN_DM While the signal gain of a differential I/O amplifier is determined by G = RF/RG, the noise gain (GN) is given to: EQUATION 4-11: 2 G N = -------------------1 + 2 Where: RF 1  1 =  2  G N = --- = 1 + -------RG  This is important to remember as the noise gain needs to be considered when calculating total errors, like offsets or noise, that need to be referred to the output. Source impedance plays a factor and needs to be considered for maintaining matching between the two sides (1 = 2). DS20006162A-page 28 4.2.3 INPUT AND OUTPUT COMMON-MODE VOLTAGES The input common-mode voltage (VIN_CM) for the differential I/O amplifier is defined as the average voltage of the two input pins IN+ and IN-. EQUATION 4-12: V IN_CM V +V IN+ IN= -------------------------------2 The input common-mode voltage of the MCP6D11 typically ranges from VCM_L = VSS - 0.25V to VCM_H = VDD - 0.9V. Because of the external resistive divider formed by the feedback and gain resistors, the effective VIN_CM range is wider than the specified range. The input common-mode range of the amplifier depends on the Gain (G), the VOCM voltage, any externally applied common-mode voltage (VICM) and the circuit configuration. For fully differential input configurations, where VIN+ = -VIN-, the common-mode input voltage can be estimated using Equation 4-13: EQUATION 4-13: RG RF V IN_CM  V OCM  ---------------------- + V ICM  ---------------------R +R R +R F G F G For single-ended input configurations there will be an additional input signal component (either on VIN+ or VIN-, depending on how the source is connected) to the input common-mode voltage, as there is no out-of-phase signal applied to the other input. Applying the signal to VIN+ (connecting VIN- to ground), the common-mode input voltage can be approximated using Equation 4-14: EQUATION 4-14: V IN+ RF RG V IN_CM  V OCM  ---------------------- +  V ICM + -------------  ---------------------  2 R F + RG RF + R G Note: Here the input voltage VIN+ is equal to the peak input signal, VIN_P = VIN_PP/2. The result yields the upper value for the input common-mode voltage. To estimate the lower value use the negative term: -VIN+ in Equation 4-14.  2019 Microchip Technology Inc. MCP6D11 The output common-mode voltage (VOUT_CM) for the differential I/O amplifier is defined as the average of the two output voltages VOUT+ and VOUT-; see Figure 4-2. The output common-mode voltage VOUT_CM is primarily determined by the voltage at the VOCM pin, but they are not identical due to an offset component, the common-mode offset voltage. EQUATION 4-15: V 4.2.4 OUT_CM V +V  OUT+ OUT= -------------------------------------------------  V OCM 2 COMMON-MODE OFFSET VOLTAGE The common-mode offset voltage (VOS_CM) is defined as the difference between the output common-mode voltage and the VOCM voltage. EQUATION 4-16: V OS_CM =  V OUT_CM – VOCM  4.2.5 OUTPUT HEADROOM Once the VOCM voltage has been defined for a given amplifier configuration, verify that the desired maximum differential output swing (VOUT_PP) falls within the linear output voltage range of the differential I/O amplifier. As listed in the DC Electrical Characteristics table, the MCP6D11 requires a minimum output headroom (VOH, VOL) of 150 mV. EQUATION 4-17: V OUTpp V OUTmax = V OCM + --------------------4 EQUATION 4-18: V OUTpp V OUTmin = V OCM – --------------------4  2019 Microchip Technology Inc. 4.2.6 OUTPUT BALANCE An ideal differential output signal implies the two outputs of the differential amplifier should be exactly equal in amplitude and shifted 180° in phase. Hence, any imbalance in amplitude or phase between the two output signals results in an undesirable common-mode signal on the output. The Output Balance error is the measure of how well the outputs are balanced and is defined as the ratio of the output common-mode voltage (VOUT_CM) to the output differential signal (VOUT_DM). It is generally expressed as dB in logarithmic scale: EQUATION 4-19: VOUT_CM Output Balance error = 20log ---------------------V OUT_DM The function of the internal common-mode feedback loop circuit drives the output common-mode voltage (VOUT_CM) to equal the voltage level present at the VOCM pin. This ensures a very good output balance over a wide bandwidth (see Figure 3-17). Note that this figure is derived from simulation results to show the best-case output balance of the MCP6D11 itself. For this, the resistor tolerance was set to zero. However, at lower frequencies, the dominant contribution to the output balance error comes from the resistor tolerance of the external feedback network (1 ≠ 2) as the imbalance creates a common-mode to differential conversion (see Section 5.1.3 “Mismatches and DC Errors”). At higher frequencies capacitive and parasitic effects come into play. 4.2.7 STABILITY CONSIDERATIONS One of the primary applications for differential I/O amplifiers like the MCP6D11 is as a high-bandwidth driver amplifier for Analog-to-Digital converters, as described in Section 5.3.1 “Driving High Precision ADCs”. Here, the amplifier’s stability is of particular concern as the output of the amplifier not only has to drive a fairly large capacitive load, isolated by only small value resistors, but also has to respond quickly to transient currents resulting from the ADC's sampling effects. In order to analyze the amplifier’s stability in a closed-loop configuration, the open-loop gain (AOL) and phase frequency response need to be examined. Figure 4-3 shows the simulated differential input to output open-loop gain and phase of the MCP6D11 under two loading conditions: 1k load and no-load. The no-load condition removes any effect of the open-loop output impedance interacting with any external load element. DS20006162A-page 29 MCP6D11 -60 Open-loop Gain (dB) -90 Phase 80 -120 AOL 60 -150 40 -180 No Load 20 -210 Noise Gain = 6 dB 0 -240 Open-loop Phase (deg.) VDD = 5.0V 100 1 k: Load -20 -270 10k 100k 1M 10M 100M 1G Frequency (Hz) FIGURE 4-3: Simulated Open-Loop Gain and Phase Response with No-Load and 1 k Load Condition. For the simulation to show only the amplifier's forward path signal response, the two internal 0.7 pF feedback capacitors were removed. When operating the actual device, those 0.7 pF integrated capacitors (see Figure 4-3) are part of the feedback network that sets the noise gain and phase in the specific application. This also includes any external feedback capacitors and parasitic elements which can quickly lead to stability problems, effectively a result of insufficient phase margin. In general, the phase margin can be found at the frequency where the noise gain (GN) and the open-loop gain magnitude intersect; the loop-gain equals unity (0 dB) at this point. The difference between the phase at that point and -180 degrees is defined as the phase margin. Using Figure 4-3 the extracted phase margin is about 63 degrees for the 1 k load condition. Since the MCP6D11 operates as an inverting amplifier the noise gain remains at greater or equal to 1V/V (GN = 1 + RF/RG). This allows the user to set the signal gain to a fractional gain (G = RF/RG < 1V/V), with the amplifier's phase margin at approximately ≥ 30 degrees. The effect of this reduced phase margin for a G = 0.1V/V configuration can be seen from in increased gain peaking shown in Figure 3-1 and Figure 3-2. Operating the MCP6D11 with a high loop gain will result in the lowest distortion performance. Therefore, most ADC driver applications operate the amplifier at a low signal gain of 1V/V. The loop gain will decrease as AOL decreases with higher frequencies. Equation 4-20 describes the loop gain for the differential I/O amplifier having two feedback factors (as discussed in Section 4.2.2 “Signal Gain, Noise Gain”). EQUATION 4-20: A OL Loop Gain = ----------- = A OL   AVG , GN with  1 + 2  AVG = -------------------- DS20006162A-page 30 2 Figure 4-4 shows the simulated differential open-loop output impedance of the MCP6D11. Starting at the lower frequencies, the output impedance of the rail-to-rail output stage is high, then declines with a rate of -20 dB/decade until flattening out in a mid-range frequency section. The high impedance section will be significantly reduced when the amplifier is operated in a closed-loop configuration, as shown in Figure 3-18 and Figure 3-19. At higher frequencies, the open-loop output impedance starts to increase again at a rate of +20 dB/decade, resembling a first-order inductive behavior, indicating that purely capacitive loads can lead to stability issues (see Section 5.2.4 “Capacitive Loads”). 2SHQ/RRS2XWSXW,PSHGDQFHŸ 120 10000 VDD = 5.0V 1000 100 10 1 10 100 1k 10k 100k 1M 10M 100M 1G Frequency (Hz) FIGURE 4-4: Simulated Open-Loop Differential Output Impedance. Figure 4-3 shows the effect of the output impedance interacting with the load by comparing the frequency response of the no-load condition to the 1 k load condition. The load causes the AOL curve to have its 0 dB cross-over point at lower frequencies as well as increasing the phase shift. 4.3 Operation Differential I/O Amplifiers (or Fully Differential Amplifiers) resemble standard operational amplifiers configured in an inverting gain configuration. It should be noted that the polarity signs on the inputs (VIN-, VIN+) and outputs (VOUT-, VOUT+) typically shown on differential I/O amplifiers only indicate their phase relationship, and therefore may be misleading for correctly identifying input impedances. For this, is it useful to consider that both signal inputs on the differential I/O amplifier are in fact summing junctions, indicated by the labels VSJ- and VSJ+ shown in Figure 4-5. Because of the closed-loop feedback around the amplifier, these summing junctions represent a virtual ground and the resistors RG1 and RG2 set the input impedance seen by the source. If the input configuration is for a differential signal, the input impedance analysis is as simple as it is for an inverting op amp circuit, but more difficult in the case the input configuration is for a single-ended signal. Both situations will be discussed later in Section 5.1  2019 Microchip Technology Inc. MCP6D11 “Amplifier Configuration Options”. Figure 4-5 shows the typical representation of a differential I/O amplifier with its feedback resistor network, and also showing an equivalent circuit implementation based on standard op amps that illustrates more clearly the inverting amplifier configuration for the differential signal path. RG1 VSJ- + VOCM VIN+ - VIN- VOUT+ - VSJ- VIN- RF1 RG1 RF1 - + VOUT- RG2 RF2 VOUT+ + VOUT- VOCM Equivalent VSJ+ + VIN+ RG2 VSJ+ RF2 FIGURE 4-5: Equivalent Basic Circuit Functions. Comparing the two equivalent differential circuits of Figure 4-5, the key difference is that in the case of the two inverting operational amplifiers, the common-mode voltage is controlled by the voltage applied to the non-inverting inputs. For the differential I/O amplifier the output common-mode voltage is controlled using an independent feedback loop circuit (see Figure 4-1). 4.3.1 VOCM INPUT As mentioned earlier (see Section 4.1 “Overview”), the internal feedback control-loop drives the output common-mode voltage (VOUT_CM) to be equal to the voltage present at the VOCM pin. When the pin is left open, VOCM defaults approximately to a mid-supply voltage level set by the internal resistor divider (see Figure 4-1). EQUATION 4-21:  V OUT+ + V OUT-  V OUT_CM = -------------------------------------------------  V OCM 2 The VOCM input can be connected to an external reference voltage and varied within the specified range to accommodate specific output common-mode levels and achieve tighter control and higher accuracy. Refer to Figure 3-30 and Figure 3-31 for the impact on the achievable distortion when setting the VOCM voltage away from mid-supply. In any case, an external decoupling capacitor is recommended to be added on the VOCM pin to reduce the otherwise high output noise for this high impedance node.  2019 Microchip Technology Inc. DS20006162A-page 31 MCP6D11 4.3.2 INPUT AND ESD PROTECTION The inputs have a primary and secondary ESD protection using diodes connected from each input terminal to the supply rails. The design of the MCP6D11 includes comprehensive circuitry to protect the device against ESD, overvoltage and reverse-voltage events, as shown in Figure 4-6. VDD VSS MCP6D11 ESD Clamp VDD VDD VDD 50Ω - INVDD VDD VSS OUT+ + VSS VSS IN+ VDD - + OUT- 50Ω VSS VSS VSS VDD VSS VDD VSS VOCM FIGURE 4-6: ESD and Input Protection Scheme for the MCP6D11. The input stage of the MCP6D11 is protected against differential input voltages which exceed approximately 1.4V by two pairs of series diodes connected back-to-back between the differential amplifier inputs. If the differential input voltage exceeds 1.4V, the input current should be limited to 10 mA or less to prevent damage. Moreover, all pins have clamping diodes to both power supplies. If any pin is driven to voltages which exceed either supply, the current should be limited to under 10 mA. Internal protection diodes remain present across the input pins in both the operating and Power-Down mode. Large input signals during power-down can turn on the input differential protection diodes, thus producing a load current in the supply even with the device in Power-Down mode. The MCP6D11 is protected with an edge-triggered ESD clamp between the supply pins, VDD and VSS. Care should be taken to ensure a power supply turn-on/off edge rate (dV/dt) that does not exceed the rates stated in Section “Absolute Maximum Ratings †” to avoid activating this clamp circuit. 4.3.3 POWER-DOWN FUNCTION (PD\) The design of the MCP6D11 includes a power-down function which will reduce the quiescent current (IQ) down to about 5 µA (typical). The PD\ pin is referenced to the negative supply (VSS). Therefore, when operating the MCP6D11 with a negative supply voltage ensure that the voltage applied to the PD\ pin can be pulled down to within 0.4V of the negative rail. Similarly, pulling the PD\ high to within 0.4V of the positive rail will DS20006162A-page 32 PD\ ensure normal operation. Applying voltages at intermediate levels to the PD\ pin may result in an increase in quiescent current. When this pin is pulled low ( VSS) the amplifier is disabled and placed in Power-Down mode. Tying this pin to a high potential ( VDD) will enable normal operation. There is no internal pull-up or pull-down resistor and the PD\ pin should not be left floating. Note that when disabling the amplifier the signal path is still present for the source signal through the external resistors, which results in relatively poor signal isolation from the input to output in Power-Down mode. The Power-Down circuit of the MCP6D11 offers very fast turn-on and turn-off times, with typically 1 µs for the turn-on time and only 40 ns for the turn-off time (see Figure 3-48 and Figure 3-49). 4.4 Test Circuits Since most test equipment is specified for a 50 impedance, it requires the characterization circuit for the device-under-test (DUT) to include proper input and output termination or impedance matching. In addition, most equipment also has single-ended signal inputs or outputs. The basic characterization circuit therefore has the MCP6D11 configured for a singled-ended input with matched, 50 input termination, as shown in Figure 4-7. The amplifier's differential outputs drive the load resistor, which is split in order to facilitate a differential to single-ended signal conversion using a miniature RF-transformer (or Balun) while also providing a  2019 Microchip Technology Inc. MCP6D11 50 output impedance. The 50 input impedance from the network analyzer reflects through the transformer to be in parallel with the 52.3 resistor. a slightly higher attenuation due to the transformer insertion loss. The standard output load used for most tests is 1 k which results in approximately 31.8 dB of loss. This signal loss is normalized out for the typical frequency response curves to show the gain response to the amplifier output pins. Note that the 1:1 RF-transformer acts as a bandpass filter with a usable range from about 100 kHz to 500 MHz for this application. The total load impedance seen by the differential I/O amplifier is RL = 2 x RO + (ROT || 50). Due to the voltage divider on the output formed by the load component values, the amplifier's output is attenuated, see column “Attenuation” in Table 4-1. When using a transformer as shown in Figure 4-7, the signal will see 50Ω Source Impedance (Network Analyzer Output) VDD RG1 1 kΩ - VIN- RS RT1 52.3Ω VS C1 0.1 μF RO1 487Ω PD\ VOCM + N1 VOUT+ ROT 52.3Ω MCP6D11 VOUT- - ADTL1-4-75+ N2 + RO2 487Ω VSS VIN+ RG2 RT2 1 kΩ 52.3Ω RS1 50Ω FIGURE 4-7: RF2 1 kΩ Basic Characterization Circuit Configuration for Frequency Domain Tests. Furthermore, the components on the non-signal input side match very closely the components on the signal input side. This has the advantage of closely matching the two divider networks on each side of the amplifier. Alternatively, the three resistors on the non-signal input side, RG2, RT2, RS2, can be replaced by a single resistor to ground using a standard E96 value of 1.02 k with some loss in gain balancing between the two sides. For any active channel tests the power-down pin PD\ is tied to the positive supply (VDD). Most characterization plots are based on a 1 k value for RF (RF1 = RF2). While this resistor value can be adapted for a specific application purpose, the 1 k value offers a good compromise with issues related to this resistor value, specifically: • Output loading: both feedback resistors contribute to the total load seen across the outputs; for TABLE 4-1: 50 Ω Load (Network Analyzer Input) RF1 1 kΩ example the total differential loads shown in Figure 4-7 is 1 k || 2 k = 667. The 1 k value also reduces the power dissipated in the feedback networks. • Noise contribution: appears as the (4kTRF) term and the current noise multiplied by the resistor value (see paragraph Noise Analysis). • Feedback pole at the summing junction inputs: this pole is created by the feedback resistor (RF) value and the 1.0 pF differential input capacitance, CIN_DM, (plus any PC-board parasitic) and adds a zero in the noise gain, resulting in a reduced phase margin in most cases. The two internal 0.7 pF feedback capacitors (see Figure 4-1) combine with the external feedback resistors to introduce a zero in the noise gain, reducing the effect of the feedback pole. COMPONENT VALUES FOR DIFFERENTIAL TO SINGLE-ENDED OUTPUT USING A 1:1 TRANSFORMER RL RO1, RO2 ROT Attenuation 100 25 open 6 dB 200 86.6 69.8 16.8 dB 499 237 56.2 25.5 dB 1 k 487 52.3 31.8 dB 2 k 976 51.1 37.9 dB  2019 Microchip Technology Inc. DS20006162A-page 33 MCP6D11 4.4.1 DUAL-SUPPLY VERSUS SINGLE SUPPLY CHARACTERIZATION Although most end-equipment applications use a single-supply implementation, the factory device characterization is typically done using a dual-balanced supply. For example, a 5V test uses a ±2.5V supply and a 3V test uses a ±1.5V supply with the VOCM input pin at ground. Using a dual supply keeps the input and output common-mode voltages near mid-supply with optimal headroom for the output swing and no DC bias currents for level shifting. It also avoids the need for additional DC blocking capacitors that could restrict the signal bandwidth. This setup is used for characterizations such as the frequency response, harmonic distortion, and noise plots. Some of the time domain plots are done with a single supply to obtain the correct movement of the input common-mode voltage. 4.4.2 SIMULATED CHARACTERIZATION CURVES Some of the characterization data can only be generated through simulation in order to reflect the actual performance of the device without being constrained by hardware and measurement errors. One example of such a case is the output balance plot of Figure 3-17, which shows the best-case output balance by using exact matching on the external resistors for the single-ended input to differential output configuration. As discussed earlier, in practice the output balance is being constrained by the resistor value mismatch primarily at low frequencies but will converge with the high frequency portion of Figure 3-17 due to parasitic effects. Other Performance Figures that are based on simulations are: • AOL gain and phase, see Figure 3-16. • Settling times, see Figure 3-44 and Figure 3-45. • Closed loop output impedance versus frequency, see Figure 3-18 and Figure 3-19. • CMRR vs frequency, see Figure 3-14. • PSRR vs frequency, Figure 3-15. DS20006162A-page 34  2019 Microchip Technology Inc. MCP6D11 5.0 APPLICATION INFORMATION 5.1.1 5.1 Amplifier Configuration Options A differential signal source VSDiff, with its associated output impedances RS1 and RS2, is directly connected (dc-coupled) to the differential I/O amplifier, as shown in Figure 5-1. Alternatively, two capacitors can be inserted in series with each RG resistor to achieve an ac-coupled configuration. Because the voltage between the two amplifier summing junction inputs is driven to a null by negative feedback, they are virtually connected, and the differential input resistance, ZIN_AMP, is simply RG1 + RG2. Keep in mind that there is a frequency dependency to ZIN_AMP which may require for some applications to place a termination resistor, RT, between the signal inputs, as shown in Figure 5-2. When calculating the amplifier gain the source impedance, RS (shown here split in half for symmetry) needs to be added to the RG value; assuming a balanced design the gain is given to: When applying Differential I/O amplifiers there are two common configurations: interfacing to a differential source and driving the output differentially, or interfacing to a single-ended source and converting the signal into a differential output. Depending on the nature and impedance of the source, an input impedance match needs to be implemented. Accomplishing the optimum matching will be different for each configuration. As mentioned earlier, operating the differential I/O amplifier in a lab environment often requires configuration for matched 50 single-ended inputs and outputs (see Section 4.4 “Test Circuits”). However, for many signal chain applications the issue of impedance matching may be neglected as the source has sufficiently low impedance in the bandwidth of interest. But it is generally important to understand the interaction between a source impedance and the gain-setting network of a differential I/O amplifier. INTERFACING TO A DIFFERENTIAL SOURCE EQUATION 5-1: RF G = ---------------------RG + R S ZSource ZIN_Amp RF1 RG1 VSDiff RS1 - VINVOCM RS2 + VOUT+ MCP6D11 VIN+ - + VOUT- RG2 RF2 FIGURE 5-1: Differential I/O Amplifier Driven by a Differential Source with DC-Coupling. When driving a signal over a longer distance, for example twisted-pair cables, termination with a resistor across the differential inputs may be required, as illustrated in Figure 5-2. The termination resistor, RT, will appear in parallel to the sum of the two RG resistors. The calculation of RT is shown in Equation 5-2: For example, with RG1 = RG2 = 500, and if the source impedance to match up to is RS = 100 (RS1 = RS2 = 50), then the required value for RT is 111. Hence, the amplifiers input impedance ZIN_AMP, as seen by the source will be (RT || (RG1 + RG2)) = 100. EQUATION 5-2: 1 R T = ----------------------------------------------1 1 ------ – ---------------------------------R S  R G1 + R G2   2019 Microchip Technology Inc. DS20006162A-page 35 MCP6D11 ZIN_Amp RF1 VSDiff VIN- RG1 - + RS1 RT VOCM MCP6D11 RS2 VIN+ VOUT+ - + VOUT- RG2 RF2 FIGURE 5-2: Differential I/O Amplifier in Differential Configuration with Input Termination. 5.1.2 The DC biasing levels are determined as described in Section 4.2 “Terminology and Definitions”. For a true differential input, the common-mode voltages on the summing junction inputs of the amplifier remain fixed and do not move with the input signal (unlike the single-ended input configurations where the input common-mode voltages do vary with the input signal). For the AC coupled configuration the VOCM voltage also is the input biasing voltage since there is no DC current path through the feedback and gain resistors. In either case, setting the VOCM voltage to a mid-supply value, which is the default value if this pin is not externally driven, assures symmetry and therefore maximizes the achievable output swing. ZSource INTERFACING TO A SINGLE-ENDED SOURCE If the source is single-ended and referenced to ground, the differential I/O amplifier can be used to convert a single-ended input signal into a differential output signal while also providing a DC level shift; Figure 5-3 shows the basic circuit for this configuration. Again, VS is the input source with the associated source impedance ZSource = RS. ZIN_Amp RF1 RS VS VIN- - + RG1 VOCM VOUT+ MCP6D11 VIN+ - + VOUT- RG2 RF2 FIGURE 5-3: DS20006162A-page 36 Differential I/O Amplifier Driven by a Single-Ended Source with DC-Coupling.  2019 Microchip Technology Inc. MCP6D11 As discussed earlier, any source resistance (ZSource) will change the gain of the amplifier. For a single-ended input this occurs in an unbalanced way such that it affects only one side, for example RG1 as shown in Figure 5-3. To maintain balance between the amplifier's two feedback paths the user must set RF1 = RF2 and (RG1 + RS) = RG2. The effect of resistor mismatching is discussed in Section 5.1.3 “Mismatches and DC Errors”. For the circuit shown in Figure 5-3 the differential output voltage is given by Equation 5-3: EQUATION 5-3: and R Use Equation 5-4 to determine the effective input impedance for the single-ended input configuration: EQUATION 5-4: 2V S  1 –  1  + 2V OCM   1 –  2  V OUT_DM = ---------------------------------------------------------------------------------- +   1 2 Where: resistor RG1 is reduced by the moving common-mode voltage component, therefore resulting in an increased input impedance. The amplifier's common-mode feedback loop is a critical component in developing a differential output from a single-ended input signal, as it needs to dynamically adjust the input common-mode voltage in order to maintain balanced outputs. Meanwhile, the differential loop of the amplifier is forcing the voltages at the summing junctions to remain equal. R G1 Z IN_AMP = ---------------------------------------------RF 1 – ------------------------------------2   RG1 + R F  +R S G1  1 = ------------------------------------------- , R G1 +R +R S F1 Figure 5-4 shows the single-ended input circuit of Figure 5-3 extended for matching input termination, which adds resistor RT1. The input impedance as seen by the source (ZIN) therefore becomes: R G2  2 = ---------------------------- . R G2 + R F2 In order to calculate the amplifier's input impedance, ZIN_AMP, for this single-ended configuration it is important to recognize that the impedance looking in at the VIN- point is actually higher than just the physical RG1 value, as it could be assumed based on a standard inverting op amp circuit. Compared to the differential input configuration, here the input common-mode voltage has an additional signal dependent term (see Equation 4-14) resulting in a portion of the input signal moving the summing junctions of the differential I/O amplifier in the same direction as the applied signal. This creates a signal related current flow in the non-signal side RG2 resistor and produces the inverted output signal. The current flow in the signal-side gain To retain balance between the feedback resistor networks it is necessary to adjust the gain resistor RG2 accordingly, here shown by adding RT2 which can be calculated as RT2 = RS || RT1. If the resistor ratios are matched, the ratio of single-ended input to differential output gain is given by Equation 5-5: EQUATION 5-5: VOUT_DM R F1 2RT1 Gain (G) = ---------------------- = -----------------------------------------  ---------------------VS R G1 +  R S  R T1  R T1 + R S ZIN_Amp ZIN VIN- RS RF1 - + RG1 VOCM RT1 VS ZIN = RT1 || ZIN_AMP. MCP6D11 VIN+ - + RT2 FIGURE 5-4: VOUT+ VOUT- RG2 RF2 Single-Ended Configuration with Input Termination and Resistor Ratio Matching.  2019 Microchip Technology Inc. DS20006162A-page 37 MCP6D11 For the single-ended configuration with input impedance matching to a 50 source, Table 5-1 provides the required resistors using 1% standard values. TABLE 5-1: RF, RG, RT VALUES (1%) FOR SINGLE-ENDED INPUT PER Figure 5-4 Ideal Gain (V/V) Act. Gain (V/V) RF1, RF2 () RG1 () RG1* () (Note 1) RT1 () RG2 () RT2 () (Note 2) ZIN_AMP () ZIN () 1 0.997 1000 1000 1025.5 52.3 1020 25.5 1333 50.3 1 1.010 1000 976 1001.2 51.1 1020 25.2 1307 49.2 2 1.988 1020 499 524.5 52.3 523 25.5 754 48.9 5 5.057 1000 187 214.1 59.0 215 27.1 336 50.2 10 10.009 1020 88.7 117.4 68.1 118 28.7 183 50.6 Note 1: 2: 5.1.3 Effective gain resistor value: RG1* = RG1 + (RS || RT1); RS = 50. RT2 = RT1 || RS. MISMATCHES AND DC ERRORS Compared to a standard op amp, the differential I/O amplifier has an additional output error term that arises from the effects of mismatching of the resistor values and feedback ratios, see Section 4.2.2 “Signal Gain, Noise Gain”. The user must select from standard resistor values (e.g. E96) and define the tolerance (e.g. 1% or 0.1%) suitable for the application, which will in almost all cases lead to some degree of imbalance, or difference in the feedback factors (). For example, when selecting 1% tolerance resistors the worst case gain mismatch will be +/-2%; one side of the feedback path is at +2%, the other at -2%. Any such mismatch will cause a common-mode to differential conversion creating additional differential error terms. Opting for resistor values with a 0.1% tolerance is a good compromise between DC precision and cost. The parameters that are affected and need to be considered for a DC error analysis are the VOCM voltage, any input common-mode voltage from the source (VICM), and the input bias current. The input bias current (IB) contributes two error terms, one based on the resistor tolerance (±T; for 1% T = 0.01) and the other based on the gain mismatch. Estimate the first error term by multiplying the input bias current by (±2 x T x RF_Nom), with RF_Nom being the nominal feedback resistor value, for example 1 k. To estimate the IB error due to the gain mismatch multiply IB by the average RF value times the (/AVG) is the conversion gain factor. Additional terms for a comprehensive DC error analysis should include the input offset voltage (VOS) and the input offset current (IOS). Before adding to the total differential output error the input offset voltage needs to be multiplied by the noise gain (GN), which in the case of the differential I/O amplifier is the average of the two noise gains resulting from the two mismatched RG/RF ratios. The error resulting from the input offset current is simply referred to the output by multiplying with the average feedback resistor value (RF_AVG). To estimate the error contribution from VICM and VOCM to the differential output voltage (VOUT_DM) and assuming the differential input voltage is zero, use Equation 5-6 where the term (/AVG) is the conversion gain factor to the output for the gain ratio mismatch. EQUATION 5-6:  V OUT_DM   V OCM – V ICM   -------------- AVG DS20006162A-page 38  2019 Microchip Technology Inc. MCP6D11 5.2 Noise and Distortion 5.2.1 referred differential noise (eno) of an amplifier stage the surrounding resistor network contributions need to be considered. Shown in Figure 5-5 is the general noise model of a differential I/O amplifier and its resistor feedback components. NOISE ANALYSIS The MCP6D11 features very low noise with the voltage noise density at 5.0 nV/√Hz and the current noise density at 0.6 pA/√Hz. When analyzing the total output enRG1 enRF1 RF1 RG1 inenVocm in+ - + VOCM + eno - eni enRG2 FIGURE 5-5: RG2 RF2 enRF2 Differential I/O Amplifier Noise Analysis Model. The analysis starts by identifying each voltage- and current-noise term and its corresponding multiplication factor (noise gain) in order to derive its contribution to the total output noise (eno). The individual output-referred noise terms are then squared to combine noise as powers and are subsequently combined as a root-sum-of-squares (RSS). For the Differential I/O amplifier the voltage- and current-noise terms from each feedback path result in a 2 x contribution to the total noise as shown in Equation 5-7 below. One additional term is the common-mode voltage noise (envocm), which normally reflects to the output as a common-mode term, unless the two feedback ratios are mismatched. Then a conversion from common-mode to differential will occur. For envocm the gain multiplier when referred to the output is: GN x (1 - 2), which will be zero when 1 = 2. EQUATION 5-7: e no = 2 2 2 2 2 2  2e ni  +  2i n  R eq1  +  2i n  R eq2  +  2e nvocm    1 –  2   +  2e nRG1  1 –  1   +  2e nRG2  1 –  2   2 2 ------------------------------------------------------------------------------------------------------------------------------------------------------------------------------------------------------------------------------------------------------------------------------------------- +  e nRF1  +  e nRF2  2  1 +  2  Where: R eq1 = R G1  R F2 R eq2 = R G2  R F2 If the feedback ratios, 1 and 2 (RG = RG1 = RG2, RF = RF1 = RF2) are equal, the equation simplifies considerably to an expression very similar to a common voltage-feedback op amp's noise equation, as shown in Equation 5-8. Also, the current noise terms are assumed to be equal (in+ = in-) and uncorrelated.  2019 Microchip Technology Inc. DS20006162A-page 39 MCP6D11 EQUATION 5-8: e no = R  2 R 2   2 2 F F e  1 + -------- + 2  i R  + 2  4kTR -------- + 2  4kTR  ni  n F GR  F R G  G Where: 4kT = 1.64-20J at 298K (25°C) The noise contribution from resistors RG1, RF1, RG2 and RF2 can be calculated based on the Johnson noise equation: enR = √4kTR, where k is Boltzmann's constant (1.38065 x 10-23J/K), T is the resistor's absolute temperature in Kelvin, and R is the resistor value in ohms (). The last term is the output noise resulting from both the RF and RG resistors, at again twice the value for the output noise power of each side added together. The input referred noise of the MCP6D11 can be equated to that of a 1.6 k resistor. The recommended value for the feedback resistor RF is 1k, which results in the total output referred noise to be dominated by the amplifier's voltage noise. While there is flexibility in selecting different values for RF (and similarly for RG), lowering the feedback resistor value in order to lower its noise contribution will increase the amplifier's total output load and eventually result in an increase in distortion. Scaling the resistor value up will have the opposite effect of potentially improving distortion at the expense of higher noise contribution. However, because the feedback resistor interacts with the amplifier's input capacitance large values can lead to a noticeable reduction in phase margin and cause stability issues. A typical approach is to start with the recommended feedback resistor value and set the desired gain by scaling the gain resistor (RG) accordingly; Table 5-2 shows some example resistor values and corresponding noise results. By using the noise gain (GN = 1+ RF/RG) the two resistor noise terms can be combined into a single term of (4kTRFGN) resulting in a much simplified equation for the amplifier's differential output noise: EQUATION 5-9: e no = 2 2  e ni G N  + 2  i n R F  + 2  4kTR F G N  The first term of Equation 5-9 is the differential input noise times the noise gain. The second term is the input current noise times the feedback resistor - twice, since there are two uncorrelated current noise terms. TABLE 5-2: EXAMPLE OUTPUT NOISE RESULTS FOR THE SINGLE-ENDED INPUT CONFIGURATION WITH 50 INPUT MATCHING PER Figure 5-4 Ideal Gain (V/V) Act. Gain (V/V) RF1, RF2 () RG1 () RT1 () RG2 () ZIN () Diff-Out Noise eno (nV/√Hz) Noise RTI (nV/√Hz) 1 0.997 1000 1000 52.3 1020 50.3 12.90 12.90 2 1.988 1020 499 52.3 523 48.9 18.02 9.06 5 5.057 1000 187 59.0 215 50.2 31.99 6.33 10 10.009 1020 88.7 68.1 118 50.6 52.60 5.26 5.2.2 FACTORS AFFECTING HARMONIC DISTORTION In general, an amplifier's output harmonic distortion mainly relates to the open loop linearity in the output stage corrected by the loop gain at the fundamental frequency. Reducing the total load impedance, including the effect of the feedback resistor as discussed previously, the output stage open loop linearity degrades, causing an increase in harmonic distortion. Secondly, harmonic distortion will degrade as a function of the amplifier's output swing due to fine scale open loop output stage nonlinearities. A nominal swing of 2Vpp is typically used for harmonic distortion testing where Figure 3-29 illustrates the effect of going up to an 8Vpp differential swing that is more common with SAR-type ADC converters. An increase in the DS20006162A-page 40 amplifiers' gain correspondingly reduces the available loop gain to correct errors resulting in an increase in harmonic distortion terms. The MCP6D11 has a nearly constant distortion level when the VOCM operating point is moved within the allowed range; see Figure 3-30 and Figure 3-31. Driving the VOCM voltage beyond this range or the output voltages close to the supply rails will rapidly degrade the distortion performance. The device characterization used primarily resistors with a 1% tolerance. The resulting imbalance of the feedback factors does not directly degrade the distortion performance of the amplifier, but rather DC related errors (see section Section 5.1.3 “Mismatches and DC Errors”).  2019 Microchip Technology Inc. MCP6D11 5.2.3 SIGNAL BANDWIDTH LIMITATIONS Even when the application may not require the use of series RISO resistors, good practice is to leave a footprint for the RISO components on the PC-board layout (a 0 value initially) for later adjustment in case the response appears unacceptable. 5.2.4 Isolation Resistance (Ω) Although the MCP6D11 has a unity gain bandwidth of 90 MHz, it is primarily intended as driver for lower sample rate, high-precision ADCs with baseband input signal bandwidths in the DC to 100 kHz range. The high open loop gain and bandwidth of the MCP6D11 provides ultra-low distortion and fast settling times. Maximum power bandwidth is limited by the slew rate capability, which is typically 25V/µs. Operation with input signals above 100 kHz with near full output swings will see an increase in distortion levels (see Figure 3-22). CAPACITIVE LOADS Directly connecting a capacitive load to the output pins of a closed loop amplifier such as the MCP6D11 can lead to an unstable response; see the step response plots into a capacitive load Figure 3-37 and Figure 3-38. As illustrated in Figure 4-4, the rail-to-rail output stage of the MCP6D11 exhibits an inductive characteristic in the open loop output impedance at higher frequencies. This inductive open loop output impedance will interact with any capacitance present at the amplifier's outputs causing an additional phase shift, i.e. phase margin reduction. Account for the total capacitive load by considering all contributions from sources including feedback capacitors, next-stage input capacitance and PC-board parasitics. Larger values of feedback capacitors (CF greater than 10 pF) can risk a low phase margin. Including a 10 to 15 series resistor with a feedback capacitor can be used to reduce this effect. 220 200 180 160 140 120 100 80 60 40 20 0 G = 1V/V G = 2V/V 1 10 100 Differential Load Capacitor (pF) FIGURE 5-6: Capacitor. 5.3 1,000 RISO vs. Differential Load Application Example 5.3.1 DRIVING HIGH PRECISION ADCS The MCP6D11 differential I/O amplifier was designed primarily as an integrated front-end driver amplifier solution for high resolution ADCs, such as the SAR ADC MCP33131D. The 16-bit MCP33131D is part of a family of low-power 16-/14-/12-Bit, 1Msps SAR ADCs that feature differential inputs which are preferably driven by an amplifier with differential outputs to preserve their full performance. In most cases, inserting small value resistors (RISO) in series with each of the amplifier's outputs will isolate the capacitive load and can help avoiding or eliminating stability problems. Refer to Figure 5-6 for suggested RISO values. In general, as the noise gain (GN) of the device increases the value of the RISO resistor can be reduced while still obtaining a dampened response. The circuit in Figure 5-7 shows the MCP6D11 amplifier in a dc-coupled differential-input to differential-output configuration driving the MCP33131D while operating on a single-supply. The circuit can easily be adapted for a single-ended input to differential output configuration. MCP1501 Voltage Reference VDC VREF CR 10 µF RF2 Differential Input Signal VREF VIN+ 0V 1k RG2 1 k VREF/2 + VREF VIN- RG1 1 k Gain = 1V/V FIGURE 5-7: - VOCM 0.1 µF 0V VDD + VSS 1.8V to 5.5V VREF 5.0V VREF/2 0V VREF AVDD DVIO AIN+ Riso 24 MCP6D11 - 1.8V C1 1.8 nF Riso 24 SDI MCP331x1D-XX AIN- C1 1.8 nF VREF VREF/2 0V GND CNVST SCLK SDO Host Device (PIC32MZ) RF1 1 k Circuit Example for the MCP6D11 Driving the 16-bit 1Msps SAR ADC MCP33131D.  2019 Microchip Technology Inc. DS20006162A-page 41 MCP6D11 In almost all cases, a single-pole RC low-pass filter should be placed between the driver amplifier and the ADC, as shown in Figure 5-7. The input stage of the ADC is a sample-and-hold and the internal capacitor appears as a capacitive load to the driver amplifier. The typical sampling capacitor of the MCP33131D is 62 pF (differential). As part of the sampling process, significant charge injection occurs, resulting in fast current pulses that the amplifier output needs to react to while settling from this transient load condition to the new signal value within the allowed acquisition time. Here, the RC components serve a number of purposes. The capacitor (C1) helps to dampen the charge injection effects by providing a charge reservoir for an instantaneous current transfer with the ADC's sampling capacitor. Therefore, the value of this charge capacitor (C1) needs to be several times larger than the ADC sampling capacitor, however too high values will load the amplifier and result in increased distortion. The capacitor should be a NP0- or C0G -type due to their superior electrical and temperature stability. The two series resistors (RISO) primarily serve to isolate the capacitive loading due to capacitor C1 plus the sampling capacitor from the amplifier outputs and improved stability. Their value, along with the value of C1, should be chosen to achieve the desired settling behavior. For example, settling to 15-bit accuracy will require approximately 10 RC time constants. The number of time constants may vary between ADC models, depending on how the internal Sample-and-Hold circuit operates. It is generally best to keep the resistor value as low as possible while maintaining stability. High resistor values can be detrimental and lead to increased distortion. The RC network also performs a low-pass function and sets a bandwidth limit for the noise. However, as a single-pole filter, it is of limited use as an anti-aliasing filter. To implement a higher-order anti-aliasing filter, the MCP6D11 differential I/O amplifier can be configured to perform this function, such as an MFB-type filter. Shown in Figure 5-8 is the FFT of the MCP6D11 driving the MCP33131 differentially using a 4V ADC reference voltage resulting in an 8Vpp full-scale range (FSR). The 9.674 kHz input signal is set to -1 dBFS and the second and third harmonic is down at -113.3 dBc and -111.3 dBc respectively, with THD at -104.9 dBc. FIGURE 5-8: FFT Result of the MCP6D11 driving the MCP33131D, FSR = 8Vpp, fin = 9.7 kHz at -1dBFS. Figure 5-9 shows the FFT for the same configuration as in Figure 5-8, now with the input signal at 100 kHz. The highest harmonic is HD3 at -92.8 dBc while the second harmonic is at -101.6 dBc. THD is measured at -91.3 dBc. The -3 dB frequency of the RC filter is calculated using Equation 5-10: EQUATION 5-10: 1 f -3dB = ----------------------------------------------------------2  R ISO  C1 + C Sampling  Note: The ADC input capacitance should be factored into the frequency response of the input filter. DS20006162A-page 42 FIGURE 5-9: FFT Result of the MCP6D11 Driving the MCP33131D, FSR = 8Vpp, fin = 100 kHz at -1 dBFS.  2019 Microchip Technology Inc. MCP6D11 5.4 5.4.1 Application Tips SUPPLY BYPASSING CAPACITORS When operating the MCP6D11 in a single-supply configuration (VSS = Ground), only the VDD pin will require supply bypass capacitors. Using split supplies ensure that both amplifier supply pins have similar bypass capacitors tied to a low-noise analog ground. The QFN-16 package has four pins for the VDD and VSS supply connections, which are usually tied together on the PCB and the bypass capacitor can be shared for each set of four supply pins. The primary high-frequency bypass capacitors should be placed as close to the amplifier's supply pins as possible with direct connection to a low impedance analog ground. Use a 1 nF and a 0.1 µF leadless surface mount (e.g. size 0603), ceramic capacitor in parallel for each supply. For best high-frequency decoupling, consider X2Y-type capacitors that offer a much higher self-resonance frequency over standard capacitors. Most applications benefit from adding a bulk capacitor (e.g. 2.2 µF to 10 µF, tantalum or ceramic) within approximately 20 mm (0.8 inch) of the supply pins, which can be shared among multiple MCP6D11 devices. 5.4.2 VOCM BYPASSING • When using the QFN-16 package, note that the FB+ and FB- pins are duplicates of the output pins (OUT+, OUT-) but are placed conveniently close to the corresponding inputs such that the external feedback resistor (RF) can be placed with minimum trace length. This minimizes potential parasitic capacitances affecting sensitive device pins. Note that the internal trace adds about 3 to the external feedback resistor value. • When routing differential/complementary signals, ensure a highly symmetric layout placement with identical trace length. Even small amounts of asymmetry can lead to distortion and balance errors. Routing such signal traces over a longer distance will in most cases require microstrip layout and impedance matching techniques. • Use surface mount small geometry 0603 or 0402 size resistors and capacitors to minimize parasitic capacitance effects. • The exposed thermal pad on the QFN-16 package should be soldered to a low-noise ground or power plane. While the pad is electrically isolated from the die it must be connected preferably to a ground plane (alternatively a power plane) and not be left floating. In addition to the supply decoupling, the VOCM pin should be bypassed with a 0.1 µF leadless surface mount, ceramic capacitor for either case, externally driven or left open. This will reduce the noise feedthrough to the amplifier's output from this high impedance input. 5.4.3 PCB LAYOUT While the MCP6D11 amplifier may be used to for relatively low signal frequencies (f < 500 kHz), it is critical to apply high-frequency PC-board techniques in order to preserve the low distortion and fast step response capabilities of the device. The input summing junctions and the differential outputs in particular of the MCP6D11 are very sensitive to parasitic capacitances even as low as 0.5 pF. Following are some specific recommendations: • Continuous ground planes usually work well to establish a low impedance analog ground potential. Also, multi-layer PCB designs often use power planes. When used, the user must make sure that both power and ground planes include keep-out areas under and around the device and around sensitive nodes on the feedback and gain setting components (e.g. RG, RF, CF). • The feedback resistors (RF) should be placed with minimum trace length between the amplifier's output and summing function input pins. • The gain resistors (RG) should connect to the summing junction pins with minimum trace length.  2019 Microchip Technology Inc. DS20006162A-page 43 MCP6D11 NOTES: DS20006162A-page 44  2019 Microchip Technology Inc. MCP6D11 6.0 DESIGN AIDS Microchip provides the basic design aids needed for the MCP6D11 family of op amps. 6.1 Microchip Advanced Part Selector (MAPS) MAPS is a software tool that helps efficiently identify Microchip devices that fit a particular design requirement. Available at no cost from the Microchip web site at www.microchip.com/maps, MAPS is an overall selection tool for Microchip’s product portfolio that includes Analog, Memory, MCUs and DSCs. Using this tool, a customer can define a filter to sort features for a parametric search of devices and export side-by-side technical comparison reports. Helpful links are also provided for data sheets, purchase and sampling of Microchip parts. 6.2 Application Notes The following Microchip Application Notes are available on the Microchip web site at www.microchip. com/appnotes and are recommended as supplemental reference resources. ADN003: “Select the Right Operational Amplifier for your Filtering Circuits”, DS21821 AN722: “Operational Amplifier Topologies and DC Specifications”, DS00722 AN723: “Operational Amplifier AC Specifications and Applications”, DS00723 AN884: “Driving Capacitive Loads With Op Amps”, DS00884 AN990: “Analog Sensor Conditioning Circuits – An Overview”, DS00990 AN1177: “Op Amp Precision Design: DC Errors”, DS01177 AN1228: “Op Amp Precision Design: Random Noise”, DS01228 AN1258: “Op Amp Precision Design: PCB Layout Techniques”, DS01258  2019 Microchip Technology Inc. DS20006162A-page 45 MCP6D11 NOTES: DS20006162A-page 46  2019 Microchip Technology Inc. MCP6D11 7.0 PACKAGING INFORMATION 7.1 Package Marking Information 8-Lead MSOP Example 6D11 908256 16-Lead, 3x3 mm QFN Example Part Number Legend: XX...X Y YY WW NNN e3 * Note: ACC 1908 256 Code MCP6D11-E/MG ACC MCP6D11T-E/MG ACC Customer-specific information Year code (last digit of calendar year) Year code (last 2 digits of calendar year) Week code (week of January 1 is week ‘01’) Alphanumeric traceability code Pb-free JEDEC® designator for Matte Tin (Sn) This package is Pb-free. The Pb-free JEDEC designator ( e3 ) can be found on the outer packaging for this package. In the event the full Microchip part number cannot be marked on one line, it will be carried over to the next line, thus limiting the number of available characters for customer-specific information.  2019 Microchip Technology Inc. DS20006162A-page 47 MCP6D11 Note: For the most current package drawings, please see the Microchip Packaging Specification located at http://www.microchip.com/packaging DS20006162A-page 48  2019 Microchip Technology Inc. MCP6D11 Note: For the most current package drawings, please see the Microchip Packaging Specification located at http://www.microchip.com/packaging  2019 Microchip Technology Inc. DS20006162A-page 49 MCP6D11 Note: For the most current package drawings, please see the Microchip Packaging Specification located at http://www.microchip.com/packaging DS20006162A-page 50  2019 Microchip Technology Inc. MCP6D11 Note: For the most current package drawings, please see the Microchip Packaging Specification located at http://www.microchip.com/packaging  2019 Microchip Technology Inc. DS20006162A-page 51 MCP6D11 Note: For the most current package drawings, please see the Microchip Packaging Specification located at http://www.microchip.com/packaging DS20006162A-page 52  2019 Microchip Technology Inc. MCP6D11 Note: For the most current package drawings, please see the Microchip Packaging Specification located at http://www.microchip.com/packaging  2019 Microchip Technology Inc. DS20006162A-page 53 MCP6D11 NOTES: DS20006162A-page 54  2019 Microchip Technology Inc. MCP6D11 APPENDIX A: REVISION HISTORY Revision A (February 2019) • Initial release of this Data Sheet.  2019 Microchip Technology Inc. DS20006162A-page 55 MCP6D11 NOTES:  2019 Microchip Technology Inc. DS20006162A-page 56 MCP6D11 PRODUCT IDENTIFICATION SYSTEM To order or obtain information, e.g., on pricing or delivery, refer to the factory or the listed sales office. [X](1) -X /XX Tape and Reel Option Temperature Range Package PART NO. Device Device: MCP6D11 - Low Noise, Precision, 90 MHz Differential I/O Amplifier Tape and Reel Option: Blank T = Standard packaging (tube or tray) = Tape and Reel(1) Temperature Range: E = -40C to +125C Package: MS MG = = (Extended) 8-Lead Plastic Micro Small Outline Package (MSOP) 16-Lead Quad Flat No Lead Package (QFN)  2019 Microchip Technology Inc. Examples: a) MCP6D11-E/MS Extended temperature, MSOP package b) MCP6D11-E/MG Extended temperature, QFN package c) MCP6D11T-E/MS Extended temperature, MSOP package, Tape and Reel d) MCP6D11T-E/MG Extended temperature, QFN package, Tape and Reel Note 1: Tape and Reel identifier only appears in the catalog part number description. This identifier is used for ordering purposes and is not printed on the device package. Check with your Microchip Sales Office for package availability with the Tape and Reel option. DS20006162A-page 57 MCP6D11 NOTES: DS20006162A-page 58  2019 Microchip Technology Inc. Note the following details of the code protection feature on Microchip devices: • Microchip products meet the specification contained in their particular Microchip Data Sheet. • Microchip believes that its family of products is one of the most secure families of its kind on the market today, when used in the intended manner and under normal conditions. • There are dishonest and possibly illegal methods used to breach the code protection feature. All of these methods, to our knowledge, require using the Microchip products in a manner outside the operating specifications contained in Microchip’s Data Sheets. Most likely, the person doing so is engaged in theft of intellectual property. • Microchip is willing to work with the customer who is concerned about the integrity of their code. • Neither Microchip nor any other semiconductor manufacturer can guarantee the security of their code. Code protection does not mean that we are guaranteeing the product as “unbreakable.” Code protection is constantly evolving. We at Microchip are committed to continuously improving the code protection features of our products. Attempts to break Microchip’s code protection feature may be a violation of the Digital Millennium Copyright Act. If such acts allow unauthorized access to your software or other copyrighted work, you may have a right to sue for relief under that Act. Information contained in this publication regarding device applications and the like is provided only for your convenience and may be superseded by updates. It is your responsibility to ensure that your application meets with your specifications. MICROCHIP MAKES NO REPRESENTATIONS OR WARRANTIES OF ANY KIND WHETHER EXPRESS OR IMPLIED, WRITTEN OR ORAL, STATUTORY OR OTHERWISE, RELATED TO THE INFORMATION, INCLUDING BUT NOT LIMITED TO ITS CONDITION, QUALITY, PERFORMANCE, MERCHANTABILITY OR FITNESS FOR PURPOSE. Microchip disclaims all liability arising from this information and its use. Use of Microchip devices in life support and/or safety applications is entirely at the buyer’s risk, and the buyer agrees to defend, indemnify and hold harmless Microchip from any and all damages, claims, suits, or expenses resulting from such use. No licenses are conveyed, implicitly or otherwise, under any Microchip intellectual property rights unless otherwise stated. Trademarks Microchip received ISO/TS-16949:2009 certification for its worldwide headquarters, design and wafer fabrication facilities in Chandler and Tempe, Arizona; Gresham, Oregon and design centers in California and India. The Company’s quality system processes and procedures are for its PIC® MCUs and dsPIC® DSCs, KEELOQ® code hopping devices, Serial EEPROMs, microperipherals, nonvolatile memory and analog products. In addition, Microchip’s quality system for the design and manufacture of development systems is ISO 9001:2000 certified. SQTP is a service mark of Microchip Technology Incorporated in the U.S.A. QUALITY MANAGEMENT SYSTEM CERTIFIED BY DNV The Microchip name and logo, the Microchip logo, AnyRate, AVR, AVR logo, AVR Freaks, BitCloud, chipKIT, chipKIT logo, CryptoMemory, CryptoRF, dsPIC, FlashFlex, flexPWR, Heldo, JukeBlox, KeeLoq, Kleer, LANCheck, LINK MD, maXStylus, maXTouch, MediaLB, megaAVR, MOST, MOST logo, MPLAB, OptoLyzer, PIC, picoPower, PICSTART, PIC32 logo, Prochip Designer, QTouch, SAM-BA, SpyNIC, SST, SST Logo, SuperFlash, tinyAVR, UNI/O, and XMEGA are registered trademarks of Microchip Technology Incorporated in the U.S.A. and other countries. ClockWorks, The Embedded Control Solutions Company, EtherSynch, Hyper Speed Control, HyperLight Load, IntelliMOS, mTouch, Precision Edge, and Quiet-Wire are registered trademarks of Microchip Technology Incorporated in the U.S.A. Adjacent Key Suppression, AKS, Analog-for-the-Digital Age, Any Capacitor, AnyIn, AnyOut, BodyCom, CodeGuard, CryptoAuthentication, CryptoAutomotive, CryptoCompanion, CryptoController, dsPICDEM, dsPICDEM.net, Dynamic Average Matching, DAM, ECAN, EtherGREEN, In-Circuit Serial Programming, ICSP, INICnet, Inter-Chip Connectivity, JitterBlocker, KleerNet, KleerNet logo, memBrain, Mindi, MiWi, motorBench, MPASM, MPF, MPLAB Certified logo, MPLIB, MPLINK, MultiTRAK, NetDetach, Omniscient Code Generation, PICDEM, PICDEM.net, PICkit, PICtail, PowerSmart, PureSilicon, QMatrix, REAL ICE, Ripple Blocker, SAM-ICE, Serial Quad I/O, SMART-I.S., SQI, SuperSwitcher, SuperSwitcher II, Total Endurance, TSHARC, USBCheck, VariSense, ViewSpan, WiperLock, Wireless DNA, and ZENA are trademarks of Microchip Technology Incorporated in the U.S.A. and other countries. Silicon Storage Technology is a registered trademark of Microchip Technology Inc. in other countries. GestIC is a registered trademark of Microchip Technology Germany II GmbH & Co. KG, a subsidiary of Microchip Technology Inc., in other countries. All other trademarks mentioned herein are property of their respective companies. © 2019, Microchip Technology Incorporated, All Rights Reserved. ISBN: 978-1-5224-4195-3 == ISO/TS 16949 ==  2019 Microchip Technology Inc. DS20006162A-page 59 Worldwide Sales and Service AMERICAS ASIA/PACIFIC ASIA/PACIFIC EUROPE Corporate Office 2355 West Chandler Blvd. Chandler, AZ 85224-6199 Tel: 480-792-7200 Fax: 480-792-7277 Technical Support: http://www.microchip.com/ support Web Address: www.microchip.com Australia - Sydney Tel: 61-2-9868-6733 India - Bangalore Tel: 91-80-3090-4444 China - Beijing Tel: 86-10-8569-7000 India - New Delhi Tel: 91-11-4160-8631 Austria - Wels Tel: 43-7242-2244-39 Fax: 43-7242-2244-393 China - Chengdu Tel: 86-28-8665-5511 India - Pune Tel: 91-20-4121-0141 Denmark - Copenhagen Tel: 45-4450-2828 Fax: 45-4485-2829 China - Chongqing Tel: 86-23-8980-9588 Japan - Osaka Tel: 81-6-6152-7160 Finland - Espoo Tel: 358-9-4520-820 China - Dongguan Tel: 86-769-8702-9880 Japan - Tokyo Tel: 81-3-6880- 3770 China - Guangzhou Tel: 86-20-8755-8029 Korea - Daegu Tel: 82-53-744-4301 France - Paris Tel: 33-1-69-53-63-20 Fax: 33-1-69-30-90-79 China - Hangzhou Tel: 86-571-8792-8115 Korea - Seoul Tel: 82-2-554-7200 China - Hong Kong SAR Tel: 852-2943-5100 Malaysia - Kuala Lumpur Tel: 60-3-7651-7906 China - Nanjing Tel: 86-25-8473-2460 Malaysia - Penang Tel: 60-4-227-8870 China - Qingdao Tel: 86-532-8502-7355 Philippines - Manila Tel: 63-2-634-9065 China - Shanghai Tel: 86-21-3326-8000 Singapore Tel: 65-6334-8870 China - Shenyang Tel: 86-24-2334-2829 Taiwan - Hsin Chu Tel: 886-3-577-8366 China - Shenzhen Tel: 86-755-8864-2200 Taiwan - Kaohsiung Tel: 886-7-213-7830 China - Suzhou Tel: 86-186-6233-1526 Taiwan - Taipei Tel: 886-2-2508-8600 China - Wuhan Tel: 86-27-5980-5300 Thailand - Bangkok Tel: 66-2-694-1351 China - Xian Tel: 86-29-8833-7252 Vietnam - Ho Chi Minh Tel: 84-28-5448-2100 Atlanta Duluth, GA Tel: 678-957-9614 Fax: 678-957-1455 Austin, TX Tel: 512-257-3370 Boston Westborough, MA Tel: 774-760-0087 Fax: 774-760-0088 Chicago Itasca, IL Tel: 630-285-0071 Fax: 630-285-0075 Dallas Addison, TX Tel: 972-818-7423 Fax: 972-818-2924 Detroit Novi, MI Tel: 248-848-4000 Houston, TX Tel: 281-894-5983 Indianapolis Noblesville, IN Tel: 317-773-8323 Fax: 317-773-5453 Tel: 317-536-2380 Los Angeles Mission Viejo, CA Tel: 949-462-9523 Fax: 949-462-9608 Tel: 951-273-7800 Raleigh, NC Tel: 919-844-7510 New York, NY Tel: 631-435-6000 San Jose, CA Tel: 408-735-9110 Tel: 408-436-4270 Canada - Toronto Tel: 905-695-1980 Fax: 905-695-2078 DS20006162A-page 60 China - Xiamen Tel: 86-592-2388138 China - Zhuhai Tel: 86-756-3210040 Germany - Garching Tel: 49-8931-9700 Germany - Haan Tel: 49-2129-3766400 Germany - Heilbronn Tel: 49-7131-67-3636 Germany - Karlsruhe Tel: 49-721-625370 Germany - Munich Tel: 49-89-627-144-0 Fax: 49-89-627-144-44 Germany - Rosenheim Tel: 49-8031-354-560 Israel - Ra’anana Tel: 972-9-744-7705 Italy - Milan Tel: 39-0331-742611 Fax: 39-0331-466781 Italy - Padova Tel: 39-049-7625286 Netherlands - Drunen Tel: 31-416-690399 Fax: 31-416-690340 Norway - Trondheim Tel: 47-7288-4388 Poland - Warsaw Tel: 48-22-3325737 Romania - Bucharest Tel: 40-21-407-87-50 Spain - Madrid Tel: 34-91-708-08-90 Fax: 34-91-708-08-91 Sweden - Gothenberg Tel: 46-31-704-60-40 Sweden - Stockholm Tel: 46-8-5090-4654 UK - Wokingham Tel: 44-118-921-5800 Fax: 44-118-921-5820  2019 Microchip Technology Inc. 08/15/18