MIC2125/6
28V Synchronous Buck Controllers
Featuring Adaptive ON-Time Control
Features
General Description
• Hyper Speed Control Architecture Enables:
- High delta V operation (VIN = 28V and VOUT
= 0.6V)
- Any Capacitor™ stable
• 4.5V to 28V Input Voltage
• Adjustable Output Voltage from 0.6V to 24V
• 200 kHz to 750 kHz Programmable Switching
Frequency
The MIC2125 and MIC2126 are constant-frequency
synchronous buck controllers featuring a unique
adaptive ON-time control architecture. The MIC2125/6
operate over an input voltage range from 4.5V to 28V
and can be used to supply load current up to 25A. The
output voltage is adjustable down to 0.6V with a
guaranteed accuracy of ±1%. The device operates with
programmable switching frequency from 200 kHz to
750 kHz.
Package Type
2015 Microchip Technology Inc.
16
15
14
13
12
AGND
11
NC
3
10
OVP
4
9
BST
VDD
1
PVDD
2
ILIM
EP 17
5
6
7
8
SW
DL
FB
MIC2125/6
16-Pin 3 mm x 3 mm QFN (ML)
PG
Networking/Telecom Equipment
Base Stations, Servers
Distributed Power Systems
Industrial Power Supplies
DH
•
•
•
•
EN
Applications
The MIC2125/6 offer a full suite of features to ensure
protection of the IC during fault conditions. These
include undervoltage lockout to ensure proper
operation under power-sag conditions, internal
soft-start to reduce inrush current, “hiccup” mode
short-circuit protection, and thermal shutdown.
FREQ
•
•
•
HyperLight Load® architecture provides the same high
efficiency and ultra-fast transient response as the
Hyper Speed Control® architecture under medium to
heavy loads. It also maintains high efficiency under
light load conditions by transitioning to variable
frequency, discontinuous conduction mode operation.
VIN
•
HyperLight Load® (MIC2125)
Hyper Speed Control® (MIC2126)
Enable Input and Power Good Output
Built-in 5V Regulator for Single-Supply Operation
Programmable current limit and “hiccup” mode
short-circuit protection
7 ms internal soft-start, internal compensation,
and thermal shutdown
Supports Safe Start-Up into a Prebiased Output
–40°C to +125°C Junction Temperature Range
Available in 16-pin, 3 mm × 3 mm QFN Package
PGND
•
•
•
•
•
DS20005459B-page 1
MIC2125/6
Typical Application Circuit
MIC2125/6
3x3 QFN
VIN
4.5V TO 28V
FREQ
PVDD
2.2μF
×3
VIN
VDD
220μF
BST
MIC2125/6
4.7μF
AGND
0.1μF
DH
VOUT
3.3V/20A
0.72μH
EN
EN
SW
90.9kΩ
PG
PG
470pF
10kΩ
100μF 470μF
DL
VOUT
56.2kΩ
0.1μF
OVP
PGND
10kΩ
FB
2.26kΩ
ILIM
1.2kΩ
Functional Block Diagram
MIC2125/26
PVDD
VIN
VDD
4.5V TO 28V
VIN
LDO
R19
4.7μF
FIXED TON
UVLO
220μF
R20
BST
VIN
2.2μF
×2
FREQ
ESTIMATE
VDD
MODIFIED
TOFF
HSD
DH
Q1
0.1μF
0.72μH
100kΩ
SW
VOUT
3.3V/20A
CONTROL
EN
EN
PVDD
LOGIC
470pF
TIMER
SOFT–START
LSD
DL
90.9kΩ
Q3
1.2kΩ
R1
10kΩ
100μF
470μF
0.1μF
PGND
SOFT
START
CL
DETECTION
ILIM
R2
2.26kΩ
THERMAL
SHUTDOWN
OVP
COMPENSATION
VREF
0.6V
VDD
gm EA
FB
COMP
49.9kΩ
8%
PG
PG
VREF
0.6V
AGND
92%
DS20005459B-page 2
2015 Microchip Technology Inc.
MIC2125/6
1.0
ELECTRICAL CHARACTERISTICS
Absolute Maximum Ratings †
VIN .............................................................................................................................................................. –0.3V to +30V
VDD, PVDD .................................................................................................................................................... –0.3V to +6V
VSW, VFREQ, VILIM, VEN ....................................................................................................................–0.3V to (VIN +0.3V)
VBST to VSW ................................................................................................................................................... –0.3V to 6V
VBST ............................................................................................................................................................. –0.3V to 36V
VPG................................................................................................................................................. –0.3V to (VDD + 0.3V)
VFB ................................................................................................................................................. –0.3V to (VDD + 0.3V)
PGND to AGND ........................................................................................................................................... –0.3V to +0.3V
ESD Rating(1) ............................................................................................................................................................. 2 kV
Operating Ratings ‡
Supply Voltage (VIN) ...................................................................................................................................... 4.5V to 28V
VSW, VFREQ, VILIM, VEN ......................................................................................................................................0V to VIN
† Notice: Stresses above those listed under “Absolute Maximum Ratings” may cause permanent damage to the device.
This is a stress rating only and functional operation of the device at those or any other conditions above those indicated
in the operational sections of this specification is not intended. Exposure to maximum rating conditions for extended
periods may affect device reliability.
‡ Notice: The device is not guaranteed to function outside its operating ratings.
Note 1: Devices are ESD sensitive. Handling precautions are recommended. Human body model, 1.5 kΩ in series
with 100 pF.
2015 Microchip Technology Inc.
DS20005459B-page 3
MIC2125/6
TABLE 1-1:
ELECTRICAL CHARACTERISTICS
Electrical Characteristics: VIN = 12V, VOUT = 1.2V, VBST – VSW = 5V; TA = 25°C, unless noted. Bold values indicate
–40°C ≤ TJ ≤ +125°C. (Note 1).
Parameters
Min.
Typ.
Max.
Units
Conditions
Input Voltage Range (VIN)
(Note 2)
4.5
—
5.5
V
4.5
—
28
Quiescent Supply Current
(MIC2125)
—
340
750
µA
VFB = 1.5V
Quiescent Supply Current
(MIC2126)
—
1.1
3
mA
VFB = 1.5V
Shutdown Supply Current
—
0.1
5
µA
SW unconnected, VEN = 0V
VDD Output Voltage
4.8
5.2
5.4
V
VIN = 7V to 28V, IDD = 10 mA
VDD UVLO Threshold
3.7
4.2
4.5
VDD UVLO Hysteresis
—
400
—
mV
—
Load Regulation
0.6
2
3.6
%
IDD = 0 to 40 mA
0.597
0.6
0.603
V
TJ = 25°C (±0.5%)
0.594
0.6
0.606
—
0.01
0.5
µA
VFB = 0.6V
EN Logic Level High
1.6
—
—
V
—
EN Logic Level Low
—
—
0.6
EN Hysteresis
—
120
—
mV
—
EN Bias Current
—
6
30
µA
VEN = 12V
—
750
—
kHz
VFREQ = VIN
—
375
—
Maximum Duty Cycle
—
85
—
Minimum Duty Cycle
—
0
—
Minimum On-Time
—
100
—
Minimum Off-Time
150
220
300
—
7
—
ms
—
Current-Limit Comparator
Offset
–15
–4
7
mV
VFB = 0.6V
Current-Limit Source Current
32
36
40
µA
VFB = 0.6V
Power Supply Input
VDD = VIN
—
VDD Supply
VDD rising
Reference
Feedback Reference Voltage
FB Bias Current
–40°C ≤ TJ ≤ +125°C (±1%)
Enable Control
—
Oscillator
Switching Frequency
VFREQ = 50% x VIN
%
—
VFB > 0.6V
ns
—
—
Soft-Start
Soft-Start Time
Short-Circuit Protection and OVP
Note 1:
2:
Specification for packaged product only.
The application is fully functional at low VDD (supply of the control section) if the external MOSFETs have
low voltage VTH.
DS20005459B-page 4
2015 Microchip Technology Inc.
MIC2125/6
TABLE 1-1:
ELECTRICAL CHARACTERISTICS (CONTINUED)
Electrical Characteristics: VIN = 12V, VOUT = 1.2V, VBST – VSW = 5V; TA = 25°C, unless noted. Bold values indicate
–40°C ≤ TJ ≤ +125°C. (Note 1).
Parameters
Min.
Typ.
Max.
Units
——
0.62
—
V
—
DH, DL Output Low Voltage
—
—
0.1
V
ISINK = 10 mA
DH, DL Output High Voltage
VPVDD-0.1
or
VBST-0.1
—
—
DH On-Resistance, High State
—
2.5
—
DH On-Resistance, Low State
—
1.6
—
—
DL On-Resistance, High State
—
1.9
—
—
DL On-Resistance, Low State
—
0.55
—
—
SW, BST Leakage Current
—
—
50
µA
PG Threshold Voltage
85
89
95
%VOUT
PG Hysteresis
—
6
—
PG Delay Time
—
80
—
µs
Sweep VFB from low to high
PG Low Voltage
—
60
200
mV
VFB < 90% x VNOM, IPG = 1 mA
Overtemperature Shutdown
—
150
—
°C
TJ Rising
Overtemperature Shutdown
Hysteresis
—
15
—
°C
—
Overvoltage Protection
Threshold
Conditions
FET Drivers
ISOURCE = 10 mA
Ω
—
—
Power Good (PG)
Sweep VFB from low to high
Sweep VFB from high to low
Thermal Protection
Note 1:
2:
Specification for packaged product only.
The application is fully functional at low VDD (supply of the control section) if the external MOSFETs have
low voltage VTH.
2015 Microchip Technology Inc.
DS20005459B-page 5
MIC2125/6
TEMPERATURE SPECIFICATIONS
Parameters
Sym.
Min.
Typ.
Max.
Units
Conditions
Junction Operating Temperature
TJ
–40
—
+125
°C
Note 1
Temperature Ranges
Storage Temperature Range
TS
–65
—
+150
°C
—
Junction Temperature
TJ
—
—
+150
°C
—
Lead Temperature
—
—
—
+260
°C
Soldering, 10s
JA
—
50.8
—
°C/W
—
JC
—
25.3
—
°C/W
—
Package Thermal Resistances
Thermal Resistance 3 mm x 3 mm
QFN-16LD
Note 1:
The maximum allowable power dissipation is a function of ambient temperature, the maximum allowable
junction temperature and the thermal resistance from junction to air (i.e., TA, TJ, JA). Exceeding the
maximum allowable power dissipation will cause the device operating junction temperature to exceed the
maximum +125°C rating. Sustained junction temperatures above +125°C can impact the device reliability.
DS20005459B-page 6
2015 Microchip Technology Inc.
MIC2125/6
2.0
Note:
TYPICAL PERFORMANCE CURVES
The graphs and tables provided following this note are a statistical summary based on a limited number of
samples and are provided for informational purposes only. The performance characteristics listed herein
are not tested or guaranteed. In some graphs or tables, the data presented may be outside the specified
operating range (e.g., outside specified power supply range) and therefore outside the warranted range.
Note: Unless otherwise noted, VIN = 12V, FREQ = 350 kHz.
FIGURE 2-1:
VIN Operating Supply
Current vs. Input Voltage (MIC2125).
FIGURE 2-4:
VIN Shutdown Current vs.
Input Voltage (MIC2125).
FIGURE 2-2:
Feedback Voltage vs. Input
Voltage (MIC2125).
FIGURE 2-5:
Input Voltage.
FIGURE 2-3:
Output Voltage vs. Input
Voltage (MIC2125).
FIGURE 2-6:
Switching Frequency vs.
Temperature (MIC2126).
2015 Microchip Technology Inc.
Switching Frequency vs.
DS20005459B-page 7
MIC2125/6
Note: Unless otherwise noted, VIN = 12V, FREQ = 350 kHz.
.V
FIGURE 2-7:
VDD Voltage vs. Input
Voltage (MIC2125).
FIGURE 2-10:
VIN Operating Supply
Current vs. Temperature (MIC2125).
FIGURE 2-8:
Enable Threshold vs. Input
Voltage (MIC2125).
FIGURE 2-11:
Feedback Voltage vs.
Temperature (MIC2125).
FIGURE 2-9:
Output Peak Current Limit
vs. Input Voltage (MIC2125).
FIGURE 2-12:
Load Regulation vs.
Temperature (MIC2125).
DS20005459B-page 8
2015 Microchip Technology Inc.
MIC2125/6
Note: Unless otherwise noted, VIN = 12V, FREQ = 350 kHz.
FIGURE 2-13:
VIN Shutdown Current vs.
Temperature (MIC2125).
FIGURE 2-16:
EN Bias Current vs.
Temperature (MIC2125).
FIGURE 2-14:
VDD UVLO Threshold vs.
Temperature (MIC2125).
FIGURE 2-17:
VDD Voltage vs.
Temperature (MIC2125).
FIGURE 2-15:
Enable Threshold vs.
Temperature (MIC2125).
FIGURE 2-18:
Current-Limit Source
Current vs. Temperature (MIC2125).
2015 Microchip Technology Inc.
DS20005459B-page 9
MIC2125/6
Note: Unless otherwise noted, VIN = 12V, FREQ = 350 kHz.
*Note: For Case Temperature graphs: The temperature measurement was taken at the hottest point on the MIC2125/6
case mounted on a 5 square inch PCBn. Actual results will depend upon the size of the PCB, ambient temperature and
proximity to other heat emitting components.
FIGURE 2-19:
Line Regulation vs.
Temperature (MIC2125).
FIGURE 2-22:
Output Regulation vs. Input
Voltage (MIC2125).
FIGURE 2-20:
Feedback Voltage vs.
Output Current (MIC2125).
FIGURE 2-23:
Case Temperature* vs.
Output Current (MIC2125).
FIGURE 2-21:
Line Regulation vs. Output
Current (MIC2125).
FIGURE 2-24:
Case Temperature* vs.
Output Current (MIC2125).
DS20005459B-page 10
2015 Microchip Technology Inc.
MIC2125/6
Note: Unless otherwise noted, VIN = 12V, FREQ = 350 kHz.
*Note: For Case Temperature graphs: The temperature measurement was taken at the hottest point on the MIC2125/6
case mounted on a 5 square inch PCBn. Actual results will depend upon the size of the PCB, ambient temperature and
proximity to other heat emitting components.
FIGURE 2-25:
Case Temperature* vs.
Output Current (MIC2125).
FIGURE 2-28:
Efficiency (VIN = 18V) vs.
Output Current (MIC2125).
FIGURE 2-26:
Efficiency (VIN = 5V) vs.
Output Current (MIC2125).
FIGURE 2-29:
Efficiency (VIN = 5V) vs.
Output Current (MIC2126).
FIGURE 2-27:
Efficiency (VIN = 12V) vs.
Output Current (MIC2125).
FIGURE 2-30:
Efficiency (VIN = 12V) vs.
Output Current (MIC2126).
2015 Microchip Technology Inc.
DS20005459B-page 11
MIC2125/6
Note: Unless otherwise noted, VIN = 12V, FREQ = 350 kHz.
IN
VIN
(10V/div)
VSW
(10V/div)
VIN = 12V
VOUT = 1.2V
IOUT = 0A
VPRE-BIAS = 0.5V
VOUT
(500mV/div)
Time (10ms/div)
FIGURE 2-31:
Efficiency (VIN = 18V) vs.
Output Current (MIC2126).
FIGURE 2-34:
MIC2125 VIN Start-Up with
Prebiased Output.
IN
VIN = 12V
VOUT = 1.2V
IOUT = 20A
VIN
(10V/div)
VSW
(10V/div)
VEN
(2V/div)
VOUT
(1V/div)
VOUT
(2V/div)
IL
(20A/div)
IL
(20A/div)
Time (10ms/div)
Time (10ms/div)
FIGURE 2-32:
VIN Soft Turn-On.
FIGURE 2-35:
VIN
(10V/div)
VIN = 12V
VOUT = 1.2V
IOUT = 20A
VSW
(10V/div)
VIN = 12V
VOUT = 1.2V
IOUT = 20A
VEN
(2V/div)
VOUT
(1V/div)
VOUT
(2V/div)
IL
(20A/div)
IL
(20A/div)
Time (10ms/div)
DS20005459B-page 12
Enable Turn-On/Turn-Off.
y
IN
FIGURE 2-33:
VIN = 12V
VOUT = 1.2V
IOUT = 20A
VIN Soft Turn-Off.
Time (4ms/div)
FIGURE 2-36:
Rise Time.
Enable Turn-On Delay and
2015 Microchip Technology Inc.
MIC2125/6
Note: Unless otherwise noted, VIN = 12V, FREQ = 350 kHz.
VIN = 12V
VOUT = 1.2V
IOUT = 20A
VEN
(2V/div)
VIN = 12V
VOUT = 1.2V
IOUT = Short
VEN
(2V/div)
VOUT
(500mV/div)
VOUT
(1V/div)
IL
(20A/div)
IL
(10A/div)
Time (200μs/div)
FIGURE 2-37:
Fall Time.
Time (4ms/div)
Enable Turn-Off Delay and
VIN = 12V
VOUT = 1.2V
IOUT = 20A
VEN
(1V/div)
FIGURE 2-40:
Enabled into Short.
VIN = 12V
VOUT = 1.2V
IOUT = Short
VIN
(2V/div)
VOUT
(500mV/div)
VOUT
(1V/div)
IL
(10A/div)
Time (4ms/div)
Time (10ms/div)
FIGURE 2-38:
Enable Thresholds.
FIGURE 2-41:
Power-Up into Short-Circuit.
p
VOUT = 1.2V
IOUT = 1A
VIN = 12V
VOUT = 1.2V
VIN
(2V/div)
VOUT
(500mV/div)
VOUT
(500mV/div)
IOUT
(10A/div)
Time (20ms/div)
Time (20ms/div)
FIGURE 2-39:
Rise Time.
Enable Turn-On Delay and
2015 Microchip Technology Inc.
FIGURE 2-42:
Threshold.
Output Peak Current-Limit
DS20005459B-page 13
MIC2125/6
Note: Unless otherwise noted, VIN = 12V, FREQ = 350 kHz.
VIN = 12V
VOUT = 1.2V
IOUT = 10A to Short
VOUT
(500mV/div)
VIN = 12V
VOUT = 1.2V
IOUT = 2A to 12A
VOUT
(50mV/div)
(AC-Coupled)
IOUT
(10A/div)
IL
(10A/div)
Time (8ms/div)
FIGURE 2-43:
Time (100μs/div)
Short-Circuit.
p
FIGURE 2-46:
Transient Response.
y
OUT
VIN = 12V
ILDO = 1.2V
VIN = Short to 10A
VOUT
(20mV/div)
(AC-coupled)
VIN = 12V
VOUT = 1.2V
IOUT = 0A
VOUT
(500mV/div)
VSW
(5V/div)
IL
(10A/div)
IL
(2A/div)
Time (8ms/div)
Time (8ms/div)
FIGURE 2-44:
Short-Circuit.
p
Output Recovery from
FIGURE 2-47:
MIC2125 Switching
Waveform, IOUT = 0A.
y
OUT
vin = 12V
VOUT = 1.2V
IOUT = 2.5A
VOUT
(500mV/div)
VOUT
(20mV/div)
(AC-coupled)
VIN = 12V
VOUT = 1.2V
IOUT = 0.1A
VSW
(5V/div)
IL
(2A/div)
vsw
(5V/div)
Time (2ms/div)
FIGURE 2-45:
Output Recovery from
Thermal Shutdown.
DS20005459B-page 14
Time (4μs/div)
FIGURE 2-48:
MIC2125 Switching
Waveform, IOUT = 0.1A.
2015 Microchip Technology Inc.
MIC2125/6
Note: Unless otherwise noted, VIN = 12V, FREQ = 350 kHz.
OUT
OUT
VOUT
(20mV/div)
(AC-coupled)
VIN = 12V
VOUT = 1.2V
IOUT = 10A
VIN = 12V
VOUT = 1.2V
IOUT = 0.1A
IL
(2V/div)
VSW
(10V/div)
VSW
(5V/div)
VDH
(10V/div)
IL
(10A/div)
VDL
(5V/div)
Time (4μs/div)
Time (2μs/div)
FIGURE 2-49:
10A.
Switching Waveform, IOUT =
FIGURE 2-52:
MIC2125 Switching
Waveform, IOUT = 0.1A.
IN
OUT
VIN = 12V
VOUT = 1.2V
IOUT = 0A
VOUT
(20mV/div)
(AC-Coupled)
VIN
(5V/div)
VOUT
(1V/div)
VSW
(5V/div)
VIN = 12V
VOUT = 1.2V
IOUT = 20A
IL
(10A/div)
VPG
(5V/div)
Time (2μs/div)
FIGURE 2-50:
20A.
Time (4ms/div)
Switching Waveform, IOUT =
FIGURE 2-53:
Turn-On.
IN
OUT
VIN = 12V
VOUT = 1.2V
IOUT = 0A
VIN = 12V
VOUT = 1.2V
IOUT = 0A
IL
(2A/div)
VIN
(5V/div)
VSW
(10V/div)
VOUT
(1V/div)
VDH
(10V/div)
VDL
(5V/div)
Power Good at VIN Soft
VPG
(5V/div)
Time (20ms/div)
Time (4μs/div)
FIGURE 2-51:
MIC2125 Switching
Waveform, IOUT = 0A.
2015 Microchip Technology Inc.
FIGURE 2-54:
Turn-Off.
Power Good at VIN Soft
DS20005459B-page 15
MIC2125/6
3.0
PIN DESCRIPTIONS
The descriptions of the pins are listed in Table 3-1.
TABLE 3-1:
PIN FUNCTION TABLE
Pin Number
Symbol
Description
1
VDD
Internal Linear regulator output. Connect a 4.7 μF ceramic capacitor from VDD to
AGND for decoupling. In the applications where VIN < +5.5V, VDD should be tied to
VIN to by-pass the linear regulator.
2
PVDD
5V supply input for the low-side N-channel MOSFET driver, which can be tied to
VDD externally. A 4.7 μF ceramic capacitor from PVDD to PGND is recommended for
decoupling.
3
ILIM
Current limit setting input. Connect a resistor from SW to ILIM to set the overcurrent
threshold for the converter.
Low-side gate driver output. The DL driving voltage swings from ground to VDD.
4
DL
5
PGND
Power ground. PGND is the return path for the low side gate driver. Connect PGND
pin to the source of low-side N-Channel external MOSFET.
6
FREQ
Switching frequency adjust input. Connect FREQ to the mid-point of an external
resistor divider from VIN to GND to program the switching frequency. Tie to VIN to
operate at 750 kHz frequency.
7
DH
High-side gate driver output. The DH driving voltage is floating on the switch node
voltage (VSW).
8
SW
Switch node and current-sense input. Connect the SW pin to the switch node of the
buck converter. The SW pin also senses the current by monitoring the voltage
across the low-side MOSFET during OFF time. In order to sense the current
accurately, connect the low-side MOSFET drain to the SW pin using a Kelvin
connection.
9
BST
Bootstrap Capacitor Input. Connect a ceramic capacitor with a minimum value of
0.1 μF from BST to SW.
10
OVP
Output Overvoltage Protection Input. Connect to the mid-point of an external
resistive divider from the VOUT to GND to program overvoltage limit. Connect to
AGND if the output overvoltage protection is not required.
11
NC
12
AGND
13
FB
Feedback input. Input to the transconductance amplifier of the control loop. The FB
pin is regulated to 0.6V. A resistor divider connecting the feedback to the output is
used to set the desired output voltage.
14
PG
Open-drain Power good output. Pull-up with an external pull-up resistor to VDD or to
an external power rail.
15
EN
Enable input. A logic signal to enable or disable the buck converter operation.
Logic-high enables the device; logic-low shuts down the regulator. In disable mode,
the VDD supply current for the device is minimized to 0.1 µA typically. Do not pull-up
EN pin to VDD/PVDD.
16
VIN
Supply voltage input. The VIN operating voltage range is from 4.5V to 28V. A 1 μF
ceramic capacitor from VIN to AGND is required for decoupling.
17
EP
Exposed Pad. Connect the exposed pad to the AGND copper plane to improve the
thermal performance.
DS20005459B-page 16
No connect.
Analog Ground. Connect AGND to the exposed pad.
2015 Microchip Technology Inc.
MIC2125/6
4.0
FUNCTIONAL DESCRIPTION
The MIC2125 and MIC2126 are adaptive on-time
synchronous buck controllers built for high input
voltage to low output voltage applications. They are
designed to operate over a wide input voltage range
from 4.5V to 28V and their output is adjustable with an
external resistive divider. An adaptive ON-time control
scheme is employed to obtain a constant switching
frequency and to simplify the control compensation.
Overcurrent protection is implemented when sensing
low-side MOSFET’s RDS(ON). The device features
internal soft-start, enable, UVLO, and thermal
shutdown.
4.1
Theory of Operation
The MIC2125/6 Functional Block Diagram appears on
page two. The output voltage is sensed by the
MIC2125/6 feedback pin (FB), and is compared to a
0.6V reference voltage (VREF) at the low gain
transconductance error amplifier (gm). Figure 4-1
shows the MIC2125/6 control loop timing during
steady-state operation. When the feedback voltage
decreases and the amplifier output is below 0.6V, the
comparator triggers and generates an ON-time period.
The ON-time period is predetermined by the fixed tON
estimator circuitry value from Equation 4-1:
EQUATION 4-2:
t S – t OFF MIN
220ns
D MAX = ----------------------------------- = 1 – --------------tS
tS
Where:
tS
1/fSW
It is not recommended to use MIC2125/6 with an
OFF-time close to tOFF(MIN) during steady-state
operation.
The adaptive ON-time control scheme results in a
constant switching frequency in the MIC2125/6. The
actual ON-time and resulting switching frequency
varies with the different rising and falling times of the
external MOSFETs. Also, the minimum tON results in a
lower switching frequency in high VIN to VOUT
applications.
EQUATION 4-1:
V OUT
t ON ESTIMATED = ----------------------V IN f SW
Where:
VOUT
Output Voltage
VIN
Power Stage Input Voltage
fSW
Switching Frequency
At the end of the ON-time, the internal high-side driver
turns off the high-side MOSFET and the low-side driver
turns on the low-side MOSFET. The OFF-time depends
upon the feedback voltage. When the feedback voltage
decreases and the output of the gm amplifier is below
0.6V, the ON-time period is triggered and the OFF-time
period ends. If the OFF-time period determined by the
feedback voltage is less than the minimum OFF-time
tOFF(min), which is about 220 ns, the MIC2125/6 control
logic applies the tOFF(min) instead. tOFF(min) is required
to maintain enough energy in the boost capacitor
(CBST) to drive the high-side MOSFET.
The maximum duty cycle is obtained from the 220 ns
tOFF(MIN):
2015 Microchip Technology Inc.
FIGURE 4-1:
Timing
MIC2125/6 Control Loop
Figure 4-2 shows the operation of the MIC2125/6
during load transient. The output voltage drops due to
a sudden increase in load, which results in the VFB
falling below VREF. This causes the comparator to
trigger an ON-time period. At the end of the ON-time, a
minimum OFF-time tOFF(min) is generated to charge
CBST if the feedback voltage is still below VREF. The
next ON-time is triggered immediately after the
tOFF(min) due to the low feedback voltage. This
operation results in higher switching frequency during
load transients. The switching frequency returns to the
nominal set frequency once the output stabilizes at new
load current level. The output recovery time is fast and
the output voltage deviation is small in MIC2125/6
converter due to the varying duty cycle and switching
frequency.
DS20005459B-page 17
MIC2125/6
IL CROSSES 0 AND VFB > 0.6.
DISCONTINUOUS CONDUCTION MODE STARTS.
IL
VFB > 0.8. WAKE UP FROM
DISCONTINUOUS CONDUCTION MODE.
0
VFB
VREF
ZC
VHSD
FIGURE 4-2:
Response
MIC2125/6 Load Transient
Unlike true current-mode control, the MIC2125/6 uses
the output voltage ripple to trigger an ON-time period.
In order to meet the stability requirements, the
MIC2125/6 feedback voltage ripple should be in phase
with the inductor current ripple and large enough to be
sensed by the gm amplifier. The recommended
feedback voltage ripple is 20 mV ~ 100 mV over the full
input voltage range. If a low-ESR output capacitor is
selected, then the feedback voltage ripple may be too
small to be sensed by the gm amplifier. Also, the output
voltage ripple and the feedback voltage ripple are not
necessarily in phase with the inductor current ripple if
the ESR of the output capacitor is very low. For these
applications, ripple injection is required to ensure
proper operation. Refer to the Ripple Injection section
under Application Information for details about the
ripple injection technique.
4.2
Discontinuous Conduction Mode
(MIC2125 Only)
ESTIMATED ON TIME
VLSD
FIGURE 4-3:
MIC2125 Control Loop
Timing (Discontinuous Conduction Mode)
The typical no load supply current during discontinuous
conduction mode is only about 340 μA, allowing the
MIC2125 to achieve high efficiency at light load
operation.
4.3
Soft-Start
Soft-start reduces the power supply inrush current at
startup by controlling the output voltage rise time. The
MIC2125/6 implements an internal digital soft-start by
ramping up the reference voltage VREF from 0 to 100%
in about 7 ms. Once the soft-start is completed, the
related circuitry is disabled to reduce the current
consumption.
The MIC2125 operates in discontinuous conduction
mode at light load. The MIC2125 has a zero crossing
comparator (ZC detection) that monitors the inductor
current by sensing the voltage drop across the low-side
MOSFET during its ON-time. If the VFB > 0.6V and the
inductor current goes slightly negative, the MIC2125
turns off both the high-side and low-side MOSFETs.
During this period, the efficiency is optimized by
shutting down all the non-essential circuits and the load
current is supplied by the output capacitor. The control
circuitry wakes up when the feedback voltage falls
below VREF and triggers a tON pulse. Figure 4-3 shows
the control loop timing in discontinuous conduction
mode.
DS20005459B-page 18
2015 Microchip Technology Inc.
MIC2125/6
4.4
Current Limit
The MIC2125/6 uses the low-side MOSFET RDS(ON) to
sense the inductor current.
Because MOSFET RDS(ON) varies from 30% to 40%
with temperature, it is recommended to add a 50%
margin to ICL in the previous equation to avoid false
current limiting due to increased MOSFET junction
temperature rise. It is also recommended to connect
the SW pin directly to the drain of the low-side
MOSFET to accurately sense the MOSFET’s RDS(ON).
4.5
Negative Current Limit
(MIC2126 Only)
The MIC2126 implements negative current limit by
sensing the SW voltage when the low-side FET is off.
If the SW node voltage exceeds 12 mV typical, the
device turns off the low-side FET until the next ON-time
event is triggered. The negative current limit value is
given by Equation 4-4.
EQUATION 4-4:
FIGURE 4-4:
Circuit
MIC2125/6 Current-Limiting
In each switching cycle of the MIC2125/6 converter, the
inductor current is sensed by monitoring the voltage
across the low-side MOSFET during the OFF period.
An internal current source of 36 µA generates a voltage
across the external resistor RCL. The ILIM pin voltage
V(ILIM) is the sum of the voltage across the low side
MOSFET and the voltage across the resistor (VCL).
The sensed voltage V(ILIM) is compared with the power
ground (PGND) after a blanking time of 150 ns.
If the absolute value of the voltage drop across the low
side MOSFET is greater than VCL, the current limit
event is triggered. Eight consecutive current limit
events triggers hiccup mode. The hiccup sequence,
including the soft-start, reduces the stress on the
switching FETs and protects the load and supply from
severe short conditions.
The current limit can be programmed by using
Equation 4-3.
EQUATION 4-3:
I CLIM + PP 0.5 R DS ON – V OFFSET
R CL = --------------------------------------------------------------------------------------------------------I CL
Where:
INLIM
Negative Current Limit
RDS(ON)
On-Resistance of Low-Side Power
MOSFET
4.6
Desired Current Limit
∆PP
Inductor Current Peak-to-Peak
RDS(ON)
On-Resistance of Low-Side Power
MOSFET
VOFFSET
Current-Limit Comparator Offset (Typical
Value is –4 mV per Table 1-1)
ICL
Current-Limit Source Current (Typical
Value is 36 µA, per Table 1-1)
2015 Microchip Technology Inc.
MOSFET Gate Drive
The MIC2125/6 high-side drive circuit is designed to
switch an N-Channel MOSFET. Figure 4-1 shows a
bootstrap circuit, consisting of a PMOS switch and
CBST. This circuit supplies energy to the high-side drive
circuit. Capacitor CBST is charged while the low-side
MOSFET is on and the voltage on the SW pin is
approximately 0V. When the high-side MOSFET driver
is turned on, energy from CBST is used to turn the
MOSFET on. If the bias current of the high-side driver
is less than 10 mA, a 0.1 μF capacitor is sufficient to
hold the gate voltage within minimal droop, (i.e., ∆BST
= 10 mA × 3.33 μs/0.1 μF = 333 mV). A small resistor,
RG in series with CBST, can be used to slow down the
turn-on time of the high-side N-channel MOSFET.
4.7
Where:
ICLIM
12mV
I NLIM = -------------------R DS ON
Overvoltage Protection
The MIC2125/6 includes the OVP feature to protect the
load from overshoots due to input transients and output
short to a high voltage. When the overvoltage condition
is triggered, the converter turns off immediately to allow
the output voltage to discharge. The MIC2125/6 power
should be recycled to enable it again.
DS20005459B-page 19
MIC2125/6
5.0
APPLICATION INFORMATION
5.2
5.1
Setting the Switching Frequency
Voltage rating, on-resistance, and total gate charge are
important parameters for MOSFET selection.
The
MIC2125/6
are
adjustable-frequency,
synchronous buck controllers featuring a unique
adaptive ON–time control architecture. The switching
frequency can be adjusted between 200 kHz and
750 kHz by changing the resistor divider network
consisting of R19 and R20.
MIC2125/26
AGND
The voltage rating for the high-side and low-side
MOSFETs are essentially equal to the power stage
input voltage VIN. A safety factor of 30% should be
added to the VIN(MAX) while selecting the voltage rating
of the MOSFETs to account for voltage spikes due to
circuit parasitic elements.
The power dissipated in the MOSFETs is the sum of
conduction losses (PCONDUCTION) and switching
losses (PAC).
VDD/PVDD
VDD
5V
4.7μF
MOSFET Selection
BST
EQUATION 5-2:
P SW = P CONDUCTION + P AC
VIN
VIN
SW
CS
EQUATION 5-3:
R19
2.2μF
x3
FREQ
R20
FB
2
P CONDUCTION = I SW RMS R DS ON
PGND
Where:
FIGURE 5-1:
Adjustment.
Switching Frequency
RDS(ON)
On-Resistance of the MOSFET
ISW(RMS)
RMS current of the MOSFET
Equation 5-1 gives the estimated switching frequency.
The total high-side MOSFET switching loss is:
EQUATION 5-1:
R20
f SW ADJ = f O -------------------------R19 + R20
Where:
fO
Switching Frequency when R19 is
100 kΩ and R20 is open. fO is typically
750 kHz.
For more precise setting, it is recommended to use
Figure 5-2.
EQUATION 5-4:
P AC = 0.5 V IN I LOAD t R + t F f SW
Where:
tR/tF
Switching Transition Times
ILOAD
Load Current
fSW
Switching Frequency
Turn-on and turn-off
approximated by:
transition
times
can
be
EQUATION 5-5:
Q SW HS R HSD PULL – UP + R HS GATE
t R = ----------------------------------------------------------------------------------------------------------V DD – V TH
FIGURE 5-2:
R20
DS20005459B-page 20
Switching Frequency vs.
2015 Microchip Technology Inc.
MIC2125/6
EQUATION 5-6:
Q SW HS R HSD PULL – UP + R HS GATE
t F = ----------------------------------------------------------------------------------------------------------V TH
The peak inductor current is equal to the average
output current plus one half of the peak-to-peak
inductor current ripple.
EQUATION 5-9:
Where:
RHSD(PULL-UP)
I L PK = I OUT MAX + 0.5 I L PP
High-Side Gate Driver Pull-Up
Resistance
RHSD(PULL-DOWN) High-Side Gate Driver Pull-Down
Resistance
RHS(GATE)
High-Side MOSFET Gate
Resistance
QSW(HS)
Switching Gate Charge of the
High-Side MOSFET
VTH
Gate Threshold Voltage
The saturation current rating is given by:
EQUATION 5-10:
R CL I CL – V OFFSET
I L SAT = --------------------------------------------------------R DS ON
Where:
RCL
The high-side MOSFET switching losses increase with
the switching frequency and the input voltage. The
low-side MOSFET switching losses are negligible and
can be ignored for these calculations.
5.3
Inductor Selection
Inductance value, saturation, and RMS currents are
required to select the output inductor. The input and
output voltages and the inductance value determine
the peak-to-peak inductor ripple current. Larger
peak-to-peak ripple current increases the power
dissipation in the inductor and MOSFETs. Larger
output ripple current also requires more output
capacitance to smooth out the larger ripple current.
Smaller peak-to-peak ripple current requires a larger
inductance value and therefore a larger and more
expensive inductor.
A good compromise between size, loss, and cost is to
set the inductor ripple current to be equal to 40% of the
maximum output current.
The inductance value is calculated by Equation 5-7.
EQUATION 5-7:
V OUT V IN MAX – V OUT
L = ------------------------------------------------------------------------------------V IN MAX f SW 0.4 I OUT MAX
Where:
fSW
Switching Frequency
0.4
Ratio of AC Ripple Current to DC Output
Current
VIN(MAX)
Maximum Power Stage Input Voltage
The peak-to-peak inductor current ripple is:
Current-Limit Resistor
ICL
Current-Limit Source Current
VOFFSET
Current-Limit Comparator Offset
RDS(ON)
On-Resistance of Low-Side Power
MOSFET
The RMS inductor current is used to calculate the I2R
losses in the inductor.
EQUATION 5-11:
2
I L RMS =
I OUT MAX
2
I L PP
+ --------------------12
Maximizing efficiency requires the proper selection of
core material and minimizing the winding resistance.
The high-frequency operation of the MIC2125/6
requires the use of ferrite materials. Lower cost iron
powder cores may be used, but the increase in core
loss reduces the efficiency of the power supply. This is
especially noticeable at low output power. The winding
resistance decreases efficiency at the higher output
current levels. The winding resistance must be
minimized, although this usually comes at the expense
of a larger inductor. The power dissipated in the
inductor is equal to the sum of the core and copper
losses. At higher output loads, the core losses are
usually insignificant and can be ignored. At lower
output currents, the core losses can be significant.
Core loss information is usually available from the
magnetics vendor.
The amount of copper loss in the inductor is calculated
by Equation 5-12:
EQUATION 5-12:
2
EQUATION 5-8:
P INDUCTOR CU = I L RMS R WINDING
V OUT V IN MAX – V OUT
I L PP = -------------------------------------------------------------------V IN MAX f SW L
2015 Microchip Technology Inc.
DS20005459B-page 21
MIC2125/6
5.4
Output Capacitor Selection
The type of the output capacitor is usually determined
by its equivalent series resistance (ESR). Voltage and
RMS current capability are two other important factors
for selecting the output capacitor. Recommended
capacitor types are ceramic, tantalum, low-ESR
aluminum electrolytic, OS-CON, and POSCAP. The
output capacitor’s ESR is usually the main cause of the
output ripple. The output capacitor ESR also affects the
control loop from a stability point of view. The maximum
value of ESR is calculated by Equation 5-13.
EQUATION 5-13:
ESR C
V OUT PP
--------------------------OUT
I L PP
The power dissipated in the output capacitor is:
EQUATION 5-16:
2
P DISS COUT = I COUT RMS ESR COUT
5.5
Input Capacitor Selection
The input capacitor reduces peak current drawn from
the power supply and reduces noise and voltage ripple
on the input. The input voltage ripple depends on the
input capacitance and ESR. The input capacitance and
ESR values are calculated by using Equation 5-17 and
Equation 5-18.
EQUATION 5-17:
Where:
∆VOUT(PP)
Peak-to-Peak Output Voltage Ripple
∆IL(PP)
Peak-to-Peak Inductor Current Ripple
The required output capacitance is calculated in
Equation 5-14.
EQUATION 5-14:
I L PP
C OUT = --------------------------------------------------V OUT PP f SW 8
I OUT D 1 – D
C IN = ---------------------------------------------- V IN C f SW
Where:
IOUT
ƞ
Power Conversion Efficiency
∆VIN(C)
Input Ripple Due to Capacitance Value
EQUATION 5-18:
V IN ESR
ESR CIN = ------------------------I L PK
Where:
COUT
Output Capacitance Value
fSW
Switching Frequency
As described in the Theory of Operation subsection of
the Functional Description, the MIC2125/26 requires at
least 20 mV peak-to-peak ripple at the FB pin to ensure
that the gm amplifier and the comparator behave
properly. Also, the output voltage ripple should be in
phase with the inductor current. Therefore, the output
voltage ripple caused by the output capacitors value
should be much smaller than the ripple caused by the
output capacitor ESR. If low-ESR capacitors, such as
ceramic capacitors, are selected as the output
capacitors, a ripple injection method should be applied
to provide the enough feedback voltage ripple. Refer to
the Ripple Injection subsection for details.
The voltage rating of the capacitor should be twice the
output voltage for a tantalum and 20% greater for
aluminum electrolytic or OS-CON. The output capacitor
RMS current is calculated in Equation 5-15.
EQUATION 5-15:
IC
OUT RMS
DS20005459B-page 22
Load Current
Where:
∆VIN(ESR)
Input Ripple Due to Capacitor ESR
Value
IL(PK)
Peak Inductor Current
The input capacitor should be qualified for ripple
current rating and voltage rating. The RMS value of the
input capacitor current is determined at the maximum
output current. Assuming the peak-to-peak inductor
current ripple is low:
EQUATION 5-19:
I CIN RMS I OUT MAX D 1 – D
The power dissipated in the input capacitor is:
EQUATION 5-20:
2
I L PP
= -----------------12
P DISS CIN = I CIN RMS ESR CIN
2015 Microchip Technology Inc.
MIC2125/6
5.6
Output Voltage Setting
5.7
The MIC2125/26 requires two resistors to set the
output voltage, as shown in Figure 5-3.
Output Overvoltage Limit Setting
The output overvoltage limit should be typically 20%
higher than the nominal output voltage. Set the OVP
limit by connecting a resistor divider from the output to
ground as shown in Figure 5-4.
R1
FB
gm AMP
R1
OVP
R2
R2
VREF
VREF
FIGURE 5-3:
Configuration.
Voltage-Divider
The output voltage is determined by Equation 5-21:
FIGURE 5-4:
Configuration.
OVP Voltage-Divider
Choose R2 in the range of 10 kΩ to 49.9 kΩ and
calculate R1 using Equation 5-23.
EQUATION 5-21:
R1
V OUT = V FB 1 + -------
R2
EQUATION 5-23:
V OVP
R1 = R2 ------------- – 1
0.6
Where:
VFB
0.6V
A typical value of R1 can be in the range of 3 kΩ and
15 kΩ. If R1 is too large, it may allow noise to be
introduced into the voltage feedback loop. If R1 is too
small in value, it will decrease the efficiency of the
power supply, especially at light loads. Once R1 is
selected, R2 can be calculated using Equation 5-22.
EQUATION 5-22:
V FB R1
R2 = ----------------------------V OUT – V FB
5.8
Ripple Injection
The VFB ripple required for proper operation of the
MIC2125/6 gm amplifier and comparator is 20 mV to
100 mV. However, the output voltage ripple is generally
designed as 1% to 2% of the output voltage. For low
output voltages, such as a 1V, the output voltage ripple
is only 10 mV to 20 mV, and the feedback voltage ripple
is less than 20 mV. If the feedback voltage ripple is so
small that the gm amplifier and comparator cannot
sense it, then the MIC2125/6 loses control and the
output voltage is not regulated. In order to have
sufficient VFB ripple, a ripple injection method should
be applied for low output voltage ripple applications.
The applications are divided into three situations
according to the amount of the feedback voltage ripple:
• Enough ripple at the feedback voltage due to the
large ESR of the output capacitors (Figure 5-5).
The converter is stable without any ripple
injection.
2015 Microchip Technology Inc.
DS20005459B-page 23
MIC2125/6
SW
L
MIC2125/26 FB
FIGURE 5-5:
R1
COUT
R2
ESR
R2
V FB PP = -------------------- ESR C I L PP
OUT
R1 + R2
Where:
Peak-to-Peak Value of the Inductor
Current Ripple
• Inadequate ripple at the feedback voltage due to
the small ESR of the output capacitors.
The output voltage ripple is fed into the FB pin
through a feed-forward capacitor, Cff in this situation,
as shown in Figure 5-7. The typical Cff value is
between 1 nF and 100 nF.
MIC2125/26 FB
L
R1
R2
Cff
R1
Cff
R2
COUT
ESR
Invisible Ripple at FB.
The process of sizing the ripple injection resistor and
capacitors is as follows.
EQUATION 5-24:
SW
MIC2125/26 FB RINJ
FIGURE 5-7:
Enough Ripple at FB.
The feedback voltage ripple is:
∆IL(PP)
L
CINJ
SW
COUT
ESR
• Select CINJ as 100 nF, which can be considered
as short for a wide range of the frequencies.
• Select Cff to feed all output ripples into the
feedback pin. Typical choice of Cff is 0.47 nF to
47 nF, if R1 and R2 are in the kΩ range. The Cff
value can be calculated using Equation 5-26:
EQUATION 5-26:
t S V IN D 1 – D
1
C ff » ------ ------------------------------------------------------------------------------
V IN D 1 – D – V FB PP
RP
Where:
VIN
Power Stage Input Voltage
D
Duty Cycle
tS
1/fSW
RP
(R1//R2//RINJ)
∆VFB(PP)
Feedback Ripple
• Select RINJ according to Equation 5-27.
EQUATION 5-27:
FIGURE 5-6:
Inadequate Ripple at FB.
With the feed-forward capacitor, the feedback
voltage ripple is very close to the output voltage
ripple.
V IN D 1 – D
1
R INJ = ------- --------------------------------------------
C ff V FB PP f SW
EQUATION 5-25:
V FB PP ESR I L PP
• Virtually no ripple at the FB pin voltage due to the
very low ESR of the output capacitors.
Therefore, additional ripple is injected into the FB pin
from the switching node SW via a resistor RINJ and a
capacitor CINJ, as shown in Figure 5-7.
DS20005459B-page 24
2015 Microchip Technology Inc.
MIC2125/6
6.0
PCB LAYOUT GUIDELINES
PCB layout is critical to achieve reliable, stable and
efficient performance. The following guidelines should
be followed to ensure proper operation of the
MIC2125/26 converter.
6.1
IC
• The ceramic bypass capacitors which are
connected to the VDD and PVDD pins must be
located right at the IC. Use wide traces to connect
to the VDD, PVDD and AGND, PGND pins
respectively.
• The signal ground pin (AGND) must be connected
directly to the ground planes.
• Place the IC close to the point-of-load (POL).
• Signal and power grounds should be kept
separate and connected at only one location.
6.2
Input Capacitor
• Place the input ceramic capacitors as close as
possible to the MOSFETs.
• Place several vias to the ground plane close to
the input capacitor ground terminal.
6.3
Inductor
• Keep the inductor connection to the switch node
(SW) short.
• Do not route any digital lines underneath or close
to the inductor.
• Keep the switch node (SW) away from the
feedback (FB) pin.
• The SW pin should be connected directly to the
drain of the low-side MOSFET to accurately
sense the voltage across the low-side MOSFET.
6.4
Output Capacitor
• Use a copper plane to connect the output
capacitor ground terminal to the input capacitor
ground terminal.
• The feedback trace should be separate from the
power trace and connected as close as possible
to the output capacitor. Sensing a long
high-current load trace can degrade the DC load
regulation.
6.5
MOSFETs
• MOSFET gate drive traces must be short. The
ground plane should be the connection between
the MOSFET source and PGND.
• Choose a low-side MOSFET with a high CGS/CGD
ratio and a low internal gate resistance to
minimize the effect of dv/dt inducted turn-on.
• Use a 4.5V VGS rated MOSFET. Its higher gate
threshold voltage is more immune to glitches than
a 2.5V or 3.3V rated MOSFET.
For more information about the Evaluation board layout, please contact Microchip sales.
2015 Microchip Technology Inc.
DS20005459B-page 25
MIC2125/6
7.0
PACKAGING INFORMATION
16-Lead QFN 3 mm x 3 mm Package Outline and Recommended Land Pattern
Note:
For the most current package drawings, please see the Microchip Packaging Specification located at
http://www.microchip.com/packaging
DS20005459B-page 26
2015 Microchip Technology Inc.
MIC2125/6
Note:
For the most current package drawings, please see the Microchip Packaging Specification located at
http://www.microchip.com/packaging
2015 Microchip Technology Inc.
DS20005459B-page 27
MIC2125/6
DS20005459B-page 28
2015 Microchip Technology Inc.
MIC2125/6
APPENDIX A:
REVISION HISTORY
Revision A (November 2015)
• Original Conversion of this Document.
Revision B (December 2015)
• Corrected the erroneous listing of the MIC2126
example with a 64LD package. Replaced with correct 16LD package information.
2015 Microchip Technology Inc.
DS20005459B-page 29
MIC2125/6
NOTES:
DS20005459B-page 30
2015 Microchip Technology Inc.
MIC2125/6
PRODUCT IDENTIFICATION SYSTEM
To order or obtain information, e.g., on pricing or delivery, contact your local Microchip representative or sales office.
PART NO.
XX
X
Examples:
a)
MIC2125YML:
Device Temperature Package
Device:
MIC2125:
28V, Synchronous Buck Controller featuring Adaptive On-Time Control with HyperLight Load
28V, Synchronous Buck Controller featuring Adaptive On-Time Control with Hyper
Speed Control
MIC2126:
Temperature:
Y
=
Package:
ML
=
–40°C to +125°C
16-Pin 3 mm x 3 mm QFN
2015 Microchip Technology Inc.
28V, Synchronous Buck
Controller featuring Adaptive On-Time Control with
HyperLight Load,
–40°C to +125°C junction
temperature range,
16LD QFN
b)
MIC2126YML:
28V, Synchronous Buck
Controller featuring Adaptive On-Time Control with
Hyper Speed Control,
–40°C to +125°C junction
temperature range,
16LD QFN
DS2005459B-page 31
MIC2125/6
NOTES:
DS2005459B-page 32
2015 Microchip Technology Inc.
Note the following details of the code protection feature on Microchip devices:
•
Microchip products meet the specification contained in their particular Microchip Data Sheet.
•
Microchip believes that its family of products is one of the most secure families of its kind on the market today, when used in the
intended manner and under normal conditions.
•
There are dishonest and possibly illegal methods used to breach the code protection feature. All of these methods, to our
knowledge, require using the Microchip products in a manner outside the operating specifications contained in Microchip’s Data
Sheets. Most likely, the person doing so is engaged in theft of intellectual property.
•
Microchip is willing to work with the customer who is concerned about the integrity of their code.
•
Neither Microchip nor any other semiconductor manufacturer can guarantee the security of their code. Code protection does not
mean that we are guaranteeing the product as “unbreakable.”
Code protection is constantly evolving. We at Microchip are committed to continuously improving the code protection features of our
products. Attempts to break Microchip’s code protection feature may be a violation of the Digital Millennium Copyright Act. If such acts
allow unauthorized access to your software or other copyrighted work, you may have a right to sue for relief under that Act.
Information contained in this publication regarding device
applications and the like is provided only for your convenience
and may be superseded by updates. It is your responsibility to
ensure that your application meets with your specifications.
MICROCHIP MAKES NO REPRESENTATIONS OR
WARRANTIES OF ANY KIND WHETHER EXPRESS OR
IMPLIED, WRITTEN OR ORAL, STATUTORY OR
OTHERWISE, RELATED TO THE INFORMATION,
INCLUDING BUT NOT LIMITED TO ITS CONDITION,
QUALITY, PERFORMANCE, MERCHANTABILITY OR
FITNESS FOR PURPOSE. Microchip disclaims all liability
arising from this information and its use. Use of Microchip
devices in life support and/or safety applications is entirely at
the buyer’s risk, and the buyer agrees to defend, indemnify and
hold harmless Microchip from any and all damages, claims,
suits, or expenses resulting from such use. No licenses are
conveyed, implicitly or otherwise, under any Microchip
intellectual property rights unless otherwise stated.
Trademarks
The Microchip name and logo, the Microchip logo, dsPIC,
FlashFlex, flexPWR, JukeBlox, KEELOQ, KEELOQ logo, Kleer,
LANCheck, MediaLB, MOST, MOST logo, MPLAB,
OptoLyzer, PIC, PICSTART, PIC32 logo, RightTouch, SpyNIC,
SST, SST Logo, SuperFlash and UNI/O are registered
trademarks of Microchip Technology Incorporated in the
U.S.A. and other countries.
The Embedded Control Solutions Company and mTouch are
registered trademarks of Microchip Technology Incorporated
in the U.S.A.
Analog-for-the-Digital Age, BodyCom, chipKIT, chipKIT logo,
CodeGuard, dsPICDEM, dsPICDEM.net, ECAN, In-Circuit
Serial Programming, ICSP, Inter-Chip Connectivity, KleerNet,
KleerNet logo, MiWi, motorBench, MPASM, MPF, MPLAB
Certified logo, MPLIB, MPLINK, MultiTRAK, NetDetach,
Omniscient Code Generation, PICDEM, PICDEM.net, PICkit,
PICtail, RightTouch logo, REAL ICE, SQI, Serial Quad I/O,
Total Endurance, TSHARC, USBCheck, VariSense,
ViewSpan, WiperLock, Wireless DNA, and ZENA are
trademarks of Microchip Technology Incorporated in the
U.S.A. and other countries.
SQTP is a service mark of Microchip Technology Incorporated
in the U.S.A.
Silicon Storage Technology is a registered trademark of
Microchip Technology Inc. in other countries.
GestIC is a registered trademark of Microchip Technology
Germany II GmbH & Co. KG, a subsidiary of Microchip
Technology Inc., in other countries.
All other trademarks mentioned herein are property of their
respective companies.
© 2015, Microchip Technology Incorporated, Printed in the
U.S.A., All Rights Reserved.
ISBN: 978-1-5224-0039-4
QUALITY MANAGEMENT SYSTEM
CERTIFIED BY DNV
== ISO/TS 16949 ==
2015 Microchip Technology Inc.
Microchip received ISO/TS-16949:2009 certification for its worldwide
headquarters, design and wafer fabrication facilities in Chandler and
Tempe, Arizona; Gresham, Oregon and design centers in California
and India. The Company’s quality system processes and procedures
are for its PIC® MCUs and dsPIC® DSCs, KEELOQ® code hopping
devices, Serial EEPROMs, microperipherals, nonvolatile memory and
analog products. In addition, Microchip’s quality system for the design
and manufacture of development systems is ISO 9001:2000 certified.
DS20005459B-page 33
Worldwide Sales and Service
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07/14/15
DS20005459B-page 34
2015 Microchip Technology Inc.
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