MIC2168BMM

MIC2168 1 MHz PWM Synchronous Buck Control IC Features General Description • • • • • The MIC2168 is a high-efficiency, simple-to-use 1 MHz PWM synchronous buck control IC housed in a small MSOP-10 package. The MIC2168 allows compact DC/DC solutions with a minimal external component count and cost. • • • • • • • • • 3V to 14.5V Input Voltage Range Adjustable Output Voltage Down to 0.8V Up to 95% Efficiency 1 MHz PWM Operation Adjustable Current Limit Senses High-Side N-Channel MOSFET Current No External Current Sense Resistor Adaptive Gate Drive Increases Efficiency Ultra-Fast Response with Hysteretic Transient Recovery Mode Overvoltage Protection Protects the Load in Fault Conditions Dual-Mode Current Limit Speeds up Recovery Time Hiccup Mode Short-Circuit Protection Internal Soft-Start Dual Function COMP and EN Pin Allows Low-Power Shutdown Small Size MSOP 10-Lead Package Applications • • • • • • • Point-of-Load DC/DC Conversion Set-Top Boxes Graphics Cards LCD Power Supplies Telecom Power Supplies Networking Power Supplies Cable Modems and Routers  2020 Microchip Technology Inc. The MIC2168 operates from a 3V to 14.5V input, without the need of any additional bias voltage. The output voltage can be precisely regulated down to 0.8V. The adaptive all N-Channel MOSFET drive scheme allows efficiencies over 95% across a wide load range. The MIC2168 senses current across the high-side N-Channel MOSFET, eliminating the need for an expensive and lossy current-sense resistor. Current limit accuracy is maintained by a positive temperature coefficient that tracks the increasing RDS(ON) of the external MOSFET. Further cost and space are saved by the internal in-rush-current limiting and digital soft-start. The MIC2168 is available in a 10-lead MSOP package, with a wide junction operating range of –40°C to +125°C. Package Type MIC2168 10-Lead MSOP (MM) (Top View) VIN 1 10 BST VDD 2 9 HSD CS 3 8 VSW COMP/EN 4 7 LSD FB 5 6 GND DS20006145A-page 1 MIC2168 Typical Application Circuit MIC2168 Adjustable Output 1 MHz Converter VIN = 5V SD103BWS 100μF 4.7μF 0.1μF VDD BST CS Ÿ VIN NŸ IRF7821 HSD 1000pF MIC2168 VSW 1.2μH Ÿ COMP/EN 100pF IRF7821 LSD 100nF 3.3V NŸ Ÿ 1000nF NŸ 150μF x 2 FB GND NŸ Functional Block Diagram CIN RCS VIN CS VDD D1 Current Limit Comparator VDD 5V 5V LDO High-Side Driver HSD Q1 5V Bandgap Reference BOOST Current Limit Reference 0.8V BG Valid SW Clamp & Startup Current Ramp Clock L1 Driver Logic 5V Soft-Start & Digital Delay Counter CBST 4W RSW Ÿ VOUT COUT 1000pF 5V Low-Side Driver LSD Q2 PWM Comparator Enable Error Loop 0.8V VREF +3% VREF 3% Error Amp FB Hys Comparator R3 R2 MIC2168 COMP GND C1 C2 R1 DS20006145A-page 2  2020 Microchip Technology Inc. MIC2168 1.0 ELECTRICAL CHARACTERISTICS Absolute Maximum Ratings † Supply Voltage (VIN) ............................................................................................................................................. +15.5V Bootstrapped Voltage (VBST) .............................................................................................................................. VIN + 5V Operating Ratings †† Supply Voltage (VIN) .................................................................................................................................. +3V to +14.5V Output Voltage Range.........................................................................................................................0.8V to VIN x DMAX † Notice: Absolute maximum ratings indicate limits beyond which damage to the component may occur. Electrical specifications do not apply when operating the device outside of its operating ratings. The maximum allowable power dissipation is a function of the maximum junction temperature, TJ(MAX), the junction-to-ambient thermal resistance, θJA, and the ambient temperature, TA. The maximum allowable power dissipation will result in excessive die temperature, and the regulator will go into thermal shutdown. †† Notice: Devices are ESD sensitive, handling precautions required. ELECTRICAL CHARACTERISTICS Electrical Characteristics: TJ = +25°C, VIN = 5V, unless otherwise specified. Bold values indicate –40°C < TJ < +125°C Note 1 Parameter Sym. Min. Typ. Max. 0.792 0.8 0.808 0.784 0.8 0.816 — 30 100 nA — Output Voltage Line Regulation — 0.03 — %/V — Output Voltage Load Regulation — 0.5 — % — Output Voltage Total Regulation — 0.6 — % 3V ≤ VIN ≤ 14.5V; 1A ≤ IOUT ≤ 10A; (VOUT = 2.5V), Note 2 Feedback Voltage Reference VFB Feedback Bias Current IBIAS Units V Conditions ±1% ±2% over temperature Oscillator Selection Oscillator Frequency fO 900 1000 1100 kHz — Maximum Duty Cycle DMAX — — 90 % — tON(MIN) — 30 60 ns Note 2 — 1.6 3 mA VCS = VIN – 0.25V; VFB = 0.7V (output switching but excluding external MOSFET gate current) — 50 150 μA VCOMP/EN = 0V VCOMP Shutdown Threshold 0.1 0.25 0.4 V — VCOMP Shutdown Blanking Period — 4 4.7 5 Minimum On-Time Input and VDD Supply PWM Mode Supply Current Shutdown Quiescent Current ISHDN Digital Supply Voltage Note 1: 2: VDD ms 5.3 V CCOMP = 100 nF VIN ≥ 6V Specification for packaged product only. Guaranteed by design.  2020 Microchip Technology Inc. DS20006145A-page 3 MIC2168 ELECTRICAL CHARACTERISTICS (CONTINUED) Electrical Characteristics: TJ = +25°C, VIN = 5V, unless otherwise specified. Bold values indicate –40°C < TJ < +125°C Note 1 Parameter Sym. Min. Typ. Max. Units Conditions DC Gain — 70 — dB — Transconductance — 1 — ms — — 8.5 — μA After timeout of internal timer. See Soft-Start section. 160 200 240 μA VCS = VIN – 0.25V Error Amplifier Soft-Start Soft-Start Current ISS Current Sense CS Overcurrent Trip Point Temperature Coefficient ppm/°C — 1800 Output Fault Correction Thresholds Upper Threshold VFB_OVT — +3 — % Relative to VFB Lower Threshold VFB_UVT — –3 — % Relative to VFB tr/tf — 30 — ns Into 3000 pF at VIN > 5V — — 6 — — 6 — — 10 — — 10 10 20 — Gate Drivers Rise/Fall Time Output Driver Impedance Driver Non-Overlap Time Note 1: 2: Source, VIN = 5V Ω Sink, VIN = 5V Source, VIN = 3V Sink, VIN = 3V ns Note 2 Specification for packaged product only. Guaranteed by design. DS20006145A-page 4  2020 Microchip Technology Inc. MIC2168 TEMPERATURE SPECIFICATIONS Parameters Sym. Min. Typ. Max. Units Conditions Maximum Junction Temperature Range TJ –40 — +125 °C — Storage Temperature Range TS –65 — +150 °C — JA — 180 — °C/W — Temperature Ranges Package Thermal Resistances Thermal Resistance, MSOP 10-Ld Note 1: The maximum allowable power dissipation is a function of ambient temperature, the maximum allowable junction temperature and the thermal resistance from junction to air (i.e., TA, TJ, JA). Exceeding the maximum allowable power dissipation will cause the device operating junction temperature to exceed the maximum +125°C rating. Sustained junction temperatures above +125°C can impact the device reliability.  2020 Microchip Technology Inc. DS20006145A-page 5 MIC2168 2.0 TYPICAL PERFORMANCE CURVES Note: The graphs and tables provided following this note are a statistical summary based on a limited number of samples and are provided for informational purposes only. The performance characteristics listed herein are not tested or guaranteed. In some graphs or tables, the data presented may be outside the specified operating range (e.g., outside specified power supply range) and therefore outside the warranted range. 2.9 2.7 2.5 2.3 2.1 1.9 1.7 1.5 1.3 1.1 0.9 0.7 0.5 -40 -20 0 20 40 60 80 100120140 TEMPERATURE (qC) FIGURE 2-1: vs. Temperature. PWM Mode Supply Current 0.820 0.815 0.810 0.805 VFB (V) IDD (mA) Electrical Characteristics: VIN = 5V. 0.800 0.795 0.790 0.785 0.780 -60 -30 0 30 60 90 120 150 TEMPERATURE (qC) FIGURE 2-4: 6 5 1.5 4 VDD (V) QUIESCENT CURRENT (mA) 2.0 3 2 1.0 1 0.5 0 0 5 10 SUPPLY VOLTAGE (V) 15 FIGURE 2-5: VDD REGULATOR VOLTAGE (V) 0.820 0.815 0.810 0.805 0.800 0.795 0.790 0.785 0.780 0 5 10 DS20006145A-page 6 5 10 15 15 VFB Line Regulation. VDD Line Regulation. 5.01 4.99 4.97 4.95 4.93 4.91 4.89 4.87 4.85 VIN (V) FIGURE 2-3: 0 VIN (V) FIGURE 2-2: PWM Mode Supply Current vs. Supply Voltage. VFB (V) VFB vs. Temperature. FIGURE 2-6: 0 5 10 15 20 25 LOAD CURRENT (mA) 30 VDD Load Regulation.  2020 Microchip Technology Inc. MIC2168 4 4.5 4.0 3 3.5 VOUT (V) VDD LINE REGULATION (%) 5.0 3.0 2.5 2.0 1.5 1 Top MOSFET = Si4800 1.0 RCS = 1k: 0.5 0 0.0 -60 -30 0 30 60 90 120 150 TEMPERATURE (qC) FIGURE 2-7: Temperature. VDD Line Regulation vs. 0 FIGURE 2-10: 1200 240 1150 220 1100 200 1050 ICS (PA) FREQUENCY (kHz) 2 1000 950 2 4 6 ILOAD (A) 8 10 Current Limit Foldback. 180 160 140 900 850 120 800 -60 -30 0 30 60 90 120 150 TEMPERATURE (qC) 100 -60 -30 0 30 60 90 120 150 TEMPERATURE (qC) FIGURE 2-8: Temperature. Oscillator Frequency vs. FIGURE 2-11: Temperature. Overcurrent Trip Point vs. FREQUENCY VARIATION (%) 1.5 1.0 0.5 0 -0.5 -1.0 -1.5 0 FIGURE 2-9: Supply Voltage. 5 10 SUPPLY VOLTAGE (V) 15 Oscillator Frequency vs.  2020 Microchip Technology Inc. DS20006145A-page 7 MIC2168 3.0 PIN DESCRIPTIONS The descriptions of the pins are listed in Table 3-1. TABLE 3-1: PIN FUNCTION TABLE Pin Number Pin Name 1 VIN Supply Voltage (Input): 3V to 14.5V. 2 VDD 5V Internal Linear Regulator (Output): VDD is the external MOSFET gate drive supply voltage and an internal supply bus for the IC. When VIN is < 5V, this regulator operates in dropout mode. 3 CS 4 COMP/EN Compensation (Input): Dual function pin. Pin for external compensation. If this pin is pulled below 0.2V, with the reference fully up the device shuts down (50 μA typical current draw). 5 FB Feedback (Input): Input to error amplifier. Regulates error amplifier to 0.8V. 6 GND Ground (Return). 7 LSD Low-Side Drive (Output): High-current driver output for external synchronous MOSFET. 8 VSW Switch (Return): High-side MOSFET driver return. 9 HSD High-Side Drive (Output): High-current output-driver for the high-side MOSFET. When VIN is between 3.0V to 5V, 2.5V threshold-rated MOSFETs should be used. At VIN > 5V, 5V threshold MOSFETs should be used. 10 BST Boost (Input): Provides the drive voltage for the high-side MOSFET driver. The gate-drive voltage is higher than the source voltage by VIN minus a diode drop. DS20006145A-page 8 Description Current Sense/Enable (Input): Current limit comparator noninverting input. The current limit is sensed across the MOSFET during the ON time. The current can be set by the resistor in series with the CS pin.  2020 Microchip Technology Inc. MIC2168 4.0 FUNCTIONAL DESCRIPTION The MIC2168 is a voltage mode, synchronous step-down switching regulator controller designed for high output power without the use of an external sense resistor. It includes an internal soft-start function that reduces the power supply input surge current at start-up by controlling the output voltage rise time, a PWM generator, a reference voltage, two MOSFET drivers, and short-circuit current limiting circuitry to form a complete 1 MHz switching regulator. 4.1 Theory of Operation The MIC2168 is a voltage mode step-down regulator. The Functional Block Diagram illustrates the block diagram for the voltage control loop. The output voltage variation due to load or line changes will be sensed by the inverting input of the transconductance error amplifier via the feedback resistors R3, and R2 and compared to a reference voltage at the non-inverting input. This will cause a small change in the DC voltage level at the output of the error amplifier which is the input to the PWM comparator. The other input to the comparator is a 0V to 1V triangular waveform. The comparator generates a rectangular waveform whose width tON is equal to the time from the start of the clock cycle t0 until t1, the time the triangle crosses the output waveform of the error amplifier. To illustrate the control loop, let us assume the output voltage drops due to sudden load turn-on, this would cause the inverting input of the error amplifier which is divided down version of VOUT to be slightly less than the reference voltage causing the output voltage of the error amplifier to go high. This will cause the PWM comparator to increase tON time of the top side MOSFET, causing the output voltage to go up and bringing VOUT back in regulation. 4.2 Soft-Start The COMP/EN pin is used for three functions: 1. 2. 3. Disables the part by grounding this pin. External compensation to stabilize the voltage control loop. Soft-Start. For better understanding of the soft-start feature, let’s assume VIN = 12V and the MIC2168 is allowed to power up by ungrounding the COMP/EN pin. The COMP pin has an internal 8.5 μA current source that charges the external compensation capacitor. As soon as this voltage rises to 180 mV (t = Cap_COMP x 0.18V/8.5 μA), the MIC2168 allows the internal VDD linear regulator to power up. As soon as it crosses the undervoltage lockout of 2.6V, the chip’s internal oscillator starts switching. At this point in time, the COMP pin current source increases to 40 μA and an internal 11-bit counter starts counting; this takes approximately 2 ms to complete. During counting, the  2020 Microchip Technology Inc. COMP voltage is clamped at 0.65V. After this counting cycle, the COMP current source is reduced to 8.5 μA and the COMP pin voltage rises from 0.65V to 0.95V, the bottom edge of the saw-tooth oscillator. This is the beginning of 0% duty cycle and it increases slowly, causing the output voltage to slowly rise. The MIC2168 has two hysteretic comparators that are enabled when VOUT is with ±3% of steady state. When the output voltage reaches 97% of the programmed output voltage, the gm error amplifier is enabled along with the hysteretic comparator. From this point onward, the voltage control loop (gm error amplifier) is fully in control and will regulate the output voltage. Soft-start time can be calculated approximately by adding the following four time frames: EQUATION 4-1: t1 = Cap_COMP  0.18V  8.5A t2 = 12 bit counter, appx. 2ms t3 = Cap_COMP  0.3V  8.5A V OUT Cap_COMP t4 = --------------  0.5  ------------------------------8.5A V IN Soft-Start Time  Cap_COMP = 100nF  = t1 + t2 + t3 + t4 2.1ms + 2ms + 3.5ms + 1.8ms = 10ms 4.3 Current Limit The MIC2168 uses the RDS(ON) of the top power MOSFET to measure the output current. Because it uses the drain to source the resistance of the power MOSFET, it is not very accurate. This scheme is adequate to protect the power supply and external components during a fault condition by cutting back the time the top MOSFET is on if the feedback voltage is greater than 0.67V. In case of a hard short when the feedback voltage is less than 0.67V, the MIC2168 discharges the COMP capacitor to 0.65V, resets the digital counter, automatically shuts off the top gate drive, and the gm error amplifier and the –3% hysteretic comparators are complete disabled. Then the soft-start cycle restarts. This mode of operation is called the hiccup mode and its purpose is to protect the downstream load in case of a hard shirt. the circuit in Figure 4-1 illustrates the MIC2168 current limiting circuit. DS20006145A-page 9 MIC2168 4.5 VIN C2 CIN HSD 0 Q1 MOSFET N VOUT Ÿ L1 Inductor 1000pF RCS CS LSD Q2 MOSFET N C1 COUT 200μA FIGURE 4-1: Circuit. MIC2168 Current Limiting The current limiting resistor RCS is calculated using the following equation: EQUATION 4-2: R DS  ON Q1  I L R CS = --------------------------------------200A 1 I L = I LOAD = --------------------------------------------------------------------2  Inductor Ripple Current  V IN – V OUT Inductor Ripple Current = V OUT = -----------------------------------V IN  f SW  L Where: fSW = 1 MHz 200 μA is the internal sink current to program the MIC2168 current limit. The MOSFET RDS(ON) varies 30% to 40% with temperature. Therefore, it is recommended to add a 50% margin to the load current (ILOAD) in Equation 4-2 to avoid false current limiting due to increased MOSFET junction temperature rise. It is also recommended to connect RCS resistor directly to the drain of the top MOSFET Q1, and the RSW resistor to the source of Q1 to accurately sense the MOSFETs RDS(ON). A 0.1 μF capacitor in parallel with RCS should be connected to filter some of the switching noise. 4.4 Internal VDD Supply The MIC2168 controller internally generates VDD for self-biasing and to provide power to the gate drives. This VDD supply is generated through a low-dropout regulator and generates 5V from VIN supply greater than 5V. For supply voltage less than 5V, the VDD linear regulator is approximately 200 mV in dropout. Therefore, it is recommended to short the VDD supply to the input supply through a 10Ω resistor for input supplies between 2.9V to 5V. DS20006145A-page 10 MOSFET Gate Drive The MIC2168 high-side drive circuit is designed to switch an N-Channel MOSFET. The Functional Block Diagram shows a bootstrap circuit, consisting of D2 and CBST, supplies energy to the high-side drive circuit. Capacitor CBST is charged while the low-side MOSFET is on and the voltage on the VSW pin is approximately 0V. When the high-side MOSFET driver is turned on, energy from CBST is used to turn the MOSFET on. As the MOSFET turns on, the voltage on the VSW pin increases to approximately VIN. Diode D2 is reversed biased and CBST floats high while continuing to keep the high-side MOSFET on. When the low-side switch is turned back on, CBST is recharged through D2. The drive voltage is derived from the internal 5V VDD bias supply. The nominal low-side gate drive voltage is 5V and the nominal high-side gate drive voltage is approximately 4.5V due the voltage drop across D2. An approximate 20 ns delay between the high- and low-side driver transitions is used to prevent current from simultaneously flowing unimpeded through both MOSFETs. 4.6 MOSFET Selection The MIC2168 controller works from input voltages of 3V to 13.2V and has an internal 5V regulator to provide power to turn the external N-Channel power MOSFETs for high- and low-side switches. For applications where VIN < 5V, the internal VDD regulator operates in dropout mode, and it is necessary that the power MOSFETs used are low-threshold and are in full conduction mode for VGS of 2.5V. For applications when VIN > 5V; logic-level MOSFETs, whose operation is specified at VGS = 4.5V, must be used. It is important to note the ON-resistance of a MOSFET increases with increasing temperature. A 75°C rise in junction temperature will increase the channel resistance of the MOSFET by 50% to 75% of the resistance specified at 25°C. This change in resistance must be accounted for when calculating MOSFET power dissipation and in calculating the value of current-sense (CS) resistor. Total gate charge is the charge required to turn the MOSFET on and off under specified operating conditions (VDS and VGS). The gate charge is supplied by the MIC2168 gate drive circuit. At 1 MHz switching frequency and above, the gate charge can be a significant source of power dissipation in the MIC2168. At low output load, this power dissipation is noticeable as a reduction in efficiency. The average current required to drive the high-side MOSFET is:  2020 Microchip Technology Inc. MIC2168 EQUATION 4-3: EQUATION 4-6: P SW = P CONDUCTION + P AC I G  HIGH -SIDEAVG  = Q G  f S Where: IG(HIGH-SIDEAVG) = Average high-side MOSFET gate current. QG = Total gate charge for the high-side MOSFET taken from the manufacturer’s data sheet for VGS = 5V. The low-side MOSFET is turned on and off at VDS = 0V because the freewheeling diode is conducting during this time. The switching loss for the low-side MOSFET is usually negligible. Also, the gate-drive current for the low-side MOSFET is more accurately calculated using CISS at VDS = 0V instead of gate charge. Where: PCONDUCTION = ISW(RMS)2 x RSW PAC = PAC(OFF) + PAC(ON) RSW = ON-Resistance of the MOSFET switch. D = Duty cycle (VOUT/VIN) Making the assumption that the turn-on and turn-off transition times are equal, the transition times can be approximated by: EQUATION 4-7: C ISS  V GS + C OSS  V IN t T = ---------------------------------------------------------------IG For the low-side MOSFET: EQUATION 4-4: I G  LOW -SIDEAVG  = C ISS  V GS  f S Where: CISS and COSS are measured at VDS = 0V. IG = Gate-drive current (1A for the MIC2168) Total high-side MOSFET switching loss is: Because the current from the gate drive comes from the input voltage, the power dissipated in the MIC2168 due to gate drive is: EQUATION 4-5: P GATEDRIVE = V IN  I G  HIGH -SIDEAVG  + I G  LOW -SIDEAVG   A convenient figure of merit for switching MOSFETs is the ON-resistance times the total gate charge (RDS(ON) × QG). Lower numbers translate into higher efficiency. Low gate-charge logic-level MOSFETs are a good choice for use with the MIC2168. Parameters that are important to MOSFET switch selection are: • Voltage Rating • ON-Resistance • Total Gate Charge The voltage ratings for the top and bottom MOSFET are essentially equal to the input voltage. A safety factor of 20% should be added to the VDS(MAX) of the MOSFETs to account for voltage spikes due to circuit parasitics. The power dissipated in the switching transistor is the sum of the conduction losses during the on-time (PCONDUCTION) and the switching losses that occur during the period of time when the MOSFETs turn on and off (PAC).  2020 Microchip Technology Inc. EQUATION 4-8: P AC =  V IN + V D   I PK  t T  f S Where: tT = Switching transition time (typ. 20 ns to 50 ns) VD = Freewheeling diode drop (typ. 0.5V) fS = Switching frequency (nom. 1 MHz) The low-side MOSFET switching losses are negligible and can be ignored for these calculations. 4.7 Inductor Selection Values for inductance, peak, and RMS currents are required to select the output inductor. The input and output voltages and the inductance value determine the peak-to-peak inductor ripple current. Generally, higher inductance values are used with higher input voltages. Larger peak-to-peak ripple currents will increase the power dissipation in the inductor and MOSFETs. Larger output ripple currents will also require more output capacitance to smooth out the larger ripple current. Smaller peak-to-peak ripple currents require a larger inductance value and therefore a larger and more expensive inductor. A good compromise between size, loss, and cost is to set the inductor ripple current to be equal to 20% of the maximum output current. The inductance value is calculated by the equation below. DS20006145A-page 11 MIC2168 EQUATION 4-9: EQUATION 4-13: V OUT   V IN  MAX  – V OUT  L = ------------------------------------------------------------------------------------V IN  MAX   f S  0.2  I OUT  MAX  Where: fS = Switching frequency, 1 MHz 0.2 = The ratio of AC ripple current to DC output current. VIN(MAX) = Maximum input voltage. The peak-to-peak inductor current (AC ripple current) is: EQUATION 4-10: V OUT   V IN  MAX  – V OUT  I PP = ----------------------------------------------------------------------V IN  MAX   f S  L The peak inductor current is equal to the average output current plus one half of the peak-to-peak inductor ripple current. EQUATION 4-11: 2 IP 1 I INDUCTOR  RMS  = I OUT  MAX   1 + ---  --------------------------- 3  I OUT  MAX  Maximizing efficiency requires the proper selection of core material and minimizing the winding resistance. The high frequency operation of the MIC2168 requires the use of ferrite materials for all but the most cost sensitive applications. Lower cost iron powder cores may be used but the increase in core loss will reduce the efficiency of the power supply. This is especially noticeable at low output power. The winding resistance decreases efficiency at the higher output current levels. The winding resistance must be minimized although this usually comes at the expense of a larger inductor. The power dissipated in the inductor is equal to the sum of the core and copper losses. At higher output loads, the core losses are usually insignificant and can be ignored. At lower output currents, the core losses can be a significant contributor. Core loss information is usually available from the magnetics vendor. Copper loss in the inductor is calculated by the equation below: EQUATION 4-12: 2 P INDUCTOR CU = I INDUCTOR  RMS   R WINDING R WINDING  HOT  = R WINDING  20C    1 + 0.0042   T HOT – T 20C   Where: THOT = Temperature of the wire under operating load T20°C = Ambient temperature. RWINDING(20°C) = Room temperature winding resistance (usually specified by manufacturer). 4.8 Output Capacitor Selection The output capacitor values are usually determined by the capacitor’s ESR (equivalent series resistance). Voltage and RMS current capability are two other important factors selecting the output capacitor. Recommended capacitors tantalum, low-ESR aluminum electrolytics, and POSCAPS. The output capacitor’s ESR is usually the main cause of output ripple. The output capacitor ESR also affects the overall voltage feedback loop from stability point of view. See “Feedback Loop Compensation” section for more information. The maximum value of ESR is calculated: EQUATION 4-14: V OUT R ESR  ----------------I PP Where: VOUT = Peak-to-peak output voltage ripple. IPP = Peak-to-peak inductor ripple current. The total output ripple is a combination of the ESR output capacitance. The total ripple is calculated below. EQUATION 4-15: V OUT = I PP   1 – D  2 2  -------------------------------- +  I PP  R ESR   C  f  OUT S Where: D = Duty cycle. COUT = Output capacitance. fS = Switching frequency. The voltage rating of capacitor should be twice the voltage for a tantalum and 20% greater for an aluminum electrolytic. The output capacitor RMS current is calculated below. EQUATION 4-16: The resistance of the copper wire, RWINDING, increases with temperature. The value of the winding resistance used should be at the operating temperature. DS20006145A-page 12 I PP I COUT  RMS  = --------12  2020 Microchip Technology Inc. MIC2168 The power dissipated in the output capacitor is: 4.10 Voltage Setting Components EQUATION 4-17: The MIC2168 requires two resistors to set the output voltage as shown in Figure 4-2. 2 P DISS  COUT  = I COUT  RMS   R ESR  COUT  R1 4.9 Error Amp Input Capacitor Selection The input capacitor should be selected for ripple current rating and voltage rating. Tantalum input capacitors may fail when subjected to high inrush currents, caused by turning the input supply on. Tantalum input capacitor voltage rating should be at least 2 times the maximum input voltage to maximize reliability. Aluminum electrolytic, OS-CON, and multilayer polymer film capacitors can handle the higher inrush currents without voltage derating. The input voltage ripple will primarily depend on the input capacitor’s ESR. The peak input current is equal to the peak inductor current, so: FB 7 R2 VREF 0.8V MIC2168 [adj.] FIGURE 4-2: Configuration. Voltage-Divider In this figure, VREF for MIC2168 is typically 0.8V. The output voltage is determined by Equation 4-21. EQUATION 4-21: EQUATION 4-18: R1 V OUT = V REF   1 + -------  R2 V IN = I INDUCTOR  PEAK   R ESR  CIN  The input capacitor must be rated for the input current ripple. The RMS value of input capacitor current is determined at the maximum output current. Assuming the peak-to-peak inductor ripple current is low: EQUATION 4-19: I CIN  RMS   I OUT  MAX   D   1 – D  A typical value for R1 can be between 3 kΩ and 10 kΩ. If R1 is too large, it may allow noise to be introduced into the voltage feedback loop. If R1 is too small, it will decrease the efficiency of the power supply, especially at light loads. Once R1 is selected, R2 can be calculated. EQUATION 4-22: V REF  R1 R2 = ---------------------------------V OUT – V REF The power dissipated in the input capacitor is: EQUATION 4-20: 4.11 2 P DISS  CIN  = I CIN  RMS   R ESR  CIN  External Schottky Diode An external freewheeling diode is used to keep the inductor current flow continuous while both MOSFETs are turned off. This dead time prevents current from flowing unimpeded through both MOSFETs and is typically 15 ns. The diode conducts twice during each switching cycle. Although the average current through this diode is small, the diode must be able to handle the peak current. The reverse voltage requirement of the diode is: EQUATION 4-23: V DIODE  RRM  = V IN  2020 Microchip Technology Inc. DS20006145A-page 13 MIC2168 The power dissipated by the Schottky diode is: L DCR EQUATION 4-24: VOUT P DIODE = I D  AVG   V F ESR Where: COUT VF = Forward voltage at the peak diode current. 4.12 FIGURE 4-3: The Output LC Filter in a Voltage Mode Buck Converter. EQUATION 4-25: ( G(S) = Plotting this transfer function with the following assumed values (L = 2 μH, DCR = 0.009Ω, COUT = 1000 μF, ESR = 0.050Ω) gives much insight toward why one needs to compensate on the COMP pin. Figure 4-4 and Figure 4-5 show the gain curve and phase curve for the transfer function in Equation 4-25. 30 30 7.5 Feedback Loop Compensation The MIC2168 controller comes with an internal transconductance error amplifier used for compensating the voltage feedback loop by placing a capacitor (C1) in series with a resistor (R1) and another capacitor C2 in parallel from the COMP pin to ground. See the Functional Block Diagram. 15 37.5 60 60 100 100 3 1.10 FIGURE 4-4: The power stage of a voltage mode controller has an inductor, L1, with its winding resistance (DCR) connected to the output capacitor, COUT, with its electrical series resistance (ESR) as shown in Figure 4-3. The transfer function G(s), for such a system is: 0 5 4 1 .10 f 6 1 .10 1 .10 1000000 Gain Curve for G(S). Power Stage 0 50 PHASE 4.13 ) (1 + ESR × s × C) DCR × s × C + s2 × L × C + 1 + ESR × s × C GAIN The external Schottky diode, D1, is not necessary for circuit operation because the low-side MOSFET contains a parasitic body diode. The external diode will improve efficiency and decrease high frequency noise. If the MOSFET body diode is used, it must be rated to handle the peak and average current. The body diode has a relatively slow reverse recovery time and a relatively high forward voltage drop. The power lost in the diode is proportional to the forward voltage drop of the diode. As the high-side MOSFET starts to turn on, the body diode becomes a short circuit for the reverse recovery period, dissipating additional power. The diode recovery and the circuit inductance will cause ringing during the high-side MOSFET turn-on. An external Schottky diode conducts at a lower forward voltage preventing the body diode in the MOSFET from turning on. The lower forward voltage drop dissipates less power than the body diode. The lack of a reverse recovery mechanism in a Schottky diode causes less ringing and less power loss. Depending on the circuit components and operating conditions, an external Schottky diode will give a 1/2% to 1% improvement in efficiency. 100 150 180 100 100 FIGURE 4-5: DS20006145A-page 14 3 1.10 4 1 .10 f 5 1 .10 6 1 .10 1000000 Phase Curve for G(S).  2020 Microchip Technology Inc. MIC2168 It can be seen from the transfer function G(S) and the gain curve that the output inductor and capacitor create a two pole system with a break frequency at: 0 0 50 1 f C = ----------------------------------------2   L  C OUT Therefore, fLC = 3.6 kHz. By looking at the phase curve, it can be seen that the output capacitor ESR (0.050Ω) cancels one of the two poles (LCOUT) system by introducing a zero at: EQUATION 4-27: 1 f ZERO = --------------------------------------------------2    ESR  C OUT Therefore, fZERO = 6.36 kHz. From the point of view of compensating the voltage loop, it is recommended to use higher ESR output capacitors because they provide a 90° phase gain in the power path. For comparison purposes, Figure 4-6 shows the same phase curve with an ESR value of 0.002Ω. PHASE EQUATION 4-26: 100 150 180 100 100 FIGURE 4-6: 0.002Ω. 3 1.10 4 1 .10 f 5 1 .10 6 1 .10 1000000 Phase Curve with ESR = It can be seen from Figure 4-5 that at 50 kHz, the phase is approximately –90° versus Figure 4-6 where the number is –150°. This means that the transconductance error amplifier has to provide a phase boost of about 45° to achieve a closed loop phase margin of 45° at a crossover frequency of 50 kHz for Figure 4-4, versus 105° for Figure 4-6. The simple RC and C2 compensation scheme allows a maximum error amplifier phase boost of about 90°. Therefore, it is easier to stabilize the MIC2168 voltage control loop by using high ESR value output capacitors. 4.14 gm Error Amplifier It is undesirable to have high error amplifier gain at high frequencies because high frequency noise spikes would be picked up and transmitted at large amplitude to the output, thus, gain should be permitted to fall off at high frequencies. At low frequency, it is desired to have high open-loop gain to attenuate the power line ripple. Thus, the error amplifier gain should be allowed to increase rapidly at low frequencies. The transfer function with R1, C1, and C2 for the internal gm error amplifier can be approximated by Equation 4-28. EQUATION 4-28: 1 + R1  S  C1 Error Amplifer (z) = g m  ---------------------------------------------------------------------------------------------------C1  C2  S- s   C1 + C2    1 + R1  ----------------------------- C1 + C2  The equation above can be simplified by assuming C2