MIC2168
1 MHz PWM Synchronous Buck Control IC
Features
General Description
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The MIC2168 is a high-efficiency, simple-to-use 1 MHz
PWM synchronous buck control IC housed in a small
MSOP-10 package. The MIC2168 allows compact
DC/DC solutions with a minimal external component
count and cost.
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3V to 14.5V Input Voltage Range
Adjustable Output Voltage Down to 0.8V
Up to 95% Efficiency
1 MHz PWM Operation
Adjustable Current Limit Senses High-Side
N-Channel MOSFET Current
No External Current Sense Resistor
Adaptive Gate Drive Increases Efficiency
Ultra-Fast Response with Hysteretic Transient
Recovery Mode
Overvoltage Protection Protects the Load in Fault
Conditions
Dual-Mode Current Limit Speeds up Recovery
Time
Hiccup Mode Short-Circuit Protection
Internal Soft-Start
Dual Function COMP and EN Pin Allows
Low-Power Shutdown
Small Size MSOP 10-Lead Package
Applications
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Point-of-Load DC/DC Conversion
Set-Top Boxes
Graphics Cards
LCD Power Supplies
Telecom Power Supplies
Networking Power Supplies
Cable Modems and Routers
2020 Microchip Technology Inc.
The MIC2168 operates from a 3V to 14.5V input,
without the need of any additional bias voltage. The
output voltage can be precisely regulated down to 0.8V.
The adaptive all N-Channel MOSFET drive scheme
allows efficiencies over 95% across a wide load range.
The MIC2168 senses current across the high-side
N-Channel MOSFET, eliminating the need for an
expensive and lossy current-sense resistor. Current
limit accuracy is maintained by a positive temperature
coefficient that tracks the increasing RDS(ON) of the
external MOSFET. Further cost and space are saved
by the internal in-rush-current limiting and digital
soft-start.
The MIC2168 is available in a 10-lead MSOP package,
with a wide junction operating range of –40°C to
+125°C.
Package Type
MIC2168
10-Lead MSOP (MM)
(Top View)
VIN 1
10 BST
VDD 2
9 HSD
CS 3
8 VSW
COMP/EN 4
7 LSD
FB 5
6 GND
DS20006145A-page 1
MIC2168
Typical Application Circuit
MIC2168 Adjustable Output 1 MHz Converter
VIN = 5V
SD103BWS
100μF
4.7μF
0.1μF
VDD
BST
CS
VIN
N
IRF7821
HSD
1000pF
MIC2168 VSW
1.2μH
COMP/EN
100pF
IRF7821
LSD
100nF
3.3V
N
1000nF
N
150μF x 2
FB
GND
N
Functional Block Diagram
CIN
RCS
VIN
CS
VDD
D1
Current Limit
Comparator
VDD
5V
5V LDO
High-Side
Driver
HSD
Q1
5V
Bandgap
Reference
BOOST
Current Limit
Reference
0.8V
BG Valid
SW
Clamp &
Startup
Current
Ramp
Clock
L1
Driver
Logic
5V
Soft-Start &
Digital Delay
Counter
CBST
4W
RSW
VOUT
COUT
1000pF
5V
Low-Side
Driver
LSD
Q2
PWM
Comparator
Enable
Error
Loop
0.8V
VREF +3%
VREF 3%
Error
Amp
FB
Hys
Comparator
R3
R2
MIC2168
COMP
GND
C1
C2
R1
DS20006145A-page 2
2020 Microchip Technology Inc.
MIC2168
1.0
ELECTRICAL CHARACTERISTICS
Absolute Maximum Ratings †
Supply Voltage (VIN) ............................................................................................................................................. +15.5V
Bootstrapped Voltage (VBST) .............................................................................................................................. VIN + 5V
Operating Ratings ††
Supply Voltage (VIN) .................................................................................................................................. +3V to +14.5V
Output Voltage Range.........................................................................................................................0.8V to VIN x DMAX
† Notice: Absolute maximum ratings indicate limits beyond which damage to the component may occur. Electrical
specifications do not apply when operating the device outside of its operating ratings. The maximum allowable power
dissipation is a function of the maximum junction temperature, TJ(MAX), the junction-to-ambient thermal resistance, θJA,
and the ambient temperature, TA. The maximum allowable power dissipation will result in excessive die temperature,
and the regulator will go into thermal shutdown.
†† Notice: Devices are ESD sensitive, handling precautions required.
ELECTRICAL CHARACTERISTICS
Electrical Characteristics: TJ = +25°C, VIN = 5V, unless otherwise specified.
Bold values indicate –40°C < TJ < +125°C Note 1
Parameter
Sym.
Min.
Typ.
Max.
0.792
0.8
0.808
0.784
0.8
0.816
—
30
100
nA
—
Output Voltage Line
Regulation
—
0.03
—
%/V
—
Output Voltage Load
Regulation
—
0.5
—
%
—
Output Voltage Total
Regulation
—
0.6
—
%
3V ≤ VIN ≤ 14.5V; 1A ≤ IOUT ≤ 10A;
(VOUT = 2.5V), Note 2
Feedback Voltage
Reference
VFB
Feedback Bias Current
IBIAS
Units
V
Conditions
±1%
±2% over temperature
Oscillator Selection
Oscillator Frequency
fO
900
1000
1100
kHz
—
Maximum Duty Cycle
DMAX
—
—
90
%
—
tON(MIN)
—
30
60
ns
Note 2
—
1.6
3
mA
VCS = VIN – 0.25V; VFB = 0.7V (output
switching but excluding external
MOSFET gate current)
—
50
150
μA
VCOMP/EN = 0V
VCOMP Shutdown
Threshold
0.1
0.25
0.4
V
—
VCOMP Shutdown
Blanking Period
—
4
4.7
5
Minimum On-Time
Input and VDD Supply
PWM Mode Supply
Current
Shutdown Quiescent
Current
ISHDN
Digital Supply Voltage
Note 1:
2:
VDD
ms
5.3
V
CCOMP = 100 nF
VIN ≥ 6V
Specification for packaged product only.
Guaranteed by design.
2020 Microchip Technology Inc.
DS20006145A-page 3
MIC2168
ELECTRICAL CHARACTERISTICS (CONTINUED)
Electrical Characteristics: TJ = +25°C, VIN = 5V, unless otherwise specified.
Bold values indicate –40°C < TJ < +125°C Note 1
Parameter
Sym.
Min.
Typ.
Max.
Units
Conditions
DC Gain
—
70
—
dB
—
Transconductance
—
1
—
ms
—
—
8.5
—
μA
After timeout of internal timer. See
Soft-Start section.
160
200
240
μA
VCS = VIN – 0.25V
Error Amplifier
Soft-Start
Soft-Start Current
ISS
Current Sense
CS Overcurrent Trip Point
Temperature Coefficient
ppm/°C —
1800
Output Fault Correction Thresholds
Upper Threshold
VFB_OVT
—
+3
—
%
Relative to VFB
Lower Threshold
VFB_UVT
—
–3
—
%
Relative to VFB
tr/tf
—
30
—
ns
Into 3000 pF at VIN > 5V
—
—
6
—
—
6
—
—
10
—
—
10
10
20
—
Gate Drivers
Rise/Fall Time
Output Driver Impedance
Driver Non-Overlap Time
Note 1:
2:
Source, VIN = 5V
Ω
Sink, VIN = 5V
Source, VIN = 3V
Sink, VIN = 3V
ns
Note 2
Specification for packaged product only.
Guaranteed by design.
DS20006145A-page 4
2020 Microchip Technology Inc.
MIC2168
TEMPERATURE SPECIFICATIONS
Parameters
Sym.
Min.
Typ.
Max.
Units
Conditions
Maximum Junction Temperature Range
TJ
–40
—
+125
°C
—
Storage Temperature Range
TS
–65
—
+150
°C
—
JA
—
180
—
°C/W
—
Temperature Ranges
Package Thermal Resistances
Thermal Resistance, MSOP 10-Ld
Note 1:
The maximum allowable power dissipation is a function of ambient temperature, the maximum allowable
junction temperature and the thermal resistance from junction to air (i.e., TA, TJ, JA). Exceeding the
maximum allowable power dissipation will cause the device operating junction temperature to exceed the
maximum +125°C rating. Sustained junction temperatures above +125°C can impact the device reliability.
2020 Microchip Technology Inc.
DS20006145A-page 5
MIC2168
2.0
TYPICAL PERFORMANCE CURVES
Note:
The graphs and tables provided following this note are a statistical summary based on a limited number of
samples and are provided for informational purposes only. The performance characteristics listed herein
are not tested or guaranteed. In some graphs or tables, the data presented may be outside the specified
operating range (e.g., outside specified power supply range) and therefore outside the warranted range.
2.9
2.7
2.5
2.3
2.1
1.9
1.7
1.5
1.3
1.1
0.9
0.7
0.5
-40 -20 0 20 40 60 80 100120140
TEMPERATURE (qC)
FIGURE 2-1:
vs. Temperature.
PWM Mode Supply Current
0.820
0.815
0.810
0.805
VFB (V)
IDD (mA)
Electrical Characteristics: VIN = 5V.
0.800
0.795
0.790
0.785
0.780
-60 -30 0 30 60 90 120 150
TEMPERATURE (qC)
FIGURE 2-4:
6
5
1.5
4
VDD (V)
QUIESCENT CURRENT (mA)
2.0
3
2
1.0
1
0.5
0
0
5
10
SUPPLY VOLTAGE (V)
15
FIGURE 2-5:
VDD REGULATOR VOLTAGE (V)
0.820
0.815
0.810
0.805
0.800
0.795
0.790
0.785
0.780
0
5
10
DS20006145A-page 6
5
10
15
15
VFB Line Regulation.
VDD Line Regulation.
5.01
4.99
4.97
4.95
4.93
4.91
4.89
4.87
4.85
VIN (V)
FIGURE 2-3:
0
VIN (V)
FIGURE 2-2:
PWM Mode Supply Current
vs. Supply Voltage.
VFB (V)
VFB vs. Temperature.
FIGURE 2-6:
0
5
10 15 20 25
LOAD CURRENT (mA)
30
VDD Load Regulation.
2020 Microchip Technology Inc.
MIC2168
4
4.5
4.0
3
3.5
VOUT (V)
VDD LINE REGULATION (%)
5.0
3.0
2.5
2.0
1.5
1
Top MOSFET = Si4800
1.0
RCS = 1k:
0.5
0
0.0
-60 -30 0 30 60 90 120 150
TEMPERATURE (qC)
FIGURE 2-7:
Temperature.
VDD Line Regulation vs.
0
FIGURE 2-10:
1200
240
1150
220
1100
200
1050
ICS (PA)
FREQUENCY (kHz)
2
1000
950
2
4
6
ILOAD (A)
8
10
Current Limit Foldback.
180
160
140
900
850
120
800
-60 -30 0 30 60 90 120 150
TEMPERATURE (qC)
100
-60 -30 0 30 60 90 120 150
TEMPERATURE (qC)
FIGURE 2-8:
Temperature.
Oscillator Frequency vs.
FIGURE 2-11:
Temperature.
Overcurrent Trip Point vs.
FREQUENCY VARIATION (%)
1.5
1.0
0.5
0
-0.5
-1.0
-1.5
0
FIGURE 2-9:
Supply Voltage.
5
10
SUPPLY VOLTAGE (V)
15
Oscillator Frequency vs.
2020 Microchip Technology Inc.
DS20006145A-page 7
MIC2168
3.0
PIN DESCRIPTIONS
The descriptions of the pins are listed in Table 3-1.
TABLE 3-1:
PIN FUNCTION TABLE
Pin Number
Pin Name
1
VIN
Supply Voltage (Input): 3V to 14.5V.
2
VDD
5V Internal Linear Regulator (Output): VDD is the external MOSFET gate
drive supply voltage and an internal supply bus for the IC. When VIN is < 5V,
this regulator operates in dropout mode.
3
CS
4
COMP/EN
Compensation (Input): Dual function pin. Pin for external compensation.
If this pin is pulled below 0.2V, with the reference fully up the device shuts
down (50 μA typical current draw).
5
FB
Feedback (Input): Input to error amplifier. Regulates error amplifier to 0.8V.
6
GND
Ground (Return).
7
LSD
Low-Side Drive (Output): High-current driver output for external synchronous
MOSFET.
8
VSW
Switch (Return): High-side MOSFET driver return.
9
HSD
High-Side Drive (Output): High-current output-driver for the high-side
MOSFET. When VIN is between 3.0V to 5V, 2.5V threshold-rated MOSFETs
should be used. At VIN > 5V, 5V threshold MOSFETs should be used.
10
BST
Boost (Input): Provides the drive voltage for the high-side MOSFET driver.
The gate-drive voltage is higher than the source voltage by VIN minus a
diode drop.
DS20006145A-page 8
Description
Current Sense/Enable (Input): Current limit comparator noninverting input.
The current limit is sensed across the MOSFET during the ON time. The current
can be set by the resistor in series with the CS pin.
2020 Microchip Technology Inc.
MIC2168
4.0
FUNCTIONAL DESCRIPTION
The MIC2168 is a voltage mode, synchronous
step-down switching regulator controller designed for
high output power without the use of an external sense
resistor. It includes an internal soft-start function that
reduces the power supply input surge current at
start-up by controlling the output voltage rise time, a
PWM generator, a reference voltage, two MOSFET
drivers, and short-circuit current limiting circuitry to
form a complete 1 MHz switching regulator.
4.1
Theory of Operation
The MIC2168 is a voltage mode step-down regulator.
The Functional Block Diagram illustrates the block
diagram for the voltage control loop. The output voltage
variation due to load or line changes will be sensed by
the inverting input of the transconductance error
amplifier via the feedback resistors R3, and R2 and
compared to a reference voltage at the non-inverting
input. This will cause a small change in the DC voltage
level at the output of the error amplifier which is the
input to the PWM comparator. The other input to the
comparator is a 0V to 1V triangular waveform. The
comparator generates a rectangular waveform whose
width tON is equal to the time from the start of the clock
cycle t0 until t1, the time the triangle crosses the output
waveform of the error amplifier. To illustrate the control
loop, let us assume the output voltage drops due to
sudden load turn-on, this would cause the inverting
input of the error amplifier which is divided down
version of VOUT to be slightly less than the reference
voltage causing the output voltage of the error amplifier
to go high. This will cause the PWM comparator to
increase tON time of the top side MOSFET, causing the
output voltage to go up and bringing VOUT back in
regulation.
4.2
Soft-Start
The COMP/EN pin is used for three functions:
1.
2.
3.
Disables the part by grounding this pin.
External compensation to stabilize the voltage
control loop.
Soft-Start.
For better understanding of the soft-start feature, let’s
assume VIN = 12V and the MIC2168 is allowed to
power up by ungrounding the COMP/EN pin. The
COMP pin has an internal 8.5 μA current source that
charges the external compensation capacitor. As soon
as this voltage rises to 180 mV (t = Cap_COMP x
0.18V/8.5 μA), the MIC2168 allows the internal VDD
linear regulator to power up. As soon as it crosses the
undervoltage lockout of 2.6V, the chip’s internal
oscillator starts switching. At this point in time, the
COMP pin current source increases to 40 μA and an
internal 11-bit counter starts counting; this takes
approximately 2 ms to complete. During counting, the
2020 Microchip Technology Inc.
COMP voltage is clamped at 0.65V. After this counting
cycle, the COMP current source is reduced to 8.5 μA
and the COMP pin voltage rises from 0.65V to 0.95V,
the bottom edge of the saw-tooth oscillator. This is the
beginning of 0% duty cycle and it increases slowly,
causing the output voltage to slowly rise. The MIC2168
has two hysteretic comparators that are enabled when
VOUT is with ±3% of steady state. When the output
voltage reaches 97% of the programmed output
voltage, the gm error amplifier is enabled along with the
hysteretic comparator. From this point onward, the
voltage control loop (gm error amplifier) is fully in
control and will regulate the output voltage.
Soft-start time can be calculated approximately by
adding the following four time frames:
EQUATION 4-1:
t1 = Cap_COMP 0.18V 8.5A
t2 = 12 bit counter, appx. 2ms
t3 = Cap_COMP 0.3V 8.5A
V OUT
Cap_COMP
t4 = -------------- 0.5 ------------------------------8.5A
V IN
Soft-Start Time Cap_COMP = 100nF =
t1 + t2 + t3 + t4
2.1ms + 2ms + 3.5ms + 1.8ms = 10ms
4.3
Current Limit
The MIC2168 uses the RDS(ON) of the top power
MOSFET to measure the output current. Because it
uses the drain to source the resistance of the power
MOSFET, it is not very accurate. This scheme is
adequate to protect the power supply and external
components during a fault condition by cutting back the
time the top MOSFET is on if the feedback voltage is
greater than 0.67V. In case of a hard short when the
feedback voltage is less than 0.67V, the MIC2168
discharges the COMP capacitor to 0.65V, resets the
digital counter, automatically shuts off the top gate
drive, and the gm error amplifier and the –3% hysteretic
comparators are complete disabled. Then the soft-start
cycle restarts. This mode of operation is called the
hiccup mode and its purpose is to protect the
downstream load in case of a hard shirt. the circuit in
Figure 4-1 illustrates the MIC2168 current limiting
circuit.
DS20006145A-page 9
MIC2168
4.5
VIN
C2
CIN
HSD
0
Q1
MOSFET N
VOUT
L1 Inductor
1000pF
RCS
CS
LSD
Q2
MOSFET N
C1
COUT
200μA
FIGURE 4-1:
Circuit.
MIC2168 Current Limiting
The current limiting resistor RCS is calculated using the
following equation:
EQUATION 4-2:
R DS ON Q1 I L
R CS = --------------------------------------200A
1
I L = I LOAD = --------------------------------------------------------------------2 Inductor Ripple Current
V IN – V OUT
Inductor Ripple Current = V OUT = -----------------------------------V IN f SW L
Where:
fSW = 1 MHz
200 μA is the internal sink current to program the
MIC2168 current limit.
The MOSFET RDS(ON) varies 30% to 40% with
temperature. Therefore, it is recommended to add a
50% margin to the load current (ILOAD) in Equation 4-2
to avoid false current limiting due to increased
MOSFET junction temperature rise. It is also
recommended to connect RCS resistor directly to the
drain of the top MOSFET Q1, and the RSW resistor to
the source of Q1 to accurately sense the MOSFETs
RDS(ON). A 0.1 μF capacitor in parallel with RCS should
be connected to filter some of the switching noise.
4.4
Internal VDD Supply
The MIC2168 controller internally generates VDD for
self-biasing and to provide power to the gate drives.
This VDD supply is generated through a low-dropout
regulator and generates 5V from VIN supply greater
than 5V. For supply voltage less than 5V, the VDD linear
regulator is approximately 200 mV in dropout.
Therefore, it is recommended to short the VDD supply
to the input supply through a 10Ω resistor for input
supplies between 2.9V to 5V.
DS20006145A-page 10
MOSFET Gate Drive
The MIC2168 high-side drive circuit is designed to
switch an N-Channel MOSFET. The Functional Block
Diagram shows a bootstrap circuit, consisting of D2
and CBST, supplies energy to the high-side drive
circuit. Capacitor CBST is charged while the low-side
MOSFET is on and the voltage on the VSW pin is
approximately 0V. When the high-side MOSFET driver
is turned on, energy from CBST is used to turn the
MOSFET on. As the MOSFET turns on, the voltage on
the VSW pin increases to approximately VIN. Diode D2
is reversed biased and CBST floats high while
continuing to keep the high-side MOSFET on. When
the low-side switch is turned back on, CBST is
recharged through D2. The drive voltage is derived
from the internal 5V VDD bias supply. The nominal
low-side gate drive voltage is 5V and the nominal
high-side gate drive voltage is approximately 4.5V due
the voltage drop across D2. An approximate 20 ns
delay between the high- and low-side driver transitions
is used to prevent current from simultaneously flowing
unimpeded through both MOSFETs.
4.6
MOSFET Selection
The MIC2168 controller works from input voltages of
3V to 13.2V and has an internal 5V regulator to provide
power to turn the external N-Channel power MOSFETs
for high- and low-side switches. For applications where
VIN < 5V, the internal VDD regulator operates in dropout
mode, and it is necessary that the power MOSFETs
used are low-threshold and are in full conduction mode
for VGS of 2.5V. For applications when VIN > 5V;
logic-level MOSFETs, whose operation is specified at
VGS = 4.5V, must be used.
It is important to note the ON-resistance of a MOSFET
increases with increasing temperature. A 75°C rise in
junction temperature will increase the channel
resistance of the MOSFET by 50% to 75% of the
resistance specified at 25°C. This change in resistance
must be accounted for when calculating MOSFET
power dissipation and in calculating the value of
current-sense (CS) resistor. Total gate charge is the
charge required to turn the MOSFET on and off under
specified operating conditions (VDS and VGS). The gate
charge is supplied by the MIC2168 gate drive circuit. At
1 MHz switching frequency and above, the gate charge
can be a significant source of power dissipation in the
MIC2168. At low output load, this power dissipation is
noticeable as a reduction in efficiency. The average
current required to drive the high-side MOSFET is:
2020 Microchip Technology Inc.
MIC2168
EQUATION 4-3:
EQUATION 4-6:
P SW = P CONDUCTION + P AC
I G HIGH -SIDEAVG = Q G f S
Where:
IG(HIGH-SIDEAVG) = Average high-side MOSFET gate
current.
QG = Total gate charge for the high-side MOSFET
taken from the manufacturer’s data sheet for VGS =
5V.
The low-side MOSFET is turned on and off at VDS = 0V
because the freewheeling diode is conducting during
this time. The switching loss for the low-side MOSFET
is usually negligible. Also, the gate-drive current for the
low-side MOSFET is more accurately calculated using
CISS at VDS = 0V instead of gate charge.
Where:
PCONDUCTION = ISW(RMS)2 x RSW
PAC = PAC(OFF) + PAC(ON)
RSW = ON-Resistance of the MOSFET switch.
D = Duty cycle (VOUT/VIN)
Making the assumption that the turn-on and turn-off
transition times are equal, the transition times can be
approximated by:
EQUATION 4-7:
C ISS V GS + C OSS V IN
t T = ---------------------------------------------------------------IG
For the low-side MOSFET:
EQUATION 4-4:
I G LOW -SIDEAVG = C ISS V GS f S
Where:
CISS and COSS are measured at VDS = 0V.
IG = Gate-drive current (1A for the MIC2168)
Total high-side MOSFET switching loss is:
Because the current from the gate drive comes from
the input voltage, the power dissipated in the MIC2168
due to gate drive is:
EQUATION 4-5:
P GATEDRIVE
= V IN I G HIGH -SIDEAVG + I G LOW -SIDEAVG
A convenient figure of merit for switching MOSFETs is
the ON-resistance times the total gate charge (RDS(ON)
× QG). Lower numbers translate into higher efficiency.
Low gate-charge logic-level MOSFETs are a good
choice for use with the MIC2168.
Parameters that are important to MOSFET switch
selection are:
• Voltage Rating
• ON-Resistance
• Total Gate Charge
The voltage ratings for the top and bottom MOSFET
are essentially equal to the input voltage. A safety
factor of 20% should be added to the VDS(MAX) of the
MOSFETs to account for voltage spikes due to circuit
parasitics.
The power dissipated in the switching transistor is the
sum of the conduction losses during the on-time
(PCONDUCTION) and the switching losses that occur
during the period of time when the MOSFETs turn on
and off (PAC).
2020 Microchip Technology Inc.
EQUATION 4-8:
P AC = V IN + V D I PK t T f S
Where:
tT = Switching transition time (typ. 20 ns to 50 ns)
VD = Freewheeling diode drop (typ. 0.5V)
fS = Switching frequency (nom. 1 MHz)
The low-side MOSFET switching losses are negligible
and can be ignored for these calculations.
4.7
Inductor Selection
Values for inductance, peak, and RMS currents are
required to select the output inductor. The input and
output voltages and the inductance value determine
the peak-to-peak inductor ripple current. Generally,
higher inductance values are used with higher input
voltages. Larger peak-to-peak ripple currents will
increase the power dissipation in the inductor and
MOSFETs. Larger output ripple currents will also
require more output capacitance to smooth out the
larger ripple current. Smaller peak-to-peak ripple
currents require a larger inductance value and
therefore a larger and more expensive inductor. A good
compromise between size, loss, and cost is to set the
inductor ripple current to be equal to 20% of the
maximum output current. The inductance value is
calculated by the equation below.
DS20006145A-page 11
MIC2168
EQUATION 4-9:
EQUATION 4-13:
V OUT V IN MAX – V OUT
L = ------------------------------------------------------------------------------------V IN MAX f S 0.2 I OUT MAX
Where:
fS = Switching frequency, 1 MHz
0.2 = The ratio of AC ripple current to DC output
current.
VIN(MAX) = Maximum input voltage.
The peak-to-peak inductor current (AC ripple current)
is:
EQUATION 4-10:
V OUT V IN MAX – V OUT
I PP = ----------------------------------------------------------------------V IN MAX f S L
The peak inductor current is equal to the average
output current plus one half of the peak-to-peak
inductor ripple current.
EQUATION 4-11:
2
IP
1
I INDUCTOR RMS = I OUT MAX 1 + --- ---------------------------
3 I OUT MAX
Maximizing efficiency requires the proper selection of
core material and minimizing the winding resistance.
The high frequency operation of the MIC2168 requires
the use of ferrite materials for all but the most cost
sensitive applications. Lower cost iron powder cores
may be used but the increase in core loss will reduce
the efficiency of the power supply. This is especially
noticeable at low output power. The winding resistance
decreases efficiency at the higher output current levels.
The winding resistance must be minimized although
this usually comes at the expense of a larger inductor.
The power dissipated in the inductor is equal to the sum
of the core and copper losses. At higher output loads,
the core losses are usually insignificant and can be
ignored. At lower output currents, the core losses can
be a significant contributor. Core loss information is
usually available from the magnetics vendor. Copper
loss in the inductor is calculated by the equation below:
EQUATION 4-12:
2
P INDUCTOR CU = I INDUCTOR RMS R WINDING
R WINDING HOT = R WINDING 20C
1 + 0.0042 T HOT – T 20C
Where:
THOT = Temperature of the wire under operating load
T20°C = Ambient temperature.
RWINDING(20°C) = Room temperature winding
resistance (usually specified by manufacturer).
4.8
Output Capacitor Selection
The output capacitor values are usually determined by
the capacitor’s ESR (equivalent series resistance).
Voltage and RMS current capability are two other
important factors selecting the output capacitor.
Recommended
capacitors
tantalum,
low-ESR
aluminum electrolytics, and POSCAPS. The output
capacitor’s ESR is usually the main cause of output
ripple.
The output capacitor ESR also affects the overall
voltage feedback loop from stability point of view. See
“Feedback Loop Compensation” section for more
information. The maximum value of ESR is calculated:
EQUATION 4-14:
V OUT
R ESR ----------------I PP
Where:
VOUT = Peak-to-peak output voltage ripple.
IPP = Peak-to-peak inductor ripple current.
The total output ripple is a combination of the ESR
output capacitance. The total ripple is calculated below.
EQUATION 4-15:
V OUT =
I PP 1 – D 2
2
-------------------------------- + I PP R ESR
C
f
OUT
S
Where:
D = Duty cycle.
COUT = Output capacitance.
fS = Switching frequency.
The voltage rating of capacitor should be twice the
voltage for a tantalum and 20% greater for an
aluminum electrolytic.
The output capacitor RMS current is calculated below.
EQUATION 4-16:
The resistance of the copper wire, RWINDING, increases
with temperature. The value of the winding resistance
used should be at the operating temperature.
DS20006145A-page 12
I PP
I COUT RMS = --------12
2020 Microchip Technology Inc.
MIC2168
The power dissipated in the output capacitor is:
4.10
Voltage Setting Components
EQUATION 4-17:
The MIC2168 requires two resistors to set the output
voltage as shown in Figure 4-2.
2
P DISS COUT = I COUT RMS R ESR COUT
R1
4.9
Error
Amp
Input Capacitor Selection
The input capacitor should be selected for ripple
current rating and voltage rating. Tantalum input
capacitors may fail when subjected to high inrush
currents, caused by turning the input supply on.
Tantalum input capacitor voltage rating should be at
least 2 times the maximum input voltage to maximize
reliability. Aluminum electrolytic, OS-CON, and
multilayer polymer film capacitors can handle the
higher inrush currents without voltage derating. The
input voltage ripple will primarily depend on the input
capacitor’s ESR. The peak input current is equal to the
peak inductor current, so:
FB
7
R2
VREF
0.8V
MIC2168 [adj.]
FIGURE 4-2:
Configuration.
Voltage-Divider
In this figure, VREF for MIC2168 is typically 0.8V.
The output voltage is determined by Equation 4-21.
EQUATION 4-21:
EQUATION 4-18:
R1
V OUT = V REF 1 + -------
R2
V IN = I INDUCTOR PEAK R ESR CIN
The input capacitor must be rated for the input current
ripple. The RMS value of input capacitor current is
determined at the maximum output current. Assuming
the peak-to-peak inductor ripple current is low:
EQUATION 4-19:
I CIN RMS I OUT MAX D 1 – D
A typical value for R1 can be between 3 kΩ and 10 kΩ.
If R1 is too large, it may allow noise to be introduced
into the voltage feedback loop. If R1 is too small, it will
decrease the efficiency of the power supply, especially
at light loads. Once R1 is selected, R2 can be
calculated.
EQUATION 4-22:
V REF R1
R2 = ---------------------------------V OUT – V REF
The power dissipated in the input capacitor is:
EQUATION 4-20:
4.11
2
P DISS CIN = I CIN RMS R ESR CIN
External Schottky Diode
An external freewheeling diode is used to keep the
inductor current flow continuous while both MOSFETs
are turned off. This dead time prevents current from
flowing unimpeded through both MOSFETs and is
typically 15 ns. The diode conducts twice during each
switching cycle. Although the average current through
this diode is small, the diode must be able to handle the
peak current.
The reverse voltage requirement of the diode is:
EQUATION 4-23:
V DIODE RRM = V IN
2020 Microchip Technology Inc.
DS20006145A-page 13
MIC2168
The power dissipated by the Schottky diode is:
L
DCR
EQUATION 4-24:
VOUT
P DIODE = I D AVG V F
ESR
Where:
COUT
VF = Forward voltage at the peak diode current.
4.12
FIGURE 4-3:
The Output LC Filter in a
Voltage Mode Buck Converter.
EQUATION 4-25:
(
G(S) =
Plotting this transfer function with the following
assumed values (L = 2 μH, DCR = 0.009Ω, COUT =
1000 μF, ESR = 0.050Ω) gives much insight toward
why one needs to compensate on the COMP pin.
Figure 4-4 and Figure 4-5 show the gain curve and
phase curve for the transfer function in Equation 4-25.
30
30
7.5
Feedback Loop Compensation
The MIC2168 controller comes with an internal
transconductance
error
amplifier
used
for
compensating the voltage feedback loop by placing a
capacitor (C1) in series with a resistor (R1) and another
capacitor C2 in parallel from the COMP pin to ground.
See the Functional Block Diagram.
15
37.5
60 60
100
100
3
1.10
FIGURE 4-4:
The power stage of a voltage mode controller has an
inductor, L1, with its winding resistance (DCR)
connected to the output capacitor, COUT, with its
electrical series resistance (ESR) as shown in Figure
4-3. The transfer function G(s), for such a system is:
0
5
4
1 .10
f
6
1 .10
1 .10
1000000
Gain Curve for G(S).
Power Stage
0
50
PHASE
4.13
)
(1 + ESR × s × C)
DCR × s × C + s2 × L × C + 1 + ESR × s × C
GAIN
The external Schottky diode, D1, is not necessary for
circuit operation because the low-side MOSFET
contains a parasitic body diode. The external diode will
improve efficiency and decrease high frequency noise.
If the MOSFET body diode is used, it must be rated to
handle the peak and average current. The body diode
has a relatively slow reverse recovery time and a
relatively high forward voltage drop. The power lost in
the diode is proportional to the forward voltage drop of
the diode. As the high-side MOSFET starts to turn on,
the body diode becomes a short circuit for the reverse
recovery period, dissipating additional power. The
diode recovery and the circuit inductance will cause
ringing during the high-side MOSFET turn-on. An
external Schottky diode conducts at a lower forward
voltage preventing the body diode in the MOSFET from
turning on. The lower forward voltage drop dissipates
less power than the body diode. The lack of a reverse
recovery mechanism in a Schottky diode causes less
ringing and less power loss. Depending on the circuit
components and operating conditions, an external
Schottky diode will give a 1/2% to 1% improvement in
efficiency.
100
150
180
100
100
FIGURE 4-5:
DS20006145A-page 14
3
1.10
4
1 .10
f
5
1 .10
6
1 .10
1000000
Phase Curve for G(S).
2020 Microchip Technology Inc.
MIC2168
It can be seen from the transfer function G(S) and the
gain curve that the output inductor and capacitor create
a two pole system with a break frequency at:
0
0
50
1
f C = ----------------------------------------2 L C OUT
Therefore, fLC = 3.6 kHz.
By looking at the phase curve, it can be seen that the
output capacitor ESR (0.050Ω) cancels one of the two
poles (LCOUT) system by introducing a zero at:
EQUATION 4-27:
1
f ZERO = --------------------------------------------------2 ESR C OUT
Therefore, fZERO = 6.36 kHz.
From the point of view of compensating the voltage
loop, it is recommended to use higher ESR output
capacitors because they provide a 90° phase gain in
the power path. For comparison purposes, Figure 4-6
shows the same phase curve with an ESR value of
0.002Ω.
PHASE
EQUATION 4-26:
100
150
180
100
100
FIGURE 4-6:
0.002Ω.
3
1.10
4
1 .10
f
5
1 .10
6
1 .10
1000000
Phase Curve with ESR =
It can be seen from Figure 4-5 that at 50 kHz, the
phase is approximately –90° versus Figure 4-6 where
the number is –150°. This means that the
transconductance error amplifier has to provide a
phase boost of about 45° to achieve a closed loop
phase margin of 45° at a crossover frequency of
50 kHz for Figure 4-4, versus 105° for Figure 4-6. The
simple RC and C2 compensation scheme allows a
maximum error amplifier phase boost of about 90°.
Therefore, it is easier to stabilize the MIC2168 voltage
control loop by using high ESR value output capacitors.
4.14
gm Error Amplifier
It is undesirable to have high error amplifier gain at high
frequencies because high frequency noise spikes
would be picked up and transmitted at large amplitude
to the output, thus, gain should be permitted to fall off
at high frequencies. At low frequency, it is desired to
have high open-loop gain to attenuate the power line
ripple. Thus, the error amplifier gain should be allowed
to increase rapidly at low frequencies.
The transfer function with R1, C1, and C2 for the
internal gm error amplifier can be approximated by
Equation 4-28.
EQUATION 4-28:
1 + R1 S C1
Error Amplifer (z) = g m ---------------------------------------------------------------------------------------------------C1 C2 S-
s C1 + C2 1 + R1 -----------------------------
C1 + C2
The equation above can be simplified by assuming C2
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