MIC2182
Micrel
MIC2182
High-Efficiency Synchronous Buck Controller
General Description
Features
Micrel’s MIC2182 is a synchronous buck (step-down) switching regulator controller. An all N-channel synchronous architecture and powerful output drivers allow up to a 20A output
current capabilty. The PWM and skip-mode control scheme
allows efficiency to exceed 95% over a wide range of load
current, making it ideal for battery powered applications, as
well as high current distributed power supplies.
The MIC2182 operates from a 4.5V to 32V input and can
operate with a maximum duty cycle of 86% for use in lowdropout conditions. It also features a shutdown mode that
reduces quiescent current to 0.1µA.
The MIC2182 achieves high efficiency over a wide output
current range by automatically switching between PWM and
skip mode. Skip-mode operation enables the converter to
maintain high efficiency at light loads by turning off circuitry
pertaining to PWM operation, reducing the no-load supply
current from 1.6mA to 600µA. The operating mode is internally selected according to the output load conditions. Skip
mode can be defeated by pulling the PWM pin low which
reduces noise and RF interference.
The MIC2182 is available in a 16-pin SOP (small-outline
package) and SSOP (shrink small-outline package) with an
operating range from –40°C to +85°C.
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•
•
•
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4.5V to 32V Input voltage range
1.25V to 6V Output voltage range
95% efficiency
300kHz oscillator frequency
Current sense blanking
5Ω impedance MOSFET Drivers
Drives N-channel MOSFETs
600µA typical quiescent current (skip-mode)
Logic controlled micropower shutdown (IQ < 0.1µA)
Current-mode control
Cycle-by-cycle current limiting
Built-in undervoltage protection
Adjustable undervoltage lockout
Easily synchronizable
Precision 1.245V reference output
0.6% total regulation
16-pin SOP and SSOP packages
Frequency foldback overcurrent protection
Sustained short-circuit protection at any input voltage
20A output current capability
Applications
•
•
•
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DC power distribution systems
Notebook and subnotebook computers
PDAs and mobile communicators
Wireless modems
Battery-operated equipment
Typical Application
VIN
4.5V to 30V*
D2
MIC2182-3.3BSM
10
11 SD103BWS
VIN
VDD
R7
100k
C5
0.1µF
6
2
BST
14
EN/UVLO HSD
16
PWM
VSW
15
LSD
13
PGND
12
C4
1nF
C3
0.1µF
C2
2.2nF
GND
R1
2k
1
SS
3
COMP
CSH
8
5
SYNC
VOUT
9
VREF
7
SGND
4
C9
4.7µF
16V
C6
0.1µF
C11
22uf
35V
x2
Q2*
Si4884
Q1*
Si4884
L1
10µH
R2
0.02Ω
D1
B140
VOUT
3.3V/4A
C7
220uf
10V ×2
GND
C13, 1nF
* 30V maximum input voltage limit is due
to standard 30V MOSFET selection.
C1
0.1µF
See “Application Information” section for
5V to 3.3V/10A and other circuits.
4.5V–30V* to 3.3V/4A Converter
Micrel, Inc. • 1849 Fortune Drive • San Jose, CA 95131 • USA • tel + 1 (408) 944-0800 • fax + 1 (408) 474-1000 • http://www.micrel.com
April 22, 2004
1
M9999-042204
MIC2182
Micrel
Ordering Information
Part Number
Voltage
Temperature Range
Package
Lead Finish
MIC2182BM
Adjustable
–40°C to +85°C
16-pin narrow SOP
Standard
3.3V
–40°C to +85°C
16-pin narrow SOP
Standard
MIC2182-3.3BM
MIC2182-5.0BM
5.0V
–40°C to +85°C
16-pin narrow SOP
Standard
Adjustable
–40°C to +85°C
16-pin narrow SSOP
Standard
MIC2182-3.3BSM
3.3V
–40°C to +85°C
16-pin narrow SSOP
Standard
MIC2182-5.0BSM
5.0V
–40°C to +85°C
16-pin narrow SSOP
Standard
MIC2182BSM
MIC2182YM
Adjustable
–40°C to +85°C
16-pin narrow SOP
Pb-Free
MIC2182-3.3YM
3.3V
–40°C to +85°C
16-pin narrow SOP
Pb-Free
MIC2182-5.0YM
5.0V
–40°C to +85°C
16-pin narrow SOP
Pb-Free
MIC2182YSM
Adjustable
–40°C to +85°C
16-pin narrow SSOP
Pb-Free
MIC2182-3.3YSM
3.3V
–40°C to +85°C
16-pin narrow SSOP
Pb-Free
MIC2182-5.0YSM
5.0V
–40°C to +85°C
16-pin narrow SSOP
Pb-Free
Pin Configuration
MIC2182
MIC2182-x.x
SS 1
16 HSD
SS 1
16 HSD
PWM 2
15 VSW
PWM 2
15 VSW
COMP 3
14 BST
COMP 3
14 BST
SGND 4
13 LSD
SGND 4
13 LSD
SYNC 5
12 PGND
SYNC 5
12 PGND
EN/UVLO 6
FB 7
CSH 8
EN/UVLO 6
11 VDD
VREF 7
10 VIN
CSH 8
9 VOUT
Adjustable
16-pin SOP (M)
16-Pin SSOP (SM)
M9999-042204
11 VDD
10 VIN
9 VOUT
Fixed
16-pin SOP (M)
16-Pin SSOP (SM)
2
April 22, 2004
MIC2182
Micrel
Pin Description
Pin Number
Pin Name
1
SS
2
PWM
PWM/Skip-Mode Select (Input): Low sets PWM-mode operation. 1nF
capacitor to ground sets automatic PWM/skip-mode selection.
3
COMP
Compensation (Output): Internal error amplifier output. Connect to capacitor
or series RC network to compensate the regulator control loop.
4
SGND
Small Signal Ground (Return): Route separately from other ground traces to
the (–) terminal of COUT.
5
SYNC
Frequency Synchronization (Input): Optional. Connect to external clock
signal to synchronize the oscillator. Leading edge of signal above the
threshold terminates the switching cycle. Connect to SGND if unused.
6
EN/UVLO
7 (fixed)
VREF
7 (adj)
FB
Feedback (Input): Regulates FB pin to 1.245V. See “Application Information”
for resistor divider calculations.
8
CSH
Current-Sense High (Input): Current-limit comparator noninverting input. A
built-in offset of 100mV between CSH and VOUT pins in conjunction with the
current-sense resistor set the current-limit threshold level. This is also the
positive input to the current sense amplifier.
9
VOUT
10
VIN
[Battery] Unregulated Input (Input): +4.5V to +32V supply input.
11
VDD
5V Internal Linear-Regulator (Output): VDD is the external MOSFET gate
drive supply voltage and an internal supply bus for the IC. Bypass to SGND
with 4.7µF. VDD can supply up to 5mA for external loads.
12
PGND
13
LSD
Low-Side Drive (Output): High-current driver output for external synchronous
MOSFET. Voltage swing is between ground and VDD.
14
BST
Boost (Input): Provides drive voltage for the high-side MOSFET driver. The
drive voltage is higher than the input voltage by VDD minus a diode drop.
15
VSW
Switch (Return): High-side MOSFET driver return.
16
HSD
High-Side Drive (Output): High-current driver output for high-side MOSFET.
This node voltage swing is between ground and VIN + 5V – Vdiode drop.
April 22, 2004
Pin Function
Soft-Start (External Component): Connect external capacitor to ground to
reduce inrush current by delaying and slowing the output voltage rise time.
Rise time is controlled by an internal 5µA current source that charges an
external capacitor to VDD.
Enable/Undervoltage Lockout (Input): Low-level signal powers down the
controller. Input below the 2.5V threshold disables switching and functions
as an accurate undervoltage lockout (UVLO). Input below the threshold
forces complete micropower (< 0.1µA) shutdown.
Reference Voltage (Output): 1.245V output. Requires 0.1µf capacitor to
ground.
Current-Sense Low (Input): Output voltage feedback input and inverting
input for the current limit comparator and the current sense amplifier.
MOSFET Driver Power Ground (Return): Connects to source of synchronous MOSFET and the (–) terminal of CIN
3
M9999-042204
MIC2182
Micrel
Absolute Maximum Ratings (Note 1)
Operating Ratings (Note 2)
Analog Supply Voltage (VIN) ....................................... +34V
Digital Supply Voltage (VDD) ......................................... +7V
Driver Supply Voltage (BST) .................................... VIN +7V
Sense Voltage (VOUT, CSH) ............................. 7V to –0.3V
Sync Pin Voltage (VSYNC) ................................ 7V to –0.3V
Enable Pin Voltage (VEN/UVLO) ...................................... VIN
Power Dissipation (PD)
SOP ................................................ 400mW @ TA= 85°C
SSOP ............................................. 270mW @ TA= 85°C
Ambient Storage Temperature (TS) ......... –65°C to +150°C
ESD, Note 3
Analog Supply Voltage (VIN) ........................ +4.5V to +32V
Ambient Temperature (TA) ......................... –40°C to +85°C
Junction Temperature (TJ) ....................... –40°C to +125°C
Package Thermal Resistance
SOP (θJA) .......................................................... 100°C/W
SSOP (θJA) ........................................................ 150°C/W
Electrical Characteristics
VIN = 15V; SS = open; VPWM = 0V; VSHDN = 5V; ILOAD = 0.1A; TA = 25°C, bold values indicate –40°C ≤ TA ≤ +85°C; Note 4; unless
noted
Parameter
Condition
Min
Typ
Max
Units
Feedback Voltage Reference
1.233
1.245
1.257
V
Feedback Voltage Reference
1.220
1.245
1.270
V
1.208
1.245
1.282
V
MIC2182 [Adjustable], (Note 5)
Feedback Voltage Reference
4.5V < VIN < 32V, 0 < VCSH – VOUT < 75mV
Feedback Bias Current
10
Output Voltage Range
1.25
nA
6
V
Output Voltage Line Regulation
VIN = 4.5V to 32V, VCSH – VOUT = 50mV
0.03
%/V
Output Voltage Load Regulation
25mV < (VCSH – VOUT) < 75mV (PWM mode only)
0.5
%
Output Voltage Total Regulation
0mV < (VCSH – VOUT) < 75mV (full load range) 4.5V < VIN < 32V
0.6
%
MIC2182-3.3
Output Voltage
3.267
3.3
3.333
V
Output Voltage
3.234
3.3
3.366
V
3.201
3.3
3.399
V
Output Voltage
4.5V < VIN < 32V, 0 < VCSH – VOUT < 75mV
Output Voltage Line Regulation
VIN = 4.5V to 32V, VCSH – VOUT = 50mV
0.03
%/V
Output Voltage Load Regulation
25mV < (VCSH – VOUT) < 75mV (PWM mode only)
0.5
%
Output Voltage Total Regulation
0mV < (VCSH – VOUT) < 75mV (full load range) 4.5V < VIN < 32V
0.8
%
MIC2182-5.0
Output Voltage
4.95
5.0
5.05
V
Output Voltage
4.90
5.0
5.10
V
4.85
5.0
5.150
V
Output Voltage
6.5V < VIN < 32V, 0 < VCSH – VOUT < 75mV
Output Voltage Line Regulation
VIN = 6.5V to 32V, VCSH – VOUT = 50mV
0.03
%/V
Output Voltage Load Regulation
25mV < (VCSH – VOUT) < 75mV (PWM mode only)
0.5
%
Output Voltage Total Regulation
0mV < (VCSH – VOUT) < 75mV (full load range) 6.5V < VIN < 32V
0.8
%
PWM Mode
VPWM = 0V, excluding external MOSFET gate drive current
1.6
2.5
mA
Skip Mode
IL = 0mA, VPWM floating (1nF capacitor to ground)
600
1500
µA
Shutdown Quiescent Current
VEN/UVLO = 0V
0.1
5
µA
Digital Supply Voltage (VDD)
IL = 0mA to 5mA
5.3
V
Undervoltage Lockout
VDD upper threshold (turn on threshold)
4.2
V
VDD lower threshold (turn off threshold)
4.1
V
Input and VDD Supply
M9999-042204
4.7
4
April 22, 2004
MIC2182
Parameter
Micrel
Condition
Min
Typ
Max
Units
1.220
1.245
1.270
V
Reference Output (Fixed Versions Only)
Reference Voltage
Reference Line Regulation
6V < VIN < 32V
1
mV
Reference Load Regulation
0µA < IREF < 100µA
2
mV
Enable/UVLO
Enable Input Threshold
0.6
1.1
1.6
V
UVLO Threshold
2.2
2.5
2.8
V
0.1
5
µA
–3.5
–5
–6.5
µA
75
100
135
mV
330
kHz
Enable Input Current
VEN/UVLO = 5V
Soft Start
Soft-Start Current
VSS = 0V
Current Limit
Current-Limit Threshold Voltage
VCSH = VOUT
Error Amplifier
Error Sense Amplifier Gain
20
Current Amp
Current Sense Amplifier Gain
2.0
Oscillator Section
Oscillator Frequency
270
Maximum Duty Cycle
Minimum On-Time
86
VOUT = VOUT(nominal) + 200mV
SYNC Threshold Level
SYNC Input Current
300
0.7
VSYNC = 5V
SYNC Minimum Pulse Width
%
140
250
ns
1.3
1.9
V
0.1
5
µA
200
ns
kHz
SYNC Capture Range
Note 6
330
Frequency Foldback Threshold
measured at VOUT pin
0.75
Foldback Frequency
0.95
1.15
V
60
kHz
ns
Gate Drivers
Rise/Fall Time
CL = 3000pF
60
Output Driver Impedance
source
sink
5
3.5
Driver Nonoverlap Time
8.5
6
Ω
Ω
80
ns
–10
µA
PWM Input
PWM Input Current
VPWM = 0V
Note 1.
Exceeding the absolute maximum rating may damage the device.
Note 2.
The device is not guaranteed to function outside its operating rating.
Note 3.
Devices are ESD sensitive. Handling precautions recommended. Human body model, 1.5k in series with 100pF.
Note 4.
25°C limits are 100% production tested. Limits over the operating temperature range are guaranteed by design and are not production tested.
Note 5.
VIN > 1.3 × VOUT (for the feedback voltage reference and output voltage line and total regulation).
Note 6.
See applications information for limitations on the maximum operating frequency.
April 22, 2004
5
M9999-042204
MIC2182
Micrel
Typical Characteristics
Quiescent Current
vs. Temperature
0.50
0
-0.50
0.20
0.8
Skip
0.6
0.4
0.15
0.10
0.2
0.1
8 12 16 20 24 28 32
SUPPLY VOLTAGE (V)
1.244
1.242
1.240
1.238
1.236
0
4
8 12 16 20 24 28 32
SUPPLY VOLTAGE (V)
VDD
vs. Temperature
Line Regulation
1.250
1.245
1.240
-40 -20 0 20 40 60 80 100120140
TEMPERATURE (°C)
VDD
vs. Temperature
4.96
4.94
4.92
4.90
4.88
4.86
4.84
4.82
-40 -20 0 20 40 60 80 100120140
TEMPERATURE (°C)
M9999-042204
1.248
1.246
VREF (Fixed Versions)
1.255
4.98
1.252
1.250
Skip
0
5.0
4.8
4.6
4.4
4.2
4.0
0
4
8 12 16 20 24 28 32
INPUT VOLTAGE (V)
Load Regulation
1.240
1.230
1.220
1.210
1.200
8 12 16 20 24 28 32
SUPPLY VOLTAGE (V)
0
200 400 600 800 1000
LOAD CURRENT (µA)
VDD
Load Regulation
5.00
4.95
4.90
4.85
4.80
0
5
10
15
20
LOAD CURRENT (mA)
25
Oscillator Frequency
vs. Supply Voltage
Oscillator Frequency
vs. Temperature
10
8
4
1.250
VDD REGULATOR VOLTAGE (V)
1.260
4
1.0
1.260
1.0
FREQUENCY VARIATION (%)
0
PWM
1.5
VREF (Fixed Versions)
REFERENCE VOLTAGE (V)
SHUTDOWN
(µA)
2.0
0
Line Regulation
1.256
1.254
VDD REGULATOR VOLTAGE (V)
0.5
-0.5
0.4
0.3
REFERENCE VOLTAGE (V)
0
2.5
VREF (Fixed Versions)
FREQUENCY VARIATION (%)
CURRENT (µA)
UVLO Mode
(mA)
0.5
3.0
0.5
0
-40 -20 0 20 40 60 80 100120140
TEMPERATURE (°C)
1.5
1.0
SHUTDOWN
(µA)
0.05
Quiescent Current
vs. Supply Voltage
REFERENCE VOLTAGE (V)
UVLO Mode
(mA)
CURRENT (mA)
1.2
1.0
0.2
0
-40 -20 0 20 40 60 80 100120140
TEMPERATURE (°C)
VDD REGULATOR VOLTAGE (V)
3.5
1.00
PWM
1.6
1.4
0
4.0
1.50
CURRENT (mA)
CURRENT (mA)
2.0
1.8
Quiescent Current
vs. Supply Voltage
Quiescent Current
vs. Temperature
6
4
2
0
-2
-4
-6
-8
-10
-40 -20 0 20 40 60 80 100120140
TEMPERATURE (°C)
6
0.8
0.6
0.4
0.2
0
-0.2
-0.4
-0.6
-0.8
-1.0
0
4
8 12 16 20 24 28 32
SUPPLY VOLTAGE (V)
April 22, 2004
MIC2182
Micrel
CURRENT (µA)
4.8
4.6
4.4
4.2
4.0
-40 -20 0 20 40 60 80 100120140
TEMPERATURE (°C)
April 22, 2004
Current-Limit
Foldback
5
0.12
OUTPUT VOLTAGE (V)
5.0
Overcurrent Threshold
vs. Temperature
OVERCURRENT THRESHOLD (V)
Soft-Start Current
vs. Temperature
0.11
0.10
0.09
0.08
-40 -20 0 20 40 60 80 100120140
TEMPERATURE (°C)
7
4
3
2
1
0
0
VIN = 5V
VOUT = 3.3V
RCS = 15mΩ
1 2 3 4 5 6 7
OUTPUT CURRENT (A)
8
M9999-042204
MIC2182
Micrel
Block Diagrams
VIN
CIN
VDD
EN/UVLO
6
Reference V
IN
VDD
11
4.7µF
VIN
D2
VBG
1.245V
10
SS
1
VBST
14
Control
Logic
HSD
16
Q2
CBST
L1
VSW
PWM
2
RCS
VOUT
15
LSD
13
COUT
D1
Q1
PGND
Current
Limit
12
PWM Mode
to Skip
Mode
0.024V
Skip-Mode
Current
Limit
0.07V
Low
Comp
PWM OUTPUT
–2%VBG
Hysteresis
Comp
Current
Sense
Amp
PWM
CSH
8
CORRECTIVE
RAMP
VOUT
VBG
RESET
SYNC
5
Oscillator
AV = 2
R1
Error
Amp
FB
7
COMP
CCOMP
9
3
100k
RCOMP
SGND
Gm = 0.2×10-3
4
MIC2182 [adj.]
R2
R1
VOUT = 1.245V 1 +
R2
VOUT(max) = 6.0V
Figure 2a. Adjustable Output Voltage Version
M9999-042204
8
April 22, 2004
MIC2182
Micrel
VIN
CIN
VDD
EN/UVLO
6
VDD
11
Reference V
IN
4.7µF
VIN
D2
VBG
1.245V
10
SS
1
VBST
14
Control
Logic
HSD
16
Q2
CBST
L1
VSW
PWM
2
RCS
VOUT
15
LSD
13
D1
COUT
Q1
PGND
Current
Limit
12
PWM Mode
to Skip
Mode
0.024V
Skip-Mode
Current
Limit
0.07V
Low
Comp
PWM OUTPUT
–2%VBG
Hysteresis
Comp
Current
Sense
Amp
PWM
CSH
8
VOUT
VBG
RESET
CORRECTIVE
RAMP
SYNC
5
Oscillator
Error
Amp
SGND
R2
50k
3
100k
-3
Gm = 0.2×10
RCOMP
* 82.5k for 3.3V Output
150k for 5V Output
R1*
COMP
CCOMP
9
AV = 2
4
VREF
7
MIC2182-x.x
Figure 2b. Fixed Output Voltage Versions
April 22, 2004
9
M9999-042204
MIC2182
Micrel
Control Loop
PWM and Skip Modes of Operation
The MIC2182 operates in PWM (pulse-width-modulation)
mode at heavier output load conditions. At lighter load conditions, the controller can be configured to automatically switch
to a pulse-skipping mode to improve efficiency. The potential
disadvantage of skip mode is the variable switching frequency that accompanies this mode of operation. The occurrence of switching pulses depends on component values as
well as line and load conditions. There is an external sync
function that is disabled in skip mode. In PWM mode, the
synchronous buck converter forces continuous current to
flow in the inductor. In skip mode, current through the inductor
can settle to zero, causing voltage ringing across the inductor. Pulling the PWM pin (pin 2) low will force the controller to
operate in PWM mode for all load conditions, which will
improve cross regulation of transformer-coupled, multiple
output configurations.
PWM Control Loop
The MIC2182 uses current-mode control to regulate the
output voltage. This method senses the output voltage (outer
loop) and the inductor current (inner loop). It uses inductor
current and output voltage to determine the duty cycle of the
buck converter. Sampling the inductor current removes the
inductor from the control loop, which simplifies compensation.
Functional Description
See “Applications Information” following this section for component selection information and Figure 14 and Tables 1
through 5 for predesigned circuits.
The MIC2182 is a BiCMOS, switched-mode, synchronous
step-down (buck) converter controller. Current-mode control
is used to achieve superior transient line and load regulation.
An internal corrective ramp provides slope compensation for
stable operation above a 50% duty cycle. The controller is
optimized for high-efficiency, high-performance dc-dc converter applications.
The MIC2182 block diagrams are shown in Figure 2a and
Figure 2b.
The MIC2182 controller is divided into 6 functions.
• Control loop
- PWM operation
- Skip-mode operation
• Current limit
•
•
•
•
Reference, enable, and UVLO
MOSFET gate drive
Oscillator and sync
Soft start
VIN
CIN
VDD
VDD
11
Reference V
IN
4.7µF
VIN
D2
VBG
1.245V
10
CONTROL LOGIC AND
PULSE-WIDTH MODULATOR
VBST
14
HSD
16
Q2
CBST
L1
VSW
RCS
VOUT
15
LSD
13
PWM Mode
to Skip
Mode
COUT
D1
Q1
PGND
12
Q
R
S
LOW
FORCES
SKIP MODE
0.024V
Current
Sense
Amp
PWM
COMPARATOR
CSH
8
VBG
VOUT
RESET
CORRECTIVE
RAMP
AV = 2
9
R1
Oscillator
Error
Amp
FB
7
COMP
CCOMP
3
100k
RCOMP
Gm = 0.2×10-3
R1
VOUT = 1.245V 1 +
R2
R2
MIC2182 [adj.] PWM Mode
Figure 3. PWM Operation
M9999-042204
10
April 22, 2004
MIC2182
Micrel
Skip-Mode Control Loop
A block diagram of the MIC2182 PWM current-mode control
loop is shown in Figure 3 and the PWM mode voltage and
current waveforms are shown in figure 5A. The inductor
current is sensed by measuring the voltage across the
resistor, RCS. A ramp is added to the amplified current-sense
signal to provide slope compensation, which is required to
prevent unstable operation at duty cycles greater than 50%.
A transconductance amplifier is used for the error amplifier,
which compares an attenuated sample of the output voltage
with a reference voltage. The output of the error amplifier is
the COMP (compensation) pin, which is compared to the
current-sense waveform in the PWM block. When the current
signal becomes greater than the error signal, the comparator
turns off the high-side drive. The COMP pin (pin 3) provides
access to the output of the error amplifier and allows the use
of external components to stabilize the voltage loop.
This control method is used to improve efficiency at light
output loads. At light output currents, the power drawn by the
MIC2182 is equal to the input voltage times the IC supply
current (IQ). At light output currents, the power dissipated by
the IC can be a significant portion of the total output power,
which lowers the efficiency of the power supply. The MIC2182
draws less supply current in skip mode by disabling portions
of the control and drive circuitry when the IC is not switching.
The disadvantage of this method is greater output voltage
ripple and variable switching frequency.
A block diagram of the MIC2182 skip mode is shown in Figure
4. Skip mode voltage and current waveforms are shown in
figure 5B.
VIN
CIN
VDD
VDD
11
Reference V
IN
4.7µF
VIN
D2
VBG
1.245V
10
CONTROL LOGIC AND
SKIP-MODE LOGIC
VBST
14
HSD
16
Q2
VSW
LOW-SIDE DRIVER
ONE SHOT
CBST
L1
RCS
VOUT
15
LSD
13
COUT
Q1
PGND
12
Q
R
S
Skip-Mode
Current
Limit
0.07V
Low
Comp
ONE SHOT
–2%VBG
Hysteresis
Comp
±1%
VBG
LOW
FORCES
PWM MODE
Current
Sense
Amp
CSH
8
VOUT
AV = 2
9
R1
FB
7
R1
VOUT = 1.245V 1 +
R2
MIC2182 [adj.] Skip Mode
R2
Figure 4. Skip-Mode Operation
April 22, 2004
11
M9999-042204
MIC2182
Micrel
VIN
VSW
0V
IL1
ILOAD
0A
Reset
Pulse
VDD
0V
VIN + VDD
VHSD
0V
VLSD
VDD
0V
Figure 5a. PWM-Mode Timing
VDD
VHSD
0V
VDD
VLSD
VSW
0V
VIN
VOUT
0V
ILIM(skip)
IL1
0A
Vone-shot
VDD
0V
VOUT
+1%
VNOMINAL
–1%
0V
IOUT
0A
Figure 5b. Skip-Mode Timing
Figure 6 shows the improvement in efficiency that skip mode
makes when at lower output currents.
A hysteretic comparator is used in place of the PWM error
amplifier and a current-limit comparator senses the inductor
current. A one-shot starts the switching cycle by momentarily
turning on the low side MOSFET to insure the high-side drive
boost capacitor, Cbst, is fully charged. The high-side MOSFET is turned on and current ramps up in the inductor, L1.
The high-side drive is turned off when either the peak voltage
on the input of the current-sense comparator exceeds the
threshold, typically 35mV, or the output voltage rises above
the hysteretic threshold of the output voltage comparator.
Once the high-side MOSFET is turned off, the load current
discharges the output capacitor, causing VOUT to fall. The
cycle repeats when VOUT falls below the lower threshold, –
1%.
The maximum peak inductor current depends on the skipmode current-limit threshold and the value of the currentsense resistor, RCS.
Iinductor(peak) =
M9999-042204
100
PWM
EFFICIENCY (%)
80
Skip
60
40
20
0
0.01
0.1
1
10
OUTPUT CURRENT (A)
100
Figure 6. Efficiency
35mV
R sense
12
April 22, 2004
MIC2182
Micrel
Switching from PWM to Skip Mode
rent-limit threshold is 100mV+35mV –25mV. The currentsense resistor must be sized using the minimum current-limit
threshold. The external components must be designed to
withstand the maximum current limit. The current-sense
resistor value is calculated by the equation below:
The current sense amplifier in Figure 3 monitors the average
voltage across the current-sense resistor. The controller will
switch from PWM to skip mode when the average voltage
across the current-sense resistor drops below approximately
12mV. This is shown in Figure 7b. The average output current
at this transition level for is calculated below.
RCS =
0.012
IOUT(skipmode) =
RCS
IOUT(max) =
The frequency of occurrence of the skip-mode current pulses
increase as the output current increases until the hysteretic
duty cycle reaches 100% (continuous pulses). Increasing the
current past this point will cause the output voltage will drop.
The low limit comparator senses the output voltage when it
drops below 2% of the set output and automatically switches
the converter to PWM mode.
The inductor current in skip mode is a triangular wave shape
a minimum value of 0 and a maximum value of 35mV/RCS
(see Figure 7b). The maximum average output current in skip
mode is the average value of the inductor waveform:
35mV
RCS
The capacitor on the PWM pin (pin 2) is discharged when the
IC transitions from skip to PWM mode. This forces the IC to
remain in PWM mode for a fixed period of time. The added
delay prevents unwanted switching between PWM and skip
mode. The capacitor is charged with a 10uA current source
on pin 2. The threshold on pin 2 is 2.5V. The delay for a typical
1nF capacitor is:
CPWM × Vthreshold 1nF × 2.5V
=
= 250µs
Isource
10µA
where:
CPWM = capacitor connected to pin 2
Current Limit
The current-limit circuit operates during PWM mode. The
output current is detected by the voltage drop across the
external current-sense resistor (RCS in Figure 2.). The cur-
Inductor
Current
135mV
RCS
The current-sense pins CSH (pin 8) and VOUT (pin 9) are
noise sensitive due to the low signal level and high input
impedance. The PCB traces should be short and routed close
to each other. A small (1nF to 0.1µF) capacitor across the pins
will attenuate high frequency switching noise.
When the peak inductor current exceeds the current-limit
threshold, the current-limit comparator, in Figure 2, turns off
the high-side MOSFET for the remainder of the cycle. The
output voltage drops as additional load current is pulled from
the converter. When the output voltage reaches approximately 0.95V, the circuit enters frequency-foldback mode
and the oscillator frequency will drop to 60kHz while maintaining the peak inductor current equal to the nominal 100mV
across the external current-sense resistor. This limits the
maximum output power delivered to the load under a short
circuit condition.
Reference, Enable, and UVLO Circuits
The output drivers are enabled when the following conditions
are satisfied:
• The VDD voltage (pin 11) is greater than its
undervoltage threshold (typically 4.2V).
• The voltage on the enable pin is greater than the
enable UVLO threshold (typically 2.5V)
The internal bias circuit generates a 1.245V bandgap reference voltage for the voltage error amplifier and a 5V VDD
voltage for the gate drive circuit. The reference voltage in the
fixed-output-voltage versions of the MIC2182 is buffered and
brought to pin 7. The VREF pin should be bypassed to GND
(pin 4) with a 0.1µF capacitor. The adjustable version of the
MIC2182 uses pin 7 for output voltage sensing. A decoupling
capacitor on pin 7 is not used in the adjustable output voltage
version.
Switching from Skip to PWM Mode
t delay =
IOUT(max)
The maximum output current is:
where:
0.012 = threshold voltage of the internal comparator
RCS = current-sense resistor value
IOUT(max skipmode) = 0.5 ×
75mV
35mV THRESHOLD
ACROSS RCS.
ILIM(skip)
0A
Figure 7a. Maximum Skip-Mode-Load Inductor Current
IMIN(PWM)
Inductor
Current
12mV THRESHOLD
OF AVERAGE VOLTAGE
ACROSS RCS.
0A
Figure 7b. Minimum PWM-Mode-Load Inductor Current for PWM Operation
April 22, 2004
13
M9999-042204
MIC2182
Micrel
Oscillator and Sync
The internal oscillator is free running and requires no external
components. The nominal oscillator frequency is 300kHz. If
the output voltage is below approximately 0.95V, the oscillator operates in a frequency-foldback mode and the switching
frequency is reduced to 60kHz.
The SYNC input (pin 5) allows the MIC2182 to synchronize
with an external clock signal. The rising edge of the sync
signal generates a reset signal in the oscillator, which turns
off the low-side gate drive output. The high-side drive then
turns on, restarting the switching cycle. The sync signal is
inhibited when the controller operates in skip mode or during
frequency foldback. The sync signal frequency must be
greater than the maximum specified free running frequency
of the MIC2182. If the synchronizing frequency is lower,
double pulsing of the gate drive outputs will occur. When not
used, the sync pin must be connected to ground.
Figure 8 shows the timing between the external sync signal
(trace 2), the low-side drive (trace 1) and the high-side drive
(trace R1). There is a delay of approximately 250ns between
the rising edge of the external sync signal and turnoff of the
low-side MOSFET gate drive.
Some concerns of operating at higher frequencies are:
• Higher power dissipation in the internal VDD
regulator. This occurs because the MOSFET
gates require charge to turn on the device. The
average current required by the MOSFET gate
increases with switching frequency. This increases the power dissipated by the internal
VDD regulator. Figure 10 shows the total gate
charge which can be driven by the MIC2182
over the input voltage range, for different values
of switching frequency. The total gate charge
includes both the high- and low-side MOSFETs.
The larger SOP package is capable of dissipating more power than the SSOP package and
can drive larger MOSFETs with higher gate
drive requirements.
VSS
SYNC
SIGNAL
VOUT
LOW-SIDE HIGH-SIDE
DRIVE
DRIVE
The enable pin (pin 6) has two threshold levels, allowing the
MIC2182 to shut down in a low current mode, or turn off output
switching in UVLO mode. An enable pin voltage lower than
the shutdown threshold turns off all the internal circuitry and
reduces the input current to typically 0.1µA.
If the enable pin voltage is between the shutdown and UVLO
thresholds, the internal bias, VDD, and reference voltages are
turned on. The soft-start pin is forced low by an internal
discharge MOSFET. The output drivers are inhibited from
switching and remain in a low state. Raising the enable
voltage above the UVLO threshold of 2.5V allows the softstart capacitor to charge and enables the output drivers.
Either of two UVLO conditions will pull the soft-start capacitor
low.
• When the VDD drops below 4.1V
• When the enable pin drops below the 2.5V
threshold
MOSFET Gate Drive
The MIC2182 high-side drive circuit is designed to switch an
N-channel MOSFET. Referring to the block diagram in Figure
2, a bootstrap circuit, consisting of D2 and CBST, supplies
energy to the high-side drive circuit. Capacitor CBST is
charged while the low-side MOSFET is on and the voltage on
the VSW pin (pin 15) is approximately 0V. When the high-side
MOSFET driver is turned on, energy from CBST is used to turn
the MOSFET on. As the MOSFET turns on, the voltage on the
VSW pin increases to approximately VIN. Diode D2 is reversed biased and CBST floats high while continuing to keep
the high-side MOSFET on. When the low-side switch is
turned back on, CBST is recharged through D2.
The drive voltage is derived from the internal 5V VDD bias
supply. The nominal low-side gate drive voltage is 5V and the
nominal high-side gate drive voltage is approximately 4.5V
due the voltage drop across D2. A fixed 80ns delay between
the high- and low-side driver transitions is used to prevent
current from simultaneously flowing unimpeded through both
MOSFETs.
TIME
TIME
Figure 8. Sync Waveforms
M9999-042204
Figure 9. Startup Waveforms
14
April 22, 2004
MIC2182
Micrel
• Reduced maximum duty cycle due to switching
transition times and constant delay times in the
controller. As the switching frequency increased,
the switching period decreases. The switching
transition times and constant delays in the
MIC2182 start to become noticeable. The effect
is to reduce the maximum duty cycle of the
controller. This will cause the minimum input to
output differential voltage (dropout voltage) to
increase.
The soft-start voltage is applied directly to the PWM comparator. A 5uA internal current source is used to charge up the
soft-start capacitor. The capacitor is discharged when either
the enable voltage drops below the UVLO threshold (2.5V) or
the VDD voltage drops below the UVLO level (4.1V).
The part switches at a minimum duty cycle when the soft-start
pin voltage is less than 0.4V. This maintains a charge on the
bootstrap capacitor and insures high-side gate drive voltage.
As the soft-start voltage rises above 0.4V, the duty cycle
increases from the minimum duty cycle to the operating duty
cycle. The oscillator runs at the foldback frequency of 60kHz
until the output voltage rises above 0.95V. Above 0.95V, the
switching frequency increases to 300kHz (or the sync’d
frequency), causing the output voltage to rise a greater rate.
The rise time of the output is dependent on the soft-start
capacitor, output capacitance, output voltage, and load current. The oscilloscope photo in Figure 9 show the output
voltage and the soft-start pin voltage at startup.
Minimum Pulse Width
The MIC2182 has a specified minimum pulse width. This
minimum pulse width places a lower limit on the minimum
duty cycle of the buck converter. When the MIC2182 is
operating in forced PWM mode (pin 2 low) and when the
output current is very low or zero, there is a limit on the ratio
of VOUT/VIN. If this limit is exceeded, the output voltage will
rise above the regulated voltage level. A minimum load is
required to prevent the output from rising up. This will not
occur for output voltages greater than 3V.
Figure 11 should be used as a guide when the MIC2182 is
forced into PWM-only mode. The actual maximum input
voltage will depend on the exact external components used
(MOSFETs, inductors, etc.).
100
GATE CHARGE (nC)
SOP
80
60
300kHz
40
400kHz
20
0
500kHz
0
4
8 12 16 20 24 28 32
SUPPLY VOLTAGE (V)
Figure 10a. SOP Gate Charge vs. Input Voltage
100
GATE CHARGE (nC)
SSOP
80
60
300kHz
40
400kHz
20
500kHz
0
0
4
8 12 16 20 24 28 32
SUPPLY VOLTAGE (V)
35
It is recommended that the user limits the maximum synchronized frequency to 600kHz. If a higher synchronized frequency is required, it may be possible and will be design
dependent. Please consult Micrel applications for assistance.
Soft Start
Soft start reduces the power supply input surge current at
startup by controlling the output voltage rise time. The input
surge appears while the output capacitance is charged up. A
slower output rise time will draw a lower input surge current.
Soft start may also be used for power supply sequencing.
30
April 22, 2004
INPUT VOLTAGE (V)
Figure 10b. SSOP Gate Charge vs. Input Voltage
25
20
15
10
0
1
2
3
4
5
OUTPUT VOLTAGE (V)
6
Figure 11. Max. Input Voltage in Forced-PWM Mode
This restriction does not occur when the MIC2182 is set to
automatic mode (pin 2 connected to a capacitor) since the
converter operates in skip mode at low output current.
15
M9999-042204
MIC2182
Micrel
output currents, the core losses can be a significant contributor. Core loss information is usually available from the magnetics vendor.
Copper loss in the inductor is calculated by the equation
below:
Applications Information
The following applications information includes component
selection and design guidelines. See Figure 14 and Tables 1
through 5 for predesigned circuits.
Inductor Selection
Values for inductance, peak, and RMS currents are required
to select the output inductor. The input and output voltages
and the inductance value determine the peak to peak inductor
ripple current. Generally, higher inductance values are used
with higher input voltages. Larger peak to peak ripple currents
will increase the power dissipation in the inductor and
MOSFETs. Larger output ripple currents will also require
more output capacitance to smooth out the larger ripple
current. Smaller peak to peak ripple currents require a larger
inductance value and therefore a larger and more expensive
inductor. A good compromise between size, loss and cost is
to set the inductor ripple current to be equal to 20% of the
maximum output current.
The inductance value is calculated by the equation below.
L=
Pinductor Cu = Iinductor(rms)2 × R winding
The resistance of the copper wire, Rwinding, increases with
temperature. The value of the winding resistance used should
be at the operating temperature.
(
where:
THOT = temperature of the wire
under operating load
T20°C = ambient temperature
Rwinding(20°C) is room temperature winding resistance
(usually specified by the manufacturer)
Current-Sense Resistor Selection
Low inductance power resistors, such as metal film resistors
should be used. Most resistor manufacturers make low
inductance resistors with low temperature coefficients, designed specifically for current-sense applications. Both resistance and power dissipation must be calculated before the
resistor is selected. The value of RSENSE is chosen based on
the maximum output current and the maximum threshold
level. The power dissipated is based on the maximum peak
output current at the minimum overcurrent threshold limit.
VOUT × (VIN(max) − VOUT )
VIN(max) × fS × 0.2 × IOUT(max)
where:
fS = switching frequency
0.2 = ratio of ac ripple current to dc output current
VIN(max) = maximum input voltage
The peak-to-peak inductor current (ac ripple current) is:
IPP =
VOUT × (VIN(max) − VOUT )
VIN(max) × fS × L
RSENSE =
The peak inductor current is equal to the average output
current plus one half of the peak to peak inductor ripple
current.
IOUT(max)
Iovercurrent(max) =
The RMS inductor current is used to calculate the I2·R losses
in the inductor.
135mV
RCS
The maximum power dissipated in the sense resistor is:
PD(R
2
SENSE )
= Iovercurrent(max)2 × RCS
MOSFET Selection
External N-channel logic-level power MOSFETs must be
used for the high- and low-side switches. The MOSFET gateto-source drive voltage of the MIC2182 is regulated by an
internal 5V VDD regulator. Logic-level MOSFETs, whose
operation is specified at VGS = 4.5V must be used.
It is important to note the on-resistance of a MOSFET
increases with increasing temperature. A 75°C rise in junction temperature will increase the channel resistance of the
MOSFET by 50% to 75% of the resistance specified at 25°C.
This change in resistance must be accounted for when
calculating MOSFET power dissipation.
Total gate charge is the charge required to turn the MOSFET
on and off under specified operating conditions (VDS and
VGS). The gate charge is supplied by the MIC2182 gate drive
circuit. At 300kHz switching frequency and above, the gate
Maximizing efficiency requires the proper selection of core
material and minimizing the winding resistance. The high
frequency operation of the MIC2182 requires the use of ferrite
materials for all but the most cost sensitive applications.
Lower cost iron powder cores may be used but the increase
in core loss will reduce the efficiency of the power supply. This
is especially noticeable at low output power. The winding
resistance decreases efficiency at the higher output current
levels. The winding resistance must be minimized although
this usually comes at the expense of a larger inductor.
The power dissipated in the inductor is equal to the sum of the
core and copper losses. At higher output loads, the core
losses are usually insignificant and can be ignored. At lower
M9999-042204
75mV
The maximum overcurrent threshold is:
IPK = IOUT(max) + 0.5 × IPP
1 IPP
Iinductor(rms) = IOUT(max) × 1 +
3 IOUT(max)
)
R winding(hot) = R winding(20°C) × 1 + 0.0042 × (Thot − T20°C )
16
April 22, 2004
MIC2182
Micrel
charge can be a significant source of power dissipation in the
MIC2182. At low output load this power dissipation is noticeable as a reduction in efficiency. The average current required to drive the high-side MOSFET is:
tT =
where:
IG[high-side](avg) =
average high-side MOSFET gate current
QG = total gate charge for the high-side MOSFET
taken from manufacturer’s data sheet
with VGS = 5V.
The low-side MOSFET is turned on and off at VDS = 0
because the freewheeling diode is conducting during this
time. The switching losses for the low-side MOSFET is
usually negligable. Also, the gate drive current for the lowside MOSFET is more accurately calculated using CISS at
VDS = 0 instead of gate charge.
For the low-side MOSFET:
PAC = (VIN +VD ) × IPK × t T × fS
where:
tT = switching transition time
(typically 20ns to 50ns)
VD = freewheeling diode drop, typically 0.5V.
fS it the switching frequency, nominally 300kHz
The low-side MOSFET switching losses are negligible and
can be ignored for these calculations.
RMS Current and MOSFET Power Dissipation Calculation
Under normal operation, the high-side MOSFET’s RMS
current is greatest when VIN is low (maximum duty cycle). The
low-side MOSFET’s RMS current is greatest when VIN is high
(minimum duty cycle). However, the maximum stress the
MOSFETs see occurs during short circuit conditions, where
the output current is equal to Iovercurrent(max). (See the Sense
Resistor section). The calculations below are for normal
operation. To calculate the stress under short circuit conditions, substitute Iovercurrent(max) for IOUT(max). Use the formula
below to calculate D under short circuit conditions.
IG[low-side](avg) = CISS × VGS × fS
Since the current from the gate drive comes from the input
voltage, the power dissipated in the MIC2182 due to gate
drive is:
(
)
A convenient figure of merit for switching MOSFETs is the onresistance times the total gate charge (RDS(on) × QG). Lower
numbers translate into higher efficiency. Low gate-charge
logic-level MOSFETs are a good choice for use with the
MIC2182. Power dissipation in the MIC2182 package limits
the maximum gate drive current. Refer to Figure 10 for the
MIC2182 gate drive limits.
Parameters that are important to MOSFET switch selection
are:
• Voltage rating
• On-resistance
• Total gate charge
The voltage rating of the MOSFETs are essentially equal to
the input voltage. A safety factor of 20% should be added to
the VDS(max) of the MOSFETs to account for voltage spikes
due to circuit parasitics.
The power dissipated in the switching transistor is the sum of
the conduction losses during the on-time (Pconduction) and the
switching losses that occur during the period of time when the
MOSFETs turn on and off (PAC).
Dshort circuit = 0.063 − 1.8 × 10 −3 × VIN
The RMS value of the high-side switch current is:
ISW(highside)(rms) =
I 2
D × IOUT(max)2 + PP
12
ISW(low side)(rms) =
(1− D) IOUT(max)2 +
IPP2
12
where:
D = duty cycle of the converter
D=
VOUT
η × VIN
η = efficiency of the converter.
Converter efficiency depends on component parameters,
which have not yet been selected. For design purposes, an
efficiency of 90% can be used for VIN less than 10V and 85%
can be used for VIN greater than 10V. The efficiency can be
more accurately calculated once the design is complete. If the
assumed efficiency is grossly inaccurate, a second iteration
through the design procedure can be made.
For the high-side switch, the maximum dc power dissipation
is:
PSW = Pconduction + PAC
where:
Pconduction = ISW(rms)2 × RSW
PAC = PAC(off) + PAC(on)
RSW = on-resistance of the MOSFET switch.
Making the assumption the turn-on and turnoff transition
times are equal, the transition time can be approximated by:
April 22, 2004
IG
where:
CISS and COSS are measured at VDS = 0.
IG = gate drive current (1A for the MIC2182)
The total high-side MOSFET switching loss is:
IG[high-side](avg) = QG × fS
Pgate drive = VIN IG[high-side](avg) + IG[low-side](avg)
CISS × VGS + COSS × VIN
Pswitch1(dc) = RDS(on)1 × ISW1(rms)2
17
M9999-042204
MIC2182
Micrel
For the low-side switch (N-channel MOSFET), the dc power
dissipation is:
circuit inductance will cause ringing during the high-side
MOSFET turn-on.
An external Schottky diode conducts at a lower forward
voltage preventing the body diode in the MOSFET from
turning on. The lower forward voltage drop dissipates less
power than the body diode. The lack of a reverse recovery
mechanism in a Schottky diode causes less ringing and less
power loss. Depending on the circuit components and operating conditions, an external Schottky diode will give a 1/2%
to 1% improvement in efficiency. Figure 12 illustrates the
difference in noise on the VSW pin with and without a
Schottky diode.
Output Capacitor Selection
The output capacitor values are usually determined by the
capacitors ESR (equivalent series resistance). Voltage rating
and RMS current capability are two other important factors in
selecting the output capacitor. Recommended capacitors are
tantalum, low-ESR aluminum electrolytics, and OS-CON.
The output capacitor’s ESR is usually the main cause of
output ripple. The maximum value of ESR is calculated by:
Pswitch2(dc) = RDS(on)2 × ISW 2(rms)2
Since the ac switching losses for the low side MOSFET is
near zero, the total power dissipation is:
Plow-side MOSFET(max) = Pswitch2(dc)
The total power dissipation for the high-side MOSFET is:
PhighsideMOSFET(max) = PSWITCH 1(dc) + PAC
External Schottky Diode
An external freewheeling diode is used to keep the inductor
current flow continuous while both MOSFETs are turned off.
This dead time prevents current from flowing unimpeded
through both MOSFETs and is typically 80ns The diode
conducts twice during each switching cycle. Although the
average current through this diode is small, the diode must be
able to handle the peak current.
ID(avg) = IOUT × 2 × 80ns × fS
The reverse voltage requirement of the diode is:
RESR ≤
Vdiode(rrm) = VIN
IPP
where:
VOUT = peak to peak output voltage ripple
IPP = peak to peak inductor ripple current
The total output ripple is a combination of the ESR and the
output capacitance. The total ripple is calculated below:
The power dissipated by the Schottky diode is:
Pdiode = ID(avg) × VF
where:
VF = forward voltage at the peak diode current
The external Schottky diode, D2, is not necessary for circuit
operation since the low-side MOSFET contains a parasitic
body diode. The external diode will improve efficiency and
decrease high frequency noise. If the MOSFET body diode is
used, it must be rated to handle the peak and average current.
The body diode has a relatively slow reverse recovery time
and a relatively high forward voltage drop. The power lost in
the diode is proportional to the forward voltage drop of the
diode. As the high-side MOSFET starts to turn on, the body
diode becomes a short circuit for the reverse recovery period,
dissipating additional power. The diode recovery and the
2
∆VOUT =
IPP × (1− D)
C
+ IPP × RESR
OUT × fS
(
)2
WITHOUT
WITH
FREEWHEELING DIODE FREEWHEELING DIODE
where:
D = duty cycle
COUT = output capacitance value
fS = switching frequency
The voltage rating of capacitor should be twice the output
voltage for a tantalum and 20% greater for an aluminum
electrolytic or OS-CON.
The output capacitor RMS current is calculated below:
IC
OUT
(rms) =
IPP
12
The power dissipated in the output capacitor is:
PDISS(C
OUT )
= IC
OUT
(rms)2 × RESR(C
OUT )
Input Capacitor Selection
The input capacitor should be selected for ripple current
rating and voltage rating. Tantalum input capacitors may fail
when subjected to high inrush currents, caused by turning the
input supply on. Tantalum input capacitor voltage rating
should be at least 2 times the maximum input voltage to
maximize reliability. Aluminum electrolytic, OS-CON, and
multilayer polymer film capacitors can handle the higher
inrush currents without voltage derating.
TIME
Figure 12. Switch Output Noise
With and Without Shottky Diode
M9999-042204
∆VOUT
18
April 22, 2004
MIC2182
Micrel
The input voltage ripple will primarily depend on the input
capacitors ESR. The peak input current is equal to the peak
inductor current, so:
• Supply current to the MIC2182
• MOSFET gate-charge power (included in the IC
supply current)
• Core losses in the output inductor
To maximize efficiency at light loads:
• Use a low gate-charge MOSFET or use the
smallest MOSFET, which is still adequate for
maximum output current.
• Allow the MIC2182 to run in skip mode at lower
currents.
• Use a ferrite material for the inductor core, which
has less core loss than an MPP or iron power
core.
Under heavy output loads the significant contributors to
power loss are (in approximate order of magnitude):
• Resistive on-time losses in the MOSFETs
• Switching transition losses in the MOSFETs
• Inductor resistive losses
• Current-sense resistor losses
• Input capacitor resistive losses (due to the
capacitors ESR)
To minimize power loss under heavy loads:
• Use logic-level, low on-resistance MOSFETs.
Multiplying the gate charge by the on-resistance
gives a Figure of merit, providing a good balance between low and high load efficiency.
• Slow transition times and oscillations on the
voltage and current waveforms dissipate more
power during turn-on and turnoff of the
MOSFETs. A clean layout will minimize parasitic
inductance and capacitance in the gate drive
and high current paths. This will allow the fastest
transition times and waveforms without oscillations. Low gate-charge MOSFETs will transition
faster than those with higher gate-charge
requirements.
• For the same size inductor, a lower value will
have fewer turns and therefore, lower winding
resistance. However, using too small of a value
will require more output capacitors to filter the
output ripple, which will force a smaller bandwidth, slower transient response and possible
instability under certain conditions.
• Lowering the current-sense resistor value will
decrease the power dissipated in the resistor.
However, it will also increase the overcurrent
limit and will require larger MOSFETs and
inductor components.
• Use low-ESR input capacitors to minimize the
power dissipated in the capacitors ESR.
Decoupling Capacitor Selection
The 4.7µF decoupling capacitor is used to minimize noise on
the VDD pin. The placement of this capacitor is critical to the
proper operation of the IC. It must be placed right next to the
∆VIN = Iinductor(peak) × RESR(C
IN )
The input capacitor must be rated for the input current ripple.
The RMS value of input capacitor current is determined at the
maximum output current. Assuming the peak to peak inductor ripple current is low:
IC (rms) ≈ IOUT(max) ×
IN
D × (1− D)
The power dissipated in the input capacitor is:
PDISS(C ) = IC (rms)2 × RESR(C )
IN
IN
IN
Voltage Setting Components
The MIC2182-3.3 and MIC2182-5.0 ICs contain internal
voltage dividers that set the output voltage. The MIC2182
adjustable version requires two resistors to set the output
voltage as shown in Figure 13.
R1
Error
Amp
FB
7
R2
VREF
1.245V
MIC2182 [adj.]
Figure 13. Voltage-Divider Configuration
The output voltage is determined by the equation:
R1
VO = VREF × 1 +
R2
Where: VREF for the MIC2182 is typically 1.245V.
A typical value of R1 can be between 3k and 10k. If R1 is too
large it may allow noise to be introduced into the voltage
feedback loop. If R1 is too small in value it will decrease the
efficiency of the power supply, especially at low output loads.
Once R1 is selected, R2 can be calculated using:
R2 =
VREF × R1
VO − VREF
Voltage Divider Power Dissipation
The reference voltage and R2 set the current through the
voltage divider.
V
Idivider = REF
R2
The power dissipated by the divider resistors is:
Pdivider = (R1+ R2) × Idivider 2
Efficiency Calculation and Considerations
Efficiency is the ratio of output power to input power. The
difference is dissipated as heat in the buck converter. Under
light output load, the significant contributors are:
April 22, 2004
19
M9999-042204
MIC2182
Micrel
pins and routed with a wide trace. The capacitor should be a
good quality tantalum. An additional 1µF ceramic capacitor
may be necessary when driving large MOSFETs with high
gate capacitance. Incorrect placement of the VDD decoupling
capacitor will cause jitter or oscillations in the switching
waveform and large variations in the overcurrent limit.
A 0.1µF ceramic capacitor is required to decouple the VIN.
The capacitor should be placed near the IC and connected
directly to between pin 10 (Vcc) and pin 12 (PGND).
PCB Layout and Checklist
PCB layout is critical to achieve reliable, stable and efficient
performance. A ground plane is required to control EMI and
minimize the inductance in power, signal and return paths.
The following guidelines should be followed to insure proper
operation of the circuit.
• Signal and power grounds should be kept
separate and connected at only one location.
Large currents or high di/dt signals that occur
when the MOSFETs turn on and off must be
kept away from the small signal connections.
• The connection between the current-sense
resistor and the MIC2182 current-sense inputs
(pin 8 and 9) should have separate traces,
routed from the terminals directly to the IC pins.
The traces should be routed as closely as
possible to each other and their length should be
minimized. Avoid running the traces under the
inductor and other switching components. A 1nF
to 0.1µF capacitor placed between pins 8 and 9
will help attenuate switching noise on the current
sense traces. This capacitor should be placed
close to pins 8 and 9.
M9999-042204
• When the high-side MOSFET is switched on, the
critical flow of current is from the input capacitor
through the MOSFET, inductor, sense resistor,
output capacitor, and back to the input capacitor.
These paths must be made with short, wide
pieces of trace. It is good practice to locate the
ground terminals of the input and output capacitors close to each.
• When the low-side MOSFET is switched on,
current flows through the inductor, sense
resistor, output capacitor, and MOSFET. The
source of the low-side MOSFET should be
located close to the output capacitor.
• The freewheeling diode, D1 in Figure 2, conducts current during the dead time, when both
MOSFETs are off. The anode of the diode
should be located close to the output capacitor
ground terminal and the cathode should be
located close to the input side of the inductor.
• The 4.7µF capacitor, which connects to the VDD
terminal (pin 11) must be located right at the IC.
The VDD terminal is very noise sensitive and
placement of this capacitor is very critical.
Connections must be made with wide trace. The
capacitor may be located on the bottom layer of
the board and connected to the IC with multiple
vias.
• The VIN bypass capacitor should be located
close to the IC and connected between pins 10
and 12. Connections should be made with a
ground and power plane or with short, wide
trace.
20
April 22, 2004
MIC2182
Micrel
Predesigned Circuits
A single schematic diagram, shown in Figure 14, can be used
to build power supplies ranging from 3A to 10A at the common
output voltages of 1.8V, 2.5V, 3.3V, and 5V. Components that
vary, depending upon output current and voltage, are listed
in the accompanying Tables 3 through 6.
MIC2182
VIN
VIN
VDD
C5
0.1µF
R7
100k
BST
EN/UVLO HSD
PWM
C6
0.1µF
C9
4.7µF
16V
C11
(table)
Q2
(table)
Q1
(table)
LSD
SS
R1
2k
D2
SD103BWS
L1
(table)
R2
(table)
VOUT
VSW
C4
1nF
C3
0.1µF
Power supplies larger than 10A can also be constructed
using the MIC2182 using larger power-handling components.
The “Power Supply Operating Characteristics” graphs following the component and vendor tables provide useful information about the actual performance of some of these circuits.
D1
(table)
C7
(table)
PGND
COMP
CSH
SYNC
VOUT
C12
0.1µF
50V
GND
C13, 1nF
VREF
C2
2.2nF
SGND
GND
C1
0.1µF
50V
Figure 14. Basic Circuit Diagram for Use with Tables 3 through 6
Specification
Limit
Switching frequency ripple
1% of output voltage
Maximum ambient temperature
85°C
Short-circuit capability
Continuous
Switching frequency
300kHz
Table 1. Specifications for Figure 14 and Tables 3 through 6
Manufacturer
Telephone Number (USA)
Web Address
AVX
(803) 946-0690
www.avxcorp.com
Central Semiconductor
(516) 435-1110
www.centralsemi.com
Coiltronics
(561) 241-7876
www.coiltronics.com
IRC
(704) 264-8861
IR
(310) 322-3331
www.irf.com
Micrel
(408) 944-0800
www.micrel.com
Vishay/Lite On
(diodes)
(805) 446-4800
www.vishay-liteon.com
Vishay/Siliconix
(MOSFETs)
(800) 554-5665
www.siliconix.com
Vishay/Dale
(inductors and resistors)
(800) 487-9437
www.vishaytechno.com
Sumida
(847) 956-0666
www.japanlink.com/sumida
Table 2. Component Suppliers
April 22, 2004
21
M9999-042204
MIC2182
Micrel
3A (6.5V–30V)
Part No. / Description
4A (6.5V–30V)
Part No. / Description
5A (6.5V–30V)
Part No. / Description
10A (6.5V–10V)
Part No. / Description
C7
qty: 2
TPSE227M010R0100
AVX, 220µF 10V,
0.1Ω ESR,
output filter capacitor
qty: 2
TPSE227M010R0100
AVX, 220µF 10V,
0.1Ω ESR,
output filter capacitor
qty: 2
TPSV227M010R0060
AVX, 220µF 10V,
0.06Ω ESR,
output filter capacitor
qty: 2
TPSV337M010R0060
AVX, 330µF 10V,
0.06Ω ESR,
output filter capacitor
C11
qty: 2
TPSE226M035R0300
AVX, 22µF 35V,
0.3Ω ESR,
input filter capacitor
qty: 3
TPSE226M035R0300
AVX, 22µF 35V,
0.3Ω ESR,
input filter capacitor
qty: 4
TPSE226M035R0300
AVX, 22µF 35V,
0.3Ω ESR,
input filter capacitor
qty: 4
TPSV107M020R0085
AVX, 100µF 20V,
0.06Ω ESR,
input filter capacitor
D1
qty: 1 B140, Vishay,
freewheeling diode
qty: 1 B140, Vishay,
freewheeling diode
qty: 1 B140, Vishay,
freewheeling diode
qty: 1 B330, Vishay,
freewheeling diode
L1
qty: 1 CDRH125-100,
Sumida Inductor,
10µH 4A,
output inductor
qty: 1 CDRH127-100,
Sumida Inductor,
10µH 5A,
output inductor
qty: 1 CDRH127-100
Sumida,
10µH 5A,
output inductor
qty: 1 UP4B-3R3,
Coiltronics,
3.3µH 11A,
output inductor
Q1
qty: 1 Si4800, Siliconix,
low-side MOSFET
qty: 1 Si4800, Siliconix,
low-side MOSFET
qty: 1 Si4884, Siliconix,
low-side MOSFET
qty: 2 Si4884, Siliconix
low-side MOSFET
Q2
qty: 1 Si4800, Siliconix,
high-side MOSFET
qty: 1 Si4800, Siliconix,
high-side MOSFET
qty: 1 Si4884, Siliconix,
high-side MOSFET
qty: 2 Si4884, Siliconix,
high-side MOSFET
R2
qty: 1
WSL-2010 .025 1%,
Vishay, 0.025, 1%, 0.5W,
current sense resistor
qty: 1
WSL-2010 .020 1%,
Vishay, 0.02, 1%, 0.5W,
current sense resistor
qty: 1
WSL-2512 .015 1%,
Vishay, 0.015, 1%, 1W,
current sense resistor
qty: 2
WSL-2512 .015 1% ,
Vishay, 0.015, 1%, 1W,
current sense resistor
U1
MIC2182-5.0BSM or
MIC2182-5.0BM
MIC2182-5.0BSM or
MIC2182-5.0BM
MIC2182-5.0BSM or
MIC2182-5.0BM
MIC2182-5.0BM
Reference
Table 3. Components for 5V Output
3A (4.5V–30V)
Part No. / Description
4A (4.5V–30V)
Part No. / Description
5A (4.5V–30V)
Part No. / Description
10A (4.5V–5.5V)
Part No. / Description
C7
qty: 2
TPSE227M010R0100
AVX, 220µF 10V,
0.1Ω ESR,
output filter capacitor
qty: 2
TPSE227M010R0100
AVX, 220µF 10V,
0.1Ω ESR,
output filter capacitor
qty: 2
TPSV227M010R0060
AVX, 220µF 10V,
0.06Ω ESR,
output filter capacitor
qty: 2
TPSV477M006R0055
AVX, 470µF 6.3V,
0.055Ω ESR,
output filter capacitor
C11
qty: 2
TPSE226M035R0300
AVX, 22µF 35V,
0.3Ω ESR,
input filter capacitor
qty: 2
TPSE226M035R0300
AVX, 22µF 35V,
0.3Ω ESR,
input filter capacitor
qty: 3
TPSE226M035R0300
AVX, 22µF 35V,
0.3Ω ESR,
input filter capacitor
qty: 3
TPSV227M016R0075
AVX, 220µF 16V,
0.075Ω ESR,
filter capacitor
D1
qty: 1 B140, Vishay,
freewheeling diode
qty: 1 B140, Vishay,
freewheeling diode
qty: 1 B140, Vishay,
freewheeling diode
qty: 1 B330, Vishay,
freewheeling diode
L1
qty: 1 CDRH125-100,
Sumida Inductor,
10µH 4A,
output inductor
qty: 1 CDRH127-100,
Sumida Inductor,
10µH 5A,
output inductor
qty: 1 CDRH127-100
Sumida,
10µH 5A,
output inductor
qty: 1 UP4B-3R3,
Coiltronics,
3.3µH 11A,
output inductor
Q1
qty: 1 Si4800, Siliconix,
low-side MOSFET
qty: 1 Si4800, Siliconix,
low-side MOSFET
qty: 1 Si4800, Siliconix,
low-side MOSFET
qty: 2 Si4884, Siliconix,
low-side MOSFET
Q2
qty: 1 Si4800, Siliconix,
high-side MOSFET
qty: 1 Si4800, Siliconix,
high-side MOSFET
qty: 1 Si4884, Siliconix,
high-side MOSFET
qty: 2 Si4884, Siliconix,
high-side MOSFET
R2
qty: 1
WSL-2010 .025 1%,
Vishay, 0.025, 1%, 0.5W,
current sense resistor
qty: 1
WSL-2010 .020 1%,
Vishay, 0.02, 1%, 0.5W,
current sense resistor
qty: 1
WSL-2512 .015 1%,
Vishay, 0.015, 1%, 1W,
current sense resistor
qty: 2
WSL-2512 .015 1% ,
Vishay, 0.015, 1%, 1W,
current sense resistor
U1
MIC2182-3.3BSM or
MIC2182-3.3BM
MIC2182-3.3BM or
MIC2182-3.3BSM
MIC2182-3.3BM or
MIC2182-3.3BSM
MIC2182-3.3BM
Reference
Table 4. Components for 3.3V Output
M9999-042204
22
April 22, 2004
MIC2182
Micrel
3A (4.5V–30V)
Part No. / Description
4A (4.5V–30V)
Part No. / Description
5A (4.5V–30V)
Part No. / Description
10A (4.5V–5.5V)
Part No. / Description
C7
qty: 2
TPSE227M010R0100
AVX, 220µF 10V,
0.1Ω ESR,
output filter capacitor
qty: 2
TPSE227M010R0100
AVX, 220µF 10V,
0.1Ω ESR,
output filter capacitor
qty: 2
TPSV227M010R0060
AVX, 220µF 10V,
0.06Ω ESR,
output filter capacitor
qty: 2
TPSV447M006R0055
AVX, 470µF 6.3V,
0.06Ω ESR,
output filter capacitor
C11
qty: 2
TPSE226M035R0300
AVX, 22µF 35V,
0.3Ω ESR,
input filter capacitor
qty: 2
TPSE226M035R0300
AVX, 22µF 35V,
0.3Ω ESR,
input filter capacitor
qty: 2
TPSE226M035R0300
AVX, 22µF 35V,
0.3Ω ESR,
input filter capacitor
qty: 3
TPSV227M016R0075
AVX, 220µF 16V,
0.06Ω ESR,
input filter capacitor
D1
qty: 1 B140, Vishay,
freewheeling diode
qty: 1 B140, Vishay,
freewheeling diode
qty: 1 B140, Vishay,
freewheeling diode
qty: 1 B330, Vishay,
freewheeling diode
L1
qty: 1 CDRH125-100,
Sumida Inductor,
10µH 4A,
output inductor
qty: 1 CDRH127-100,
Sumida Inductor,
10µH 5A,
output inductor
qty: 1 CDRH127-100
Sumida,
10µH 5A,
output inductor
qty: 1 UP4B-3R3,
Coiltronics,
3.3µH 11A,
output inductor
Q1
qty: 1 Si4800, Siliconix,
low-side MOSFET
qty: 1 Si4884, Siliconix,
low-side MOSFET
qty: 1 Si4884, Siliconix,
low-side MOSFET
qty: 2 Si4884, Siliconix
low-side MOSFET
Q2
qty: 1 Si4800, Siliconix,
high-side MOSFET
qty: 1 Si4800, Siliconix,
high-side MOSFET
qty: 1 Si4800, Siliconix,
high-side MOSFET
qty: 2 Si4884, Siliconix,
high-side MOSFET
R2
qty: 1
WSL-2010 .025 1%,
Vishay, 0.025, 1%, 0.5W,
current sense resistor
qty: 1
WSL-2010 .020 1%,
Vishay, 0.02, 1%, 0.5W,
current sense resistor
qty: 1
WSL-2512 .015 1%,
Vishay, 0.015, 1%, 1W,
current sense resistor
qty: 1
WSL-2512 .015 1% ,
Vishay, 0.015, 1%, 1W,
current sense resistor
U1
MIC2182BSM or
MIC2182BM
MIC2182BSM or
MIC2182BM
MIC2182BSM or
MIC2182BM
MIC2182BM
Reference
Table 5. Components for 2.5V Output
3A (4.5V–30V)
Part No. / Description
4A (4.5V–30V)
Part No. / Description
5A (4.5V–8V)
Part No. / Description
10A (4.5V–5.5V)
Part No. / Description
C7
qty: 2
TPSE227M010R0100
AVX, 220µF 10V,
0.1Ω ESR,
output filter capacitor
qty: 2
TPSE227M010R0100
AVX, 220µF 10V,
0.1Ω ESR,
output filter capacitor
qty: 2
TPSV227M010R0060
AVX, 220µF 10V,
0.06Ω ESR,
output filter capacitor
qty: 2
TPSV447M006R0055
AVX, 470µF 6.3V,
0.06Ω ESR,
output filter capacitor
C11
qty: 2
TPSE226M035R0300
AVX, 22µF 35V,
0.3Ω ESR,
input filter capacitor
qty: 2
TPSE226M035R0300
AVX, 22µF 35V,
0.3Ω ESR,
input filter capacitor
qty: 2
TPSE226M035R0300
AVX, 22µF 35V,
0.3Ω ESR,
input filter capacitor
qty: 2
TPSV227M016R0075
AVX, 220µF 16V,
0.06Ω ESR,
input filter capacitor
D1
qty: 1 B140, Vishay,
freewheeling diode
qty: 1 B140, Vishay,
freewheeling diode
qty: 1 B140, Vishay,
freewheeling diode
qty: 1 B330, Vishay,
freewheeling diode
L1
qty: 1 CDRH125-100,
Sumida Inductor,
10µH 4A,
output inductor
qty: 1 CDRH127-100,
Sumida Inductor,
10µH 5A,
output inductor
qty: 1 CDRH127-100
Sumida,
10µH 5A,
output inductor
qty: 1 UP4B-3R3,
Coiltronics,
3.3µH 11A,
output inductor
Q1
qty: 1 Si4800, Siliconix,
low-side MOSFET
qty: 1 Si4884, Siliconix,
low-side MOSFET
qty: 1 Si4884, Siliconix,
low-side MOSFET
qty: 2 Si4884, Siliconix
low-side MOSFET
Q2
qty: 1 Si4800, Siliconix,
high-side MOSFET
qty: 1 Si4800, Siliconix,
high-side MOSFET
qty: 1 Si4800, Siliconix,
high-side MOSFET
qty: 2 Si4884, Siliconix,
high-side MOSFET
R2
qty: 1
WSL-2010 .025 1%,
Vishay, 0.025, 1%, 0.5W,
current sense resistor
qty: 1
WSL-2010 .020 1%,
Vishay, 0.02, 1%, 0.5W,
current sense resistor
qty: 1
WSL-2512 .015 1%,
Vishay, 0.015, 1%, 1W,
current sense resistor
qty: 2
WSL-2512 .015 1% ,
Vishay, 0.015, 1%, 1W,
current sense resistor
U1
MIC2182BSM or
MIC2182BM
MIC2182BSM or
MIC2182BM
MIC2182BSM or
MIC2182BM
MIC2182BM
Reference
Table 6. Components for 1.8V Output
April 22, 2004
23
M9999-042204
MIC2182
Micrel
Power Supply Operating Characteristics
Effect of Soft-Start Capacitor (CSS) Value
On Output Voltage Waveforms
During Turn-On
(4A Power Supply Configuration)
Normal (300kHz Switching Frequency) and
Output Short-Circuit (60kHz) Conditions
Switch Node (Pin 15) Waveforms
Converter Waveforms
PIN 16
IL1
(2A/div)
VGS
LOW-SIDE
MOSFET
VGS
HIGH-SIDE
MOSFET
VSW+HSD
VSW
PIN 15
Effect of Soft-Start Capacitor (CSS) Value
On Output Voltage Waveforms
During Turn-On
(10A Power Supply Configuration)
HIGH-SIDE
DRIVE VOLTAGE
REFERENCED TO GROUND
HIGH-SIDE MOSFET
GATE-TO-SOURCE VOLTAGE
LOW-SIDE MOSFET
GATE-TO-SOURCE VOLTAGE
INDUCTOR CURRENT
VIN = 7V
L1 = 3.3µH
VOUT = 3.3V
IOUT = 10A
QTY: 2
Si4884
HIGH-SIDE
MOSFETS
QTY: 2
Si4884
LOW-SIDE
MOSFETS
10Amps
Typical PWM-Mode Waveforms
IL1
(0.5A/div)
IL1
(0.5A/div)
VSW
Pin 15
VSW
Pin 15
VOUT
VOUT
Typical Skip-Mode Waveforms
M9999-042204
SWITCH-NODE
VOLTAGE
24
April 22, 2004
MIC2182
Micrel
Load Transient Response
and Bode Plot
(10A Power Supply Configuration)
VOUT
VOUT
Load Transient Response
and Bode Plot
(4A Power Supply Configuration)
VIN = 12V
VOUT = 3.3V
L1 = 10µH
R2 = 20mΩ
90
300x103
0
100x103
-40
10x103
30
1x103
60
100x100
0
-20
EFFICIENCY (%)
120
PHASE
20
PHASE (°)
40
10x100
GAIN (dB)
GAIN
80
180
60
60
40 VIN = 5V
R2 = 15mΩ
L1 = 10µH
20
1 high-side MOSFET: Si4800
1 low-side MOSFET: Si4800
0
0.01
0.1
1
4
OUTPUT CURRENT (A)
40
120
20
90
0
60
PHASE
-20
-40
FREQUENCY (Hz)
30
0
FREQUENCY (Hz)
12V Efficiency
(4A Power Supply Configuration)
24V Efficiency
(4A Power Supply Configuration)
Efficiency
(10A Power Supply Configuration)
100
100
100
80
80
60
40 VIN = 12V
R2 = 15mΩ
L1 = 10µH
20
1 high-side MOSFET: Si4800
1 low-side MOSFET: Si4800
0
0.01
0.1
1
4
OUTPUT CURRENT (A)
April 22, 2004
Skip
PWM
PWM
Skip
60
40 VIN = 24V
R2 = 15mΩ
L1 = 10µH
20
1 high-side MOSFET: Si4800
1 low-side MOSFET: Si4800
0
0.01
0.1
1
4
OUTPUT CURRENT (A)
25
EFFICIENCY (%)
80 Skip
EFFICIENCY (%)
EFFICIENCY (%)
150
GAIN
PHASE (°)
PWM
80
210
300x103
150
100
100x103
180
60
Skip
GAIN (dB)
80
100
10x103
210
Bode Plot
(10A Power Supply Configuration)
10x100
100
5V Efficiency
(4A Power Supply Configuration)
100x100
Bode Plot
(4A Power Supply Configuration)
1x103
IOUT
2A/div
IOUT
5A/div
VIN = 6V
VOUT = 3.3V
L1 = 3.3µH
R2 = 7.5mΩ
PWM
60
40
20
R2 = 7.5mΩ
L1 = 3.3µH
2 high-side MOSFETs: Si4884
2 low-side MOSFETs: Si4884
0
0.01
0.1
1
OUTPUT CURRENT (A)
10
M9999-042204
MIC2182
Micrel
Package Information
PIN 1
0.157 (3.99)
0.150 (3.81)
DIMENSIONS:
INCHES (MM)
0.020 (0.51)
REF
0.050 (1.27)
BSC
0.0648 (1.646)
0.0434 (1.102)
0.020 (0.51)
0.013 (0.33) 0.0098 (0.249)
0.0040 (0.102)
0.394 (10.00)
0.386 (9.80)
SEATING
PLANE
45°
0°–8°
0.050 (1.27)
0.016 (0.40)
0.244 (6.20)
0.228 (5.79)
16-pin SOP (M)
5.40 (0.213)
5.20 (0.205)
7.90 (0.311)
7.65 (0.301)
DIMENSIONS:
MM (INCH)
0.875
(0.034) REF
6.33 (0.239)
6.07 (0.249)
0.38 (0.015)
0.25 (0.010)
2.00 (0.079)
1.73 (0.068)
0.21 (0.008)
0.05 (0.002)
0.65 (0.0260) COPLANARITY:
0.10 (0.004) MAX
BSC
10°
4°
0°
–8°
0.22 (0.009)
0.13 (0.005)
1.25 (0.049) REF
0.95 (0.037)
0.55 (0.022)
16-Pin SSOP (SM)
M9999-042204
26
April 22, 2004
MIC2182
April 22, 2004
Micrel
27
M9999-042204
MIC2182
Micrel
MICREL, INC. 1849 FORTUNE DRIVE SAN JOSE, CA 95131
TEL
+ 1 (408) 944-0800
FAX
+ 1 (408) 474-1000
WEB
USA
http://www.micrel.com
The information furnished by Micrel in this data sheet is believed to be accurate and reliable. However, no responsibility is assumed by Micrel for its use.
Micrel reserves the right to change circuitry and specifications at any time without notification to the customer.
Micrel Products are not designed or authorized for use as components in life support appliances, devices or systems where malfunction of a product can
reasonably be expected to result in personal injury. Life support devices or systems are devices or systems that (a) are intended for surgical implant into
the body or (b) support or sustain life, and whose failure to perform can be reasonably expected to result in a significant injury to the user. A Purchaser’s
use or sale of Micrel Products for use in life support appliances, devices or systems is at Purchaser’s own risk and Purchaser agrees to fully indemnify
Micrel for any damages resulting from such use or sale.
© 2004 Micrel, Incorporated.
M9999-042204
28
April 22, 2004
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