AD1671SQ

a FEATURES Conversion Time: 800 ns 1.25 MHz Throughput Rate Complete: On-Chip Sample-and-Hold Amplifier and Voltage Reference Low Power Dissipation: 570 mW No Missing Codes Guaranteed Signal-to-Noise Plus Distortion Ratio fIN = 100 kHz: 70 dB Pin Configurable Input Voltage Ranges Twos Complement or Offset Binary Output Data 28-Pin DIP and 28-Pin Surface Mount Package Out of Range Indicator AIN1 AIN2 Complete 12-Bit 1.25 MSPS Monolithic A/D Converter AD1671 FUNCTIONAL BLOCK DIAGRAM SHA OUT 5k UPO/BPO ENCODE VCC ACOM VEE VLOGIC DCOM S/H RANGE SELECT 5k X4 3-BIT FLASH DAC 3-BIT FLASH DAC COARSE 4-BIT FLASH 4 8-BIT LADDER MATRIX 3 REF IN REF OUT 2.5V REF 3 CORRECTION LOGIC 8 FINE 4-BIT FLASH 4 LATCHES AD1671 REF COM OTR MSB 12 BIT 1 –12 DAV PRODUCT DESCRIPTION The AD1671 is a monolithic 12-bit, 1.25 MSPS analog-todigital converter with an on-board, high performance sampleand-hold amplifier (SHA) and voltage reference. The AD1671 guarantees no missing codes over the full operating temperature range. The combination of a merged high speed bipolar/ CMOS process and a novel architecture results in a combination of speed and power consumption far superior to previously available hybrid implementations. Additionally, the greater reliability of monolithic construction offers improved system reliability and lower costs than hybrid designs. The fast settling input SHA is equally suited for both multiplexed systems that switch negative to positive full-scale voltage levels in successive channels and sampling inputs at frequencies up to and beyond the Nyquist rate. The AD1671 provides both reference output and reference input pins, allowing the on-board reference to serve as a system reference. An external reference can also be chosen to suit the dc accuracy and temperature drift requirements of the application. The AD1671 uses a subranging flash conversion technique, with digital error correction for possible errors introduced in the first part of the conversion cycle. An on-chip timing generator provides strobe pulses for each of the four internal flash cycles. A single ENCODE pulse is used to control the converter. The digital output data is presented in twos complement or offset binary output format. An out-of-range signal indicates an overflow condition. It can be used with the most significant bit to determine low or high overflow. The performance of the AD1671 is made possible by using high speed, low noise bipolar circuitry in the linear sections and low power CMOS for the logic sections. Analog Devices’ ABCMOS-1 process provides both high speed bipolar and 2-micron CMOS devices on a single chip. Laser trimmed thin-film resistors are used to provide accuracy and temperature stability. The AD1671 is available in two performance grades and three temperature ranges. The AD1671J and K grades are available over the 0°C to +70°C temperature range. The AD1671A grade is available over the –40°C to +85°C temperature range. The AD1671S grade is available over the –55°C to +125°C temperature range. PRODUCT HIGHLIGHTS The AD1671 offers a complete single chip sampling 12-bit, 1.25 MSPS analog-to-digital conversion function in a 28-pin package. The AD1671 at 570 mW consumes a fraction of the power of currently available hybrids. An OUT OF RANGE output bit indicates when the input signal is beyond the AD1671’s input range. Input signal ranges are 0 V to +5 V unipolar or ± 5 V bipolar, selected by pin strapping, with an input resistance of 10 kΩ. The input signal range can also be pin strapped for 0 V to +2.5 V unipolar or ± 2.5 V bipolar with an input resistance of 10 MΩ. Output data is available in unipolar, bipolar offset or bipolar twos complement binary format. REV. B Information furnished by Analog Devices is believed to be accurate and reliable. However, no responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other rights of third parties which may result from its use. No license is granted by implication or otherwise under any patent or patent rights of Analog Devices. One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A. Tel: 617/329-4700 Fax: 617/326-8703 AD1671–SPECIFICATIONS DC SPECIFICATIONS (T Parameter RESOLUTION CONVERSION TIME ACCURACY Integral Nonlinearity (INL) (S Grade) Differential Nonlinearity (DNL) No Missing Codes Unipolar Offsets1 (+25°C) Bipolar Zero1 (+25°C) Gain Error1, 2 (+25°C) TEMPERATURE COEFFICIENTS3 Unipolar Offset (S Grade) Bipolar Zero (S Grade) Gain Error3 (S Grade) Gain Error4 POWER SUPPLY REJECTION5 VCC (+5 V ± 0.25 V) (S Grade) VLOGIC (+5 V ± 0.25 V) (S Grade) VEE (–5 V ± 0.25 V) (S Grade) ANALOG INPUT Input Ranges Bipolar Unipolar Input Resistance (0 V to +2.5 V or ± 2.5 V Range) (0 V to +5.0 V or ± 5 V Range) Input Capacitance Aperture Delay Aperture Jitter INTERNAL VOLTAGE REFERENCE Output Voltage Output Current Unipolar Mode Bipolar Mode LOGIC INPUTS High Level Input Voltage, VIH Low Level Input Voltage, VIL High Level Input Current, IIH (VIN = VLOGIC) Low Level Input Current, ILL (VIN = 0 V) Input Capacitance, CIN LOGIC OUTPUTS High Level Output Voltage, VOH (IOH = 0.5 mA) Low Level Output Voltage, VOL (IOL = 1.6 mA) POWER SUPPLIES Operating Voltages VCC VLOGIC VEE Operating Current ICC ILOGIC6 IEE POWER CONSUMPTION TEMPERATURE RANGE (SPECIFIED) J/K A S MIN to TMAX with VCC = +5 V Min 12 5%, VLOGIC = +5 V AD1671J/A/S Typ 10%, VEE = –5 V Max 800 Min 12 5%, unless otherwise noted) AD1671K Typ Max 800 ± 0.7 ± 2.5 Units Bits ns LSB Bits ±9 ± 10 0.35 ± 25 ± 25 ± 30 ± 20 ±4 ±4 ±4 LSB LSB % FSR ppm/°C ppm/°C ppm/°C ppm/°C LSB LSB LSB ± 1.5 11 11 Bits Guaranteed 0.1 ± 2.5 ± 3.0 ±9 ± 10 0.35 ± 25 ± 25 ± 25 ± 30 ± 30 ± 40 ± 20 ±4 ±5 ±4 ±5 ±4 ±5 12 12 Bits Guaranteed 0.1 –2.5 –5.0 0 0 8 10 10 10 15 20 2.5 +2.5 +5.0 +2.5 +5.0 12 –2.5 –5.0 0 0 8 10 10 10 15 20 2.5 +2.5 +5.0 +2.5 +5.0 12 Volts Volts Volts Volts MΩ kΩ pF ns ps Volts mA mA Volts Volts µA µA pF Volts Volts 2.475 2.525 +2.5 +1.0 2.475 2.525 +2.5 +1.0 2.0 –10 –10 5 2.4 0.4 0.8 +10 +10 2.0 –10 –10 5 2.4 0.4 0.8 +10 +10 +4.75 +4.5 –4.75 55 3 –55 570 0 –40 –55 +5.25 +5.5 –5.25 68 5 –68 750 +70 +85 +125 +4.75 +4.5 –4.75 55 3 –55 570 0 –40 –55 +5.25 +5.5 –5.25 68 5 –68 750 +70 +85 +125 Volts Volts Volts mA mA mA mW °C °C °C NOTES 1 Adjustable to zero with external potentiometers. 2 Includes internal voltage reference error. 3 +25°C to TMIN and +25°C to TMAX 4 Excludes internal reference drift. 5 Change in gain error as a function of the dc supply voltage. 6 Tested under static conditions. See Figure 15 for typical curve of ILOGIC vs. load capacitance at maximum tC. Specifications subject to change without notice. –2– REV. B AD1671 AC SPECIFICATIONS f Parameter SIGNAL-TO-NOISE PLUS DISTORTION RATIO (S/N + D) –0.5 dB Input –20 dB Input EFFECTIVE NUMBER OF BITS (ENOB) TOTAL HARMONIC DISTORTION (THD) PEAK SPURIOUS OR PEAK HARMONIC COMPONENT SMALL SIGNAL BANDWIDTH FULL POWER BANDWIDTH INTERMODULATION DISTORTION (IMD) 2nd Order Products 3rd Order Products 2 (TMIN to TMAX with VCC = +5 V 5%, VLOGIC = +5 V 1 lNPUT = 1OO kHz, unless otherwise noted) Min 10%, VEE = –5 V 5%, fSAMPLE = 1 MSPS, AD1671K Typ Max AD1671J/A/S Typ Max Min Units 68 11.2 70 50 68 11.2 71 51 dB dB Bits –80 –80 12 2 –80 –85 –75 –77 –83 –81 12 2 –75 –77 dB dB MHz MHz –75 –75 –80 –85 –75 –75 dB dB NOTES 1 fIN amplitude = –0.5 dB (9.44 V p-p) bipolar mode full scale unless otherwise indicated. All measurements referred to a 0 dB ( ± 5 V) input signal, unless otherwise indicated. 2 fA = 99 kHz, fB = 100 kHz with fSAMPLE = 1 MSPS. Specifications subject to change without notice. SWITCHING SPECIFICATIONS V Parameters (For all grades TMIN to TMAX with VCC = +5 V 5%, VLO61C = +5 V 10%, 5%; VIL = 0.8 V, VIH = 2.0 V, VOL = 0.4 V and VOH = 2.4 V) EE = –5 V Symbol tC FS tENC tENCL tDAV tF tR tDD1 tSS2 Min Typ Max 800 1.25 50 300 Units ns MSPS ns ns ns ns ns ns ns Conversion Time Sample Rate ENCODE Pulse Width High (Figure 1a) ENCODE Pulse Width Low (Figure 1b) DAV Pulse Width ENCODE Falling Edge Delay Start New Conversion Delay Data and OTR Delay from DAV Falling Edge Data and OTR Valid before DAV Rising Edge 20 20 150 0 0 20 20 75 75 NOTES 1 tDD is measured from when the falling edge of DAV crosses 0.8 V to when the output crosses 0.4 V or 2.4 V with a 25 pF load capacitor on each output pin. 2 tSS is measured from when the outputs cross 0.4 V or 2.4 V to when the rising edge of DAV crosses 2.4 V with a 25 pF load capacitor on each output pin. Specifications subject to change without notice. t ENC ENCODE tC tC t DAV t ENCL tR ENCODE tF t DAV DAV tR DAV t DD BIT 1–12 MSB, OTR DATA 0 (PREVIOUS) t SS DATA 1 t DD BIT 1–12 MSB, OTR DATA 0 (PREVIOUS) t SS DATA 1 Figure 1a. Encode Pulse HIGH Figure 1b. Encode Pulse LOW REV. B –3– AD1671 PIN DESCRIPTION Symbol ACOM AIN Pin No. 27 22, 23 Type P AI Name and Function Analog Ground. Analog Inputs, AIN1 and AIN2. The AD1671 can be pin strapped for four input ranges: Range 0 to +2.5 V, ± 2.5 V 0 to +5 V, ± 5 V Pin Strap Connect AIN1 to AIN2 Connect AIN1 or AIN2 to ACOM Signal Input AIN1 or AIN2 AIN1 or AIN2 BIT 1 (MSB) BIT 12 (LSB) BPO/UPO DAV 13 2 26 16 DO DO DO AI DO Most Significant Bit. Data Bits 2 through 11. Least Significant Bit. Bipolar or Unipolar Configuration Pin. See section on Input Range Connections for details. Data Available Output. The rising edge of DAV indicates an end of conversion and can be used to latch current data into an external register. The falling edge of DAV can be used to latch previous dam into an external register. Digital Ground. The analog input is sampled on the rising edge of ENCODE. Inverted Most Significant Bit. Provides twos complement output data format. Out of Range is Active HIGH when the analog input is out of range. See Output Data Format, Table III. REF COM is the internal reference ground pin. REF COM should be connected as indicated in the Grounding and Decoupling Rules and Optional External Reference Connection Sections. REF IN is the external 2.5 V reference input. REF OUT is the internal 2.5 V reference output. No Connect for bipolar input ranges. Connect SHA OUT to BPO/UPO for unipolar input ranges. +5 V Analog Power. –5 V Analog Power. +5 V Digital Power. BIT 2–BIT 11 12-3 DCOM ENCODE MSB OTR REF COM REF IN REF OUT SHA OUT VCC VEE VLOGIC 19 17 14 15 20 24 21 25 28 1 18 P DI DO DO AI AI AO AO P P P TYPE: AI = Analog Input; AO = Analog Output; DI = Digital Input; DO = Digital Outputs; P = Power. PIN CONFIGURATION VEE BIT 12 (LSB) BIT 11 BIT 10 BIT 9 BIT 8 BIT 7 BIT 6 BIT 5 28 VCC 27 26 25 24 23 ACOM BPO/UPO SHA OUT REF IN AIN1 AIN2 REF OUT 1 2 3 4 5 6 7 8 9 AD1671 TOP VIEW (Not to Scale) 22 21 20 REF COM 19 DCOM 18 VLOGIC 17 ENCODE 16 DAV BIT 4 10 BIT 3 11 BIT 2 12 BIT 1 (MSB) 13 MSB 14 15 OTR –4– REV. B AD1671 ABSOLUTE MAXIMUM RATINGS* ORDERING GUIDE Parameter With Respect to Min –0 5 –6.5 –0.5 –1.0 –6.5 –0.5 –0.5 –11.0 –0.5 –65 Max +6.5 +0.5 +6.5 +1.0 +6.5 VLOGIC + 0.5 VCC + 0.5 +11.0 VCC + 0.5 +150 +150 +300 Units Volts Volts Volts Volts Volts Volts Volts Volts Volts °C °C °C Model1 AD1671JQ AD1671KQ AD1671JP AD1671KP AD1671AQ AD1671AP AD1671SQ Linearity ± 2.5 LSB ± 2 LSB ± 2.5 LSB ± 2 LSB ± 2.5 LSB ± 2.5 LSB ± 3 LSB VCC ACOM VEE ACOM VLOGIC DCOM ACOM DCOM VCC VLOGIC ENCODE DCOM REF IN ACOM AIN ACOM BPO/UPO ACOM Junction Temperature Storage Temperature Lead Temperature (10 sec) Temperature Range 0°C to +70°C 0°C to +70°C 0°C to +70°C 0°C to +70°C –40°C to +85°C –40°C to +85°C –55°C to +125°C Package Option2, 3 Q-28 Q-28 P-28A P-28A Q-28 P-28A Q-28 *Stresses above those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. This is a stress rating only and functional operation of the device at these or any other conditions above those indicated in the operational sections of this specification is not implied. Exposure to absolute maximum ratings for extended periods may effect device reliability. NOTES 1 For details on grade and package offerings screened in accordance with MIL-STD-883, refer to Analog Devices’ Military Products Databook or current AD1671/883 data sheet. 2 P = Plastic Leaded Chip Carrier, Q = Cerdip. 3 Analog Devices reserves the right to ship side brazed ceramic packages in lieu of cerdip. CAUTION ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily accumulate on the human body and test equipment and can discharge without detection. Although the AD1671 features proprietary ESD protection circuitry, permanent damage may occur on devices subjected to high energy electrostatic discharges. Therefore, proper ESD precautions are recommended to avoid performance degradation or loss of functionality. WARNING! ESD SENSITIVE DEVICE REV. B –5– AD1671 DEFINITIONS OF SPECIFICATIONS INTEGRAL NONLINEARITY (INL) DYNAMIC SPECIFICATIONS SIGNAL-TO-NOISE PLUS DISTORTION (S/ N+D) RATIO Integral nonlinearity refers to the deviation of each individual code from a line drawn from “zero” through “full scale.” The point used as “zero” occurs 1/2 LSB (1.22 mV for a 10 V span) before the first code transition (all zeros to only the LSB on). “Full-scale” is defined as a level 1 1/2 LSB beyond the last code transition (to all ones). The deviation is measured from the low side transition of each particular code to the true straight line. DIFFERENTIAL LINEARITY ERROR (NO MISSING CODES) S/N+D is the ratio of the rms value of the measured input signal to the rms sum of all other spectral components, including harmonics but excluding dc. The value for S/N+D is expressed in decibels. EFFECTIVE NUMBER OF BITS (ENOB) ENOB is calculated from the expression (S/N+D) = 6.02N + 1.76 dB, where N is equal to the effective number of bits. TOTAL HARMONIC DISTORTION (THD) An ideal ADC exhibits code transitions that are exactly 1 LSB apart. DNL is the deviation from the ideal value. Thus every code has a finite width. Guaranteed no missing codes to 11- or 12-bit resolution indicates that all 2048 and 4096 codes, respectively, must be present over all operating ranges. No missing codes to 11 bits (in the case of a 12-bit resolution ADC) also means that no two consecutive codes are missing. UNIPOLAR OFFSET THD is the ratio of the rms sum of the first six harmonic components to the rms value of the measured input signal and is expressed as a percentage or in decibels. INTERMODULATION DISTORTION (IMD) The first transition should occur at a level 1/2 LSB above analog common. Unipolar offset is defined as the deviation of the actual from that point. This offset can be adjusted as discussed later. The unipolar offset temperature coefficient specifies the maximum change of the transition point over temperature, with or without external adjustments. BIPOLAR ZERO In the bipolar mode the major carry transition (0111 1111 1111 to 1000 0000 0000) should occur for an analog value 1/2 LSB below analog common. The bipolar offset error and temperature coefficient specify the initial deviation and maximum change in the error over temperature. GAIN ERROR With inputs consisting of sine waves at two frequencies, fa and fb, any device with nonlinearities will create distortion products of order (m + n), at sum and difference frequencies of mfa ± nfb, where m, n = 0, 1, 2, 3. . . . Intermodulation terms are those for which m or n is not equal to zero. For example, the second order terms are (fa + fb) and (fa – fb), and the third order terms are (2 fa + fb), (2 fa – fb), (fa + 2 fb) and (2fb – fa). The IMD products are expressed as the decibel ratio of the rms sum of the measured input signals to the rms sum of the distortion terms. The two signals are of equal amplitude and the peak value of their sum is –0.5 dB from full scale. The IMD products are normalized to a 0 dB input signal. PEAK SPURIOUS OR PEAK HARMONIC COMPONENT The peak spurious or peak harmonic component is the largest spectral component, excluding the input signal and dc. This value is expressed in decibels relative to the rms value of a fullscale input signal. APERTURE DELAY The last transition (from 1111 1111 1110 to 1111 1111 1111) should occur for an analog value 1 1/2 LSB below the nominal full scale (4.9963 volts for 5.000 volts full scale). The gain error is the deviation of the actual level at the last transition from the ideal level. The gain error can be adjusted to zero as shown in Figures 4 through 7. TEMPERATURE COEFFICIENTS Aperture delay is the difference between thc switch delay and the analog delay of the SHA. This delay represents the point in time, relative to the rising edge of ENCODE input, that the analog input is sampled. APERTURE JITTER The temperature coefficients for unipolar offset, bipolar zero and gain error specify the maximum change from the initial (+25°C) value to the value at TMIN or TMAX. POWER SUPPLY REJECTION Aperture jitter is the variation in aperture delay for successive samples. FULL POWER BANDWIDTH One of the effects of power supply error on the performance of the device will be a small change in gain. The specifications show the maximum full-scale change from the initial value with the supplies at the various limits. The input frequency at which the amplitude of the reconstructed fundamental is reduced by 3 dB for a full-scale input. –6– REV. B AD1671 THEORY OF OPERATION The AD1671 uses a successive subranging architecture. The analog-to-digital conversion takes place in four independent steps or flashes. The sampled analog input signal is subranged to an intermediate residue voltage for the final 12-bit result by utilizing multiple flashes with subtraction DACs (see the AD1671 functional block diagram). The AD1671 can be configured to operate with unipolar (0 V to +5 V, 0 V to +2.5 V) or bipolar (± 5 V, ± 2.5 V) inputs by connecting AIN (Pins 22, 23), SHA OUT (Pin 25) and BPO/UPO (Pin 26) as shown in Figure 2. 0 TO +2.5V AIN1 5k SHA AIN2 5k –2.5V TO +2.5V AIN1 5k SHA AIN2 5k an input range that is configured with one bit of overlap with the previous DAC. The overlap allows for errors during the flash conversion. The first residue voltage is connected to the second 3-bit flash and to the noninverting input of a high speed, differential, gain of eight amplifier. The second flash result is passed to the correction logic register and to the second segmented current output DAC. The output of the second DAC is connected to the inverting input of the differential amplifier. The differential amplifier output is connected to a two-step, backend, 8-bit flash. This 8-bit flash consists of coarse and fine flash converters. The result of the coarse 4-bit flash converter, also configured to overlap one bit of DAC 2, is connected to the correction logic register and selects one of 16 resistors from which the fine 4-bit flash will establish its span voltage. The fine 4-bit flash is connected directly to the output latches. The internal timing generator automatically places the SHA into the acquire mode when DAV goes LOW. Upon completion of conversion (when DAV is set HIGH), the SHA has acquired the analog input to the specified level of accuracy and will remain in the sample mode until the next ENCODE command. The AD1671 will flag an out-of-range condition when the input voltage exceeds the analog input range. OTR (Pin 15) is active HIGH when an out-of-range high or low condition exists. Bits 1–12 are HIGH when the analog input voltage is greater than the selected input range and LOW when the analog input is less than the selected input range. AD1671 DYNAMIC PERFORMANCE SHA OUT SHA OUT AD1671 BPO/UPO AD1671 BPO/UPO REF IN REF IN REF OUT REF OUT a. 0 V to +2.5V Input Range 0 TO +5V AIN1 5k SHA AIN2 5k ±5V b. ± 2.5 V Input Range AIN1 5k SHA AIN2 5k SHA OUT SHA OUT AD1671 BPO/UPO AD1671 BPO/UPO REF IN REF IN REF OUT REF OUT c. 0 V to +5 V Input Range d. ± 5 V Input Range The AD1671 is specified for dc and dynamic performance. A sampling converter’s dynamic performance reflects both quantizer and sample-and-hold amplifier (SHA) performance. Quantizer nonlinearities, such as INL and DNL, can degrade dynamic performance. However, a SHA is the critical element which has to accurately sample fast slewing analog input signals. The AD1671’s high performance, low noise, patented on-chip SHA minimizes distortion and noise specifications. Nonlinearities are minimized by using a fast slewing, low noise architecture and subregulation of the sampling switch to provide constant offsets (therefore reducing input signal dependent nonlinearities). Figure 3 is a typical 2k point Fast Fourier Transform (FFT) plot of a 100 kHz input signal sampled at 1 MHz. The fundamental amplitude is set at –0.5 dB to avoid input signal clipping of offset or gain errors. Note the total harmonic distortion is approximately –81 dB, signal to noise plus distortion is 71 dB and the spurious free dynamic range is 84 dB. 0 Figure 2. AD1671 Input Range Connections SIGNAL AMPLITUDE – dB The AD1671 conversion cycle begins by simply providing an active HIGH level on the ENCODE pin (Pin 17). The rising edge of the ENCODE pulse starts the conversion. The falling edge of the ENCODE pulse is specified to operate within a window of time, less than 50 ns after the rising edge of ENCODE or after the falling edge of DAV. The time window prevents digitally coupled noise from being introduced during the final stages of conversion. An internal timing generator circuit accurately controls SHA, flash and DAC timing. Upon receipt of an ENCODE command the input voltage is held by the front-end SHA and the first 3-bit flash converts the analog input voltage. The 3-bit result is passed to a correction logic register and a segmented current output DAC. The DAC output is connected through a resistor (within the Range/Span Select Block) to SHA OUT. A residue voltage is created by subtracting the DAC output from SHA OUT, which is less than one eighth of the full-scale analog input. The second flash has –25 –50 –75 –100 0 FREQUENCY Figure 3. AD1671 FFT Plot, fIN = 100 kHz, fSAMPLE = 1 MHz REV. B –7– AD1671 Figure 4 plots both S/(N+D) and Effective Number of Bits (ENOB) for a 100 kHz input signal sampled from 666 kHz to 1.25 MHz. 72.5 72 71.5 11.75 SPURIOUS FREE DYNAMIC RANGE – dB 85 80 75 70 65 60 55 50 45 40 –50 –45 –40 –35 –30 –25 –20 –15 –10 ANALOG INPUT – dB 71 70.5 70 69.5 69 68.5 68 666 714 769 833 909 1000 1111 11.50 11.25 EFFECTIVE NUMBER OF BITS S/(N+D) – dB –5 0 11.00 1250 Figure 7. Spurious Free Dynamic Range vs. Input Amplitude, fIN = 250 kHz APPLYING THE AD1671 GROUNDING AND DECOUPLING RULES SAMPLING FREQUENCY – kHz Figure 4. S/(N/D) vs. Sampling Frequency, fIN = 100 kHz Figure 5 is a THD plot for a full-scale 100 kHz input signal with the sample frequency swept from 666 kHz to 1.25 MHz. –68 –70 –72 –74 THD – dB –76 –78 –80 –82 –84 –86 666 714 769 833 909 1000 1111 1250 SAMPLING FREQUENCY – kHz Figure 5. THD vs. Sampling Rate, fIN = 100 kHz Proper grounding and decoupling should be a primary design objective in any high speed, high resolution system. The AD1671 separates analog and digital grounds to optimize the management of analog and digital ground currents in a system. The AD1671 is designed to minimize the current flowing from REF COM (Pin 20) by directing the majority of the current from VCC (+5 V–Pin 28) to VEE (–5 V–Pin 1). Minimizing analog ground currents hence reduces the potential for large ground voltage drops. This can be especially true in systems that do not utilize ground planes or wide ground runs. REF COM is also configured to be code independent, therefore reducing input dependent analog ground voltage drops and errors. Code dependent ground current is diverted to ACOM (Pin 27). Also critical in any high speed digital design is the use of proper digital grounding techniques to avoid potential CMOS “ground bounce.” Figure 3 is provided to assist in the proper layout, grounding and decoupling techniques. +5V 0.1µF 10µF 0.1µF –5V 10µF +5V 0.1µF 10µF The AD1671’s SFDR performance is ideal for use in communication systems such as high speed modems and digital radios. The SFDR is better than 84 dB with sample rates up to 1.11 MHz and increases as the input signal amplitude is attenuated by approximately 3 dB. Note also the SFDR is typically better than 80 dB with input signals attenuated by up to –7 dB. SPURIOUS FREE DYNAMIC RANGE – dB –68 –70 –72 –74 –76 –78 –80 –82 –84 –86 –88 –90 666 714 769 833 909 1000 SAMPLING FREQUENCY – kHz 1111 1250 28 VCC 23 AIN1 1 18 VEE VLOGIC BIT 1 13 AD1671 VIN (±5V) 22 AIN2 20 REF COM 27 ACOM 19 DCOM 25 SHA OUT OTR 15 26 BPO/UPO MSB 14 24 REF IN 21 REF OUT BIT 12 2 AGP* DGP* ENCODE 17 DAV 16 Figure 6. Spurious Free Dynamic Range vs. Sampling Rate, fIN = 100 kHz 1µF *GROUND PLANE RECOMMENDED Figure 8. AD1671 Grounding and Decoupling –8– REV. B AD1671 Table I is a list of grounding and decoupling rules that should be reviewed before laying out a printed circuit board. Table I. Grounding and Decoupling Guidelines The gain trim is done by applying a signal 1 1/2 LSBs below the nominal full scale (4.998 V for a 5 V range). Trim R2 to give the last transition (1111 1111 1110 to 1111 1111 1111). This circuit will give approximately ± 0.5% FS of adjustment range. VIN 0 TO +5V +5V OFFSET R1 ADJ 10k –5V GAIN ADJ 50k 25Ω AIN1 Power Supply Decoupling Capacitor Values Comment 0.1 µF (Ceramic) and 1 µF (Tantalum) Surface Mount Chip Capacitors Recommended to Reduce Lead Inductance Directly at Positive and Negative Supply Pins to Common Ground Plane 5k SHA 5k AIN2 R2 50Ω SHA OUT AD1671 BPO/UPO REF IN Capacitor Locations Reference (REF OUT) Capacitor Value Grounding Analog Ground Ground Plane or Wide Ground Return Connected to the Analog Power Supply Critical Common Connections Should be Star Connected to REF COM (as Shown in Figure 8) Ground Plane or Wide Ground Return Connected to the Digital Power Supply 1 µF (Tantalum) to ACOM 1µF REF OUT Figure 9. Unipolar (0 V to +5 V) Calibration BIPOLAR ( 5 V) CALIBRATION The connections for the bipolar ± 5 V input range is shown in Figure 10. VIN –5V TO +5V +5V OFFSET R1 ADJ 10k –5V GAIN ADJ 50k Reference Ground (REF COM) Digital Ground 25Ω AIN1 5k SHA 5k AIN2 Analog and Digital Ground Connected Together Once at the AD1671 UNIPOLAR (0 V TO +5 V) CALIBRATION R2 50Ω SHA OUT AD1671 BPO/UPO REF IN The AD1671 is factory trimmed to minimize offset, gain and linearity errors. In some applications the offset and gain errors of the AD1671 need to be externally adjusted to zero. This is accomplished by trimming the voltage at AIN2 (Pin 22). The circuit in Figure 9 is recommended for calibrating offset and gain errors of the AD1671 when configured in the 0 V to +5 V input range. If the offset trim resistor R1 is used, it should be trimmed as follows, although a different offset can be set for a particular system requirement. This circuit will give approximately ± 5 mV of offset trim range. Nominally the AD1671 is intended to have a 1/2 LSB offset so that the exact analog input for a given code will be in the middle of that code (halfway between the transitions to the codes above it and below it). Thus, the first transition (from 0000 0000 0000 to 0000 0000 0001) will occur for an input level of +1/2 LSB (0.61 mV for 5 V range). REF OUT 1µF Figure 10. Bipolar (± 5 V) Calibration Bipolar calibration is similar to unipolar calibration. First, a signal 1/2 LSB above negative full scale (–4.9988 V) is applied and R1 is trimmed to give the first transition (0000 0000 0000 to 0000 0000 0001). Then a signal 1 1/2 LSB below positive full scale (+4.9963 V) is applied and R2 is trimmed to give the last transition (1111 1111 1110 to 1111 1111 1111). REV. B –9– AD1671 UNIPOLAR (0 V TO +2.5 V) CALIBRATION The connections for the 0 V to +2.5 V input range calibration is shown in Figure 11. Figure 11 shows an example of how the offset error can be trimmed in front of the AD1671. The procedure for trimming the offset and gain errors is the same as for the unipolar 5 V range. +15V 390Ω OFFSET ADJ R1 1kΩ GAIN R2 ADJ 2k AIN1 5k SHA AIN2 1k SHA OUT 5k and tSS minimum). To satisfy the requirements of the 574 type latch the recommended logic families are S, AS, ALS, F or BCT. New data from the AD1671 is latched on the rising edge of the DAV (Pin 16) output pulse. Previous data can be latched by inverting the DAV output with a 7404 type inverter. 74HC574 BIT 1 BIT 2 BIT 3 BIT 4 BIT 5 BIT 6 BIT 7 BIT 8 DAV 1D 2D 3D 4D 5D 6D 7D 8D CLOCK 1Q 2Q 3Q 4Q 5Q 6Q 7Q 8Q OC DATA BUS VIN 0 TO +2.5V AD845 10k 74HC574 BIT 9 BIT 10 BIT 11 BIT 12 1D 2D 3D 4D 5D 6D 7D 8D CLOCK 1Q 2Q 3Q 4Q 5Q 6Q 7Q 8Q OC 10k AD1671 BPO/UPO REF IN AD1671 REF OUT 1µF 3-STATE CONTROL Figure 13. AD1671 to Output Latches OUT OF RANGE Figure 11. Unipolar (0 V to +2.5 V) Calibration The connections for the bipolar ± 2.5 V input range is shown in Figure 12. +15V 390Ω OFFSET ADJ R1 1kΩ GAIN R2 ADJ 2k 10k 1k SHA OUT AIN1 5k SHA AIN2 5k BIPOLAR ( 2.5 V) CALIBRATION VIN –2.5V TO +2.5V AD845 An out-of-range condition exists when the analog input voltage is beyond the input range (0 V to +2.5 V, 0 V to +5 V, ± 2.5 V, ± 5 V) of the converter OTR (Pin 15) is set low when the analog input voltage is within the analog input range. OTR is set HIGH and will remain HIGH when the analog input voltage exceeds the input range by typically 1/2 LSB (OTR transition is tested to ± 6 LSBs of accuracy) from the center of the ± full-scale output codes. OTR will remain HIGH until the analog input is within the input range and another conversion is completed. By logical ANDing OTR with the MSB and its complement, overrange high or underrange low conditions can be detected. Table II is a truth table for the over/under range circuit in Figure 14. Systems requiring programmable gain conditioning prior to the AD1671 can immediately detect an out-of-range condition, thus eliminating gain selection iterations. Table II. Out-of-Range Truth Table 10k AD1671 BPO/UPO REF IN OTR 0 0 1 1 MSB MSB 0 1 0 1 Analog Input Is In Range In Range Underrange Overrange REF OUT 1µF Figure 12. Bipolar (± 2.5 V) Calibration OUTPUT LATCHES Figure 13 shows the AD1671 connected to the 74HC574 octal D-type edge-triggered latches with 3-state outputs. The latch can drive highly capacitive loads (i.e., bus lines, I/O ports) while maintaining the data signal integrity. The maximum setup and hold times of the 574 type latch must be less than 20 ns (tDD OVER = "1" OTR UNDER = "1" MSB Figure 14. Overrange or Underrange Logic –10– REV. B AD1671 Table III. Output Data Format Input Range 0 V to +2.5 V Coding Straight Binary Analog Inputl ≤ –0.0003 V 0V +2.5 V ≥ +2.5003 V Digital Output 0000 0000 0000 0000 0000 0000 1111 1111 1111 1111 1111 1111 0000 0000 0000 0000 0000 0000 1111 1111 1111 1111 1111 1111 0000 0000 0000 0000 0000 0000 1111 1111 1111 1111 1111 1111 0000 0000 0000 0000 0000 0000 1111 1111 1111 1111 1111 1111 1000 0000 0000 1000 0000 0000 0111 1111 1111 0111 1111 1111 1000 0000 0000 1000 0000 0000 0111 1111 1111 0111 1111 1111 OTR2 1 0 0 1 1 0 0 1 1 0 0 1 1 0 0 1 1 0 0 1 1 0 0 1 0 V to +5 V Straight Binary ≤ –0.0006 V 0V +5 V ≥ +5.0006 V ≤ –2.5006 V –2.5 V +2.5 V ≥ +2.4994 V ≤ –5.0012 V –5 V +5 V ≥ +4.9988 V –2.5 V to +2.5 V Offset Binary –5 V to +5 V Offset Binary –2.5 V to +2.5 V Twos Complement ≤ –2.5006 V (Using MSB) –2.5 V +2.5 V ≥ +2.4994 V –5 V to +5 V Twos Complement ≤ –5.0012 V (Using MSB) –5 V +5 V ≥ +4.9988 V NOTES 1 Voltages listed are with offset and gain errors adjusted to zero. 2 Typical performance. OUTPUT DATA FORMAT The AD1671 provides both MSB and MSB outputs, delivering data in positive true straight binary for unipolar input ranges and positive true offset binary or twos complement for bipolar input ranges. Straight binary coding is used for systems that accept positive-only signals. If straight binary coding is used with bipolar input signals, a 0 V input would result in a binary output of 2048. The application software would have to subtract 2048 to determine the true input voltage. Host registers typically perform math on signed integers and assume data is in that format. Twos complement format minimizes software overhead which is especially important in high speed data transfers, such as a DMA operation. The CPU is not bogged down performing data conversion steps, hence the total system throughput is increased. OPTIONAL EXTERNAL REFERENCE connected to +5 V. It is possible to connect REF OUT to +5 V due to its output circuit implementation which shuts down the reference. ILOGIC VS. CONVERSION RATE Figure 15 is the typical logic supply current vs. conversion rate for various capacitor loads on the digital outputs. 6.5 6.0 5.5 5.0 4.5 4.0 mA 3.5 3.0 2.5 2.0 1.5 1.0 0.5 1k 10k 100k CONVERSION RATE – Hz CL = 30pF CL = 50pF The AD1671 includes an onboard +2.5 V reference. The reference input pin (REF IN) can be connected to reference output pin (REF OUT) or a standard external +2.5 V reference can be selected to meet specific system requirements. Fast switching input dependent currents are modulated at the reference input. The reference input voltage can be held with the use of a capacitor. To prevent the AD1671’s onboard reference from oscillating when not connected to REF IN, REF OUT must be CL = 0pF 1M Figure 15. ILOGIC vs. Conversion Rate for Various Capacitive Loads on the Digital Outputs REV. B –11– AD1671 APPLICATIONS AD1671 TO ADSP-2100A AD1671 TO ADSP-2101/2102 Figure 16 demonstrates the AD1671 to ADSP-2100A interface. The 2100A with a clock frequency of 12.5 MHz can execute an instruction in one 80 ns cycle. The AD1671 is configured to perform continuous time sampling. The DAV output of the AD1671 is asserted at the end of each conversion. DAV can be used to latch the conversion result into the two 574 octal D-latches. The falling edge of the sampling clock is used to generate an interrupt (IRQ3) for the processor. Upon interrupt, the ADSP-2100A starts a data memory read by providing an address on the DMA bus. The decoded address generates OE for the latches and the processor reads their output over the DMA bus. The conversion result is read within a single processor cycle. DMRD DMA0:13 ADDRESS BUS DECODE 8 Figure 17 is identical to the 2100A interface except the sampling clock is used to generate an interrupt (IRQ2) for the processor. Upon interrupt the ADSP-2100A starts a data memory read by providing an address on the address (A) bus. The decode address generates OE for the D-latches and the processor reads their output over the Data (D) bus. Reading the conversion result is thus completed within a single processor cycle. RD A0:13 ADDRESS BUS 8 DAV OE 574 Q0:7 8 D0:7 OE AD1671 ADSP-2101 DECODE 16 OE DAV BIT1:12 4 4 ENCODE D0:15 DATA BUS 8 574 D0:3 Q0:7 D0:7 574 Q0:7 8 AD1671 IRQ2 ADSP2100A DMA0:15 DMACK D0:7 SAMPLING CLOCK 16 DATA BUS 8 +5V SAMPLING CLOCK OE BIT1:12 4 574 D0:3 Q0:7 4 D0:7 Figure 17. AD1671 to ADSP-2101/ADSP-2102 Interface IRQ3 ENCODE Figure 16. AD1671 to ADSP-2100A Interface –12– REV. B AD1671 COMPONENT LIST Parts List Reference Designator R1, R2 R3, R4, R5 R6 R7 R8 R9, R11 R10 R12 R13 R14 R15–R28 C1, C3, C5 C2, C4, C6, C8, C10 C7, C9, C15, C16 C11, C12, C13, C14, C17 C18 C19–C22 C23 C24 U1 U2 U3 U4–U5 U6 W1–W3 J1–J15 S1 S2 S3 SW1–SW3 TP1, TP2, TP4–TP6 TP3, TP7, TP10, TP13 TP8, TP9, TP11, TP12, TP14 P1 Type Description Resistor, 5%, 0.5 W, 100 Ω Resistor, 1%, 49.9 Ω 100 Ω Trim Potentiometer Resistor 1%, 4.99 kΩ Optional X Ω Trim Potentiometer, Optional Resistor, 1%, 4.99 kΩ Resistor, 1%, 10 kΩ Resistor, 1%, 2.49 kΩ Resistor, 1%, 787 Ω Resistor, 1%, 249 Ω Resistor, 5%, 22 Ω Cap, Tantalum, 22 µF Cap, Ceramic, 0.01 µF Cap, Tantalum, 10 µF Cap, Ceramic, 0.1 µF Cap, Ceramic, 1.0 µF Cap, Ceramic, 0.1 µF Cap, Mica, 100 pF Cap, Ceramic, 0.001 µF 78L05 +5 V Regulator 79L05 –5 V Regulator AD1671 74HC573 Drivers AD568 BNC Jacks Jumpers and Headers Metal Binding Posts Wide 28-Pin Socket Narrow 20-Pin Socket Narrow 24-Pin Socket SECMA SPDT Switch Test Point, Red Test Point, Black Test Point, White 40-Pin Connector Male + Hooks REV. B –13– AD1671 Figure 18. AD1671/EB PCB Layout—Silkscreen Layer –14– REV. B AD1671 Figure 19. AD1671/EB PCB Layout—Component Side Figure 20. AD1671/EB PCB Layout—Solder Side REV. B –15– AD1671 OUTLINE DIMENSIONS Dimensions shown in inches and (mm). 28-Lead PLCC (P-28A) Package C1616a–10–10/93 28-Pin Cerdip (Q-28) Package –16– REV. B PRINTED IN U.S.A.