AD625ADZ

a FEATURES User Programmed Gains of 1 to 10,000 Low Gain Error: 0.02% Max Low Gain TC: 5 ppm/C Max Low Nonlinearity: 0.001% Max Low Offset Voltage: 25 V Low Noise 4 nV/√Hz (at 1 kHz) RTI Gain Bandwidth Product: 25 MHz 16-Lead Ceramic or Plastic DIP Package, 20-Terminal LCC Package Standard Military Drawing Available MlL-Standard Parts Available Low Cost Programmable Gain Instrumentation Amplifier AD625 FUNCTIONAL BLOCK DIAGRAM 50
–INPUT – + – AD625 + –GAIN SENSE 10k
SENSE –GAIN DRIVE 10k
VB +GAIN DRIVE – OUTPUT + 10k
REFERENCE +GAIN SENSE 50
+INPUT 10k
– + – + PRODUCT DESCRIPTION PRODUCT HIGHLIGHTS The AD625 is a precision instrumentation amplifier specifically designed to fulfill two major areas of application: 1) Circuits requiring nonstandard gains (i.e., gains not easily achievable with devices such as the AD524 and AD624). 2) Circuits requiring a low cost, precision software programmable gain amplifier. 1. The AD625 affords up to 16-bit precision for user selected fixed gains from 1 to 10,000. Any gain in this range can be programmed by 3 external resistors. For low noise, high CMRR, and low drift the AD625JN is the most cost effective instrumentation amplifier solution available. An additional three resistors allow the user to set any gain from 1 to 10,000. The error contribution of the AD625JN is less than 0.05% gain error and under 5 ppm/°C gain TC; performance limitations are primarily determined by the external resistors. Common-mode rejection is independent of the feedback resistor matching. A software programmable gain amplifier (SPGA) can be configured with the addition of a CMOS multiplexer (or other switch network), and a suitable resistor network. Because the ON resistance of the switches is removed from the signal path, an AD625 based SPGA will deliver 12-bit precision, and can be programmed for any set of gains between 1 and 10,000, with completely user selected gain steps. 2. A 12-bit software programmable gain amplifier can be configured using the AD625, a CMOS multiplexer and a resistor network. Unlike previous instrumentation amplifier designs, the ON resistance of a CMOS switch does not affect the gain accuracy. 3. The gain accuracy and gain temperature coefficient of the amplifier circuit are primarily dependent on the user selected external resistors. 4. The AD625 provides totally independent input and output offset nulling terminals for high precision applications. This minimizes the effects of offset voltage in gain-ranging applications. 5. The proprietary design of the AD625 provides input voltage noise of 4 nV/√Hz at 1 kHz. 6. External resistor matching is not required to maintain high common-mode rejection. For the highest precision the AD625C offers an input offset voltage drift of less than 0.25 µV/°C, output offset drift below 15 µV/°C, and a maximum nonlinearity of 0.001% at G = 1. All grades exhibit excellent ac performance; a 25 MHz gain bandwidth product, 5 V/µs slew rate and 15 µs settling time. The AD625 is available in three accuracy grades (A, B, C) for industrial (–40°C to +85°C) temperature range, two grades (J, K) for commercial (0°C to +70°C) temperature range, and one (S) grade rated over the extended (–55°C to +125°C) temperature range. REV. D Information furnished by Analog Devices is believed to be accurate and reliable. However, no responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other rights of third parties which may result from its use. No license is granted by implication or otherwise under any patent or patent rights of Analog Devices. One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A. Tel: 781/329-4700 World Wide Web Site: http://www.analog.com Fax: 781/326-8703 © Analog Devices, Inc., 2000 AD625–SPECIFICATIONS Model GAIN Gain Equation Gain Range Gain Error1 Nonlinearity, Gain = 1-256 Gain>256 Gain vs. Temp. Gain 5 + VB – 50A 50A +VS FD333 A1 A2 C3 10k
C4 FD333 SENSE +IN 2k
10k
GAIN DRIVE 50
Q1, Q3 –IN 50A GAIN DRIVE RF RF RG GAIN SENSE 10k
50
Q2, Q4 GAIN SENSE 10k
RF 2N5952 VO RG REF AD625 VOUT RF +IN –IN 2k
FD333 2N5952 50A FD333 –VS –VS Figure 26c. Input Protection Circuit Figure 25. Simplified Circuit of the AD625 –8– REV. D AD625 Any resistors in series with the inputs of the AD625 will degrade the noise performance. For this reason the circuit in Figure 26b should be used if the gains are all greater than 5. For gains less than 5, either the circuit in Figure 26a or in Figure 26c can be used. The two 1.4 kΩ resistors in Figure 26a will degrade the noise performance to: 4 kTRext +(4 nV/ Hz )2 = 7.9 nV / Hz 300 200 100 10k 20k 30k 40k 50k 60k FEEDBACK RESISTANCE –
RESISTOR PROGRAMMABLE GAIN AMPLIFIER In the resistor-programmed mode (Figure 27), only three external resistors are needed to select any gain from 1 to 10,000. Depending on the application, discrete components or a pretrimmed network can be used. The gain accuracy and gain TC are primarily determined by the external resistors since the AD625C contributes less than 0.02% to gain error and under 5 ppm/°C gain TC. The gain sense current is insensitive to common-mode voltage, making the CMRR of the resistor programmed AD625 independent of the match of the two feedback resistors, RF. RTO OFFSET VOLTAGE DRIFT RG +GAIN SENSE RTI NULL 1 16 2 15 3 RTO 14 NULL 4 13 +VS RTI NULL +GAIN DRIVE A1 5 A2 NC 6 REF –VS 8 10k
AD625 A3 10k
RTO NULL –GAIN DRIVE VOUT 10 9 +VS Figure 27. AD625 in Fixed Gain Configuration A list of standard resistors which can be used to set some common gains is shown in Table I. 20k
50k
10k 1 10 100 1k FEEDBACK RESISTANCE –
RF RG 20 kΩ 19.6 kΩ 20 kΩ 20 kΩ 20 kΩ 19.6 kΩ 20 kΩ 20.5 kΩ 19.6 kΩ 19.6 kΩ 20 kΩ 19.6 kΩ 20 kΩ 19.6 kΩ 20 kΩ 20 kΩ 19.6 kΩ 19.6 kΩ 19.6 kΩ ∞ 39.2 kΩ 10 kΩ 4.42 kΩ 2.1 kΩ 806 Ω 402 Ω 205 Ω 78.7 Ω 39.2 Ω 13.3 kΩ 5.62 kΩ 2.67 kΩ 1.27 kΩ 634 Ω 316 Ω 154 Ω 76.8 Ω 38.3 Ω SENSE TERMINAL The sense terminal is the feedback point for the AD625 output amplifier. Normally it is connected directly to the output. If heavy load currents are to be drawn through long leads, voltage drops through lead resistance can cause errors. In these instances the sense terminal can be wired to the load thus putting For single gain applications, only one offset null adjust is necessary; in these cases the RTI null should be used. REV. D FREQUENCY – Hz MULTIPLYING FACTOR 1 2 5 10 20 50 100 200 500 1000 4 8 16 32 64 128 256 512 1024 –GAIN SENSE 10k
2 GAIN 11 10k
7 12 3 10k
Table I. Common Gains Nominally Within 0.5% Error Using Standard 1% Resistors RF –INPUT 100k 4 BANDWIDTH Figure 28. RTO Noise, Offset, Drift and Bandwidth vs. Feedback Resistance Normalized to 20 kΩ 2RF +1 RG +INPUT 2 1M 1 10k 20k 30k 40k 50k 60k FEEDBACK RESISTANCE –
As previously stated each RF provides feedback to the input stage and sets the unity gain transconductance. These feedback resistors are provided by the user. The AD625 is tested and specified with a value of 20 kΩ for RF. Since the magnitude of RTO errors increases with increasing feedback resistance, values much above 20 kΩ are not recommended (values below 10 kΩ for RF may lead to instability). Refer to the graph of RTO noise, offset, drift, and bandwidth (Figure 28) when selecting the feedback resistors. The gain resistor (RG) is determined by the formula RG = 2 RF/(G – l). RF 5 3 10k 20k 30k 40k 50k 60k FEEDBACK RESISTANCE –
6 Selecting Resistor Values G= RTO OFFSET VOLTAGE MULTIPLYING FACTOR VOLTAGE NOISE – nV Hz RTO NOISE –9– AD625 the I × R drops “inside the loop” and virtually eliminating this error source. GND VDD VSS Typically, IC instrumentation amplifiers are rated for a full ± 10 volt output swing into 2 kΩ. In some applications, however, the need exists to drive more current into heavier loads. Figure 29 shows how a high-current booster may be connected “inside the loop” of an instrumentation amplifier. By using an external power boosting circuit, the power dissipated by the AD625 will remain low, thereby, minimizing the errors induced by selfheating. The effects of nonlinearities, offset and gain inaccuracies of the buffer are reduced by the loop gain of the AD625’s output amplifier. +VS A1 EN SENSE AD7502 VS 39k
SENSE AD589 VOUT –VS REFERENCE VREF +VS 1.2V RF RG AD625 –IN +VS VIN+ +IN A0 0.01F R3 20k
RFB AD625 X1 RF DATA INPUTS RI VIN– MSB LSB OUT 1 OUT 2 AD7524 8-BIT DAC CS REFERENCE +VS C1 1/2 AD712 R4 10k
1/2 AD712 WR –VS R5 2k
5k
–VS Figure 29. AD625 /Instrumentation Amplifier with Output Current Booster Figure 30. Software Controllable Offset REFERENCE TERMINAL An instrumentation amplifier can be turned into a voltage-tocurrent converter by taking advantage of the sense and reference terminals as shown in Figure 31. The reference terminal may be used to offset the output by up to ± 10 V. This is useful when the load is “floating” or does not share a ground with the rest of the system. It also provides a direct means of injecting a precise offset. However, it must be remembered that the total output swing is ± 10 volts, from ground, to be shared between signal and reference offset. VIN+ SENSE RF The AD625 reference terminal must be presented with nearly zero impedance. Any significant resistance, including those caused by PC layouts or other connection techniques, will increase the gain of the noninverting signal path, thereby, upsetting the common-mode rejection of the in-amp. Inadvertent thermocouple connections created in the sense and reference lines should also be avoided as they will directly affect the output offset voltage and output offset voltage drift. RG AD625 +VX– R1 RF VIN– IL AD711 LOAD Figure 31. Voltage-to-Current Converter In the AD625 a reference source resistance will unbalance the CMR trim by the ratio of 10 kΩ/RREF. For example, if the reference source impedance is 1 Ω, CMR will be reduced to 80 dB (10 kΩ/1 Ω = 80 dB). An operational amplifier may be used to provide the low impedance reference point as shown in Figure 30. The input offset voltage characteristics of that amplifier will add directly to the output offset voltage performance of the instrumentation amplifier. By establishing a reference at the “low” side of a current setting resistor, an output current may be defined as a function of input voltage, gain and the value of that resistor. Since only a small current is demanded at the input of the buffer amplifier A1, the forced current IL will largely flow through the load. Offset and drift specifications of A2 must be added to the output offset and drift specifications of the In-Amp. The circuit of Figure 30 also shows a CMOS DAC operating in the bipolar mode and connected to the reference terminal to provide software controllable offset adjustments. The total offset range is equal to ± (VREF/2 × R5/R4), however, to be symmetrical about 0 V R3 = 2 × R4. The offset per bit is equal to the total offset range divided by 2N, where N = number of bits of the DAC. The range of offset for Figure 30 is ± 120 mV, and the offset is incremented in steps of 0.9375 mV/LSB. INPUT AND OUTPUT OFFSET VOLTAGE Offset voltage specifications are often considered a figure of merit for instrumentation amplifiers. While initial offset may be adjusted to zero, shifts in offset voltage due to temperature variations will cause errors. Intelligent systems can often correct for this factor with an autozero cycle, but this requires extra circuitry. –10– REV. D AD625 Offset voltage and offset voltage drift each have two components: input and output. Input offset is that component of offset that is generated at the input stage. Measured at the output it is directly proportional to gain, i.e., input offset as measured at the output at G = 100 is 100 times greater than that measured at G = 1. Output offset is generated at the output and is constant for all gains. in distributed stray capacitances. In many applications shielded cables are used to minimize noise. This technique can create +VS +INPUT SENSE RF 100
AD711 The input offset and drift are multiplied by the gain, while the output terms are independent of gain, therefore, input errors dominate at high gains and output errors dominate at low gains. The output offset voltage (and drift) is normally specified at G = 1 (where input effects are insignificant), while input offset (and drift) is given at a high gain (where output effects are negligible). All input-related parameters are specified referred to the input (RTI) which is to say that the effect on the output is “G” times larger. Offset voltage vs. power supply is also specified as an RTI error. RF –INPUT REFERENCE –VS Figure 32. Common-Mode Shield Driver common-mode rejection errors unless the shield is properly driven. Figures 32 and 33 show active data guards which are configured to improve ac common-mode rejection by “bootstrapping” the capacitances of the input cabling, thus minimizing differential phase shift. By separating these errors, one can evaluate the total error independent of the gain. For a given gain, both errors can be combined to give a total error referred to the input (RTI) or output (RTO) by the following formula: +INPUT +VS AD712 100
SENSE RF Total Error RTI = input error + (output error/gain) AD625 RG Total Error RTO = (Gain × input error) + output error 100
The AD625 provides for both input and output offset voltage adjustment. This simplifies nulling in very high precision applications and minimizes offset voltage effects in switched gain applications. In such applications the input offset is adjusted first at the highest programmed gain, then the output offset is adjusted at G = 1. If only a single null is desired, the input offset null should be used. The most additional drift when using only the input offset null is 0.9 µV/°C, RTO. COMMON-MODE REJECTION Common-mode rejection is a measure of the change in output voltage when both inputs are changed by equal amounts. These specifications are usually given for a full-range input voltage change and a specified source imbalance. In an instrumentation amplifier, degradation of common-mode rejection is caused by a differential phase shift due to differences RF –VS –VS Figure 33. Differential Shield Driver GROUNDING In order to isolate low level analog signals from a noisy digital environment, many data-acquisition components have two or more ground pins. These grounds must eventually be tied together at one point. It would be convenient to use a single ground line, however, current through ground wires and pc runs of the circuit card can cause hundreds of millivolts of error. Therefore, separate ground returns should be provided to minimize the current flow from the sensitive points to the system ground (see Figure 34). Since the AD625 output voltage is developed with respect to the potential on the reference terminal, it can solve many grounding problems. STATUS AD583 SAMPLE AND HOLD AD625 –VS –VS +VS ANALOG OUT AD574A +VS HOLD CAP A/D CONVERTER VLOGIC +VS –VS +VS –VS DIGITAL COMMON ANALOG POWER GROUND Figure 34. Basic Grounding Practice for a Data Acquisition System REV. D VOUT REFERENCE –INPUT AD7502 INPUT SIGNAL VOUT AD625 RG –11– AD625 GROUND RETURNS FOR BIAS CURRENTS Input bias currents are those currents necessary to bias the input transistors of a dc amplifier. There must be a direct return path for these currents, otherwise they will charge external capacitances, causing the output to drift uncontrollably or saturate. Therefore, when amplifying “floating” input sources such as transformers, or ac-coupled sources, there must be a dc path from each input to ground as shown in Figure 35. high thermoelectric potential (about 35 µV°C). This means that care must be taken to insure that all connections (especially those in the input circuit of the AD625) remain isothermal. This includes the input leads (1, 16) and the gain sense lines (2, 15). These pins were chosen for symmetry, helping to desensitize the input circuit to thermal gradients. In addition, the user should also avoid air currents over the circuitry since slowly fluctuating GND VDD VSS +VS SENSE RF RG AD625 +VS VOUT 15 RF 16 LOAD AD7502 REFERENCE TO POWER SUPPLY GROUND –VS Figure 35a. Ground Returns for Bias Currents with Transformer Coupled Inputs 10 VOUT AD625 9 14 + VIN – 0.1F LOW LEAKAGE 13 1k
–VS 11 12 AD711 +VS SENSE RF RG AD625 VDD VOUT VSS RF LOAD AD7510DIKD GND REFERENCE 200s TO POWER SUPPLY GROUND –VS ZERO PULSE A1 A2 A3 A4 Figure 36. Auto-Zero Circuit Figure 35b. Ground Returns for Bias Currents with Thermocouple Input +VS SENSE RF RG AD625 VOUT RF LOAD REFERENCE 100k
100k
–VS TO POWER SUPPLY GROUND Figure 35c. Ground Returns for Bias Currents with AC Coupled Inputs thermocouple voltages will appear as “flicker” noise. In SPGA applications relay contacts and CMOS mux leads are both potential sources of additional thermocouple errors. The base emitter junction of an input transistor can rectify out of band signals (i.e., RF interference). When amplifying small signals, these rectified voltages act as small dc offset errors. The AD625 allows direct access to the input transistors’ bases and emitters enabling the user to apply some first order filtering to these unwanted signals. In Figure 37, the RC time constant should be chosen for desired attenuation of the interfering signals. In the case of a resistive transducer, the capacitance alone working against the internal resistance of the transducer may suffice. RG RF R FILTER CAP AUTOZERO CIRCUITS In many applications it is necessary to maintain high accuracy. At room temperature, offset effects can be nulled by the use of offset trimpots. Over the operating temperature range, however, offset nulling becomes a problem. For these applications the autozero circuit of Figure 36 provides a hardware solution. C +IN +GAIN SENSE RTI NULL RF R +IN OTHER CONSIDERATIONS +GAIN DRIVE One of the more overlooked problems in designing ultralowdrift dc amplifiers is thermocouple induced offset. In a circuit comprised of two dissimilar conductors (i.e., copper, kovar), a current flows when the two junctions are at different temperatures. When this circuit is broken, a voltage known as the “Seebeck” or thermocouple emf can be measured. Standard IC lead material (kovar) and copper form a thermocouple with a 16 2 15 3 RTO 14 NULL 4 13 A1 5 A2 NC 6 REF C –IN –GAIN SENSE 1 +V RTI NULL FILTER CAP –IN 11 10k
7 –VS 8 12 10k
AD625 RTO NULL –GAIN DRIVE SENSE 10k
A3 10k
VOUT 10 VOUT 9 +VS Figure 37. Circuit to Attenuate RF Interference –12– REV. D AD625 These capacitances may also be incorporated as part of the external input protection circuit (see section on Input Protection). As a general practice every effort should be made to match the extraneous capacitance at Pins 15 and 2, and Pins 1 and 16, to preserve high ac CMR. –GAIN SENSE 20k
CS-OUT COUT An SPGA provides the ability to externally program precision gains from digital inputs. Historically, the problem in systems requiring electronic switching of gains has been the ON resistance (RON) of the multiplexer, which appears in series with the gain setting resistor RG. This can result in substantial gain errors and gain drifts. The AD625 eliminates this problem by making the gain drive and gain sense pins available (Pins 2, 15, 5, 12; see Figure 39). Consequently the multiplexer’s ON resistance is removed from the signal current path. This transforms the ON resistance error into a small nullable offset error. To clarify this point, an error budget analysis has been performed in Table II based on the SPGA configuration shown in Figure 39. AD7502 TTL/DTL TO CMOS LEVEL TRANSLATOR A0 DECODER/DRIVER VDD A1 GND IOUT – VIN + 3.9k
975k
12-BIT DAS VS 650k
975k
RON IOUT 10k
10k
IS CS CS-OUT COUT –GAIN DRIVE 15.6k
RON SOFTWARE PROGRAMMABLE GAIN AMPLIFIER VSS AD625 –INPUT 10k
IS CS 3.9k
15.6k
+GAIN DRIVE 10k
20k
+GAIN SENSE +INPUT Figure 39. SPGA with Multiplexer Error Sources Figure 39 shows a complete SPGA feeding a 12-bit DAS with a 0 V–10 V input range. This configuration was used in the error budget analysis shown in Table II. The gain used for the RTI calculations is set at 16. As the gain is changed, the ON resistance of the multiplexer and the feedback resistance will change, which will slightly alter the values in the table. EN Table II. Errors Induced by Multiplexer to an SPGA 3.9k
975
20k
650
975
3.9k
15.6k
15.6k
+INPUT +GAIN SENSE RTI NULL 20k
–INPUT 1 16 2 15 3 14 4 13 –GAIN SENSE RTO NULL Specifications AD625C AD7520KN Calculation RTI Offset Voltage Gain Sense Switch Offset Resistance Current 170 Ω 40 nA 40 nA × 170 Ω = 6.8 µV 6.8 µV RTI Offset Voltage Gain Sense Differential Current Switch 60 nA Resistance 6.8 Ω 60 nA × 6.8 Ω = 0.41 µV 0.41 µV –VS +VS RTI NULL +GAIN DRIVE A1 5 A2 NC 6 REF 12 RTO NULL –GAIN DRIVE 11 10k
7 –VS 8 10k
AD625 10k
10k
VOUT A3 9 RG +1= Differential 2 (0.2 nA × 20 kΩ) Leakage = 8 µV/16 Current (IS)2 +0.2 nA –0.2 nA RTO Offset Feedback Voltage Resistance 20 kΩ1 Differential Leakage Current (IOUT)2 +1 nA –1 nA +VS Figure 38 shows an AD625 based SPGA with possible gains of 1, 4, 16, 64. RG equals the resistance between the gain sense lines (Pins 2 and 15) of the AD625. In Figure 38, RG equals the sum of the two 975 Ω resistors and the 650 Ω resistor, or 2600 Ω. RF equals the resistance between the gain sense and the gain drive pins (Pins 12 and 15, or Pins 2 and 5), that is RF equals the 15.6 kΩ resistor plus the 3.9 kΩ resistor, or 19.5 kΩ. The gain, therefore equals: 2RF RTO Offset Feedback Voltage Resistance 20 kΩ1 10 Figure 38. SPGA in a Gain of 16 2(19.5 kΩ) +1=16 (2.6 kΩ) As the switches of the differential multiplexer proceed synchronously, RG and RF change, resulting in the various programmed gain settings. REV. D Induced Error 2 (1 nA × 20 kΩ) = 40 µV/16 Total error induced by a typical CMOS multiplexer to an SPGA at +25°C Voltage Offset Induced RTI 0.5 µV 2.5 µV 10.21 A NOTES 1 The resistor for this calculation is the user-provided feedback resistance (R F). 20 kΩ is recommended value (see Resistor Programmable Gain Amplifier section). 2 The leakage currents (I S and IOUT ) will induce an offset voltage, however, the offset will be determined by the difference between the leakages of each “half’’ of the differential multiplexer. The differential leakage current is multiplied by the feedback resistance (see Note 1), to determine offset voltage. Because differential leakage current is not a parameter specified on multiplexer data sheets, the most extreme difference (one most positive and one most negative) was used for the calculations in Table II. Typical performance will be much better. **The frequency response and settling will be affected by the ON resistance and internal capacitance of the multiplexer. Figure 40 shows the settling time vs. ON resistance at different gain settings for an AD625 based SPGA. **Switch resistance and leakage current errors can be reduced by using relays. –13– AD625 3) Begin all calculations with G0 = 1 and RF0 = 0. 1000 800 RF1 = (20 kΩ – RF0) (1–1/4): RF0 = 0 ∴ RF1 = 15 kΩ 400 RON = 1k
RF2 = [20 kΩ – (RF0 + RF1)] (1–4/16): SETTLING TIME – s 200 RF0 + RF1 = 15 kΩ ∴ RF2 = 3.75 kΩ 100 80 RF3 = [20 kΩ – (RF0 + RF1 + RF2)] (1–16/64): RON = 500
40 RF0 + RF1 + RF2 = 18.75 kΩ ∴ RF3 = 937.5 Ω RON = 200
20 10 8 4) The center resistor (RG of the highest gain setting), is determined last. Its value is the remaining resistance of the 40 kΩ string, and can be calculated with the equation: RON = 0
4 2 1 M 1 4 16 64 GAIN 256 1024 RG = (40 kΩ – 2 ∑ RF j ) j =0 4096 RG = 40 kΩ – 2 (RF0 + RF1 + RF2 + RF3 ) 40 kΩ – 39.375 kΩ = 625 Ω Figure 40. Time to 0.01% of a 20 V Step Input for SPGA with AD625 DETERMINING SPGA RESISTOR NETWORK VALUES The individual resistors in the gain network can be calculated sequentially using the formula given below. The equation determines the resistors as labeled in Figure 41. The feedback resistors and the gain setting resistors are interactive, therefore; the formula must be a series where the present term is dependent on the preceding term(s). The formula RFi + 1 = (20 kΩ – 1 Gi j =0 Gi = 1 ∑ RF j ) (1 – ) G0 = 1 RF0 = 0 can be used to calculate the necessary feedback resistors for any set of gains. This formula yields a network with a total resistance of 40 kΩ. A dummy variable (j) serves as a counter to keep a running total of the preceding feedback resistors. To illustrate how the formula can be applied, an example similar to the calculation used for the resistor network in Figure 38 is examined below. 5) If different resistor values are desired, all the resistors in the network can be scaled by some convenient factor. However, raising the impedance will increase the RTO errors, lowering the total network resistance below 20 kΩ can result in amplifier instability. More information on this phenomenon is given in the RPGA section of the data sheet. The scale factor will not affect the unity gain feedback resistors. The resistor network in Figure 38 has a scaling factor of 650/625 = 1.04, if this factor is used on RF1, RF2, RF3, and RG, then the resistor values will match exactly. 6) Round off errors can be cumulative, therefore, it is advised to carry as many significant digits as possible until all the values have been calculated. AD75xx TO GAIN SENSE (PIN 2) 1) Unity gain is treated as a separate case. It is implemented with separate 20 kΩ feedback resistors as shown in Figure 41. It is then ignored in further calculations. 2) Before making any calculations it is advised to draw a resistor network similar to the network in Figure 41. The network will have (2 × M) + 1 resistors, where M = number of gains. For Figure 38 M = 3 (4, 16, 64), therefore, the resistor string will have seven resistors (plus the two 20 kΩ “side” resistors for unity gain). –14– RF2 20k
RFN RFG RFN TO GAIN SENSE (PIN 15) RF2 20k
RF1 CONNECT IF UNITY GAIN IS DESIRED TO GAIN DRIVE (PIN 5) CONNECT IF UNITY GAIN IS DESIRED TO GAIN DRIVE (PIN 12) Figure 41. Resistors for a Gain Setting Network REV. D AD625 OUTLINE DIMENSIONS Dimensions shown in inches and (mm). 0.430 (10.922) 0.755 (19.18) 0.745 (18.93) 16 9 1 8 0.040R 16 0.310 0.01 7.874 0.254) 0.26 (6.61) 0.24 (6.1) PIN 1 0.17 (4.32) MAX 0.175 (4.45) SEATING PLANE 0.12 (3.05) 0.02 (0.508) 0.015 (2.67) 0.065 (1.66) 0.015 (0.381) 0.095 (2.42) 0.045 (1.15) 9 0.265 0.290 0.010 (6.73) (7.37 0.254) 1 0.306 (7.78) 0.294 (7.47) 8 PIN 1 0.800 0.010 20.32 0.254 0.14 (3.56) 0.12 (3.05) 0.095 (2.41) 0.012 (0.305) 0.008 (0.203) 0.125 (3.175) MIN 0.047 0.007 (1.19 0.18) 0.035 0.01 (0.889 0.254) 0.300 (7.62) REF 0.085 (2.159) C00780c–0–6/00 (rev. D) 16-Lead Ceramic DIP (D-16) 16-Lead Plastic DIP (N-16) 0.180 0.03 (4.57 0.762) +0.003 –0.002 +0.076 0.43 –0.05 0.017 0.700 (17.78) BSC SEATING PLANE 0.010 0.002 (0.254 0.05) 0.100 (254) BSC 20-Terminal Leadless Chip Carrier (E-20A) 0.082 0.018 (2.085 0.455) 0.350 0.008 (8.89 0.20) SQ 3 4 19 18 20 1 0.050 (1.27) BOTTOM VIEW 14 13 0.20  45° (0.51  45°) REF 0.025 0.003 (0.635 0.075) 8 9 PRINTED IN U.S.A. 0.040  45° (1.02  45°) REF 3 PLCS REV. D –15–