AD711KNZ

a FEATURES Enhanced Replacement for LF411 and TL081 AC PERFORMANCE Settles to ⴞ0.01% in 1.0 ␮s 16 V/␮s min Slew Rate (AD711J) 3 MHz min Unity Gain Bandwidth (AD711J) DC PERFORMANCE 0.25 mV max Offset Voltage: (AD711C) 3 ␮V/ⴗC max Drift: (AD711C) 200 V/mV min Open-Loop Gain (AD711K) 4 ␮V p-p max Noise, 0.1 Hz to 10 Hz (AD711C) Available in Plastic Mini-DIP, Plastic SOIC, Hermetic Cerdip, and Hermetic Metal Can Packages MIL-STD-883B Parts Available Available in Tape and Reel in Accordance with EIA-481A Standard Surface Mount (SOIC) Dual Version: AD712 PRODUCT DESCRIPTION The AD711 is a high speed, precision monolithic operational amplifier offering high performance at very modest prices. Its very low offset voltage and offset voltage drift are the results of advanced laser wafer trimming technology. These performance benefits allow the user to easily upgrade existing designs that use older precision BiFETs and, in many cases, bipolar op amps. The superior ac and dc performance of this op amp makes it suitable for active filter applications. With a slew rate of 16 V/ms and a settling time of 1 ms to ± 0.01%, the AD711 is ideal as a buffer for 12-bit D/A and A/D Converters and as a high-speed integrator. The settling time is unmatched by any similar IC amplifier. The combination of excellent noise performance and low input current also make the AD711 useful for photo diode preamps. Common-mode rejection of 88 dB and open loop gain of 400 V/mV ensure 12-bit performance even in high-speed unity gain buffer circuits. The AD711 is pinned out in a standard op amp configuration and is available in seven performance grades. The AD711J and AD711K are rated over the commercial temperature range of 0∞C to 70∞C. The AD711A, AD711B and AD711C are rated over the industrial temperature range of –40∞C to +85∞C. The AD711S and AD711T are rated over the military temperature range of –40∞C to +125∞C and are available processed to MILSTD-883B, REV. E. Precision, Low Cost, High Speed, BiFET Op Amp AD711 CONNECTION DIAGRAMS OFFSET NULL NC +VS INVERTING INPUT OUTPUT AD711 NON INVERTING INPUT OFFSET NULL –VS 10k⍀ NC = NO CONNECT –15V NOTE PIN 4 CONNECTED TO CASE OFFSET NULL INVERTING INPUT NONINVERTING INPUT –VS VOS TRIM 1 8 NC 2 7 +VS 3 6 4 5 OUTPUT OFFSET NULL AD711 NC = NO CONNECT Extended reliability PLUS screening is available, specified over the commercial and industrial temperature ranges. PLUS screening includes 168 hour burn-in, as well as other environmental and physical tests. The AD711 is available in an 8-pin plastic mini-DIP, small outline, cerdip, TO-99 metal can, or in chip form. PRODUCT HIGHLIGHTS 1. The AD711 offers excellent overall performance at very competitive prices. 2. Analog Devices’ advanced processing technology and 100% testing guarantee a low input offset voltage (0.25 mV max, C grade, 2 mV max, J grade). Input offset voltage is specified in the warmed-up condition. Analog Devices’ laser wafer drift trimming process reduces input offset voltage drifts to 3 mV/∞C max on the AD711C. 3. Along with precision dc performance, the AD711 offers excellent dynamic response. It settles to ± 0.01% in 1 ms and has a 100% tested minimum slew rate of 16 V/ms. Thus this device is ideal for applications such as DAC and ADC buffers which require a combination of superior ac and dc performance. 4. The AD711 has a guaranteed and tested maximum voltage noise of 4 mV p-p, 0.1 to 10 Hz (AD711C). 5. Analog Devices’ well-matched, ion-implanted JFETs ensure a guaranteed input bias current (at either input) of 25 pA max (AD711C) and an input offset current of 10 pA max (AD711C). Both input bias current and input offset current are guaranteed in the warmed-up condition. REV. E Information furnished by Analog Devices is believed to be accurate and reliable. However, no responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other rights of third parties that may result from its use. No license is granted by implication or otherwise under any patent or patent rights of Analog Devices. One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A. Tel: 781/329-4700 www.analog.com Fax: 781/326-8703 © Analog Devices, Inc., 2002 AD711–SPECIFICATIONS (V =
15 V @ T = 25C, unless otherwise noted.) S Parameter Min INPUT OFFSET VOLTAGE1 Initial Offset TMIN to TMAX vs. Temp vs. Supply 76 TMIN to TMAX 76/76/76 Long-Term Stability INPUT BIAS CURRENT2 VCM = 0 V VCM = 0 V @ TMAX VCM = ± 10 V 3.0 16 Current POWER SUPPLY Rated Performance Operating Range Quiescent Current Min 2/1/1 3/2/2 20/20/20 K/B/T Typ 0.2 5 100 80 80 50 1.1/3.2/51 100 15 20 25 0.6/1.6/26 5 3.4 4.0 200 20 1.0 0.0003 18 1.2 Min 0.5 1.0 10 C Typ 0.10 86 86 2 110 Max Unit 0.25 0.45 5 mV mV mV/∞C dB dB mV/Month 25 1.6 50 pA nA pA 10 0.65 pA nA 15 50 1.1/3.2/51 100 15 20 25 0.6/1.6/26 5 3.4 18 1.2 4.0 200 20 1.0 0.0003 1.2 MHz kHz V/ms ms % 3 ¥ 1012储5.5 3 ¥ 1012储5.5 3 ¥ 1012储5.5 3 ¥ 1012储5.5 3 ¥ 1012储5.5 3 ¥ 1012储5.5 W储pF W储pF ± 20 +14.5, –11.5 ± 20 +14.5, –11.5 ± 20 +14.5, –11.5 V +VS – 2 88 84 84 80 –VS + 4 80 80 76 74 +VS – 2 88 84 84 80 0.01 +13, –12.5 +13.9, –13.3 ± 12/± 12/± 12 +13.8, –13.1 25 +13, –12.5 +13.9, –13.3 ± 12 +13.8, –13.1 25 2.5 86 86 76 74 ± 18 3.4 400 ± 4.5 ± 15 2.5 +V – 2 V 94 90 90 84 2 45 22 18 16 0.01 200 100 ± 15 –VS + 4 2 45 22 18 16 150 400 100/100/100 ± 4.5 Max 15 2 45 22 18 16 INPUT CURRENT NOISE OUTPUT CHARACTERISTICS Voltage 4.0 200 20 1.0 0.0003 –VS + 4 76 76/76/76 70 70/70/70 Max 15 10 INPUT VOLTAGE NOISE OPEN-LOOP GAIN 7 95 20 INPUT IMPEDANCE Differential Common Mode INPUT VOLTAGE RANGE Differential3 Common-Mode Voltage4 TMIN to TMAX Common-Mode Rejection Ratio VCM = ± 10 V TMIN to TMAX VCM = ± 11 V TMIN to TMAX 0.3 15 INPUT OFFSET CURRENT VCM = 0 V VCM = 0 V @ TMAX FREQUENCY RESPONSE Small Signal Bandwidth Full Power Response Slew Rate Settling Time to 0.01% Total Harmonic Distortion J/A/S Typ A 200 100 dB dB dB dB 4 0.01 pA/÷Hz 400 V/mV V/mV +13, –12.5 +13.9, –13.3 ± 12 +13.8, –13.1 25 ± 18 3.0 ± 4.5 mV p-p nV/÷Hz nV/÷Hz nV/÷Hz nV/÷Hz ± 15 2.5 V V mA ± 18 2.8 V V mA NOTES 1 Input Offset Voltage specifications are guaranteed after 5 minutes of operation at T A = 25∞C. 2 Bias Current specifications are guaranteed maximum at either input after 5 minutes of operation at T A = 25∞C. For higher temperatures, the current doubles every 10∞C. 3 Defined as voltage between inputs, such that neither exceeds ± 10 V from ground. 4 Typically exceeding –14.1 V negative common-mode voltage on either input results in an output phase reversal. Specifications subject to change without notice. –2– REV. E AD711 ABSOLUTE MAXIMUM RATINGS 1 Supply Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . ± 18 V Internal Power Dissipation2 . . . . . . . . . . . . . . . . . . . . . 500 mW Input Voltage3 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . ± 18 V Output Short Circuit Duration . . . . . . . . . . . . . . . . . Indefinite Differential Input Voltage . . . . . . . . . . . . . . . . . . +VS and –VS Storage Temperature Range (Q, H) . . . . . . . –65∞C to +150∞C Storage Temperature Range (N) . . . . . . . . . . –65∞C to +125∞C Operating Temperature Range AD711J/K . . . . . . . . . . . . . . . . . . . . . . . . . . . 0∞C to +70∞C AD711A/B/C . . . . . . . . . . . . . . . . . . . . . . . . –40∞C to +85∞C AD711S/T . . . . . . . . . . . . . . . . . . . . . . . . . –55∞C to +125∞C Lead Temperature Range (Soldering 60 sec) . . . . . . . . . 300∞C NOTES 1 Stresses above those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. This is a stress rating only and functional operation of the device at these or any other conditions above those indicated in the operational section of this specification is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability. 2 Thermal Characteristics: 8-Pin Plastic Package: qJC = 33∞C/Watt; qJA = 100∞C/Watt 8-Pin Cerdip Package: qJC = 22∞C/Watt; qJA = 110∞C/Watt 8-Pin Metal Can Package: qJC = 65∞C/Watt; qJA = 150∞C/Watt 8-Pin SOIC Package: qJC = 43∞C/Watt; qJA = 160∞C/Watt 3 For supply voltages less than ± 18 V, the absolute maximum input voltage is equal to the supply voltage. ORDERING GUIDE Model *AD711AH AD711AQ *AD711BQ *AD711CH AD711JN AD711JR AD711JR-REEL AD711JR-REEL7 AD711KN AD711KR AD711KR-REEL AD711KR-REEL7 *AD711SQ/883B *AD711TQ/883B Temperature Range –40∞C to +85∞C –40∞C to +85∞C –40∞C to +85∞C –40∞C to +85∞C 0∞C to 70∞C 0∞C to 70∞C 0∞C to 70∞C 0∞C to 70∞C 0∞C to 70∞C 0∞C to 70∞C 0∞C to 70∞C 0∞C to 70∞C –55∞C to +125∞C –55∞C to +125∞C 8-Pin Metal Can 8-Pin Ceramic DIP 8-Pin Ceramic DIP 8-Pin Metal Can 8-Pin Plastic DIP 8-Pin Plastic SOIC 8-Pin Plastic SOIC 8-Pin Plastic SOIC 8-Pin Plastic DIP 8-Pin Plastic SOIC 8-Pin Plastic SOIC 8-Pin Plastic SOIC 8-Pin Ceramic DIP 8-Pin Ceramic DIP Package Option* H-08A Q-8 Q-8 H-08A N-8 RN-8 RN-8 RN-8 N-8 RN-8 RN-8 RN-8 Q-8 Q-8 *Not for new design, obsolete April 2002 CAUTION ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily accumulate on the human body and test equipment and can discharge without detection. Although the AD711 features proprietary ESD protection circuitry, permanent damage may occur on devices subjected to high-energy electrostatic discharges. Therefore, proper ESD precautions are recommended to avoid performance degradation or loss of functionality. REV. E Package Description –3– WARNING! ESD SENSITIVE DEVICE AD711–Typical Performance Characteristics 10 RL = 2k 25C 5 0 5 10 15 SUPPLY VOLTAGE –
Volts +VOUT 10 RL = 2k 25C 5 0 –VOUT 0 5 10 15 SUPPLY VOLTAGE –
Volts –6 –8 10 5 10 15 SUPPLY VOLTAGE –
Volts 20 –9 10 –10 0.01 0.01 1k 0 20 40 60 80 100 120 140 TEMPERATURE – C TPC 5. Input Bias Current vs. Temperature SHORT CIRCUIT CURRENT LIMIT – mA INPUT BIAS CURRENT – pA 1 MAX J GRADE LIMIT 25 –5 0 5 COMMON MODE VOLTAGE – Volts 10 TPC 7. Input Bias Current vs. Common Mode Voltage 10k 100k 1M FREQUENCY – Hz 10M TPC 6. Output Impedance vs. Frequency 26 50 0 –10 –12 –60 –40 –20 100 75 10k –11 10 TPC 4. Quiescent Current vs. Supply Voltage VS =
15V 25C 100 1k LOAD RESISTANCE –  10 5.0 24 +OUTPUT CURRENT 22 20 18 –OUTPUT CURRENT 16 14 12 10 –60 –40 –20 0 20 40 60 80 100 120 140 AMBIENT TEMPERATURE – C TPC 8. Short Circuit Current Limit vs. Temperature –4– UNITY GAIN BANDWIDTHT – MHz 0 5 AVCL = 1 10 1.75 10 –7 10 10 2.00
15V SUPPLIES 15 100 OUTPUT IMPEDANCE –  INPUT BIAS CURRENT (VCM = 0) – Amps 2.25 20 TPC 3. Output Voltage Swing vs. Load Resistance 10 2.50 25 0 10 20 TPC 2. Output Voltage Swing vs. Supply Voltage 2.75 QUIESCENT CURRENT – mA 15 20 TPC 1. Input Voltage Swing vs. Supply Voltage OUTPUT VOLTAGE SWING – Volts p-p 15 0 30 20 OUTPUT VOLTAGE SWING – Volts INPUT VOLTAGE SWING – Volts 20 4.5 4.0 3.5 3.0 –60 –40 –20 0 20 40 60 80 100 120 140 TEMPERATURE – C TPC 9. Unity Gain Bandwidth vs. Temperature REV. E AD711 80 120 GAIN 40 40 20 20 RL = 2k C = 100pF –20 10 100 1k 10k 100k FREQUENCY – Hz 1M –20 10M 95 OUTPUT VOLTAGE – Volts p-p CMR – dB 40 20 100 1k 10k 100k FREQUENCY – Hz 1M INPUT NOISE VOLTAGE – nV/ Hz –90 –100 –110 –120 100k TPC 16. Total Harmonic Distortion vs. Frequency 40 VS =
15 SUPPLIES WITH 1V p-p SINE WAVE 25 C 20 10 100 1k 10k 10k SUPPLY MODULATION FREQUENCY – Hz 2 25 20 15 10 5 1M INPUT FREQUENCY – Hz 10M 8 6 4 2 1% 0.1% 0.01% 0 ERROR –2 1% 0.1% 0.01% –4 –6 –8 –10 0.5 0.6 0.7 0.8 0.9 SETTLING TIME – s 1.0 TPC 15. Output Swing and Error vs. Settling Time 25 20 100 10 15 10 5 1 1k 10k FREQUENCY – Hz –SUPPLY 60 TPC 12. Power Supply Rejection vs. Frequency 1k 3V RMS RL = 2k CL = 100pF +SUPPLY 80 1 TPC 14. Large Signal Frequency Response –70 100 0 20 RL = 2k 25 C VS =
15V 0 100k TPC 13. Common Mode Rejection vs. Frequency THD – dB 5 10 15 SUPPLY VOLTAGE –
Volts 30 60 REV. E 0 TPC 11. Open-Loop Gain vs. Supply Voltage VS =
15V VCM = 1V p-p 25 C 80 –130 100 105 100 100 –80 110 0 TPC 10. Open-Loop Gain and Phase Margin vs. Frequency 0 10 115 OUTPUT SWING FRIM 0V TO
Volts 0 POWER SUPPLY REJECTION – dB 60 60 110 RL = 2k 25C SLEW RATE – Vs OPEN LOOP GAIN – dB 125 OPEN-LOOP GAIN – dB PHASE 80 100 PHASE MARGIN – Degrees 100 1 10 100 1k FREQUENCY – Hz 10k TPC 17. Input Noise Voltage Spectral Density –5– 100k 0 0 100 200 300 400 500 600 700 800 900 INPUT ERROR SIGNAL – mV (AT SUMMING JUNCTION) TPC 18. Slew Rate vs. Input Error Signal AD711 25 24 SLEW RATE – V/s 23 22 +VS 0.1F +VS +VS 0.1F 1.3Mk 0.1F 21 10k 20 19 INPUT 18 AD711 AD711 OUTPUT 0.1F 2k 100pF AD711 17 0.1F 16 0.1F 10k –VS –VS 15 –60 –40 –20 0 20 40 60 80 100 120 140 TEMPERATURE – C TPC 19. Slew Rate vs. Temperature +VS 0.1F SQUARE WAVE INPUT TPC 21. Offset Null Configurations 0.1F AD711 VIN TPC 20. T.H.D. Test Circuit –VS RL 2k VOUT CL 100pF –VS TPC 22a. Unity Gain Follower TPC 22b. Unity Gain Follower Pulse Response (Large Signal) TPC 22c. Unity Gain Follower Pulse Response (Small Signal) TPC 23b. Unity Gain Inverter Pulse Response (Large Signal) TPC 23c. Unity Gain Inverter Pulse Response (Small Signal) 5k +VS VIN 0.1F 5k AD711 0.1F SQUARE WAVE INPUT RL 2k VOUT CL 100pF –VS TPC 23a. Unity Gain Inverter –6– REV. E AD711 In addition to a significant improvement in settling time, the low offset voltage, low offset voltage drift, and high open-loop gain of the AD711 family assures 12-bit accuracy over the full operating temperature range. OPTIMIZING SETTLING TIME Most bipolar high-speed D/A converters have current outputs; therefore, for most applications, an external op amp is required for current-to-voltage conversion. The settling time of the converter/op amp combination depends on the settling time of the DAC and output amplifier. A good approximation is: t S Total = (t S DAC )2 + (t S AMP )2 The excellent high-speed performance of the AD711 is shown in the oscilloscope photos of Figure 2. Measurements were taken using a low input capacitance amplifier connected directly to the summing junction of the AD711 – both photos show the worst case situation: a full-scale input transition. The DAC’s 4 kW [10 kW储8 kW = 4.4 kW] output impedance together with a 10 kW feedback resistor produce an op amp noise gain of 3.25. The current output from the DAC produces a 10 V step at the op amp output (0 to –10 V Figure 2a, –10 V to 0 V Figure 2b.) (1) The settling time of an op amp DAC buffer will vary with the noise gain of the circuit, the DAC output capacitance, and with the amount of external compensation capacitance across the DAC output scaling resistor. Therefore, with an ideal op amp, settling to ± 1/2 LSB (± 0.01%) requires that 375 mV or less appears at the summing junction. This means that the error between the input and output (that voltage which appears at the AD711 summing junction) must be less than 375 mV. As shown in Figure 2, the total settling time for the AD711/AD565 combination is 1.2 microseconds. Settling time for a bipolar DAC is typically 100 ns to 500 ns. Previously, conventional op amps have required much longer settling times than have typical state-of-the-art DACs; therefore, the amplifier settling time has been the major limitation to a high-speed voltage-output D-to-A function. The introduction of the AD711/712 family of op amps with their 1 ms (to ± 0.01% of final value) settling time now permits the full high-speed capabilities of most modern DACs to be realized. 0.1F BIPOLAR OFFSET ADJUST REF OUT R2 GAIN 100 ADJUST 10V 20k –VEE 20V SPAN POWER GND 10pF 5k 9.95k 10V SPAN 0.5mA IREF 0.1F BIPOLAR OFF AD565A 19.95k REF IN REF GND R1 100 VCC 5k DAC IOUT = 4  IREF  CODE IO 5k +15V 0.1F DAC OUT AD711K 0.1F OUTPUT –10V TO +10V –15V MSB LSB Figure 1. ± 10 V Voltage Output Bipolar DAC a. (Full-Scale Negative Transition) b. (Full-Scale Positive Transition) Figure 2. Settling Characteristics for AD711 with AD565A REV. E –7– AD711 OP AMP SETTLING TIME—A MATHEMATICAL MODEL The design of the AD711 gives careful attention to optimizing individual circuit components; in addition, a careful tradeoff was made: the gain bandwidth product (4 MHz) and slew rate (20 V/ms) were chosen to be high enough to provide very fast settling time but not too high to cause a significant reduction in phase margin (and therefore stability). Thus designed, the AD711 settles to ± 0.01%, with a 10 V output step, in under 1 ms, while retaining the ability to drive a 100 pF load capacitance when operating as a unity gain follower. If an op amp is modeled as an ideal integrator with a unity gain crossover frequency of wo/2p, Equation 1 will accurately describe the small signal behavior of the circuit of Figure 3a, consisting of an op amp connected as an I-to-V converter at the output of a bipolar or CMOS DAC. This equation would completely describe the output of the system if not for the op amp’s finite slew rate and other nonlinear effects. VO –R = R(C f = CX ) 2 Ê GN I IN ˆ s +Á + RC f ˜ s + 1 wo ¯ Ë wo (3) where: wo =op amp’s unity gain frequency 2 op amp is being simulated or it is the combined capacitance of the DAC output and the op amp input if the DAC buffer is being modeled. AD711 RIN VIN RL R Figure 3b. Simplified Model of the AD711 Used as an Inverter In either case, the capacitance CX causes the system to go from a one-pole to a two-pole response; this additional pole increases settling time by introducing peaking or ringing in the op amp output. Since the value of CX can be estimated with reasonable accuracy, Equation 2 can be used to choose a small capacitor, CF, to cancel the input pole and optimize amplifier response. Figure 4 is a graphical solution of Equation 2 for the AD711 with R = 4 kW. 60 R ˆ GN = “noise” gain of circuit ÁË1 + R ˜¯ O GN = 4.0 50 GN = 3.0 This equation may then be solved for Cf: GN = 2.0 40 2 - GN 2 RC X w o + (1 - GN ) + Rw o Rw o (3) CX Cf = CL CX p Ê VOUT CF 30 GN = 1.5 In these equations, capacitor CX is the total capacitor appearing the inverting terminal of the op amp. When modeling a DAC buffer application, the Norton equivalent circuit of Figure 3a can be used directly; capacitance CX is the total capacitance of the output of the DAC plus the input capacitance of the op amp (since the two are in parallel). 20 GN = 1.0 10 0 0 10 20 30 CF 40 50 60 Figure 4. Value of Capacitor CF vs. Value of CX AD711 VOUT CF RL The photos of Figures 5a and 5b show the dynamic response of the AD711 in the settling test circuit of Figure 6. CL The input of the settling time fixture is driven by a flat-top pulse generator. The error signal output from the false summing node of A1 is clamped, amplified by A2 and then clamped again. The error signal is thus clamped twice: once to prevent overloading amplifier A2 and then a second time to avoid overloading the oscilloscope preamp. The Tektronix oscilloscope preamp type 7A26 was carefully chosen because it does not overload with these input levels. Amplifier A2 needs to be a very high speed FET-input op amp; it provides a gain of 10, amplifying the error signal output of A1. R IO RO CX Figure 3a. Simplified Model of the AD711 Used as a Current-Out DAC Buffer When RO and IO are replaced with their Thevenin VIN and RIN equivalents, the general purpose inverting amplifier of Figure 26b is created. Note that when using this general model, capacitance CX is either the input capacitance of the op amp if a simple inverting –8– REV. E AD711 current-to-voltage converters. The use of a guarding technique such as that shown in Figure 7, in printed circuit board layout and construction is critical to minimize leakage currents. The guard ring is connected to a low impedance potential at the same level as the inputs. High impedance signal lines should not be extended for any unnecessary length on the printed circuit board. 4 4 5 5 3 6 2 3 2 6 Figure 5a. Settling Characteristics 0 to +10 V Step Upper Trace: Output of AD711 Under Test (5 V/Div) Lower Trace: Amplified Error Voltage (0.01%/Div) 1 8 1 7 7 8 Figure 7. Board Layout for Guarding Inputs D/A CONVERTER APPLICATIONS The AD711 is an excellent output amplifier for CMOS DACs. It can be used to perform both 2-quadrant and 4-quadrant operation. The output impedance of a DAC using an inverted R-2R ladder approaches R for codes containing many 1s, 3R for codes containing a single 1, and for codes containing all zero, the output impedance is infinite. For example, the output resistance of the AD7545 will modulate between 11 kW and 33 kW. Therefore, with the DAC’s internal feedback resistance of 11 kW, the noise gain will vary from 2 to 4/3. This changing noise gain modulates the effect of the input offset voltage of the amplifier, resulting in nonlinear DAC amplifier performance. Figure 5b. Settling Characteristics 0 to –10 V Step Upper Trace: Output of AD711 Under Test (5 V/Div) Lower Trace: Amplified Error Voltage (0.01%/Div) The AD711K with guaranteed 500 mV offset voltage minimizes this effect to achieve 12-bit performance. GUARDING The low input bias current (15 pA) and low noise characteristics of the AD711 BiFET op amp make it suitable for electrometer applications such as photo diode preamplifiers and picoampere 5pF AD3554 HP2835 205 VERROR  5 HP2835 0.47F 4.99k 4.99k 200k DATA DYNAMICS 5109 (OR EQUIVALENT FLAT TOP PULSE GENERATOR) –15V 5-18pF VIN 1.1k 10k 10k 0.47F +15V 10k 0.2-0.0pF VOUT AD711 5k 10pF 0.1F 0.1F –15V +15V Figure 6. Settling Time Test Circuit REV. E –9– TEXTRONIX 7A26 OSCILLOSCOPE PREAMP INPUT SELECTION 1M 20pF AD711 compared to a series of switched trial currents. The comparison point is diode clamped but may deviate several hundred millivolts resulting in high frequency modulation of A/D input current. Figures 8 and 9 show the AD711 and AD7545 (12-bit CMOS DAC) configured for unipolar binary (2-quadrant multiplication) or bipolar (4-quadrant multiplication) operation. Capacitor C1 provides phase compensation to reduce overshoot and ringing. Figures 10a and 10b show the settling time characteristics of the AD711 when used as a DAC output buffer for the AD7545. R2* +15 VDD GAIN ADJUST VIN R1* VDD VREF C1 33pF 0.1F RFB OUT1 AD711K AD7545 VOUT DGND AGND CF 0.1F ANALOG COMMON DB11-DB0 *FOR VALUES R1 AND R2, REFER TO TABLE 1 –15 a. Full-Scale Positive Transition Figure 8. Unipolar Binary Operation Figure 10. Settling Characteristics for AD711 with AD7545 R1 and R2 calibrate the zero offset and gain error of the DAC. Specific values for these resistors depend upon the grade of AD7545 and are shown below. compared to a series of switched trial currents. The comparison point is diode clamped but may deviate several hundred millivolts resulting in high frequency modulation of A/D input current. The output impedance of a feedback amplifier is made artificially low by the loop gain. At high frequencies, where the loop gain is low, the amplifier output impedance can approach its open loop value. Most IC amplifiers exhibit a minimum open loop output impedance of 25 W due to current limiting resistors. A few hundred microamps reflected from the change in converter loading can introduce errors in instantaneous input Table I. Recommended Trim Resistor Values vs. Grades of the AD7545 for VDD = 5 V TRIM RESISTOR JN/AQ/SD KN/BQ/TD LN/CQ/UD GLN/GCQ/GUD R1 R2 500 W 150 W 200 W 68 W 100 W 33 W b. Full-Scale Negative Transition 20 W 6.8 W NOISE CHARACTERISTICS 12/8 The random nature of noise, particularly in the 1/f region, makes it difficult to specify in practical terms. At the same time, designers of precision instrumentation require certain guaranteed maximum noise levels to realize the full accuracy of their equipment. R/C GAIN ADJUST R2 100 0.1F OFFSET ADJUST An op amp driving the analog input of an A/D converter, such as that shown in Figure 11, must be capable of maintaining a constant output voltage under dynamically changing load conditions. In successive-approximation converters, the input current is VIN R1* CE MIDDLE BITS REF IN REF OUT BIP OFF 10VIN ANA COM LOW BITS +5V +15V –15V DIG COM ANALOG COM Figure 11. AD711 as ADC Unity Gain Buffer C1 33pF R4 20k 1% +15V 0.1F OUT1 VREF AD7545 AGND DGND DB11-DB0 0.1F –15V R2* RFB AD574 20VIN AD711
10V ANALOG INPUT DRIVING THE ANALOG INPUT OF AN A/D CONVERTER VDD R1 100 +15V All other grades of the AD711 are sample-tested on an AQL basis to a limit of 6 mV p-p, 0.1 to 10 Hz. GAIN ADJUST HIGH BITS AO The AD711C grade is specified at a maximum level of 4.0 mV p-p, in a 0.1 Hz to 10 Hz bandwidth. Each AD711C receives a 100% noise test for two 10-second intervals; devices with any excursion in excess of 4.0 mV are rejected. The screened lot is then submitted to Quality Control for verification on an AQL basis. VDD STS CS AD711K R5 20k 1% +15V R3 10k 1% 0.1F 0.1F AD711K VOUT 0.1F 12 –15V *FOR VALUES R1 AND R2, REFER TO TABLE 1 DATA INPUT ANALOG COMMON –15V Figure 9. Bipolar Operation –10– REV. E AD711 voltage. If the A/D conversion speed is not excessive and the bandwidth of the amplifier is sufficient, the amplifier’s output will return to the nominal value before the converter makes its comparison. However, many amplifiers have relatively narrow bandwidth yielding slow recovery from output transients. The AD711 is ideally suited to drive high speed A/D converters since it offers both wide bandwidth and high open-loop gain. large value input resistors, bias currents flowing through these resistors will also generate an offset voltage. In addition, at higher frequencies, an op amp’s dynamics must be carefully considered. Here, slew rate, bandwidth, and open-loop gain play a major role in op amp selection. The slew rate must be fast as well as symmetrical to minimize distortion. The amplifier’s bandwidth in conjunction with the filter’s gain will dictate the frequency response of the filter. The use of a high performance amplifier such a s the AD711 will minimize both dc and ac errors in all active filter applications. SECOND ORDER LOW PASS FILTER a. Source Current = 2 mA b. Sink Current = 1 mA Figure 12. ADC Input Unity Gain Buffer Recovery Times Figure 15 depicts the AD711 configured as a second order Butterworth low pass filter. With the values as shown, the corner frequency will be 20 kHz; however, the wide bandwidth of the AD711 permits a corner frequency as high as several hundred kilohertz. Equations for component selection are shown below. R1 = R2 = user selected (typical values: 10 kW – 100 kW) DRIVING A LARGE CAPACITIVE LOAD The circuit in Figure 13 employs a 100 W isolation resistor which enables the amplifier to drive capacitive loads exceeding 1500 pF; the resistor effectively isolates the high frequency feedback from the load and stabilizes the circuit. Low frequency feedback is returned to the amplifier summing junction via the low pass filter formed by the 100 W series resistor and the load capacitance, CL. Figure 14 shows a typical transient response for this connection. C1= 1.414 0.707 , C2 = (2 p)( f cutoff )(R1) (2 p)( f cutoff )(R1) (5) Where: C1 and C2 are in farads. C1 560pF +15V 0.1F R1 20k 4.99k 30pF +VS (4) VIN 0.1F R2 20k C2 280pF AD711 VOUT 0.1F INPUT 4.99k TYPICAL CAPACITANCE LIMIT FOR VARIOUS LOAD RESISTORS RL 2k 10k 20k CL UP TO 1500pF 1500pF 1000pF AD711 0.1F 100 CL OUTPUT –15V RL –VS Figure 13. Circuit for Driving a Large Capacitive Load Figure 15. Second Order Low Pass Filter An important property of filters is their out-of-band rejection. The simple 20 kHz low pass filter shown in Figure 15, might be used to condition a signal contaminated with clock pulses or sampling glitches which have considerable energy content at high frequencies. The low output impedance and high bandwidth of the AD711 minimize high frequency feedthrough as shown in Figure 16. The upper trace is that of another low-cost BiFET op amp showing 17 dB more feedthrough at 5 MHz. Figure 14. Transient Response RL = 2 kW, CL = 500 pF ACTIVE FILTER APPLICATIONS In active filter applications using op amps, the dc accuracy of the amplifier is critical to optimal filter performance. The amplifier’s offset voltage and bias current contribute to output error. Offset voltage will be passed by the filter and may be amplified to produce excessive output offset. For low frequency applications requiring REV. E Figure 16. –11– AD711 2-pole response; for a total of 8 poles. The 9th pole consists of a 0.001 mF capacitor and a 124 kW resistor at Pin 3 of amplifier A2. Figure 18 depicts the circuits for each FDNR with the proper selection of R. To achieve optimal performance, the 0.001 mF capacitors must be selected for 1% or better matching and all resistors should have 1% or better tolerance. 9-POLE CHEBYCHEV FILTER Figure 17 shows the AD711 and its dual counterpart, the AD712, as a 9-pole Chebychev filter using active frequency dependent negative resistors (FDNR). With a cutoff frequency of 50 kHz and better than 90 dB rejection, it may be used as an anti-aliasing filter for a 12-bit data acquisition system with 100 kHz throughput. As shown in Figure 17, the filter is comprised of four FDNRs (A, B, C, D) having values of 4.9395 ⫻ 10–15 and 5.9276 ⫻ 10–15 farad-seconds. Each FDNR active network provides a +15V 0.1F +15V 0.1F VIN 0.001F A1 2800 6190 6490 6190 2800 AD711 –15 4.9395E 0.1F –15 5.9276E –15 5.9276E A2 –15 4.9395E A B C D * * * * AD711 VOUT 0.1F –15V 0.001 F 100k *SEE TEXT 4.99k 124k –15V 4.99k Figure 17. 9-Pole Chebychev Filter +15V 0.1F 0.001F 1/2 R 0.1F 1/2 AD712 AD712 0.001F –15V 1k R: 24.9k FOR 4.9395E–15 29.4k FOR 5.9276E–15 4.99k Figure 18. FDNR for 9-Pole Chebychev Filter Figure 19. High Frequency Response for 9-Pole Chebychev Filter –12– REV. E AD711 OUTLINE DIMENSIONS 8-Lead Plastic Dual-in-Line Package [PDIP] (N-8) 8-Lead Ceramic Dip – Glass Hermetic Seal [CERDIP] (Q-8) Dimensions shown in inches and (millimeters) Dimensions shown in inches and (millimeters) 0.005 (0.13) MIN 0.375 (9.53) 0.365 (9.27) 0.355 (9.02) 8 8 5 0.295 (7.49) 0.285 (7.24) 0.275 (6.98) 4 1 0.310 (7.87) 0.220 (5.59) 1 0.100 (2.54) BSC 0.320 (8.13) 0.290 (7.37) 0.405 (10.29) MAX 0.060 (1.52) 0.015 (0.38) 0.200 (5.08) MAX 0.150 (3.81) MIN 0.200 (5.08) 0.125 (3.18) 0.015 (0.38) 0.010 (0.25) 0.008 (0.20) SEATING PLANE 0.060 (1.52) 0.050 (1.27) 0.045 (1.14) 4 0.100 (2.54) BSC 0.150 (3.81) 0.135 (3.43) 0.120 (3.05) 0.015 (0.38) MIN 0.150 (3.81) 0.130 (3.30) 0.110 (2.79) 0.022 (0.56) 0.018 (0.46) 0.014 (0.36) 5 PIN 1 0.325 (8.26) 0.310 (7.87) 0.300 (7.62) 0.180 (4.57) MAX 0.055 (1.40) MAX 0.023 (0.58) 0.014 (0.36) SEATING 0.070 (1.78) PLANE 0.030 (0.76) 0.015 (0.38) 0.008 (0.20) 15 0 CONTROLLING DIMENSIONS ARE IN INCH; MILLIMETERS DIMENSIONS (IN PARENTHESES) ARE ROUNDED-OFF MILLIMETER EQUIVALENTS FOR REFERENCE ONLY AND ARE NOT APPROPRIATE FOR USE IN DESIGN COMPLIANT TO JEDEC STANDARDS MO-095AA CONTROLLING DIMENSIONS ARE IN INCHES; MILLIMETERS DIMENSIONS (IN PARENTHESES) 8-Lead Standard Small Outline Package [SOIC] Narrow Body (RN-8) 8-Lead Metal Can [TO-99] (H-8) Dimensions shown in inches and (millimeters) Dimensions shown in millimeters and (inches) REFERENCE PLANE 5.00 (0.1968) 4.80 (0.1890) 5 1 4 1.27 (0.0500) BSC 0.25 (0.0098) 0.10 (0.0040) COPLANARITY SEATING 0.10 PLANE 6.20 (0.2440) 5.80 (0.2284) 0.50 (0.0196) ⴛ 45ⴗ 0.25 (0.0099) 8ⴗ 0.25 (0.0098) 0ⴗ 1.27 (0.0500) 0.41 (0.0160) 0.19 (0.0075) 0.0400 (1.02) MAX 0.0400 (1.02) 0.0100 (0.25) COMPLIANT TO JEDEC STANDARDS MS-012AA CONTROLLING DIMENSIONS ARE IN MILLIMETERS; INCH DIMENSIONS (IN PARENTHESES) ARE ROUNDED-OFF MILLIMETER EQUIVALENTS FOR REFERENCE ONLY AND ARE NOT APPROPRIATE FOR USE IN DESIGN REV. E 0.1000 (2.54) BSC 0.1600 (4.06) 0.1400 (3.56) 5 1.75 (0.0688) 1.35 (0.0532) 0.51 (0.0201) 0.33 (0.0130) 0.2500 (6.35) MIN 0.0500 (1.27) MAX 0.3700 (9.40) 0.3350 (8.51) 0.3350 (8.51) 0.3050 (7.75) 4.00 (0.1574) 3.80 (0.1497) 8 0.1850 (4.70) 0.1650 (4.19) 0.5000 (12.70) MIN 6 4 0.2000 (5.08) BSC 3 7 2 0.0190 (0.48) 0.0160 (0.41) 0.1000 (2.54) BSC 0.0210 (0.53) 0.0160 (0.41) 0.0450 (1.14) 0.0270 (0.69) 8 1 0.0340 (0.86) 0.0280 (0.71) 45 BSC BASE & SEATING PLANE COMPLIANT TO JEDEC STANDARDS MO-002AK CONTROLLING DIMENSIONS ARE IN INCHES; MILLIMETERS DIMENSIONS (IN PARENTHESES) ARE ROUNDED-OFF EQUIVALENTS FOR REFERENCE ONLY AND ARE NOT APPROPRIATE FOR USE IN DESIGN –13– AD711 Revision History Location Page 10/02—Data Sheet changed from REV. D to REV. E. Edits to ABSOLUTE MAXIMUM RATINGS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 10/02—Data Sheet changed from REV. C to REV. D. Edits to CONNECTION DIAGRAMS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1 Updated OUTLINE DIMENSIONS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 13 5/02—Data Sheet changed from REV. B to REV. C. Change from Small Outline Package (R-8) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 13 Edits to ORDERING GUIDE . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 Deleted METALLIZATION PHOTOGRAPH . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 –14– REV. E –15– –16– PRINTED IN U.S.A. C00832–0–10/02(E)