AD7856ARSZ

a FUNCTIONAL BLOCK DIAGRAM ..... AIN1 I/P MUX AD7856 T/H DVDD AIN8 4.096V REFERENCE COMP REFIN/REFOUT APPLICATIONS Battery-Powered Systems (Personal Digital Assistants, Medical Instruments, Mobile Communications) Pen Computers Instrumentation and Control Systems High Speed Modems AGND AVDD .... FEATURES Single 5␣ V Supply 285 kSPS Throughput Rate Self- and System Calibration with Autocalibration on Power-Up Eight Single-Ended or Four Pseudo-Differential Inputs Low Power: 60 mW Typ Automatic Power-Down After Conversion (2.5␣ ␮W Typ) Flexible Serial Interface: 8051/SPI™/QSPI™/␮P Compatible 24-Lead DIP, SOIC and SSOP Packages 5 V Single Supply, 8-Channel 14-Bit 285 kSPS Sampling ADC AD7856 DGND BUF CREF1 CHARGE REDISTRIBUTION DAC CLKIN CREF2 CAL SAR + ADC CONTROL CALIBRATION MEMORY AND CONTROLLER CONVST BUSY SLEEP GENERAL DESCRIPTION The AD7856 is a high speed, low power, 14-bit ADC that operates from a single 5 V power supply. The ADC powers up with a set of default conditions at which time it can be operated as a read only ADC. The ADC contains self-calibration and system calibration options to ensure accurate operation over time and temperature and it has a number of power-down options for low power applications. The AD7856 is capable of 285 kHz throughput rate. The input track-and-hold acquires a signal in 500 ns and features a pseudodifferential sampling scheme. The AD7856 voltage range is 0 to VREF with straight binary output coding. Input signal range is to the supply and the part is capable of converting full power signals to 10 MHz. CMOS construction ensures low power dissipation of typically 60 mW for normal operation and 5.1 mW in power-down mode at 10 kSPS throughput rate. The part is available in 24-lead, 0.3 inch-wide dual in-line package (DIP), 24-lead small outline (SOIC) and 24-lead small shrink outline (SSOP) packages. Please see page 31 for data sheet index. SERIAL INTERFACE/CONTROL REGISTER SYNC DIN DOUT SCLK PRODUCT HIGHLIGHTS 1. Single 5 V supply. 2. Automatic calibration on power-up. 3. Flexible power management options including automatic power-down after conversion. 4. Operates with reference voltages from 1.2 V to VDD. 5. Analog input range from 0 V to VDD. 6. Eight single-ended or four pseudo-differential input channels. 7. Self- and system calibration. 8. Versatile serial I/O port (SPI/QSPI/8051/µP). SPI and QSPI are trademarks of Motorola, Inc. REV. A Information furnished by Analog Devices is believed to be accurate and reliable. However, no responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other rights of third parties which may result from its use. No license is granted by implication or otherwise under any patent or patent rights of Analog Devices. One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A. Tel: 781/329-4700 World Wide Web Site: http://www.analog.com Fax: 781/326-8703 © Analog Devices, Inc., 1998 AD7856–SPECIFICATIONS1, 2 A Grade: fCLKIN = 6 MHz, (–40ⴗC to +105ⴗC), fSAMPLE = 285 kHz; K Grade: fCLKIN = 4 MHz, (0ⴗC to +105ⴗC), fSAMPLE = 102 kHz; (AVDD = DVDD = +5.0 V ⴞ 5%, REFIN/REFOUT = 4.096 V External Reference unless otherwise noted, SLEEP = Logic High; TA = TMIN to TMAX, unless otherwise noted.) Specifications apply for Mode 2 operation, standard 3-wire SPI interface; refer to Detailed Timing section for Mode 1 Specifications. A Version1 K Version1 Units Test Conditions/Comments 78 –86 –87 78 –86 –87 dB min dB max dB max fIN = 10 kHz 79.5 dB typ –95 dB typ –95 dB typ –86 –86 –90 –90 –90 –90 dB typ dB typ dB typ fa = 9.983 kHz, fb = 10.05 kHz fa = 9.983 kHz, fb = 10.05 kHz VIN = 25 kHz 14 14 ±2 ±2 Bits LSB max LSB typ LSB max LSB typ LSB max LSB typ LSB max LSB max LSB typ LSB max 0 to VREF 0 to VREF Volts ±1 20 ±1 20 µA max pF typ REFERENCE INPUT/OUTPUT REFIN Input Voltage Range Input Impedance REFOUT Output Voltage REFOUT Tempco 4.096/VDD 150 3.696/4.496 20 2.3/VDD 150 3.696/4.496 20 V min/max kΩ typ V min/max ppm/°C typ LOGIC INPUTS Input High Voltage, VINH Input Low Voltage, VINL Input Current, IIN Input Capacitance, CIN4 VDD – 1.0 0.4 ±1 10 VDD – 1.0 0.4 ±1 10 V min V max µA max pF max LOGIC OUTPUTS Output High Voltage, VOH Output Low Voltage, VOL Floating-State Leakage Current Floating-State Output Capacitance4 Output Coding VDD – 0.4 VDD – 0.4 0.4 0.4 ±1 ±1 10 10 Straight (Natural) Binary V min V max µA max pF max ISOURCE = 200 µA ISINK = 0.8 mA CONVERSION RATE Conversion Time Track/Hold Acquisition Time 3.5 0.33 µs max µs min 21 CLKIN Cycles Parameter DYNAMIC PERFORMANCE Signal to Noise + Distortion Ratio3 (SNR) Total Harmonic Distortion (THD) Peak Harmonic or Spurious Noise Intermodulation Distortion (IMD) Second Order Terms Third Order Terms Channel-to-Channel Isolation DC ACCURACY Resolution Integral Nonlinearity Any Channel ±2 Differential Nonlinearity ±2 Offset Error ± 10 Offset Error Match Positive Full-Scale Error ± 10 Positive Full-Scale Error Match ANALOG INPUT Input Voltage Ranges Leakage Current Input Capacitance ±2 ± 10 ±5 ±3 ± 10 5.25 0.5 –2– 4.096 V External Reference, VDD = 5 V Guaranteed No Missed Codes to 13 Bits. i.e., AIN(+) – AIN(–) = 0 to VREF, AIN(–) Can Be Biased Up, but AIN(+) Cannot Go Below AIN(–) Functional from 1.2 V Resistor Connected to Internal Reference Node Typically 10 nA, VIN = 0 V or VDD REV. A AD7856 Parameter POWER PERFORMANCE AVDD, DVDD IDD Normal Mode5 Sleep Mode6 With External Clock On With External Clock Off Normal Mode Power Dissipation Sleep Mode Power Dissipation With External Clock On With External Clock Off SYSTEM CALIBRATION Offset Calibration Span7 Gain Calibration Span7 A Version1 K Version1 Units +4.75/+5.25 +4.75/+5.25 V min/max 17 17 mA max AVDD = DVDD = 4.75 V to 5.25 V. Typically 12 mA 30 10 µA typ 400 500 µA typ 5 5 µA max 200 200 µA typ 89.25 89.25 mW max Full Power-Down. Power Management Bits in Control Register Set as PMGT1 = 1, PMGT0 = 0 Partial Power-Down. Power Management Bits in Control Register Set as PMGT1 = 1, PMGT0 = 1 Typically 0.5 µA. Full Power-Down. Power Management. Bits in Control Register Set as PMGT1 = 1, PMGT0 = 0 Partial Power-Down. Power Management Bits in Control Register Set as PMGT1 = 1, PMGT0 = 1 VDD = 5.25 V. Typically 60 mW; SLEEP = VDD 52.5 26.25 52.5 26.25 µW typ µW max VDD = 5.25 V. SLEEP = 0 V VDD = 5.25 V. Typically 5.25 µW; SLEEP = 0 V V max/min V max/min Allowable Offset Voltage Span for Calibration Allowable Full-Scale Voltage Span for Calibration +0.0375 × VREF/–0.0375 × VREF +1.01875 × VREF/–0.98125 × VREF Test Conditions/Comments NOTES 1 Temperature ranges as follows: A Version: –40°C to +105°C. K Version: 0°C to +105°C. 2 Specifications apply after calibration. 3 SNR calculation includes distortion and noise components. 4 Sample tested @ +25°C to ensure compliance. 5 All digital inputs @ DGND except for CONVST, SLEEP, CAL and SYNC @ DVDD. No load on the digital outputs. Analog inputs @ AGND. 6 CLKIN @ DGND when external clock off. All digital inputs @ DGND except for CONVST, SLEEP, CAL, and SYNC @ DVDD. No load on the digital outputs. Analog inputs @ AGND. 7 The Offset and Gain Calibration Spans are defined as the range of offset and gain errors that the AD7856 can calibrate. Note also that these are voltage spans and are not absolute voltages (i.e., the allowable system offset voltage presented at AIN(+) for the system offset error to be adjusted out will be AIN(–) ± 0.0375 × VREF, and the allowable system full-scale voltage applied between AIN(+) and AIN(–) for the system full-scale voltage error to be adjusted out will be V REF ± 0.01875 × VREF). This is explained in more detail in the Calibration section of the data sheet. Specifications subject to change without notice. REV. A –3– AD7856 TIMING SPECIFICATIONS1 (V DD = 5 V; TA = TMIN to TMAX, unless otherwise noted. A Grade: fCLKIN = 6 MHz; K Grade: fCLKIN = 4 MHz.) ␣ ␣ ␣ Limit at TMIN, TMAX A Version K Version Units Description 500 4 4 100 50 5.25 –0.4 tSCLK ± 0.4 tSCLK 50 50 75 40 20 0.4 tSCLK 0.4 tSCLK 30 30/0.4 tSCLK 50 90 50 2.5 tCLKIN 2.5 tCLKIN 62.5 kHz min MHz max MHz max ns min ns max µs max ns min ns min/max ns max ns max ns max ns min ns min ns min ns min ns min ns min/max ns max ns max ns max ns max ns max ms typ Master Clock Frequency t125 t13 t146 t15 t16 tCAL 500 6 6 100 50 3.5 –0.4 tSCLK ± 0.4 tSCLK 30 30 45 30 20 0.4 tSCLK 0.4 tSCLK 30 30/0.4 tSCLK 50 90 50 2.5 tCLKIN 2.5 tCLKIN 41.7 tCAL1 37.04 55.5 ms typ tCAL2 4.63 6.94 ms typ Parameter fCLKIN2 fSCLK t1 3 t2 tCONVERT t3 t4 4 t5 4 t6 4 t7 t8 t9 t10 t11 CONVST Pulsewidth CONVST↓ to BUSY↑ Propagation Delay Conversion Time = 20 tCLKIN SYNC↓ to SCLK↓ Setup Time (Noncontinuous SCLK Input) SYNC↓ to SCLK↓ Setup Time (Continuous SCLK Input) Delay from SYNC↓ Until DOUT 3-State Disabled Delay from SYNC↓ Until DIN 3-State Disabled Data Access Time After SCLK↓ Data Setup Time Prior to SCLK↑ Data Valid to SCLK Hold Time SCLK High Pulsewidth SCLK Low Pulsewidth SCLK↑ to SYNC↑ Hold Time (Noncontinuous SCLK) (Continuous SCLK) Delay from SYNC↑ Until DOUT 3-State Enabled Delay from SCLK↑ to DIN Being Configured as Output Delay from SCLK↑ to DIN Being Configured as Input CAL↑ to BUSY↑ Delay CONVST↓ to BUSY↑ Delay in Calibration Sequence Full Self-Calibration Time, Master Clock Dependent (250026 tCLKIN) Internal DAC Plus System Full-Scale Cal Time, Master Clock Dependent (222228 tCLKIN) System Offset Calibration Time, Master Clock Dependent (27798 tCLKIN) NOTES 1 Sample tested at +25°C to ensure compliance. All input signals are specified with tr = tf = 5 ns (10% to 90% of V DD) and timed from a voltage level of 1.6 V. See Table X and timing diagrams for different interface modes and calibration. 2 Mark/Space ratio for the master clock input is 40/60 to 60/40. 3 The CONVST pulsewidth here only applies for normal operation. When the part is in power-down mode, a different CONVST pulsewidth will apply (see Power-Down section). 4 Measured with the load circuit of Figure 1 and defined as the time required for the output to cross 0.8 V or 2.4 V. 5 t12 is derived form the measured time taken by the data outputs to change 0.5 V when loaded with the circuit of Figure 1. The measured number is then extrapolated back to remove the effects of charging or discharging the 100 pF capacitor. This means that the time, t 12, quoted in the timing characteristics is the true bus relinquish time of the part and is independent of the bus loading. 6 t14 is derived form the measured time taken by the data outputs to change 0.5 V when loaded with the circuit of Figure 1. The measured number is then extrapolated back to remove the effects of charging or discharging the 100 pF capacitor. This means that the time, t 14, quoted in the Timing Characteristics is the true delay of the part in turning off the output drivers and configuring the DIN line as an input. Once this time has elapsed the user can drive the DIN line knowing that a bus conflict will not occur. Specifications subject to change without notice. –4– REV. A AD7856 TYPICAL TIMING DIAGRAMS 1.6mA Figures 2 and 3 show typical read and write timing diagrams for serial Interface Mode 2. The reading and writing occurs after conversion in Figure 2, and during conversion in Figure 3. To attain the maximum sample rate of 285 kHz, reading and writing must be performed during conversion as in Figure 3. At least 330 ns acquisition time must be allowed (the time from the falling edge of BUSY to the next rising edge of CONVST) before the next conversion begins to ensure that the part is settled to the 14-bit level. If the user does not want to provide the CONVST signal, the conversion can be initiated in software by writing to the control register. TO OUTPUT PIN IOL +2.1V CL 100pF 200mA IOL Figure 1. Load Circuit for Digital Output Timing Specifications tCONVERT = 3.5ms MAX, 5.25ms MAX FOR K VERSION t1 = 100ns MIN, t4 = 30/50ns MAX A/K, t7 = 30/40ns MIN A/K t1 CONVST (I/P) tCONVERT t2 BUSY (O/P) SYNC (I/P) t3 SCLK (I/P) 5 1 t4 THREE-STATE DOUT (O/P) 6 t6 t6 DB15 DB11 16 t10 t12 DB0 THREESTATE t8 t7 DB15 DIN (I/P) t11 t9 DB0 DB11 Figure 2. Timing Diagram for Interface Mode 2 (Reading/Writing After Conversion) tCONVERT = 3.5ms MAX, 5.25ms MAX FOR K VERSION t1 = 100ns MIN, t4 = 30/50ns MAX A/K, t7 = 30/40ns MIN A/K t1 CONVST (I/P) tCONVERT t2 BUSY (O/P) SYNC (I/P) t3 SCLK (I/P) 5 1 t4 DOUT (O/P) t6 THREE-STATE 6 t6 16 t10 t12 DB0 DB11 DB15 t7 DIN (I/P) t11 t9 THREESTATE t8 DB11 DB15 DB0 Figure 3. Timing Diagram for Interface Mode 2 (Reading/Writing During Conversion) REV. A –5– AD7856 ABSOLUTE MAXIMUM RATINGS 1 ORDERING GUIDE (TA = +25°C unless otherwise noted) AVDD to AGND . . . . . . . . . . . . . . . . . . . . . . . ␣ –0.3 V to +7 V DVDD to DGND . . . . . . . . . . . . . . . . . . . . . . . ␣ –0.3 V to +7 V AVDD to DVDD␣ . . . . . . . . . . . . . . . . . . . . . . . –0.3 V to +0.3 V Analog Input Voltage to AGND . . . . ␣ –0.3 V to AVDD + 0.3 V Digital Input Voltage to DGND . . . . –0.3 V to DVDD + 0.3 V Digital Output Voltage to DGND . . . –0.3 V to DVDD + 0.3 V REFIN/REFOUT to AGND . . . . . . . . . ␣ –0.3 V to AVDD + 0.3 V Input Current to Any Pin Except Supplies2␣ . . . . . . . ± 10 mA Operating Temperature Range Commercial A Version . . . . . . . . . . . . . . . . . . . . . . . . . –40°C to +105°C K Version . . . . . . . . . . . . . . . . . . . . . . . . . . . 0°C to +105°C Storage Temperature Range . . . . . . . . . . . . –65°C to +150°C Junction Temperature . . . . . . . . . . . . . . . . . . . . . . . . . +150°C Plastic DIP Package, Power Dissipation . . . . . . . . . . 450 mW θJA Thermal Impedance . . . . . . . . . . . . . . . . . . . . 105°C/W θJC Thermal Impedance . . . . . . . . . . . . . . . . . . . . 34.7°C/W Lead Temperature, (Soldering, 10 secs) . . . . . . . . . +260°C SOIC, SSOP Package, Power Dissipation . . . . . . . . . 450 mW θJA Thermal Impedance . 75°C/W (SOIC) 115°C/W (SSOP) θJC Thermal Impedance . . 25°C/W (SOIC) 35°C/W (SSOP) Lead Temperature, Soldering Vapor Phase (60 secs) . . . . . . . . . . . . . . . . . . . . . . . +215°C Infrared (15 secs) . . . . . . . . . . . . . . . . . . . . . . . . . . . +220°C ESD . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . >1 kV Model AD7856AN AD7856AR AD7856KR AD7856ARS EVAL-AD7856CB3 EVAL-CONTROL BOARD4 Linearity Error (LSB)1 Package Options2 ± 2 typ ± 2 typ ±2 ± 2 typ N-24 R-24 R-24 RS-24 NOTES 1 Linearity error here refers to integral linearity error. 2 N = Plastic DIP; R = SOIC; RS = SSOP. 3 This can be used as a stand-alone evaluation board or in conjunction with the EVAL-CONTROL BOARD for evaluation/demonstration purposes. 4 This board is a complete unit allowing a PC to control and communicate with all Analog Devices evaluation boards ending in the CB designators. PIN CONFIGURATIONS (DIP, SOIC AND SSOP) NOTES 1 Stresses above those listed under Absolute Maximum Ratings may cause permanent damage to the device. This is a stress rating only; functional operation of the device at these or any other conditions above those listed in the operational sections of this specification is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability. 2 Transient currents of up to 100 mA will not cause SCR latch-up. CONVST 1 24 SYNC BUSY 2 23 SCLK SLEEP 3 22 CLKIN REFIN/REFOUT 4 AVDD 5 20 DOUT AGND 6 TOP VIEW 19 DGND CREF1 7 (Not to Scale) 18 DVDD CREF2 8 17 CAL AIN1 9 16 AIN8 AIN2 10 15 AIN7 AIN3 11 14 AIN6 AIN4 12 13 AIN5 CAUTION ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily accumulate on the human body and test equipment and can discharge without detection. Although the AD7856 features proprietary ESD protection circuitry, permanent damage may occur on devices subjected to high energy electrostatic discharges. Therefore, proper ESD precautions are recommended to avoid performance degradation or loss of functionality. –6– 21 DIN AD7856 WARNING! ESD SENSITIVE DEVICE REV. A AD7856 PIN FUNCTION DESCRIPTIONS Pin Mnemonic Description 1 CONVST 2 BUSY 3 SLEEP 4 REFIN/REFOUT 5 6 7 AVDD AGND CREF1 8 CREF2 9–16 AIN1–AIN8 17 CAL 18 19 20 21 DVDD DGND DOUT DIN 22 CLKIN 23 24 SCLK SYNC Convert Start. Logic Input. A low to high transition on this input puts the track/hold into its hold mode and starts conversion. When this input is not used, it should be tied to DVDD. Busy Output. The busy output is triggered high by the falling edge of␣ CONVST or rising edge of CAL, and remains high until conversion is completed. BUSY is also used to indicate when the AD7856 has completed its on-chip calibration sequence. Sleep Input/Low Power Mode. A Logic 0 initiates a sleep, and all circuitry is powered down, including the internal voltage reference, provided there is no conversion or calibration being performed. Calibration data is retained. A Logic 1 results in normal operation. See Power-Down section for more details. Reference Input/Output. This pin is connected to the internal reference through a series resistor and is the reference source for the analog-to-digital converter. The nominal reference voltage is 4.096 V and this appears at the pin. This pin can be overdriven by an external reference or can be taken as high as AVDD. When this pin is tied to AVDD, or when an externally applied reference approaches AVDD, the CREF1 pin should also be tied to AVDD. Analog Positive Supply Voltage, +5.0 V ± 5%. Analog Ground. Ground reference for track/hold, reference and DAC. Reference Capacitor (0.1 µF Multilayer Ceramic in parallel with a 470 nF NPO type). This external capacitor is used as a charge source for the internal DAC. The capacitor should be tied between the pin and AGND. Reference Capacitor (0.01 µF Multilayer Ceramic). This external capacitor is used in conjunction with the on-chip reference. The capacitor should be tied between the pin and AGND. Analog Inputs. Eight analog inputs that can be used as eight single-ended inputs (referenced to AGND) or four pseudo-differential inputs. Channel configuration is selected by writing to the control register. Both the positive and negative inputs cannot go below AGND or above AVDD at any time. Also the positive input cannot go below the negative input. See Table III for channel selection. Calibration Input. This pin has an internal pull-up current source of 0.15 µA. A falling edge on this pin resets all calibration control logic and initiates a calibration on its rising edge. There is the option of connecting a 10 nF capacitor from this pin to DGND to allow for an automatic self-calibration on power-up. This input overrides all other internal operations. If the autocalibration is not required, this pin should be tied to a logic high. Digital Supply Voltage, +5.0 V ± 5%. Digital Ground. Ground reference point for digital circuitry. Serial Data Output. The data output is supplied to this pin as a 16-bit serial word. Serial Data Input. The data to be written is applied to this pin in serial form (16-bit word). This pin can act as an input pin or as a I/O pin depending on the serial interface mode the part is in (see Table X). Master clock signal for the device (A Grade: 6 MHz; K Grade: 4 MHz). Sets the conversion and calibration times. Serial Port Clock. Logic Input. The user must provide a serial clock on this input. Frame Sync. Logic Input. This pin is level triggered active low and frames the serial clock for the read and write operations (see Table IX). REV. A –7– AD7856 TERMINOLOGY1 Integral Nonlinearity Total Harmonic Distortion Total Harmonic Distortion (THD) is the ratio of the rms sum of harmonics to the fundamental. For the AD7856, it is defined as: This is the maximum deviation from a straight line passing through the endpoints of the ADC transfer function. The endpoints of the transfer function are zero scale, a point 1/2 LSB below the first code transition, and full scale, a point 1/2 LSB above the last code transition. 2 2 2 2 2 V 2 +V 3 +V 4 +V 5 +V 6 V1 where V1 is the rms amplitude of the fundamental and V2, V3, V4, V5 and V6 are the rms amplitudes of the second through the sixth harmonics. THD (dB) = 20 log Differential Nonlinearity This is the difference between the measured and the ideal 1 LSB change between any two adjacent codes in the ADC. Peak Harmonic or Spurious Noise Total Unadjusted Error This is the deviation of the first code transition (00␣ .␣ .␣ .␣ 000 to 00␣ .␣ .␣ .␣ 001) from the ideal AIN(+) voltage (AIN(–) + 1/2 LSB). Peak harmonic or spurious noise is defined as the ratio of the rms value of the next largest component in the ADC output spectrum (up to fS/2 and excluding dc) to the rms value of the fundamental. Normally, the value of this specification is determined by the largest harmonic in the spectrum, but for parts where the harmonics are buried in the noise floor, it will be a noise peak. Positive Full-Scale Error Intermodulation Distortion This is the deviation of the last code transition from the ideal AIN(+) voltage (AIN(–) + Full Scale – 1.5 LSB) after the offset error has been adjusted out. With inputs consisting of sine waves at two frequencies, fa and fb, any active device with nonlinearities will create distortion products at sum and difference frequencies of mfa ± nfb where m, n = 0, 1, 2, 3, etc. Intermodulation distortion terms are those for which neither m nor n are equal to zero. For example, the second order terms include (fa + fb) and (fa – fb), while the third order terms include (2fa + fb), (2fa – fb), (fa + 2fb) and (fa – 2fb). This is the deviation of the actual code from the ideal code taking all errors into account (Gain, Offset, Integral Nonlinearity and other errors) at any point along the transfer function. Unipolar Offset Error Channel-to-Channel Isolation Channel-to-channel isolation is a measure of crosstalk between the channels. It is measured by applying a full-scale 25 kHz signal to the other seven channels and determining how much that signal is attenuated in the channel of interest. The figure given is the worst case for all channels. Track/Hold Acquisition Time The track/hold amplifier returns into track mode and the end of conversion. Track/Hold acquisition time is the time required for the output of the track/hold amplifier to reach its final value, within ± 1/2 LSB, after the end of conversion. Signal to (Noise + Distortion) Ratio This is the measured ratio of signal to (noise + distortion) at the output of the A/D converter. The signal is the rms amplitude of the fundamental. Noise is the sum of all nonfundamental signals up to half the sampling frequency (fS/2), excluding dc. The ratio is dependent on the number of quantization levels in the digitization process; the more levels, the smaller the quantization noise. The theoretical signal to (noise + distortion) ratio for an ideal N-bit converter with a sine wave input is given by: Testing is performed using the CCIF standard where two input frequencies near the top end of the input bandwidth are used. In this case, the second order terms are usually distanced in frequency from the original sine waves, while the third order terms are usually at a frequency close to the input frequencies. As a result, the second and third order terms are specified separately. The calculation of the intermodulation distortion is as per the THD specification where it is the ratio of the rms sum of the individual distortion products to the rms amplitude of the sum of the fundamentals expressed in dBs. Full Power Bandwidth The Full Power Bandwidth (FPBW) of the AD7856 is that frequency at which the amplitude of the reconstructed (using FFTs) fundamental (neglecting harmonics and SNR) is reduced by 3 dB for a full-scale input. Signal to (Noise + Distortion) = (6.02 N + 1.76) dB Thus for a 14-bit converter, this is 86 dB. NOTE 1 AIN(+) refers to the positive input of the pseudo-differential pair, and AIN(–) refers to the negative analog input of the pseudo-differential pair or to AGND depending on the channel configuration. –8– REV. A AD7856 ON-CHIP REGISTERS The AD7856 powers up with a set of default conditions. The only writing that is required is to select the channel configuration. Without performing any other write operations the AD7856 still retains the flexibility for performing a full power-down, and a full self-calibration. Extra features and flexibility such as performing different power-down options, different types of calibrations including system calibration, and software conversion start can be selected by further writing to the part. The AD7856 contains a Control Register, ADC Output Data Register, Status Register, Test Register and ten Calibration Registers. The control register is write only, the ADC output data register and the status register are read only, and the test and calibration registers are both read/write registers. The Test Register is used for testing the part and should not be written to. Addressing the On-Chip Registers Writing A write operation to the AD7856 consists of 16 bits. The two MSBs, ADDR0 and ADDR1, are decoded to determine which register is addressed, and the subsequent 14 bits of data are written to the addressed register. It is not until all 16 bits are written that the data is latched into the addressed registers. Table I shows the decoding of the address bits while Figure 4 shows the overall write register hierarchy. Table I. Write Register Addressing ADDR1 ADDR0 Comment 0 0 0 1 1 0 1 1 This combination does not address any register so the subsequent 14 data bits are ignored. This combination addresses the TEST REGISTER. The subsequent 14 data bits are written to the test register. This combination addresses the CALIBRATION REGISTERS. The subsequent 14 data bits are written to the selected calibration register. This combination addresses the CONTROL REGISTER. The subsequent 14 data bits are written to the control register. Reading To read from the various registers the user must first write to Bits 6 and 7 in the Control Register, RDSLT0 and RDSLT1. These bits are decoded to determine which register is addressed during a read operation. Table II shows the decoding of the read address bits while Figure 5 shows the overall read register hierarchy. The power-up status of these bits is 00 so that the default read will be from the ADC output data register. Once the read selection bits are set in the Control Register, all subsequent read operations that follow will be from the selected register until the read selection bits are changed in the Control Register. Table II. Read Register Addressing RDSLT1 RDSLT0 Comment 0 0 0 1 1 1 0 1 All successive read operations will be from ADC OUTPUT DATA REGISTER. This is the powerup default setting. There will always be two leading zeros when reading from the ADC Output Data Register. All successive read operations will be from TEST REGISTER. All successive read operations will be from CALIBRATION REGISTERS. All successive read operations will be from STATUS REGISTER. ADDR1, ADDR0 DECODE 01 10 TEST REGISTER GAIN(1) OFFSET(1) DAC(8) CALSLT1, CALSLT0 DECODE 00 RDSLT1, RDSLT0 DECODE 11 CALIBRATION REGISTERS GAIN(1) OFFSET(1) 01 OFFSET(1) 10 00 CONTROL REGISTER ADC OUTPUT DATA REGISTER 11 CALSLT1, CALSLT0 DECODE 10 TEST REGISTER GAIN(1) OFFSET(1) DAC(8) GAIN(1) Figure 4. Write Register Hierarchy/Address Decoding REV. A 01 00 11 CALIBRATION REGISTERS GAIN(1) OFFSET(1) 01 OFFSET(1) 10 STATUS REGISTER GAIN(1) 11 Figure 5. Read Register Hierarchy/Address Decoding –9– AD7856 CONTROL REGISTER The arrangement of the Control Register is shown below. The control register is a write only register and contains 14 bits of data. The control register is selected by putting two 1s in ADDR1 and ADDR0. The function of the bits in the control register are described below. The power-up status of all bits is 0. MSB SGL/DIFF CH2 CH1 CH0 PMGT1 PMGT0 RDSLT1 RDSLT0 2/3 MODE CONVST CALMD CALSLT1 CALSLT0 STCAL LSB CONTROL REGISTER BIT FUNCTION DESCRIPTION Bit Mnemonic Comment 13 SGL/DIFF 12 11 10 9 8 7 6 5 CH2 CH1 CH0 PMGT1 PMGT0 RDSLT1 RDSLT0 2/3 MODE A 0 in this bit position configures the input channels in pseudo-differential mode. A 1 in this bit position configures the input channels in single-ended mode (see Table III). These three bits are used to select the channel on which the conversion is performed. The channels can be configured as eight single-ended channels or four pseudo-differential channels. The default selection is AIN1 for the positive input and AIN2 for the negative input (see Table III for channel selection). Power Management Bits. These two bits are used with the SLEEP pin for putting the part into various power-down modes (see Power-Down section for more details). These two bits determine which register is addressed for the read operations (see Table II). 4 CONVST 3 2 1 0 CALMD CALSLT1 CALSLT0 STCAL Interface Mode Select Bit. With this bit set to 0, Interface Mode 2 is enabled. With this bit set to 1, Interface Mode 1 is enabled where DIN is used as an output as well as an input. This bit is set to 0 by default after every read cycle; thus when using the Two-Wire Interface Mode, this bit needs to be set to 1 in every write cycle. Conversion Start Bit. A logic one in this bit position starts a single conversion, and this bit is automatically reset to 0 at the end of conversion. This bit may also used in conjunction with system calibration (see Calibration section.) Calibration Mode Bit. A 0 here selects self-calibration, and a 1 selects a system calibration (see Table IV). Calibration Selection Bits and Start Calibration Bit. These bits have two functions. With the STCAL bit set to 1 the CALSLT1 and CALSLT0 bits determine the type of calibration performed by the part (see Table IV). The STCAL bit is automatically reset to 0 at the end of calibration. With the STCAL bit set to 0 the CALSLT1 and CALSLT0 bits are decoded to address the calibration register for read/write of calibration coefficients (see section on the Calibration Registers for more details). –10– REV. A AD7856 Table III. Channel Selection SGL/DIFF CH2 CH1 CH0 AIN(+)* AIN(–)* 0 0 0 0 0 0 0 0 0 0 0 0 1 1 1 1 0 0 1 1 0 0 1 1 0 1 0 1 0 1 0 1 AIN1 AIN3 AIN5 AIN7 AIN2 AIN4 AIN6 AIN8 AIN2 AIN4 AIN6 AIN8 AIN1 AIN3 AIN5 AIN7 1 1 1 1 1 1 1 1 0 0 0 0 1 1 1 1 0 0 1 1 0 0 1 1 0 1 0 1 0 1 0 1 AIN1 AIN3 AIN5 AIN7 AIN2 AIN4 AIN6 AIN8 AGND AGND AGND AGND AGND AGND AGND AGND *AIN(+) refers to the positive input seen by the AD7856 sample and hold circuit. AIN(–) refers to the negative input seen by the AD7856 sample and hold circuit. Table IV. Calibration Selection CALMD CALSLT1 CALSLT0 Calibration Type 0 0 0 0 0 1 0 0 1 1 1 0 0 1 0 1 0 1 1 1 1 1 0 1 A Full Internal Calibration is initiated where the Internal DAC is calibrated followed by the Internal Gain Error, and finally the Internal Offset Error is calibrated out. This is the default setting. Here the Internal Gain Error is calibrated out followed by the Internal Offset Error calibrated out. This calibrates out the Internal Offset Error only. This calibrates out the Internal Gain Error only. A Full System Calibration is initiated here where first the Internal DAC is calibrated followed by the System Gain Error, and finally the System Offset Error is calibrated out. Here the System Gain Error is calibrated out followed by the System Offset Error. This calibrates out the System Offset Error only. This calibrates out the System Gain Error only. REV. A –11– AD7856 STATUS REGISTER The arrangement of the Status Register is shown below. The status register is a read only register and contains 16 bits of data. The status register is selected by first writing to the control register and putting two 1s in RDSLT1 and RDSLT0. The function of the bits in the status register are described below. The power-up status of all bits is 0. START WRITE TO CONTROL REGISTER SETTING RDSLT0 = RDSLT1 = 1 READ STATUS REGISTER Figure 6. Flowchart for Reading the Status Register MSB ZERO BUSY SGL/DIFF CH2 CH1 CH0 PMGT1 PMGT0 RDSLT1 RDSLT0 2/3 MODE X CALMD CALSLT1 CALSLT0 STCAL LSB STATUS REGISTER BIT FUNCTION DESCRIPTION Bit Mnemonic Comment 15 14 ZERO BUSY This bit is always 0. Conversion/Calibration Busy Bit. When this bit is 1 it indicates that there is a conversion or calibration in progress. When this bit is 0, there is no conversion or calibration in progress. 13 12 11 10 9 8 SGL/DIFF CH2 CH1 CH0 PMGT1 PMGT0 These four bits indicate the channel which is selected for conversion (see Table III). 7 6 RDSLT1 RDSLT0 Both of these bits are always 1, indicating it is the status register which is being read (see Table II). 5 2/3 MODE 4 X Interface Mode Select Bit. With this bit at 0, the device is in Interface Mode 2. With this bit at 1, the device is in Interface Mode 1. This bit is reset to 0 after every read cycle. Don’t care bit. 3 CALMD 2 1 0 CALSLT1 CALSLT0 STCAL Power management bits. These bits, along with the SLEEP pin, will indicate if the part is in a power-down mode or not. See Table VI for description. Calibration Mode Bit. A 0 in this bit indicates a self calibration is selected, and a 1 in this bit indicates a system calibration is selected (see Table IV). Calibration Selection Bits and Start Calibration Bit. The STCAL bit is read as a 1 if a calibration is in progress and as a 0 if there is no calibration in progress. The CALSLT1 and CALSLT0 bits indicate which of the calibration registers are addressed for reading and writing (see section on the Calibration Registers for more details). –12– REV. A AD7856 CALIBRATION REGISTERS CALIBRATION REGISTERS The AD7856 has ten calibration registers in all, eight for the DAC, one for the offset and one for gain. Data can be written to or read from all ten calibration registers. In self- and system calibration the part automatically modifies the calibration registers; only if the user needs to modify the calibration registers should an attempt be made to read from and write to the calibration registers. CAL REGISTER ADDRESS POINTER . . . . . . . (10) When reading from the calibration registers there will always be two leading zeros for each of the registers. When operating in Serial Interface Mode 1 the read operations to the calibration registers cannot be aborted. The full number of read operations must be completed (see section on Serial Interface Mode 1 Timing for more detail). CALSLT0 Comment 0 0 0 1 1 0 This combination addresses the Gain (1), Offset (1) and DAC Registers (8). Ten registers in total. This combination addresses the Gain (1) and Offset (1) Registers. Two registers in total. This combination addresses the Offset Register. One register in total. This combination addresses the Gain Register. One register in total. START WRITE TO CONTROL REGISTER SETTING STCAL = 0 AND CALSLT1, CALSLT0 = 00, 01, 10, 11 CAL REGISTER POINTER IS AUTOMATICALLY RESET WRITE TO CAL REGISTER (ADDR1 = 1, ADDR0 = 0) CAL REGISTER POINTER IS AUTOMATICALLY INCREMENTED Writing to/Reading from the Calibration Registers For writing to the calibration registers a write to the control register is required to set the CALSLT0 and CALSLT1 bits. For reading from the calibration registers a write to the control register is required to set the CALSLT0 and CALSLT1 bits, but also to set the RDSLT1 and RDSLT0 bits to 10 (this addresses the calibration registers for reading). The calibration register pointer is reset upon writing to the control register setting the CALSLT1 and CALSLT0 bits, or upon completion of all the calibration register write/read operations. When reset, it points to the first calibration register in the selected write/ read sequence. The calibration register pointer will point to the gain calibration register upon reset in all but one case, this case being where the offset calibration register is selected on its own (CALSLT1 = 1, CALSLT0 = 0). Where more than one calibration register is being accessed the calibration register pointer will be automatically incremented after each calibration register write/read operation. The order in which the ten calibration registers are arranged is shown in Figure 7. The user may abort at any time before all the calibration register write/read operations are completed, and the next control register write operation will reset the calibration register pointer. The flowchart in Figure 8 shows the sequence for writing to the calibration registers and Figure 9 for reading. REV. A (3) . . . . . . . . . . . . . . Figure 7. Calibration Register Arrangements CALSLT1 1 (2) DAC 1ST MSB REGISTER DAC 8TH MSB REGISTER Table V. Calibration Register Addressing 1 (1) CALIBRATION REGISTER ADDRESS POINTER POSITION IS DETERMINED BY THE NUMBER OF CALIBRATION REGISTERS ADDRESSED AND THE NUMBER OF READ/WRITE OPERATIONS Addressing the Calibration Registers The calibration selection bits in the control register CALSLT1 and CALSLT0 determine which of the calibration registers are addressed (see Table V). The addressing applies to both the read and write operations for the calibration registers. The user should not attempt to read from and write to the calibration registers at the same time. GAIN REGISTER OFFSET REGISTER –13– LAST REGISTER WRITE OPERATION OR ABORT ? NO YES FINISHED Figure 8. Flowchart for Writing to the Calibration Registers AD7856 of the 14 data bits in the offset register is binary weighted: the MSB has a weighting of 5% of the reference voltage, the MSB-1 has a weighting of 2.5%, the MSB-2 has a weighting of 1.25%, and so on down to the LSB, which has a weighting of 0.0006%. This gives a resolution of approximately ± 0.0006% of VREF. More accurately the resolution is ± (0.05 × VREF )/213 volts = ± 0.015 mV, with a 2.5 V reference. The maximum specified offset that can be compensated for is ± 3.75% of the reference voltage but is typically ± 5%, which equates to ± 125 mV with a 2.5 V reference and ± 250 mV with a 5 V reference. START WRITE TO CONTROL REGISTER SETTING STCAL = 0, RDSLT1 = 1, RDSLT0 = 0, AND CALSLT1, CALSLT0 = 00, 01, 10, 11 CAL REGISTER POINTER IS AUTOMATICALLY RESET READ CAL REGISTER Q. If a +20 mV offset is present in the analog input signal and the reference voltage is 2.5 V, what code needs to be written to the offset register to compensate for the offset? CAL REGISTER POINTER IS AUTOMATICALLY INCREMENTED LAST REGISTER READ OPERATION OR ABORT ? A. 2.5 V reference implies that the resolution in the offset register is 5% × 2.5 V/213 = 0.015 mV. +20 mV/0.015 mV = 1310.72; rounding to the nearest number gives 1311. In binary terms this is 0101 0001 1111. Therefore, decrease the offset register by 0101 0001 1111. NO This method of compensating for offset in the analog input signal allows for fine tuning the offset compensation. If the offset on the analog input signal is known, there will be no need to apply the offset voltage to the analog input pins and do a system calibration. The offset compensation can take place in software. YES FINISHED Figure 9. Flowchart for Reading from the Calibration Registers Adjusting the Gain Calibration Register Adjusting the Offset Calibration Register The offset calibration register contains 16 bits, two leading zeros and 14 data bits. By changing the contents of the offset register different amounts of offset on the analog input signal can be compensated for. Increasing the number in the offset calibration register compensates for negative offset on the analog input signal, and decreasing the number in the offset calibration register compensates for positive offset on the analog input signal. The default value of the offset calibration register is approximately 0010 0000 0000 0000. This is not an exact value, but the value in the offset register should be close to this value. Each The gain calibration register contains 16 bits, two leading 0s and 14 data bits. The data bits are binary weighted as in the offset calibration register. The gain register value is effectively multiplied by the analog input to scale the conversion result over the full range. Increasing the gain register compensates for a smaller analog input range and decreasing the gain register compensates for a larger input range. The maximum analog input range for which the gain register can compensate is 1.01875 times the reference voltage; the minimum input range is 0.98125 times the reference voltage. –14– REV. A AD7856 CIRCUIT INFORMATION The AD7856 is a fast, 14-bit single supply A/D converter. The part requires an external 6 MHz/4 MHz master clock (CLKIN), two CREF capacitors, a CONVST signal to start conversion and power supply decoupling capacitors. The part provides the user with track/hold, on-chip reference, calibration features, A/D converter and serial interface logic functions on a single chip. The A/D converter section of the AD7856 consists of a conventional successive-approximation converter based around a capacitor DAC. The AD7856 accepts an analog input range of 0 to +VDD where the reference can be tied to VDD. The reference input to the part is buffered on-chip. A major advantage of the AD7856 is that a conversion can be initiated in software as well as applying a signal to the CONVST pin. Another innovative feature of the AD7856 is self-calibration on power-up, which is initiated having a 0.01 µF capacitor from the CAL pin to DGND, to give superior dc accuracy. The part should be allowed 150 ms after power up to perform this automatic calibration before any reading or writing takes place. The part is available in a 24-pin SSOP package and this offers the user considerable spacing saving advantages over alternative solutions. this CLKIN falling edge. If the 10 ns setup time is not met, the conversion will take 21 CLKIN periods. The maximum specified conversion time is 3.5 µs (6 MHz) 5.25 µs (4 MHz) for the AD7856. When a conversion is completed, the BUSY output goes low, and then the result of the conversion can be read by accessing the data through the serial interface. To obtain optimum performance from the part, the read operation should not occur during the conversion or 500␣ ns prior to the next CONVST rising edge. However, the maximum throughput rates are achieved by reading/writing during conversion, and reading/writing during conversion is likely to degrade the Signal to (Noise + Distortion) by only 0.5 dBs. The AD7856 can operate at throughput rates up to 285 kHz. For the AD7856 a conversion takes 21 CLKIN periods; two CLKIN periods are needed for the acquisition time, giving a full cycle time of 3.66 µs (= 260 kHz, CLKIN = 6 MHz). When using the software conversion start for maximum throughput the user must ensure the control register write operation extends beyond the falling edge of BUSY. The falling edge of BUSY resets the CONVST bit to 0 and allows it to be reprogrammed to 1 to start the next conversion. TYPICAL CONNECTION DIAGRAM CONVERTER DETAILS The master clock for the part must be applied to the CLKIN pin. Conversion is initiated on the AD7856 by pulsing the CONVST input or by writing to the control register and setting the CONVST bit to 1. On the rising edge of CONVST (or at the end of the control register write operation), the on-chip track/hold goes from track to hold mode. The falling edge of the CLKIN signal that follows the rising edge of the CONVST signal initiates the conversion, provided the rising edge of CONVST occurs at least 10 ns typically before this CLKIN edge. The conversion cycle will take 20 CLKIN periods from Figure 10 shows a typical connection diagram for the AD7856. The AGND and DGND pins are connected together at the device for good noise suppression. The CAL pin has a 0.01 µF capacitor to enable an automatic self-calibration on power-up. The conversion result is output in a 16-bit word with two leading zeros followed by the MSB of the 14-bit result. Note that after the AVDD and DVDD power-up the part will require 150 ms for the internal reference to settle and for the automatic calibration on power-up to be completed. For applications where power consumption is a major concern the SLEEP pin can be connected to DGND. See Power-Down section for more detail on low power applications. 6MHz/4MHz OSCILLATOR 285kHz/148kHz PULSE GENERATOR ANALOG SUPPLY +5V 10mF MASTER CLOCK INPUT 0.1mF 0.1mF CONVERSION START INPUT AVDD DVDD 0V TO 4.096V INPUT AIN(–) 470nF 0.1mF 0.01mF CH1 SCLK CREF1 CREF2 OSCILLOSCOPE CLKIN AIN(+) SERIAL CLOCK INPUT CONVST AD7856 CH2 SYNC CH3 FRAME SYNC INPUT DVDD SLEEP SERIAL DATA INPUT CH5 DOUT 0.01mF SERIAL DATA OUTPUT AGND AUTO CAL ON POWER-UP CH4 DIN CAL DGND REFIN/REFOUT INTERNAL/ 0.1mF EXTERNAL REFERENCE AD780/ OPTIONAL REF-198 EXTERNAL REFERENCE DATA GENERATOR PULSE GENERATOR Figure 10. Typical Circuit REV. A –15– 2 LEADING ZEROS FOR ADC DATA AD7856 The equivalent circuit of the analog input section is shown in Figure 11. During the acquisition interval the switches are both in the track position and the AIN(+) charges the 20 pF capacitor through the 125 Ω resistance. On the rising edge of CONVST switches SW1 and SW2 go into the hold position retaining charge on the 20 pF capacitor as a sample of the signal on AIN(+). The AIN(–) is connected to the 20 pF capacitor, and this unbalances the voltage at node A at the input of the comparator. The capacitor DAC adjusts during the remainder of the conversion cycle to restore the voltage at node A to the correct value. This action transfers a charge, representing the analog input signal, to the capacitor DAC which in turn forms a digital representation of the analog input signal. The voltage on the AIN(–) pin directly influences the charge transferred to the capacitor DAC at the hold instant. If this voltage changes during the conversion period, the DAC representation of the analog input voltage will be altered. Therefore it is most important that the voltage on the AIN(–) pin remains constant during the conversion period. Furthermore, it is recommended that the AIN(–) pin is always connected to AGND or to a fixed dc voltage. distortion (THD) that can be tolerated. The THD will increase as the source impedance increases and performance will degrade. Figure 12 shows a graph of the total harmonic distortion versus analog input signal frequency for different source impedances. With the setup as in Figure 13, the THD is at the –90␣ dB level. With a source impedance of 1␣ kΩ and no capacitor on the AIN(+) pin, the THD increases with frequency. –50 THD VS. FREQUENCY FOR DIFFERENT SOURCE IMPEDANCES –60 –70 THD – dB ANALOG INPUT –80 AIN(+) AIN(–) RIN = 10V, 10nF AS IN FIGURE 13 –90 –100 –110 125V RIN = 560V 1 10 20 TRACK 125V CAPACITOR DAC SW1 NODE A SW2 TRACK HOLD COMPARATOR CREF2 Figure 11. Analog Input Equivalent Circuit Acquisition Time The track and hold amplifier enters its tracking mode on the falling edge of the BUSY signal. The time required for the track and hold amplifier to acquire an input signal will depend on how quickly the 20 pF input capacitance is charged. The acquisition time is calculated using the formula: 166 In a single supply application (5 V), the V+ and V– of the op amp can be taken directly from the supplies to the AD7856 which eliminates the need for extra external power supplies. When operating with rail-to-rail inputs and outputs, at frequencies greater than 10 kHz care must be taken in selecting the particular op amp for the application. In particular for single supply applications the input amplifiers should be connected in a gain of –1 arrangement to get the optimum performance. Figure 13 shows the arrangement for a single supply application with a 50 Ω and 10 nF low-pass filter (cutoff frequency 320 kHz) on the AIN(+) pin. Note that the 10 nF is a capacitor with good linearity to ensure good ac performance. Recommended single supply op amp is the AD820. tACQ = 10 × (RIN + 125 Ω) × 20 pF +5V 0.1mF 10mF where RIN is the source impedance of the input signal, and 125 Ω, 20 pF is the input RC. 10kV VIN (0 TO VREF) DC/AC Applications For dc applications high source impedances are acceptable provided there is enough acquisition time between conversions to charge the 20 pF capacitor. The acquisition time can be calculated from the above formula for different source impedances. For example, with RIN = 5 kΩ the required acquisition time will be 1025 ns. 10kV 10kV VREF V+ 50V IC1 V– 10kV AD820 10nF (NPO) TO AIN(+) OF AD7856 Figure 13. Analog Input Buffering Input Range For ac applications, removing high frequency components from the analog input signal is recommended by use of an RC lowpass filter on the AIN(+) pin as shown in Figure 13. In applications where harmonic distortion and signal-to-noise ratio are critical, the analog input should be driven from a low impedance source. Large source impedances will significantly affect the ac performance of the ADC. This may necessitate the use of an input buffer amplifier. The choice of the op amp will be a function of the particular application. When no amplifier is used to drive the analog input the source impedance should be limited to low values. The maximum source impedance will depend on the amount of total harmonic 140 Figure 12. THD vs. Analog Input Frequency 20pF HOLD 100 120 50 80 INPUT FREQUENCY – kHz The analog input range for the AD7856 is 0 V to VREF. The AIN(–) pin on the AD7856 can be biased up above AGND, if required. The advantage of biasing the lower end of the analog input range away from AGND is that the user does not need to have the analog input swing all the way down to AGND. This has the advantage in true single supply applications that the input amplifier does not need to swing all the way down to AGND. The upper end of the analog input range is shifted up by the same amount. Care must be taken so that the bias applied does not shift the upper end of the analog input above the AVDD supply. In the case where the reference is the supply, AVDD, the AIN(–) must be tied to AGND. –16– REV. A AD7856 AIN(+) VIN = 0 TO VREF TRACK AND HOLD AMPLIFIER DOUT AIN(–) ANALOG SUPPLY +5V 10V 10mF 0.01mF STRAIGHT BINARY FORMAT 0.1mF AVDD DVDD CREF1 AD7856 0.1mF 470nF AD7856 CREF2 0.01mF Figure 14. 0 to VREF Input Configuration REFIN/REFOUT 0.1mF Transfer Function For the AD7856 input range the designed code transitions occur midway between successive integer LSB values (i.e., 1/2 LSB, 3/2 LSBs, 5/2 LSBs␣ .␣ .␣ .␣ FS – 3/2 LSBs). The output coding is straight binary, with 1 LSB = FS/16384 = 4.096 V/16384 = 0.25 mV when VREF = 4.096 V. The ideal input/output transfer characteristic is shown in Figure 15. OUTPUT CODE 111...111 111...110 111...101 111...100 000...011 1LSB = Figure 16. Relevant Connections When Using Internal Reference The other option is that the REFIN/REFOUT pin be overdriven by connecting it to an external reference. This is possible due to the series resistance from the REFIN/REFOUT pin to the internal reference. This external reference can have a range that includes AVDD. When using AVDD as the reference source or when an externally applied reference approaches AVDD, the 100 nF capacitor from the REFIN/REFOUT pin to AGND should be as close as possible to the REFIN/REFOUT pin, and also the CREF1 pin should be connected to AVDD to keep this pin at the same level as the reference. The connections for this arrangement are shown in Figure 17. When using AVDD it may be necessary to add a resistor in series with the AVDD supply. This will have the effect of filtering the noise associated with the AVDD supply. FS 16384 ANALOG SUPPLY +5V 000...010 000...001 000...000 10V 0.1mF 0.01mF 10mF 0V 1LSB +FS –1LSB VIN = (AIN(+) – AIN(–)), INPUT VOLTAGE AVDD DVDD CREF1 0.1mF Figure 15. Transfer Characteristic 470nF 10V REFERENCE SECTION For specified performance, it is recommended that when using an external reference this reference should be between 4 V and the analog supply AVDD. The connections for the relevant reference pins are shown in the typical connection diagrams. If the internal reference is being used, the REFIN/REFOUT pin should have a 100 nF capacitor connected to AGND very close to the REFIN/REFOUT pin. These connections are shown in Figure 16. AD7856 0.01mF REFIN/REFOUT 0.1mF Figure 17. Relevant Connections When Using AVDD as the Reference If the internal reference is required for use external to the ADC, it should be buffered at the REFIN/REFOUT pin and a 100 nF capacitor connected from this pin to AGND. The typical noise performance for the internal reference, with 5␣ V supplies is 150␣ nV/√Hz @ 1␣ kHz and dc noise is 100 µV p-p. REV. A CREF2 –17– AD7856 PERFORMANCE CURVES –72 The following performance curves apply to Mode 2 operation only. If a conversion is initiated in software, then a slight degradation in SNR can be expected when in Mode 2 operation. As the sampling instant cannot be guaranteed internally, nonequidistant sampling will occur, resulting in a rise in the noise floor. Initiating conversions in software is not recommended for Mode 1 operation. –74 –76 AVDD = DVDD = 5.0V 100mV p-p SINEWAVE ON AVDD REFIN = 4.098 EXT REFERENCE PSRR – dB –78 Figure 18 shows a typical FFT plot for the AD7856 at 190 kHz sample rate and 10 kHz input frequency. –80 –82 –84 –86 –88 4096 POINT FFT FSAMPLE = 190.476 kHz FIN = 10.091 kHz SNR = 79.2dB –15 –90 0.91 –35 –95 10 20 30 40 50 60 FREQUENCY –kHz 70 80 90 Figure 18. FFT Plot Figure 19 shows the SNR vs. Frequency for 5 V supply and a 4.096 external reference (5 V reference is typically 1 dB better performance). 79 78 S(N+D) RATIO – dB 63.5 74.8 38.3 50.3 INPUT FREQUENCY – kHz 87.4 100 POWER-DOWN OPTIONS –75 77 76 75 0 25.7 Figure 20. PSRR vs. Frequency –55 –115 0 13.4 10 20 80 100 50 120 INPUT FREQUENCY – kHz 140 166 Figure 19. SNR vs. Frequency Figure 20 shows the Power Supply Rejection Ratio versus Frequency for the part. The Power Supply Rejection Ratio is defined as the ratio of the power in ADC output at frequency f to the power of a full-scale sine wave. PSRR (dB) = 10 log (Pf/Pfs) The AD7856 provides flexible power management to allow the user to achieve the best power performance for a given throughput rate. The power management options are selected by programming the power management bits, PMGT1 and PMGT0, in the control register and by use of the SLEEP pin. Table VI summarizes the power-down options that are available and how they can be selected by using either software, hardware or a combination of both. The AD7856 can be fully or partially powered down. When fully powered down, all the on-chip circuitry is powered down and IDD is 1 µA typ. If a partial powerdown is selected, then all the on-chip circuitry except the reference is powered down and IDD is 400 µA typ. The choice of full or partial power-down does not give any significant improvement in throughput with a power-down between conversions. This is discussed in the next section–Power-Up Times. However, a partial power-down does allow the on-chip reference to be used externally even though the rest of the AD7856 circuitry is powered down. It also allows the AD7856 to be powered up faster after a long power-down period when using the on-chip reference (See Power-Up Times–Using On-Chip Reference). When using the SLEEP pin, the power management bits PMGT1 and PMGT0 should be set to zero (default status on power-up). Bringing the SLEEP pin logic high ensures normal operation, and the part does not power down at any stage. This may be necessary if the part is being used at high throughput rates when it is not possible to power down between conversions. If the user wishes to power down between conversions at lower throughput rates (i.e.