AD8013AR-14-REEL7

a FEATURES Three Video Amplifiers in One Package Drives Large Capacitive Load Excellent Video Specifications (RL = 150 ) Gain Flatness 0.1 dB to 60 MHz 0.02% Differential Gain Error 0.06 ° Differential Phase Error Low Power Operates on Single +5 V to +13 V Power Supplies 4 mA/Amplifier Max Power Supply Current High Speed 140 MHz Unity Gain Bandwidth (3 dB) Fast Settling Time of 18 ns (0.1%) 1000 V/ s Slew Rate High Speed Disable Function per Channel Turn-Off Time 30 ns Easy to Use 95 mA Short Circuit Current Output Swing to Within 1 V of Rails APPLICATIONS LCD Displays Video Line Driver Broadcast and Professional Video Computer Video Plug-In Boards Consumer Video RGB Amplifier in Component Systems PRODUCT DESCRIPTION Single Supply, Low Power, Triple Video Amplifier AD8013 PIN CONFIGURATION 14-Pin DIP & SOIC Package DISABLE 1 DISABLE 2 DISABLE 3 +VS +IN 1 –IN 1 OUT 1 1 2 3 4 5 6 7 14 OUT 2 13 –IN 2 12 +IN 2 AD8013 11 –VS 10 +IN 3 9 8 –IN 3 OUT 3 differential gain and phase error of 0.02% and 0.06°. This makes the AD8013 ideal for broadcast and professional video electronics. The AD8013 offers low power of 4 mA per amplifier max and runs on a single +5 V to +13 V power supply. The outputs of each amplifier swing to within one volt of either supply rail to easily accommodate video signals. The AD8013 is unique among current feedback op amps by virtue of its large capacitive load drive. Each op amp is capable of driving large capacitive loads while still achieving rapid settling time. For instance it can settle in 18 ns driving a resistive load, and achieves 40 ns (0.1%) settling while driving 200 pF. The outstanding bandwidth of 140 MHz along with 1000 V/µs of slew rate make the AD8013 useful in many general purpose high speed applications where a single +5 V or dual power supplies up to ± 6.5 V are required. Furthermore the AD8013’s high speed disable function can be used to power down the amplifier or to put the output in a high impedance state. This can then be used in video multiplexing applications. The AD8013 is available in the industrial temperature range of –40°C to +85°C. 500mV 500ns The AD8013 is a low power, single supply, triple video amplifier. Each of the three amplifiers has 30 mA of output current, and is optimized for driving one back terminated video load (150 Ω) each. Each amplifier is a current feedback amplifier and features gain flatness of 0.1 dB to 60 MHz while offering G = +2 RL = 150Ω 0.2 0.1 NORMALIZED GAIN – dB 0 –0.1 –0.2 –0.3 –0.4 –0.5 1M VS = +5V VS = ± 5V 100 9 0 1 0 0% 5V 100M 10M FREQUENCY – Hz 1G Fine-Scale Gain Flatness vs. Frequency, G = +2, RL = 150 Ω Channel Switching Characteristics for a 3:1 Mux REV. A Information furnished by Analog Devices is believed to be accurate and reliable. However, no responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other rights of third parties which may result from its use. No license is granted by implication or otherwise under any patent or patent rights of Analog Devices. © Analog Devices, Inc., 1995 One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A. Tel: 617/329-4700 Fax: 617/326-8703 AD8013–SPECIFICATIONS (@ T = +25 C, R A LOAD = 150 VS , unless otherwise noted) Min 100 110 AD8013A Typ Max 125 140 50 60 400 1000 18 40 Units MHz MHz MHz MHz V/µs V/µs ns ns Model Conditions DYNAMIC PERFORMANCE Bandwidth (3 dB) Bandwidth (0.1 dB) Slew Rate Settling Time to 0.1% No Peaking, G = +2 No Peaking, G = +2 No Peaking, G = +2 No Peaking, G = +2 2 V Step 6 V Step 0 V to +2 V 4.5 V Step, CLOAD = 200 pF RLOAD > 1 kΩ, RFB = 4 kΩ fC = 5 MHz, RL = 1 k fC = 5 MHz, RL = 150 Ω f = 10 kHz f = 10 kHz (–IIN) f = 3.58 MHz, G = +2 f = 3.58 MHz, G = +2 +5 V ±5 V +5 V ±5 V +5 V ±5 V ±5 V ±6 V 600 NOISE/HARMONIC PERFORMANCE Total Harmonic Distortion Input Voltage Noise Input Current Noise Differential Gain (RL = 150 Ω) Differential Phase (RL = 150 Ω) DC PERFORMANCE Input Offset Voltage Offset Drift Input Bias Current (–) Input Bias Current (+) Open-Loop Transresistance ±5 V ±5 V +5 V, ± 5 V +5 V, ± 5 V +5 V1 ±5 V +5 V1 ±5 V +5 V, ± 5 V +5 V, ± 5 V +5 V, ± 5 V +5 V ±5 V –76 –66 3.5 12 0.05 0.02 0.06 0.06 2 7 2 3 800 1.1 M 650 200 150 2 3.8 1.2 52 52 56 56 0.2 5 0.05 0.12 5 10 15 dBc dBc nV/√Hz pA/√Hz % % Degrees Degrees mV µV/°C µA µA kΩ kΩ Ω kΩ kΩ Ω pF ±V +V dB dB µA/V µA/V TMIN to TMAX TMIN to TMAX TMIN to TMAX TMIN to TMAX 650 550 800 k INPUT CHARACTERISTICS Input Resistance Input Capacitance Input Common-Mode Voltage Range Common-Mode Rejection Ratio Input Offset Voltage Input Offset Voltage –Input Current +Input Current OUTPUT CHARACTERISTICS Output Voltage Swing RL = 1 kΩ RL = 150 Ω Output Current Short-Circuit Current Capacitive Load Drive MATCHING CHARACTERISTICS Dynamic Crosstalk Gain Flatness Match DC Input Offset Voltage –Input Bias Current +Input –Input ±5 V ±5 V ±5 V ±5 V +5 V +5 V ±5 V +5 V, ± 5 V +5 V, ± 5 V 3.8 0.4 7 VOL–VEE VCC–VOH VOL–VEE VCC–VOH +5 V ±5 V ±5 V ±5 V 25 0.8 0.8 1.1 1.1 30 30 95 1000 1.0 1.0 1.3 1.3 V V V V mA mA mA pF G = +2, f = 5 MHz f = 20 MHz +5 V, ± 5 V ±5 V +5 V, ± 5 V +5 V, ± 5 V 70 0.1 0.3 1.0 dB dB mV µA –2– REV. A AD8013 Model Conditions POWER SUPPLY Operating Range Quiescent Current/Amplifier Quiescent Current/Amplifier Power Supply Rejection Ratio Input Offset Voltage –Input Current +Input Current DISABLE CHARACTERISTICS Off Isolation Off Output Impedance Turn-On Time Turn-Off Time Switching Threshold Power Down VS = ± 2.5 V to ± 5 V Single Supply Dual Supply +5 V ±5 V ± 6.5 V +5 V ±5 V +5 V, ± 5 V +5 V, ± 5 V +5 V, ± 5 V +5 V, ± 5 V –VS + xV 1.3 70 VS Min +4.2 ± 2.1 3.0 3.4 3.5 0.25 0.3 76 0.03 0.07 –70 12 50 30 1.6 AD8013A Typ Max +13 ± 6.5 3.5 4.0 0.35 0.4 0.2 1.0 Units V V mA mA mA mA mA dB µA/V µA/V dB pF ns ns V f = 6 MHz G = +1 1.9 NOTES 1 The test circuit for differential gain and phase measurements on a +5 V supply is ac coupled. Specifications subject to change without notice. ABSOLUTE MAXIMUM RATINGS 1 Maximum Power Dissipation Supply Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . 13.2 V Total Internal Power Dissipation2 Plastic (N) . . . . . . . . . 1.6 Watts (Observe Derating Curves) Small Outline (R) . . . . 1.0 Watts (Observe Derating Curves) Input Voltage (Common Mode) . . Lower of ± VS or ± 12.25 V Differential Input Voltage . . . . . . . . Output ± 6 V (Clamped) Output Voltage Limit Maximum . . . . . . . . . Lower of (+12 V from –VS) or (+VS) Minimum . . . . . . . . . Higher of (–12.5 V from +VS) or (–VS) Output Short Circuit Duration . . . . . . . . . . . . . . . . . . . . Observe Power Derating Curves Storage Temperature Range N and R Package . . . . . . . . . . . . . . . . . . . –65°C to +125°C Operating Temperature Range AD8013A . . . . . . . . . . . . . . . . . . . . . . . . . . –40°C to +85°C Lead Temperature Range (Soldering 10 sec) . . . . . . . . +300°C NOTES 1 Stresses above those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. This is a stress rating only and functional operation of the device at these or any other conditions above those indicated in the operational section of this specification is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability. 2 Specification is for device in free air: 14-Pin Plastic DIP Package: θJA = 75°C/Watt 14-Pin SOIC Package: θJA = 120°C/Watt The maximum power that can be safely dissipated by the AD8013 is limited by the associated rise in junction temperature. The maximum safe junction temperature for the plastic encapsulated parts is determined by the glass transition temperature of the plastic, about 150°C. Exceeding this limit temporarily may cause a shift in parametric performance due to a change in the stresses exerted on the die by the package. Exceeding a junction temperature of 175°C for an extended period can result in device failure. While the AD8013 is internally short circuit protected, this may not be enough to guarantee that the maximum junction temperature is not exceeded under all conditions. To ensure proper operation, it is important to observe the derating curves. It must also be noted that in (noninverting) gain configurations (with low values of gain resistor), a high level of input overdrive can result in a large input error current, which may result in a significant power dissipation in the input stage. This power must be included when computing the junction temperature rise due to total internal power. 2.5 TJ = +150°C MAXIMUM POWER DISSIPATION – Watts 2.0 14-PIN DIP PACKAGE ORDERING GUIDE Model AD8013AN AD8013AR-14 AD8013AR-14-REEL AD8013AR-14-REEL7 AD8013ACHIPS Temperature Range –40°C to +85°C –40°C to +85°C –40°C to +85°C –40°C to +85°C –40°C to +85°C Package Description 14-Pin Plastic DIP 14-Pin Plastic SOIC 14-Pin Plastic SOIC 14-Pin Plastic SOIC Die Form Package Options N-14 R-14 R-14 R-14 1.5 14-PIN SOIC 1.0 0.5 –50 –40 –30 –20 –10 0 10 20 30 40 50 60 70 AMBIENT TEMPERATURE – °C 80 90 Maximum Power Dissipation vs. Ambient Temperature REV. A –3– AD8013 METALIZATION PHOTO Contact factory for latest dimensions. Dimensions shown in inches and (mm). +IN1 5 –IN1 6 +vs 4 DISABLE 3 3 2 DISABLE 2 OUT1 7 1 DISABLE 1 0.044 (1.13) 14 OUT 2 OUT3 8 –IN3 9 10 +IN3 0.071 (1.81) 11 –VS 12 +IN2 13 –IN2 CAUTION ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily accumulate on the human body and test equipment and can discharge without detection. Although the AD8013 features proprietary ESD protection circuitry, permanent damage may occur on devices subjected to high energy electrostatic discharges. Therefore, proper ESD precautions are recommended to avoid performance degradation or loss of functionality. WARNING! ESD SENSITIVE DEVICE 6 12 COMMON-MODE VOLTAGE RANGE – ± Volts OUTPUT VOLTAGE SWING – V p-p 5 10 NO LOAD 8 RL = 150Ω 6 4 3 2 4 1 2 0 1 2 3 4 5 SUPPLY VOLTAGE – ± Volts 6 7 0 1 2 3 4 5 SUPPLY VOLTAGE – ± Volts 6 7 Figure 1. Input Common-Mode Voltage Range vs. Supply Voltage Figure 2. Output Voltage Swing vs. Supply Voltage –4– REV. A AD8013 10 VS = ±5V INPUT BIAS CURRENT – µA 3 OUTPUT VOLTAGE SWING – V p-p 8 2 1 6 0 –IB –1 4 VS = +5V 2 –2 +IB 0 10 1k 100 LOAD RESISTANCE – Ω 10k –3 –60 –40 –20 0 20 40 60 80 100 JUNCTION TEMPERATURE – °C 120 140 Figure 3. Output Voltage Swing vs. Load Resistance Figure 6. Input Bias Current vs. Junction Temperature 12 2 SUPPLY CURRENT – mA VS = ± 5V 10 INPUT OFFSET VOLTAGE – mV 11 1 0 9 VS = +5V 8 –1 VS = +5V –2 VS = ±5V –3 7 6 –60 –40 –20 0 20 40 60 80 100 JUNCTION TEMPERATURE – °C 120 140 –4 –60 –40 –20 0 20 40 60 80 100 JUNCTION TEMPERATURE – °C 120 140 Figure 4. Total Supply Current vs. Junction Temperature Figure 7. Input Offset Voltage vs. Junction Temperature 11 140 VS = ± 5V SHORT CIRCUIT CURRENT – mA TA = +25°C SUPPLY CURRENT – mA 130 SOURCE 120 SINK 100 10 9 8 90 7 1 2 3 4 5 SUPPLY VOLTAGE – ± Volts 6 7 80 –60 –40 –20 0 20 40 60 80 100 JUNCTION TEMPERATURE – °C 120 140 Figure 5. Supply Current vs. Supply Voltage Figure 8. Short Circuit Current vs. Junction Temperature REV. A –5– AD8013 1k CLOSED-LOOP OUTPUT RESISTANCE – Ω 70 G = +2 100 COMMON-MODE REJECTION – dB 60 VCM R R R R 50 10 40 1 VS = ±5V 30 0.1 20 0.01 100k 1M 10M FREQUENCY – Hz 100M 1G 10 100k 1M 10M FREQUENCY – Hz 100M 1G Figure 9. Closed-Loop Output Resistance vs. Frequency Figure 12. Common-Mode Rejection vs. Frequency 100k 80 70 VS = ±5V 10k POWER SUPPLY REJECTION – dB OUTPUT RESISTANCE – Ω 60 50 +PSR 40 30 20 10 –PSR 1k 100 10 1M 100M 10M FREQUENCY – Hz 1G 0 100k 1M 10M FREQUENCY – Hz 100M 1G Figure 10. Output Resistance vs. Frequency, Disabled State Figure 13. Power Supply Rejection Ratio vs. Frequency 1k 1k VS = ±5V RL = 1k –45 –90 –135 VOLTAGE NOISE nV/ √ Hz CURRENT NOISE pA/ √ Hz 120 100 100 TRANSIMPEDANCE – dB 100 –180 NONINVERTING I 10 INVERTING I 10 80 60 VNOISE 1 100 1 1k 10k FREQUENCY – Hz 100k 1M 40 10k 100k 1M 10M FREQUENCY – Hz 100M 1G Figure 11. Input Current and Voltage Noise vs. Frequency Figure 14. Open-Loop Transimpedance vs. Frequency (Relative to 1 Ω) –6– REV. A PHASE – Degrees 140 0 AD8013 –40 HARMONIC DISTORTION – dBc –50 –60 –70 –80 –90 –100 –110 –120 1k G = +2 VO = 2V p-p VS = ±5V +1 0 2nd RL = 150Ω PHASE VS = +5V G = +1 RL = 150Ω VS = ±5V 0 –90 –180 –270 CLOSED-LOOP GAIN (NORMALIZED) – dB GAIN VS = ±5V –1 –2 –3 VS = +5V –4 –5 2nd RL = 1kΩ 3rd RL = 150Ω 10k 100k 1M FREQUENCY – Hz 3rd RL = 1kΩ 10M 100M –6 1M 100M 10M FREQUENCY – Hz 1G Figure 15. Harmonic Distortion vs. Frequency Figure 18. Closed-Loop Gain and Phase vs. Frequency, G = +1, RL = 150 Ω 1800 1600 1400 VS = ±5V RL = 500Ω 2000 1800 G = +10 SLEW RATE – V/µs G = +10 1600 1400 G = –1 1200 G = +2 1000 800 G = +1 600 400 200 SLEW RATE – V/µs 1200 G = –1 1000 800 600 400 200 1 2 3 4 5 6 OUTPUT STEP SIZE – V p-p 7 8 G = +2 G = +1 1.5 2.5 3.5 4.5 5.5 SUPPLY VOLTAGE – ±Volts 6.5 7.5 Figure 16. Slew Rate vs. Output Step Size Figure 19. Maximum Slew Rate vs. Supply Voltage 2V 100 20ns 100 500mV VIN 90 20ns VIN 90 VOUT 10 0% VOUT 10 0% 2V 500mV Figure 17. Large Signal Pulse Response, Gain = +1, (RF = 2 kΩ, RL = 150 Ω, VS = ± 5 V) Figure 20. Small Signal Pulse Response, Gain = +1, (RF = 2 kΩ, RL = 150 Ω, VS = ± 5 V) REV. A –7– PHASE SHIFT – Degrees –30 AD8013 50mV 100 20ns 100 2V VIN 90 20ns VIN 90 VOUT 10 0% VOUT 10 0% 500mV 2V Figure 21. Large Signal Pulse Response, Gain = +10, RF = 301 Ω, RL = 150 Ω, VS = ± 5 V) Figure 24. Large Signal Pulse Response, Gain = –1, (RF = 698 Ω, RL = 150 Ω, VS = ± 5 V) PHASE SHIFT – Degrees PHASE VS = +5V +1 G = +10 RL = 150Ω VS = ±5V 0 –90 –180 PHASE VS = +5V G = –1 RL = 150Ω VS = ±5V 180 90 0 –90 +1 CLOSED-LOOP GAIN (NORMALIZED) – dB 0 –1 –2 –3 –4 –5 –6 1M CLOSED-LOOP GAIN (NORMALIZED) – dB GAIN –270 0 –1 –2 –3 –4 –5 GAIN VS = +5V VS = ±5V VS = ±5V VS = +5V 10M 100M FREQUENCY – Hz 1G –6 1M 10M 100M FREQUENCY – Hz 1G Figure 22. Closed-Loop Gain and Phase vs. Frequency, G = +10, RL = 150 Ω Figure 25. Closed-Loop Gain and Phase vs. Frequency, G = –1, RL = 150 Ω 50mV 100 20ns 100 500mV VIN 90 20ns VIN 90 VOUT 10 0% VOUT 10 0% 500mV 500mV Figure 23. Small Signal Pulse Response, Gain = +10, (RF = 301 Ω, RL = 150 Ω, VS = ± 5 V) Figure 26. Small Signal Pulse Response, Gain = –1, (RF = 698 Ω, RL = 150 Ω, VS = ± 5 V) –8– REV. A PHASE SHIFT – Degrees AD8013 G = –10 RL = 150Ω VS = ±5V +1 VS = +5V 180 90 0 –90 PHASE CLOSED-LOOP GAIN (NORMALIZED) – dB 0 –1 –2 –3 GAIN VS = +5V –4 –5 –6 1M 100M 10M FREQUENCY – Hz VS = ±5V To estimate the –3 dB bandwidth for closed-loop gains of 2 or greater, for feedback resistors not listed in the following table, the following single pole model for the AD8013 may be used: G ACL 1 + SC ( R + Gn rin ) T F where: CT = transcapacitance 1 pF RF = feedback resistor G = ideal closed loop gain  RF  Gn = 1 + R  = noise gain  G rin = inverting input resistance 150 Ω ACL = closed loop gain The –3 dB bandwidth is determined from this model as: 1 f3 2 π CT ( RF + Gn rin ) This model will predict –3 dB bandwidth to within about 10% to 15% of the correct value when the load is 150 Ω and VS = ± 5 V. For lower supply voltages there will be a slight decrease in bandwidth. The model is not accurate enough to predict either the phase behavior or the frequency response peaking of the AD8013. It should be noted that the bandwidth is affected by attenuation due to the finite input resistance. Also, the open-loop output resistance of about 12 Ω reduces the bandwidth somewhat when driving load resistors less than about 250 Ω. (Bandwidths will be about 10% greater for load resistances above a few hundred ohms.) Table I. –3 dB Bandwidth vs. Closed-Loop Gain and Feedback Resistor, RL = 150 Ω (SOIC) VS – Volts ±5 Gain +1 +2 +10 –1 –10 +1 +2 +10 –1 –10 RF – Ohms 2000 845 (931) 301 698 (825) 499 2000 887 (931) 301 698 (825) 499 BW – MHz 230 150 (135) 80 140 (130) 85 180 120 (130) 75 130 (120) 80 1G Figure 27. Closed-Loop Gain and Phase vs. Frequency, G = –10, RL = 150 Ω General The AD8013 is a wide bandwidth, triple video amplifier that offers a high level of performance on less than 4.0 mA per amplifier of quiescent supply current. The AD8013 uses a proprietary enhancement of a conventional current feedback architecture, and achieves bandwidth in excess of 200 MHz with low differential gain and phase errors, making it an extremely efficient video amplifier. The AD8013’s wide phase margin coupled with a high output short circuit current make it an excellent choice when driving any capacitive load. High open-loop gain and low inverting input bias current enable it to be used with large values of feedback resistor with very low closed-loop gain errors. It is designed to offer outstanding functionality and performance at closed-loop inverting or noninverting gains of one or greater. Choice of Feedback & Gain Resistors Because it is a current feedback amplifier, the closed-loop bandwidth of the AD8013 may be customized using different values of the feedback resistor. Table I shows typical bandwidths at different supply voltages for some useful closed-loop gains when driving a load of 150 Ω. The choice of feedback resistor is not critical unless it is important to maintain the widest, flattest frequency response. The resistors recommended in the table are those (chip resistors) that will result in the widest 0.1 dB bandwidth without peaking. In applications requiring the best control of bandwidth, 1% resistors are adequate. Package parasitics vary between the 14-pin plastic DIP and the 14-pin plastic SOIC, and may result in a slight difference in the value of the feedback resistor used to achieve the optimum dynamic performance. Resistor values and widest bandwidth figures are shown in parenthesis for the SOIC where they differ from those of the DIP. Wider bandwidths than those in the table can be attained by reducing the magnitude of the feedback resistor (at the expense of increased peaking), while peaking can be reduced by increasing the magnitude of the feedback resistor. Increasing the feedback resistor is especially useful when driving large capacitive loads as it will increase the phase margin of the closed-loop circuit. (Refer to the section on driving capacitive loads for more information.) REV. A –9– PHASE SHIFT – Degrees +5 Driving Capacitive Loads When used in combination with the appropriate feedback resistor, the AD8013 will drive any load capacitance without oscillation. The general rule for current feedback amplifiers is that the higher the load capacitance, the higher the feedback resistor required for stable operation. Due to the high open-loop transresistance and low inverting input current of the AD8013, the use of a large feedback resistor does not result in large closedloop gain errors. Additionally, its high output short circuit current makes possible rapid voltage slewing on large load capacitors. For the best combination of wide bandwidth and clean pulse response, a small output series resistor is also recommended. Table II contains values of feedback and series resistors which result in the best pulse responses. Figure 29 shows the AD8013 driving a 300 pF capacitor through a large voltage step with virtually no overshoot. (In this case, the large and small signal pulse responses are quite similar in appearance.) AD8013 RF +VS RG 4 1.0µF 0.1µF 15Ω 1.0µF 0.1µF RS VO CL AD8013 VIN RT –VS 11 As noted in the warning under “Maximum Power Dissipation,” a high level of input overdrive in a high noninverting gain circuit can result in a large current flow in the input stage. Though this current is internally limited to about 30 mA, its effect on the total power dissipation may be significant. High Performance Video Line Driver Figure 28. Circuit for Driving a Capacitive Load Table II. Recommended Feedback and Series Resistors vs. Capacitive Load and Gain C L – pF 20 50 100 200 300 ≥500 RF – Ohms 2k 2k 3k 4k 6k 7k 500mV 100 At a gain of +2, the AD8013 makes an excellent driver for a back terminated 75 Ω video line (Figures 31, 32, and 33). Low differential gain and phase errors and wide 0.1 dB bandwidth can be realized. The low gain and group delay matching errors ensure excellent performance in RGB systems. Figures 34 and 35 show the worst case matching. RG RF +VS 4 RS – Ohms G=2 G≥3 25 25 20 15 15 15 15 15 15 15 15 15 75Ω CABLE VIN 0.1µF 75Ω 75Ω CABLE VOUT 0.1µF 75Ω AD8013 11 75Ω –VS 50ns Figure 31. A Video Line Driver Operating at a Gain of +2 (RF = RG from Table I) G = +2 RL = 150Ω VS = ±5V +1 VS = +5V 0 –90 –180 –270 VS = ±5V VIN 90 PHASE 0% CLOSED-LOOP GAIN (NORMALIZED) – dB VOUT 10 0 –1 –2 –3 GAIN 1V Figure 29. Pulse Response Driving a Large Load Capacitor. CL = 300 pF, G = +2, RF = 6k, RS = 15 Ω Overload Recovery VS = +5V –4 –5 –6 1M 100M 10M FREQUENCY – Hz 1G NORMALIZED GAIN – dB The three important overload conditions are: input commonmode voltage overdrive, output voltage overdrive, and input current overdrive. When configured for a low closed-loop gain, the amplifier will quickly recover from an input commonmode voltage overdrive; typically in under 25 ns. When configured for a higher gain, and overloaded at the output, the recovery time will also be short. For example, in a gain of +10, with 15% overdrive, the recovery time of the AD8013 is about 20 ns (see Figure 30). For higher overdrive, the response is somewhat slower. For 6 dB overdrive, (in a gain of +10), the recovery time is about 65 ns. 500mV 100 Figure 32. Closed-Loop Gain & Phase vs. Frequency for the Line Driver G = +2 RL = 150Ω +0.2 +0.1 0 –0.1 –0.2 –0.3 –0.4 –0.5 1M 10M 100M FREQUENCY – Hz 1G VS = +5V VS = ±5V 50ns VIN 90 VOUT 10 0% 5V Figure 33. Fine-Scale Gain Flatness vs. Frequency, G = +2, RL = 150 Ω Figure 30. 15% Overload Recovery, G = +10 (RF = 300 Ω, RL = 1 kΩ, VS = ± 5 V) –10– REV. A PHASE SHIFT – Degrees AD8013 1.5 1.0 0.5 G = +2 RL = 150Ω VI 8k +5V GAIN MATCHING – dB TO DISABLE PIN 4k 10k –5V VS = +5V 0 –0.5 VS = ±5V –1.0 V I HIGH => AMPLIFIER ENABLED V I LOW => AMPLIFIER DISABLED Figure 36. Level Shifting to Drive Disable Pins on Dual Supplies –1.5 –2.0 1M 100M 10M FREQUENCY – Hz 1G The AD8013’s input stages include protection from the large differential input voltages that may be applied when disabled. Internal clamps limit this voltage to about ± 3 V. The high input to output isolation will be maintained for voltages below this limit. 3:1 Video Multiplexer Figure 34. Closed-Loop Gain Matching vs. Frequency 10 8 6 GROUP DELAY – ns 4 2 1.0 0.5 0 –0.5 –1.0 100k DELAY MATCHING 10M 1M FREQUENCY – Hz VS = +5V G = +2 RL = 150Ω VS = ±5V VS = ±5V VS = +5V G = +2 RL = 150Ω Wiring the amplifier outputs together will form a 3:1 mux with excellent switching behavior. Figure 37 shows a recommended configuration which results in –0.1 dB bandwidth of 35 MHz and OFF channel isolation of 60 dB at 10 MHz on ± 5 V supplies. The time to switch between channels is about 50 ns. Switching time is virtually unaffected by signal level. DELAY 665Ω 845Ω +VS 6 4 7 84Ω VIN1 5 1 100M DISABLE 1 75Ω Figure 35. Group Delay and Group Delay Matching vs. Frequency, G = +2, RL = 150 Ω Disable Mode Operation VIN2 665Ω 845Ω 75Ω CABLE VOUT 75Ω 13 14 12 2 84Ω Pulling the voltage on any one of the Disable pins about 1.6 V up from the negative supply will put the corresponding amplifier into a disabled, powered down, state. In this condition, the amplifier’s quiescent current drops to about 0.3 mA, its output becomes a high impedance, and there is a high level of isolation from input to output. In the case of the gain of two line driver for example, the impedance at the output node will be about the same as for a 1.6 kΩ resistor (the feedback plus gain resistors) in parallel with a 12 pF capacitor and the input to output isolation will be about 66 dB at 5 MHz. Leaving the Disable pin disconnected (floating) will leave the corresponding amplifier operational, in the enabled state. The input impedance of the disable pin is about 40 kΩ in parallel with a few picofarads. When driven to 0 V, with the negative supply at –5 V, about 100 µA flows into the disable pin. When the disable pins are driven by complementary output CMOS logic, on a single 5 V supply, the disable and enable times are about 50 ns. When operated on dual supplies, level shifting will be required from standard logic outputs to the Disable pins. Figure 36 shows one possible method which results in a negligible increase in switching time. 75Ω DISABLE 2 665Ω 845Ω 9 8 11 84Ω 3 VIN3 10 75Ω –VS DISABLE 3 Figure 37. A Fast Switching 3:1 Video Mux (Supply Bypassing Not Shown) 500mV 100 90 200ns 10 0% 5V Figure 38. Channel Switching Characteristic for the 3:1 Mux REV. A –11– AD8013 2:1 Video Multiplexer 698Ω 698Ω +5V 6 5 13 R1 2kΩ 2k 13 4 1 7 15Ω VOUT 2 VINA 12 2 14 R3 10Ω DISABLE 5 100Ω VIN 12 50Ω 14 DIS 1 1k 10 1k 9 DIS 3 3 11 –5V 845Ω 845Ω 8 VINB 10 R4 10Ω 1 6 7 VOUT 3 9 3 8 R2 2kΩ DISABLE R5 845Ω R6 845Ω Figure 39. 2:1 Mux with High Isolation and Low Differential Gain and Phase Errors 2 1 GAIN 0 CLOSED-LOOP GAIN – dB 100 90 Figure 41. Circuit to Switch Between Gains of –1 and +1 500mV 500mV 200ns –1 –2 –3 –4 –5 FEEDTHROUGH –6 –7 –8 1M 100M 10M FREQUENCY – Hz –60 –70 –80 1G –30 –40 –50 FEEDTHROUGH – dB 10 0% 5V Figure 42. Switching Characteristic for Circuit of Figure 41 Figure 40. 2:1 Mux ON Channel Gain and Mux OFF Channel Feedthrough vs. Frequency Gain Switching OUTLINE DIMENSIONS Dimensions shown in inches and (mm). 14-Lead Plastic DIP (N-14) 0.795 (20.19) 0.725 (18.42) 14 1 8 7 14-Lead SOIC (R-14) 0.3444 (8.75) 0.3367 (8.55) 0.280 (7.11) 0.240 (6.10) 0.325 (8.25) 0.300 (7.62) 0.195 (4.95) 0.115 (2.93) 0.1574 (4.00) 0.1497 (3.80) 14 1 8 7 0.2440 (6.20) 0.2284 (5.80) PIN 1 0.210 (5.33) MAX 0.160 (4.06) 0.115 (2.93) 0.022 (0.558) 0.014 (0.356) 0.060 (1.52) 0.015 (0.38) 0.130 (3.30) MIN 0.100 0.070 (1.77) (2.54) 0.045 (1.15) BSC SEATING PLANE PIN 1 0.0098 (0.25) 0.0040 (0.10) 0.0688 (1.75) 0.0532 (1.35) 0.0196 (0.50) x 45° 0.0099 (0.25) 0.015 (0.381) 0.008 (0.204) SEATING PLANE 0.0500 (1.27) BSC 0.0192 (0.49) 0.0138 (0.35) 0.0098 (0.25) 0.0075 (0.19) 8° 0° 0.0500 (1.27) 0.0160 (0.41) –12– REV. A PRINTED IN U.S.A. C2084–18–10/95 Configuring two amplifiers as unity gain followers and using the third to set the gain results in a high performance 2:1 mux (Figures 39 and 40). This circuit takes advantage of the very low crosstalk between Channels 2 and 3 to achieve the OFF channel isolation shown in Figure 40. This circuit can achieve differential gain and phase of 0.03% and 0.07° respectively. The AD8013 can be used to build a circuit for switching between any two arbitrary gains while maintaining a constant input impedance. The example of Figure 41 shows a circuit for switching between a noninverting gain of 1 and an inverting gain of 1. The total time for channel switching and output voltage settling is about 80 ns.